WO2020019741A1 - 一种n相n+1桥臂逆变器及其调制方法 - Google Patents

一种n相n+1桥臂逆变器及其调制方法 Download PDF

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WO2020019741A1
WO2020019741A1 PCT/CN2019/078987 CN2019078987W WO2020019741A1 WO 2020019741 A1 WO2020019741 A1 WO 2020019741A1 CN 2019078987 W CN2019078987 W CN 2019078987W WO 2020019741 A1 WO2020019741 A1 WO 2020019741A1
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bridge arm
phase
voltage
bridge
inverter
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PCT/CN2019/078987
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English (en)
French (fr)
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蒋栋
李安
刘自程
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华中科技大学
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Priority to US17/040,553 priority Critical patent/US11342879B2/en
Publication of WO2020019741A1 publication Critical patent/WO2020019741A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H02P25/22—Multiple windings; Windings for more than three phases
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/26—Rotor flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation

Definitions

  • the invention belongs to the field of AC motors and drive control, and more particularly, relates to an N-phase N + 1 bridge arm inverter topology and a modulation method thereof.
  • half-bridge inverter topology is the basic topology of traditional multi-phase inverters.
  • stator winding of the motor has a neutral point and the neutral point is suspended, the same component (ie, the zero-axis current component) in the stator current cannot flow.
  • This topology does not have a zero-axis current path, which reduces the degree of freedom in controlling the stator current. In some multi-phase motors that require zero-axis current, this topology cannot be adopted.
  • the maximum fundamental voltage amplitude that can be generated in the output voltage is about half of the DC bus voltage, which results in a narrow speed range of the motor and limits the motor's operating range.
  • it is necessary to increase the DC bus voltage which will increase the cost of the entire drive system and reduce reliability and power density.
  • the present invention provides an N-phase N + 1 bridge arm inverter topology and a modulation method thereof, the purpose of which is to reduce the number of power devices in the inverter topology and reduce the capacity of the inverter system. Reduce drive system cost and increase power density.
  • the present invention provides an N-phase N + 1 bridge arm inverter, including: N + 1 bridge arms, each bridge arm including: an upper bridge arm power switch device and a lower bridge arm power switch device, each bridge arm
  • the upper node of the upper arm power switching device is connected to the DC bus voltage
  • the lower node of the lower arm power switching device is connected to the power ground
  • the lower node of the upper arm power switching device is connected to the upper node of the lower arm power switching device
  • all the stator windings are arranged in the order of the electrical angle interval ⁇ n ⁇ and connected end to end, thereby leading to N + 1 motor winding nodes; ⁇ n is the number of arrangement intervals, and ⁇ n may be 1 to (N-1) / 2 is an integer that is coprime with N. Each value of ⁇ n corresponds to the connection order of a phase winding.
  • the N + 1 motor winding nodes are sequentially connected with the N + 1 bridge arm output nodes in sequence.
  • the present invention also provides a modulation method of an N-phase N + 1 bridge arm inverter, including the following steps:
  • V p (i) V l (k) -V l (k + 1) to obtain the N + 1 bridge arm voltage values
  • the real-time duty cycle is obtained by comparing the N + 1 bridge arm voltage values with the carrier triangle wave, and the square-wave drive signals of each bridge arm power device are generated according to the real-time duty cycle, and the carrier PWM voltage modulation of the common-mode voltage injection is achieved.
  • the maximum fundamental phase voltage amplitude generated by the N-phase N + 1 bridge arm inverter is the DC bus voltage U dc .
  • the N-phase N + 1 bridge arm inverter has a high DC voltage utilization rate and can be applied to DC In the application scenarios where the bus voltage is limited and high DC voltage utilization is required, such as electric vehicles, ship propulsion systems, etc.
  • the N-phase N + 1 bridge arm inverter topology has good fault-tolerance performance. If one phase winding is disconnected, it will not affect the operation of other windings.
  • the N-phase N + 1 bridge arm inverter topology has a zero-axis current path and has the ability to control the zero-axis current, while the N-phase half-bridge inverter cannot flow and control the zero-axis current.
  • the N-phase half-bridge topology cannot be used, and the N-phase N + 1 bridge arm inverter Topologies do not have this limitation.
  • the application field of N-phase N + 1 bridge arm inverter topology is wider.
  • Figure 1 is the electrical angle distribution of the stator voltage of each phase of a five-phase six-pole permanent magnet synchronous motor
  • FIG. 2 is a topology diagram of a five-phase six-bridge arm inverter provided by the present invention
  • FIG. 3 is a structural diagram of a voltage modulation algorithm of a five-phase six-bridge arm inverter provided by the present invention
  • Figure 4 is a flowchart of a general-purpose multi-phase motor vector control system
  • FIG. 5 is a structural diagram of a five-phase motor drive control system provided by the present invention.
  • the present invention can reduce the number of power devices in the inverter topology, reduce the capacity of the inverter system, reduce the cost of the drive system, increase the power density, improve the utilization rate of the DC voltage, improve the fault tolerance of the topology, and realize the stator Control of all degrees of current.
  • N is an odd number
  • N is 3 or more.
  • Each bridge arm contains an upper bridge arm power switch device and a lower bridge arm power switch device (the power switch device can be a two-phase limited full-control power device such as an IGBT or MOSFET with anti-parallel diodes).
  • the upper node of the bridge arm power switching device is connected to the DC bus voltage
  • the lower node of the lower arm power switching device is connected to the power ground
  • the lower node of the upper arm power switching device is connected to the upper node of the lower arm power switching device as a bridge arm.
  • V p (1) , V p (2) , ..., V p (n) respectively correspond to the first, second, second, second, n-phase stator windings of the motor. Voltage at the winding end.
  • Set V l (1) , V l (2) , ..., V l (n + 1) respectively correspond to the output node voltages of the 1st to n + 1 bridge arms.
  • ⁇ 2 ⁇ / N be the electrical angle between adjacent voltage phasors.
  • ⁇ n be the number of arrangement intervals.
  • the possible values of ⁇ n are integers that are coprime with N from 1 to (N-1) / 2.
  • Each value of ⁇ n corresponds to the connection order of a phase winding.
  • the number of values of ⁇ n is not the same, that is, there are multiple ways to connect the phase windings of multi-phase motors.
  • the 6 nodes that are drawn out are connected to the bridge arms 1 to 6 in sequence, and two types of five-phase and six bridge arm inverter topologies are obtained.
  • the present invention provides a carrier PWM voltage modulation method for common mode voltage injection of the N-phase N + 1 bridge arm inverter topology provided in the first aspect, the modulation method is applicable to any phase topology and any The values of ⁇ n are all valid, including:
  • the N-dimensional phase voltage vector generated by the N-phase N + 1 bridge arm inverter is [V p (1) , V p (2) , ..., V p (n) ] T , N + 1 dimension
  • the voltage vectors of the bridge arms are [V l (1) , V l (2) , ..., V l (n + 1) ] T.
  • the output voltage vectors of N + 1 bridge arms and N The relationship of the phase voltage vector is:
  • V p (i) V l (k) -V l (k + 1) i, k ⁇ (1,2, ..., n) ... (1)
  • the N-dimensional reference phase voltage vector to be modulated in the stationary coordinate system can be converted into N reference phase voltage sub-vectors in the N-dimensional synchronous coordinate system:
  • the bridge arm voltage vector can be expressed as a linear combination of (N-1) / 2 two-dimensional plane dq-axis phase voltage sub-vectors and a zero-axis phase voltage sub-vector as:
  • the linear combination of the dq-axis phase voltage component vectors in each plane of the bridge arm voltage in the generalized synchronous coordinate system is recorded as the dq-axis bridge arm voltage component vector in the plane.
  • the dq-axis bridge arm can be directly obtained from the dq-axis phase voltage sub-vector in each plane Voltage vector:
  • equation (4) can be rewritten as a modified inverse PARK transformation formula:
  • the injected common-mode voltage component can be obtained as:
  • the five-phase stator currents are i p (1) , i p (2) , i p (3) , i p (4) , and i p (5) , respectively, and the five-phase stator voltages are V p (1) , V p (2) , V p (3) , V p (4) , V p (5) .
  • Figure 2 is a phasor distribution diagram of the phase frequency of the stator fundamental frequency of a five-phase motor.
  • the six motor winding nodes are sequentially connected to the output nodes of the six bridge arms of the five-phase six-bridge inverter in sequence, so as to obtain the topology diagram of the five-phase six-bridge inverter shown in FIG. 3.
  • the voltage modulation algorithm provided by the present invention needs to be adopted. By controlling the complementary conduction of the power switching devices in the six bridge arms, the five-phase stator voltage required by the motor is generated.
  • the 5-phase stator voltage can be converted into d-axis and q-axis phase voltage sub-vectors in two two-dimensional planes (fundamental plane: V d (1) and V q (1 ) , The third harmonic plane: V d (3) and V q (3) ) and a zero-axis phase voltage sub-vector V 0 .
  • the voltage vectors of the six bridge arms are obtained from the four bridge arm voltage sub-vectors and a zero-axis phase voltage sub-vector in the generalized synchronous coordinate system:
  • the common-mode voltage component is added to all 6 bridge-arm voltages to obtain the voltage values of the 6 bridge arms after the common-mode voltage component is injected.
  • the driving square wave signal of the power switching device in each bridge arm can be obtained.
  • the driving square wave signal is input into the driving circuit, and the driving circuit can drive the power switching device to generate the required motor stator phase voltage.
  • the common-mode voltage-injected carrier-comparison PWM strategy has undergone two coordinate transformations. Because the transformation matrix is more complex and requires a large amount of calculation, it can be found that in the two coordinate transformation matrices, the trigonometric functions can correspond one-to-one, so in one control In the period, the value of the transformation matrix only needs to be calculated once, and does not need to be calculated twice, which reduces the calculation amount by half.
  • the PWM modulation method can be combined with the vector control technology of the multi-phase motor to simplify the calculation.
  • Figure 4 is a general-purpose multi-phase motor vector control system flowchart. In the vector control of general-purpose multi-phase motors, the currents of each phase of the motor are collected.
  • the d-axis and q-axis currents and the zero-axis current in the generalized park coordinate system can be obtained.
  • Each current component is realized by a current regulator. Closed-loop control, to obtain the corresponding d-axis and q-axis voltage and the reference value of the zero-axis voltage, and then to obtain the command value of the phase voltage through generalized PARK inverse transformation, and to obtain the drive signal of the inverter switching device through the voltage modulation link.
  • the coordinate transformation of the middle two times can be directly simplified to obtain the structure diagram of the five-phase motor vector control system shown in FIG. 5.
  • the vector control system of the five-phase motor only needs to pass two coordinate transformations instead of four times, which greatly simplifies the calculation complexity.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

本发明公开了一种N相N+1桥臂逆变器，包括：N+1个桥臂，每个桥臂包括：上桥臂功率开关器件和下桥臂功率开关器件，每个桥臂中的上桥臂功率开关器件的上节点连接直流母线电压，下桥臂功率开关器件的下节点连接电源地，上桥臂功率开关器件的下节点与下桥臂功率开关器件的上节点连接，作为桥臂的输出节点；其中N为电机的总相数，N为奇数，且N大于等于3。本发明可以减少逆变拓扑中的功率器件数量，减少逆变器***容量，降低驱动***成本，提高功率密度，提高直流电压利用率，提高拓扑容错能力，并实现定子电流全部自由度的控制。

Description

一种N相N+1桥臂逆变器及其调制方法 【技术领域】
本发明属于交流电机与驱动控制领域，更具体地，涉及一种N相N+1桥臂逆变器拓扑结构及其调制方法。
【背景技术】
近年来，由于多相电机具有转矩脉动小，功率器件容量要求低，容错能力强等诸多优点，多相电机及其控制技术越来越引起人们的重视。在多相电机驱动的研究中，半桥逆变拓扑是传统的多相逆变器基本拓扑结构。
在半桥逆变器结构中，多相电机定子绕组的一端连接相桥臂，另一端星型连接形成中性点。该拓扑结构存在以下一些缺陷：
(1)由于电机定子绕组具有中性点，且中性点悬空，因此定子电流中相同的成分(即零轴电流成分)无法流通。该拓扑结构不具备零轴电流通路，减少了控制定子电流的自由度。在某些需要零轴电流的多相电机中，该拓扑结构无法被采用。
(2)在该拓扑中，输出电压中所能生成的最大基波电压幅值为直流母线电压的一半左右，这导致了电机的调速范围窄，限制了电机的运行范围。若要改善电机的运行范围，则需要提高直流母线电压，这将提高整个驱动***的成本，且可靠性和功率密度降低。
(3)由于存在中性点，各相电压之间相互影响，当电机某相定子绕组故障时，会影响其他正常定子相绕组的电压，进而使电机不能正常运行，容错性能较差。
现有技术中，存在一种N相全桥逆变器拓扑结构，每个H桥逆变控制一相定子绕组，可以克服上述N相半桥逆变器拓扑的缺点，但同时带来了功率器件多，***成本高，功率密度低，运行损耗大的缺点。因此，在多 相电机驱动领域，电机驱动器的性能、成本以及功率密度始终存在难以调和的矛盾，难以实现最优的匹配。
【发明内容】
针对现有技术的缺陷，本发明提供了一种N相N+1桥臂逆变器拓扑结构及其调制方法，其目的在于减少逆变拓扑中的功率器件数量，减少逆变器***容量，降低驱动***成本，提高功率密度。
本发明提供了一种N相N+1桥臂逆变器，包括：N+1个桥臂，每个桥臂包括：上桥臂功率开关器件和下桥臂功率开关器件，每个桥臂中的上桥臂功率开关器件的上节点连接直流母线电压，下桥臂功率开关器件的下节点连接电源地，上桥臂功率开关器件的下节点与下桥臂功率开关器件的上节点连接，作为桥臂的输出节点；其中N为电机的总相数，N为奇数，且N大于等于3。
其中，当电机定子绕组对称分布时，相邻相对应的电动势之间的相位差α＝2π/N。
其中，从第一相开始，将所有的定子绕组按照电角度间隔Δnα的顺序进行排列并首尾相连，从而引出N+1个电机绕组节点；Δn为排列间隔数，Δn可能的取值为1到(N-1)/2中与N互质的整数，每一种Δn的取值对应了一种相绕组的连接顺序。这N+1个电机绕组节点与N+1个桥臂输出节点依次顺序连接。
其中，N+1个桥臂的输出电压矢量与N个相电压矢量的关系为相电压等于其右节点连接的桥臂电压减去其左节点连接的桥臂电压V p(i)＝V l(k)-V l(k+1)i,k∈(1,2,…,n)；其中，N维相电压矢量为[V p(1)，V p (2)，……，V p(n)] T，N+1维桥臂电压矢量为[V l(1)，V l(2)，……，V l(n+1)] T。
本发明还提供了一种N相N+1桥臂逆变器的调制方法，包括下述步骤：
根据N+1个桥臂的输出电压矢量与N个相电压矢量的关系 V p(i)＝V l(k)-V l(k+1)获得N+1个桥臂电压值；
将N+1个桥臂电压值与载波三角波进行比较后获得实时占空比，并根据实时占空比产生各个桥臂功率器件的方波驱动信号，实现共模电压注入的载波PWM电压调制。
其中，当排列间隔数Δn＝(N-1)/2时，N相N+1桥臂逆变器产生的最大的基波相电压幅值为直流母线电压U dc。
其中，在广义同步坐标系下根据N+1个桥臂的输出电压矢量与N个相电压矢量的关系V p(i)＝V l(k)-V l(k+1)获得N+1个桥臂电压值。
总体而言，通过本发明所构思的以上技术方案与现有技术相比，具有以下有益效果：
(1)相比于N相半桥逆变器拓扑，在驱动不需要零轴电流的多相电机负载时，N相N+1桥臂逆变器的直流电压利用率高，可以应用在直流母线电压有限，需要高直流电压利用率的应用场景中，譬如电动汽车、船舶推动***等领域。
(2)相比N相半桥，N相N+1桥臂逆变器拓扑结构的容错性能好，某相绕组如果断路，不会影响其他绕组的工作。
(3)N相N+1桥臂逆变器拓扑存在零轴电流通路且具有对零轴电流的控制能力，而N相半桥逆变器不能流通和控制零轴电流。在需要零轴电流的电机负载中(如直流励磁电机，开关磁阻电机和电流谐波注入多相电机等)，N相半桥拓扑不能被采用，而N相N+1桥臂逆变器拓扑则不存在该限制。N相N+1桥臂逆变器拓扑的应用领域更广阔。
(4)在N相N+1桥臂逆变器中，除桥臂1和桥臂N+1两个桥臂外，其他桥臂均不流通零轴电流。因此，在N+1桥臂N相逆变器中，桥臂功率器件的电流应力低，进一步降低器件成本，且***的运行效率高。
【附图说明】
图1为五相六极永磁同步电机各相定子电压的电角度分布；
图2为本发明提供的五相六桥臂逆变器拓扑结构图；
图3为本发明提供的五相六桥臂逆变器电压调制算法结构图；
图4为通用多相电机矢量控制***流程图；
图5为本发明提供的五相电机驱动控制***结构图。
【具体实施方式】
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅仅用以解释本发明，并不用于限定本发明。
针对现有技术的缺陷，本发明可以减少逆变拓扑中的功率器件数量，减少逆变器***容量，降低驱动***成本，提高功率密度，提高直流电压利用率，提高拓扑容错能力，并实现定子电流全部自由度的控制。
为实现上述目的，第一方面，本发明提供一种N相N+1桥臂逆变器拓扑，包括：桥臂1至桥臂n+1，共N+1个桥臂，其中N(＝n)为电机的总相数，N为奇数，且N大于等于3。当电机定子绕组对称分布时，相邻相对应的电动势之间的相位差为α＝2π/N。
每个桥臂包含一个上桥臂功率开关器件和一个下桥臂功率开关器件(功率开关器件可以为带反并联二极管的IGBT或MOSFET等两相限全控功率器件)，每个桥臂的上桥臂功率开关器件的上节点连接直流母线电压，下桥臂功率开关器件的下节点连接电源地，上桥臂功率开关器件的下节点与下桥臂功率开关器件的上节点连接，作为桥臂的输出节点；
N相N+1桥臂逆变拓扑中，设定V p(1)，V p(2)，……，V p(n)分别对应电机第1，2，……，n相定子绕组中的绕组端电压。设定V l(1)，V l(2)，……，V l(n+1)分别对应第1至n+1桥臂的输出节点电压。设α＝2π/N为相邻电压相量间的电角度。
以电机的第1相为参考相，定子绕组的左节点对应绕组端电压的正极， 绕组的右节点对应绕组端电压的负极。记Δn为排列间隔数，Δn可能的取值为1到(N-1)/2中与N互质的整数，每一种Δn的取值对应了一种相绕组的连接顺序。对不同相数的多相电机来说，Δn的取值个数也不尽相同，即多相电机的相绕组连接顺序有多种方式，由于1和(N-1)/2一定与N互质，因此多相电机的相绕组至少存在着两种连接顺序。在Δn取某一特定值时，其对应的具体相绕组连接顺序和相绕组与桥臂的连接方式阐述如下：
从第一相开始，将所有的定子绕组按照电角度间隔Δnα的顺序进行排列(1，1+Δn，1+2Δn，……)并首尾相连，从而可以引出N+1个电机绕组节点。将这N+1个电机绕组节点与N+1个桥臂输出节点依次顺序连接，从而得到了一种N相N+1桥臂逆变器拓扑结构。譬如在五相电机中，N＝5且(N-1)/2＝2，故Δn可以等于1和2，则五相绕组排列顺序有两种，为：(-1-2-3-4-5-)和(-1-3-5-2-4-)，其中“-”为节点，因此可以引出6个节点。将引出的6个节点与桥臂1至6顺序相连，即得到了两种五相六桥臂逆变器拓扑结构。
第二方面，本发明提供一种针对上述第一方面提供的N相N+1桥臂逆变器拓扑结构的共模电压注入的载波PWM电压调制方法，该调制方法对任意相拓扑以及任意取值的Δn均成立，包括：
要求所述的N相N+1桥臂逆变器产生的N维相电压矢量为[V p(1)，V p (2)，……，V p(n)] T，N+1维桥臂电压矢量为[V l(1)，V l(2)，……，V l(n+1)] T，根据拓扑结构可以得到，N+1个桥臂的输出电压矢量与N个相电压矢量的关系为：
V p(i)＝V l(k)-V l(k+1)  i,k∈(1,2,…,n)……(1)
在采用载波PWM策略对相电压调制时，则需要在相电压矢量确定时，确定桥臂的输出电压矢量的大小，由于上述关系表达式中，桥臂电压矢量的未知个数为N+1，而方程数为N，因此理论上有无穷个解。以下提出一种在广义同步坐标系下求取桥臂电压矢量的方法，阐述如下：
首先根据广义PARK变换，可以将静止坐标系下的需要调制的N维参考相电压矢量转换为N维同步坐标系下的N个参考相电压分矢量：
Figure PCTCN2019078987-appb-000001
其中，
Figure PCTCN2019078987-appb-000002
为了调制出所需的相电压，可以将桥臂电压矢量用(N-1)/2个2维平面dq轴相电压分矢量和1个零轴相电压分矢量的线性组合表示为：
Figure PCTCN2019078987-appb-000003
其中，β＝Δn·α。容易证明将桥臂电压矢量用公式(4)表示时，相电压 与桥臂电压之间的要求关系始终是满足的，即公式(1)是始终成立的。
为了计算的方便，将桥臂电压在广义同步坐标系下的各平面内的dq轴相电压分矢量的线性组合记为该平面内的dq轴桥臂电压分矢量。在广义同步坐标系下的第m平面内(m＝1，3，5，…，n-2)，根据上述分析计算，可以直接由各平面内的dq轴相电压分矢量得到dq轴桥臂电压分矢量：
Figure PCTCN2019078987-appb-000004
因此，式(4)可以重写为修改的逆PARK变换公式：
Figure PCTCN2019078987-appb-000005
其中，
Figure PCTCN2019078987-appb-000006
假设V l(max)和V l(min)为式(7)求出的N+1维桥臂电压矢量中的最大值和最小值，则可以得到注入的共模电压分量为：
Figure PCTCN2019078987-appb-000007
把所有的由公式(7)求出的N+1个桥臂电压均加上该共模电压分量，得到注入共模电压分量后的N+1个桥臂的电压值。将这N+1个桥臂电压值与载波三角波比较，即可以得到实时占空比，产生各个桥臂功率器件的方波驱动信号，从而实现共模电压注入的载波PWM电压调制方法。
可以证明，采用以上共模电压注入的载波PWM电压调制方法调制的N相N+1桥臂逆变器，无论N为多少，当Δn＝(N-1)/2时，其可以产生的最大的基波相电压幅值为直流母线电压U dc，直流电压利用率高。
为了更进一步的说明本发明实施例提供的N相N+1桥臂逆变器及其调制方法，现结合附图以及具体实例详述如下：
在静止坐标系下，五相定子电流分别为i p(1)、i p(2)、i p(3)、i p(4)、i p(5)，五相定子电压分别为V p(1)、V p(2)、V p(3)、V p(4)、V p(5)。
图2为五相电机定子基频相电压的相量分布图，相邻相间相差的电角度为α＝360°/5＝72°。根据相量分布图，取Δn＝(5-1)/2＝2，可以得到定子绕组排列顺序为(-1-3-5-2-4-)，并引出6个电机定子绕组节点。将该将这6个电机绕组节点与五相六桥臂逆变器的6个桥臂的输出节点依次顺序连接，从而得到如图3所示的五相六桥臂逆变器拓扑结构图。
采用五相六桥臂逆变器拓扑驱动五相电机时，需要采用本发明所提供的电压调制算法。通过控制6个桥臂中功率开关器件的互补导通，产生电机所需要的五相定子电压。
根据广义PARK变换矩阵，在5相***中可以将5相定子电压转换为2个二维平面下的d轴和q轴相电压分矢量(基波平面：V d(1)和V q(1)，三次谐波平面：V d(3)和V q(3))和1个零轴相电压分矢量V 0。
Figure PCTCN2019078987-appb-000008
根据得到的2个二维平面下的d轴和q轴相电压分矢量求出2个二维平面下的d轴和q轴桥臂电压分矢量：
Figure PCTCN2019078987-appb-000009
其中，β＝Δn·α＝2×72°＝144°。
根据式(7)、(8)作修改的PARK逆变换，由广义同步坐标系下的4个桥臂电压分矢量和一个零轴相电压分矢量得到6个桥臂的电压矢量：
Figure PCTCN2019078987-appb-000010
比较这6个桥臂电压，找到其中的最大值和最小值V l(max)和V l(min)，计算共模电压分量：
Figure PCTCN2019078987-appb-000011
把所有的6个桥臂电压均加上该共模电压分量，得到注入共模电压分量后的6个桥臂的电压值。
将实际输出的6个桥臂电压与三角载波比较，桥臂电压大于载波信号时，上桥臂开通，下桥臂关断；桥臂电压小于载波信号时，下桥臂开通，上桥臂关断。从而可以得到各个桥臂中功率开关器件的驱动方波信号，将驱动方波信号输入到驱动电路中，驱动电路就可以驱动功率开关器件动作，产生需要的电机定子相电压。
该共模电压注入的载波比较PWM策略经过了两次坐标变换，由于变换矩阵比较复杂，计算量较大，但可以发现两次坐标变换的矩阵中，三角函数可以一一对应，因此在一个控制周期内，只需要计算一次变换矩阵的值，不需要计算两次，减少了一半的计算量。另外该PWM调制方法可以与多相电机的矢量控制技术相结合，实现简化计算。如图4所示为通用多相电机 矢量控制***流程图。在通用多相电机的矢量控制中，电机各相电流被采集，通过广义的PARK变换，可以得到广义PARK坐标系下的d轴与q轴电流以及零轴电流，各电流分量通过电流调节器实现闭环控制，得到对应的d轴与q轴电压以及零轴电压的参考值，再通过广义PARK逆变换得到相电压的指令值，通过电压调制环节得到逆变器开关器件的驱动信号。
当采用本发明实现的调制算法时，可以直接化简中间两次的坐标转换得到图5所示的五相电机矢量控制***结构图。此时五相电机的矢量控制***只需要通过两次坐标转换，不需要四次，大大简化了计算复杂度。
本领域的技术人员容易理解，以上所述仅为本发明的较佳实施例而已，并不用以限制本发明，凡在本发明的精神和原则之内所作的任何修改、等同替换和改进等，均应包含在本发明的保护范围之内。

Claims (7)

  1. 一种N相N+1桥臂逆变器，其特征在于，包括：N+1个桥臂，每个桥臂包括：上桥臂功率开关器件和下桥臂功率开关器件，每个桥臂中的上桥臂功率开关器件的上节点连接直流母线电压，下桥臂功率开关器件的下节点连接电源地，上桥臂功率开关器件的下节点与下桥臂功率开关器件的上节点连接，作为桥臂的输出节点；
    其中N为电机的总相数，N为奇数，且N大于等于3。
  2. 如权利要求1所述的N相N+1桥臂逆变器，其特征在于，当电机定子绕组对称分布时，相邻相对应的电动势之间的相位差α＝2π/N。
  3. 如权利要求1或2所述的N相N+1桥臂逆变器，其特征在于，从第一相开始，将所有的定子绕组按照电角度间隔Δnα的顺序进行排列并首尾相连，从而引出N+1个电机绕组节点；Δn为排列间隔数，Δn可能的取值为1到(N-1)/2中与N互质的整数，每一种Δ n的取值对应了一种相绕组的连接顺序。这N+1个电机绕组节点与N+1个桥臂输出节点依次顺序连接。
  4. 如权利要求1-3任一项所述的N相N+1桥臂逆变器，其特征在于，N+1个桥臂的输出电压矢量与N个相电压矢量的关系为相电压等于其右节点连接的桥臂电压减去其左节点连接的桥臂电压V p(i)＝V l(k)-V l(k+1)i,k∈(1,2,…,n)；其中，N维相电压矢量为[V p(1)，V p (2)，……，V p(n)] T，N+1维桥臂电压矢量为[V l(1)，V l(2)，……，V l(n+1)] T。
  5. 一种N相N+1桥臂逆变器的调制方法，其特征在于，包括下述步骤：
    根据N+1个桥臂的输出电压矢量与N个相电压矢量的关系：相电压等于其右节点连接的桥臂电压减去其左节点连接的桥臂电压V p(i)＝V l(k)-V l(k+1)获得N+1个桥臂电压值；
    将N+1个桥臂电压值与载波三角波进行比较后获得实时占空比，并根 据实时占空比产生各个桥臂功率器件的方波驱动信号，实现共模电压注入的载波PWM电压调制。
  6. 如权利要求5所述的调制方法，其特征在于，当排列间隔数Δn＝(N-1)/2时，N相N+1桥臂逆变器产生的最大的基波相电压幅值为直流母线电压U dc。
  7. 如权利要求5或6所述的调制方法，其特征在于，在广义同步坐标系下根据N+1个桥臂的输出电压矢量与N个相电压矢量的关系V p(i)＝V l(k)-V l(k+1)获得N+1个桥臂电压值。
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