WO2019244212A1 - Variable-speed motor device - Google Patents

Variable-speed motor device Download PDF

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Publication number
WO2019244212A1
WO2019244212A1 PCT/JP2018/023139 JP2018023139W WO2019244212A1 WO 2019244212 A1 WO2019244212 A1 WO 2019244212A1 JP 2018023139 W JP2018023139 W JP 2018023139W WO 2019244212 A1 WO2019244212 A1 WO 2019244212A1
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WO
WIPO (PCT)
Prior art keywords
phase
leg
current supply
motor device
pwm
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Application number
PCT/JP2018/023139
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French (fr)
Japanese (ja)
Inventor
田中　正一
Original Assignee
田中　正一
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Application filed by 田中　正一 filed Critical 田中　正一
Priority to PCT/JP2018/023139 priority Critical patent/WO2019244212A1/en
Priority to JP2020525094A priority patent/JPWO2019244212A1/ja
Priority to PCT/JP2019/008756 priority patent/WO2019244418A1/en
Priority to JP2020525258A priority patent/JP7027024B2/en
Priority to PCT/JP2019/022803 priority patent/WO2019244680A1/en
Priority to JP2020525537A priority patent/JP7191951B2/en
Priority to US16/973,478 priority patent/US20210288506A1/en
Publication of WO2019244212A1 publication Critical patent/WO2019244212A1/en
Priority to JP2022021672A priority patent/JP7218460B2/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H02P25/22—Multiple windings; Windings for more than three phases
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters

Definitions

  • the present invention relates to a PWM-driven variable speed motor device, and more particularly to a battery-powered variable speed PWM motor device.
  • a three-phase inverter that drives a star-connected three-phase coil is employed as an inverter of an electric vehicle (EV).
  • EV electric vehicle
  • a double inverter that drives a double-ended three-phase coil has also been proposed.
  • a six-phase inverter for driving a six-phase coil has also been proposed.
  • a six-phase coil is essentially equivalent to two three-phase coils.
  • Patent Document 1 proposes one of the double inverter type motor devices. According to Patent Literature 1, two legs connected to one phase coil are driven by PWM signals having opposite phases. Patent Document 2 proposes injecting a third harmonic current into a stator coil of a double inverter type motor device.
  • Patent Document 3 proposes a double inverter type motor device in which a three-phase rectifier and a short-circuit transistor are further added. According to this motor device, the first terminals of the three-phase coils are connected to the first three-phase inverter, and the second terminals of the three-phase coils are connected to the second three-phase inverter and the three-phase full-wave rectifier. Both are connected in parallel.
  • the short-circuit transistor short-circuits a pair of DC terminals of the three-phase rectifier.
  • the resistance of an EV battery increases due to its deterioration. As battery loss increases in proportion to this resistance, battery temperature rises. Battery deterioration is accelerated by the high temperature of the battery.
  • an inverter of an electric vehicle employs a high-voltage type transistor.
  • a high-voltage transistor has a higher on-resistance than a low-voltage transistor. Therefore, the power loss of the EV inverter is increased as compared with the general-purpose low-voltage inverter.
  • the three-phase inverter 3A connected to the unillustrated star-connected three-phase coil includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W.
  • FIG. 1 shows a state in which the upper arm transistor 3UU of the leg 3U, the lower arm transistor 3VL of the leg 3V, and the lower arm transistor 3WL of the leg 3W are on.
  • Inverter 3A is connected to battery 5 and smoothing capacitor 6.
  • the transistors 3UU, 3VL, and 3WL have an on-resistance r.
  • the battery 5 has a resistance value r5, and the capacitor 6 has a resistance value r6.
  • FIG. 2 is one schematic wiring diagram of the motor device of FIG.
  • the DC power supply 7 having an equivalent resistance value r7 includes a battery 5 and a capacitor 6.
  • Each of the six arms of inverter 3A has two transistors 300 connected in parallel.
  • the stator coil 1A connected to the inverter 3A includes three phase coils 1U0, 1V0, and 1W0 connected in a star shape.
  • a three-phase inverter for driving a three-phase motor executes sine-wave control for outputting a sine-wave three-phase voltage or rectangular wave control for outputting a three-phase voltage having a rectangular waveform.
  • the sine wave control employs a pulse width modulation method (PWM method).
  • PWM method each leg of the inverter has a current supply period and a freewheeling period every PWM cycle period.
  • the current supply period means a period during which the DC power supply supplies a power supply current to the leg.
  • the free wheeling period means a period during which the DC power supply does not supply a power supply current to the leg.
  • PWM method a triangular wave PWM method and a space vector PWM (SVPWM) method are well known.
  • the SVPWM method uses six voltage vectors, each essentially corresponding to a current supply period.
  • Sine wave control outputs an output voltage vector that rotates along one circle.
  • This output voltage vector is a vector sum (composite vector) of two adjacent voltage vectors.
  • the length of the voltage vector essentially corresponds to the current supply period.
  • two types of current supply periods corresponding to two types of voltage vectors are set in each PWM cycle period.
  • a problem with sine wave inverters is the switching losses of the inverter.
  • Square wave control outputs an output voltage vector that jumps from one voltage vector to another adjacent voltage vector.
  • the switching loss of the inverter is reduced as compared with the sine wave control.
  • the output voltage vector is composed of one of the above-mentioned six voltage vectors that are switched every 60 electrical degrees of the rotating magnetic field. That is, according to the rectangular wave control, the PWM method of synthesizing an arbitrary output voltage vector by the vector sum of two voltage vectors is not executed to save the number of switching times. However, noise and iron loss increase.
  • Patent Document 4 proposes switching between sine wave PWM control and rectangular wave control. Further, Patent Document 4 discloses that six sub-voltage vectors for stopping supply of a power supply current to one phase coil are used in rectangular wave control. Further, Patent Document 4 discloses that twelve voltage vectors are used in rectangular wave control.
  • Patent Document 1 which is essentially the same as the conventional 120-degree conduction method or 180-degree conduction method of the rectangular wave control, six or twelve voltage vectors have an electrical angle of the rotating magnetic field of 60 degrees or Output in order every 30 degrees. Therefore, a PWM cycle period is not required in this rectangular wave control.
  • the inverter is driven by a sequential space vector pulse width modulation (sequential SVPWM) method.
  • sequential SVPWM space vector pulse width modulation
  • three or more types of current supply periods for applying different phase voltages to the stator coil are formed in each PWM cycle period. Furthermore, these current supply periods do not overlap each other under partial load conditions. Thereby, the resistance loss of the DC power supply that supplies the power supply current to the stator coil can be reduced.
  • the current supply periods when the sum of all the current supply periods in the PWM cycle period is longer than the PWM cycle period, the current supply periods have an arrangement in which the overlap period is the shortest. Thereby, the resistance loss of the DC power supply can be reduced.
  • a plurality of current supply periods except the longest current supply period are preferentially overlapped. Thereby, the resistance loss of the DC power supply can be reduced.
  • the end operation of one current supply period and the start operation of the next current supply period are executed in essentially the same period. Thereby, the surge voltage can be reduced.
  • the relatively long current supply period is arranged before the relatively short current supply period.
  • the surge voltage can be reduced.
  • the reduction in the surge voltage realizes a reduction in the on-resistance of the transistor of the inverter.
  • the power loss of the inverter is reduced.
  • the longest current supply period is located between two relatively short current supply periods. Thereby, high-frequency current loss can be reduced.
  • the inverter consists of two three-phase inverters connected to two star-connected three-phase coils.
  • the four types of current supply periods can be sequentially arranged within the PWM cycle period.
  • the inverter comprises two three-phase inverters connected to a double-ended three-phase coil.
  • three types of current supply periods can be arranged in order within the PWM cycle period.
  • the inverter consists of four three-phase inverters connected to two double-ended three-phase coils.
  • the six types of current supply periods can be sequentially arranged within the PWM cycle period.
  • the star-connected three-phase coils are driven by a space vector PWM method.
  • This space vector PWM method employs at least six sub-voltage vectors to supply phase current to only two of the three phase coils. Thereby, the power loss of the motor device can be reduced in the partial load mode.
  • the two three-phase inverters connected to the double-ended three-phase coil comprise a PWM leg that is PWM modulated and a fixed potential leg that is fixed at the DC link voltage.
  • Each of the three phase coils is connected to a PWM leg and a fixed potential leg.
  • the fixed potential leg and the PWM leg are switched every 180 electrical degrees or 360 electrical degrees.
  • FIG. 1 is a wiring diagram showing a conventional three-phase motor device.
  • FIG. 2 is a schematic wiring diagram of the motor device shown in FIG.
  • FIG. 3 is a wiring diagram illustrating the motor device according to the first embodiment.
  • FIG. 4 is a vector diagram showing the SVPWM method.
  • FIG. 5 is a timing chart showing an example of the sequential SVPWM method.
  • FIG. 6 is a vector diagram showing the SVPWM method of the second embodiment.
  • FIG. 7 is a timing chart showing another example of the sequential SVPWM method.
  • FIG. 8 is a vector diagram showing the SVPWM method of the third embodiment.
  • FIG. 9 is a timing chart showing another example of the sequential SVPWM method.
  • FIG. 10 is a wiring diagram illustrating a motor device according to a fourth embodiment.
  • FIG. 10 is a wiring diagram illustrating a motor device according to a fourth embodiment.
  • FIG. 11 is a vector diagram showing the six-phase voltages shown in FIG.
  • FIG. 12 is a timing chart showing the sequential SVPWM method of the fourth embodiment.
  • FIG. 13 is a wiring diagram illustrating a motor device according to a fifth embodiment.
  • FIG. 14 is a timing chart showing an average waveform of three phase voltage differences applied to three phase coils.
  • FIG. 15 is a schematic wiring diagram illustrating a current supply period of the U-phase H-bridge.
  • FIG. 16 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge.
  • FIG. 17 is a schematic wiring diagram showing a current supply period of the U-phase H-bridge.
  • FIG. 18 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge.
  • FIG. 15 is a schematic wiring diagram illustrating a current supply period of the U-phase H-bridge.
  • FIG. 16 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge.
  • FIG. 17
  • FIG. 19 is a schematic wiring diagram showing a current supply period of the U-phase H-bridge.
  • FIG. 20 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge.
  • FIG. 21 is a schematic wiring diagram showing a current supply period of the U-phase H-bridge.
  • FIG. 22 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge.
  • FIG. 23 is a timing chart for explaining an example of the alternating PWM method.
  • FIG. 24 is a schematic wiring diagram showing a conventional three-phase inverter for driving a star-connected three-phase coil.
  • FIG. 25 is a schematic wiring diagram showing the motor device of this embodiment.
  • FIG. 26 is a wiring diagram illustrating the motor device of the sixth embodiment.
  • FIG. 27 is a schematic wiring diagram showing a U-phase freewheeling current.
  • FIG. 28 is a timing chart for explaining another example of the alternating PWM method.
  • FIG. 29 is a timing chart for explaining the sequential VPWM method.
  • FIG. 30 is a wiring diagram showing a motor device according to the seventh embodiment.
  • FIG. 31 is a timing chart showing a three-phase average voltage waveform output from one three-phase inverter.
  • FIG. 32 is a timing chart for explaining the sequential SVPWM method.
  • FIG. 33 is a schematic wiring diagram showing the motor device of the eighth embodiment.
  • FIG. 34 is a schematic diagram showing an electric vehicle having two motors.
  • FIG. 35 is a schematic wiring diagram showing the serial mode of the ninth embodiment.
  • FIG. 36 is a schematic wiring diagram for illustrating the parallel mode of the ninth embodiment.
  • FIG. 37 is a schematic wiring diagram showing a starter generator according to a tenth embodiment.
  • FIG. 3 is a wiring diagram of the traction motor device for EV.
  • the motor of this motor device has a stator coil composed of three-phase coils 1 and 2 of a star connection type, respectively.
  • the first three-phase coil 1 includes a U-phase coil 1U, a V-phase coil 1V, and a W-phase coil 1W.
  • the second three-phase coil 2 includes a U-phase coil 2U, a V-phase coil 2V, and a W-phase coil 2W.
  • the three-phase coil 1 is connected to a three-phase inverter 3, and the three-phase coil 2 is connected to a three-phase inverter 4.
  • the inverter 3 includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W.
  • the inverter 4 includes a U-phase leg 4U, a V-phase leg 4V, and a W-phase leg 4W.
  • Leg 3U is connected to phase coil 1U
  • leg 3V is connected to phase coil 1V
  • leg 3W is connected to phase coil 1W.
  • Leg 4U is connected to phase coil 2U
  • leg 4V is connected to phase coil 2V
  • leg 4W is connected to phase coil 2W.
  • the inverters 3 and 4 are powered by a DC power supply 7 having an internal resistance r7.
  • the controller 100 controls the three-phase inverters 3 and 4 according to the sequential SVPWM method.
  • leg 3U outputs U-phase voltage VU1, leg 3V outputs V-phase voltage VV1, and leg 3W outputs W-phase voltage VW1.
  • leg 4U outputs a U-phase voltage VU2
  • leg 4V outputs a V-phase voltage VV2
  • leg 4W outputs a W-phase voltage VW2.
  • FIG. 4 is a vector diagram showing the sequential SVPWM method for controlling the inverters 3 and 4.
  • the required output voltage vector is formed by at least one of the six voltage vectors (100, 110, 010, 011, 001, 101) and the two zero vectors (000, 111).
  • the numeral '1' indicates a state where the leg outputs a high-level voltage value
  • the numeral '0' indicates a state where the leg outputs a low-level voltage value.
  • the inverter outputs an output voltage vector (100).
  • the region where the output voltage vector is formed is divided into six phase regions (R1-R6) by the six voltage vectors.
  • An output voltage vector that means a composite vector is formed by a vector sum of two adjacent voltage vectors sandwiching the output voltage vector. Each of these two adjacent voltage vectors is essentially proportional to the length of a current supply period during which a power supply current is supplied from the DC power supply 7 to the inverters 3 and 4.
  • one cycle of a PWM carrier signal having a predetermined PWM carrier frequency is called a PWM cycle period TC.
  • each PWM cycle period TC is 100 microseconds.
  • Each PWM cycle period is generally divided into one or two types of current supply periods and at least one freewheeling period Tf.
  • the freewheeling period Tf during which the DC power supply 7 does not supply a phase current to the stator coil through the inverter 3 corresponds to a zero vector (000 or 111).
  • the magnetic energy stored in the inductance of the phase coil during the current supply period causes a phase current to flow in the direction opposite to the back electromotive force of the phase coil during the free wheeling period Tf.
  • FIG. 5 is a timing chart showing one PWM cycle period TC in the phase region R1 shown in FIG.
  • Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order.
  • the current supply periods T1 and T2 are the current supply periods of the inverter 3, and the current supply periods T3 and T4 are the current supply periods of the inverter 4.
  • the time point t0-t4 is substantially equal to the start time point or the end time point of the current supply period (T1-T4). In the following description, the phase delay of the phase current due to the inductance of the three-phase coils 1 and 2 is ignored.
  • the phase voltage VU1 is 1, the phase voltage VV1 is 1, and the phase voltage VW1 is 0.
  • DC power supply 7 supplies -W phase current -IW to three-phase coil 1.
  • the phase voltage VU1 is 1, the phase voltage VV1 is 0, and the phase voltage VW1 is 0.
  • DC power supply 7 supplies U-phase current IU to three-phase coil 1.
  • the phase voltage VU2 is 1, the phase voltage VV2 is 1, and the phase voltage VW2 is 0.
  • DC power supply 7 supplies -W phase current -IW to three-phase coil 2.
  • the phase voltage VU2 is 1, the phase voltage VV2 is 0, and the phase voltage VW2 is 0.
  • DC power supply 7 supplies U-phase current IU to three-phase coil 2.
  • inverters 3 and 4 output three-phase average voltages in phase with each other.
  • the current supply periods T1 and T2 of the inverter 3 are temporally different from the current supply periods T3 and T4 of the inverter 4.
  • the four current supply periods T1-T4 of the three-phase inverters 3 and 4 are sequentially arranged in each PWM cycle period TC.
  • the current supply periods (T1-T4) overlap.
  • the overlap period is set as short as possible.
  • Each of the three phase coils 1U0, 1V0, and 1W0 shown in FIG. 2 has a winding value N and an electric resistance value r.
  • Each of the six phase coils (1U, 1V, 1W, 2U, 2V, and 2W) shown in FIG. 3 has a winding value N and an electrical resistance value 2r.
  • each phase coil shown in FIG. 3 has a conductor cross-sectional area that is half that of each phase coil shown in FIG. Therefore, the motor shown in FIG. 3 has substantially the same size as the conventional motor shown in FIG.
  • inverters 3 and 4 shown in FIG. 3 have substantially the same circuit scale as inverter 3A shown in FIG. Inverters 3 and 4 shown in FIG. 3 each output half the phase current of inverter 3A shown in FIG. In the following description, it is assumed that inverters 3A, 3, and 4 output an output voltage vector (100).
  • the phase currents IU1 and IU2 are each half of the phase current IU. After all, the sequential SVPWM method is used. According to the above, the power loss Pdc of the DC power supply 7 is halved.
  • the current flowing through one transistor 300 shown in FIG. 2 is equal to the current flowing through one transistor 300 shown in FIG.
  • the upper arm transistor of the inverter 3A is composed of two transistors 300 connected in parallel. Each of the upper arm transistors of the inverters 3 and 4 includes one transistor 300.
  • the surge voltage generated during the transition period during which the upper arm transistor of any of inverters 3 and 4 is turned off is half of the surge voltage generated during the transition period during which the upper arm transistor of inverter 3A is turned off. Therefore, low breakdown voltage type transistors having low on-resistance can be employed for inverters 3 and 4. As a result, inverter loss can be further reduced.
  • FIG. 6 is a vector diagram showing another sequential SVPWM method employed by the inverters 3 and 4 shown in FIG.
  • new voltage vectors (1D0, D10, 01D, 0D1) called sub-voltage vectors are used instead of the six voltage vectors (100, 110, 010, 011, 001, 101) shown in FIG. 4
  • new voltage vectors (1D0, D10, 01D, 0D1) called sub-voltage vectors are used.
  • D01, and 10D are employed.
  • An output voltage vector for generating a rotating magnetic field is formed by at least one of these six sub-voltage vectors and two zero vectors (000, 111).
  • the letter D indicates a state where both the upper arm transistor and the lower arm transistor of the leg are turned off.
  • This leg called the rest leg, cuts off the power supply current of the DC power supply 7.
  • freewheeling current can circulate through the anti-parallel diode in the idle leg. For example, when the U-phase leg outputs "1", the V-phase leg is a pause leg, and the W-phase leg outputs "0", the inverter outputs the sub voltage vector (1D0) as an output voltage vector.
  • Each sub-voltage vector indicates a phase voltage applied to two phase coils connected in series to each other among the three phase coils of the star-connected three-phase coils.
  • a region corresponding to an electrical angle of 360 degrees is divided into six phase regions (R7-R12) by the six sub-voltage vectors described above.
  • An arbitrary output voltage vector is composed of a vector sum of two sub-voltage vectors adjacent to the output voltage vector.
  • the sub voltage vector is substantially proportional to the length of the current supply period during which the DC power supply 7 supplies the power supply current.
  • FIG. 7 is a timing chart showing one PWM cycle period TC in the phase region R7.
  • Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order.
  • the current supply periods T1 and T2 are the current supply periods of the inverter 3
  • the current supply periods T3 and T4 are the current supply periods of the inverter 4.
  • phase voltage VU1 is 1, the phase voltage VV1 is in the rest state (D), and the phase voltage VW1 is 0.
  • Inverter 3 outputs a phase voltage (VU1-VW1).
  • the phase voltage VU1 is 1, the phase voltage VV1 is 0, and the phase voltage VW1 is in the rest state (D).
  • Inverter 3 outputs a phase voltage (VU1-VV1).
  • the phase voltage VU2 is 1, the phase voltage VV2 is in the rest state (D), and the phase voltage VW2 is 0.
  • Inverter 4 outputs a phase voltage (VU2-VW2).
  • the phase voltage VU2 is 1, the phase voltage VV2 is 0, and the phase voltage VW2 is in the rest state (D).
  • Inverter 4 outputs a phase voltage (VU2-VV2).
  • DC power supply 7 does not supply useless phase current to inverters 3 and 4 under partial load conditions. Therefore, it is possible to further reduce the power loss of the DC power supply 7, the inverters 3 and 4, and the stator coil.
  • FIG. 8 is a vector diagram showing another sequential SVPWM method applied to the inverters 3 and 4 shown in FIG.
  • the six voltage vectors (100, 110, 010, 011, 001, 101) shown in FIG. 4 and the six sub-voltage vectors (1D0, D10, 01D, 0D1,. D01, 10D) are used.
  • the six voltage vectors (100, 110, 010, 011, 001, 101) are called main voltage vectors.
  • An output voltage vector for forming a rotating magnetic field is formed using at least one of these twelve voltage vectors and two zero vectors (000, 111).
  • the sub-voltage vector is separated from two adjacent main voltage vectors by an electrical angle of 30 degrees.
  • a region corresponding to an electrical angle of 360 degrees is divided into twelve phase regions by twelve voltage vectors.
  • An arbitrary output voltage vector is composed of a vector sum of one main voltage vector and one sub voltage vector adjacent to the output voltage vector.
  • output voltage vector 1000 shown in FIG. 8 is composed of a vector sum of main voltage vector V110 and sub-voltage vector V1D0.
  • FIG. 9 is a timing chart showing one PWM cycle period TC in one phase region shown in FIG.
  • Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order.
  • the current supply periods T1 and T2 are the current supply periods of the inverter 3
  • the current supply periods T3 and T4 are the current supply periods of the inverter 4.
  • the phase voltage VU1 and the phase voltage VV1 are 1, and the phase voltage VW1 is 0.
  • the inverter 3 supplies a phase current (-IW).
  • the phase voltage VU1 is 1, the phase voltage VV1 is in the rest state (D), and the phase voltage VW1 is 0.
  • Inverter 3 outputs a phase voltage (VU1-VV1).
  • the phase voltage VU2 and the phase voltage VV2 are 1, and the phase voltage VW2 is 0.
  • the inverter 4 supplies a phase current (-IW).
  • the phase voltage VU2 is 1, the phase voltage VV2 is in the rest state (D), and the phase voltage VW2 is 0.
  • Inverter 4 outputs a phase voltage (VU2-VV2).
  • inverters 3 and 4 output output voltage vector 1000 shown in FIG.
  • This embodiment has the same advantages as the second embodiment. Furthermore, according to the sequential SVPWM method of this embodiment, the DC power supply 7 does not supply useless phase currents to the inverters 3 and 4 under partial load conditions. Therefore, it is possible to further reduce the power loss of the DC power supply 7, the inverters 3 and 4, and the stator coil.
  • FIG. 10 A motor device according to a fourth embodiment will be described with reference to FIG.
  • the three-phase coils 1 and 2 shown in FIG. 10 form a star-connected six-phase coil.
  • this star-connected six-phase coil is equivalent to the circuit to which the neutral points of the two three-phase coils 1 and 2 shown in FIG. 3 are connected.
  • Inverters 3 and 4 output two three-phase voltages 180 electrical degrees apart from each other.
  • Leg 2U outputs -U phase voltage VU2
  • leg 2V outputs -V phase voltage VV2
  • leg 2W outputs -W phase voltage VW2.
  • FIG. 11 is a vector diagram showing these six phase voltages.
  • FIG. 12 is a timing chart showing one PWM cycle period TC.
  • Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order.
  • the current supply periods T1 and T2 are the current supply periods of the inverter 3
  • the current supply periods T3 and T4 are the current supply periods of the inverter 4.
  • the time point (t0-t5) is substantially equal to the start time point or the end time point of each current supply period.
  • DC power supply 7 supplies -W phase current -IW to three-phase coil 1.
  • the phase voltage VU1 is 1, the phase voltage VV1 is 0, and the phase voltage VW1 is 0.
  • DC power supply 7 supplies U-phase current IU to three-phase coil 1.
  • the phase voltage VU2 and the phase voltage VV2 are 0, and the phase voltage VW2 is 1.
  • DC power supply 7 supplies W-phase current IW to three-phase coil 2.
  • the phase voltage VU2 is 0, and the phase voltages VV2 and VW2 are 1.
  • DC power supply 7 supplies -U phase current -IU to three-phase coil 2.
  • the inverters 3 and 4 output three-phase average voltages having mutually opposite phases.
  • the current supply periods T1 and T2 have different time positions than the current supply periods T3 and T4.
  • the four current supply periods T1-T4 are sequentially arranged in each PWM cycle period TC.
  • the current supply periods (T1-T4) overlap.
  • FIG. 13 is a wiring diagram showing a double-ended three-phase coil type motor device used as a traction motor of an EV.
  • This traction motor consisting of a three-phase synchronous motor or a three-phase induction motor, has a stator coil 1 consisting of a double-ended three-phase coil.
  • the stator coil 1 connected to the three-phase inverters 3 and 4 includes a U-phase coil 1U, a V-phase coil 1V, and a W-phase coil 1W.
  • Inverter 3 comprises legs 3U, 3V and 3W.
  • Inverter 4 comprises legs 4U, 4V and 4W.
  • the U-phase coil 1U connects the AC terminals of the legs 3U and 4U.
  • the V-phase coil 1V connects the AC terminals of the legs 3V and 4V.
  • the W-phase coil 1W connects the AC terminals of the legs 3W and 4W.
  • the three phase coils 1U, 1V, and 1W are separately connected to three H-bridges each consisting of two legs.
  • FIG. 14 is a timing chart showing an average waveform of three phase voltage differences (VU1-VU2, VV1-VV2, and VW1-VW2) applied to the phase coils 1U, 1V, and 1W.
  • the U-phase voltage difference (VU1-VU2) is applied to the phase coil 1U
  • the V-phase voltage difference (VV1-VV2) is applied to the phase coil 1V
  • the W-phase voltage difference (VW1-VW2) is applied to the phase coil 1W.
  • the controller 100 controls the inverters 3 and 4 according to the sequential SVPWM method.
  • each PWM cycle period TC has a plurality of current supply periods Ts.
  • each PWM cycle period TC has a free wheeling period Tf.
  • the on-duty ratio of the H-bridge is (Ts / (Ts + Tf)).
  • the DC power supply supplies a phase current to the phase coil through the H bridge in the current supply period Ts. During the freewheeling period Tf, the freewheeling current circulates through the phase coil and the H-bridge.
  • FIGS. 15 to 22 are schematic wiring diagrams showing the PWM operation of the U-phase H bridge composed of the legs 3U and 4U.
  • the leg 3U has an upper arm transistor 3UU and a lower arm transistor 3UL.
  • the leg 4U has an upper arm transistor 4UU and a lower arm transistor 4UL.
  • Each of these four transistors has an anti-parallel diode.
  • the operation of legs 3V and 4V and the operation of legs 3W and 4W are essentially the same as the operation of legs 3U and 4U.
  • FIGS. 15 to 18 show the positive half-wave periods P1 and P2 in which the U-phase current flows from the leg 3U to the U-phase coil 1U
  • FIGS. 19 to 22 show the U-phase current flowing from the leg 4U to the U-phase coil 1U.
  • the negative half-wave periods P3 and P4 are shown.
  • positive half-wave periods P1 and P2 are equal to an angle range of 0 to 180 electrical degrees
  • negative half-wave periods P3 and P4 are equal to an angle range of 180 to 0 electrical degrees
  • 15 and 16 show the period P1.
  • Leg 3U is a PWM leg
  • leg 4U is a fixed potential leg.
  • the lower arm transistor 4UL of the leg 4U is always turned on.
  • FIG. 15 shows a current supply period Ts in which the upper arm transistor 3UU is turned on
  • FIG. 16 shows a free wheeling period Tf in which the lower arm transistor 3UL is turned on.
  • FIG. 17 and 18 show the period P2.
  • Leg 4U is a PWM leg and leg 3U is a fixed potential leg.
  • Upper arm transistor 3UU of leg 3U is always turned on.
  • FIG. 17 shows a current supply period Ts in which the lower arm transistor 4UL is turned on
  • FIG. 18 shows a free wheeling period Tf in which the upper arm transistor 4UU is turned on.
  • FIG. 19 and FIG. 20 show the period P3.
  • Leg 4U is a PWM leg and leg 3U is a fixed potential leg.
  • the lower arm transistor 3UL of the leg 3U is always turned on.
  • FIG. 19 shows a current supply period Ts in which the upper arm transistor 4UU is turned on
  • FIG. 20 shows a free wheeling period Tf in which the lower arm transistor 4UL is turned on.
  • FIG. 21 and FIG. 22 show the period P4.
  • Leg 3U is a PWM leg
  • leg 4U is a fixed potential leg.
  • Upper arm transistor 4UU of leg 4U is always turned on.
  • FIG. 21 shows a current supply period Ts during which the lower arm transistor 3UL is turned on
  • FIG. 22 shows a free wheeling period Tf during which the upper arm transistor 3UU is turned on.
  • FIG. 23 is a timing chart showing the state of the six legs 3U-4W.
  • the six legs 3U-4W become PWM legs in odd-numbered periods equal to 360 electrical degrees, and become fixed potential legs in even-numbered periods equal to 360 electrical degrees.
  • the upper arm transistor and the lower arm transistor of the fixed potential leg are alternately turned on every 180 electrical degrees. In other words, the fixed potential leg alternately outputs the DC link voltage Vd and the DC link voltage 0V every 180 electrical degrees.
  • FIG. 24 is a schematic wiring diagram showing a conventional three-phase inverter 3A for driving a conventional star-connected three-phase coil 1A.
  • the motor device shown in FIG. 24 is essentially the same as the motor device shown in FIG.
  • the three-phase inverter 3A includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W.
  • the three-phase coil 1A includes a U-phase coil 34, a V-phase coil 35, and a W-phase coil 36.
  • Each of the phase coils 34-36 includes two coils 200 connected in parallel.
  • Each coil 200 has a predetermined winding value N.
  • Each of the six arms of inverter 3A has two transistors 300 connected in parallel.
  • FIG. 25 is a schematic wiring diagram showing the motor device of this embodiment.
  • This motor device is essentially the same as the motor device shown in FIG.
  • Each of the phase coils 1U-1W is composed of two coils 200 connected in series.
  • Inverters 3 and 4 can output a three-phase voltage having a voltage amplitude twice that of the conventional three-phase inverter shown in FIG.
  • the phase current supplied to each coil 200 shown in FIG. 24 is equal to the phase current supplied to each coil 200 shown in FIG. Therefore, each of phase coils 1U-1W shown in FIG. 25 has twice the number of turns as each of phase coils 34-36 shown in FIG.
  • the double-ended three-phase coil shown in FIG. 25 requires two three-phase inverters 3 and 4.
  • three-phase inverters 3 and 4 each have half the current capacity of three-phase inverter 3A shown in FIG.
  • the two inverters 3 and 4 have essentially the same circuit scale as the three-phase inverter 3A.
  • inverters 3 and 4 have lower power losses compared to inverter 3A. This is because half of the transistors of the three-phase inverters 3 and 4 have a fixed potential leg having no switching loss. Further, according to this alternate PWM driving method, the transistor 300 can have substantially the same temperature by switching between the PWM leg and the fixed potential leg.
  • FIG. 26 is a wiring diagram of this motor device.
  • This motor device is essentially the same as the motor device of the fifth embodiment shown in FIG.
  • the antiparallel connected diodes on the upper arm side of the six legs 3U-4W are respectively connected to the positive electrode 50 of the battery 5 and the positive electrode 60 of the smoothing capacitor 6 through the feedback line 56.
  • Positive electrode 50 of battery 5 is connected to DC terminals 57 of inverters 3 and 4 via DC link line 54.
  • the positive electrode 60 of the smoothing capacitor 6 is connected to a DC terminal 57 through a capacitor line 55.
  • FIG. 27 is a detailed wiring diagram of the U-phase H bridge.
  • Lines 54-56 have wiring inductances 51-53.
  • Leg 3U has an upper arm transistor 301 and a lower arm transistor 302.
  • Leg 4U has an upper arm transistor 303 and a lower arm transistor 304.
  • the anode electrode of the diode 305 is connected to one end of the phase coil 1U.
  • the anode electrode of the diode 306 is connected to the other end of the phase coil 1U.
  • the cathode electrodes of the diodes 305 and 306 are connected to the positive electrodes 50 and 60 of the battery 5 and the smoothing capacitor 6 through the feedback line 56.
  • diode 305 is anti-parallel connected to transistor 301 through lines 54-56.
  • diode 306 is anti-parallel connected to transistor 303 through lines 54-56.
  • FIG. 27 shows a state immediately after the lower arm transistor 304 of the leg 4U is turned off.
  • Upper arm transistor 301 of leg 3U continues to be on.
  • the power supply current supplied to the transistor 301 is cut off. Due to the strong inductance of phase coil 1U, freewheeling current If circulates through diode 306, lines 54-56, and transistor 301. Therefore, when the power supply current to the phase coil 1U is cut off, the wiring inductances 51 and 53 do not generate a surge voltage.
  • the freewheeling current If returns to the positive electrode of the battery 5 and the smoothing capacitor 6 through the anti-parallel diode 305 or 306.
  • the current in the DC link line 54 and the capacitor line 55 is maintained, and the lines 54 and 55 do not generate a surge voltage.
  • a low-voltage transistor having a low on-resistance value can be used as the transistors 301 and 303. Lines 54, 55, and 56 can be adjacent to one another.
  • FIG. 28 is a timing chart showing the state of the six legs 3U-4W.
  • FIG. 28 is essentially the same as FIG. However, according to this embodiment, only the period P2 shown in FIGS. 17 and 18 and the period P4 shown in FIGS. 21 and 22 are used. In other words, the period P2 is used instead of the period P1 shown in FIG. 23, and the period P4 is used instead of the period P3 shown in FIG.
  • each of the upper arm transistors alternately turns on and off every 180 electrical degrees.
  • Each of the lower arm transistors alternately performs PWM switching and off at every electrical angle of 180 degrees.
  • the lower arm transistor of the PWM leg is switched by the PWM method.
  • the upper arm transistor of the PWM leg can be kept off.
  • This embodiment is suitable for an upper arm transistor made of an IGBT.
  • FIG. 29 is a timing chart showing one PWM cycle period TC.
  • a period TU is a current supply period of the phase coil 1U
  • a period TV is a current supply period of the phase coil 1V
  • a period TW is a current supply period of the phase coil 1W.
  • This PWM driving method is called a sequential SVPWM method.
  • the resistance loss of the DC power supply is reduced by reducing the amplitude of the power supply current.
  • the end of one current supply period overlaps with the start of the next current supply period. Thereby, the high-frequency current component included in the power supply current of the DC power supply can be reduced. Further, the surge voltage of the DC link line 54 is reduced.
  • the plurality of current supply periods overlap.
  • the current supply periods TU, TV, and TW are arranged within each PWM cycle period such that the overlap period is shortest.
  • the other two current supply periods except the longest current supply period are preferentially overlapped.
  • the longest current supply period means that the amplitude of the phase current is large. Therefore, an overlap between two relatively short current supply periods can suppress an increase in the amplitude of the power supply current. As a result, an increase in battery loss is suppressed.
  • part (A) shows the arrangement of the current supply periods TU, TV, and TW when the current supply period TV is the longest.
  • Part (B) shows the arrangement of the current supply periods TU, TV, and TW when the current supply period TU is the longest.
  • Part (C) shows the arrangement of the current supply periods TU, TV, and TW when the current supply period TW is the longest.
  • the longest current supply period among the three current supply periods TU, TV, and TW is sandwiched on the time axis from the other two current supply periods.
  • the wiring inductance 52 reduces the so-called recovery loss of the antiparallel diodes on the upper arm side of the inverters 3 and 4.
  • the anti-parallel diode 306 allows a recovery current to flow in a direction opposite to the freewheeling current If.
  • the back electromotive force of the wiring inductance 52 reduces the recovery current.
  • a coil element having an appropriate inductance value instead of the wiring inductance 52.
  • a pair of this coil element and backflow prevention diode 306 connected in series connects DC terminal 57 and phase coil 1U. 27, an independent coil element connected to the DC link line 54 can be used instead of the wiring inductance 52 in order to reduce the recovery loss of the diodes 305 and 306.
  • the upper arm transistors of inverters 3 and 4 carry both current and freewheeling current.
  • the lower arm transistors of inverters 3 and 4 allow only the power supply current to flow.
  • the upper arm transistor has only conduction loss.
  • the lower arm transistor has conduction loss and switching loss. Eventually, the upper arm transistor can have approximately the same loss as the lower arm transistor.
  • the sequential SVPWM method in which the current supply periods TU, TV, and TW do not overlap each other and the simultaneous SVPWM method in which the three-phase currents IU, IV, and IW flow simultaneously are compared. It is assumed that the phase current IU is 1 and the phase currents IV and IW are each 0.5.
  • the DC power supply 7 having the power supply resistance r7 has a power loss (1.5 * r7) in the sequential SVPWM method and has a power loss (4 * r7) in the simultaneous SVPWM method.
  • the sequential SVPWM method can greatly reduce the power loss of the DC power supply compared to the simultaneous SVPWM method.
  • FIG. 30 is a wiring diagram showing a motor device suitable for a large traction motor for an EV bus or an EV truck.
  • the stator coil of this motor device is composed of two double-ended three-phase coils 1 and 2. In other words, the stator coil is divided into two double-ended three-phase coils 1 and 2. Double-ended three-phase coil 1 is connected to three-phase inverters 3 and 4. Double-ended three-phase coil 2 is connected to three-phase inverters 8 and 9.
  • the three-phase coil 1 and the inverters 3 and 4 are the same as in the fifth embodiment shown in FIG.
  • the three-phase coil 2 includes three phase coils 2X, 2Y, and 2W.
  • the phase coil 2X has the same phase as the phase coil 1U
  • the phase coil 2Y has the same phase as the phase coil 1Y
  • the phase coil 2Z has the same phase as the phase coil 1W.
  • phase coil 2X has a phase opposite to phase coil 1U
  • phase coil 2Y has a phase opposite to phase coil 1Y
  • phase coil 2Z has a phase opposite to phase coil 1W.
  • the stator coil of the second example is a double-ended six-phase coil.
  • the six phase coils 1U-2Z have equal winding values.
  • FIG. 31 is a timing chart showing an average waveform of three phase voltage differences (VX-VX2), (VY1-VY2), and (VZ1-VZ2).
  • the controller 100 controls the four inverters 3, 4, 8, and 9 according to the sequential SVPWM method.
  • FIG. 32 is a timing chart for explaining the sequential SVPWM method.
  • the U-phase coil 1U has a current supply period TU
  • the V-phase coil 1V has a current supply period TV
  • the W-phase coil 1W has a current supply period TW.
  • the X-phase coil 1X has a current supply period TX
  • the Y-phase coil 1Y has a current supply period TY
  • the Z-phase coil 1Z has a current supply period TZ.
  • FIG. 32 six current supply periods (TU, TV, TW, TX, TY, and TZ) are arranged in order.
  • the sum of the six current supply periods TU, TV, and TW is longer than the PWM cycle period TC, the plurality of current supply periods overlap.
  • the six current supply periods TU-TZ are arranged in each PWM cycle period TC such that the total length of the overlap period is the shortest. Thereby, the amplitude of the power supply current is reduced, so that the resistance loss of DC power supply 7 is reduced.
  • the end point of the preceding current supply period is substantially equal to the start point of the next current supply period.
  • the transition period in which the power supply current decreases due to the end of the preceding current supply period overlaps with the transition period in which the power supply current increases due to the start of the next current supply period.
  • a relatively long current supply period is sandwiched by relatively short current supply periods.
  • high-frequency current loss of the DC power supply can be reduced.
  • short current supply periods TU and TX are arranged before long current supply periods TV and TY
  • short current supply periods TW and TZ are arranged after current supply periods TV and TY.
  • the AC power loss of the DC power supply can be reduced by reducing the high-frequency current component included in the power supply current.
  • a surge voltage generated at the end of the last current supply period can be reduced.
  • FIG. 33 is essentially the same as FIG.
  • double-ended three-phase coil 1 is accommodated in left-wheel drive motor 11 of electric vehicle 10
  • double-ended three-phase coil 2 is accommodated in right-wheel drive motor 12 of the EV.
  • Each of the motors 11 and 12 is an in-wheel motor.
  • the electric vehicle 10 has front wheels 13 and 14 together with in-wheel motors 11 and 12.
  • the four inverters (3, 4, 8, and 9) are controlled by the sequential SVPWM method described in the sixth embodiment.
  • the power supply current supplied to the motors 11 and 12 is substantially equal except for the vehicle turning period.
  • FIG. 35 is a wiring diagram showing a double-ended three-phase coil type motor device used as a traction motor for a large EV.
  • the U-phase coils 1U and 2U shown in FIG. 35 are realized by dividing the U-phase coil 1U shown in FIG.
  • V-phase coils 1V and 2V shown in FIG. 35 are realized by dividing V-phase coil 1V shown in FIG.
  • the W-phase coils 1W and 2W shown in FIG. 35 are realized by dividing the W-phase coil 1W shown in FIG.
  • Each of the six phase coils 1U-2W has an equal winding value.
  • the three-phase inverter 3 includes legs 3U, 3V, and 3W.
  • the three-phase inverter 4 includes legs 4U, 4V, and 3W.
  • Phase coils 1U and 2U connected in series are connected to legs 3U and 4U.
  • Phase coils 1V and 2V connected in series are connected to legs 3V and 4V.
  • Phase coils 1W and 2W connected in series are connected to legs 3W and 4W.
  • Leg 3U outputs a phase voltage VU1, and leg 4U outputs a phase voltage VU2.
  • Leg 3V outputs a phase voltage VV1, and leg 4V outputs a phase voltage VV2.
  • Leg 3W outputs phase voltage VW1, and leg 4W outputs phase voltage VW2.
  • a third three-phase inverter 8 including a U-phase leg 8U, a V-phase leg 8V, and a W-phase leg 8W is added.
  • the AC terminal 81 of the leg 8U is connected to a connection point between the phase coils 1U and 2U.
  • the AC terminal 82 of the leg 8V is connected to a connection point of the phase coils 1V and 2V.
  • the AC terminal 83 of the leg 8W is connected to a connection point between the phase coils 1W and 2W.
  • Leg 8U outputs a phase voltage VU3
  • leg 8V outputs a phase voltage VV3
  • leg 8W outputs a phase voltage VW3.
  • the controller 100 has a serial mode and a parallel mode. All transistors of the inverter 8 are turned off in the series mode. Therefore, the serial mode is executed by inverters 3 and 4. This series mode is the same as the operation of the motor device of the fifth embodiment. Therefore, in the serial mode, the sequential SVPWM method and the alternating PWM method described in the fifth embodiment can be executed.
  • phase coils 1U and 2U In the series mode shown in FIG. 35, U-phase current IU flows through phase coils 1U and 2U, V-phase current IV flows through phase coils 1V and 2V, and W-phase current IW flows through phase coils 1W and 2W.
  • Each of the six phase coils 1U-2W has a back electromotive force in the opposite direction to the phase current flowing through itself.
  • the parallel mode will be described with reference to FIG. FIG. 36 is essentially the same as FIG. 35 except for the directions of the three phase currents IU, IV, and IW flowing through the three-phase coil 1.
  • the three phase coils 1U, 1V, and 1W of the three-phase coil 1 have counter electromotive forces in the opposite directions as compared with the series mode. Therefore, the phase current IU1 of the phase coil 1U flows in the opposite direction as compared with the phase current IU2 of the phase coil 2U.
  • the phase current IV1 of the phase coil 1V flows in a direction opposite to the phase current IV2 of the phase coil 2V.
  • the phase current IW1 of the phase coil 1W flows in a direction opposite to the phase current IW2 of the phase coil 2W.
  • the three-phase motor shown in FIG. 35 is changed to a six-phase motor shown in FIG.
  • the direction switching of the phase currents IU1, ⁇ IV1, and ⁇ IW1 will be described with reference to FIG.
  • the U-phase H bridge composed of the legs 8U and 4U supplies the U-phase current IU2 to the phase coil 2U.
  • the V-phase H bridge composed of the legs 8V and 4V supplies the V-phase current IV2 to the phase coil 2V.
  • the W-phase H bridge including the legs 8W and 4W supplies the W-phase current IW2 to the phase coil 2W.
  • a U-phase H bridge composed of leg 8U and leg 3U supplies U-phase current IU1 to phase coil 1U.
  • a V-phase H bridge consisting of leg 8V and leg 3V supplies V-phase current IV1 to phase coil 1V.
  • the W-phase H bridge composed of the leg 8W and the leg 3W supplies the W-phase current IW1 to the phase coil 1W.
  • the parallel mode has essentially the same operation as the fifth embodiment. Therefore, in the parallel mode, the sequential SVPWM method and the alternating PWM method described in the fifth embodiment can be executed.
  • each of the six H-bridges consists of a PWM leg and a fixed potential leg.
  • Legs 8U, 8V, and 8W are always fixed potential legs, and the other six legs 3U-4W are always PWM legs.
  • the upper arm transistor of leg 8U is always turned on, and legs 3U and 4U are driven by the PWM method.
  • the lower arm transistor of leg 8U is always turned on, and legs 3U and 4U are driven by the PWM method.
  • the positive half-wave period and the negative half-wave period each correspond to an electrical angle of 180 degrees.
  • the inverter 8 In the parallel mode, the inverter 8 must supply twice the current to the three-phase coils 1 and 2 as compared to the inverter 3 or 4. However, each leg of the inverter 8 is always driven as a fixed potential leg. As a result, the switching loss of the inverter 8 is greatly reduced.
  • Each current supply period of the six PWM legs (3U, 3V, 3W, 4U, 4V, 4W) can be sequentially arranged within one PWM cycle period.
  • these six current supply periods can have the arrangement shown in FIG.
  • the serial mode the number of turns of the stator coil is twice as large as in the parallel mode. Therefore, the series mode can reduce battery loss in a high current region. It is also possible to switch the number of stator poles by switching the serial mode to the parallel mode.
  • This stator pole number switching method is suitable for an induction motor that does not require switching of the number of rotor poles.
  • FIG. 37 is a wiring diagram illustrating a double-ended three-phase coil motor device as a starter generator of a vehicle.
  • FIG. 37 is essentially the same as FIG. However, FIG. 37 employs a three-phase diode rectifier 8 instead of the three-phase inverter 8 shown in FIG.
  • the motor device of FIG. 37 has a motor mode and a generator mode.
  • the motor mode is the same as the serial mode of the ninth embodiment.
  • the generator mode is essentially the same as the parallel mode of the ninth embodiment. However, in generator mode, the U-phase voltages of phase coils 1U and 2U are rectified in parallel.
  • the V-phase voltages of the phase coils 1V and 2V are rectified in parallel.
  • the W-phase voltages of the phase coils 1W and 2W are rectified in parallel. Therefore, the copper loss of the stator coil in the generator mode is approximately 25% as compared with the motor mode.
  • the starter generator becomes an alternator.
  • the alternator has a low-speed power generation mode essentially equal to the series mode and a high-speed power generation mode essentially equal to the parallel mode. This high-speed power generation mode has 1/4 copper loss compared to the low-speed power generation mode.

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Abstract

Provided is a variable-speed motor device in which a sequential SVPWM method is performed that sequentially arranges three or more types of current supply periods within each PWM cycle period. Stator coils comprise star-connected three-phase coils or double-ended three-phase coils. Phase currents are supplied to only two phase coils among the star-connected three-phase coils in two current supply periods formed in each PWM cycle period. The double-ended three-phase coils are connected to the PWM leg and fixed potential leg of a three-phase inverter. The PWM leg and the fixed potential leg are switched to each other at every electrical angle of 180° or 360°.

Description

可変速モータ装置Variable speed motor device
本発明は、PWM駆動される可変速モータ装置に関し、特に、バッテリ給電可変速PWMモータ装置に関する。 The present invention relates to a PWM-driven variable speed motor device, and more particularly to a battery-powered variable speed PWM motor device.
一般に、星形接続３相コイルを駆動する３相インバータが電気自動車（EV)のインバータとして採用される。また、ダブルエンデッド３相コイルを駆動するダブルインバータも提案されている。さらに、６相コイルを駆動する６相インバータも提案されている。６相コイルは２つの３相コイルと本質的に等しい。 Generally, a three-phase inverter that drives a star-connected three-phase coil is employed as an inverter of an electric vehicle (EV). A double inverter that drives a double-ended three-phase coil has also been proposed. Further, a six-phase inverter for driving a six-phase coil has also been proposed. A six-phase coil is essentially equivalent to two three-phase coils.
特許文献１は、ダブルインバータ式モータ装置の１つを提案している。特許文献１によれば、１つの相コイルに接続される２個のレグは、互いに反対位相をもつPWM信号により駆動される。特許文献２は、ダブルインバータ式モータ装置のステータコイルに第３高調波電流を注入することを提案している。 Patent Document 1 proposes one of the double inverter type motor devices. According to Patent Literature 1, two legs connected to one phase coil are driven by PWM signals having opposite phases. Patent Document 2 proposes injecting a third harmonic current into a stator coil of a double inverter type motor device.
特許文献３は、３相整流器及び短絡トランジスタがさらに追加されたダブルインバータ式モータ装置を提案する。このモータ装置によれば、３つの相コイルの各第１端子は第１の３相インバータに接続され、３つの相コイルの各第２端子は第２の３相インバータ及び３相全波整流器の両方に並列に接続される。短絡トランジスタは３相整流器の一対の直流端子を短絡する。 Patent Document 3 proposes a double inverter type motor device in which a three-phase rectifier and a short-circuit transistor are further added. According to this motor device, the first terminals of the three-phase coils are connected to the first three-phase inverter, and the second terminals of the three-phase coils are connected to the second three-phase inverter and the three-phase full-wave rectifier. Both are connected in parallel. The short-circuit transistor short-circuits a pair of DC terminals of the three-phase rectifier.
バッテリ、インバータ、及びトラクションモータからなるEVパワートレインの重要な問題は電力損失の低減である。従来のEVのバッテリ及びインバータは冷却装置により強力に冷却されねばならない。しかし、この冷却装置は追加の電力損失を発生する。 An important issue in EV powertrains consisting of batteries, inverters and traction motors is reducing power loss. Conventional EV batteries and inverters must be strongly cooled by cooling devices. However, this cooling device causes additional power loss.
EV用バッテリの抵抗値はその劣化により増加する。この抵抗値と比例するバッテリ損失が増加する時、バッテリ温度が上昇する。バッテリ劣化はバッテリの高温により促進される。 The resistance of an EV battery increases due to its deterioration. As battery loss increases in proportion to this resistance, battery temperature rises. Battery deterioration is accelerated by the high temperature of the battery.
一般に、電気自動車（EV）のインバータは高電圧タイプのトランジスタを採用する。しかし、高電圧タイプのトランジスタは低電圧タイプのトランジスタと比べて高いオン抵抗値をもつ。したがって、EVインバータの電力損失は一般用の低電圧インバータと比べて増加される。 Generally, an inverter of an electric vehicle (EV) employs a high-voltage type transistor. However, a high-voltage transistor has a higher on-resistance than a low-voltage transistor. Therefore, the power loss of the EV inverter is increased as compared with the general-purpose low-voltage inverter.
従来のEV用インバータのための直流電源の電力損失が図１を参照して説明される。図略の星形接続３相コイルに接続される３相インバータ３Aは、U相レグ３U、V相レグ３V、及びW相レグ３Wからなる。図１は、レグ３Uの上アームトランジスタ３UU、レグ３Vの下アームトランジスタ３VL、及びレグ３Wの下アームトランジスタ３WLがオンしている状態を示す。インバータ３Aはバッテリ５及び平滑キャパシタ６に接続されている。トランジスタ３UU、３VL、及び３WLはオン抵抗値rをもつ。バッテリ５は抵抗値r５をもち、キャパシタ６は抵抗値r６をもつ。 The power loss of a DC power supply for a conventional EV inverter will be described with reference to FIG. The three-phase inverter 3A connected to the unillustrated star-connected three-phase coil includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W. FIG. 1 shows a state in which the upper arm transistor 3UU of the leg 3U, the lower arm transistor 3VL of the leg 3V, and the lower arm transistor 3WL of the leg 3W are on. Inverter 3A is connected to battery 5 and smoothing capacitor 6. The transistors 3UU, 3VL, and 3WL have an on-resistance r. The battery 5 has a resistance value r5, and the capacitor 6 has a resistance value r6.
図２は、図１のモータ装置の１つの模式配線図である。等価抵抗値r７をもつ直流電源７はバッテリ５及びキャパシタ６からなる。インバータ３Aの６個のアームはそれぞれ、並列接続された２つのトランジスタ３００をもつ。インバータ３Aに接続されるステータコイル１Aは星形接続された３つの相コイル１U０、１V０、及び１W０からなる。 FIG. 2 is one schematic wiring diagram of the motor device of FIG. The DC power supply 7 having an equivalent resistance value r7 includes a battery 5 and a capacitor 6. Each of the six arms of inverter 3A has two transistors 300 connected in parallel. The stator coil 1A connected to the inverter 3A includes three phase coils 1U0, 1V0, and 1W0 connected in a star shape.
一般に、３相モータを駆動するための３相インバータは、正弦波形の３相電圧を出力するための正弦波制御、又は、矩形波形の３相電圧を出力するための矩形波制御を実行する。一般に、正弦波制御はパルス幅変調方式（PWM法）を採用する。このPWM法において、インバータの各レグは、PWMサイクル周期毎に電流供給期間及びフリーホィーリング期間をもつ。電流供給期間は、直流電源がレグに電源電流を供給する期間を意味する。フリーホィーリング期間は直流電源がレグに電源電流を供給しない期間を意味する。PWM法として、三角波PWM方式及び空間ベクトルPWM（SVPWM）方式が良く知られている。SVPWM法は、それぞれ電流供給期間に本質的に相当する６個の電圧ベクトルを使用する。 In general, a three-phase inverter for driving a three-phase motor executes sine-wave control for outputting a sine-wave three-phase voltage or rectangular wave control for outputting a three-phase voltage having a rectangular waveform. Generally, the sine wave control employs a pulse width modulation method (PWM method). In this PWM method, each leg of the inverter has a current supply period and a freewheeling period every PWM cycle period. The current supply period means a period during which the DC power supply supplies a power supply current to the leg. The free wheeling period means a period during which the DC power supply does not supply a power supply current to the leg. As the PWM method, a triangular wave PWM method and a space vector PWM (SVPWM) method are well known. The SVPWM method uses six voltage vectors, each essentially corresponding to a current supply period.
正弦波制御は、１つの円に沿って回転する出力電圧ベクトルを出力する。この出力電圧ベクトルは、隣接する２つの電圧ベクトルのベクトル和（合成ベクトル）となる。電圧ベクトルの長さは電流供給期間に本質的に相当する。正弦波インバータによれば、２種類の電圧ベクトルに相当する２種類の電流供給期間が各PWMサイクル周期内に設定される。正弦波インバータの問題は、インバータのスイッチング損失である。 Sine wave control outputs an output voltage vector that rotates along one circle. This output voltage vector is a vector sum (composite vector) of two adjacent voltage vectors. The length of the voltage vector essentially corresponds to the current supply period. According to the sine wave inverter, two types of current supply periods corresponding to two types of voltage vectors are set in each PWM cycle period. A problem with sine wave inverters is the switching losses of the inverter.
矩形波制御は、１つの電圧ベクトルからもう１つの隣接電圧ベクトルへジャンプする出力電圧ベクトルを出力する。正弦波制御と比べてインバータのスイッチング損失を低減する。矩形波制御において、出力電圧ベクトルは、回転磁界の電気角６０度ごとに切り替えられる上記６個の電圧ベクトルの１つからなる。すなわち、矩形波制御によれば、２つの電圧ベクトルのベクトル和により任意の出力電圧ベクトルを合成するというPWM法はスイッチング回数節約のために実行されない。けれども、騒音や鉄損が増加する。 Square wave control outputs an output voltage vector that jumps from one voltage vector to another adjacent voltage vector. The switching loss of the inverter is reduced as compared with the sine wave control. In the rectangular wave control, the output voltage vector is composed of one of the above-mentioned six voltage vectors that are switched every 60 electrical degrees of the rotating magnetic field. That is, according to the rectangular wave control, the PWM method of synthesizing an arbitrary output voltage vector by the vector sum of two voltage vectors is not executed to save the number of switching times. However, noise and iron loss increase.
特許文献４は、正弦波PWM制御と矩形波制御とを切り替えることを提案する。さらに、特許文献４は、１つの相コイルへの電源電流の供給を停止する６個のサブ電圧ベクトルを矩形波制御において使用することを開示する。さらに、特許文献４は、１２個の電圧ベクトルを矩形波制御において使用することを開示する。 Patent Document 4 proposes switching between sine wave PWM control and rectangular wave control. Further, Patent Document 4 discloses that six sub-voltage vectors for stopping supply of a power supply current to one phase coil are used in rectangular wave control. Further, Patent Document 4 discloses that twelve voltage vectors are used in rectangular wave control.
しかしながら、従来の矩形波制御の１２０度通電方式又は１８０度通電方式と本質的に同じである特許文献１の矩形波制御において、６個又は１２個の電圧ベクトルが回転磁界の電気角６０度又は３０度毎に順番に出力される。このため、PWMサイクル周期はこの矩形波制御において必要とされない。 However, in the rectangular wave control of Patent Document 1 which is essentially the same as the conventional 120-degree conduction method or 180-degree conduction method of the rectangular wave control, six or twelve voltage vectors have an electrical angle of the rotating magnetic field of 60 degrees or Output in order every 30 degrees. Therefore, a PWM cycle period is not required in this rectangular wave control.
特開２００９-３０３２９８号公報JP 2009-303298 A 特開２０１５-１２２９５０号公報JP 2015-122950 A 特開２０１４-５４０９４号公報JP 2014-54094 A 特開２００５-１６０１８３公報JP 2005-160183 A
本発明の一つの目的は、電力損失の低減が可能なPWM駆動可変速モータ装置を提供することである。本発明の他の目的は電力損失の低減が可能なバッテリ給電PWM駆動可変速モータ装置を提供することである。 An object of the present invention is to provide a PWM drive variable speed motor device capable of reducing power loss. Another object of the present invention is to provide a battery-powered PWM drive variable speed motor device capable of reducing power loss.
本発明の第１の様相によれば、インバータは、シーケンシャル空間ベクトルパルス幅変調（シーケンシャルSVPWM)法で駆動される。このシーケンシャルSVPWM法によれば、それぞれ異なる相電圧をステータコイルに印加するための３種類以上の電流供給期間が各PWMサイクル期間内に形成される。さらに、これらの電流供給期間は、部分負荷条件において互いにオーバーラップしない。これにより、ステータコイルに電源電流を供給する直流電源の抵抗損失を低減することができる。 According to a first aspect of the invention, the inverter is driven by a sequential space vector pulse width modulation (sequential SVPWM) method. According to the sequential SVPWM method, three or more types of current supply periods for applying different phase voltages to the stator coil are formed in each PWM cycle period. Furthermore, these current supply periods do not overlap each other under partial load conditions. Thereby, the resistance loss of the DC power supply that supplies the power supply current to the stator coil can be reduced.
１つの好適態様によれば、PWMサイクル期間内の全ての電流供給期間の和がPWMサイクル期間より長い時、電流供給期間は、オーバーラップ期間が最短となる配列をもつ。これにより、直流電源の抵抗損失を低減することができる。もう１つの好適態様によれば、PWMサイクル期間内の全ての電流供給期間の和がPWMサイクル期間より長い時、最も長い電流供給期間を除く複数の電流供給期間を優先的にオーバーラップさせられる。これにより、直流電源の抵抗損失を低減することができる。もう１つの好適態様によれば、１つの電流供給期間の終了動作と次の電流供給期間の開始動作は本質的に同じ期間に実行される。これにより、サージ電圧を低減することができる。 According to one preferred embodiment, when the sum of all the current supply periods in the PWM cycle period is longer than the PWM cycle period, the current supply periods have an arrangement in which the overlap period is the shortest. Thereby, the resistance loss of the DC power supply can be reduced. According to another preferred embodiment, when the sum of all the current supply periods in the PWM cycle period is longer than the PWM cycle period, a plurality of current supply periods except the longest current supply period are preferentially overlapped. Thereby, the resistance loss of the DC power supply can be reduced. According to another preferred embodiment, the end operation of one current supply period and the start operation of the next current supply period are executed in essentially the same period. Thereby, the surge voltage can be reduced.
もう１つの好適態様によれば、相対的に長い電流供給期間は相対的に短い電流供給期間の前に配置される。これにより、サージ電圧を低減することができる。このサージ電圧の低減は、インバータのトランジスタのオン抵抗の低減を実現する。これにより、インバータの電力損失が低減される。もう１つの好適態様によれば、最も長い電流供給期間は２つの相対的に短い電流供給期間の間に配置される。これにより、高周波電流損失を低減することができる。 According to another preferred embodiment, the relatively long current supply period is arranged before the relatively short current supply period. Thereby, the surge voltage can be reduced. The reduction in the surge voltage realizes a reduction in the on-resistance of the transistor of the inverter. Thereby, the power loss of the inverter is reduced. According to another preferred embodiment, the longest current supply period is located between two relatively short current supply periods. Thereby, high-frequency current loss can be reduced.
もう１つの好適態様によれば、インバータは、２つの星形接続３相コイルに接続される２つの３相インバータからなる。これにより、４種類の電流供給期間をPWMサイクル期間内に順番に配置することができる。もう１つの好適態様によれば、インバータは、ダブルエンデッド３相コイルに接続される２つの３相インバータからなる。これにより、３種類の電流供給期間をPWMサイクル期間内に順番に配置することができる。もう１つの好適態様によれば、インバータは、２つのダブルエンデッド３相コイルに接続される４つの３相インバータからなる。これにより、６種類の電流供給期間をPWMサイクル期間内に順番に配置することができる。 According to another preferred embodiment, the inverter consists of two three-phase inverters connected to two star-connected three-phase coils. Thus, the four types of current supply periods can be sequentially arranged within the PWM cycle period. According to another preferred embodiment, the inverter comprises two three-phase inverters connected to a double-ended three-phase coil. Thus, three types of current supply periods can be arranged in order within the PWM cycle period. According to another preferred embodiment, the inverter consists of four three-phase inverters connected to two double-ended three-phase coils. Thus, the six types of current supply periods can be sequentially arranged within the PWM cycle period.
本発明の第２の様相によれば、星形接続３相コイルは空間ベクトルPWM法で駆動される。この空間ベクトルPWM法は、３つの相コイルのうちの２つにだけに相電流を供給するための６個のサブ電圧ベクトルを少なくとも採用する。これにより、部分負荷モードにおいてモータ装置の電力損失を低減することができる。 According to a second aspect of the invention, the star-connected three-phase coils are driven by a space vector PWM method. This space vector PWM method employs at least six sub-voltage vectors to supply phase current to only two of the three phase coils. Thereby, the power loss of the motor device can be reduced in the partial load mode.
本発明の第３の様相によれば、ダブルエンデッド３相コイルに接続される２つの３相インバータは、PWM変調されるPWMレグと、DCリンク電圧に固定される固定電位レグとからなる。３つの相コイルはそれぞれ、PWMレグと固定電位レグとに接続される。固定電位レグ及びPWMレグは、電気角１８０度又は３６０度毎に切り替えられる。これにより、インバータの電源電圧利用率が倍増され、インバータの電力損失が低減される。 According to a third aspect of the invention, the two three-phase inverters connected to the double-ended three-phase coil comprise a PWM leg that is PWM modulated and a fixed potential leg that is fixed at the DC link voltage. Each of the three phase coils is connected to a PWM leg and a fixed potential leg. The fixed potential leg and the PWM leg are switched every 180 electrical degrees or 360 electrical degrees. As a result, the power supply voltage utilization rate of the inverter is doubled, and the power loss of the inverter is reduced.
図１は、従来の３相モータ装置を示す配線図である。FIG. 1 is a wiring diagram showing a conventional three-phase motor device. 図２は、図１に示されるモータ装置の模式配線図である。FIG. 2 is a schematic wiring diagram of the motor device shown in FIG. 図３は、第１実施例のモータ装置を示す配線図である。FIG. 3 is a wiring diagram illustrating the motor device according to the first embodiment. 図４は、SVPWM法を示すベクトル図である。FIG. 4 is a vector diagram showing the SVPWM method. 図５は、シーケンシャルSVPWM法の一例を示すタイミングチャートである。FIG. 5 is a timing chart showing an example of the sequential SVPWM method. 図６は、第２実施例のSVPWM法を示すベクトル図である。FIG. 6 is a vector diagram showing the SVPWM method of the second embodiment. 図７は、シーケンシャルSVPWM法のもう一つの例を示すタイミングチャートである。FIG. 7 is a timing chart showing another example of the sequential SVPWM method. 図８は、第３実施例のSVPWM法を示すベクトル図である。FIG. 8 is a vector diagram showing the SVPWM method of the third embodiment. 図９は、シーケンシャルSVPWM法のもう一つの例を示すタイミングチャートである。FIG. 9 is a timing chart showing another example of the sequential SVPWM method. 図１０は、第４実施例のモータ装置を示す配線図である。FIG. 10 is a wiring diagram illustrating a motor device according to a fourth embodiment. 図１１は、図１０に示される６相電圧を示すベクトル図である。FIG. 11 is a vector diagram showing the six-phase voltages shown in FIG. 図１２は第４実施例のシーケンシャルSVPWM法を示すタイミングチャートである。FIG. 12 is a timing chart showing the sequential SVPWM method of the fourth embodiment. 図１３は、第５実施例のモータ装置を示す配線図である。FIG. 13 is a wiring diagram illustrating a motor device according to a fifth embodiment. 図１４は、３つの相コイルに印加される３つの相電圧差の平均波形をを示すタイミングチャートである。FIG. 14 is a timing chart showing an average waveform of three phase voltage differences applied to three phase coils. 図１５は、U相Hブリッジの電流供給期間を示す模式配線図である。FIG. 15 is a schematic wiring diagram illustrating a current supply period of the U-phase H-bridge. 図１６は、U相Hブリッジのフリーホィーリング期間を示す模式配線図である。FIG. 16 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge. 図１７は、U相Hブリッジの電流供給期間を示す模式配線図である。FIG. 17 is a schematic wiring diagram showing a current supply period of the U-phase H-bridge. 図１８は、U相Hブリッジのフリーホィーリング期間を示す模式配線図である。FIG. 18 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge. 図１９は、U相Hブリッジの電流供給期間を示す模式配線図である。FIG. 19 is a schematic wiring diagram showing a current supply period of the U-phase H-bridge. 図２０は、U相Hブリッジのフリーホィーリング期間を示す模式配線図である。FIG. 20 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge. 図２１は、U相Hブリッジの電流供給期間を示す模式配線図である。FIG. 21 is a schematic wiring diagram showing a current supply period of the U-phase H-bridge. 図２２は、U相Hブリッジのフリーホィーリング期間を示す模式配線図である。FIG. 22 is a schematic wiring diagram showing a free wheeling period of the U-phase H bridge. 図２３は、交互PWM法の一例を説明するためのタイミングチャートである。FIG. 23 is a timing chart for explaining an example of the alternating PWM method. 図２４は、従来の星形接続３相コイルを駆動する３相インバータを示す模式配線図である。FIG. 24 is a schematic wiring diagram showing a conventional three-phase inverter for driving a star-connected three-phase coil. 図２５は、この実施例のモータ装置を示す模式配線図である。FIG. 25 is a schematic wiring diagram showing the motor device of this embodiment. 図２６は、第６実施例のモータ装置を示す配線図である。FIG. 26 is a wiring diagram illustrating the motor device of the sixth embodiment. 図２７は、U相フリーホィーリング電流を示す模式配線図である。FIG. 27 is a schematic wiring diagram showing a U-phase freewheeling current. 図２８は、交互PWM法の他例を説明するためのタイミングチャートである。FIG. 28 is a timing chart for explaining another example of the alternating PWM method. 図２９は、シーケンシャルVPWM法を説明するためのタイミングチャートである。FIG. 29 is a timing chart for explaining the sequential VPWM method. 図３０は第７実施例のモータ装置を示す配線図である。FIG. 30 is a wiring diagram showing a motor device according to the seventh embodiment. 図３１は１つの３相インバータが出力する３相平均電圧波形を示すタイミングチャートである。FIG. 31 is a timing chart showing a three-phase average voltage waveform output from one three-phase inverter. 図３２はシーケンシャルSVPWM法を説明するためのタイミングチャートである。FIG. 32 is a timing chart for explaining the sequential SVPWM method. 図３３は第８実施例のモータ装置を示す模式配線図である。FIG. 33 is a schematic wiring diagram showing the motor device of the eighth embodiment. 図３４は２つのモータをもつ電気自動車を示す模式図である。FIG. 34 is a schematic diagram showing an electric vehicle having two motors. 図３５は第９実施例の直列モードを示すための模式配線図である。FIG. 35 is a schematic wiring diagram showing the serial mode of the ninth embodiment. 図３６は第９実施例の並列モードを示すための模式配線図である。FIG. 36 is a schematic wiring diagram for illustrating the parallel mode of the ninth embodiment. 図３７は第１０実施例のスタータジエネレータを示す模式配線図である。FIG. 37 is a schematic wiring diagram showing a starter generator according to a tenth embodiment.
　　　　　　　　　　第１実施例
第１実施例のモータ装置が図３を参照して説明される。図３は、EV用トラクションモータ装置の配線図である。このモータ装置のモータは、それぞれ星形接続タイプの３相コイル１及び２からなるステータコイルをもつ。第１の３相コイル１はU相コイル１U、V相コイル１V、及びW相コイル１Wからなる。第２の３相コイル２はU相コイル２U、V相コイル２V、及びW相コイル２Wからなる。３相コイル１は３相インバータ３に接続され、３相コイル２は３相インバータ４に接続されている。 First Embodiment A motor device according to a first embodiment will be described with reference to FIG. FIG. 3 is a wiring diagram of the traction motor device for EV. The motor of this motor device has a stator coil composed of three- phase coils 1 and 2 of a star connection type, respectively. The first three-phase coil 1 includes a U-phase coil 1U, a V-phase coil 1V, and a W-phase coil 1W. The second three-phase coil 2 includes a U-phase coil 2U, a V-phase coil 2V, and a W-phase coil 2W. The three-phase coil 1 is connected to a three-phase inverter 3, and the three-phase coil 2 is connected to a three-phase inverter 4.
インバータ３はU相レグ３U、V相レグ３V、及びW相レグ３Wからなる。インバータ４はU相レグ４U、V相レグ４V、及びW相レグ４Wからなる。レグ３Uは相コイル１Uに接続され、レグ３Vは相コイル１Vに接続され、レグ３Wは相コイル１Wに接続されている。レグ４Uは相コイル２Uに接続され、レグ４Vは相コイル２Vに接続され、レグ４Wは相コイル２Wに接続されている。インバータ３および４は、内部抵抗r７をもつ直流電源７によって給電されている。コントローラ１００は、シーケンシャルSVPWM法により３相インバータ３および４を制御する。レグ３UはU相電圧VU１を出力し、レグ３VはV相電圧VV１を出力し、レグ３WはW相電圧VW１を出力する。同様に、レグ４UはU相電圧VU２を出力し、レグ４VはV相電圧VV２を出力し、レグ４WはW相電圧VW２を出力する。 The inverter 3 includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W. The inverter 4 includes a U-phase leg 4U, a V-phase leg 4V, and a W-phase leg 4W. Leg 3U is connected to phase coil 1U, leg 3V is connected to phase coil 1V, and leg 3W is connected to phase coil 1W. Leg 4U is connected to phase coil 2U, leg 4V is connected to phase coil 2V, and leg 4W is connected to phase coil 2W. The inverters 3 and 4 are powered by a DC power supply 7 having an internal resistance r7. The controller 100 controls the three- phase inverters 3 and 4 according to the sequential SVPWM method. Leg 3U outputs U-phase voltage VU1, leg 3V outputs V-phase voltage VV1, and leg 3W outputs W-phase voltage VW1. Similarly, leg 4U outputs a U-phase voltage VU2, leg 4V outputs a V-phase voltage VV2, and leg 4W outputs a W-phase voltage VW2.
図４は、インバータ３及び４を制御するシーケンシャルSVPWM法を示すベクトル図である。要求される出力電圧ベクトルが６つの電圧ベクトル（１００、１１０、０１０、０１１、００１、１０１）と、２つのゼロベクトル（０００、１１１）の少なくとも１つにより形成される。数字’１’はレグがハイレベル電圧値を出力する状態を示し、数字’０’は、レグがローレベル電圧値を出力する状態を示す。たとえば、U相レグが’１’を出力し、V相レグ及びW相レグが’０’を出力する時、インバータは出力電圧ベクトル（１００）を出力する。 FIG. 4 is a vector diagram showing the sequential SVPWM method for controlling the inverters 3 and 4. The required output voltage vector is formed by at least one of the six voltage vectors (100, 110, 010, 011, 001, 101) and the two zero vectors (000, 111). The numeral '1' indicates a state where the leg outputs a high-level voltage value, and the numeral '0' indicates a state where the leg outputs a low-level voltage value. For example, when the U-phase leg outputs '1' and the V-phase leg and W-phase leg output '0', the inverter outputs an output voltage vector (100).
出力電圧ベクトルが形成される領域は、上記６個の電圧ベクトルにより６個の位相領域（R１-R６）に分割される。合成ベクトルを意味する出力電圧ベクトルは、この出力電圧ベクトルを挟む２つの隣接電圧ベクトルのベクトル和からなる。これら２つの隣接電圧ベクトルはそれぞれ、直流電源７からインバータ３および４へ電源電流を供給する期間である電流供給期間の長さに本質的に比例する。 The region where the output voltage vector is formed is divided into six phase regions (R1-R6) by the six voltage vectors. An output voltage vector that means a composite vector is formed by a vector sum of two adjacent voltage vectors sandwiching the output voltage vector. Each of these two adjacent voltage vectors is essentially proportional to the length of a current supply period during which a power supply current is supplied from the DC power supply 7 to the inverters 3 and 4.
SVPWM法によれば、所定のPWMキャリヤ周波数をもつPWMキャリヤ信号の１周期はPWMサイクル期間TCと呼ばれる。たとえばPWMキャリヤ周波数が１０kHzである時、各PWMサイクル期間TCはそれぞれ１００マイクロ秒となる。各PWMサイクル期間は、１つ又は２つの種類の電流供給期間と少なくとも１つのフリーホィーリング期間Tfとに一般に分割される。直流電源７がインバータ３を通じてステータコイルに相電流を供給しないフリーホィーリング期間Tfは、ゼロベクトル（０００又は１１１）に相当する。電流供給期間において相コイルのインダクタンスに蓄積された磁気エネルギーは、フリーホィーリング期間Tfにおいてこの相コイルの逆起電力と反対向きに相電流を流す。 According to the SVPWM method, one cycle of a PWM carrier signal having a predetermined PWM carrier frequency is called a PWM cycle period TC. For example, when the PWM carrier frequency is 10 kHz, each PWM cycle period TC is 100 microseconds. Each PWM cycle period is generally divided into one or two types of current supply periods and at least one freewheeling period Tf. The freewheeling period Tf during which the DC power supply 7 does not supply a phase current to the stator coil through the inverter 3 corresponds to a zero vector (000 or 111). The magnetic energy stored in the inductance of the phase coil during the current supply period causes a phase current to flow in the direction opposite to the back electromotive force of the phase coil during the free wheeling period Tf.
図５は、図４に示される位相領域R１における１つのPWMサイクル期間TCを示すタイミングチャートである。各PWMサイクル期間TCは、順番に配置された４種類の電流供給期間（T１-T４）をもつ。電流供給期間T１及びT２はインバータ３の電流供給期間であり、電流供給期間T３及びT４はインバータ４の電流供給期間である。時点t０-t４は電流供給期間（T１-T４）の開始時点又は終了時点にほぼ等しい。以下の説明において、３相コイル１及び２のインダクタンスによる相電流の位相遅れは無視される。 FIG. 5 is a timing chart showing one PWM cycle period TC in the phase region R1 shown in FIG. Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order. The current supply periods T1 and T2 are the current supply periods of the inverter 3, and the current supply periods T3 and T4 are the current supply periods of the inverter 4. The time point t0-t4 is substantially equal to the start time point or the end time point of the current supply period (T1-T4). In the following description, the phase delay of the phase current due to the inductance of the three- phase coils 1 and 2 is ignored.
電流供給期間T１において、相電圧VU１は１であり、相電圧VV１は１であり、相電圧VW１は０である。直流電源７は-W相電流-IWを３相コイル１に供給する。電流供給期間T２において、相電圧VU１は１であり、相電圧VV１は０であり、相電圧VW１は０である。直流電源７はU相電流IUを３相コイル１に供給する。電流供給期間T３において、相電圧VU２は１であり、相電圧VV２は１であり、相電圧VW２は０である。直流電源７は-W相電流-IWを３相コイル２に供給する。電流供給期間T４において、相電圧VU２は１であり、相電圧VV２は０であり、相電圧VW２は０である。直流電源７はU相電流IUを３相コイル２に供給する。 In the current supply period T1, the phase voltage VU1 is 1, the phase voltage VV1 is 1, and the phase voltage VW1 is 0. DC power supply 7 supplies -W phase current -IW to three-phase coil 1. In the current supply period T2, the phase voltage VU1 is 1, the phase voltage VV1 is 0, and the phase voltage VW1 is 0. DC power supply 7 supplies U-phase current IU to three-phase coil 1. In the current supply period T3, the phase voltage VU2 is 1, the phase voltage VV2 is 1, and the phase voltage VW2 is 0. DC power supply 7 supplies -W phase current -IW to three-phase coil 2. In the current supply period T4, the phase voltage VU2 is 1, the phase voltage VV2 is 0, and the phase voltage VW2 is 0. DC power supply 7 supplies U-phase current IU to three-phase coil 2.
結局、インバータ３及び４は互いに同相の３相平均電圧を出力する。しかし、インバータ３の電流供給期間T１及びT２は、インバータ４の電流供給期間T３及びT４と時間的に異なる。図５によれば、３相インバータ３及び４の４つの電流供給期間T１-T４は各PWMサイクル期間TC内において順番に配置される。けれども、高いトルク値が要求される時、電流供給期間（T１-T４）はオーバーラップする。言い換えれば、電流供給期間（T１-T４）の和がPWMサイクル期間TCより長い時、電流供給期間（T１-T４）はオーバーラップする。けれども、オーバーラップ期間は、できるだけ短く設定される。PWMサイクル期間TC内に配置された３個以上の電流供給期間が部分負荷条件において互いにオーバーラップしないこのSVPWM法は、この明細書においてシーケンシャルSVPWM法と呼ばれる。 Eventually, inverters 3 and 4 output three-phase average voltages in phase with each other. However, the current supply periods T1 and T2 of the inverter 3 are temporally different from the current supply periods T3 and T4 of the inverter 4. According to FIG. 5, the four current supply periods T1-T4 of the three- phase inverters 3 and 4 are sequentially arranged in each PWM cycle period TC. However, when a high torque value is required, the current supply periods (T1-T4) overlap. In other words, when the sum of the current supply periods (T1-T4) is longer than the PWM cycle period TC, the current supply periods (T1-T4) overlap. However, the overlap period is set as short as possible. This SVPWM method in which three or more current supply periods arranged in the PWM cycle period TC do not overlap each other under partial load conditions is referred to as a sequential SVPWM method in this specification.
シーケンシャルSVPWM法の１つの利点が図２及び図３を参照して説明される。図２に示される３つの相コイル１U０、１V０、及び１W０はそれぞれ、巻数値Nと電気抵抗値rをもつ。図３に示される６つの相コイル（１U、１V、１W、２U、２V、及び２W）はそれぞれ、巻数値Nと電気抵抗値２rをもつ。たとえば、図３に示される各相コイルは、図２に示される各相コイルと比べて半分の導体断面積をもつ。したがって、図３に示されるモータは、図２に示される従来のモータとほぼ等しいサイズをもつ。 One advantage of the sequential SVPWM method is described with reference to FIGS. Each of the three phase coils 1U0, 1V0, and 1W0 shown in FIG. 2 has a winding value N and an electric resistance value r. Each of the six phase coils (1U, 1V, 1W, 2U, 2V, and 2W) shown in FIG. 3 has a winding value N and an electrical resistance value 2r. For example, each phase coil shown in FIG. 3 has a conductor cross-sectional area that is half that of each phase coil shown in FIG. Therefore, the motor shown in FIG. 3 has substantially the same size as the conventional motor shown in FIG.
図３に示される２つのインバータ３及び４の１２個のアームはそれぞれ、１個のトランジスタ３００をもつ。したがって、図３に示されるインバータ３及び４は、図２に示されるインバータ３Aとほぼ等しい回路規模をもつ。図３に示されるインバータ３及び４はそれぞれ、図２に示されるインバータ３Aの半分の相電流を出力する。以下の説明において、インバータ３A、３、及び４は、出力電圧ベクトル（１００）を出力することが仮定される。図２に示される直流電源７は電力損失Pdc（＝r７*IU*IU）を発生する。図３に示される直流電源７は電力損失Pdc（＝r７*（IU１*IU１+IU２*IU２）を発生する。相電流IU１及びIU２はそれぞれ、相電流IUの半分である。結局、シーケンシャルSVPWM法によれば、直流電源７の電力損失Pdcは半分となる。 Each of the twelve arms of the two inverters 3 and 4 shown in FIG. 3 has one transistor 300. Therefore, inverters 3 and 4 shown in FIG. 3 have substantially the same circuit scale as inverter 3A shown in FIG. Inverters 3 and 4 shown in FIG. 3 each output half the phase current of inverter 3A shown in FIG. In the following description, it is assumed that inverters 3A, 3, and 4 output an output voltage vector (100). The DC power supply 7 shown in FIG. 2 generates a power loss Pdc (= r7 * IU * IU). 3 generates a power loss Pdc (= r7 * (IU1 * IU1 + IU2 * IU2). The phase currents IU1 and IU2 are each half of the phase current IU. After all, the sequential SVPWM method is used. According to the above, the power loss Pdc of the DC power supply 7 is halved.
このシーケンシャルSVPWM法のもう１つの利点が説明される。図２に示される１つのトランジスタ３００を流れる電流は、図３に示される１つのトランジスタ３００を流れる電流と等しい。インバータ３Aの上アームトランジスタは、並列接続された２つのトランジスタ３００からなる。インバータ３及び４の上アームトランジスタはそれぞれ、１つのトランジスタ３００からなる。 Another advantage of this sequential SVPWM method is explained. The current flowing through one transistor 300 shown in FIG. 2 is equal to the current flowing through one transistor 300 shown in FIG. The upper arm transistor of the inverter 3A is composed of two transistors 300 connected in parallel. Each of the upper arm transistors of the inverters 3 and 4 includes one transistor 300.
したがって、インバータ３及び４のいずれかの上アームトランジスタがオフされる過渡期間に発生するサージ電圧は、インバータ３Aの上アームトランジスタがオフされる過渡期間に発生するサージ電圧の半分となる。したがって、インバータ３および４は、低いオン抵抗をもつ低耐圧タイプのトランジスタを採用することができる。その結果、インバータ損失をさらに低減することができる。 Therefore, the surge voltage generated during the transition period during which the upper arm transistor of any of inverters 3 and 4 is turned off is half of the surge voltage generated during the transition period during which the upper arm transistor of inverter 3A is turned off. Therefore, low breakdown voltage type transistors having low on-resistance can be employed for inverters 3 and 4. As a result, inverter loss can be further reduced.
　　　　　　　　　　第２実施例
第２実施例のモータ装置が図６を参照して説明される。図６は図３に示されるインバータ３及び４により採用されるもう１つのシーケンシャルSVPWM法を示すベクトル図である。この実施例によれば、図４に示される６つの電圧ベクトル（１００、１１０、０１０、０１１、００１、１０１）の代わりに、サブ電圧ベクトルと呼ばれる新たな電圧ベクトル（１D０、D１０、０１D、０D１、D０１、１０D）が採用される。回転磁界形成のための出力電圧ベクトルは、これらの６つのサブ電圧ベクトル及び２つのゼロベクトル（０００、１１１）の少なくとも１つで形成される。 Second Embodiment A motor device according to a second embodiment will be described with reference to FIG. FIG. 6 is a vector diagram showing another sequential SVPWM method employed by the inverters 3 and 4 shown in FIG. According to this embodiment, instead of the six voltage vectors (100, 110, 010, 011, 001, 101) shown in FIG. 4, new voltage vectors (1D0, D10, 01D, 0D1) called sub-voltage vectors are used. , D01, and 10D) are employed. An output voltage vector for generating a rotating magnetic field is formed by at least one of these six sub-voltage vectors and two zero vectors (000, 111).
このシーケンシャルSVPWM法において、文字Dは、レグの上アームトランジスタ及び下アームトランジスタの両方がオフとなる状態を示す。休止レグと呼ばれるこのレグは、直流電源７の電源電流を遮断する。しかし、フリーホィーリング電流は休止レグの逆並列ダイオードを通じて循環することができる。たとえば、U相レグが’１’を出力し、V相レグが休止レグであり、W相レグが’０’を出力する時、インバータはサブ電圧ベクトル（１D０）を出力電圧ベクトルとして出力する。 In the sequential SVPWM method, the letter D indicates a state where both the upper arm transistor and the lower arm transistor of the leg are turned off. This leg, called the rest leg, cuts off the power supply current of the DC power supply 7. However, freewheeling current can circulate through the anti-parallel diode in the idle leg. For example, when the U-phase leg outputs "1", the V-phase leg is a pause leg, and the W-phase leg outputs "0", the inverter outputs the sub voltage vector (1D0) as an output voltage vector.
各サブ電圧ベクトルは、星形接続３相コイルの３つの相コイルのうち、互いに直列接続された２つの相コイルに印加される相電圧を示す。電気角３６０度に相当する領域は、上記された６個のサブ電圧ベクトルにより６個の位相領域（R７-R１２）に分割される。任意の出力電圧ベクトルは、この出力電圧ベクトルに隣接する２つのサブ電圧ベクトルのベクトル和からなる。サブ電圧ベクトルは、直流電源７が電源電流を供給する電流供給期間の長さにほぼ比例する。 Each sub-voltage vector indicates a phase voltage applied to two phase coils connected in series to each other among the three phase coils of the star-connected three-phase coils. A region corresponding to an electrical angle of 360 degrees is divided into six phase regions (R7-R12) by the six sub-voltage vectors described above. An arbitrary output voltage vector is composed of a vector sum of two sub-voltage vectors adjacent to the output voltage vector. The sub voltage vector is substantially proportional to the length of the current supply period during which the DC power supply 7 supplies the power supply current.
図７は、位相領域R７における１つのPWMサイクル期間TCを示すタイミングチャートである。各PWMサイクル期間TCは、順番に配置された４種類の電流供給期間（T１-T４）をもつ。電流供給期間T１及びT２はインバータ３の電流供給期間であり、電流供給期間T３及びT４はインバータ４の電流供給期間である。 FIG. 7 is a timing chart showing one PWM cycle period TC in the phase region R7. Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order. The current supply periods T1 and T2 are the current supply periods of the inverter 3, and the current supply periods T3 and T4 are the current supply periods of the inverter 4.
電流供給期間T１において、相電圧VU１は１であり、相電圧VV１は休止状態（D)であり、相電圧VW１は０である。インバータ３は相電圧（VU１-VW１）を出力する。電流供給期間T２において、相電圧VU１は１であり、相電圧VV１は０であり、相電圧VW１は休止状態（D)である。インバータ３は相電圧（VU１-VV１）を出力する。電流供給期間T３において、相電圧VU２は１であり、相電圧VV２は休止状態（D)であり、相電圧VW２は０である。インバータ４は相電圧（VU２-VW２）を出力する。電流供給期間T４において、相電圧VU２は１であり、相電圧VV２は０であり、相電圧VW２は休止状態（D)である。インバータ４は相電圧（VU２-VV２）を出力する。 In the current supply period T1, the phase voltage VU1 is 1, the phase voltage VV1 is in the rest state (D), and the phase voltage VW1 is 0. Inverter 3 outputs a phase voltage (VU1-VW1). In the current supply period T2, the phase voltage VU1 is 1, the phase voltage VV1 is 0, and the phase voltage VW1 is in the rest state (D). Inverter 3 outputs a phase voltage (VU1-VV1). In the current supply period T3, the phase voltage VU2 is 1, the phase voltage VV2 is in the rest state (D), and the phase voltage VW2 is 0. Inverter 4 outputs a phase voltage (VU2-VW2). In the current supply period T4, the phase voltage VU2 is 1, the phase voltage VV2 is 0, and the phase voltage VW2 is in the rest state (D). Inverter 4 outputs a phase voltage (VU2-VV2).
この実施例は第１実施例と同じ利点をもつ。さらに、この実施例によれば、直流電源７が部分負荷条件において無駄な相電流をインバータ３および４へ供給しない。このため、直流電源７、インバータ３及び４、及びステータコイルの電力損失をさらに低減することができる。 This embodiment has the same advantages as the first embodiment. Further, according to this embodiment, DC power supply 7 does not supply useless phase current to inverters 3 and 4 under partial load conditions. Therefore, it is possible to further reduce the power loss of the DC power supply 7, the inverters 3 and 4, and the stator coil.
　　　　　　　　　　第３実施例
第３実施例のモータ装置が図８を参照して説明される。図８は図３に示されるインバータ３及び４に適用されるもう１つのシーケンシャルSVPWM法を示すベクトル図である。この実施例によれば、図４に示される６つの電圧ベクトル（１００、１１０、０１０、０１１、００１、１０１）と、図６に示される６つのサブ電圧ベクトル（１D０、D１０、０１D、０D１、D０１、１０D）からなる１２個の電圧ベクトルが使用される。６つの電圧ベクトル（１００、１１０、０１０、０１１、００１、１０１）はメイン電圧ベクトルと呼ばれる。回転磁界形成のための出力電圧ベクトルは、これらの１２個の電圧ベクトルと２つのゼロベクトル（０００、１１１）の少なくとも１つを用いて形成される。 Third Embodiment A motor device according to a third embodiment will be described with reference to FIG. FIG. 8 is a vector diagram showing another sequential SVPWM method applied to the inverters 3 and 4 shown in FIG. According to this embodiment, the six voltage vectors (100, 110, 010, 011, 001, 101) shown in FIG. 4 and the six sub-voltage vectors (1D0, D10, 01D, 0D1,. D01, 10D) are used. The six voltage vectors (100, 110, 010, 011, 001, 101) are called main voltage vectors. An output voltage vector for forming a rotating magnetic field is formed using at least one of these twelve voltage vectors and two zero vectors (000, 111).
サブ電圧ベクトルは２つの隣接メイン電圧ベクトルから電気角３０度だけ離れている。電気角３６０度に相当する領域は、１２個の電圧ベクトルにより１２個の位相領域に分割される。任意の出力電圧ベクトルは、この出力電圧ベクトルに隣接する１つのメイン電圧ベクトルと１つのサブ電圧ベクトルのベクトル和からなる。たとえば、図８に示される出力電圧ベクトル１０００は、メイン電圧ベクトルV１１０とサブ電圧ベクトルV１D０のベクトル和からなる。 The sub-voltage vector is separated from two adjacent main voltage vectors by an electrical angle of 30 degrees. A region corresponding to an electrical angle of 360 degrees is divided into twelve phase regions by twelve voltage vectors. An arbitrary output voltage vector is composed of a vector sum of one main voltage vector and one sub voltage vector adjacent to the output voltage vector. For example, output voltage vector 1000 shown in FIG. 8 is composed of a vector sum of main voltage vector V110 and sub-voltage vector V1D0.
図９は、図８に示される１つの位相領域における１つのPWMサイクル期間TCを示すタイミングチャートである。各PWMサイクル期間TCは、順番に配置された４種類の電流供給期間（T１-T４）をもつ。電流供給期間T１及びT２はインバータ３の電流供給期間であり、電流供給期間T３及びT４はインバータ４の電流供給期間である。 FIG. 9 is a timing chart showing one PWM cycle period TC in one phase region shown in FIG. Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order. The current supply periods T1 and T2 are the current supply periods of the inverter 3, and the current supply periods T3 and T4 are the current supply periods of the inverter 4.
電流供給期間T１において、相電圧VU１及び相電圧VV１は１であり、相電圧VW１は０である。インバータ３は相電流（-IW）を供給する。電流供給期間T２において、相電圧VU１は１であり、相電圧VV１は休止状態（D)であり、相電圧VW１は０である。インバータ３は相電圧（VU１-VV１）を出力する。電流供給期間T３において、相電圧VU２及び相電圧VV２は１であり、相電圧VW２は０である。インバータ４は相電流（-IW）を供給する。電流供給期間T４において、相電圧VU２は１であり、相電圧VV２は休止状態（D)であり、相電圧VW２は０である。インバータ４は相電圧（VU２-VV２）を出力する。結局、インバータ３および４は、図８に示される出力電圧ベクトル１０００を出力する。 In the current supply period T1, the phase voltage VU1 and the phase voltage VV1 are 1, and the phase voltage VW1 is 0. The inverter 3 supplies a phase current (-IW). In the current supply period T2, the phase voltage VU1 is 1, the phase voltage VV1 is in the rest state (D), and the phase voltage VW1 is 0. Inverter 3 outputs a phase voltage (VU1-VV1). In the current supply period T3, the phase voltage VU2 and the phase voltage VV2 are 1, and the phase voltage VW2 is 0. The inverter 4 supplies a phase current (-IW). In the current supply period T4, the phase voltage VU2 is 1, the phase voltage VV2 is in the rest state (D), and the phase voltage VW2 is 0. Inverter 4 outputs a phase voltage (VU2-VV2). Eventually, inverters 3 and 4 output output voltage vector 1000 shown in FIG.
この実施例は第２実施例と同じ利点をもつ。さらに、この実施例のシーケンシャルSVPWM法によれば、直流電源７が部分負荷条件において無駄な相電流をインバータ３および４へ供給しない。このため、直流電源７、インバータ３及び４、及びステータコイルの電力損失をさらに低減することができる。 This embodiment has the same advantages as the second embodiment. Furthermore, according to the sequential SVPWM method of this embodiment, the DC power supply 7 does not supply useless phase currents to the inverters 3 and 4 under partial load conditions. Therefore, it is possible to further reduce the power loss of the DC power supply 7, the inverters 3 and 4, and the stator coil.
　　　　　　　　　　第４実施例
第４実施例のモータ装置が図１０を参照して説明される。図１０に示される３相コイル１及び２は星形接続６相コイルを形成する。言い換えれば、この星形接続６相コイルは、図３に示される２つの３相コイル１及び２の中性点が接続されている回路に等しい。インバータ３及び４は互いに電気角１８０度離れた２つの３相電圧を出力する。レグ２Uは-U相電圧VU２を出力し、レグ２Vは-V相電圧VV２を出力し、レグ２Wは-W相電圧VW２を出力する。図１１は、これらの６個の相電圧を示すベクトル図である。 Fourth Embodiment A motor device according to a fourth embodiment will be described with reference to FIG. The three- phase coils 1 and 2 shown in FIG. 10 form a star-connected six-phase coil. In other words, this star-connected six-phase coil is equivalent to the circuit to which the neutral points of the two three- phase coils 1 and 2 shown in FIG. 3 are connected. Inverters 3 and 4 output two three-phase voltages 180 electrical degrees apart from each other. Leg 2U outputs -U phase voltage VU2, leg 2V outputs -V phase voltage VV2, and leg 2W outputs -W phase voltage VW2. FIG. 11 is a vector diagram showing these six phase voltages.
図１２は、１つのPWMサイクル期間TCを示すタイミングチャートである。各PWMサイクル期間TCは、順番に配置された４種類の電流供給期間（T１-T４）をもつ。電流供給期間T１及びT２はインバータ３の電流供給期間であり、電流供給期間T３及びT４はインバータ４の電流供給期間である。時点（t０-t５）は各電流供給期間の開始時点又は終了時点にほぼ等しい。 FIG. 12 is a timing chart showing one PWM cycle period TC. Each PWM cycle period TC has four types of current supply periods (T1-T4) arranged in order. The current supply periods T1 and T2 are the current supply periods of the inverter 3, and the current supply periods T3 and T4 are the current supply periods of the inverter 4. The time point (t0-t5) is substantially equal to the start time point or the end time point of each current supply period.
電流供給期間T１において、相電圧VU１及び相電圧VV１は１であり、相電圧VW１は０である。直流電源７は-W相電流-IWを３相コイル１に供給する。電流供給期間T２において、相電圧VU１は１であり、相電圧VV１は０であり、相電圧VW１は０である。直流電源７はU相電流IUを３相コイル１に供給する。電流供給期間T３において、相電圧VU２及び相電圧VV２は０であり、相電圧VW２は１である。直流電源７はW相電流IWを３相コイル２に供給する。電流供給期間T４において、相電圧VU２は０であり、相電圧VV２及び相電圧VW２は１である。直流電源７は-U相電流-IUを３相コイル２に供給する。 In the current supply period T1, the phase voltage VU1 and the phase voltage VV1 are 1, and the phase voltage VW1 is 0. DC power supply 7 supplies -W phase current -IW to three-phase coil 1. In the current supply period T2, the phase voltage VU1 is 1, the phase voltage VV1 is 0, and the phase voltage VW1 is 0. DC power supply 7 supplies U-phase current IU to three-phase coil 1. In the current supply period T3, the phase voltage VU2 and the phase voltage VV2 are 0, and the phase voltage VW2 is 1. DC power supply 7 supplies W-phase current IW to three-phase coil 2. In the current supply period T4, the phase voltage VU2 is 0, and the phase voltages VV2 and VW2 are 1. DC power supply 7 supplies -U phase current -IU to three-phase coil 2.
結局、インバータ３及び４は互いに逆相の３相平均電圧を出力する。しかし、電流供給期間T１及びT２は電流供給期間T３及びT４と比べて異なる時間位置をもつ。４つの電流供給期間T１-T４は各PWMサイクル期間TC内において順番に配置される。電流供給期間（T１-T４）の和がPWMサイクル期間TCより長い時、電流供給期間（T１-T４）はオーバーラップする。この実施例は、第１実施例と本質的に同じ利点をもつ。 As a result, the inverters 3 and 4 output three-phase average voltages having mutually opposite phases. However, the current supply periods T1 and T2 have different time positions than the current supply periods T3 and T4. The four current supply periods T1-T4 are sequentially arranged in each PWM cycle period TC. When the sum of the current supply periods (T1-T4) is longer than the PWM cycle period TC, the current supply periods (T1-T4) overlap. This embodiment has essentially the same advantages as the first embodiment.
　　　　　　　　　　第５実施例
第５実施例のモータ装置が図１３を参照して説明される。図１３は、EVのトラクションモータとして用いられるダブルエンデッド３相コイル式モータ装置を示す配線図である。３相同期モータ又は３相誘導モータからなるこのトラクションモータはダブルエンデッド３相コイルからなるステータコイル１をもつ。３相インバータ３及び４に接続されるステータコイル１は、U相コイル１U、V相コイル１V、及びW相コイル１Wからなる。 Fifth Embodiment A motor device according to a fifth embodiment will be described with reference to FIG. FIG. 13 is a wiring diagram showing a double-ended three-phase coil type motor device used as a traction motor of an EV. This traction motor, consisting of a three-phase synchronous motor or a three-phase induction motor, has a stator coil 1 consisting of a double-ended three-phase coil. The stator coil 1 connected to the three- phase inverters 3 and 4 includes a U-phase coil 1U, a V-phase coil 1V, and a W-phase coil 1W.
インバータ３はレグ３U、３V及び３Wからなる。インバータ４はレグ４U、４V及び４Wからなる。U相コイル１Uはレグ３U及び４Uの交流端子を接続する。V相コイル１Vはレグ３V及び４Vの交流端子を接続する。W相コイル１Wはレグ３W及び４Wの交流端子を接続する。結局、３つの相コイル１U、１V、及び１Wは、それぞれ２つのレグからなる３つのHブリッジに別々に接続される。 Inverter 3 comprises legs 3U, 3V and 3W. Inverter 4 comprises legs 4U, 4V and 4W. The U-phase coil 1U connects the AC terminals of the legs 3U and 4U. The V-phase coil 1V connects the AC terminals of the legs 3V and 4V. The W-phase coil 1W connects the AC terminals of the legs 3W and 4W. Eventually, the three phase coils 1U, 1V, and 1W are separately connected to three H-bridges each consisting of two legs.
DCリンク電圧Vdがインバータ３及び４に印加されている。レグ３Uは相電圧VU１を出力し、レグ３Vは相電圧VV１を出力し、レグ３W１は相電圧VW１を出力する。レグ４Uは相電圧VU２を出力し、レグ４Vは相電圧VV２を出力し、レグ４Wは相電圧VW２を出力する。図１４は、相コイル１U、１V、及び１Wに印加される３つの相電圧差（VU１-VU２、VV１-VV２、及びVW１-VW２）の平均波形を示すタイミングチャートである。U相電圧差（VU１-VU２）が相コイル１Uに印加され、V相電圧差（VV１-VV２）が相コイル１Vに印加され、W相電圧差（VW１-VW２）が相コイル１Wに印加される。コントローラ１００はインバータ３及び４をシーケンシャルSVPWM法により制御する。 A DC link voltage Vd is applied to inverters 3 and 4. Leg 3U outputs phase voltage VU1, leg 3V outputs phase voltage VV1, and leg 3W1 outputs phase voltage VW1. Leg 4U outputs a phase voltage VU2, leg 4V outputs a phase voltage VV2, and leg 4W outputs a phase voltage VW2. FIG. 14 is a timing chart showing an average waveform of three phase voltage differences (VU1-VU2, VV1-VV2, and VW1-VW2) applied to the phase coils 1U, 1V, and 1W. The U-phase voltage difference (VU1-VU2) is applied to the phase coil 1U, the V-phase voltage difference (VV1-VV2) is applied to the phase coil 1V, and the W-phase voltage difference (VW1-VW2) is applied to the phase coil 1W. You. The controller 100 controls the inverters 3 and 4 according to the sequential SVPWM method.
この実施例によれば、各Hブリッジの２つのレグの１つだけがPWM制御される。このレグはPWMレグと呼ばれる。各Hブリッジの２つのレグの他の１つはDCリンク電圧Vd又は０Vを出力する。このレグは固定電位レグと呼ばれる。電源電圧を等価的に倍増可能なこの動作方式は交互PWM法と呼ばれる。各PWMサイクル期間TCはそれぞれ、複数の電流供給期間Tsをもつ。一般に、各PWMサイクル期間TCはフリーホィーリング期間Tfをもつ。Hブリッジのオンデユーティ比は（Ts/（Ts＋Tf））となる。直流電源は、電流供給期間TsにおいてHブリッジを通じて相コイルに相電流を供給する。フリーホィーリング期間Tfにおいて、フリーホィーリング電流が相コイル及びHブリッジを通じて循環する。 According to this embodiment, only one of the two legs of each H-bridge is PWM controlled. This leg is called the PWM leg. The other one of the two legs of each H-bridge outputs a DC link voltage Vd or 0V. This leg is called the fixed potential leg. This operation method capable of equivalently doubling the power supply voltage is called an alternating PWM method. Each PWM cycle period TC has a plurality of current supply periods Ts. Generally, each PWM cycle period TC has a free wheeling period Tf. The on-duty ratio of the H-bridge is (Ts / (Ts + Tf)). The DC power supply supplies a phase current to the phase coil through the H bridge in the current supply period Ts. During the freewheeling period Tf, the freewheeling current circulates through the phase coil and the H-bridge.
交互PWM法が図１５-図２２を参照して説明される。図１５-図２２は、レグ３U及び４UからなるU相HブリッジのPWM動作を示す模式配線図である。レグ３Uは上アームトランジスタ３UU及び下アームトランジスタ３ULをもつ。レグ４Uは上アームトランジスタ４UU及び下アームトランジスタ４ULをもつ。これら４つのトランジスタはそれぞれ、逆並列ダイオードを有している。レグ３V及び４Vの動作及びレグ３W及び４Wの動作は、レグ３U及び４Uの動作と本質的に同じである。 The alternate PWM method will be described with reference to FIGS. FIGS. 15 to 22 are schematic wiring diagrams showing the PWM operation of the U-phase H bridge composed of the legs 3U and 4U. The leg 3U has an upper arm transistor 3UU and a lower arm transistor 3UL. The leg 4U has an upper arm transistor 4UU and a lower arm transistor 4UL. Each of these four transistors has an anti-parallel diode. The operation of legs 3V and 4V and the operation of legs 3W and 4W are essentially the same as the operation of legs 3U and 4U.
図１５-図１８は、レグ３UからU相コイル１UへU相電流を流す正半波期間P１及びP２を示し、図１９-図２２は、レグ４UからU相コイル１UへU相電流を流す負半波期間P３及びP４を示す。図１４において、正半波期間P１及びP２は電気角０度-１８０度の角度範囲に等しく、負半波期間P３及びP４は電気角１８０度-０度の角度範囲に等しい。図１５及び図１６は期間P１を示す。レグ３UはPWMレグとなり、レグ４Uは固定電位レグとなる。レグ４Uの下アームトランジスタ４ULが常にオンされる。図１５は上アームトランジスタ３UUがオンされる電流供給期間Tsを示し、図１６は下アームトランジスタ３ULがオンされるフリーホィーリング期間Tfを示す。 FIGS. 15 to 18 show the positive half-wave periods P1 and P2 in which the U-phase current flows from the leg 3U to the U-phase coil 1U, and FIGS. 19 to 22 show the U-phase current flowing from the leg 4U to the U-phase coil 1U. The negative half-wave periods P3 and P4 are shown. In FIG. 14, positive half-wave periods P1 and P2 are equal to an angle range of 0 to 180 electrical degrees, and negative half-wave periods P3 and P4 are equal to an angle range of 180 to 0 electrical degrees. 15 and 16 show the period P1. Leg 3U is a PWM leg, and leg 4U is a fixed potential leg. The lower arm transistor 4UL of the leg 4U is always turned on. FIG. 15 shows a current supply period Ts in which the upper arm transistor 3UU is turned on, and FIG. 16 shows a free wheeling period Tf in which the lower arm transistor 3UL is turned on.
図１７及び図１８は、期間P２を示す。レグ４UはPWMレグとなり、レグ３Uは固定電位レグとなる。レグ３Uの上アームトランジスタ３UUが常にオンされる。図１７は下アームトランジスタ４ULがオンされる電流供給期間Tsを示し、図１８は上アームトランジスタ４UUがオンされるフリーホィーリング期間Tfを示す。 17 and 18 show the period P2. Leg 4U is a PWM leg and leg 3U is a fixed potential leg. Upper arm transistor 3UU of leg 3U is always turned on. FIG. 17 shows a current supply period Ts in which the lower arm transistor 4UL is turned on, and FIG. 18 shows a free wheeling period Tf in which the upper arm transistor 4UU is turned on.
図１９及び図２０は、期間P３を示す。レグ４UはPWMレグとなり、レグ３Uは固定電位レグとなる。レグ３Uの下アームトランジスタ３ULが常にオンされる。図１９は上アームトランジスタ４UUがオンされる電流供給期間Tsを示し、図２０は下アームトランジスタ４ULがオンされるフリーホィーリング期間Tfを示す。 FIG. 19 and FIG. 20 show the period P3. Leg 4U is a PWM leg and leg 3U is a fixed potential leg. The lower arm transistor 3UL of the leg 3U is always turned on. FIG. 19 shows a current supply period Ts in which the upper arm transistor 4UU is turned on, and FIG. 20 shows a free wheeling period Tf in which the lower arm transistor 4UL is turned on.
図２１及び図２２は、期間P４を示す。レグ３UはPWMレグとなり、レグ４Uは固定電位レグとなる。レグ４Uの上アームトランジスタ４UUが常にオンされる。図２１は下アームトランジスタ３ULがオンされる電流供給期間Tsを示し、図２２は上アームトランジスタ３UUがオンされるフリーホィーリング期間Tfを示す。 FIG. 21 and FIG. 22 show the period P4. Leg 3U is a PWM leg, and leg 4U is a fixed potential leg. Upper arm transistor 4UU of leg 4U is always turned on. FIG. 21 shows a current supply period Ts during which the lower arm transistor 3UL is turned on, and FIG. 22 shows a free wheeling period Tf during which the upper arm transistor 3UU is turned on.
図２３は、６個のレグ３U-４Wの状態を示すタイミングチャートである。６個のレグ３U-４Wは、電気角３６０度に等しい奇数番目の期間においてPWMレグとなり、電気角３６０度に等しい偶数番目の期間において固定電位レグとなる。固定電位レグの上アームトランジスタ及び下アームトランジスタは、電気角１８０度毎に交互にオンされる。言い換えれば、固定電位レグは、電気角１８０度毎にDCリンク電圧VdとDCリンク電圧０Vを交互に出力する。 FIG. 23 is a timing chart showing the state of the six legs 3U-4W. The six legs 3U-4W become PWM legs in odd-numbered periods equal to 360 electrical degrees, and become fixed potential legs in even-numbered periods equal to 360 electrical degrees. The upper arm transistor and the lower arm transistor of the fixed potential leg are alternately turned on every 180 electrical degrees. In other words, the fixed potential leg alternately outputs the DC link voltage Vd and the DC link voltage 0V every 180 electrical degrees.
交互PWM駆動法の１つの利点が図２４及び図２５を参照して説明される。図２４は従来の星形接続３相コイル１Aを駆動する従来の３相インバータ３Aを示す模式配線図である。図２４に示されるモータ装置は、本質的に図２に示されるモータ装置と同じである。３相インバータ３Aは、U相レグ３U、V相レグ３V、及びW相レグ３Wからなる。３相コイル１Aは、U相コイル３４、V相コイル３５、及びW相コイル３６からなる。相コイル３４-３６はそれぞれ、並列接続された２つのコイル２００からなる。各コイル２００はそれぞれ、所定の巻数値Nをもつ。インバータ３Aの６個のアームはそれぞれ、並列接続された２つのトランジスタ３００をもつ。 One advantage of the alternating PWM drive method is described with reference to FIGS. FIG. 24 is a schematic wiring diagram showing a conventional three-phase inverter 3A for driving a conventional star-connected three-phase coil 1A. The motor device shown in FIG. 24 is essentially the same as the motor device shown in FIG. The three-phase inverter 3A includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W. The three-phase coil 1A includes a U-phase coil 34, a V-phase coil 35, and a W-phase coil 36. Each of the phase coils 34-36 includes two coils 200 connected in parallel. Each coil 200 has a predetermined winding value N. Each of the six arms of inverter 3A has two transistors 300 connected in parallel.
図２５は、この実施例のモータ装置を示す模式配線図である。このモータ装置は、図１３に示されるモータ装置と本質的に同じである。相コイル１U-１Wはそれぞれは、直列接続された２つのコイル２００からなる。インバータ３及び４は、図２４に示される従来の３相インバータと比べて２倍の電圧振幅をもつ３相電圧を出力することができる。図２４に示される各コイル２００に供給される相電流は、図２５に示される各コイル２００に供給される相電流と等しくなる。したがって、図２５に示される相コイル１U-１Wはそれぞれ、図２４に示される相コイル３４-３６のそれぞれと比べて２倍の巻数値をもつ。 FIG. 25 is a schematic wiring diagram showing the motor device of this embodiment. This motor device is essentially the same as the motor device shown in FIG. Each of the phase coils 1U-1W is composed of two coils 200 connected in series. Inverters 3 and 4 can output a three-phase voltage having a voltage amplitude twice that of the conventional three-phase inverter shown in FIG. The phase current supplied to each coil 200 shown in FIG. 24 is equal to the phase current supplied to each coil 200 shown in FIG. Therefore, each of phase coils 1U-1W shown in FIG. 25 has twice the number of turns as each of phase coils 34-36 shown in FIG.
図２５に示されるダブルエンデッド３相コイルは、２つの３相インバータ３及び４を必要とする。しかし、３相インバータ３及び４はそれぞれ、図２４に示される３相インバータ３Aの半分の電流容量をもつ。結局、２つのインバータ３及び４は３相インバータ３Aと本質的に等しい回路規模をもつ。 The double-ended three-phase coil shown in FIG. 25 requires two three- phase inverters 3 and 4. However, three- phase inverters 3 and 4 each have half the current capacity of three-phase inverter 3A shown in FIG. As a result, the two inverters 3 and 4 have essentially the same circuit scale as the three-phase inverter 3A.
このモータ装置の重要な利点は、インバータ３および４がインバータ３Aと比べてより低い電力損失をもつことである。なぜなら３相インバータ３及び４のトランジスタの半分は、いわゆるスイッチングロスをもたない固定電位レグとなるからである。さらに、この交互PWM駆動法によれば、トランジスタ３００はPWMレグと固定電位レグとの切替により互いにほぼ等しい温度をもつことができる。 An important advantage of this motor arrangement is that inverters 3 and 4 have lower power losses compared to inverter 3A. This is because half of the transistors of the three- phase inverters 3 and 4 have a fixed potential leg having no switching loss. Further, according to this alternate PWM driving method, the transistor 300 can have substantially the same temperature by switching between the PWM leg and the fixed potential leg.
　　　　　　　　　　第６実施例
第６実施例のモータ装置が図２６-図２９を参照して説明される。図２６は、このモータ装置の配線図である。このモータ装置は、図１３に示される第５実施例のモータ装置と本質的に同じである。しかし、６個のレグ３U-４Wの上アーム側の逆並列接続ダイオードはそれぞれ、帰還線５６を通じてバッテリ５の正極５０及び平滑キャパシタ６の正極６０に接続されている。バッテリ５の正極５０はDCリンク線５４を通じてインバータ３および４の直流端子５７に接続されている。平滑キャパシタ６の正極６０はキャパシタ線５５を通して直流端子５７に接続されている。 Sixth Embodiment A motor device according to a sixth embodiment will be described with reference to FIGS. FIG. 26 is a wiring diagram of this motor device. This motor device is essentially the same as the motor device of the fifth embodiment shown in FIG. However, the antiparallel connected diodes on the upper arm side of the six legs 3U-4W are respectively connected to the positive electrode 50 of the battery 5 and the positive electrode 60 of the smoothing capacitor 6 through the feedback line 56. Positive electrode 50 of battery 5 is connected to DC terminals 57 of inverters 3 and 4 via DC link line 54. The positive electrode 60 of the smoothing capacitor 6 is connected to a DC terminal 57 through a capacitor line 55.
図２７は、U相Hブリッジの詳細な配線図である。線５４ー５６は配線インダクタンス５１ー５３をもつ。レグ３Uは上アームトランジスタ３０１及び下アームトランジスタ３０２をもつ。レグ４Uは上アームトランジスタ３０３及び下アームトランジスタ３０４をもつ。ダイオード３０５のアノード電極が相コイル１Uの一端に接続されている。ダイオード３０６のアノード電極が相コイル１Uの他端に接続されている。ダイオード３０５及び３０６のカソード電極は帰還線５６を通じてバッテリ５及び平滑キャパシタ６の正極５０及び６０に接続されている。言い換えれば、ダイオード３０５は、線５４ー５６を通じてトランジスタ３０１と逆並列接続されている。同様に、ダイオード３０６は線５４ー５６を通じてトランジスタ３０３と逆並列接続されている。 FIG. 27 is a detailed wiring diagram of the U-phase H bridge. Lines 54-56 have wiring inductances 51-53. Leg 3U has an upper arm transistor 301 and a lower arm transistor 302. Leg 4U has an upper arm transistor 303 and a lower arm transistor 304. The anode electrode of the diode 305 is connected to one end of the phase coil 1U. The anode electrode of the diode 306 is connected to the other end of the phase coil 1U. The cathode electrodes of the diodes 305 and 306 are connected to the positive electrodes 50 and 60 of the battery 5 and the smoothing capacitor 6 through the feedback line 56. In other words, diode 305 is anti-parallel connected to transistor 301 through lines 54-56. Similarly, diode 306 is anti-parallel connected to transistor 303 through lines 54-56.
図２７は、レグ４Uの下アームトランジスタ３０４がオフされた直後の状態を示す。レグ３Uの上アームトランジスタ３０１はオン状態を継続している。トランジスタ３０４がオフされる時、トランジスタ３０１に供給されていた電源電流は遮断される。相コイル１Uの強力なインダクタンスにより、フリーホィーリング電流Ifが、ダイオード３０６、線５４-５６、トランジスタ３０１を通じて循環する。したがって、相コイル１Uへの電源電流が遮断される時、配線インダクタンス５１及び５３はサージ電圧を発生しない。 FIG. 27 shows a state immediately after the lower arm transistor 304 of the leg 4U is turned off. Upper arm transistor 301 of leg 3U continues to be on. When the transistor 304 is turned off, the power supply current supplied to the transistor 301 is cut off. Due to the strong inductance of phase coil 1U, freewheeling current If circulates through diode 306, lines 54-56, and transistor 301. Therefore, when the power supply current to the phase coil 1U is cut off, the wiring inductances 51 and 53 do not generate a surge voltage.
言い換えれば、この実施例によれば、Hブリッジの下アームトランジスタがオフされる時、フリーホィーリング電流Ifは、逆並列ダイオード３０５又は３０６を通じてバッテリ５及び平滑キャパシタ６の正極に戻る。その結果、DCリンク線５４及びキャパシタ線５５の電流が維持され、線５４及び５５はサージ電圧を発生しない。その結果、トランジスタ３０１及び３０３は、低オン抵抗値をもつ低電圧タイプのトランジスタを採用することができる。線５４、５５、及び５６は、互いに隣接することができる。 In other words, according to this embodiment, when the lower arm transistor of the H-bridge is turned off, the freewheeling current If returns to the positive electrode of the battery 5 and the smoothing capacitor 6 through the anti-parallel diode 305 or 306. As a result, the current in the DC link line 54 and the capacitor line 55 is maintained, and the lines 54 and 55 do not generate a surge voltage. As a result, a low-voltage transistor having a low on-resistance value can be used as the transistors 301 and 303. Lines 54, 55, and 56 can be adjacent to one another.
この実施例の交互PWM法が図２８を参照して説明される。図２８は、６個のレグ３U-４Wの状態を示すタイミングチャートである。図２８は図２３と本質的に同じである。けれども、この実施例によれば、図１７及び図１８に示される期間P２と、図２１及び図２２に示される期間P４だけが使用される。言い換えれば、図２３に示される期間P１の代わりに期間P２が採用され、図２３に示される期間P３の代わりに期間P４が採用される。これにより、この実施例の交互PWM法によれば、上アームトランジスタはそれぞれ、電気角１８０度毎にオン及びオフを交互に行う。下アームトランジスタはそれぞれ、電気角１８０度毎にPWMスイッチングとオフを交互に行う。 The alternate PWM method of this embodiment will be described with reference to FIG. FIG. 28 is a timing chart showing the state of the six legs 3U-4W. FIG. 28 is essentially the same as FIG. However, according to this embodiment, only the period P2 shown in FIGS. 17 and 18 and the period P4 shown in FIGS. 21 and 22 are used. In other words, the period P2 is used instead of the period P1 shown in FIG. 23, and the period P4 is used instead of the period P3 shown in FIG. Thus, according to the alternate PWM method of this embodiment, each of the upper arm transistors alternately turns on and off every 180 electrical degrees. Each of the lower arm transistors alternately performs PWM switching and off at every electrical angle of 180 degrees.
この実施例の交互PWM法によれば、PWMレグの下アームトランジスタはPWM法でスイッチングされる。好適には、PWMレグの上アームトランジスタは、オフ状態を維持することができる。この態様は、IGBTからなる上アームトランジスタに好適である。 According to the alternate PWM method of this embodiment, the lower arm transistor of the PWM leg is switched by the PWM method. Preferably, the upper arm transistor of the PWM leg can be kept off. This embodiment is suitable for an upper arm transistor made of an IGBT.
第５実施例及び第６実施例のシーケンシャルSVPWM法が図２９を参照して説明される。図２９は、１つのPWMサイクル期間TCを示すタイミングチャートである。期間TUは相コイル１Uの電流供給期間であり、期間TVは相コイル１Vの電流供給期間であり、期間TWは相コイル１Wの電流供給期間である。期間TU-TWの合計がPWMサイクル期間TCよりも短い時、互いに隣接する３つの電流供給期間TU、TV、及びTWはPWMサイクル期間TC内に順番に配置される。このPWM駆動法は、シーケンシャルSVPWM法と呼ばれる。直流電源の抵抗損失は電源電流の振幅低減により低減される。 The sequential SVPWM method according to the fifth and sixth embodiments will be described with reference to FIG. FIG. 29 is a timing chart showing one PWM cycle period TC. A period TU is a current supply period of the phase coil 1U, a period TV is a current supply period of the phase coil 1V, and a period TW is a current supply period of the phase coil 1W. When the total of the periods TU-TW is shorter than the PWM cycle period TC, the three current supply periods TU, TV, and TW adjacent to each other are sequentially arranged in the PWM cycle period TC. This PWM driving method is called a sequential SVPWM method. The resistance loss of the DC power supply is reduced by reducing the amplitude of the power supply current.
さらに、１つの電流供給期間の終了は次の電流供給期間の開始と重なる。これにより、直流電源の電源電流に含まれる高周波電流成分を低減することができる。さらに、DCリンク線５４のサージ電圧が低減される。 Further, the end of one current supply period overlaps with the start of the next current supply period. Thereby, the high-frequency current component included in the power supply current of the DC power supply can be reduced. Further, the surge voltage of the DC link line 54 is reduced.
３つの電流供給期間TU、TV、及びTWの和がPWMサイクル周期TCより長い時、複数の電流供給期間はオーバーラップする。しかし、電流供給期間TU、TV、及びTWは、このオーバーラップ期間が最短となるように各PWMサイクル期間内に配置される。さらに、この実施例によれば、３つの電流供給期間TU、TV、及びTWのうち、最も長い電流供給期間を除く他の２つの電流供給期間が優先的にオーバーラップされる。最も長い電流供給期間は、相電流の振幅が大きいことを意味する。したがって、相対的に短い２つの電流供給期間のオーバーラップは、電源電流の振幅増大を抑制することができる。その結果、バッテリ損失の増加が抑制される。 When the sum of the three current supply periods TU, TV, and TW is longer than the PWM cycle period TC, the plurality of current supply periods overlap. However, the current supply periods TU, TV, and TW are arranged within each PWM cycle period such that the overlap period is shortest. Further, according to this embodiment, of the three current supply periods TU, TV, and TW, the other two current supply periods except the longest current supply period are preferentially overlapped. The longest current supply period means that the amplitude of the phase current is large. Therefore, an overlap between two relatively short current supply periods can suppress an increase in the amplitude of the power supply current. As a result, an increase in battery loss is suppressed.
図２９において、パート（A)は、電流供給期間TVが最も長い時の電流供給期間TU、TV、及びTWの配置を示す。パート（B)は電流供給期間TUが最も長い時の電流供給期間TU、TV、及びTWの配置を示す。パート（C)は電流供給期間TWが最も長い時の電流供給期間TU、TV、及びTWの配置を示す。言い換えれば、３つの電流供給期間TU、TV、及びTWのうち最も長い電流供給期間が、他の２つの電流供給期間より時間軸上において挟まれる。これにより、バッテリ５及び平滑キャパシタ６からなる直流電源の電源電流に含まれる高周波電流成分を低減することができる。このため、この高周波電流成分の抵抗損失を低減することができる。さらに、DCリンク線５４のサージ電圧が低減される。 In FIG. 29, part (A) shows the arrangement of the current supply periods TU, TV, and TW when the current supply period TV is the longest. Part (B) shows the arrangement of the current supply periods TU, TV, and TW when the current supply period TU is the longest. Part (C) shows the arrangement of the current supply periods TU, TV, and TW when the current supply period TW is the longest. In other words, the longest current supply period among the three current supply periods TU, TV, and TW is sandwiched on the time axis from the other two current supply periods. Thereby, a high-frequency current component included in the power supply current of the DC power supply including the battery 5 and the smoothing capacitor 6 can be reduced. Therefore, the resistance loss of the high-frequency current component can be reduced. Further, the surge voltage of the DC link line 54 is reduced.
この実施例によれば、配線インダクタンス５２がインバータ３および４の上アーム側の逆並列ダイオードのいわゆるリカバリ損失を低減する。たとえば、図２７において、フリーホィーリング電流Ifがゼロとなった直後の非常に短いリカバリ期間において、逆並列ダイオード３０６は、フリーホィーリング電流Ifと逆方向へリカバリ電流を流す。しかし、このリカバリ電流が急増する時、配線インダクタンス５２の逆起電力は、リカバリ電流を低減する。適切なインダクタンス値をもつコイル素子を配線インダクタンス５２の代わりに採用することも可能である。たとえば、直列接続されたこのコイル素子と逆流防止ダイオード３０６とのペアはDCターミナル５７と相コイル１Uとを接続する。図２７において、ダイオード３０５及び３０６のリカバリ損失を低減するために、DCリンク線５４に接続される独立のコイル素子を配線インダクタンス５２の代わりに採用することができる。 According to this embodiment, the wiring inductance 52 reduces the so-called recovery loss of the antiparallel diodes on the upper arm side of the inverters 3 and 4. For example, in FIG. 27, in a very short recovery period immediately after the freewheeling current If becomes zero, the anti-parallel diode 306 allows a recovery current to flow in a direction opposite to the freewheeling current If. However, when the recovery current increases rapidly, the back electromotive force of the wiring inductance 52 reduces the recovery current. It is also possible to employ a coil element having an appropriate inductance value instead of the wiring inductance 52. For example, a pair of this coil element and backflow prevention diode 306 connected in series connects DC terminal 57 and phase coil 1U. 27, an independent coil element connected to the DC link line 54 can be used instead of the wiring inductance 52 in order to reduce the recovery loss of the diodes 305 and 306.
この実施例によれば、インバータ３および４の上アームトランジスタは電流電流及びフリーホィーリング電流の両方を流す。しかし、インバータ３および４の下アームトランジスタは電源電流のみを流す。上アームトランジスタは導通損失のみをもつ。下アームトランジスタは、導通損失及びスイッチング損失をもつ。結局、上アームトランジスタは、下アームトランジスタとほぼ等しい損失をもつことができる。 According to this embodiment, the upper arm transistors of inverters 3 and 4 carry both current and freewheeling current. However, the lower arm transistors of inverters 3 and 4 allow only the power supply current to flow. The upper arm transistor has only conduction loss. The lower arm transistor has conduction loss and switching loss. Eventually, the upper arm transistor can have approximately the same loss as the lower arm transistor.
図２６に示されるモータ装置において、電流供給期間TU、TV、及びTWが互いにオーバーラップしないシーケンシャルSVPWM法と、３相電流IU、IV、IWが同時に流れる同時SVPWM法が比較される。相電流IUが１であり、相電流IV及びIWがそれぞれ０.５であると仮定される。電源抵抗r７をもつ直流電源７は、シーケンシャルSVPWM法において電力損失（１.５＊r７）をもち、同時SVPWM法において電力損失（４＊r７）をもつ。結局、シーケンシャルSVPWM法は、同時SVPWM法と比べて直流電源の電力損失を大幅に低減することができる。 In the motor device shown in FIG. 26, the sequential SVPWM method in which the current supply periods TU, TV, and TW do not overlap each other and the simultaneous SVPWM method in which the three-phase currents IU, IV, and IW flow simultaneously are compared. It is assumed that the phase current IU is 1 and the phase currents IV and IW are each 0.5. The DC power supply 7 having the power supply resistance r7 has a power loss (1.5 * r7) in the sequential SVPWM method and has a power loss (4 * r7) in the simultaneous SVPWM method. As a result, the sequential SVPWM method can greatly reduce the power loss of the DC power supply compared to the simultaneous SVPWM method.
　　　　　　　　　　第７実施例
第７実施例のモータ装置が図３０を参照して説明される。図３０は、EVバスやEVトラックのための大型トラクションモータに好適なモータ装置を示す配線図である。このモータ装置のステータコイルは、２つのダブルエンデッド３相コイル１及び２からなる。言い換えれば、ステータコイルは、２つのダブルエンデッド３相コイル１及び２に分割される。ダブルエンデッド３相コイル１は、３相インバータ３及び４に接続される。ダブルエンデッド３相コイル２は、３相インバータ８及び９に接続される。 Seventh Embodiment A motor device according to a seventh embodiment will be described with reference to FIG. FIG. 30 is a wiring diagram showing a motor device suitable for a large traction motor for an EV bus or an EV truck. The stator coil of this motor device is composed of two double-ended three- phase coils 1 and 2. In other words, the stator coil is divided into two double-ended three- phase coils 1 and 2. Double-ended three-phase coil 1 is connected to three- phase inverters 3 and 4. Double-ended three-phase coil 2 is connected to three- phase inverters 8 and 9.
３相コイル１、インバータ３および４は、図１３に示される第５実施例と同じある。３相コイル２は、３つの相コイル２X、２Y、及び２Wからなる。第１例において、相コイル２Xは相コイル１Uと同相となり、相コイル２Yは相コイル１Yと同相となり、相コイル２Zは相コイル１Wと同相となる。第２例において、相コイル２Xは相コイル１Uと逆相となり、相コイル２Yは相コイル１Yと逆相となり、相コイル２Zは相コイル１Wと逆相となる。言い換えれば、第２例のステータコイルはダブルエンデッド６相コイルとなる。６つの相コイル１U-２Zは互いに等しい巻数値をもつ。 The three-phase coil 1 and the inverters 3 and 4 are the same as in the fifth embodiment shown in FIG. The three-phase coil 2 includes three phase coils 2X, 2Y, and 2W. In the first example, the phase coil 2X has the same phase as the phase coil 1U, the phase coil 2Y has the same phase as the phase coil 1Y, and the phase coil 2Z has the same phase as the phase coil 1W. In the second example, phase coil 2X has a phase opposite to phase coil 1U, phase coil 2Y has a phase opposite to phase coil 1Y, and phase coil 2Z has a phase opposite to phase coil 1W. In other words, the stator coil of the second example is a double-ended six-phase coil. The six phase coils 1U-2Z have equal winding values.
インバータ３は３つの相電圧VU１、VV１、及びVW１を出力する。インバータ４は３つの相電圧VU２、VV２、及びVW２を出力する。インバータ８は３つの相電圧VX１、VY１、及びVZ１を出力する。インバータ９は３つの相電圧VX２、VY２、及びVZ２を出力する。図３１は３つの相電圧差（VX-VX２)、（VY１-VY２）、及び（VZ１-VZ２）の平均波形を示すタイミングチャートである。 Inverter 3 outputs three phase voltages VU1, VV1, and VW1. Inverter 4 outputs three phase voltages VU2, VV2, and VW2. The inverter 8 outputs three phase voltages VX1, VY1, and VZ1. Inverter 9 outputs three phase voltages VX2, VY2, and VZ2. FIG. 31 is a timing chart showing an average waveform of three phase voltage differences (VX-VX2), (VY1-VY2), and (VZ1-VZ2).
コントローラ１００は４つのインバータ３、４、８、及び９をシーケンシャルSVPWM法により制御する。図３２は、このシーケンシャルSVPWM法を説明するためのタイミングチャートである。U相コイル１Uは電流供給期間TUをもち、V相コイル１Vは電流供給期間TVをもち、W相コイル１Wは電流供給期間TWをもつ。同様に、X相コイル１Xは電流供給期間TXをもち、Y相コイル１Yは電流供給期間TYをもち、Z相コイル１Zは電流供給期間TZをもつ。 The controller 100 controls the four inverters 3, 4, 8, and 9 according to the sequential SVPWM method. FIG. 32 is a timing chart for explaining the sequential SVPWM method. The U-phase coil 1U has a current supply period TU, the V-phase coil 1V has a current supply period TV, and the W-phase coil 1W has a current supply period TW. Similarly, the X-phase coil 1X has a current supply period TX, the Y-phase coil 1Y has a current supply period TY, and the Z-phase coil 1Z has a current supply period TZ.
図３２において、６個の電流供給期間（TU、TV、TW、TX、TY、及びTZ）は順番に配置される。６つの電流供給期間TU、TV、及びTWの和がPWMサイクル周期TCより長い時、複数の電流供給期間はオーバーラップする。しかし、６個の電流供給期間TU-TZは、オーバーラップ期間の全長が最短となるように各PWMサイクル期間TC内に配置される。これにより、電源電流の振幅が低減されるので、直流電源７の抵抗損失が低減される。 In FIG. 32, six current supply periods (TU, TV, TW, TX, TY, and TZ) are arranged in order. When the sum of the six current supply periods TU, TV, and TW is longer than the PWM cycle period TC, the plurality of current supply periods overlap. However, the six current supply periods TU-TZ are arranged in each PWM cycle period TC such that the total length of the overlap period is the shortest. Thereby, the amplitude of the power supply current is reduced, so that the resistance loss of DC power supply 7 is reduced.
さらに、先行する電流供給期間の終了時点は、次の電流供給期間の開始時点とほぼ等しい。言い換えれば、先行する電流供給期間の終了により電源電流が減少する過渡期間は、次の電流供給期間の開始により電源電流が増加する過渡期間とオーバーラップする。これにより、電源電流の変動が抑制され、サージ電圧が低減される。 Further, the end point of the preceding current supply period is substantially equal to the start point of the next current supply period. In other words, the transition period in which the power supply current decreases due to the end of the preceding current supply period overlaps with the transition period in which the power supply current increases due to the start of the next current supply period. Thereby, the fluctuation of the power supply current is suppressed, and the surge voltage is reduced.
さらに、この実施例によれば、電流供給期間TU-TZのうち、相対的に長い電流供給期間は、相対的に短い電流供給期間により挟まれる。これにより、直流電源の高周波電流損失を低減することができる。図３２において、短い電流供給期間TU、TX、は、長い電流供給期間TV及びTYの前に配置され、短い電流供給期間TW、TZは、電流供給期間TV及びTYの後に配置されている。その結果、電源電流に含まれる高周波電流成分の低減により、直流電源の交流電力損失を低減することができる。さらに、最後の電流供給期間の終了時点において発生するサージ電圧を低減することができる。 Further, according to this embodiment, of the current supply periods TU-TZ, a relatively long current supply period is sandwiched by relatively short current supply periods. Thus, high-frequency current loss of the DC power supply can be reduced. In FIG. 32, short current supply periods TU and TX are arranged before long current supply periods TV and TY, and short current supply periods TW and TZ are arranged after current supply periods TV and TY. As a result, the AC power loss of the DC power supply can be reduced by reducing the high-frequency current component included in the power supply current. Further, a surge voltage generated at the end of the last current supply period can be reduced.
　　　　　　　　　　第８実施例
第８実施例のモータ装置が図３３を参照して説明される。図３３は図３０と本質的に同じである。図３４に示されるように、ダブルエンデッド３相コイル１は電気自動車１０の左輪駆動用モータ１１に収容され、ダブルエンデッド３相コイル２はEVの右輪駆動用モータ１２に収容されている。モータ１１及び１２はそれぞれインホィールモータからなる。電気自動車１０はインホィールモータ１１及び１２とともに前輪１３及び１４を有している。４つのインバータ（３、４、８、及び９）は、第６実施例で説明されたシーケンシャルSVPWM法により制御される。モータ１１及び１２に供給される電源電流は、車両旋回期間を除いてほぼ等しい。 Eighth Embodiment A motor device according to an eighth embodiment will be described with reference to FIG. FIG. 33 is essentially the same as FIG. As shown in FIG. 34, double-ended three-phase coil 1 is accommodated in left-wheel drive motor 11 of electric vehicle 10, and double-ended three-phase coil 2 is accommodated in right-wheel drive motor 12 of the EV. . Each of the motors 11 and 12 is an in-wheel motor. The electric vehicle 10 has front wheels 13 and 14 together with in- wheel motors 11 and 12. The four inverters (3, 4, 8, and 9) are controlled by the sequential SVPWM method described in the sixth embodiment. The power supply current supplied to the motors 11 and 12 is substantially equal except for the vehicle turning period.
　　　　　　　　　　第９実施例
第９実施例のモータ装置が図３５を参照して説明される。図３５は、大型EV用のトラクションモータとして用いられるダブルエンデッド３相コイル式モータ装置を示す配線図である。図３５に示されるU相コイル１U及び２Uは、図１３に示されるU相コイル１Uを分割することにより実現される。同様に、図３５に示されるV相コイル１V及び２Vは、図１３に示されるV相コイル１Vを分割することにより実現される。図３５に示されるW相コイル１W及び２Wは、図１３に示されるW相コイル１Wを分割することにより実現される。６つの相コイル１U-２Wはそれぞれ、等しい巻数値をもつ。 Ninth Embodiment A motor device according to a ninth embodiment will be described with reference to FIG. FIG. 35 is a wiring diagram showing a double-ended three-phase coil type motor device used as a traction motor for a large EV. The U-phase coils 1U and 2U shown in FIG. 35 are realized by dividing the U-phase coil 1U shown in FIG. Similarly, V-phase coils 1V and 2V shown in FIG. 35 are realized by dividing V-phase coil 1V shown in FIG. The W-phase coils 1W and 2W shown in FIG. 35 are realized by dividing the W-phase coil 1W shown in FIG. Each of the six phase coils 1U-2W has an equal winding value.
３相インバータ３はレグ３U、３V、及び３Wからなる。３相インバータ４はレグ４U、４V、及び３Wからなる。直列接続された相コイル１U及び２Uは、レグ３U及び４Uに接続されている。直列接続された相コイル１V及び２Vは、レグ３V及び４Vに接続されている。直列接続された相コイル１W及び２Wは、レグ３W及び４Wに接続されている。レグ３Uは相電圧VU１を出力し、レグ４Uは相電圧VU２を出力する。レグ３Vは相電圧VV１を出力し、レグ４Vは相電圧VV２を出力する。レグ３Wは相電圧VW１を出力し、レグ４Wは相電圧VW２を出力する。 The three-phase inverter 3 includes legs 3U, 3V, and 3W. The three-phase inverter 4 includes legs 4U, 4V, and 3W. Phase coils 1U and 2U connected in series are connected to legs 3U and 4U. Phase coils 1V and 2V connected in series are connected to legs 3V and 4V. Phase coils 1W and 2W connected in series are connected to legs 3W and 4W. Leg 3U outputs a phase voltage VU1, and leg 4U outputs a phase voltage VU2. Leg 3V outputs a phase voltage VV1, and leg 4V outputs a phase voltage VV2. Leg 3W outputs phase voltage VW1, and leg 4W outputs phase voltage VW2.
さらに、U相レグ８U、V相レグ８V、及びW相レグ８Wからなる第３の３相インバータ８が追加される。レグ８Uの交流端子８１は、相コイル１U及び２Uの接続点に接続されている。レグ８Vの交流端子８２は、相コイル１V及び２Vの接続点に接続されている。レグ８Wの交流端子８３は、相コイル１W及び２Wの接続点に接続されている。レグ８Uは相電圧VU３を出力し、レグ８Vは相電圧VV３を出力し、レグ８Wは相電圧VW３を出力する。 Further, a third three-phase inverter 8 including a U-phase leg 8U, a V-phase leg 8V, and a W-phase leg 8W is added. The AC terminal 81 of the leg 8U is connected to a connection point between the phase coils 1U and 2U. The AC terminal 82 of the leg 8V is connected to a connection point of the phase coils 1V and 2V. The AC terminal 83 of the leg 8W is connected to a connection point between the phase coils 1W and 2W. Leg 8U outputs a phase voltage VU3, leg 8V outputs a phase voltage VV3, and leg 8W outputs a phase voltage VW3.
このモータ装置の動作が説明される。コントローラ１００は直列モード及び並列モードをもつ。インバータ８の全てのトランジスタは直列モードにおいてオフされる。したがって、直列モードはインバータ３および４により実行される。この直列モードは第５実施例のモータ装置の動作と同じである。したがって、直列モードにおいて、第５実施例で説明されたシーケンシャルSVPWM法及び交互PWM法を実行することができる。 The operation of the motor device will be described. The controller 100 has a serial mode and a parallel mode. All transistors of the inverter 8 are turned off in the series mode. Therefore, the serial mode is executed by inverters 3 and 4. This series mode is the same as the operation of the motor device of the fifth embodiment. Therefore, in the serial mode, the sequential SVPWM method and the alternating PWM method described in the fifth embodiment can be executed.
図３５に示される直列モードにおいて、U相電流IUが相コイル１U及び２Uを流れ、V相電流IVが相コイル１V及び２Vを流れ、W相電流IWが相コイル１W及び２Wを流れる。６個の相コイル１U-２Wはそれぞれ、自己を流れる相電流と反対方向向きの逆起電力をもつ。 In the series mode shown in FIG. 35, U-phase current IU flows through phase coils 1U and 2U, V-phase current IV flows through phase coils 1V and 2V, and W-phase current IW flows through phase coils 1W and 2W. Each of the six phase coils 1U-2W has a back electromotive force in the opposite direction to the phase current flowing through itself.
並列モードが図３６を参照して説明される。図３６は、３相コイル１を流れる３つの相電流IU、IV、及びIWの方向を除いて図３５と本質的に同じである。この並列モードによれば、３相コイル１の３つの相コイル１U、１V、１Wは、直列モードと比べて反対方向のの逆起電力くをもつ。したがって、相コイル１Uの相電流IU１は、相コイル２Uの相電流IU２と比べて反対方向へ流れる。同様に、相コイル１Vの相電流IV１は、相コイル２Vの相電流IV２と比べて反対方向へ流れる。相コイル１Wの相電流IW１は、相コイル２Wの相電流IW２と比べて反対方向へ流れる。結局、図３５に示される３相モータは、図３６に示される６相モータに変更される。 The parallel mode will be described with reference to FIG. FIG. 36 is essentially the same as FIG. 35 except for the directions of the three phase currents IU, IV, and IW flowing through the three-phase coil 1. According to the parallel mode, the three phase coils 1U, 1V, and 1W of the three-phase coil 1 have counter electromotive forces in the opposite directions as compared with the series mode. Therefore, the phase current IU1 of the phase coil 1U flows in the opposite direction as compared with the phase current IU2 of the phase coil 2U. Similarly, the phase current IV1 of the phase coil 1V flows in a direction opposite to the phase current IV2 of the phase coil 2V. The phase current IW1 of the phase coil 1W flows in a direction opposite to the phase current IW2 of the phase coil 2W. Eventually, the three-phase motor shown in FIG. 35 is changed to a six-phase motor shown in FIG.
相電流IU１, IV１, IW１の方向切替が図３６を参照して説明される。レグ８U及びレグ４UからなるU相HブリッジはU相電流IU２を相コイル２Uに供給する。レグ８V及びレグ４VからなるV相HブリッジはV相電流IV２を相コイル２Vに供給する。レグ８W及びレグ４WからなるW相HブリッジはW相電流IW２を相コイル２Wに供給する。 The direction switching of the phase currents IU1, ΔIV1, and ΔIW1 will be described with reference to FIG. The U-phase H bridge composed of the legs 8U and 4U supplies the U-phase current IU2 to the phase coil 2U. The V-phase H bridge composed of the legs 8V and 4V supplies the V-phase current IV2 to the phase coil 2V. The W-phase H bridge including the legs 8W and 4W supplies the W-phase current IW2 to the phase coil 2W.
同様に、レグ８U及びレグ３UからなるU相HブリッジはU相電流IU１を相コイル１Uに供給する。レグ８V及びレグ３VからなるV相HブリッジはV相電流IV１を相コイル１Vに供給する。レグ８W及びレグ３WからなるW相HブリッジはW相電流IW１を相コイル１Wに供給する。並列モードは、本質的に第５実施例と同じ動作をもつ。したがって、並列モードは、第５実施例で説明されたシーケンシャルSVPWM法及び交互PWM法を実行することができる。 Similarly, a U-phase H bridge composed of leg 8U and leg 3U supplies U-phase current IU1 to phase coil 1U. A V-phase H bridge consisting of leg 8V and leg 3V supplies V-phase current IV1 to phase coil 1V. The W-phase H bridge composed of the leg 8W and the leg 3W supplies the W-phase current IW1 to the phase coil 1W. The parallel mode has essentially the same operation as the fifth embodiment. Therefore, in the parallel mode, the sequential SVPWM method and the alternating PWM method described in the fifth embodiment can be executed.
並列モードで実行される交互PWM法がさらに説明される。この交互PWM法において、６つの上記Hブリッジはそれぞれ、PWMレグ及び固定電位レグからなる。レグ８U、８V、及び８Wは常に固定電位レグとなり、他の６つのレグ３U-４Wは常にPWMレグとなる。 The alternate PWM method implemented in parallel mode is further described. In this alternate PWM method, each of the six H-bridges consists of a PWM leg and a fixed potential leg. Legs 8U, 8V, and 8W are always fixed potential legs, and the other six legs 3U-4W are always PWM legs.
たとえば、実質的に並列接続された相コイル１U及び２Uに印加されるU相電圧の正半波期間において、レグ８Uの上アームトランジスタが常にオンされ、レグ３U及び４UがPWM法で駆動される。U相電圧の負半波期間において、レグ８Uの下アームトランジスタが常にオンされ、レグ３U及び４UがPWM法で駆動される。正半波期間及び負半波期間はそれぞれ電気角１８０度に相当する。 For example, during the positive half-wave period of the U-phase voltage applied to phase coils 1U and 2U substantially connected in parallel, the upper arm transistor of leg 8U is always turned on, and legs 3U and 4U are driven by the PWM method. . During the negative half-wave period of the U-phase voltage, the lower arm transistor of leg 8U is always turned on, and legs 3U and 4U are driven by the PWM method. The positive half-wave period and the negative half-wave period each correspond to an electrical angle of 180 degrees.
並列モードにおいて、インバータ８はインバータ３又は４と比べて２倍の電流を３相コイル１及び２に供給しなければならない。しかし、インバータ８の各レグは常に固定電位レグとして駆動される。その結果、インバータ８のスイッチング損失は大幅に低減される。 In the parallel mode, the inverter 8 must supply twice the current to the three- phase coils 1 and 2 as compared to the inverter 3 or 4. However, each leg of the inverter 8 is always driven as a fixed potential leg. As a result, the switching loss of the inverter 8 is greatly reduced.
並列モードで実行されるシーケンシャルSVPWMモードがさらに説明される。６個のPWMレグ（３U、３V、３W、４U、４V、４W）の各電流供給期間は１つのPWMサイクル期間内に順番に配置されることができる。たとえば、これら６個の電流供給期間は、図３２に示される配置をもつことができる。直列モードにおいて、ステータコイルの巻数は並列モードと比べて２倍となる。このため、直列モードは高電流領域においてバッテリの損失を低減することができる。直列モードを並列モードに切替えることによりステータ極数を切り替えることも可能である。このステータ極数切替方式は、ロータ極数の切替を必要としない誘導モータに好適である。 The sequential SVPWM mode executed in parallel mode is further described. Each current supply period of the six PWM legs (3U, 3V, 3W, 4U, 4V, 4W) can be sequentially arranged within one PWM cycle period. For example, these six current supply periods can have the arrangement shown in FIG. In the serial mode, the number of turns of the stator coil is twice as large as in the parallel mode. Therefore, the series mode can reduce battery loss in a high current region. It is also possible to switch the number of stator poles by switching the serial mode to the parallel mode. This stator pole number switching method is suitable for an induction motor that does not require switching of the number of rotor poles.
　　　　　　　　　　第１０実施例
第１０実施例のモータ装置が図３７を参照して説明される。図３７は、車両のスタータジエネレータとしてダブルエンデッド３相コイル式モータ装置を示す配線図である。図３７は図３５と本質的に同じである。しかし、図３７は、図３５に示される３相インバータ８の代わりに３相ダイオード整流器８を採用する。図３７のモータ装置はモータモードと発電機モードをもつ。モータモードは、第９実施例の直列モードと同じである。発電機モードは、第９実施例の並列モードと本質的に同じである。しかし、発電機モードにおいて、相コイル１U及び２UのU相電圧は並列に整流される。相コイル１V及び２VのV相電圧は並列に整流される。相コイル１W及び２WのW相電圧は並列に整流される。したがって、発電機モードにおけるステータコイルの銅損はモータモードと比べてほぼ２５％となる。図３７において、インバータ３および４の代わりに２つの三相整流器が採用される時、このスタータジエネレータはオルタネータとなる。このオルタネータは、直列モードと本質的に等しい低速発電モードと、並列モードと本質的に等しい高速発電モードをもつ。この高速発電モードは低速発電モードと比べて１/４の銅損をもつ。 Tenth Embodiment A motor device according to a tenth embodiment will be described with reference to FIG. FIG. 37 is a wiring diagram illustrating a double-ended three-phase coil motor device as a starter generator of a vehicle. FIG. 37 is essentially the same as FIG. However, FIG. 37 employs a three-phase diode rectifier 8 instead of the three-phase inverter 8 shown in FIG. The motor device of FIG. 37 has a motor mode and a generator mode. The motor mode is the same as the serial mode of the ninth embodiment. The generator mode is essentially the same as the parallel mode of the ninth embodiment. However, in generator mode, the U-phase voltages of phase coils 1U and 2U are rectified in parallel. The V-phase voltages of the phase coils 1V and 2V are rectified in parallel. The W-phase voltages of the phase coils 1W and 2W are rectified in parallel. Therefore, the copper loss of the stator coil in the generator mode is approximately 25% as compared with the motor mode. In FIG. 37, when two three-phase rectifiers are employed instead of the inverters 3 and 4, the starter generator becomes an alternator. The alternator has a low-speed power generation mode essentially equal to the series mode and a high-speed power generation mode essentially equal to the parallel mode. This high-speed power generation mode has 1/4 copper loss compared to the low-speed power generation mode.
　　　　　　　　　　他の実施例
次に、他の実施例が説明される。EVのバッテリ温度が低い時、シーケンシャルSVPWMモードの電流供給期間が互いに重なるオーバーラップ期間は延長される。これにより、バッテリの電力損失が増加し、バッテリ温度上昇が加速される。PWMサイクル周期内の電流供給期間の和がPWMサイクル周期を超える時、相対的に短い複数の電流供給期間のオーバーラップが優先的に実行される。これにより、バッテリ損失増加が低減される。トルク増加が要求される時、第３高調波電流成分がダブルエンデッド３相コイルに追加される。これにより、トルクを増加することができる。上記各実施例は、EVトラクションモータ装置の代わりに他の可変速モータ装置に適用されることができる。 Another Embodiment Next, another embodiment will be described. When the battery temperature of the EV is low, the overlap period in which the current supply periods in the sequential SVPWM mode overlap each other is extended. As a result, the power loss of the battery increases, and the battery temperature rise is accelerated. When the sum of the current supply periods within the PWM cycle period exceeds the PWM cycle period, overlapping of the relatively short current supply periods is preferentially executed. Thereby, an increase in battery loss is reduced. When a torque increase is required, a third harmonic current component is added to the double-ended three-phase coil. Thereby, the torque can be increased. Each of the above embodiments can be applied to another variable speed motor device instead of the EV traction motor device.

Claims (23)

  1. 　モータのステータコイルに接続される３相インバータと、直流電源から前記ステータコイルへ電源電流を供給するための電流供給期間をパルス幅変調（PWM）法により形成するために前記インバータを制御するコントローラとを備える可変速モータ装置において、
    　前記ステータコイルは、複数の前記３相インバータに接続される複数の３相コイルを有し、
    　前記コントローラは、２よりも多い種類の前記電流供給期間を各PWMサイクル期間内に形成し、
    　前記電流供給期間はそれぞれ、所定の部分負荷条件において互いにオーバーラップしないことを特徴とする可変速モータ装置。     A three-phase inverter connected to a stator coil of a motor, and a controller for controlling the inverter to form a current supply period for supplying power current from the DC power supply to the stator coil by a pulse width modulation (PWM) method. In a variable speed motor device comprising
    The stator coil has a plurality of three-phase coils connected to the plurality of three-phase inverters,
    The controller forms more than two types of the current supply periods within each PWM cycle period;
    The variable-speed motor device according to claim 1, wherein the current supply periods do not overlap with each other under a predetermined partial load condition.
  2. 　前記コントローラは、前記PWMサイクル期間内の前記各電流供給期間の和が前記PWMサイクル期間より長い時、最も長い前記電流供給期間を除く複数種類の前記電流供給期間のオーバーラップを優先的に実行する請求項１記載の可変速モータ装置。 When the sum of the current supply periods in the PWM cycle period is longer than the PWM cycle period, the controller preferentially executes a plurality of types of overlaps of the current supply periods excluding the longest current supply period. The variable speed motor device according to claim 1.
  3. 　前記コントローラは、前記電流供給期間の１つの終了動作と前記電流供給期間の他の１つの開始動作とを同時的に実行する請求項１記載の可変速モータ装置。 The variable speed motor device according to claim 1, wherein the controller simultaneously executes one end operation of the current supply period and another start operation of the current supply period.
  4. 　前記コントローラは、相対的に短い前記電流供給期間を相対的に長い前記電流供給期間の直後に実行する請求項４記載の可変速モータ装置。 The variable speed motor device according to claim 4, wherein the controller executes the relatively short current supply period immediately after the relatively long current supply period.
  5. 　前記コントローラは、最も長い前記電流供給期間を相対的に短い前記電流供給期間の間に配置する請求項４記載の可変速モータ装置。 The variable speed motor device according to claim 4, wherein the controller arranges the longest current supply period between the relatively short current supply periods.
  6. 　前記ステータコイルは、２つの前記３相インバータに接続される２つの星形接続３相コイルからなり、
    　前記コントローラは、前記各PWMサイクル期間内に４種類の前記電流供給期間を順番に形成する請求項１記載の可変速モータ装置。     Said stator coil comprises two star-connected three-phase coils connected to two said three-phase inverters;
    The variable speed motor device according to claim 1, wherein the controller sequentially forms the four types of the current supply periods within each of the PWM cycle periods.
  7. 　前記インバータは、前記３相コイルの３つの相コイルのうちの２つだけに相電流を同時に供給するための６個のサブ電圧ベクトルを形成し、
    　前記サブ電圧ベクトルは、前記電流供給期間に相当する請求項６記載の可変速モータ装置。     The inverter forms six sub-voltage vectors for simultaneously supplying phase current to only two of the three phase coils of the three phase coil;
    The variable speed motor device according to claim 6, wherein the sub-voltage vector corresponds to the current supply period.
  8. 　前記インバータはさらに、前記３つの相コイルに相電流を同時に供給するための６個のメイン電圧ベクトルを形成し、
    　前記メインベクトルは、前記電流供給期間に相当する請求項７記載の可変速モータ装置。     The inverter further forms six main voltage vectors for simultaneously supplying phase currents to the three phase coils;
    The variable speed motor device according to claim 7, wherein the main vector corresponds to the current supply period.
  9. 　前記インバータは、ダブルエンデッド３相コイルに接続される２つの３相インバータを含み、
    　前記コントローラは、前記各PWMサイクル期間内に３種類の前記電流供給期間を順番に形成する請求項１記載の可変速モータ装置。     The inverter includes two three-phase inverters connected to a double-ended three-phase coil;
    The variable speed motor device according to claim 1, wherein the controller sequentially forms the three types of the current supply periods within each of the PWM cycle periods.
  10. 　前記２つの３相インバータは、PWM法により駆動されるPWMレグと、出力電圧が２種類のDCリンク電圧のどちらかに固定される固定電位レグとからなり、
    　前記ダブルエンデッド３相コイルの３つの相コイルはそれぞれ、前記PWMレグと前記固定電位レグとに接続され、
    　前記コントローラは、前記固定電位レグ及び前記PWMレグを電気角１８０度又は電気角３６０度毎に交代する請求項９記載の可変速モータ装置。     The two three-phase inverters include a PWM leg driven by a PWM method and a fixed potential leg whose output voltage is fixed to one of two types of DC link voltages.
    The three phase coils of the double-ended three-phase coil are respectively connected to the PWM leg and the fixed potential leg,
    The variable speed motor device according to claim 9, wherein the controller alternates the fixed potential leg and the PWM leg every 180 electrical degrees or 360 electrical degrees.
  11. 　前記コントローラは、前記固定電位レグの上アームトランジスタを継続的にオンする請求項１０記載の可変速モータ装置。 11. The variable speed motor device according to claim 10, wherein the controller continuously turns on the upper arm transistor of the fixed potential leg.
  12. 　前記コントローラは、前記PWMレグの上アームトランジスタを継続的にオフする請求項１１記載の可変速モータ装置。 12. The variable speed motor device according to claim 11, wherein the controller continuously turns off an upper arm transistor of the PWM leg.
  13. 　前記２つの３相インバータは、上アームトランジスタに接続される逆並列ダイオードを有し、
    　前記逆並列ダイオードは、フリーホィーリング電流を循環させるための帰還線を通じて直流電源の正極に接続される請求項１０記載の可変速モータ装置。     The two three-phase inverters have anti-parallel diodes connected to the upper arm transistors;
    The variable speed motor device according to claim 10, wherein the anti-parallel diode is connected to a positive electrode of a DC power supply through a feedback line for circulating a free wheeling current.
  14. 　前記ダブルエンデッド３相コイルの各相コイルは、互いに直列接続される２つのサブ相コイルからなり、
    　前記２つのサブ相コイルの接続点は、第３のレグに接続され、
    　前記コントローラは、前記２つのサブ相コイルの一方の逆起電力の方向を変更する請求項１０記載の可変速モータ装置。     Each phase coil of the double-ended three-phase coil includes two sub-phase coils connected in series with each other,
    The connection point of the two sub-phase coils is connected to a third leg,
    The variable speed motor device according to claim 10, wherein the controller changes the direction of the back electromotive force of one of the two sub-phase coils.
  15. 　前記インバータは、２つのダブルエンデッド３相コイルに接続される４つの３相インバータからなり、
    　前記コントローラは、前記各PWMサイクル期間内に３種類より多い前記電流供給期間を順番に形成する請求項１記載の可変速モータ装置。     The inverter comprises four three-phase inverters connected to two double-ended three-phase coils,
    2. The variable speed motor device according to claim 1, wherein the controller sequentially forms more than three types of the current supply periods within each PWM cycle period.
  16. 　前記複数の３相コイルは、電気自動車の複数の車輪を別々に駆動するための複数のモータに別々に収容されている請求項１記載の可変速モータ装置。 The variable speed motor device according to claim 1, wherein the plurality of three-phase coils are separately housed in a plurality of motors for separately driving a plurality of wheels of the electric vehicle.
  17. 　モータのステータコイルに接続される３相インバータと、直流電源から前記ステータコイルへ電源電流を供給するための電流供給期間をパルス幅変調（PWM）法により形成するために前記インバータを制御するコントローラとを備える可変速モータ装置において、
    　前記ステータコイルは、星形接続タイプの３相コイルを有し、
    　前記コントローラは、各PWMサイクル期間内に２種類の前記電流供給期間を順番に形成し、
    　前記３相インバータは、前記直流電源から前記３相コイルの２つの相コイルのみに相電流を供給するための６個のサブ電圧ベクトルを形成し、
    　前記サブ電圧ベクトルは、前記電流供給期間に相当することを特徴とする可変速モータ装置。     A three-phase inverter connected to a stator coil of a motor, and a controller for controlling the inverter to form a current supply period for supplying power current from the DC power supply to the stator coil by a pulse width modulation (PWM) method. In a variable speed motor device comprising
    The stator coil has a star connection type three-phase coil,
    The controller sequentially forms the two types of the current supply periods within each PWM cycle period,
    The three-phase inverter forms six sub-voltage vectors for supplying a phase current from the DC power supply to only two phase coils of the three-phase coil,
    The variable speed motor device according to claim 1, wherein the sub-voltage vector corresponds to the current supply period.
  18. 　前記３相インバータはさらに、前記直流電源から前記星形接続タイプの３相コイルのすべての相コイルに相電流を供給するための６個のメイン電圧ベクトルを形成し、
    　前記サブ電圧ベクトル及び前記メイン電圧ベクトルはそれぞれ、前記電流供給期間の１つに相当する請求項１７記載の可変速モータ装置。     The three-phase inverter further forms six main voltage vectors for supplying a phase current from the DC power supply to all phase coils of the star connection type three-phase coil,
    The variable speed motor device according to claim 17, wherein the sub voltage vector and the main voltage vector each correspond to one of the current supply periods.
  19. 　モータのステータコイルに接続される３相インバータと、直流電源から前記ステータコイルへ電源電流を供給するための電流供給期間をパルス幅変調（PWM）法により形成するために前記インバータを制御するコントローラとを備える可変速モータ装置において、
    　前記ステータコイルは、２つの前記３相インバータに接続されるダブルエンデッド３相コイルを有し、
    　前記２つの３相インバータは、PWM変調されるPWMレグと、２種類のDCリンク電圧のどちらかに固定される固定電位レグとからなり、
    　前記ダブルエンデッド３相コイルの３つの相コイルはそれぞれ、前記PWMレグと前記固定電位レグとに接続され、
    　前記コントローラは、前記固定電位レグ及び前記PWMレグを電気角１８０度又は電気角３６０度毎に交代することを特徴とする可変速モータ装置。     A three-phase inverter connected to a stator coil of a motor, and a controller for controlling the inverter to form a current supply period for supplying power current from the DC power supply to the stator coil by a pulse width modulation (PWM) method. In a variable speed motor device comprising
    The stator coil has a double-ended three-phase coil connected to the two three-phase inverters,
    The two three-phase inverters include a PWM leg that is PWM-modulated and a fixed potential leg that is fixed to one of two types of DC link voltages.
    The three phase coils of the double-ended three-phase coil are respectively connected to the PWM leg and the fixed potential leg,
    The variable speed motor device, wherein the controller alternates the fixed potential leg and the PWM leg every 180 electrical degrees or 360 electrical degrees.
  20. 　前記コントローラは、前記固定電位レグの上アームトランジスタを継続的にオンする請求項１９記載の可変速モータ装置。 20. The variable speed motor device according to claim 19, wherein the controller continuously turns on the upper arm transistor of the fixed potential leg.
  21. 　前記コントローラは、前記PWMレグの上アームトランジスタを継続的にオフする請求項２０記載の可変速モータ装置。 21. The variable speed motor device according to claim 20, wherein the controller continuously turns off an upper arm transistor of the PWM leg.
  22. 　前記２つの３相インバータは、上アームトランジスタに接続される逆並列ダイオードを有し、
    　前記逆並列ダイオードは、フリーホィーリング電流を循環させるための帰還線を通じて直流電源の正極に接続される請求項１９記載の可変速モータ装置。     The two three-phase inverters have anti-parallel diodes connected to the upper arm transistors;
    20. The variable speed motor device according to claim 19, wherein the anti-parallel diode is connected to a positive electrode of a DC power supply through a feedback line for circulating a free wheeling current.
  23. 　前記ダブルエンデッド３相コイルの各相コイルは、互いに直列接続される２つのサブ相コイルからなり、
    　前記２つのサブ相コイルの接続点は、第３のレグに接続され、
    　前記コントローラは、前記２つのサブ相コイルの一方の逆起電力の方向を変更する請求項１９記載の可変速モータ装置。     Each phase coil of the double-ended three-phase coil includes two sub-phase coils connected in series with each other,
    The connection point of the two sub-phase coils is connected to a third leg,
    20. The variable speed motor device according to claim 19, wherein the controller changes a direction of a back electromotive force of one of the two sub-phase coils.
PCT/JP2018/023139 2018-06-18 2018-06-18 Variable-speed motor device WO2019244212A1 (en)

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PCT/JP2019/008756 WO2019244418A1 (en) 2018-06-18 2019-03-06 Three-phase motor drive device
JP2020525258A JP7027024B2 (en) 2018-06-18 2019-03-06 3-phase motor drive
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JP2014187747A (en) * 2013-03-22 2014-10-02 Ntn Corp Motor drive
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Publication number Priority date Publication date Assignee Title
WO2022242794A1 (en) * 2021-05-18 2022-11-24 Schaeffler Technologies AG & Co. KG Controller for actuating a redundant actuator comprising two sub-actuators

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