WO2019008907A1 - Linear motor system - Google Patents

Linear motor system Download PDF

Info

Publication number
WO2019008907A1
WO2019008907A1 PCT/JP2018/018667 JP2018018667W WO2019008907A1 WO 2019008907 A1 WO2019008907 A1 WO 2019008907A1 JP 2018018667 W JP2018018667 W JP 2018018667W WO 2019008907 A1 WO2019008907 A1 WO 2019008907A1
Authority
WO
WIPO (PCT)
Prior art keywords
linear motor
voltage
command value
voltage command
mover
Prior art date
Application number
PCT/JP2018/018667
Other languages
French (fr)
Japanese (ja)
Inventor
鈴木　尚礼
渉 初瀬
小山　昌喜
小林　寛
Original Assignee
日立オートモティブシステムズ株式会社
株式会社日立産機システム
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 日立オートモティブシステムズ株式会社, 株式会社日立産機システム filed Critical 日立オートモティブシステムズ株式会社
Publication of WO2019008907A1 publication Critical patent/WO2019008907A1/en

Links

Images

Classifications

    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F04—POSITIVE - DISPLACEMENT MACHINES FOR LIQUIDS; PUMPS FOR LIQUIDS OR ELASTIC FLUIDS
    • F04B—POSITIVE-DISPLACEMENT MACHINES FOR LIQUIDS; PUMPS
    • F04B27/00—Multi-cylinder pumps specially adapted for elastic fluids and characterised by number or arrangement of cylinders
    • F04B27/02—Multi-cylinder pumps specially adapted for elastic fluids and characterised by number or arrangement of cylinders having cylinders arranged oppositely relative to main shaft
    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F04—POSITIVE - DISPLACEMENT MACHINES FOR LIQUIDS; PUMPS FOR LIQUIDS OR ELASTIC FLUIDS
    • F04B—POSITIVE-DISPLACEMENT MACHINES FOR LIQUIDS; PUMPS
    • F04B35/00—Piston pumps specially adapted for elastic fluids and characterised by the driving means to their working members, or by combination with, or adaptation to, specific driving engines or motors, not otherwise provided for
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/06—Linear motors
    • H02P25/064—Linear motors of the synchronous type

Definitions

  • the present invention relates to a linear motor system, and more particularly to a linear motor system including a linear motor having a mover to which an elastic body is connected.
  • Patent Document 1 discloses a configuration in which the phase of the induced voltage from the search coil is detected, the phase difference with the current phase flowing through the linear motor is detected, and the resonance frequency of the piston is made to coincide with the phase difference. There is. Further, Patent Document 1 describes a configuration in which the voltage value of the output voltage is corrected by a value corresponding to the frequency of the output voltage to hold the stroke of the piston constant.
  • Patent document 1 equips the coil
  • a search coil not only makes the wiring complicated but also susceptible to noise, so it is not easy to control the resonance frequency with high accuracy.
  • the voltage value of the output voltage is corrected by a value corresponding to the frequency of the output voltage, and the specific configuration for keeping the stroke of the piston of the linear compressor constant is not described at all, and control to a desired stroke Is not easy. Therefore, the present invention provides a linear motor system capable of efficiently driving a linear motor at a mechanical resonance frequency including a load without requiring a sensor that detects an induced voltage that changes in accordance with a change in load. .
  • a linear motor system concerning the present invention is provided with a linear motor which has a mover to which a winding to which at least AC voltage is applied and an elastic body are connected, and the load of the elastic body and the linear motor
  • a linear motor system in which a resonance frequency of a mover fluctuates comprising: a linear motor drive device for adjusting a phase difference between the AC voltage and a current flowing in the winding so that a drive frequency of the AC voltage becomes a resonance frequency.
  • a linear motor system includes a linear motor having at least a winding to which an alternating voltage is applied and a mover to which an elastic body is connected, and a resonant frequency of the mover due to a load of the elastic body and the linear motor.
  • the linear motor system is characterized by comprising: a linear motor drive device which increases the phase difference between the alternating current voltage and the current flowing through the winding according to the increase of the load.
  • the linear motor system according to the present invention includes a linear motor having at least a winding to which an alternating voltage is applied and a mover to which an elastic body is connected, and the elastic body and the load of the linear motor resonate the mover.
  • a linear motor system having a variable frequency, wherein a fundamental wave amplitude of a current flowing through the winding is detected, and a phase difference between the alternating current voltage and the current flowing through the winding is large according to an increase in the fundamental wave amplitude. And a linear motor drive device.
  • a linear motor system capable of driving a linear motor at a mechanical resonance frequency including a load with high efficiency without requiring a sensor for detecting an induced voltage which changes in response to a change in the load. It becomes possible. Problems, configurations, and effects other than those described above will be apparent from the description of the embodiments below.
  • Example 1 It is a whole schematic block diagram of the linear motor system of Example 1 which concerns on one Example of this invention. It is a perspective view of the example of composition of an armature. It is a schematic diagram which shows the longitudinal cross section of a magnetic pole, and the flow of magnetic flux. It is explanatory drawing of the polarity generate
  • FIG. 7 is an entire schematic configuration diagram of a linear motor system of a second embodiment. It is explanatory drawing which shows the structural example of the load current detector which comprises the control part shown in FIG.
  • FIG. 24 is an explanatory view showing a configuration example of a voltage drop component creator that constitutes the control unit shown in FIG. 23;
  • FIG. 24 is an explanatory drawing showing an example of the configuration of a voltage command value generator that constitutes the control unit shown in FIG. 23.
  • It is a circuit diagram of the air suspension system of Example 3 concerning other examples of the present invention.
  • FIG. 28 is a schematic view of a vehicle equipped with the air suspension system shown in FIG. 27. It is an explanatory view showing an example of composition of a load current detector.
  • the terms “front-rear direction, left-right direction, and up-down direction orthogonal to each other” are used, but the gravity direction does not have to be parallel to the lower direction. It can be parallel to other directions.
  • FIG. 1 is a schematic view of the entire configuration of a linear motor system according to a first embodiment of the present invention.
  • the linear motor system 100 includes a linear motor drive device 101 and a linear motor 104. As described later, the linear motor 104 has an armature 9 and a mover 6 which move relative to each other.
  • the linear motor drive device 101 includes a current detector 107, a control unit 102, and a power conversion circuit 105.
  • the mover 6 moves in the vertical direction, but the armature 9 and the mover 6 (field element) may move relative to each other, and the armature 9 may move in the vertical direction.
  • Control unit 102 outputs an output voltage command value to power conversion circuit 105 or a drive signal (pulse signal) for driving power conversion circuit 105 according to the detection result of current detector 107. Details of the control unit 102 will be described later. Although the details will be described later, the power conversion circuit 105 is an example of a power conversion unit that converts the voltage of the DC voltage source 120 (FIG. 21) and outputs an AC voltage. A direct current source may be used instead of the direct current voltage source 120.
  • FIG. 2 is a perspective view of the linear motor 104 (a perspective view of a configuration example of an armature).
  • the linear motor 104 of the present embodiment has a mover 6 which can move relative to the armature 9 in the direction (longitudinal direction) in which the permanent magnets 2 (2a, 2b) are arranged.
  • the armature 9 has two magnetic poles 7 opposed in the vertical direction via an air gap, and a winding 8 wound around the magnetic poles 7.
  • the mover 6 is disposed in this air gap.
  • the magnetic pole 7 has magnetic pole teeth 70 (also referred to as teeth) as end surfaces facing the mover 6.
  • the armature 9 can apply a force in the front-rear direction (hereinafter referred to as thrust) to the mover 6.
  • thrust can be controlled so that the mover 6 reciprocates in the front-rear direction.
  • the mover 6 has two flat permanent magnets 2 (2a, 2b) magnetized in the vertical direction.
  • the rear permanent magnet 2a and the front permanent magnet 2b are magnetized in opposite directions.
  • the rear permanent magnet 2a has an N pole on the upper side
  • the front permanent magnet 2b has an S pole on the upper side.
  • the permanent magnets 2 a and 2 b are illustrated, but the mover 6 is not illustrated.
  • the control unit 102 outputs a drive signal so that the mover 6 reciprocates in a range in which the permanent magnets 2 a and 2 b face the armature 9.
  • FIG. 3 is a cross-sectional view taken along the line A-A 'of FIG. 2 (A-A' cross-sectional view).
  • the magnetic pole 7 and the yoke 7e are integrally formed of, for example, a magnetic substance such as iron, and constitute a magnetic circuit.
  • Arrow lines in FIG. 3 indicate an example of magnetic flux lines when current flows through the two windings 8. The direction of flow of the magnetic flux can be reversed depending on the direction of the current flowing through the winding 8 and thus is not limited to the illustrated one.
  • the magnetic pole teeth 70 are magnetized by the magnetic flux lines.
  • FIG. 4 is a diagram for explaining the thrust that the mover 6 receives due to the magnetization of the magnetic pole teeth 70.
  • the polarities of the magnetic pole teeth 70 generated by the current flowing through the winding 8 are represented by "N" and "S” attached near the magnetic pole teeth 70 in the figure.
  • the white arrow indicates the direction of the current flowing through the winding 8.
  • the left figure in FIG. 4 shows that the mover 6 is forced forward by magnetizing the upper magnetic pole teeth 70 a to “S” and the lower magnetic pole teeth 70 b to “N” by the current flowing through the winding 8.
  • An example is shown in which the mover 6 has moved forward.
  • the current flowing through the winding 8 magnetizes the upper magnetic pole teeth 70a to "N” and the lower magnetic pole teeth 70b to "S", thereby moving the mover 6 backward in force.
  • An example is shown in which the mover 6 moves backward.
  • magnetic flux can be supplied to the magnetic circuit including the two magnetic poles 7 to magnetize the two opposing magnetic pole teeth 70 (magnetic pole tooth set).
  • an alternating voltage or current such as a sine wave or a rectangular wave (square wave) as the voltage or current, it is possible to give a thrust for reciprocating the mover 6. Thereby, the motion of the mover 6 can be controlled.
  • the thrust applied to the mover 6 can be changed by changing the amplitude of the applied alternating current or alternating voltage.
  • the displacement of the mover 6 can be changed as desired by appropriately changing the thrust applied to the mover 6 using a known method.
  • the mover 6 reciprocates (e.g., a motion generated in the mover 6 by sequentially repeating magnetization of the magnetic pole teeth 70 as shown in the left and right views of FIG. 4), it changes in an alternating waveform.
  • the amount of change in displacement of the mover 6 is called a stroke.
  • the magnetic pole teeth 70 are magnetic members, a magnetic attraction force for attracting the permanent magnet 2 acts.
  • the two magnetic pole teeth 70 are disposed to face each other with a space between them so as to sandwich the mover 6, the total force of the magnetic attraction force acting on the mover 6 can be reduced.
  • FIG. 5 is an explanatory view of an external mechanism connected to the mover 6.
  • an external mechanism constituted by a resonance spring 23 (assist spring) which is a coil spring is connected to one end of the mover 6, and its spring force It is a figure explaining the mechanism by which the needle
  • One end of the resonant spring 23 is connected to the mover 6 via the intermediate portion 24, and the other end is fixed to the base 25.
  • a side portion 26 which extends substantially in parallel with the extending direction of the resonant spring 23 and guides or supports the resonant spring 23 is provided.
  • the mover 6 (field element 6) is configured as a mover (field element) moving type that moves in the vertical direction, but instead of the mover 6, an elastic body is connected to the armature 9 Then, it may be configured as an armature movement type in which the armature 9 is moved in the vertical direction.
  • FIG. 6 is an explanatory view of the relationship between the drive frequency and the stroke, showing the relationship between the drive frequency of the AC voltage on the horizontal axis and the stroke of the mover 6 on the vertical axis.
  • the amplitude of the AC voltage at each drive frequency is the same.
  • the stroke of the mover 6 steeply increases near the resonance frequency, and the stroke decreases as the distance from the resonance frequency decreases.
  • the resonant frequency is given by the square root of the value obtained by dividing the spring constant k of the resonant spring 23 by the mass m of the mover 6, but depending on the system of the linear motor 104, this value is an approximate value.
  • oscillation can be performed with a large stroke (large energy). That is, when controlling the linear motor 104 in which an elastic body such as the resonance spring 23 is added to the mover 6, it is important to detect or estimate the resonance frequency of the mover 6. Even when the stroke of the mover 6 is controlled as desired, it is important to detect or estimate the resonance frequency of the mover 6.
  • FIG. 7 is an explanatory view of the phase relationship between the position of the mover and the velocity of the mover, and the phase relationship between the applied voltage and the motor current.
  • the upper diagram of FIG. 7 shows the time change of the position and velocity of the mover 6
  • the lower diagram of FIG. 7 shows the relationship between the applied voltage waveform and the time change of the current flowing through the linear motor 104.
  • the upper and lower diagrams in FIG. 7 are waveforms at the same timing.
  • FIG. 8 is a diagram showing the AC waveform of FIG. 7 as a vector.
  • the velocity of the mover 6 is a time derivative of displacement, it changes like a cosine wave. Therefore, they can be shown as vectors on two orthogonal axes. From FIGS. 7 and 8, it can be seen that the speed of the mover 6, the applied voltage, and the motor current are substantially in phase.
  • the resonance spring 23 is added to the mover 6 and the mover 6 is reciprocated at the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant, the phase of the position of the mover 6 is the winding 8
  • the resonance spring 23 is added to the mover 6 and the mover 6 is reciprocated at the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant
  • the phase of the position of the mover 6 is the winding 8 It is known that there is a phase difference of 90 degrees with respect to each of the phase of the applied voltage Vm, the motor current Im, and the speed of the mover 6. That is, when any one of these relationships is established, it can be estimated that driving is performed at the resonance frequency.
  • the phase relationship of the applied voltage Vm to the winding 8, the motor current Im, and the speed of the mover 6 is Since the phase difference is not necessarily 90 degrees, it is desirable to include a control unit that changes according to conditions. At this time, in particular, it is desirable to consider the change in motor current Im due to the change in load. If the mass of the mover 6 deviates from the assumption due to manufacturing variations, or if the mass connected to the resonant spring 23 changes due to the load element added to the mover 6, the resonance frequency changes. In addition, in the case where the load element added to the mover 6 has position dependency, the resonance frequency changes during driving.
  • control unit 102 is configured such that a phase difference estimated value dlt ⁇ ⁇ (hereinafter simply referred to as a phase difference), which is an output of the phase difference detector 130 and the phase difference detector 130, is a phase difference command value dlt ⁇ * .
  • a phase difference estimated value dlt ⁇ ⁇ (hereinafter simply referred to as a phase difference)
  • a phase difference command value dlt ⁇ * is a phase difference command value dlt ⁇ * .
  • Drive frequency regulator 131 that adjusts drive frequency command value ⁇ * to follow, integrator 140, load current detector 136, induced voltage component creator 135 that creates induced voltage component, voltage drop that creates voltage drop component component generator 137, the voltage command value generator 103 outputs a voltage command value Vm *, and, by comparing the voltage command value Vm * and the triangular wave carrier signal, a drive signal for driving the power conversion circuit 105 which outputs a voltage It comprises a PWM signal generator 134 for outputting.
  • the control unit 102 receives the motor current Im from the current detector 107.
  • the motor current Im is an AC waveform.
  • the motor current Im input to the control unit 102 is input to the phase difference detector 130 and the load current detector 136 that constitute the control unit 102.
  • the phase difference detector 130 outputs a phase difference dlt ⁇ ⁇ between a reference phase ⁇ * , which is a phase command value described later in detail, and the motor current Im.
  • the phase difference dlt ⁇ ⁇ output from the phase difference detector 130 is input to the drive frequency adjuster 131.
  • the drive frequency adjuster 131 outputs a frequency command value ⁇ * based on the input phase difference dlt ⁇ ⁇ .
  • An applied voltage Vm based on the frequency command value ⁇ * is output to the linear motor 104.
  • the reference phase ⁇ * which is the phase command value of this embodiment, is obtained by integrating the drive frequency command value ⁇ * , which is the output of the drive frequency adjuster 131 in FIG. 1, by the integrator 140 as a reference phase generator. That is, the reference phase theta * applied voltage Vm ( ⁇ *) is the phase theta * at each time of the wave having a driving frequency command value omega * corresponding to the target frequency.
  • reference the phase theta * as is used driving frequency command value of the drive frequency regulator 131 omega * may be fixed to the mechanical resonance frequency of the vibration body including the movable member 6 in the present embodiment.
  • the reference phase ⁇ * is a sawtooth wave having, for example, [- ⁇ , ⁇ ], [0, 2 ⁇ ], or a range wider than this for each time while the drive frequency command value ⁇ * is constant. Alternatively, it may increase linearly with time. As described later, when the drive frequency command value ⁇ * changes, the shape of the sawtooth wave or the linear increase changes accordingly (the slope changes).
  • ⁇ Phase difference detector 130> When the mover 6 reciprocates, the position xm and velocity of the mover 6 and the motor current Im have a periodic function. Since the periodic function can be expressed by a Fourier series, when the position xm of the mover 6 is expressed using a Fourier transform equation, it can be defined as the following equation (1).
  • x 0 is a direct current offset value
  • a n and b n are n-th order Fourier coefficients, which are obtained by the following equations (2) and (3).
  • T 0 is the period of the fundamental wave (the period in which the mover 6 reciprocates), that is, the reciprocal of the primary frequency (drive frequency).
  • a so-called spring-mass system in which a resonant spring 23 is added to the mover 6, a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant is a dominant component. Therefore, one should pay attention to the fundamental wave.
  • the position xm of the mover 6 with respect to the sinusoidal applied voltage Vm can be determined by the following equation (4).
  • the integration range is ⁇ 2 ⁇ to 0. This is because when the phase difference detector 130 is realized by a semiconductor integrated circuit or the like such as a microcomputer or a DSP (Digital Signal Processor), only past information can be acquired.
  • a semiconductor integrated circuit or the like such as a microcomputer or a DSP (Digital Signal Processor)
  • FIG. 9 is an explanatory view showing a configuration example of the phase difference detector 130 constituting the control unit 102 shown in FIG. 1, and an explanatory view in the case where the equation (5) is shown in a block diagram.
  • a reference phase ⁇ * which is a phase command value, is input to each of a sine calculator 81 that outputs the sine of the input value and a cosine calculator 82 that outputs the cosine of the input value, with respect to the reference phase ⁇ * (phase command value).
  • Get sine and cosine A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92.
  • the outputs are integrated by integrators 94a and 94b, respectively, to obtain first-order Fourier coefficients of sine and cosine respectively. That is, since the frequency components higher than the drive frequency ⁇ of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise.
  • the outputs of the integrators 94a and 94b are input to the inverse tangent unit 86.
  • the arctangent device 86 outputs an arctangent value based on the input sine and cosine components.
  • the inverse tangent unit 86 of this embodiment outputs the inverse tangent value of the phase with the numerator as the output of the integrator 94a and the denominator as the output of the integrator 94b, but even if the numerator and the denominator are inverted, good.
  • FIG. 10 is an explanatory diagram of the relationship between the drive frequency and the output from the phase difference detector 130. As shown in FIG. That is, it is a figure which shows the relationship between the drive frequency (horizontal axis) of an alternating voltage, and the output value (phase difference dlt (theta) ⁇ ((theta) pos ⁇ )) (vertical axis
  • the value output from the inverse tangent device 86 is larger than 0 ° when the drive frequency is higher than the resonant frequency, and smaller than 0 ° when the drive frequency is lower than the resonant frequency.
  • the motor current Im input AC signal to the phase difference detector 130 with respect to the reference phase theta * can obtain the phase difference Dltshita ⁇ of primary frequency components of the reference phase theta *, resonance It is possible to estimate the frequency. Then, for example, when the phase difference dlt ⁇ ⁇ is more than 0 °, control is performed so that the phase difference dlt ⁇ ⁇ decreases, and when it is less than 0 °, control may be performed so as to increase. It is preferable to control the phase difference dlt ⁇ ⁇ when the reference phase ⁇ * and the fundamental frequency ⁇ have the same value as the target value dlt ⁇ * .
  • an incomplete integrator can be used instead of the integrators 94a and 94b.
  • the incomplete integrator is a type of low pass filter (low pass filter) and can be configured the same as the first order delay filter.
  • the low pass filter may be configured with another known configuration such as a second order lag filter, not limited to the first order lag filter.
  • a high pass filter (not shown) may be provided in front of the integrators 94a and 94b (or the incomplete integrators) instead of or in addition to the incomplete integrators.
  • the cutoff frequency of the high pass filter can be, for example, 10 Hz or 5 Hz or less.
  • phase difference detector 130 determines the phase ⁇ of motor current Im with respect to AC voltage command value V * using the inverse tangent of the ratio of the first-order Fourier coefficients of the drive frequency components
  • phase difference detector 130 There is great sensitivity only to the primary frequency component of the motor current Im, which is an input AC signal to the input. That is, for example, even when DC offset and high-order noise are superimposed on the motor current Im, the primary frequency component of the motor current Im, which is an AC signal input to the phase difference detector 130 with respect to the reference phase ⁇ * .
  • the phase dlt ⁇ can be determined more accurately.
  • the high pass filter is provided as described above, it can be configured robustly to a frequency smaller than the drive frequency ⁇ .
  • FIG. 11 is an explanatory view showing a configuration example of the drive frequency adjuster 131 which constitutes the control unit 102 shown in FIG.
  • the driving frequency adjuster 131 obtains the difference between the phase difference command value dlt ⁇ * (for example, 0 °) and the phase difference dlt ⁇ ⁇ obtained by the phase difference detector 130 by the subtractor 91, and the multiplier 92b calculates the difference by the proportional gain Kp_adtr.
  • the multiplication result is proportionally controlled and the integral gain Ki_adtr is multiplied by the multiplier 92c, and the result is integrated by the integrator 94c.
  • the addition result is added by the adder 90, and the frequency command is further added to the frequency command
  • the drive frequency command value ⁇ * is output by adding the initial value ( ⁇ 0 ).
  • the frequency command initial value ( ⁇ 0 ) may be obtained from an upper controller (not shown), or may be set to, for example, 0 ° in advance.
  • the drive frequency adjuster 131 of the present embodiment is a configuration of proportional integral control, other control configurations such as proportional control and integral control can also be applied.
  • phase difference detector 130 and the drive frequency adjuster 131 in the case where the linear motor 104 is driven at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant will be described.
  • the actual resonance frequency is lower than the design value. That is, when the initial value of the drive frequency is determined using the mass design value of the mover 6 (when the initial value of the drive frequency command value ⁇ * is determined using the design value), based on the actual resonance frequency Will also drive at a high frequency.
  • the phase difference dlt ⁇ ⁇ obtained by the phase difference detector 130 is a value larger than the phase difference command value dlt ⁇ * .
  • the drive frequency adjuster 131 executes control to decrease the drive frequency command value ⁇ *, and as a result, the drive frequency command value ⁇ * matches the actual resonance frequency. Therefore, the speed energy of the mover 6 can be effectively used, and the linear motor 104 can be driven with high efficiency.
  • FIG. 12 is an explanatory view showing a configuration example of the induced voltage component generator 135 which constitutes the control unit 102 shown in FIG.
  • the induced voltage component generator 135 generates a voltage command value V m1 * corresponding to the induced voltage generated according to the speed of the mover 6.
  • the upper diagram of FIG. 12 is a so-called open command type configuration example that does not constitute a stroke controller.
  • the induced voltage component generator 135 integrates a stroke command value l * obtained from a host controller (not shown) or the like, and a drive frequency command value ⁇ * which is an output of the drive frequency adjuster 131 as a reference phase generator.
  • a reference phase ⁇ * obtained by integration in the unit 140 is input.
  • the reference phase ⁇ * is input to a cosine calculator 82b that outputs the cosine of the input value, and the cosine output, the stroke command value l *, and the reference phase ⁇ * are multiplied by a multiplier 92d to obtain a speed command value vm. Get * .
  • the speed command value vm * is multiplied by the induced voltage constant Ke * in the multiplier 92e to generate a voltage command value (V m1 * ) corresponding to the induced voltage.
  • the velocity command value vm * can also be obtained from the derivative of the command value xm * at the position of the mover 6, and an arithmetic expression as a basis of the configuration of FIG. 12 is shown in the following equation (6).
  • the speed command value vm * can be represented by a sine wave whose amplitude is the stroke command value l * . Since the sine wave becomes a cosine wave when time-differentiated, the differential operation can be omitted if the configuration shown in the upper part of FIG. 12 is adopted, and the mounting on the control unit 102 becomes easy.
  • the lower part of FIG. 12 is a configuration example of a so-called close command type that constitutes a stroke controller.
  • the induced voltage component generator 135a feeds back a stroke detection value or a stroke estimation value by a known position sensor, and controls a stroke command value by the stroke controller 153.
  • FIG. 13 is an explanatory view showing a configuration example of the stroke controller 153 which constitutes the induced voltage component generator 135a shown in the lower part of FIG.
  • the stroke controller 153 obtains the difference between the stroke command value l * and the stroke detection value or the stroke estimation value with a subtractor 91a, and multiplies this by a proportional gain Kp_astr in a multiplier 92d to be proportional
  • the control result of the control is multiplied by the integral gain Ki_astr in the multiplier 92e, and the result of the integration is integrated in the integrator 94d.
  • the result of integration control is added by the adder 90a, and the adjusted stroke command value l ** is output.
  • the stroke controller 153 of the present embodiment is a configuration of proportional integral control, other control configurations such as proportional control and integral control can also be applied.
  • the position estimation value xm ⁇ is obtained, for example, by the following equation (7) using the voltage Vm applied to the linear motor 104 and the current Im flowing through the linear motor 104.
  • Vm * is a voltage command value Vm * applied to the linear motor 104.
  • FIG. 14 is an explanatory view showing a configuration example of the position estimation unit, and an explanatory view in the case where the equation (7) is shown by a block diagram.
  • position estimation unit 308 receives voltage command value Vm * and motor current Im, and multiplies the motor current Im by winding resistance value Rm * of linear motor 104 to obtain voltage command value Vm *. The result is calculated by an integrator. Further, a result obtained by multiplying the motor current Im by the winding inductance value Lm * of the linear motor 104 is obtained, and the result is subtracted from the result calculated by the above integrator, and this reduced result is divided by the induced voltage constant Ke * Outputs the position estimation value xm ⁇ .
  • the position estimation part 308 can apply the position estimation method of the synchronous type motor well-known besides the above.
  • a stroke can be calculated based on the position estimation value xm ⁇ and input to the troke detector 153 shown in FIG. 13 to control to a desired stroke.
  • FIG. 15 is an explanatory view showing a configuration example of the load current detector 136 which constitutes the control unit 102 shown in FIG.
  • the load current detector 136 of this embodiment extracts the amplitude of the load current component using a Fourier transform equation. Similar to the configuration of the phase difference detector 130 shown in FIG. 9, the load current detector 136 has a sine calculator 81 that outputs a sine of the input value and a cosine calculator 82 that outputs a cosine of the input value.
  • a reference phase ⁇ * which is a phase command value is input to obtain sine and cosine with respect to the reference phase ⁇ * .
  • a value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92, respectively.
  • the outputs are integrated by integrators 94e and 94f, respectively, to obtain first-order Fourier coefficients of sine and cosine respectively. That is, since the frequency components higher than the drive frequency ⁇ of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise.
  • the outputs of the integrators 94 e and 94 f are squared and input to the square root calculator 96. That is, the amplitude of the fundamental current is obtained by obtaining the root sum square root of the sine component and the cosine component, which are first-order Fourier coefficients of the sine and the cosine, respectively. Since the amplitude of the fundamental current also increases as the load increases, the configuration of FIG. 15 can detect the load current Im_ld.
  • FIG. 16 is an explanatory view showing another configuration example of the load current detector constituting the control unit 102 shown in FIG.
  • the load current detector 136b inputs the reference phase ⁇ * , which is a phase command value, to the cosine calculator 82 that outputs the cosine of the input value, and obtains a cosine with respect to the reference phase ⁇ * .
  • the cos component (Im_cos) of the fundamental frequency of the motor current Im is extracted.
  • the cos component (Im_cos) of the fundamental frequency of the motor current Im is low-pass filtered (low-pass filter) processed by the first-order lag filter 141 and output as a load current Im_ld.
  • FIG. 16 shows a configuration example of the first-order lag filter
  • the low-pass filter may be configured with another known configuration, such as a second-order lag filter, instead of the first-order lag filter.
  • the configuration of FIG. 16 can detect the load current by filtering the extracted cosine component.
  • the configuration of FIG. 16 is effective in reducing the calculation load.
  • FIG. 17 is an explanatory view showing a configuration example of the voltage drop component creator 137 which constitutes the control unit 102 shown in FIG.
  • the voltage drop component creator 137 inputs the load current Im_ld detected by the load current detector 136 and the reference phase ⁇ *, and a voltage command value (V m2 corresponding to the voltage drop due to the resistance and inductance of the linear motor 104 Output * ).
  • the multiplier 92 multiplies the load current Im_ld by the resistance value Rm * set or estimated in advance and the inductance value Lm * .
  • the product of the load current Im_ld and the resistance value Rm * is multiplied by the output of the cosine calculator 82 that outputs the cosine of the reference phase ⁇ * that is the input value, and the load current Im_ld and the inductance value Lm * are multiplied.
  • the value multiplied by 92 is multiplied by the output of the sine calculator 81a that outputs the negative sine of the reference phase ⁇ * that is the input value. Furthermore, these are added together by the adder 90 and output as a voltage command value (V m2 * ) corresponding to the voltage drop.
  • FIG. 18 is an explanatory view showing a configuration example of the voltage command value creation unit 103 configuring the control unit 102 shown in FIG.
  • voltage command value creator 103 receives a voltage command value (V m1 * ) corresponding to the induced voltage input from induced voltage component creator 135 and a voltage drop component creator 137.
  • the voltage command value (V m2 * ) corresponding to the voltage drop is added by the adder 90 and is output as an AC voltage command value (V m * ). Since both of the two voltage command values (V m1 * and V m2 * ) to be input are AC waveforms, the voltage command value generator 103 is equivalent to outputting a vector sum (vector addition).
  • Voltage command value generator 103 includes a voltage phase calculator 138, and outputs is the output AC voltage command values V m * phase AC voltage command phase theta Vm *.
  • FIG. 19 is a vector diagram showing vector sums in the voltage command value creator at light load and heavy load, and is an explanatory view showing vector addition at the voltage command value creator 103 in a vector diagram.
  • the left diagram of FIG. 19 is a vector diagram at light load, that is, when the load current is small
  • the right diagram of FIG. 19 is a vector diagram at heavy load, that is, when the load current is large.
  • the voltage command value (V m1 * ) corresponding to the induced voltage is the same vector.
  • the counterclockwise direction is positive.
  • the voltage command value (V m2 * ) corresponding to the voltage drop due to the resistance and inductance is vector added to the voltage command value (V m1 * ) corresponding to the induced voltage.
  • the amplitude of the voltage command value Vm * is increased, minute AC voltage command phase (theta Vm *), the phase advances, the AC voltage command value of the single-phase V m * is output. That is, the voltage command value V m * is appropriately controlled in accordance with the load condition even if the stroke of the mover 6 is the same, in other words, even if the speed command value is the same.
  • the voltage Vm * applied to the linear motor 104 can be adjusted by changing one of the stroke command value l * , the phase command value ⁇ * , and the speed command value vm * . Therefore, by adjusting the amplitude and frequency of the applied voltage, it is possible to control the drive frequency to the resonance frequency and to control the stroke.
  • the voltage phase calculator 138 outputs an AC voltage command phase ( ⁇ Vm * ) using, for example, an inverse tangent calculator or the like.
  • the voltage phase There are various definitions of the voltage phase, but in this embodiment, as shown in FIG. 19, the vertical axis rotated 90 degrees counterclockwise from the mover position (horizontal axis in FIG. 19) which changes sinusoidally (FIG. 19). Phase relative to the vertical axis of That is, in the no-load state, the AC voltage command phase ( ⁇ Vm * ) is close to zero, and in the heavy load state, it has a positive value.
  • phase difference command value dlt ⁇ * is set to, for example, 0 ° for simplification, but in the present embodiment, as shown in FIG.
  • the AC voltage command phase ( ⁇ V m * ), which is an output of the above, is input to the drive frequency adjuster 131 as a phase difference command value dlt ⁇ * .
  • the amplitude of voltage command value V m * increases according to a change in load (load current), and the drive frequency is higher than the resonance frequency even if the phase advances by the AC voltage command phase ( ⁇ V m * ). It is possible to drive at high efficiency even in a wide range of load conditions without being highly controlled.
  • the output of the voltage phase calculator 138 is multiplied by -1 by the multiplier 92 in FIG. 18, this is not necessary depending on the definition of the voltage phase.
  • FIG. 20 is an explanatory view showing time change of AC voltage command phase and drive frequency command value, and change of AC voltage command phase ( ⁇ V m * ) and drive frequency command value when load is temporally changed FIG.
  • the AC voltage command phase ( ⁇ V m * ) in FIG. 20 is based on the vertical axis (vertical axis in FIG. 19) rotated 90 degrees counterclockwise from the mover position (horizontal axis in FIG. 19) that changes sinusoidally.
  • the AC voltage command phase and the drive frequency command value are each ⁇ 1 ⁇ ⁇ 2 ⁇ ⁇ 3 according to the load changing L1 ⁇ L2 ⁇ L3 (L2>L1> L3). ( ⁇ 2> ⁇ 1> ⁇ 3) and f1 ⁇ f2 ⁇ f3 (f2>f1> f3).
  • the PWM signal generator 134 constituting the control unit 102 shown in FIG. 1 uses known pulse width modulation by comparing the carrier signal of the triangular wave and the voltage command value Vm *, and drives according to the voltage command value Vm * A signal is generated, and the generated drive signal is output to power conversion circuit 105.
  • FIG. 21 is a diagram showing a configuration example of the power conversion circuit 105 that constitutes the linear motor drive device 101 shown in FIG.
  • the full bridge circuit 126 switches the DC voltage source 120 according to the drive signal input by the control unit 102, and outputs a voltage to the linear motor 104.
  • the full bridge circuit 126 includes four switching elements 122, and includes first and second upper and lower arms (hereinafter referred to as U phase) having switching elements 122a and 122b connected in series, and second upper and lower arms having switching elements 122c and 122d.
  • An arm hereinafter, referred to as a V phase
  • U phase first and second upper and lower arms having switching elements 122a and 122b connected in series
  • V phase An arm
  • Switching element 122 performs switching operation according to pulse-like gate signals (124 a to 124 d) output from gate driver circuit 123 based on voltage command value Vm * generated by control unit 102 and a drive signal by pulse width modulation. it can. By controlling the conduction state (on / off) of switching element 122, a voltage corresponding to an AC voltage of DC voltage source 120 can be output to winding 8.
  • a direct current source may be used instead of the direct current voltage source 120.
  • the switching element 122 for example, a semiconductor switching element such as an IGBT or a MOS-FET can be employed.
  • the linear motor 104 is connected between the switching elements 122 a and 122 b of the first upper and lower arms and between the switching elements 122 c and 122 d of the second upper and lower arms of the power conversion circuit 105.
  • FIG. 21 shows an example in which the windings 8 of the upper and lower armatures 9 are connected in parallel, the windings 8 may be connected in series.
  • the U-phase lower arm and the V-phase lower arm may be provided with a current detector 107 such as a CT (current transformer).
  • a current detector 107 such as a CT (current transformer).
  • CT current transformer
  • the current detector 107 for example, in place of the CT, a shunt resistor 125 is added to the lower arm of the power conversion circuit 105, and a phase shunt current method is employed to detect the current flowing to the linear motor 104 from the current flowing to the shunt resistor 125 it can.
  • a single shunt current detection method for detecting the current on the alternating current side of the power conversion circuit 105 from the direct current flowing in the shunt resistor 125 added to the direct current side of the power conversion circuit 105 may be adopted.
  • the single shunt current detection method utilizes that the current flowing in the shunt resistor 125 temporally changes depending on the conduction state of the switching element 122 that constitutes the power conversion circuit 105.
  • the linear motor can be driven with high efficiency at a mechanical resonance frequency including the load without requiring a sensor that detects an induced voltage that changes according to a change in the load. It becomes possible to provide a linear motor system. Specifically, by controlling the voltage amplitude and voltage phase according to the load, and adjusting the drive frequency in consideration of the voltage phase, the linear motor can be driven at a mechanical resonance frequency including the load, and the efficiency is high.
  • the linear motor system can be configured.
  • the configuration of this embodiment can be the same as that of Embodiment 1 except for the following points.
  • the present embodiment relates to a hermetic compressor 50 as an example of a device equipped with a linear motor system 200 described later.
  • FIG. 22 is a longitudinal sectional view of a hermetic compressor according to a second embodiment of the present invention, and is an example of a longitudinal sectional view of the hermetic compressor 50 having the linear motor 104.
  • the hermetic compressor 50 is a reciprocating compressor in which the compression element 20 and the electric element 30 are disposed in the hermetic container 3.
  • the compression element 20 and the motor element 30 are elastically supported in the closed container 3 by a support spring 49.
  • the motor element 30 includes the mover 6 and the armature 9.
  • the compression element 20 comprises a cylinder block 1 forming a cylinder 1 a, a cylinder head 16 assembled on the end face of the cylinder block 1, and a head cover 17 forming a discharge chamber space.
  • the working fluid supplied into the cylinder 1a is compressed by the reciprocating motion of the piston 4, and the compressed working fluid is sent to a discharge pipe (not shown) communicating with the outside of the compressor.
  • a discharge pipe (not shown) communicating with the outside of the compressor.
  • the working fluid for example, air or a refrigerant of a refrigeration cycle can be adopted.
  • a piston 4 is attached to one end of the mover 6.
  • the working fluid is compressed and expanded by reciprocating the mover 6 and the piston 4.
  • the work and the like required for the compression and expansion correspond to the fluctuating load.
  • a compression element 20 is disposed at one end of the motorized element 30.
  • the cylinder block 1 has a guide rod for guiding the reciprocating motion of the mover 6 along the longitudinal direction.
  • a resonance spring 23 (not shown in FIG. 22) is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, the resonance by the compression element 20 The effect on frequency also needs to be considered.
  • the frequency at which the resonance state is obtained changes. That is, when the pressure of the cylinder 1a is high, it is equivalent to the spring constant of the resonance spring 23 added to the mover 6 being high, and the resonance frequency becomes high. On the other hand, when the pressure in the cylinder 1a is low, the spring constant of the resonant spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant.
  • the resonance frequency changes depending on the conditions of the compression element 20 (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency in accordance with changes in load and resonance frequency.
  • the influence that the resonance frequency changes can be taken from the phase relationship of the applied voltage Vm to the winding 8, the motor current Im, and the speed of the mover 6. Therefore, by controlling based on these phase relationships, a highly efficient linear motor system can be configured.
  • FIG. 23 is an overall schematic configuration diagram of a linear motor system 200 of the present embodiment.
  • the linear motor system 200 has the same configuration as the linear motor system 100 in 1 in the above-described embodiment, but a load current detector 136a and a voltage drop component creator 137a that creates a voltage drop component.
  • the voltage command value generator 103 a that outputs the voltage command value Vm * is different.
  • the load current detector 136a, the voltage drop component generator 137a, and the voltage command value generator 103a will be described below.
  • the other configuration is the same as that of the linear motor system 100 according to the first embodiment described above, and therefore the description thereof is omitted.
  • FIG. 24 is an explanatory view showing a configuration example of the load current detector 136a that constitutes the control unit 202 of the linear motor system 200 shown in FIG.
  • the load current detector 150a receives the motor current Im from the current detector 107 and the reference phase ⁇ * which is a phase command value, and the cos component of the fundamental frequency of the motor current Im (Im_cos And sin components (Im_sin) are respectively extracted and output.
  • a sine calculator 81 which outputs the sine of the input value
  • each of the cosine calculator 82 which outputs the cosine of the input value, receives a reference phase theta * a phase command value, the sine and cosine with respect to the reference phase theta * obtain.
  • a value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92, respectively.
  • the output is low-pass filtered (low-pass filter) with a first-order lag filter 141 to obtain first-order Fourier coefficients of sine and cosine respectively, and load current Im_sin as a sin component (Im_sin) of the fundamental frequency of motor current Im.
  • the load current Im_cos is output as the cos component (Im_cos) of the fundamental frequency of the motor current Im to the voltage drop component creator 137a. That is, since the frequency components higher than the drive frequency ⁇ of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise. In the case of the configuration of FIG. 24, calculation of vector sums described later can be facilitated, which is desirable for mounting on the control unit 202.
  • FIG. 25 is an explanatory view showing a configuration example of the voltage drop component creator 137a that configures the control unit 202 of the linear motor system 200 shown in FIG.
  • the voltage drop component generator 137a receives the load current Im_sin and the load current Im_cos detected by the load current detector 136a, and a voltage command corresponding to the voltage drop due to the resistance and inductance of the linear motor 104.
  • a voltage command value (V m2 * Im_cos) corresponding to the value (V m2 * Im_sin) and the voltage drop is output.
  • the voltage drop component creator 137a multiplies the load current Im_cos by the resistance value Rm * of the linear motor 104, which has been set or estimated in advance, by the multiplier 92, and sets the voltage command value creator 103a as a voltage command value (V m2 * _cos). Output to Further, the voltage drop component creator 137a multiplies the load current Im_sin by the inductance value Lm * of the linear motor 104 preset or estimated by the multiplier 92 to create a voltage command value as a voltage command value (V m2 * _sin). Output to the controller 103a.
  • FIG. 26 is an explanatory view showing a configuration example of a voltage command value creation unit 103a that configures the control unit 202 of the linear motor system 200 shown in FIG.
  • the voltage command value generator 103a generates a voltage command value V m1 * corresponding to the induced voltage generated according to the speed of the mover 6 output from the induced voltage component generator 135, and a voltage drop component
  • the voltage command value (V m2 * _ cos) output from the generator 137 a is added by the adder 90 and output.
  • adding the cos component and the sin component separately is equivalent to outputting the vector sum (vector addition) in the voltage command value generator 103 a.
  • the voltage phase in this embodiment, as shown in FIG. 19 in the above-mentioned first embodiment, in the counterclockwise direction 90 degrees from the mover position (horizontal axis in FIG. 19) which changes sinusoidally.
  • the phase is based on the rotated vertical axis (vertical axis in FIG. 19). Therefore, the output value of induced voltage component generator 135 (voltage command value V m1 * corresponding to the induced voltage generated according to the speed of mover 6) and the voltage command value (V) output of voltage drop component generator 137a
  • the adder 90 adds m2 * _cos).
  • the voltage command value generator 103 a has a voltage phase calculator 138 and a square root calculator 96.
  • the square root calculator 96 obtains the square root of the voltage command value (V m2 * _sin) output from the addition result by the adder 90 and the voltage drop component generator 137 a, and generates an AC voltage command value (V m * ) as a PWM signal. Output to the output unit 134.
  • the voltage phase calculator 138 also receives the addition result of the adder 90 and the voltage command value (V m2 * _sin) output from the voltage drop component creator 137 a, and uses, for example, an AC tangent
  • the voltage command phase ( ⁇ V m * ) is output to the drive frequency adjuster 131.
  • the voltage amplitude and the voltage phase are controlled according to the load, and the drive frequency is adjusted in consideration of the voltage phase.
  • a linear motor can be driven at a mechanical resonance frequency including a load, and a highly efficient linear motor system can be configured.
  • the configuration of this embodiment can be the same as that of Embodiment 1 or 2 except for the following points.
  • the present embodiment relates to an air suspension system 300 as an example of a device mounted with a linear motor system.
  • FIG. 27 is a circuit diagram of an air suspension system 300 according to a third embodiment of the present invention
  • FIG. 28 is a schematic view of a vehicle equipped with the air suspension system 300 shown in FIG. However, in FIG. 28, only the distribution point 309N to be described later and the components on the air suspension 301, 302 side therefrom are illustrated.
  • the air suspension system 300 includes two air suspensions 301 and 302, a compressor 303 driven by the linear motor 104, an intake filter 304, a first tank 305, an air dryer 307, and valves.
  • the three check valves 308, 315, 317, the supply / discharge switching valve 310, the two suspension control valves 311, 312, the return passage on / off valve 314, and the exhaust passage on / off valve 319 are provided.
  • the air suspension system 300 connects these by a passage through which air can flow.
  • the air suspension system 300 is, for example, a system which is mounted on a vehicle 400 and controls air pressure in air chambers 301C and 302C (FIG. 27) of the air suspensions 301 and 302, as shown in FIG.
  • air suspensions 301 and 302 are provided between the wheel 410 side and the vehicle body 430 side, such as between the left wheel 410L and the right wheel 410R and the vehicle body 430, or between the hub and the vehicle body 430,
  • the vehicle height can be adjusted by controlling the air pressure in the 302C.
  • the air suspensions 301 and 302 may be attached between the axle 420 on the wheel 410 side and the vehicle body 430 of the vehicle 400 as shown in FIG. 28 and an arm of a suspension connecting the wheel 410 and the vehicle body 430 It may be attached between the class (wheel 410 side) and the vehicle body 430 or between the hub (wheel 410 side) of the wheel 410 and the vicinity of the vehicle body 430 mounting portion of the upper arm of the suspension (vehicle body 430 side).
  • the air suspensions 301 and 302 may be provided to support the wheels 410 and the vehicle body 430.
  • the air suspensions 301 and 302 can be provided between the wheels 410 and the vehicle body 430 in the vertical direction. It is not restricted to the aspect attached to 430.
  • an air suspension system 300 having two air suspensions will be described, but the number of air suspensions included in the air suspension system 300 is not particularly limited as long as it is one or more.
  • the number of air suspensions can, for example, be equal to the number of wheels.
  • a total of four air suspensions, two on the two front wheels and two on the two rear wheels, can be disposed.
  • cylinders 301A and 302A for shock absorbing and the air chambers 301C and 302C serving as air springs are integrated, but the cylinder for shock absorbing as well known on the large vehicle and rear suspension side (Hydraulic shock absorber) 301A, 302A and an air spring may be provided independently.
  • air chambers 301C and 302C are formed between the buffer cylinders 301A and 302A and the piston rods 301B and 302B, respectively, to form an air spring. ing.
  • a passage described later is connected to each of the air chambers 301C and 302C, and the pressure and the vehicle height are controlled by the operation of the air suspension system 300.
  • the compressor 303 can compress the air sucked from the suction port 303C and discharge the compressed air from the discharge port 303D.
  • the compressor 303 includes a compressor body 303A and a linear motor 104.
  • a pressure sensor is provided to measure the pressure of the suction port 303C or the discharge port 303D or both ports.
  • the resonance spring 23 is added to the mover 6 of the linear motor 104 and the mover 6 is reciprocated at the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant, it is shown in FIG.
  • the influence of the compression element 20 on the resonant frequency also needs to be considered. That is, the spring-like action of the working fluid is exerted by the suction pressure of the compression element 20 and the pressure of the discharge space, so that the frequency at which the resonance state occurs is changed. That is, when the pressure of the cylinder 1a is high, it is equivalent to the spring constant of the resonance spring 23 added to the mover 6 being high, and the resonance frequency becomes high.
  • the spring constant of the resonant spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant.
  • the resonance frequency changes depending on the conditions of the compression element (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency in accordance with changes in load and resonance frequency.
  • FIG. 29 is a diagram showing a configuration example of the load current detector 136b.
  • the load current detector 150 b receives the motor current Im, extracts and outputs a cos component (Im_cos) and a sin component (Im_sin) of the fundamental frequency of the motor current Im.
  • a reference phase ⁇ * which is a phase command value, is input to each of a sine calculator 81 that outputs a sine of an input value and a cosine calculator 82 that outputs a cosine at the input value, and sine and cosine with respect to the reference phase ⁇ * Get A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92, respectively.
  • the output is low-pass filtered (low-pass filter) with a first-order lag filter 141a to obtain sine and cosine first-order Fourier coefficients. That is, since the frequency components higher than the drive frequency ⁇ of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise.
  • the filter time constant (T_ld) or cutoff frequency of the first-order lag filter 141a can be changed from the outside.
  • the filter time constant (T_ld) or cutoff frequency of the first-order lag filter 141a can be changed according to the pressure of the suction port 303C or discharge port 303D of the compressor 303. ing. As a result, the change in pressure can also reduce the influence on load current detection.
  • the machine including the load is controlled by controlling the voltage amplitude and the voltage phase according to the load and adjusting the drive frequency in consideration of the voltage phase.
  • the linear motor can be driven at a typical resonance frequency, and a highly efficient linear motor system can be configured.
  • each of the configurations, functions, processing units, processing procedures, and the like described above may be realized by hardware, for example, by designing part or all of them with an integrated circuit.
  • each configuration, function, and the like described above may be realized by software by a processor interpreting and executing a program that realizes each function.
  • the power conversion circuit 105 may output a current.
  • a current command value creator may be provided instead of the voltage command value creator 103.
  • head cover 20: compression element, 23: resonance spring (assist spring), 30: electric element, 50: sealed compressor, 100, 200: linear motor system, 101, 201: linear motor drive, 102, 202: control unit , 103: voltage command value generator, 104: linear motor, 105: power conversion circuit, 107: current detector, 126: full bridge circuit, 130: phase difference detector, 131: driving frequency adjuster, 134: PWM signal Creator, 135: induced voltage component creator, 136: load current detector, 137: voltage drop component creator, 140: integrator

Landscapes

  • Engineering & Computer Science (AREA)
  • Mechanical Engineering (AREA)
  • General Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Linear Motors (AREA)
  • Compressors, Vaccum Pumps And Other Relevant Systems (AREA)

Abstract

Provided is a linear motor system that can highly efficiently drive a linear motor at a mechanical resonant frequency that incorporates a load, without the need for a sensor that detects induced voltage that changes in accordance with fluctuations in the load. A linear motor system 100 that comprises a linear motor 104 that has a winding 8 to which at least alternating current voltage is applied and a mobile element 6 to which an elastic body 23 is connected. The resonant frequency of the mobile element 6 is dependent on the elastic body 23 and the load of the linear motor 104. The linear motor system 100 also has a linear motor drive device 102 that detects a fundamental wave amplitude for the current flowing in the winding 8 and increases the phase difference between the alternating current voltage Vm and the current Im flowing in the winding 8 in accordance with increases in the fundamental wave amplitude.

Description

リニアモータシステムLinear motor system
　本発明は、リニアモータシステムに係り、特に、弾性体を接続させた可動子を有するリニアモータを備えるリニアモータシステムに関する。 The present invention relates to a linear motor system, and more particularly to a linear motor system including a linear motor having a mover to which an elastic body is connected.
　弾性体を接続させた可動子を、弾性体及び可動子の系における機械的な共振周波数で駆動させるリニアモータが知られている。機械的な共振周波数は、可動子の摩擦や、可動子に接続する負荷に応じて変動するため、共振周波数を効果的に推定することが望まれる。 There is known a linear motor which drives a mover to which an elastic body is connected at a mechanical resonant frequency in a system of the elastic body and the mover. Since the mechanical resonance frequency fluctuates according to the friction of the mover and the load connected to the mover, it is desirable to effectively estimate the resonance frequency.
　例えば、特許文献１は、サーチコイルからの誘起電圧の位相を検出し、リニアモータに流れる電流位相との位相差を検出し、位相差に応じてピストンの共振周波数に一致させる構成が開示されている。また、特許文献１には、出力電圧の周波数に応じた値だけ出力電圧の電圧値を補正してピストンのストロークを一定に保持する構成が記載されている。 For example, Patent Document 1 discloses a configuration in which the phase of the induced voltage from the search coil is detected, the phase difference with the current phase flowing through the linear motor is detected, and the resonance frequency of the piston is made to coincide with the phase difference. There is. Further, Patent Document 1 describes a configuration in which the voltage value of the output voltage is corrected by a value corresponding to the frequency of the output voltage to hold the stroke of the piston constant.
特開平１１－３５１１４３号公報Japanese Patent Application Laid-Open No. 11-351143
　特許文献１は、リニアコンプレッサーの巻線にサーチコイルを具備し、誘起電圧の位相を検出している。しかしながら、サーチコイルを用いると配線が煩雑になるだけでなく、ノイズの影響を受けやすくなるため、高精度に共振周波数に制御することは容易ではない。また、出力電圧の周波数に応じた値だけ出力電圧の電圧値を補正し、リニアコンプレッサーのピストンのストロークを一定に保持する具体的構成については何ら記載されておらず、所望のストロークに制御することは容易ではない。　
　そこで、本発明は、負荷の変動に応じて変化する誘起電圧を検出するセンサを要することなく、負荷を含めた機械的な共振周波数でリニアモータを高効率に駆動し得るリニアモータシステムを提供する。 Patent document 1 equips the coil | winding of a linear compressor with a search coil, and detects the phase of induced voltage. However, using a search coil not only makes the wiring complicated but also susceptible to noise, so it is not easy to control the resonance frequency with high accuracy. Moreover, the voltage value of the output voltage is corrected by a value corresponding to the frequency of the output voltage, and the specific configuration for keeping the stroke of the piston of the linear compressor constant is not described at all, and control to a desired stroke Is not easy.
Therefore, the present invention provides a linear motor system capable of efficiently driving a linear motor at a mechanical resonance frequency including a load without requiring a sensor that detects an induced voltage that changes in accordance with a change in load. .
　上記課題を解決するため、本発明に係るリニアモータシステムは、少なくとも交流電圧が印加される巻線及び弾性体が接続する可動子を有するリニアモータを備え、前記弾性体と前記リニアモータの負荷によって可動子の共振周波数が変動するリニアモータシステムであって、前記交流電圧の駆動周波数が共振周波数となるよう、前記交流電圧と前記巻線に流れる電流の位相差を調整するリニアモータ駆動装置を有することを特徴とする。　
　また、本発明に係るリニアモータシステムは、少なくとも交流電圧が印加される巻線及び弾性体が接続する可動子を有するリニアモータを備え、前記弾性体と前記リニアモータの負荷によって可動子の共振周波数が変動するリニアモータシステムであって、前記負荷の増加に応じて、前記交流電圧と前記巻線に流れる電流の位相差を大きくするリニアモータ駆動装置を有することを特徴とする。　
　更にまた、本発明に係るリニアモータシステムは、少なくとも交流電圧が印加される巻線及び弾性体が接続する可動子を有するリニアモータを備え、前記弾性体と前記リニアモータの負荷によって可動子の共振周波数が変動するリニアモータシステムであって、前記巻線に流れる電流の基本波振幅を検出し、前記基本波振幅の増加に応じて、前記交流電圧と前記巻線に流れる電流の位相差を大きくするリニアモータ駆動装置を有することを特徴とする。 In order to solve the above-mentioned subject, a linear motor system concerning the present invention is provided with a linear motor which has a mover to which a winding to which at least AC voltage is applied and an elastic body are connected, and the load of the elastic body and the linear motor A linear motor system in which a resonance frequency of a mover fluctuates, comprising: a linear motor drive device for adjusting a phase difference between the AC voltage and a current flowing in the winding so that a drive frequency of the AC voltage becomes a resonance frequency. It is characterized by
Further, a linear motor system according to the present invention includes a linear motor having at least a winding to which an alternating voltage is applied and a mover to which an elastic body is connected, and a resonant frequency of the mover due to a load of the elastic body and the linear motor. The linear motor system is characterized by comprising: a linear motor drive device which increases the phase difference between the alternating current voltage and the current flowing through the winding according to the increase of the load.
Furthermore, the linear motor system according to the present invention includes a linear motor having at least a winding to which an alternating voltage is applied and a mover to which an elastic body is connected, and the elastic body and the load of the linear motor resonate the mover. A linear motor system having a variable frequency, wherein a fundamental wave amplitude of a current flowing through the winding is detected, and a phase difference between the alternating current voltage and the current flowing through the winding is large according to an increase in the fundamental wave amplitude. And a linear motor drive device.
　本発明によれば、負荷の変動に応じて変化する誘起電圧を検出するセンサを要することなく、負荷を含めた機械的な共振周波数でリニアモータを高効率に駆動し得るリニアモータシステムを提供することが可能となる。　
　上記した以外の課題、構成及び効果は、以下の実施形態の説明により明らかにされる。 According to the present invention, there is provided a linear motor system capable of driving a linear motor at a mechanical resonance frequency including a load with high efficiency without requiring a sensor for detecting an induced voltage which changes in response to a change in the load. It becomes possible.
Problems, configurations, and effects other than those described above will be apparent from the description of the embodiments below.
本発明の一実施例に係る実施例１のリニアモータシステムの全体概略構成図である。BRIEF DESCRIPTION OF THE DRAWINGS It is a whole schematic block diagram of the linear motor system of Example 1 which concerns on one Example of this invention. 電機子の構成例の斜視図である。It is a perspective view of the example of composition of an armature. 磁極の縦断面と磁束の流れを示す模式図である。It is a schematic diagram which shows the longitudinal cross section of a magnetic pole, and the flow of magnetic flux. 磁極歯に発生する極性の説明図である。It is explanatory drawing of the polarity generate | occur | produced in a magnetic pole tooth. 可動子に接続される外部機構の説明図である。It is explanatory drawing of the external mechanism connected to a needle | mover. 駆動周波数とストロークの関係の説明図である。It is explanatory drawing of the relationship between a drive frequency and a stroke. 可動子の位置と可動子の速度との位相関係、及び印加電圧とモータ電流の位相関係の説明図である。It is explanatory drawing of the phase relationship of the position of a needle | mover, and the velocity of a needle | mover, and the phase relationship of an applied voltage and a motor current. 印加電圧と電流のベクトル図である。It is a vector diagram of applied voltage and current. 図１に示す制御部を構成する位相差検出器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the phase difference detector which comprises the control part shown in FIG. 駆動周波数と位相差検出器出力の関係の説明図である。It is explanatory drawing of the relationship between a drive frequency and a phase difference detector output. 図１に示す制御部を構成する駆動周波数調整器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the drive frequency regulator which comprises the control part shown in FIG. 図１に示す制御部を構成する誘起電圧成分作成器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the induced voltage component creation device which comprises the control part shown in FIG. 図１２に示す誘起電圧成分作成器を構成するストローク制御器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the stroke controller which comprises the induced voltage component creation device shown in FIG. 位置推定部の構成例を示す説明図である。It is explanatory drawing which shows the structural example of a position estimation part. 図１に示す制御部を構成する負荷電流検出器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the load current detector which comprises the control part shown in FIG. 図１に示す制御部を構成する負荷電流検出器の他の構成例を示す説明図である。It is explanatory drawing which shows the other structural example of the load current detector which comprises the control part shown in FIG. 図１に示す制御部を構成する電圧降下成分作成器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the voltage drop component creation device which comprises the control part shown in FIG. 図１に示す制御部を構成する電圧指令値作成器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the voltage command value creation part which comprises the control part shown in FIG. 軽負荷時及び重負荷時における電圧指令値作成器でのベクトル和を示すベクトル図である。It is a vector diagram which shows the vector sum in the voltage command value creator at the time of light load and heavy load. 交流電圧指令位相と駆動周波数指令値の時間変化を示す説明図である。It is explanatory drawing which shows the time change of alternating voltage command phase and drive frequency command value. 図１に示すリニアモータ駆動装置を構成する電力変換回路の構成例を示す図である。It is a figure which shows the structural example of the power inverter circuit which comprises the linear motor drive shown in FIG. 本発明の他の実施例に係る実施例２の密閉型圧縮機の縦断面図である。It is a longitudinal cross-sectional view of the closed type compressor of Example 2 which concerns on the other Example of this invention. 実施例２のリニアモータシステムの全体概略構成図である。FIG. 7 is an entire schematic configuration diagram of a linear motor system of a second embodiment. 図２３に示す制御部を構成する負荷電流検出器の構成例を示す説明図である。It is explanatory drawing which shows the structural example of the load current detector which comprises the control part shown in FIG. 図２３に示す制御部を構成する電圧降下成分作成器の構成例を示す説明図である。FIG. 24 is an explanatory view showing a configuration example of a voltage drop component creator that constitutes the control unit shown in FIG. 23; 図２３に示す制御部を構成する電圧指令値作成器の構成例を示す説明図である。FIG. 24 is an explanatory drawing showing an example of the configuration of a voltage command value generator that constitutes the control unit shown in FIG. 23. 本発明の他の実施例に係る実施例３のエアサスペンションシステムの回路図である。It is a circuit diagram of the air suspension system of Example 3 concerning other examples of the present invention. 図２７に示すエアサスペンションシステムを搭載した車両の概略図である。FIG. 28 is a schematic view of a vehicle equipped with the air suspension system shown in FIG. 27. 負荷電流検出器の構成例を示す説明図である。It is an explanatory view showing an example of composition of a load current detector.
　以下、添付の図面を参照しつつ本発明の実施例を詳細に説明する。同様の構成要素には同様の符号を付し、重複する説明を省略する。　
　本発明の各種の構成要素は、必ずしも個々に独立した存在である必要はなく、複数の構成要素が一個の部材として形成されていること、一つの構成要素が複数の部材で形成されていること、或る構成要素が他の構成要素の一部であること、或る構成要素の一部と他の構成要素の一部とが重複していること、等を許容する。 Hereinafter, embodiments of the present invention will be described in detail with reference to the attached drawings. The same components are denoted by the same reference numerals and redundant description will be omitted.
The various components of the present invention do not necessarily have to be independent entities, but a plurality of components are formed as one member, and one component is formed of a plurality of members. , Allow one component to be part of another component, overlap between part of one component and part of another component, etc.
　本実施例では、説明の便宜上、互いに直交する前後方向、左右方向、及び上下方向という語を用いるが、重力方向は必ずしも下方向に平行である必要はなく、前後方向、左右方向、上下方向又はそれ以外の方向に平行にすることができる。 In this embodiment, for convenience of explanation, the terms “front-rear direction, left-right direction, and up-down direction orthogonal to each other” are used, but the gravity direction does not have to be parallel to the lower direction. It can be parallel to other directions.
　＜リニアモータシステム１００＞　
　図１は、本発明の一実施例に係る実施例１のリニアモータシステムの全体概略構成図である。リニアモータシステム１００は、リニアモータ駆動装置１０１及びリニアモータ１０４から構成される。後述するようにリニアモータ１０４は、相対移動する電機子９及び可動子６を有する。　
　リニアモータ駆動装置１０１は、電流検出器１０７、制御部１０２、及び電力変換回路１０５を有する。　
　本実施例では、可動子６が鉛直方向に移動するが、電機子９及び可動子６（界磁子）が相対移動すれば良く、電機子９が鉛直方向に移動する態様でも良い。なお、以下では、可動子６が鉛直方向に往復運動する場合を一例として説明するが、往復運動の方向は鉛直方向に限られるものではない。例えば、可動子６が水平方向に往復運動するよう構成しても良く、また、鉛直方向に対し任意の角度を有する方向に可動子６が往復運動する構成としても良い。また、これらは、電機子９につても同様である。　
　制御部１０２は、電流検出器１０７の検出結果に応じて、電力変換回路１０５への出力電圧指令値、又は電力変換回路１０５を駆動するドライブ信号（パルス信号）を出力する。制御部１０２の詳細は後述する。　
　詳細は後述するが、電力変換回路１０５は、直流電圧源１２０（図２１）の電圧を変換して交流電圧を出力する電力変換部の一例である。なお、直流電圧源１２０に代えて直流電流源を用いても良い。 <Linear motor system 100>
FIG. 1 is a schematic view of the entire configuration of a linear motor system according to a first embodiment of the present invention. The linear motor system 100 includes a linear motor drive device 101 and a linear motor 104. As described later, the linear motor 104 has an armature 9 and a mover 6 which move relative to each other.
The linear motor drive device 101 includes a current detector 107, a control unit 102, and a power conversion circuit 105.
In the present embodiment, the mover 6 moves in the vertical direction, but the armature 9 and the mover 6 (field element) may move relative to each other, and the armature 9 may move in the vertical direction. Although the case where the mover 6 reciprocates in the vertical direction will be described as an example below, the direction of the reciprocation is not limited to the vertical direction. For example, the mover 6 may be configured to reciprocate in the horizontal direction, or may be configured to reciprocate in a direction having an arbitrary angle with respect to the vertical direction. Also, these are the same for the armature 9.
Control unit 102 outputs an output voltage command value to power conversion circuit 105 or a drive signal (pulse signal) for driving power conversion circuit 105 according to the detection result of current detector 107. Details of the control unit 102 will be described later.
Although the details will be described later, the power conversion circuit 105 is an example of a power conversion unit that converts the voltage of the DC voltage source 120 (FIG. 21) and outputs an AC voltage. A direct current source may be used instead of the direct current voltage source 120.
　＜リニアモータ１０４＞　
　図２はリニアモータ１０４の斜視図（電機子の構成例の斜視図）である。本実施例のリニアモータ１０４は、電機子９に対して、永久磁石２（２ａ，２ｂ）が並んだ方向（前後方向）に相対移動可能な可動子６を有する。電機子９は空隙を介して上下方向に対向する２つの磁極７と、磁極７に捲回された巻線８とを有している。可動子６は、この空隙に配置されている。磁極７は、可動子６に対向する端面としての磁極歯７０（ティースとも称される）を有している。　
　電機子９は、可動子６に対して前後方向の力（以下、推力と称する）を付与できる。例えば、後述するように、可動子６が前後方向に往復運動するように推力を制御できる。　　可動子６は、上下方向に磁化した２つの平板状の永久磁石２（２ａ，２ｂ）を有している。後側の永久磁石２ａ及び前側の永久磁石２ｂは、互いに反対方向に磁化されている。
本実施例では、後側の永久磁石２ａは上側にＮ極を有し、前側の永久磁石２ｂは上側にＳ極を有している。図２では、永久磁石２ａ，２ｂは図示しているが、可動子６は図示していない。可動子６としては、例えば、平板状の永久磁石２を固定した平板状のものを採用できる。　
　制御部１０２は、可動子６を永久磁石２ａ，２ｂが電機子９に対向する範囲で往復運動させるようにドライブ信号を出力する。 <Linear motor 104>
FIG. 2 is a perspective view of the linear motor 104 (a perspective view of a configuration example of an armature). The linear motor 104 of the present embodiment has a mover 6 which can move relative to the armature 9 in the direction (longitudinal direction) in which the permanent magnets 2 (2a, 2b) are arranged. The armature 9 has two magnetic poles 7 opposed in the vertical direction via an air gap, and a winding 8 wound around the magnetic poles 7. The mover 6 is disposed in this air gap. The magnetic pole 7 has magnetic pole teeth 70 (also referred to as teeth) as end surfaces facing the mover 6.
The armature 9 can apply a force in the front-rear direction (hereinafter referred to as thrust) to the mover 6. For example, as described later, the thrust can be controlled so that the mover 6 reciprocates in the front-rear direction. The mover 6 has two flat permanent magnets 2 (2a, 2b) magnetized in the vertical direction. The rear permanent magnet 2a and the front permanent magnet 2b are magnetized in opposite directions.
In the present embodiment, the rear permanent magnet 2a has an N pole on the upper side, and the front permanent magnet 2b has an S pole on the upper side. In FIG. 2, the permanent magnets 2 a and 2 b are illustrated, but the mover 6 is not illustrated. As the mover 6, for example, a flat plate in which a flat permanent magnet 2 is fixed can be adopted.
The control unit 102 outputs a drive signal so that the mover 6 reciprocates in a range in which the permanent magnets 2 a and 2 b face the armature 9.
　図３は、図２のＡ－Ａ’線に沿った平面での断面図である（Ａ―Ａ’断面矢視図）。図３に示すように、磁極７及びヨーク７ｅは、例えば鉄などの磁性体で一体的に形成され、磁気回路を構成している。図３の矢印線は、２つの巻線８に電流を流したときの磁束線の一例を示している。磁束の流れの向きは、巻線８に流れる電流の向きにより逆方向になり得るため、図に示す限りではない。この磁束線により、磁極歯７０が磁化される。 FIG. 3 is a cross-sectional view taken along the line A-A 'of FIG. 2 (A-A' cross-sectional view). As shown in FIG. 3, the magnetic pole 7 and the yoke 7e are integrally formed of, for example, a magnetic substance such as iron, and constitute a magnetic circuit. Arrow lines in FIG. 3 indicate an example of magnetic flux lines when current flows through the two windings 8. The direction of flow of the magnetic flux can be reversed depending on the direction of the current flowing through the winding 8 and thus is not limited to the illustrated one. The magnetic pole teeth 70 are magnetized by the magnetic flux lines.
　［可動子６に付与する推力］　
　図４は磁極歯７０の磁化により、可動子６が受ける推力を説明する図である。巻線８に流れる電流により生じる磁極歯７０の極性を、図中の磁極歯７０近傍に付した「Ｎ」、「Ｓ」で表している。また、図４において白抜き矢印は巻線８を流れる電流の向きを示している。図４の左図は、巻線８を流れる電流により、上側の磁極歯７０ａが「Ｓ」、下部の磁極歯７０ｂが「Ｎ」に磁化されることにより、可動子６が前方向に力を受け、可動子６が前に移動した例を示している。図４の右図は、巻線８を流れる電流により、上部の磁極歯７０ａが「Ｎ」、下部の磁極歯７０ｂが「Ｓ」に磁化されることにより、可動子６が後ろ方向に力を受け、可動子６が後ろに移動した例を示している。 [Thrust to be given to mover 6]
FIG. 4 is a diagram for explaining the thrust that the mover 6 receives due to the magnetization of the magnetic pole teeth 70. The polarities of the magnetic pole teeth 70 generated by the current flowing through the winding 8 are represented by "N" and "S" attached near the magnetic pole teeth 70 in the figure. Further, in FIG. 4, the white arrow indicates the direction of the current flowing through the winding 8. The left figure in FIG. 4 shows that the mover 6 is forced forward by magnetizing the upper magnetic pole teeth 70 a to “S” and the lower magnetic pole teeth 70 b to “N” by the current flowing through the winding 8. An example is shown in which the mover 6 has moved forward. In the right view of FIG. 4, the current flowing through the winding 8 magnetizes the upper magnetic pole teeth 70a to "N" and the lower magnetic pole teeth 70b to "S", thereby moving the mover 6 backward in force. An example is shown in which the mover 6 moves backward.
　このように、巻線８に電圧や電流を印加することで、２つの磁極７を含む磁気回路に磁束を供給して、対向する２つの磁極歯７０（磁極歯組）を磁化できる。電圧や電流として、例えば正弦波や矩形波（方形波）といった交流の電圧や電流を与えることで、可動子６を往復運動させる推力を与えることができる。これにより可動子６の運動を制御できる。 As described above, by applying voltage or current to the winding 8, magnetic flux can be supplied to the magnetic circuit including the two magnetic poles 7 to magnetize the two opposing magnetic pole teeth 70 (magnetic pole tooth set). By applying an alternating voltage or current such as a sine wave or a rectangular wave (square wave) as the voltage or current, it is possible to give a thrust for reciprocating the mover 6. Thereby, the motion of the mover 6 can be controlled.
　なお、可動子６に付与する推力は、印加する交流電流や交流電圧の振幅を変更することで変えられる。また、可動子６に付与する推力を既知の方法を用いて適切に変更することで、可動子６の変位を所望に変えられる。ここで、可動子６が往復運動（例えば、図４の左図及び右図のような磁極歯７０の磁化を順次繰り返すことで可動子６に生じる運動）をする場合、交流波形的に変化する可動子６の変位の変化量をストロークと呼ぶ。　
　磁極歯７０は磁性体であるため、永久磁石２を吸引する磁気吸引力が作用する。本実施例では可動子６を挟むよう間隙を介して２つの磁極歯７０を対向配置しているため、可動子６に作用する磁気吸引力の合力を低減できる。 The thrust applied to the mover 6 can be changed by changing the amplitude of the applied alternating current or alternating voltage. Also, the displacement of the mover 6 can be changed as desired by appropriately changing the thrust applied to the mover 6 using a known method. Here, when the mover 6 reciprocates (e.g., a motion generated in the mover 6 by sequentially repeating magnetization of the magnetic pole teeth 70 as shown in the left and right views of FIG. 4), it changes in an alternating waveform. The amount of change in displacement of the mover 6 is called a stroke.
Since the magnetic pole teeth 70 are magnetic members, a magnetic attraction force for attracting the permanent magnet 2 acts. In the present embodiment, since the two magnetic pole teeth 70 are disposed to face each other with a space between them so as to sandwich the mover 6, the total force of the magnetic attraction force acting on the mover 6 can be reduced.
　[可動子６外部の機構]　
　図５は、可動子６に接続される外部機構の説明図であり、例えば、コイルバネである共振バネ２３（アシストバネ）によって構成される外部機構を可動子６の一端に接続し、そのバネ力により可動子６が戻される機構を説明する図である。共振バネ２３は、一端が中間部２４を介して可動子６に接続し、他端が基部２５に固定されている。また、共振バネ２３の延在方向と略平行に延在し、共振バネ２３を案内又は支持する側部２６が設けられている。リニアモータ１０４を往復運動させる場合、可動子６の運動方向が変わる度に、加速と減速を繰り返す。減速時は、可動子６の速度エネルギーが電気エネルギーに変換される（回生動作）が、リニアモータ１０４への配線の抵抗によって損失が生じる。一方、図５のように、可動子６に共振バネ２３（アシストバネ）を付加し、可動子６の質量とバネ定数から決まる機械的な共振周波数で、可動子６を往復運動させる場合、可動子６の速度エネルギーを有効活用でき、高効率なリニアモータ駆動システムを構成することができる。共振バネ２３に代えて、例えば、板バネ、或は、適度なヤング率を有しコイルバネを用いた場合と同様に伸縮するゴム等の弾性体を用いても良い。このように構成すると、可動子６（界磁子６）が鉛直方向に移動する可動子（界磁子）移動型として構成されるが、可動子６に代えて電機子９に弾性体を接続して電機子９を鉛直方向に移動させる電機子移動型として構成しても良い。 [Mechanism outside the mover 6]
FIG. 5 is an explanatory view of an external mechanism connected to the mover 6. For example, an external mechanism constituted by a resonance spring 23 (assist spring) which is a coil spring is connected to one end of the mover 6, and its spring force It is a figure explaining the mechanism by which the needle | mover 6 is returned by this. One end of the resonant spring 23 is connected to the mover 6 via the intermediate portion 24, and the other end is fixed to the base 25. Further, a side portion 26 which extends substantially in parallel with the extending direction of the resonant spring 23 and guides or supports the resonant spring 23 is provided. When the linear motor 104 is reciprocated, acceleration and deceleration are repeated whenever the direction of movement of the mover 6 changes. At the time of deceleration, the velocity energy of the mover 6 is converted into electric energy (regeneration operation), but a loss occurs due to the resistance of the wiring to the linear motor 104. On the other hand, as shown in FIG. 5, in the case where the resonance spring 23 (assist spring) is added to the mover 6 and the mover 6 is reciprocated at the mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, The speed energy of the rotor 6 can be effectively utilized, and a highly efficient linear motor drive system can be configured. Instead of the resonance spring 23, for example, an elastic body such as a leaf spring or rubber which has an appropriate Young's modulus and can be expanded and contracted as in the case of using a coil spring may be used. In this configuration, the mover 6 (field element 6) is configured as a mover (field element) moving type that moves in the vertical direction, but instead of the mover 6, an elastic body is connected to the armature 9 Then, it may be configured as an armature movement type in which the armature 9 is moved in the vertical direction.
　図６は、駆動周波数とストロークの関係の説明図であり、交流電圧の駆動周波数を横軸に、可動子６のストロークを縦軸にとり、これらの関係を示す図である。各駆動周波数における交流電圧の振幅は同一である。図６から分かるように、共振周波数付近で可動子６のストロークが急峻に大きくなり、共振周波数から離れるとストロークが小さくなる特性を示す。共振周波数は、共振バネ２３のバネ定数ｋを可動子６の質量ｍで除した値の平方根で与えられるが、リニアモータ１０４の系によっては、この値は近似値となる。　
　このように、共振周波数又はこの近傍の駆動周波数の交流電圧を印加することで、大きなストローク（大きなエネルギー）で振動させることができる。つまり、可動子６に共振バネ２３等の弾性体を付加したリニアモータ１０４を制御する場合には、可動子６の共振周波数を検出あるいは推定することが重要である。可動子６のストロークを所望に制御する場合においても可動子６の共振周波数を検出あるいは推定することが重要である。 FIG. 6 is an explanatory view of the relationship between the drive frequency and the stroke, showing the relationship between the drive frequency of the AC voltage on the horizontal axis and the stroke of the mover 6 on the vertical axis. The amplitude of the AC voltage at each drive frequency is the same. As can be seen from FIG. 6, the stroke of the mover 6 steeply increases near the resonance frequency, and the stroke decreases as the distance from the resonance frequency decreases. The resonant frequency is given by the square root of the value obtained by dividing the spring constant k of the resonant spring 23 by the mass m of the mover 6, but depending on the system of the linear motor 104, this value is an approximate value.
As described above, by applying an AC voltage at a resonance frequency or a driving frequency near this, oscillation can be performed with a large stroke (large energy). That is, when controlling the linear motor 104 in which an elastic body such as the resonance spring 23 is added to the mover 6, it is important to detect or estimate the resonance frequency of the mover 6. Even when the stroke of the mover 6 is controlled as desired, it is important to detect or estimate the resonance frequency of the mover 6.
　[駆動時の位相関係]　
　図７は、可動子の位置と可動子の速度との位相関係、及び印加電圧とモータ電流の位相関係の説明図である。リニアモータ１０４を駆動した際の、図７の上図に可動子６の位置と速度の時間変化、図７の下図に印加電圧波形とリニアモータ１０４に流れる電流の時間変化の関係を示している。なお、図７の上図と下図は、同じタイミングの波形である。図８は、図７の交流波形をベクトルとして示した図である。共振バネ２３を有する外部機構を接続した可動子６を往復運動させると、可動子６の変位は正弦波状に変化する。可動子６の速度は変位の時間微分であるため、余弦波状に変化する。そのため、これらは直交２軸上にベクトルとして示すことができる。図７および図８より、可動子６の速度、印加電圧、およびモータ電流はほぼ同位相であることがわかる。　
　また、可動子６に共振バネ２３を付加し、可動子６の質量とバネ定数から定まる機械的な共振周波数で可動子６を往復運動させる場合、可動子６の位置の位相は、巻線８への印加電圧Ｖｍ、モータ電流Ｉｍ、及び可動子６の速度の位相それぞれに対して９０度の位相差となることが知られている。すなわち、これらの何れかの関係が成立している時は、共振周波数で駆動していると推定できる。 [Phase relationship during driving]
FIG. 7 is an explanatory view of the phase relationship between the position of the mover and the velocity of the mover, and the phase relationship between the applied voltage and the motor current. When the linear motor 104 is driven, the upper diagram of FIG. 7 shows the time change of the position and velocity of the mover 6, and the lower diagram of FIG. 7 shows the relationship between the applied voltage waveform and the time change of the current flowing through the linear motor 104. . Note that the upper and lower diagrams in FIG. 7 are waveforms at the same timing. FIG. 8 is a diagram showing the AC waveform of FIG. 7 as a vector. When the mover 6 to which the external mechanism having the resonance spring 23 is connected is reciprocated, the displacement of the mover 6 changes sinusoidally. Since the velocity of the mover 6 is a time derivative of displacement, it changes like a cosine wave. Therefore, they can be shown as vectors on two orthogonal axes. From FIGS. 7 and 8, it can be seen that the speed of the mover 6, the applied voltage, and the motor current are substantially in phase.
In addition, when the resonance spring 23 is added to the mover 6 and the mover 6 is reciprocated at the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant, the phase of the position of the mover 6 is the winding 8 It is known that there is a phase difference of 90 degrees with respect to each of the phase of the applied voltage Vm, the motor current Im, and the speed of the mover 6. That is, when any one of these relationships is established, it can be estimated that driving is performed at the resonance frequency.
　リニアモータ１０４の巻線抵抗や巻線インダクタンスの値、あるいは可動子６に付加された負荷要素によっては、巻線８への印加電圧Ｖｍ、モータ電流Ｉｍ、及び可動子６の速度の位相関係は必ずしも９０度の位相差とは限らないため、条件によって変更する制御部を備えることが望ましい。この時、特に、負荷の変動に起因するモータ電流Ｉｍの変化を考慮するのが望ましい。　
　製造バラつきによって可動子６の質量が想定からずれている場合や、可動子６に付加された負荷要素によって、共振バネ２３に接続される質量が変化する場合は、共振周波数が変化してしまう。また、可動子６に付加された負荷要素が位置依存性を有する場合においては、駆動中に共振周波数が変化してしまう。このような場合においても所望のストロークを得るためには、条件によって変化する共振周波数を高精度に検出あるいは推定することが好ましい。以下、共振周波数の検出又は推定方法、駆動周波数の制御、および巻線８に印加する電圧や電流の制御について説明する。 Depending on the value of the winding resistance or winding inductance of the linear motor 104 or the load element added to the mover 6, the phase relationship of the applied voltage Vm to the winding 8, the motor current Im, and the speed of the mover 6 is Since the phase difference is not necessarily 90 degrees, it is desirable to include a control unit that changes according to conditions. At this time, in particular, it is desirable to consider the change in motor current Im due to the change in load.
If the mass of the mover 6 deviates from the assumption due to manufacturing variations, or if the mass connected to the resonant spring 23 changes due to the load element added to the mover 6, the resonance frequency changes. In addition, in the case where the load element added to the mover 6 has position dependency, the resonance frequency changes during driving. Even in such a case, in order to obtain a desired stroke, it is preferable to detect or estimate the resonance frequency which changes depending on conditions with high accuracy. Hereinafter, a method of detecting or estimating the resonance frequency, control of the drive frequency, and control of the voltage or current applied to the winding 8 will be described.
　＜制御部１０２＞　
　図１に示すように、制御部１０２は、位相差検出器１３０、位相差検出器１３０の出力である位相差推定値ｄｌｔθ＾（以下、単に位相差と称する）が位相差指令値ｄｌｔθ＊に追従するように駆動周波数指令値ω＊を調整する駆動周波数調整器１３１、積分器１４０、負荷電流検出器１３６、誘起電圧成分を作成する誘起電圧成分作成器１３５、電圧降下成分を作成する電圧降下成分作成器１３７、電圧指令値Ｖｍ＊を出力する電圧指令値作成器１０３、及び、電圧指令値Ｖｍ＊と三角波キャリア信号を比較して、電圧を出力する電力変換回路１０５を駆動するドライブ信号を出力するＰＷＭ信号作成器１３４から構成される。 <Control unit 102>
As shown in FIG. 1, the control unit 102 is configured such that a phase difference estimated value dltθ ^ (hereinafter simply referred to as a phase difference), which is an output of the phase difference detector 130 and the phase difference detector 130, is a phase difference command value dltθ * . Drive frequency regulator 131 that adjusts drive frequency command value ω * to follow, integrator 140, load current detector 136, induced voltage component creator 135 that creates induced voltage component, voltage drop that creates voltage drop component component generator 137, the voltage command value generator 103 outputs a voltage command value Vm *, and, by comparing the voltage command value Vm * and the triangular wave carrier signal, a drive signal for driving the power conversion circuit 105 which outputs a voltage It comprises a PWM signal generator 134 for outputting.
　制御部１０２は、電流検出器１０７によるモータ電流Ｉｍを入力する。モータ電流Ｉｍは交流波形である。制御部１０２に入力されたモータ電流Ｉｍは、制御部１０２を構成する位相差検出器１３０及び負荷電流検出器１３６に入力される。位相差検出器１３０は、詳細後述する位相指令値である基準位相θ＊とモータ電流Ｉｍの位相差ｄｌｔθ＾を出力する。位相差検出器１３０より出力される位相差ｄｌｔθ＾は駆動周波数調整器１３１に入力される。駆動周波数調整器１３１は、入力された位相差ｄｌｔθ＾に基づき周波数指令値ω＊を出力する。周波数指令値ω＊に基づく印加電圧Ｖｍが、リニアモータ１０４へ出力される。 The control unit 102 receives the motor current Im from the current detector 107. The motor current Im is an AC waveform. The motor current Im input to the control unit 102 is input to the phase difference detector 130 and the load current detector 136 that constitute the control unit 102. The phase difference detector 130 outputs a phase difference dltθ ^ between a reference phase θ * , which is a phase command value described later in detail, and the motor current Im. The phase difference dltθ ^ output from the phase difference detector 130 is input to the drive frequency adjuster 131. The drive frequency adjuster 131 outputs a frequency command value ω * based on the input phase difference dltθ ^. An applied voltage Vm based on the frequency command value ω * is output to the linear motor 104.
　可動子６の質量や共振バネ２３のバネ定数が想定からずれている場合、共振周波数が想定値から変化してしまうが、その影響は巻線８への印加電圧Ｖｍ、モータ電流Ｉｍ、及び可動子６の速度の位相関係から見て取れるため、これらの位相関係を基に制御することで、高効率なリニアモータシステムを構成することができる。可動子６に付加された負荷要素が位置依存性を有する場合においても、巻線８への印加電圧Ｖｍ、モータ電流Ｉｍ、及び可動子６の速度の位相関係を基に制御することで、高効率なリニアモータシステムを構成することができる。　
　以下、制御部１０２を構成する上述の各部の構成或いは動作について説明する。 If the mass of the mover 6 or the spring constant of the resonance spring 23 deviates from the assumption, the resonance frequency changes from the assumed value, but the influence is the voltage Vm applied to the winding 8, the motor current Im, and the movable Since it can be seen from the phase relationship of the velocity of the rotor 6, a highly efficient linear motor system can be configured by controlling based on these phase relationships. Even in the case where the load element added to the mover 6 has position dependency, the control is performed based on the phase relationship of the applied voltage Vm to the winding 8, the motor current Im, and the speed of the mover 6 An efficient linear motor system can be configured.
Hereinafter, configurations or operations of the above-described units constituting the control unit 102 will be described.
　＜基準位相作成器＞　
　本実施例の位相指令値である基準位相θ＊は、図１の駆動周波数調整器１３１の出力である駆動周波数指令値ω＊を基準位相作成器としての積分器１４０で積分することで得る。すなわち、基準位相θ＊は、印加電圧Ｖｍ（θ＊）の目標周波数に相当する駆動周波数指令値ω＊を持つ波動の各時刻の位相θ＊である。このように、本実施例では基準位相θ＊として駆動周波数調整器１３１の駆動周波数指令値ω＊を用いているが、例えば可動子６を含む振動体の機械共振周波数に固定しても良い。　
　基準位相θ＊は、駆動周波数指令値ω＊が一定の間は、例えば、各時刻に対して［－π，π］、［０，２π］、又はこれより広い範囲を値域とする、のこぎり波としたり、時刻に対して線形に増加するようにしても良い。後述するように駆動周波数指令値ω＊が変動した場合は、これに応じてのこぎり波や線形な増加の形状が変動する（傾きが変化する）。 <Reference phase generator>
The reference phase θ *, which is the phase command value of this embodiment, is obtained by integrating the drive frequency command value ω * , which is the output of the drive frequency adjuster 131 in FIG. 1, by the integrator 140 as a reference phase generator. That is, the reference phase theta * applied voltage Vm (θ *) is the phase theta * at each time of the wave having a driving frequency command value omega * corresponding to the target frequency. Thus, reference the phase theta * as is used driving frequency command value of the drive frequency regulator 131 omega *, for example, may be fixed to the mechanical resonance frequency of the vibration body including the movable member 6 in the present embodiment.
The reference phase θ * is a sawtooth wave having, for example, [-π, π], [0, 2π], or a range wider than this for each time while the drive frequency command value ω * is constant. Alternatively, it may increase linearly with time. As described later, when the drive frequency command value ω * changes, the shape of the sawtooth wave or the linear increase changes accordingly (the slope changes).
　＜位相差検出器１３０＞　
　可動子６が往復運動している場合、可動子６の位置ｘｍ及び速度、モータ電流Ｉｍは周期関数となる。周期関数はフーリエ級数で表せるため、フーリエ変換式を用いて可動子６の位置ｘｍを表すと、次式（１）のように定義できる。 <Phase difference detector 130>
When the mover 6 reciprocates, the position xm and velocity of the mover 6 and the motor current Im have a periodic function. Since the periodic function can be expressed by a Fourier series, when the position xm of the mover 6 is expressed using a Fourier transform equation, it can be defined as the following equation (1).
Figure JPOXMLDOC01-appb-M000001
 　ここで、ｘ０は直流オフセット値、ａｎおよびｂｎはｎ次のフーリエ係数であり、次式（２）及び式（３）で求められる。　
Figure JPOXMLDOC01-appb-M000001
 Here, x 0 is a direct current offset value, and a n and b n are n-th order Fourier coefficients, which are obtained by the following equations (2) and (3).
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000003
 　ここで、Ｔ０は基本波の周期（可動子６の往復運動する周期）、つまり１次周波数（駆動周波数）の逆数である。可動子６に共振バネ２３が付加された、いわゆるバネマス系においては、可動子６の質量とバネ定数から決まる機械的な共振周波数が支配的な成分となる。そのため、基本波に注目すれば良い。
Figure JPOXMLDOC01-appb-M000003
 Here, T 0 is the period of the fundamental wave (the period in which the mover 6 reciprocates), that is, the reciprocal of the primary frequency (drive frequency). In a so-called spring-mass system in which a resonant spring 23 is added to the mover 6, a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant is a dominant component. Therefore, one should pay attention to the fundamental wave.
　可動子６を共振周波数で駆動させようと制御する場合、高次成分は重要ではなく、１次成分、つまり駆動周波数成分に注目すれば良い。特に、可動子６の位置ｘｍの１次周波数成分（駆動周波数成分）の位相が重要である。１次のフーリエ係数の逆正接により、正弦波状の印加電圧Ｖｍに対する可動子６の位置ｘｍを次式（４）で求めることができる。　　 When the mover 6 is controlled to be driven at the resonance frequency, the high order component is not important, and attention may be paid to the first order component, that is, the drive frequency component. In particular, the phase of the primary frequency component (drive frequency component) of the position xm of the mover 6 is important. By the inverse tangent of the first-order Fourier coefficient, the position xm of the mover 6 with respect to the sinusoidal applied voltage Vm can be determined by the following equation (4).
Figure JPOXMLDOC01-appb-M000004
 　式（４）では、積分範囲は、－２π～０となっている。これは、位相差検出器１３０をマイコンやＤＳＰ（Ｄｉｇｉｔａｌ　Ｓｉｇｎａｌ　Ｐｒｏｃｅｓｓｏｒ）等の半導体集積回路等で実現する場合に、過去の情報しか取得できないためである。
Figure JPOXMLDOC01-appb-M000004
 In the equation (4), the integration range is −2π to 0. This is because when the phase difference detector 130 is realized by a semiconductor integrated circuit or the like such as a microcomputer or a DSP (Digital Signal Processor), only past information can be acquired.
　上述の式（１）～式（４）では、基準位相θ＊に対する可動子６の位置ｘｍの位相を算出しているが、同様にモータ電流Ｉｍを用いることにより、次式（５）に示すように基準位相θ＊に対するモータ電流Ｉｍの位相を算出できる。　 In the above equations (1) to (4), the phase of the position xm of the mover 6 with respect to the reference phase θ * is calculated, but by using the motor current Im in the same manner, Thus, the phase of the motor current Im with respect to the reference phase θ * can be calculated.
Figure JPOXMLDOC01-appb-M000005
 　図９は、図１に示す制御部１０２を構成する位相差検出器１３０の構成例を示す説明図であって、式（５）をブロック図で示した場合の説明図である。入力値の正弦を出力する正弦演算器８１及び入力値の余弦を出力する余弦演算器８２のそれぞれに、位相指令値である基準位相θ＊を入力し、基準位相θ＊（位相指令値）に対する正弦及び余弦を得る。
正弦及び余弦それぞれをモータ電流Ｉｍと乗算した値が乗算器９２から出力される。その出力をそれぞれ積分器９４ａ，９４ｂで積分すると、正弦及び余弦それぞれの１次のフーリエ係数を得る。すなわち、フーリエ展開の駆動周波数ωより高次の周波数成分を消去できるので、高次のノイズに対してロバストに構成できる。　
　積分器９４ａ，９４ｂの出力を逆正接器８６に入力する。逆正接器８６は、入力された正弦及び余弦成分を基に逆正接値を出力する。本実施例の逆正接器８６は、分子を積分器９４ａの出力、分母を積分器９４ｂの出力とした位相の逆正接値を出力するが、分子と分母を逆にした値を出力しても良い。
Figure JPOXMLDOC01-appb-M000005
 FIG. 9 is an explanatory view showing a configuration example of the phase difference detector 130 constituting the control unit 102 shown in FIG. 1, and an explanatory view in the case where the equation (5) is shown in a block diagram. A reference phase θ * , which is a phase command value, is input to each of a sine calculator 81 that outputs the sine of the input value and a cosine calculator 82 that outputs the cosine of the input value, with respect to the reference phase θ * (phase command value). Get sine and cosine.
A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92. The outputs are integrated by integrators 94a and 94b, respectively, to obtain first-order Fourier coefficients of sine and cosine respectively. That is, since the frequency components higher than the drive frequency ω of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise.
The outputs of the integrators 94a and 94b are input to the inverse tangent unit 86. The arctangent device 86 outputs an arctangent value based on the input sine and cosine components. The inverse tangent unit 86 of this embodiment outputs the inverse tangent value of the phase with the numerator as the output of the integrator 94a and the denominator as the output of the integrator 94b, but even if the numerator and the denominator are inverted, good.
　図１０は、駆動周波数と位相差検出器１３０からの出力の関係の説明図である。すなわち、交流電圧の駆動周波数（横軸）と、逆正接器８６の出力値（位相差ｄｌｔθ＾（θｐｏｓ＾））（縦軸）の関係を示す図である。図１０から分かるように、本実施例では、駆動周波数が共振周波数である場合、０°が逆正接器８６から出力される。逆正接器８６から出力される値は、駆動周波数が共振周波数より高い場合は０°より大きく、駆動周波数が共振周波数より低い場合は、０°より小さい。これにより、基準位相θ＊に対する位相差検出器１３０への入力交流信号（本実施例では、モータ電流Ｉｍ）の１次周波数成分の基準位相θ＊に対する位相差ｄｌｔθ＾を求めることができ、共振周波数の推定が可能になる。そして例えば、位相差ｄｌｔθ＾が０°超の場合は、位相差ｄｌｔθ＾が低下するように制御し、０°未満の場合は、増加するように制御すれば良い。基準位相θ＊と基本周波数ωとが同値となるときの位相差ｄｌｔθ＾を目標値ｄｌｔθ＊として制御することが好ましい。 FIG. 10 is an explanatory diagram of the relationship between the drive frequency and the output from the phase difference detector 130. As shown in FIG. That is, it is a figure which shows the relationship between the drive frequency (horizontal axis) of an alternating voltage, and the output value (phase difference dlt (theta) ^ ((theta) pos ^)) (vertical axis | shaft) of the inverse tangent device 86. As can be seen from FIG. 10, in the present embodiment, when the drive frequency is a resonance frequency, 0 ° is output from the inverse tangent device 86. The value output from the inverse tangent device 86 is larger than 0 ° when the drive frequency is higher than the resonant frequency, and smaller than 0 ° when the drive frequency is lower than the resonant frequency. Thus, (in this embodiment, the motor current Im) input AC signal to the phase difference detector 130 with respect to the reference phase theta * can obtain the phase difference Dltshita ^ of primary frequency components of the reference phase theta *, resonance It is possible to estimate the frequency. Then, for example, when the phase difference dltθ ^ is more than 0 °, control is performed so that the phase difference dltθ ^ decreases, and when it is less than 0 °, control may be performed so as to increase. It is preferable to control the phase difference dltθ ^ when the reference phase θ * and the fundamental frequency ω have the same value as the target value dltθ * .
　なお、積分器９４ａ、９４ｂに代えて、不完全積分器を用いることができる。不完全積分器はローパスフィルタ（低域通過フィルタ）の一種で、１次遅れフィルタと同様の構成にできる。もちろん、１次遅れフィルタに限らず、２次遅れフィルタ等、他の既知の構成でローパスフィルタを構成しても良い。その他、不完全積分器に代えて、又は追加して、積分器９４ａ，９４ｂ（又は不完全積分器）より前段に、ハイパスフィルタ（図示せず）を設けることができる。ハイパスフィルタの遮断周波数としては、例えば１０Ｈｚ又は５Ｈｚ以下にすることができる。 Note that an incomplete integrator can be used instead of the integrators 94a and 94b. The incomplete integrator is a type of low pass filter (low pass filter) and can be configured the same as the first order delay filter. Of course, the low pass filter may be configured with another known configuration such as a second order lag filter, not limited to the first order lag filter. Alternatively, a high pass filter (not shown) may be provided in front of the integrators 94a and 94b (or the incomplete integrators) instead of or in addition to the incomplete integrators. The cutoff frequency of the high pass filter can be, for example, 10 Hz or 5 Hz or less.
　このように、位相差検出器１３０は、駆動周波数成分の１次のフーリエ係数の比の逆正接を用い、交流電圧指令値Ｖ＊に対するモータ電流Ｉｍの位相θを求めるとき、位相差検出器１３０への入力交流信号であるモータ電流Ｉｍの１次周波数成分のみに大きな感度を有する。つまり、例えば、モータ電流Ｉｍに、直流オフセットや高次のノイズが重畳された場合においても、基準位相θ＊に対する位相差検出器１３０への入力交流信号であるモータ電流Ｉｍの１次周波数成分の位相ｄｌｔθをより正確に求められる。また、ハイパスフィルタを上記のように設ける場合は、さらに駆動周波数ωより小さな周波数に対してもロバストに構成できる。　
　したがって、モータ電流Ｉｍの検出方法として、ノイズが重畳され易いシステム、例えばインダクタンスの可動子位置依存性が大きいシステムや、近傍に別の機器が存在するシステムを採用する場合に、特に有効な制御を実現できる。このように、高精度に共振周波数を検出あるいは推定し、高効率なリニアモータ駆動を実現することができる。 Thus, when phase difference detector 130 determines the phase θ of motor current Im with respect to AC voltage command value V * using the inverse tangent of the ratio of the first-order Fourier coefficients of the drive frequency components, phase difference detector 130 There is great sensitivity only to the primary frequency component of the motor current Im, which is an input AC signal to the input. That is, for example, even when DC offset and high-order noise are superimposed on the motor current Im, the primary frequency component of the motor current Im, which is an AC signal input to the phase difference detector 130 with respect to the reference phase θ * . The phase dltθ can be determined more accurately. Further, when the high pass filter is provided as described above, it can be configured robustly to a frequency smaller than the drive frequency ω.
Therefore, particularly effective control is adopted when adopting a system in which noise is easily superimposed, for example, a system in which the mover position dependency of the inductance is large, or a system in which another device is present in the vicinity as a method of detecting the motor current Im. realizable. Thus, the resonance frequency can be detected or estimated with high accuracy, and highly efficient linear motor drive can be realized.
　＜駆動周波数調整器１３１＞　
　図１１は、図１に示す制御部１０２を構成する駆動周波数調整器１３１の構成例を示す説明図である。駆動周波数調整器１３１は、位相差指令値ｄｌｔθ＊（例えば、０°）と位相差検出器１３０で求めた位相差ｄｌｔθ＾の差を減算器９１で求め、これに乗算器９２ｂで比例ゲインＫｐ＿ａｄｔｒを乗じて比例制御した演算結果と、乗算器９２ｃで積分ゲインＫｉ＿ａｄｔｒを乗じ、その結果を積分器９４ｃで積分する積分制御した演算結果とを加算器９０で加算し、当該加算結果に更に周波数指令初期値（ω０）を加算することで駆動周波数指令値ω＊を出力する。　
　なお、周波数指令初期値（ω０）は、上位の制御器（図示せず）から得ても良いし、予め例えば０°と設定しても良い。また、本実施例の駆動周波数調整器１３１は、比例積分制御の構成であるが、比例制御や積分制御など、他の制御構成も適用できる。 <Drive frequency adjuster 131>
FIG. 11 is an explanatory view showing a configuration example of the drive frequency adjuster 131 which constitutes the control unit 102 shown in FIG. The driving frequency adjuster 131 obtains the difference between the phase difference command value dltθ * (for example, 0 °) and the phase difference dltθ ^ obtained by the phase difference detector 130 by the subtractor 91, and the multiplier 92b calculates the difference by the proportional gain Kp_adtr. The multiplication result is proportionally controlled and the integral gain Ki_adtr is multiplied by the multiplier 92c, and the result is integrated by the integrator 94c. The addition result is added by the adder 90, and the frequency command is further added to the frequency command The drive frequency command value ω * is output by adding the initial value (ω 0 ).
The frequency command initial value (ω 0 ) may be obtained from an upper controller (not shown), or may be set to, for example, 0 ° in advance. Further, although the drive frequency adjuster 131 of the present embodiment is a configuration of proportional integral control, other control configurations such as proportional control and integral control can also be applied.
　[高効率駆動の実現]　
　リニアモータ１０４を可動子６の質量とバネ定数から定まる機械的な共振周波数で駆動する場合の位相差検出器１３０と駆動周波数調整器１３１の動作を説明する。　
　例えば、可動子６の質量が設計値よりも重かった場合、実際の共振周波数は、設計値よりも低くなる。つまり、可動子６の質量設計値を用いて駆動周波数の初期値を決めた場合（設計値を利用して駆動周波数指令値ω＊の初期値を決めた場合）には、実際の共振周波数よりも高い周波数で駆動することになる。この時、位相差検出器１３０で求めた位相差ｄｌｔθ＾は、位相差指令値ｄｌｔθ＊よりも大きい値となる。そのため、駆動周波数調整器１３１は、駆動周波数指令値ω＊を減少させる制御を実行し、その結果、駆動周波数指令値ω＊が実際の共振周波数に一致する。したがって、可動子６の速度エネルギーを有効活用でき、高効率にリニアモータ１０４を駆動することができる。 [Realization of high efficiency drive]
The operation of the phase difference detector 130 and the drive frequency adjuster 131 in the case where the linear motor 104 is driven at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant will be described.
For example, when the mass of the mover 6 is heavier than the design value, the actual resonance frequency is lower than the design value. That is, when the initial value of the drive frequency is determined using the mass design value of the mover 6 (when the initial value of the drive frequency command value ω * is determined using the design value), based on the actual resonance frequency Will also drive at a high frequency. At this time, the phase difference dltθ ^ obtained by the phase difference detector 130 is a value larger than the phase difference command value dltθ * . Therefore, the drive frequency adjuster 131 executes control to decrease the drive frequency command value ω *, and as a result, the drive frequency command value ω * matches the actual resonance frequency. Therefore, the speed energy of the mover 6 can be effectively used, and the linear motor 104 can be driven with high efficiency.
　＜誘起電圧成分作成器１３５＞　
　図１２は、図１に示す制御部１０２を構成する誘起電圧成分作成器１３５の構成例を示す説明図である。誘起電圧成分作成器１３５は、可動子６の速度に応じて生じる誘起電圧に相当する電圧指令値Ｖｍ１ ＊を生成する。　
　図１２の上図は、ストローク制御器を構成しない、いわゆるオープン指令型の構成例である。誘起電圧成分作成器１３５は、上位の制御器（図示せず）などから得るストローク指令値ｌ＊と、駆動周波数調整器１３１の出力である駆動周波数指令値ω＊を基準位相作成器としての積分器１４０で積分することで得られる基準位相θ＊を入力する。　
　入力値の余弦を出力する余弦演算器８２ｂに基準位相θ＊を入力し、その余弦出力と、ストローク指令値ｌ＊と、基準位相θ＊とを乗算器９２ｄにて乗算し、速度指令値ｖｍ＊を得る。速度指令値ｖｍ＊に、乗算器９２ｅにて誘起電圧定数Ｋｅ＊を乗じ、誘起電圧に相当する電圧指令値（Ｖｍ１ ＊）を生成する。 <Induced Voltage Component Creator 135>
FIG. 12 is an explanatory view showing a configuration example of the induced voltage component generator 135 which constitutes the control unit 102 shown in FIG. The induced voltage component generator 135 generates a voltage command value V m1 * corresponding to the induced voltage generated according to the speed of the mover 6.
The upper diagram of FIG. 12 is a so-called open command type configuration example that does not constitute a stroke controller. The induced voltage component generator 135 integrates a stroke command value l * obtained from a host controller (not shown) or the like, and a drive frequency command value ω * which is an output of the drive frequency adjuster 131 as a reference phase generator. A reference phase θ * obtained by integration in the unit 140 is input.
The reference phase θ * is input to a cosine calculator 82b that outputs the cosine of the input value, and the cosine output, the stroke command value l *, and the reference phase θ * are multiplied by a multiplier 92d to obtain a speed command value vm. Get * . The speed command value vm * is multiplied by the induced voltage constant Ke * in the multiplier 92e to generate a voltage command value (V m1 * ) corresponding to the induced voltage.
　速度指令値ｖｍ＊は、可動子６の位置の指令値ｘｍ＊の微分から求めることもでき、図１２の構成の基となる演算式を次式（６）に示す。　 The velocity command value vm * can also be obtained from the derivative of the command value xm * at the position of the mover 6, and an arithmetic expression as a basis of the configuration of FIG. 12 is shown in the following equation (6).
Figure JPOXMLDOC01-appb-M000006
 　速度指令値ｖｍ＊は、ストローク指令値ｌ＊を振幅とした正弦波で表すことができる。
正弦波を時間微分すると余弦波になるため、図１２の上図に示す構成とすると微分演算を省略することができ、制御部１０２への実装が容易になる。
Figure JPOXMLDOC01-appb-M000006
 The speed command value vm * can be represented by a sine wave whose amplitude is the stroke command value l * .
Since the sine wave becomes a cosine wave when time-differentiated, the differential operation can be omitted if the configuration shown in the upper part of FIG. 12 is adopted, and the mounting on the control unit 102 becomes easy.
　図１２の下図は、ストローク制御器を構成する、いわゆるクローズ指令型の構成例である。誘起電圧成分作成器１３５ａは、既知の位置センサによるストローク検出値、またはストローク推定値をフィードバックし、ストローク制御器１５３でストローク指令値を制御する。 The lower part of FIG. 12 is a configuration example of a so-called close command type that constitutes a stroke controller. The induced voltage component generator 135a feeds back a stroke detection value or a stroke estimation value by a known position sensor, and controls a stroke command value by the stroke controller 153.
　＜ストローク制御器１５３＞　
　図１３は、図１２の下図に示す誘起電圧成分作成器１３５ａを構成するストローク制御器１５３の構成例を示す説明図である。図１３に示すように、ストローク制御器１５３は、ストローク指令値ｌ＊と、ストローク検出値またはストローク推定値との差を減算器９１ａで求め、これに乗算器９２ｄで比例ゲインＫｐ＿ａｓｔｒを乗じて比例制御した演算結果と、乗算器９２ｅで積分ゲインＫｉ＿ａｓｔｒを乗じ、その結果を積分器９４ｄで積分する積分制御した演算結果とを加算器９０ａで加算し、調整後のストローク指令値ｌ＊＊を出力する。本実施例のストローク制御器１５３は比例積分制御の構成であるが、比例制御や積分制御など、他の制御構成も適用できる。 <Stroke controller 153>
FIG. 13 is an explanatory view showing a configuration example of the stroke controller 153 which constitutes the induced voltage component generator 135a shown in the lower part of FIG. As shown in FIG. 13, the stroke controller 153 obtains the difference between the stroke command value l * and the stroke detection value or the stroke estimation value with a subtractor 91a, and multiplies this by a proportional gain Kp_astr in a multiplier 92d to be proportional The control result of the control is multiplied by the integral gain Ki_astr in the multiplier 92e, and the result of the integration is integrated in the integrator 94d. The result of integration control is added by the adder 90a, and the adjusted stroke command value l ** is output. Do. Although the stroke controller 153 of the present embodiment is a configuration of proportional integral control, other control configurations such as proportional control and integral control can also be applied.
　なお、位置推定をする場合は、リニアモータ１０４に印加する電圧Ｖｍと、リニアモータ１０４に流れる電流Ｉｍを利用して、例えば次式（７）で位置推定値ｘｍ＾を求める。　 When position estimation is performed, the position estimation value xm ^ is obtained, for example, by the following equation (7) using the voltage Vm applied to the linear motor 104 and the current Im flowing through the linear motor 104.
Figure JPOXMLDOC01-appb-M000007
 　式（７）中、Ｖｍ＊はリニアモータ１０４に印加する電圧指令値Ｖｍ＊である。
Figure JPOXMLDOC01-appb-M000007
 In the equation (7), Vm * is a voltage command value Vm * applied to the linear motor 104.
　図１４は、位置推定部の構成例を示す説明図であって、式（７）をブロック図で示した場合の説明図である。図１４に示すように、位置推定部３０８は、電圧指令値Ｖｍ＊及びモータ電流Ｉｍを入力し、モータ電流Ｉｍにリニアモータ１０４の巻線抵抗値Ｒｍ＊を乗じた結果を電圧指令値Ｖｍ＊より減じて、この減じた結果を積分器にて演算した結果を得る。また、モータ電流Ｉｍにリニアモータ１０４の巻線インダクタンス値Ｌｍ＊を乗じた結果を得て、上記積分器にて演算した結果より減じて、この減じた結果を誘起電圧定数Ｋｅ＊にて除することで位置推定値ｘｍ＾を出力する。なお、位置推定部３０８には、上記以外にも既知の同期式モータの位置推定方法を適用することができる。位置推定値ｘｍ＾を基にストロークを演算し、図１３に示すトローク検出器１５３に入力すれば、所望のストロークに制御することができる。 FIG. 14 is an explanatory view showing a configuration example of the position estimation unit, and an explanatory view in the case where the equation (7) is shown by a block diagram. As shown in FIG. 14, position estimation unit 308 receives voltage command value Vm * and motor current Im, and multiplies the motor current Im by winding resistance value Rm * of linear motor 104 to obtain voltage command value Vm *. The result is calculated by an integrator. Further, a result obtained by multiplying the motor current Im by the winding inductance value Lm * of the linear motor 104 is obtained, and the result is subtracted from the result calculated by the above integrator, and this reduced result is divided by the induced voltage constant Ke * Outputs the position estimation value xm ^. In addition, the position estimation part 308 can apply the position estimation method of the synchronous type motor well-known besides the above. A stroke can be calculated based on the position estimation value xm ^ and input to the troke detector 153 shown in FIG. 13 to control to a desired stroke.
　＜負荷電流検出器１３６＞　
　図１５は、図１に示す制御部１０２を構成する負荷電流検出器１３６の構成例を示す説明図である。本実施例の負荷電流検出器１３６は、フーリエ変換式を用いて負荷電流成分の振幅を抽出する。　
　負荷電流検出器１３６は、図９に示した位相差検出器１３０の構成と同様に、入力値の正弦を出力する正弦演算器８１と、入力値の余弦を出力する余弦演算器８２のそれぞれに、位相指令値である基準位相θ＊を入力し、基準位相θ＊に対する正弦及び余弦を得る。
正弦及び余弦それぞれをモータ電流Ｉｍと乗算した値が乗算器９２からそれぞれ出力される。その出力をそれぞれ積分器９４ｅ，９４ｆで積分すると、正弦及び余弦それぞれの１次のフーリエ係数を得る。すなわち、フーリエ展開の駆動周波数ωより高次の周波数成分を消去できるので、高次のノイズに対してロバストに構成できる。 <Load current detector 136>
FIG. 15 is an explanatory view showing a configuration example of the load current detector 136 which constitutes the control unit 102 shown in FIG. The load current detector 136 of this embodiment extracts the amplitude of the load current component using a Fourier transform equation.
Similar to the configuration of the phase difference detector 130 shown in FIG. 9, the load current detector 136 has a sine calculator 81 that outputs a sine of the input value and a cosine calculator 82 that outputs a cosine of the input value. , A reference phase θ * which is a phase command value is input to obtain sine and cosine with respect to the reference phase θ * .
A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92, respectively. The outputs are integrated by integrators 94e and 94f, respectively, to obtain first-order Fourier coefficients of sine and cosine respectively. That is, since the frequency components higher than the drive frequency ω of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise.
　積分器９４ｅ，９４ｆの出力を二乗し、平方根演算器９６に入力する。すなわち、正弦及び余弦それぞれの１次のフーリエ係数となる、正弦成分及び余弦成分の二乗和平方根を得て、基本波電流の振幅を得る。負荷の増加に伴い、基本波電流の振幅も増加するため、図１５の構成により、負荷電流Ｉｍ＿ｌｄを検出することができる。 The outputs of the integrators 94 e and 94 f are squared and input to the square root calculator 96. That is, the amplitude of the fundamental current is obtained by obtaining the root sum square root of the sine component and the cosine component, which are first-order Fourier coefficients of the sine and the cosine, respectively. Since the amplitude of the fundamental current also increases as the load increases, the configuration of FIG. 15 can detect the load current Im_ld.
　図１６は、図１に示す制御部１０２を構成する負荷電流検出器の他の構成例を示す説明図である。負荷電流検出器１３６ｂは、入力値の余弦を出力する余弦演算器８２に位相指令値である基準位相θ＊を入力し、基準位相θ＊に対する余弦を得る。得られた余弦とモータ電流Ｉｍを乗算器９２にて乗算することで、モータ電流Ｉｍの基本周波数のｃｏｓ成分（Ｉｍ＿ｃｏｓ）を抽出する。次に、モータ電流Ｉｍの基本周波数のｃｏｓ成分（Ｉｍ＿ｃｏｓ）を一次遅れフィルタ１４１でローパスフィルタ（低域通過フィルタ）処理し、負荷電流Ｉｍ＿ｌｄとして出力する。　
　なお、図１６には１次遅れフィルタの構成例を示したが、１次遅れフィルタに限らず、２次遅れフィルタ等、他の既知の構成でローパスフィルタを構成しても良い。 FIG. 16 is an explanatory view showing another configuration example of the load current detector constituting the control unit 102 shown in FIG. The load current detector 136b inputs the reference phase θ * , which is a phase command value, to the cosine calculator 82 that outputs the cosine of the input value, and obtains a cosine with respect to the reference phase θ * . By multiplying the obtained cosine and the motor current Im by the multiplier 92, the cos component (Im_cos) of the fundamental frequency of the motor current Im is extracted. Next, the cos component (Im_cos) of the fundamental frequency of the motor current Im is low-pass filtered (low-pass filter) processed by the first-order lag filter 141 and output as a load current Im_ld.
Although FIG. 16 shows a configuration example of the first-order lag filter, the low-pass filter may be configured with another known configuration, such as a second-order lag filter, instead of the first-order lag filter.
　図７および図８で示したように、可動子の位置を正弦波とすると、負荷電流は余弦成分が支配的になる。そのため、図１６の構成は抽出した余弦成分を、フィルタ処理をすることで、負荷電流を検出することができる。図１６の構成とした場合、演算負荷低減に有効である。 As shown in FIGS. 7 and 8, assuming that the position of the mover is a sine wave, the load current is dominated by the cosine component. Therefore, the configuration of FIG. 16 can detect the load current by filtering the extracted cosine component. The configuration of FIG. 16 is effective in reducing the calculation load.
　＜電圧降下成分作成器１３７＞　
　図１７は、図１に示す制御部１０２を構成する電圧降下成分作成器１３７の構成例を示す説明図である。電圧降下成分作成器１３７は、負荷電流検出器１３６で検出した負荷電流Ｉｍ＿ｌｄと、基準位相θ＊とを入力し、リニアモータ１０４の抵抗及びインダクタンスによる電圧降下分に相当する電圧指令値（Ｖｍ２ ＊）を出力する。 <Voltage drop component creator 137>
FIG. 17 is an explanatory view showing a configuration example of the voltage drop component creator 137 which constitutes the control unit 102 shown in FIG. The voltage drop component creator 137 inputs the load current Im_ld detected by the load current detector 136 and the reference phase θ *, and a voltage command value (V m2 corresponding to the voltage drop due to the resistance and inductance of the linear motor 104 Output * ).
　負荷電流Ｉｍ＿ｌｄに、予め設定あるいは推定した抵抗値Ｒｍ＊とインダクタンス値Ｌｍ＊を乗算器９２で乗ずる。次に、負荷電流Ｉｍ＿ｌｄと抵抗値Ｒｍ＊を乗じた値に、入力値である基準位相θ＊の余弦を出力する余弦演算器８２の出力を乗じ、負荷電流Ｉｍ＿ｌｄとインダクタンス値Ｌｍ＊を乗算器９２にて乗じた値に、入力値である基準位相θ＊の負の正弦を出力する正弦演算器８１ａの出力を乗じる。更に、これらを加算器９０で足し合わせ、電圧降下分に相当する電圧指令値（Ｖｍ２ ＊）として出力する。 The multiplier 92 multiplies the load current Im_ld by the resistance value Rm * set or estimated in advance and the inductance value Lm * . Next, the product of the load current Im_ld and the resistance value Rm * is multiplied by the output of the cosine calculator 82 that outputs the cosine of the reference phase θ * that is the input value, and the load current Im_ld and the inductance value Lm * are multiplied. The value multiplied by 92 is multiplied by the output of the sine calculator 81a that outputs the negative sine of the reference phase θ * that is the input value. Furthermore, these are added together by the adder 90 and output as a voltage command value (V m2 * ) corresponding to the voltage drop.
　＜電圧指令値作成器１０３＞　
　図１８は、図１に示す制御部１０２を構成する電圧指令値作成器１０３の構成例を示す説明図である。図１８に示すように、電圧指令値作成器１０３は、誘起電圧成分作成器１３５から入力される誘起電圧に相当する電圧指令値（Ｖｍ１ ＊）と、電圧降下成分作成器１３７から入力される電圧降下分に相当する電圧指令値（Ｖｍ２ ＊）とを加算器９０で加算して、交流電圧指令値（Ｖｍ ＊）として出力する。入力する２つの電圧指令値（Ｖｍ１ ＊及びＶｍ２ ＊）はどちらも交流波形となるため、電圧指令値作成器１０３ではベクトル和（ベクトル加算）を出力しているのと等価である。　
　電圧指令値作成器１０３は電圧位相算出器１３８を有し、出力する交流電圧指令値Ｖｍ ＊の位相である交流電圧指令位相θＶｍ ＊を出力する。 <Voltage command value generator 103>
FIG. 18 is an explanatory view showing a configuration example of the voltage command value creation unit 103 configuring the control unit 102 shown in FIG. As shown in FIG. 18, voltage command value creator 103 receives a voltage command value (V m1 * ) corresponding to the induced voltage input from induced voltage component creator 135 and a voltage drop component creator 137. The voltage command value (V m2 * ) corresponding to the voltage drop is added by the adder 90 and is output as an AC voltage command value (V m * ). Since both of the two voltage command values (V m1 * and V m2 * ) to be input are AC waveforms, the voltage command value generator 103 is equivalent to outputting a vector sum (vector addition).
Voltage command value generator 103 includes a voltage phase calculator 138, and outputs is the output AC voltage command values V m * phase AC voltage command phase theta Vm *.
　図１９は、軽負荷時及び重負荷時における電圧指令値作成器でのベクトル和を示すベクトル図であり、電圧指令値作成器１０３でのベクトル加算をベクトル図で示した説明図である。図１９の左図は軽負荷時、すなわち負荷電流が小さい時のベクトル図で、図１９の右図は重負荷時、すなわち負荷電流が大きい時のベクトル図である。図１９の左図及び右図とも、ストローク指令値は同じ値のため、誘起電圧に相当する電圧指令値（Ｖｍ１ ＊）は、同じベクトルとなっている。なお、図１９の左図及び右図において、反時計回りを正としている。　
　このように、誘起電圧に相当する電圧指令値（Ｖｍ１ ＊）に対し、抵抗及びインダクタンスによる電圧降下分に相当する電圧指令値（Ｖｍ２ ＊）をベクトル加算することで、負荷電流に応じて電圧指令値Ｖｍ＊の振幅が増加すると共に、交流電圧指令位相（θＶｍ＊）の分、位相が進んだ、単相の交流電圧指令値Ｖｍ ＊が出力される。つまり、可動子６のストロークが同じ、換言すれば、速度指令値が同じでも、負荷条件に応じて電圧指令値Ｖｍ ＊を適切に制御する。 FIG. 19 is a vector diagram showing vector sums in the voltage command value creator at light load and heavy load, and is an explanatory view showing vector addition at the voltage command value creator 103 in a vector diagram. The left diagram of FIG. 19 is a vector diagram at light load, that is, when the load current is small, and the right diagram of FIG. 19 is a vector diagram at heavy load, that is, when the load current is large. In both the left and right views of FIG. 19, since the stroke command value is the same value, the voltage command value (V m1 * ) corresponding to the induced voltage is the same vector. In the left and right views of FIG. 19, the counterclockwise direction is positive.
Thus, according to the load current, the voltage command value (V m2 * ) corresponding to the voltage drop due to the resistance and inductance is vector added to the voltage command value (V m1 * ) corresponding to the induced voltage. the amplitude of the voltage command value Vm * is increased, minute AC voltage command phase (theta Vm *), the phase advances, the AC voltage command value of the single-phase V m * is output. That is, the voltage command value V m * is appropriately controlled in accordance with the load condition even if the stroke of the mover 6 is the same, in other words, even if the speed command value is the same.
　ストローク指令値ｌ＊、位相指令値θ＊、速度指令値ｖｍ＊のいずれかを変更することにより、リニアモータ１０４に印加する電圧Ｖｍ＊を調整することができる。そのため、印加電圧の振幅及び周波数を調整することで、駆動周波数を共振周波数に制御することやストロークを制御することが可能となる。 The voltage Vm * applied to the linear motor 104 can be adjusted by changing one of the stroke command value l * , the phase command value θ * , and the speed command value vm * . Therefore, by adjusting the amplitude and frequency of the applied voltage, it is possible to control the drive frequency to the resonance frequency and to control the stroke.
　＜電圧位相算出器１３８＞　
　電圧位相算出器１３８は、例えば、逆正接演算器などを用いて、交流電圧指令位相（θＶｍ ＊）を出力する。電圧位相の定義は種々あるが、本実施例では、図１９に示すように、正弦波状に変化する可動子位置（図１９の水平軸）から９０度反時計方向に回転した垂直軸（図１９の垂直軸）を基準とした位相とする。　
　すなわち、無負荷状態では交流電圧指令位相（θＶｍ ＊）はゼロ近傍となり、重負荷状態では正値の値となる。 <Voltage phase calculator 138>
The voltage phase calculator 138 outputs an AC voltage command phase (θ Vm * ) using, for example, an inverse tangent calculator or the like. There are various definitions of the voltage phase, but in this embodiment, as shown in FIG. 19, the vertical axis rotated 90 degrees counterclockwise from the mover position (horizontal axis in FIG. 19) which changes sinusoidally (FIG. 19). Phase relative to the vertical axis of
That is, in the no-load state, the AC voltage command phase (θ Vm * ) is close to zero, and in the heavy load state, it has a positive value.
　［位相差指令値の与え方］　
　駆動周波数調整器１３１の構成を説明する際において、簡略化のために位相差指令値ｄｌｔθ＊を、例えば、０°としたが、本実施例は図１に示す通り、電圧指令値作成器１０３の出力である交流電圧指令位相（θＶｍ ＊）を位相差指令値ｄｌｔθ＊として、駆動周波数調整器１３１に入力する。　
　仮に、電圧指令値作成器１０３で電圧降下分に相当する電圧指令値（Ｖｍ２ ＊）に相当する電圧指令値をベクトル加算したのに関わらず、位相差指令値ｄｌｔθ＊を０°に固定した場合、減駆動周波数調整器１３１を構成する減算器９１（図１１）で求める位相差指令値ｄｌｔθ＊と位相差検出器１３０で求めた位相差ｄｌｔθ＾の差は、負値となり、駆動周波数調整器１３１は、駆動周波数指令値ω＊を増加させる制御を実行する。その結果、可動子６の質量とバネ定数から決まる機械的な共振周波数よりも高い周波数で収束してしまう。 [How to give phase difference command value]
When describing the configuration of the drive frequency adjuster 131, the phase difference command value dltθ * is set to, for example, 0 ° for simplification, but in the present embodiment, as shown in FIG. The AC voltage command phase (θV m * ), which is an output of the above, is input to the drive frequency adjuster 131 as a phase difference command value dltθ * .
If, despite the voltage command value corresponding to the voltage command value corresponding to the voltage drop in the voltage command value generator 103 (V m2 *) to the vector addition, and the phase difference command value Dltshita * fixed at 0 ° In this case, the difference between the phase difference command value dltθ * determined by the subtractor 91 (FIG. 11) constituting the reduced drive frequency adjuster 131 and the phase difference dltθ ^ determined by the phase difference detector 130 becomes a negative value, and the drive frequency adjustment The controller 131 executes control to increase the drive frequency command value ω *. As a result, convergence occurs at a frequency higher than the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant.
　一方、本実施例の通り、電圧指令値作成器１０３の出力である交流電圧指令位相（θＶｍ ＊）の負値を位相差指令値ｄｌｔθ＊として、駆動周波数調整器１３１に入力した場合、ベクトル加算の結果、進んだ電圧位相を考慮するため、駆動周波数調整器１３１は、駆動周波数指令値ω＊を増加させる制御は実行されず、可動子６の質量とバネ定数から決まる機械的な共振周波数で収束する。 On the other hand, as in the present embodiment, when the negative value of the AC voltage command phase (θV m * ) which is the output of voltage command value generator 103 is input to drive frequency adjuster 131 as phase difference command value dltθ * As a result of the addition, in order to take account of the advanced voltage phase, the drive frequency adjuster 131 does not execute control to increase the drive frequency command value ω * , and a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant. Converge at.
　これにより、負荷の変化（負荷電流）に応じて電圧指令値Ｖｍ ＊の振幅が増加すると共に、交流電圧指令位相（θＶｍ ＊）の分、位相が進んでも、駆動周波数が共振周波数よりも高く制御されること無く、幅広い負荷条件においても高効率で駆動することが可能になる。　
　なお、図１８では、電圧位相算出器１３８の出力に、乗算器９２で－１を乗じているが、電圧位相の定義によっては不要である。 Thereby, the amplitude of voltage command value V m * increases according to a change in load (load current), and the drive frequency is higher than the resonance frequency even if the phase advances by the AC voltage command phase (θ V m * ). It is possible to drive at high efficiency even in a wide range of load conditions without being highly controlled.
Although the output of the voltage phase calculator 138 is multiplied by -1 by the multiplier 92 in FIG. 18, this is not necessary depending on the definition of the voltage phase.
　［負荷への追従性］　
　図２０は、交流電圧指令位相と駆動周波数指令値の時間変化を示す説明図であり、時間的に負荷が変わっていった時の交流電圧指令位相（θＶｍ ＊）と駆動周波数指令値の変化を説明する図である。図２０の交流電圧指令位相（θＶｍ ＊）は、正弦波状に変化する可動子位置（図１９の水平軸）から９０度反時計方向に回転した垂直軸（図１９の垂直軸）を基準とした位相として示している。　
　このように、リニアモータ１０４が駆動中に、負荷がＬ１→Ｌ２→Ｌ３と変化（Ｌ２＞Ｌ１＞Ｌ３）するのに応じて、交流電圧指令位相と駆動周波数指令値がそれぞれθ１→θ２→θ３（θ２＞θ１＞θ３）とｆ１→ｆ２→ｆ３（ｆ２＞ｆ１＞ｆ３）のように変化する。 [Followability to load]
FIG. 20 is an explanatory view showing time change of AC voltage command phase and drive frequency command value, and change of AC voltage command phase (θV m * ) and drive frequency command value when load is temporally changed FIG. The AC voltage command phase (θ V m * ) in FIG. 20 is based on the vertical axis (vertical axis in FIG. 19) rotated 90 degrees counterclockwise from the mover position (horizontal axis in FIG. 19) that changes sinusoidally. It is shown as a phase that
Thus, while the linear motor 104 is driven, the AC voltage command phase and the drive frequency command value are each θ1 → θ2 → θ3 according to the load changing L1 → L2 → L3 (L2>L1> L3). (Θ2>θ1> θ3) and f1 → f2 → f3 (f2>f1> f3).
　＜ＰＷＭ信号作成器１３４＞　
　図１に示す制御部１０２を構成するＰＷＭ信号作成器１３４には、三角波のキャリア信号と電圧指令値Ｖｍ＊を比較することによる既知のパルス幅変調を用い、電圧指令値Ｖｍ＊に応じたドライブ信号が生成され、生成されたドライブ信号は電力変換回路１０５へ出力される。 <PWM signal generator 134>
The PWM signal generator 134 constituting the control unit 102 shown in FIG. 1 uses known pulse width modulation by comparing the carrier signal of the triangular wave and the voltage command value Vm *, and drives according to the voltage command value Vm * A signal is generated, and the generated drive signal is output to power conversion circuit 105.
　＜電力変換回路１０５＞　
　図２１は、図１に示すリニアモータ駆動装置１０１を構成する電力変換回路１０５の構成例を示す図である。フルブリッジ回路１２６は、制御部１０２により入力されたドライブ信号に応じて直流電圧源１２０をスイッチングして、リニアモータ１０４に電圧を出力する。フルブリッジ回路１２６は４つのスイッチング素子１２２を備えており、直列接続されたスイッチング素子１２２ａ，１２２ｂを持つ第一上下アーム（以下、Ｕ相と称する）と、スイッチング素子１２２ｃ，１２２ｄを持つ第二上下アーム（以下、Ｖ相と称する）と、を構成している。スイッチング素子１２２は、制御部１０２で生成される電圧指令値Ｖｍ＊やパルス幅変調によるドライブ信号を基に、ゲートドライバ回路１２３が出力するパルス状のゲート信号（１２４ａ～１２４ｄ）に応じてスイッチング動作できる。　
　スイッチング素子１２２の導通状態（オン／オフ）を制御することにより、直流電圧源１２０の直流電圧を交流電圧に相当する電圧を巻線８に出力できる。なお、直流電圧源１２０に代えて直流電流源を用いても良い。スイッチング素子１２２としては、例えば、ＩＧＢＴやＭＯＳ－ＦＥＴなどの半導体スイッチング素子を採用できる。 <Power conversion circuit 105>
FIG. 21 is a diagram showing a configuration example of the power conversion circuit 105 that constitutes the linear motor drive device 101 shown in FIG. The full bridge circuit 126 switches the DC voltage source 120 according to the drive signal input by the control unit 102, and outputs a voltage to the linear motor 104. The full bridge circuit 126 includes four switching elements 122, and includes first and second upper and lower arms (hereinafter referred to as U phase) having switching elements 122a and 122b connected in series, and second upper and lower arms having switching elements 122c and 122d. An arm (hereinafter, referred to as a V phase) is configured. Switching element 122 performs switching operation according to pulse-like gate signals (124 a to 124 d) output from gate driver circuit 123 based on voltage command value Vm * generated by control unit 102 and a drive signal by pulse width modulation. it can.
By controlling the conduction state (on / off) of switching element 122, a voltage corresponding to an AC voltage of DC voltage source 120 can be output to winding 8. A direct current source may be used instead of the direct current voltage source 120. As the switching element 122, for example, a semiconductor switching element such as an IGBT or a MOS-FET can be employed.
　[リニアモータ１０４との結線]　
　電力変換回路１０５の第一上下アームのスイッチング素子１２２ａ，１２２ｂ間および第二上下アームのスイッチング素子１２２ｃ，１２２ｄ間それぞれが、リニアモータ１０４に接続されている。図２１では、上側及び下側の電機子９の巻線８が並列に接続されている例を示しているが、巻線８を直列に接続することもできる。 [Wire connection to linear motor 104]
The linear motor 104 is connected between the switching elements 122 a and 122 b of the first upper and lower arms and between the switching elements 122 c and 122 d of the second upper and lower arms of the power conversion circuit 105. Although FIG. 21 shows an example in which the windings 8 of the upper and lower armatures 9 are connected in parallel, the windings 8 may be connected in series.
　[電流検出器１０７]　
　Ｕ相下アームとＶ相下アームには、例えばＣＴ（カレントトランス）等の電流検出器１０７を設けることができる。これにより、リニアモータ１０４の巻線８に流れる電流Ｉｍを検出できる。　
　電流検出器１０７として、例えば、ＣＴに代えて、電力変換回路１０５の下アームにシャント抵抗１２５を付加し、シャント抵抗１２５に流れる電流からリニアモータ１０４に流れる電流を検出する相シャント電流方式を採用できる。電流検出器１０７に代えて又は追加して、電力変換回路１０５の直流側に付加されたシャント抵抗１２５に流れる直流電流から、電力変換回路１０５の交流側の電流を検出するシングルシャント電流検出方式を採用しても良い。シングルシャント電流検出方式は、電力変換回路１０５を構成するスイッチング素子１２２の通電状態によって、シャント抵抗１２５に流れる電流が時間的に変化することを利用している。 [Current detector 107]
The U-phase lower arm and the V-phase lower arm may be provided with a current detector 107 such as a CT (current transformer). Thus, the current Im flowing through the winding 8 of the linear motor 104 can be detected.
As the current detector 107, for example, in place of the CT, a shunt resistor 125 is added to the lower arm of the power conversion circuit 105, and a phase shunt current method is employed to detect the current flowing to the linear motor 104 from the current flowing to the shunt resistor 125 it can. Instead of or in addition to the current detector 107, a single shunt current detection method for detecting the current on the alternating current side of the power conversion circuit 105 from the direct current flowing in the shunt resistor 125 added to the direct current side of the power conversion circuit 105 It may be adopted. The single shunt current detection method utilizes that the current flowing in the shunt resistor 125 temporally changes depending on the conduction state of the switching element 122 that constitutes the power conversion circuit 105.
　以上のように、本実施例によれば、負荷の変動に応じて変化する誘起電圧を検出するセンサを要することなく、負荷を含めた機械的な共振周波数でリニアモータを高効率に駆動し得るリニアモータシステムを提供することが可能となる。具体的には、負荷に応じて電圧振幅と電圧位相を制御し、電圧位相を考慮して駆動周波数を調整することにより、負荷を含めた機械的な共振周波数でリニアモータを駆動でき、高効率なリニアモータシステムを構成することができる。 As described above, according to the present embodiment, the linear motor can be driven with high efficiency at a mechanical resonance frequency including the load without requiring a sensor that detects an induced voltage that changes according to a change in the load. It becomes possible to provide a linear motor system. Specifically, by controlling the voltage amplitude and voltage phase according to the load, and adjusting the drive frequency in consideration of the voltage phase, the linear motor can be driven at a mechanical resonance frequency including the load, and the efficiency is high. The linear motor system can be configured.
　本実施例の構成は、下記の点を除き実施例１と同様にできる。本実施例は、後述するリニアモータシステム２００を搭載した機器の一例としての密閉型圧縮機５０に関する。 The configuration of this embodiment can be the same as that of Embodiment 1 except for the following points. The present embodiment relates to a hermetic compressor 50 as an example of a device equipped with a linear motor system 200 described later.
　＜密閉型圧縮機５０＞　
　図２２は、本発明の他の実施例に係る実施例２の密閉型圧縮機の縦断面図であり、リニアモータ１０４を有する密閉型圧縮機５０の縦断面図の一例である。密閉型圧縮機５０は、圧縮要素２０と電動要素３０とが密閉容器３内に配置されたレシプロ圧縮機である。圧縮要素２０及び電動要素３０は支持ばね４９によって密閉容器３内に弾性的に支持されている。電動要素３０は、可動子６及び電機子９を含む。　
　圧縮要素２０、はシリンダ１ａを形成するシリンダブロック１と、シリンダブロック１の端面に組み立てられるシリンダヘッド１６と、吐出室空間を形成するヘッドカバー１７とを備えている。シリンダ１ａ内に供給された作動流体はピストン４の往復動によって圧縮され、圧縮された作動流体は圧縮機外部に連通する吐出管（図示せず）へと送られる。
なお、作動流体は、例えば、空気や冷凍サイクルの冷媒などを採用できる。 <Sealed compressor 50>
FIG. 22 is a longitudinal sectional view of a hermetic compressor according to a second embodiment of the present invention, and is an example of a longitudinal sectional view of the hermetic compressor 50 having the linear motor 104. The hermetic compressor 50 is a reciprocating compressor in which the compression element 20 and the electric element 30 are disposed in the hermetic container 3. The compression element 20 and the motor element 30 are elastically supported in the closed container 3 by a support spring 49. The motor element 30 includes the mover 6 and the armature 9.
The compression element 20 comprises a cylinder block 1 forming a cylinder 1 a, a cylinder head 16 assembled on the end face of the cylinder block 1, and a head cover 17 forming a discharge chamber space. The working fluid supplied into the cylinder 1a is compressed by the reciprocating motion of the piston 4, and the compressed working fluid is sent to a discharge pipe (not shown) communicating with the outside of the compressor.
As the working fluid, for example, air or a refrigerant of a refrigeration cycle can be adopted.
　可動子６の一端にはピストン４が取り付けられている。本実施例では、可動子６及びピストン４が往復運動することで、作動流体を圧縮及び膨張させる。この圧縮及び膨張に要する仕事等が変動する負荷に相当する。電動要素３０の片端には圧縮要素２０を配置してある。シリンダブロック１は、可動子６の往復運動を案内するガイドロッドを前後方向に沿って有している。　
　可動子６に共振バネ２３（図２２中では図示せず）を付加し、可動子６の質量とバネ定数から決まる機械的な共振周波数で可動子６を往復運動させる場合、圧縮要素２０による共振周波数への影響も考慮する必要がある。すなわち、圧縮要素２０の吸込圧力や吐出空間の圧力によって、作動流体のバネ的な作用が加わるため、共振状態となる周波数が変化する。すなわち、シリンダ１ａの圧力が高い場合には、可動子６に付加された共振バネ２３のバネ定数が高いのと等価であり、共振周波数は高くなる。反対に、シリンダ１ａの圧力が低い場合には、可動子６に付加された共振バネ２３のバネ定数が支配的となり、共振周波数は、可動子６の質量とバネ定数から決まる機械的な共振周波数に近い。 A piston 4 is attached to one end of the mover 6. In the present embodiment, the working fluid is compressed and expanded by reciprocating the mover 6 and the piston 4. The work and the like required for the compression and expansion correspond to the fluctuating load. A compression element 20 is disposed at one end of the motorized element 30. The cylinder block 1 has a guide rod for guiding the reciprocating motion of the mover 6 along the longitudinal direction.
When a resonance spring 23 (not shown in FIG. 22) is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, the resonance by the compression element 20 The effect on frequency also needs to be considered. That is, since the spring-like action of the working fluid is added by the suction pressure of the compression element 20 and the pressure of the discharge space, the frequency at which the resonance state is obtained changes. That is, when the pressure of the cylinder 1a is high, it is equivalent to the spring constant of the resonance spring 23 added to the mover 6 being high, and the resonance frequency becomes high. On the other hand, when the pressure in the cylinder 1a is low, the spring constant of the resonant spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant. Close to
　このように、リニアモータ１０４を圧縮要素２０の動力とする場合は、圧縮要素２０の条件（吸込圧力、吐出圧力、吸込と吐出の圧力差等）によって共振周波数が変化してしまう。そのため、負荷や共振周波数の変化に合わせて駆動周波数を変化させることが必要である。 As described above, when the linear motor 104 is used as the power of the compression element 20, the resonance frequency changes depending on the conditions of the compression element 20 (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency in accordance with changes in load and resonance frequency.
　共振周波数が変化してしまう影響は、巻線８への印加電圧Ｖｍ、モータ電流Ｉｍ、及び可動子６の速度の位相関係から見て取れる。そのため、これらの位相関係を基に制御することで、高効率なリニアモータシステムを構成することができる。 The influence that the resonance frequency changes can be taken from the phase relationship of the applied voltage Vm to the winding 8, the motor current Im, and the speed of the mover 6. Therefore, by controlling based on these phase relationships, a highly efficient linear motor system can be configured.
　図２３は、本実施例のリニアモータシステム２００の全体概略構成図である。図２３に示すように、リニアモータシステム２００は、上述の実施例に１におけるリニアモータシステム１００と同様の構成であるが、負荷電流検出器１３６ａ、電圧降下成分を作成する電圧降下成分作成器１３７ａ、及び電圧指令値Ｖｍ＊を出力する電圧指令値作成器１０３ａが異なる。以下では、これら、負荷電流検出器１３６ａ、電圧降下成分作成器１３７ａ、及び電圧指令値作成器１０３ａについて説明する。その他の構成は上述の実施例１におけるリニアモータシステム１００と同様であるため説明を省略する。 FIG. 23 is an overall schematic configuration diagram of a linear motor system 200 of the present embodiment. As shown in FIG. 23, the linear motor system 200 has the same configuration as the linear motor system 100 in 1 in the above-described embodiment, but a load current detector 136a and a voltage drop component creator 137a that creates a voltage drop component. And the voltage command value generator 103 a that outputs the voltage command value Vm * is different. The load current detector 136a, the voltage drop component generator 137a, and the voltage command value generator 103a will be described below. The other configuration is the same as that of the linear motor system 100 according to the first embodiment described above, and therefore the description thereof is omitted.
　＜負荷電流検出器１３６ａ＞　
　図２４は、図２３に示すリニアモータシステム２００の制御部２０２を構成する負荷電流検出器１３６ａの構成例を示す説明図である。図２４に示すように、負荷電流検出器１５０ａは、電流検出器１０７よりモータ電流Ｉｍを入力すると共に位相指令値である基準位相θ＊を入力し、モータ電流Ｉｍの基本周波数のｃｏｓ成分（Ｉｍ＿ｃｏｓ）及びｓｉｎ成分（Ｉｍ＿ｓｉｎ）をそれぞれ抽出して出力する。　
　入力値の正弦を出力する正弦演算器８１と、入力値の余弦を出力する余弦演算器８２のそれぞれに、位相指令値である基準位相θ＊を入力し、基準位相θ＊に対する正弦及び余弦を得る。正弦及び余弦それぞれをモータ電流Ｉｍと乗算した値が乗算器９２からそれぞれ出力される。その出力を一次遅れフィルタ１４１でローパスフィルタ（低域通過フィルタ）処理し、正弦及び余弦それぞれの１次のフーリエ係数を得て、モータ電流Ｉｍの基本周波数のｓｉｎ成分（Ｉｍ＿ｓｉｎ）として負荷電流Ｉｍ＿ｓｉｎ、及びモータ電流Ｉｍの基本周波数のｃｏｓ成分（Ｉｍ＿ｃｏｓ）として負荷電流Ｉｍ＿ｃｏｓを、それぞれ電圧降下成分作成器１３７ａへ出力する。すなわち、フーリエ展開の駆動周波数ωより高次の周波数成分を消去できるので、高次のノイズに対してロバストに構成できる。図２４の構成とした場合、後述するベクトル和の演算を容易化でき、制御部２０２への実装に望ましい。 <Load current detector 136a>
FIG. 24 is an explanatory view showing a configuration example of the load current detector 136a that constitutes the control unit 202 of the linear motor system 200 shown in FIG. As shown in FIG. 24, the load current detector 150a receives the motor current Im from the current detector 107 and the reference phase θ * which is a phase command value, and the cos component of the fundamental frequency of the motor current Im (Im_cos And sin components (Im_sin) are respectively extracted and output.
A sine calculator 81 which outputs the sine of the input value, each of the cosine calculator 82 which outputs the cosine of the input value, receives a reference phase theta * a phase command value, the sine and cosine with respect to the reference phase theta * obtain. A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92, respectively. The output is low-pass filtered (low-pass filter) with a first-order lag filter 141 to obtain first-order Fourier coefficients of sine and cosine respectively, and load current Im_sin as a sin component (Im_sin) of the fundamental frequency of motor current Im. The load current Im_cos is output as the cos component (Im_cos) of the fundamental frequency of the motor current Im to the voltage drop component creator 137a. That is, since the frequency components higher than the drive frequency ω of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise. In the case of the configuration of FIG. 24, calculation of vector sums described later can be facilitated, which is desirable for mounting on the control unit 202.
　＜電圧降下成分作成器１３７ａ＞
　図２５は、図２３に示すリニアモータシステム２００の制御部２０２を構成する電圧降下成分作成器１３７ａの構成例を示す説明図である。図２５に示すように、電圧降下成分作成器１３７ａは、負荷電流検出器１３６ａで検出した負荷電流Ｉｍ＿ｓｉｎ及び負荷電流Ｉｍ＿ｃｏｓを入力し、リニアモータ１０４の抵抗及びインダクタンスによる電圧降下分に相当する電圧指令値（Ｖｍ２ ＊Ｉｍ＿ｓｉｎ）及び電圧降下分に相当する電圧指令値（Ｖｍ２ ＊Ｉｍ＿ｃｏｓ）を出力する。 <Voltage drop component creator 137a>
FIG. 25 is an explanatory view showing a configuration example of the voltage drop component creator 137a that configures the control unit 202 of the linear motor system 200 shown in FIG. As shown in FIG. 25, the voltage drop component generator 137a receives the load current Im_sin and the load current Im_cos detected by the load current detector 136a, and a voltage command corresponding to the voltage drop due to the resistance and inductance of the linear motor 104. A voltage command value (V m2 * Im_cos) corresponding to the value (V m2 * Im_sin) and the voltage drop is output.
　電圧降下成分作成器１３７ａは、負荷電流Ｉｍ＿ｃｏｓに、予め設定あるいは推定したリニアモータ１０４の抵抗値Ｒｍ＊を乗算器９２にて乗じ、電圧指令値（Ｖｍ２ ＊＿ｃｏｓ）として電圧指令値作成器１０３ａへ出力する。また、電圧降下成分作成器１３７ａは、負荷電流Ｉｍ＿ｓｉｎに、予め設定あるいは推定したリニアモータ１０４のインダクタンス値Ｌｍ＊を乗算器９２にて乗じ、電圧指令値（Ｖｍ２ ＊＿ｓｉｎ）として電圧指令値作成器１０３ａへ出力する。 The voltage drop component creator 137a multiplies the load current Im_cos by the resistance value Rm * of the linear motor 104, which has been set or estimated in advance, by the multiplier 92, and sets the voltage command value creator 103a as a voltage command value (V m2 * _cos). Output to Further, the voltage drop component creator 137a multiplies the load current Im_sin by the inductance value Lm * of the linear motor 104 preset or estimated by the multiplier 92 to create a voltage command value as a voltage command value (V m2 * _sin). Output to the controller 103a.
　＜電圧指令値作成器１０３ａ＞
  図２６は、図２３に示すリニアモータシステム２００の制御部２０２を構成する電圧指令値作成器１０３ａの構成例を示す説明図である。図２６に示すように、電圧指令値作成器１０３ａは、誘起電圧成分作成器１３５より出力される可動子６の速度に応じて生じる誘起電圧に相当する電圧指令値Ｖｍ１ ＊と、電圧降下成分作成器１３７ａより出力される電圧指令値（Ｖｍ２ ＊＿ｃｏｓ）とを加算器９０にて加算して出力する。このようにｃｏｓ成分とｓｉｎ成分に分けて加算することで、電圧指令値作成器１０３ａではベクトル和（ベクトル加算）を出力しているのと等価である。 <Voltage command value generator 103a>
FIG. 26 is an explanatory view showing a configuration example of a voltage command value creation unit 103a that configures the control unit 202 of the linear motor system 200 shown in FIG. As shown in FIG. 26, the voltage command value generator 103a generates a voltage command value V m1 * corresponding to the induced voltage generated according to the speed of the mover 6 output from the induced voltage component generator 135, and a voltage drop component The voltage command value (V m2 * _ cos) output from the generator 137 a is added by the adder 90 and output. As described above, adding the cos component and the sin component separately is equivalent to outputting the vector sum (vector addition) in the voltage command value generator 103 a.
　電圧位相の定義は種々あるが、本実施例では、上述の実施例１において図１９に示したように、正弦波状に変化する可動子位置（図１９の水平軸）から９０度反時計方向に回転した垂直軸（図１９の垂直軸）を基準とした位相とする。そのため、誘起電圧成分作成器１３５の出力値（可動子６の速度に応じて生じる誘起電圧に相当する電圧指令値Ｖｍ１ ＊）と、電圧降下成分作成器１３７ａの出力である電圧指令値（Ｖｍ２ ＊＿ｃｏｓ）とを加算器９０で加算する。 Although there are various definitions of the voltage phase, in this embodiment, as shown in FIG. 19 in the above-mentioned first embodiment, in the counterclockwise direction 90 degrees from the mover position (horizontal axis in FIG. 19) which changes sinusoidally. The phase is based on the rotated vertical axis (vertical axis in FIG. 19). Therefore, the output value of induced voltage component generator 135 (voltage command value V m1 * corresponding to the induced voltage generated according to the speed of mover 6) and the voltage command value (V) output of voltage drop component generator 137a The adder 90 adds m2 * _cos).
　電圧指令値作成器１０３ａは、電圧位相算出器１３８及び平方根演算器９６を有する。
平方根演算器９６は、加算器９０による加算結果及び電圧降下成分作成器１３７ａから出力される電圧指令値（Ｖｍ２ ＊＿ｓｉｎ）の平方根を得て、交流電圧指令値（Ｖｍ＊）をＰＷＭ信号作成器１３４へ出力する。また、電圧位相算出器１３８は、加算器９０による加算結果及び電圧降下成分作成器１３７ａから出力される電圧指令値（Ｖｍ２ ＊＿ｓｉｎ）を入力し、例えば、逆正接演算器などを用いて交流電圧指令位相（θＶｍ ＊）を駆動周波数調整器１３１へ出力する。 The voltage command value generator 103 a has a voltage phase calculator 138 and a square root calculator 96.
The square root calculator 96 obtains the square root of the voltage command value (V m2 * _sin) output from the addition result by the adder 90 and the voltage drop component generator 137 a, and generates an AC voltage command value (V m * ) as a PWM signal. Output to the output unit 134. The voltage phase calculator 138 also receives the addition result of the adder 90 and the voltage command value (V m2 * _sin) output from the voltage drop component creator 137 a, and uses, for example, an AC tangent The voltage command phase (θV m * ) is output to the drive frequency adjuster 131.
　以上のように、本実施例によれば、密閉型圧縮機５０のリニアモータ１０４において、負荷に応じて電圧振幅と電圧位相を制御し、電圧位相を考慮して駆動周波数を調整することにより、負荷を含めた機械的な共振周波数でリニアモータを駆動でき、高効率なリニアモータシステムを構成することができる。 As described above, according to the present embodiment, in the linear motor 104 of the hermetic compressor 50, the voltage amplitude and the voltage phase are controlled according to the load, and the drive frequency is adjusted in consideration of the voltage phase. A linear motor can be driven at a mechanical resonance frequency including a load, and a highly efficient linear motor system can be configured.
　本実施例の構成は、下記の点を除き実施例１又は２と同様にできる。本実施例は、リニアモータシステム搭載した機器の一例としてのエアサスペンションシステム３００に関する。　
　図２７は本発明の他の実施例に係る実施例３のエアサスペンションシステム３００の回路図であり、図２８は図２７に示すエアサスペンションシステム３００を搭載した車両の概略図である。但し、図２８においては、後述する分配点３０９Ｎ及びこれよりエアサスペンション３０１，３０２側の構成要素のみを図示している。 The configuration of this embodiment can be the same as that of Embodiment 1 or 2 except for the following points. The present embodiment relates to an air suspension system 300 as an example of a device mounted with a linear motor system.
FIG. 27 is a circuit diagram of an air suspension system 300 according to a third embodiment of the present invention, and FIG. 28 is a schematic view of a vehicle equipped with the air suspension system 300 shown in FIG. However, in FIG. 28, only the distribution point 309N to be described later and the components on the air suspension 301, 302 side therefrom are illustrated.
　図２７に示すように、エアサスペンションシステム３００は、２つのエアサスペンション３０１，３０２、リニアモータ１０４を駆動源とするコンプレッサ３０３、吸気フィルタ３０４、第１タンク３０５、及びエアドライヤ３０７、並びに、弁として、３つのチェック弁３０８，３１５，３１７、給排切換弁３１０、２つのサスペンション制御弁３１１，３１２、戻り通路開閉弁３１４、及び排気通路開閉弁３１９、を有している。エアサスペンションシステム３００は、空気が流通可能な通路によってこれらを接続している。 As shown in FIG. 27, the air suspension system 300 includes two air suspensions 301 and 302, a compressor 303 driven by the linear motor 104, an intake filter 304, a first tank 305, an air dryer 307, and valves. The three check valves 308, 315, 317, the supply / discharge switching valve 310, the two suspension control valves 311, 312, the return passage on / off valve 314, and the exhaust passage on / off valve 319 are provided. The air suspension system 300 connects these by a passage through which air can flow.
　エアサスペンションシステム３００は、図２８に示すように、例えば車両４００に搭載され、エアサスペンション３０１，３０２のエア室３０１Ｃ，３０２Ｃ（図２７）内の空気圧の制御を行うシステムである。例えば、車両４００の左車輪４１０Ｌ及び右車輪４１０Ｒには、これらのハブ等同士を繋ぐ車軸４２０が設けられている。例えば、左車輪４１０Ｌ及び右車輪４１０Ｒそれぞれと車体４３０との間や、ハブと車体４３０との間といった、車輪４１０側と車体４３０側との間にエアサスペンション３０１、３０２を設け、エア室３０１Ｃ，３０２Ｃ内の空気圧を制御することで、車高の調整を行える。 The air suspension system 300 is, for example, a system which is mounted on a vehicle 400 and controls air pressure in air chambers 301C and 302C (FIG. 27) of the air suspensions 301 and 302, as shown in FIG. For example, on the left wheel 410L and the right wheel 410R of the vehicle 400, an axle 420 connecting these hubs and the like is provided. For example, air suspensions 301 and 302 are provided between the wheel 410 side and the vehicle body 430 side, such as between the left wheel 410L and the right wheel 410R and the vehicle body 430, or between the hub and the vehicle body 430, The vehicle height can be adjusted by controlling the air pressure in the 302C.
　エアサスペンション３０１，３０２は、図２８に示すように、車輪４１０側の車軸４２０と車両４００の車体４３０との間に取り付けられてもよく、また、車輪４１０と車体４３０とを連結するサスペンションのアーム類(車輪４１０側）と車体４３０との間や車輪４１０のハブ（車輪４１０側）とサスペンションのアッパーアームの車体４３０取付部近傍（車体４３０側）との間に取付けてもよい。このように、エアサスペンション３０１，３０２は、車輪４１０と車体４３０を支えるように設けられれば良く、例えば、上下方向について車輪４１０と車体４３０との間に設けることができ、直接、車輪４１０や車体４３０に取り付ける態様には限られない。 The air suspensions 301 and 302 may be attached between the axle 420 on the wheel 410 side and the vehicle body 430 of the vehicle 400 as shown in FIG. 28 and an arm of a suspension connecting the wheel 410 and the vehicle body 430 It may be attached between the class (wheel 410 side) and the vehicle body 430 or between the hub (wheel 410 side) of the wheel 410 and the vicinity of the vehicle body 430 mounting portion of the upper arm of the suspension (vehicle body 430 side). As described above, the air suspensions 301 and 302 may be provided to support the wheels 410 and the vehicle body 430. For example, the air suspensions 301 and 302 can be provided between the wheels 410 and the vehicle body 430 in the vertical direction. It is not restricted to the aspect attached to 430.
　本実施例では、エアサスペンションを２つ有するエアサスペンションシステム３００について説明するが、エアサスペンションシステム３００が含むエアサスペンションの個数は１つ以上であれば特に制限されない。エアサスペンションの個数は、例えば車輪の個数に等しくすることができる。例えば４輪自動車の場合には、２つの前輪側に２個、２つの後輪側に２個の、合計４個のエアサスペンションを配設できる。なお、本実施例では、緩衝用のシリンダ３０１Ａ，３０２Ａとエアばねとなるエア室３０１Ｃ，３０２Ｃとを一体にした例を示したが、大型車やリヤサスペンション側で既知のように緩衝用のシリンダ（油圧緩衝器）３０１Ａ，３０２Ａとエアばねとを独立に設けてもよい。 In the present embodiment, an air suspension system 300 having two air suspensions will be described, but the number of air suspensions included in the air suspension system 300 is not particularly limited as long as it is one or more. The number of air suspensions can, for example, be equal to the number of wheels. For example, in the case of a four-wheeled vehicle, a total of four air suspensions, two on the two front wheels and two on the two rear wheels, can be disposed. In the present embodiment, an example is shown in which the cylinders 301A and 302A for shock absorbing and the air chambers 301C and 302C serving as air springs are integrated, but the cylinder for shock absorbing as well known on the large vehicle and rear suspension side (Hydraulic shock absorber) 301A, 302A and an air spring may be provided independently.
　図２７に示すように、エアサスペンション３０１，３０２には、緩衝用のシリンダ３０１Ａ，３０２Ａそれぞれとピストンロッド３０１Ｂ，３０２Ｂそれぞれとの間にエア室３０１Ｃ，３０２Ｃが形成されており、エアばねを構成している。エア室３０１Ｃ，３０２Ｃそれぞれには後述する通路が接続されており、エアサスペンションシステム３００の動作によって圧力及び車高が制御されている。 As shown in FIG. 27, in the air suspensions 301 and 302, air chambers 301C and 302C are formed between the buffer cylinders 301A and 302A and the piston rods 301B and 302B, respectively, to form an air spring. ing. A passage described later is connected to each of the air chambers 301C and 302C, and the pressure and the vehicle height are controlled by the operation of the air suspension system 300.
　コンプレッサ３０３は、吸入ポート３０３Ｃから吸入した空気を圧縮して吐出ポート３０３Ｄから吐出することができる。コンプレッサ３０３は、コンプレッサ本体３０３Ａ及びリニアモータ１０４から構成される。吸入ポート３０３Ｃまたは吐出ポート３０３Ｄ、もしくは両ポートの圧力を測定する圧力センサを設けている。 The compressor 303 can compress the air sucked from the suction port 303C and discharge the compressed air from the discharge port 303D. The compressor 303 includes a compressor body 303A and a linear motor 104. A pressure sensor is provided to measure the pressure of the suction port 303C or the discharge port 303D or both ports.
　リニアモータ１０４の可動子６に共振バネ２３を付加し、可動子６の質量とバネ定数から決まる機械的な共振周波数で可動子６を往復動させる場合、上述の実施例２において図２２に示したように、圧縮要素２０による共振周波数への影響も考慮する必要がある。すなわち、圧縮要素２０の吸込圧力や吐出空間の圧力によって、作動流体のバネ的な作用が加わるため、共振状態となる周波数が変化する。つまり、シリンダ１ａの圧力が高い場合には、可動子６に付加された共振バネ２３のバネ定数が高いのと等価であり、共振周波数は高くなる。反対に、シリンダ１ａの圧力が低い場合には、可動子６に付加された共振バネ２３のバネ定数が支配的となり、共振周波数は、可動子６の質量とバネ定数から決まる機械的な共振周波数に近い。 In the case where the resonance spring 23 is added to the mover 6 of the linear motor 104 and the mover 6 is reciprocated at the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant, it is shown in FIG. As such, the influence of the compression element 20 on the resonant frequency also needs to be considered. That is, the spring-like action of the working fluid is exerted by the suction pressure of the compression element 20 and the pressure of the discharge space, so that the frequency at which the resonance state occurs is changed. That is, when the pressure of the cylinder 1a is high, it is equivalent to the spring constant of the resonance spring 23 added to the mover 6 being high, and the resonance frequency becomes high. On the other hand, when the pressure in the cylinder 1a is low, the spring constant of the resonant spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant. Close to
　このように、リニアモータ１０４を圧縮要素の動力とする場合は、圧縮要素の条件（吸込圧力、吐出圧力、吸込と吐出の圧力差等）によって共振周波数が変化してしまう。そのため、負荷や共振周波数の変化に合わせて駆動周波数を変化させることが必要である。 As described above, when the linear motor 104 is used as the power of the compression element, the resonance frequency changes depending on the conditions of the compression element (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency in accordance with changes in load and resonance frequency.
　＜負荷電流検出器１３６ｂ＞　
　図２９は、負荷電流検出器１３６ｂの構成例を示す図である。負荷電流検出器１５０ｂは、モータ電流Ｉｍを入力し、モータ電流Ｉｍの基本周波数のｃｏｓ成分（Ｉｍ＿ｃｏｓ）とｓｉｎ成分（Ｉｍ＿ｓｉｎ）をそれぞれ抽出して出力する。 <Load current detector 136b>
FIG. 29 is a diagram showing a configuration example of the load current detector 136b. The load current detector 150 b receives the motor current Im, extracts and outputs a cos component (Im_cos) and a sin component (Im_sin) of the fundamental frequency of the motor current Im.
　入力値の正弦を出力する正弦演算器８１と、入力値での余弦を出力する余弦演算器８２のそれぞれに、位相指令値である基準位相θ＊を入力し、基準位相θ＊に対する正弦及び余弦を得る。正弦及び余弦それぞれをモータ電流Ｉｍと乗算した値が乗算器９２からそれぞれ出力される。その出力を一次遅れフィルタ１４１ａでローパスフィルタ（低域通過フィルタ）処理し、正弦及び余弦それぞれの１次のフーリエ係数を得る。すなわち、フーリエ展開の駆動周波数ωより高次の周波数成分を消去できるので、高次のノイズに対してロバストに構成できる。一次遅れフィルタ１４１ａのフィルタ時定数（Ｔ＿ｌｄ）若しくは遮断周波数は、外部から変更できるようになっている。 A reference phase θ * , which is a phase command value, is input to each of a sine calculator 81 that outputs a sine of an input value and a cosine calculator 82 that outputs a cosine at the input value, and sine and cosine with respect to the reference phase θ * Get A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92, respectively. The output is low-pass filtered (low-pass filter) with a first-order lag filter 141a to obtain sine and cosine first-order Fourier coefficients. That is, since the frequency components higher than the drive frequency ω of the Fourier expansion can be eliminated, the configuration can be robust against high-order noise. The filter time constant (T_ld) or cutoff frequency of the first-order lag filter 141a can be changed from the outside.
　図２９に示す負荷電流検出器１３６ｂの構成では、一次遅れフィルタ１４１ａのフィルタ時定数（Ｔ＿ｌｄ）若しくは遮断周波数は、コンプレッサ３０３の吸入ポート３０３Ｃまたは吐出ポート３０３Ｄの圧力に応じて、変更できるようになっている。これにより、圧力の変化によっても負荷電流検出への影響を軽減できる。 In the configuration of the load current detector 136b shown in FIG. 29, the filter time constant (T_ld) or cutoff frequency of the first-order lag filter 141a can be changed according to the pressure of the suction port 303C or discharge port 303D of the compressor 303. ing. As a result, the change in pressure can also reduce the influence on load current detection.
　以上のように、本実施例によれば、エアサスペンションシステム３００において、負荷に応じて電圧振幅と電圧位相を制御し、電圧位相を考慮して駆動周波数を調整することにより、負荷を含めた機械的な共振周波数でリニアモータを駆動でき、高効率なリニアモータシステムを構成することができる。 As described above, according to the present embodiment, in the air suspension system 300, the machine including the load is controlled by controlling the voltage amplitude and the voltage phase according to the load and adjusting the drive frequency in consideration of the voltage phase. The linear motor can be driven at a typical resonance frequency, and a highly efficient linear motor system can be configured.
　本発明は上記した実施例に限定されるものではなく、様々な変形例が含まれる。例えば、上記した実施例は本発明を分かりやすく説明するために詳細に説明したものであり、必ずしも説明した全ての構成を備えるものに限定されるものではない。　
　また、上記の各構成、機能、処理部、処理手続き等は、それらの一部または全部を、例えば集積回路で設計する等によりハードウェアで実現しても良い。また、上記の各構成や機能等は、プロセッサがそれぞれの機能を実現するプログラムを解釈し、実行することによりソフトウェアで実現しても良い。 The present invention is not limited to the embodiments described above, but includes various modifications. For example, the embodiments described above are described in detail in order to explain the present invention in an easy-to-understand manner, and are not necessarily limited to those having all the configurations described.
Further, each of the configurations, functions, processing units, processing procedures, and the like described above may be realized by hardware, for example, by designing part or all of them with an integrated circuit. In addition, each configuration, function, and the like described above may be realized by software by a processor interpreting and executing a program that realizes each function.
　実施例１乃至実施例３において、リニアモータ１０４を単相機として説明したが、三相機であっても本発明の構成を適用することができ、同様の効果を得られる。　
　電力変換回路１０５は、電流を出力する態様であってもよい。この場合は、電圧指令値作成器１０３に代えて電流指令値作成器を設ければよい。 Although the linear motor 104 has been described as a single-phase machine in the first to third embodiments, the configuration of the present invention can be applied to a three-phase machine, and similar effects can be obtained.
The power conversion circuit 105 may output a current. In this case, a current command value creator may be provided instead of the voltage command value creator 103.
１…シリンダブロック、１ａ…シリンダ、２…永久磁石、３…密閉容器、４…ピストン、６…可動子、７…磁極、８…巻線、９…電機子、１６…シリンダヘッド、１７…ヘッドカバー、２０…圧縮要素、２３…共振バネ（アシストバネ）、３０…電動要素、５０…密閉型圧縮機、１００，２００…リニアモータシステム、１０１，２０１…リニアモータ駆動装置、１０２，２０２…制御部、１０３…電圧指令値作成器、１０４…リニアモータ、１０５…電力変換回路、１０７…電流検出器、１２６…フルブリッジ回路、１３０…位相差検出器、１３１…駆動周波数調整器、１３４…ＰＷＭ信号作成器、１３５…誘起電圧成分作成器、１３６…負荷電流検出器、１３７…電圧降下成分作成器、１４０…積分器 DESCRIPTION OF SYMBOLS 1 ... cylinder block, 1a ... cylinder, 2 ... permanent magnet, 3 ... closed container, 4 ... piston, 6 ... mover, 7 ... magnetic pole, 8 ... winding, 9 ... armature, 16 ... cylinder head, 17 ... head cover , 20: compression element, 23: resonance spring (assist spring), 30: electric element, 50: sealed compressor, 100, 200: linear motor system, 101, 201: linear motor drive, 102, 202: control unit , 103: voltage command value generator, 104: linear motor, 105: power conversion circuit, 107: current detector, 126: full bridge circuit, 130: phase difference detector, 131: driving frequency adjuster, 134: PWM signal Creator, 135: induced voltage component creator, 136: load current detector, 137: voltage drop component creator, 140: integrator

Claims (14)

  1. 　少なくとも交流電圧が印加される巻線及び弾性体が接続する可動子を有するリニアモータを備え、
    　前記弾性体と前記リニアモータの負荷によって可動子の共振周波数が変動するリニアモータシステムであって、
    　前記交流電圧の駆動周波数が共振周波数となるよう、前記交流電圧と前記巻線に流れる電流の位相差を調整するリニアモータ駆動装置を有することを特徴とするリニアモータシステム。     A linear motor having a mover to which a winding to which at least an alternating voltage is applied and an elastic body are connected;
    A linear motor system, in which a resonance frequency of a mover is varied by a load of the elastic body and the linear motor,
    A linear motor system comprising: a linear motor drive device for adjusting a phase difference between the alternating current voltage and the current flowing through the winding so that a drive frequency of the alternating current voltage becomes a resonance frequency.
  2. 　少なくとも交流電圧が印加される巻線及び弾性体が接続する可動子を有するリニアモータを備え、
    　前記弾性体と前記リニアモータの負荷によって可動子の共振周波数が変動するリニアモータシステムであって、
    　前記負荷の増加に応じて、前記交流電圧と前記巻線に流れる電流の位相差を大きくするリニアモータ駆動装置を有することを特徴とするリニアモータシステム。     A linear motor having a mover to which a winding to which at least an alternating voltage is applied and an elastic body are connected;
    A linear motor system, in which a resonance frequency of a mover is varied by a load of the elastic body and the linear motor,
    A linear motor system comprising: a linear motor drive device which increases the phase difference between the AC voltage and the current flowing through the winding according to the increase of the load.
  3. 　少なくとも交流電圧が印加される巻線及び弾性体が接続する可動子を有するリニアモータを備え、
    　前記弾性体と前記リニアモータの負荷によって可動子の共振周波数が変動するリニアモータシステムであって、
    　前記巻線に流れる電流の基本波振幅を検出し、前記基本波振幅の増加に応じて、前記交流電圧と前記巻線に流れる電流の位相差を大きくするリニアモータ駆動装置を有することを特徴とするリニアモータシステム。     A linear motor having a mover to which a winding to which at least an alternating voltage is applied and an elastic body are connected;
    A linear motor system, in which a resonance frequency of a mover is varied by a load of the elastic body and the linear motor,
    A linear motor drive device for detecting a fundamental wave amplitude of the current flowing through the winding, and increasing a phase difference between the alternating voltage and the current flowing through the winding according to an increase in the fundamental wave amplitude; Linear motor system.
  4. 　請求項３に記載のリニアモータシステムにおいて、
    　前記リニアモータ駆動装置は、
    　前記基本波振幅を用いて前記巻線の抵抗及びインダクタンスの電圧降下分に相当する第１の電圧指令値を出力する電圧降下成分作成器と、
    　前記可動子の速度に応じて生じる誘起電圧に相当する第２の電圧指令値を出力する誘起電圧成分作成器と、
    　前記第１の電圧指令値及び第２の電圧指令値をベクトル加算し第３の電圧指令値を出力する電圧指令値作成器と、を備え、
    　前記第３の電圧指令値に相当する電圧を前記巻線に印加することを特徴とするリニアモータシステム。     In the linear motor system according to claim 3,
    The linear motor drive device
    A voltage drop component creator which outputs a first voltage command value corresponding to a voltage drop of the resistance and inductance of the winding using the fundamental wave amplitude;
    An induced voltage component generator that outputs a second voltage command value corresponding to the induced voltage generated according to the speed of the mover;
    A voltage command value generator that adds a vector of the first voltage command value and the second voltage command value and outputs a third voltage command value;
    A linear motor system characterized in that a voltage corresponding to the third voltage command value is applied to the winding.
  5. 　請求項３に記載のリニアモータシステムにおいて、
    　前記リニアモータ駆動装置は、
    　前記巻線に流れる基本波電流の可動子位置に直交する成分である基本波余弦成分を検出し、前記基本波余弦成分の大きさを用いて前記巻線の抵抗及びインダクタンスの電圧降下分に相当する第１の電圧指令値を出力する電圧降下成分作成器と、
    　前記可動子の速度に応じて生じる誘起電圧に相当する第２の電圧指令値を出力する誘起電圧成分作成器と、
    　前記第１の電圧指令値及び第２の電圧指令値をベクトル加算し第３の電圧指令値を出力する電圧指令値作成器と、を備え、
    　前記第３の電圧指令値に相当する電圧を前記巻線に印加することを特徴とするリニアモータシステム。     In the linear motor system according to claim 3,
    The linear motor drive device
    A fundamental wave cosine component which is a component orthogonal to the mover position of the fundamental wave current flowing through the winding is detected, and the magnitude of the fundamental wave cosine component is used to correspond to the voltage drop of the resistance and inductance of the winding. A voltage drop component generator for outputting a first voltage command value
    An induced voltage component generator that outputs a second voltage command value corresponding to the induced voltage generated according to the speed of the mover;
    A voltage command value generator that adds a vector of the first voltage command value and the second voltage command value and outputs a third voltage command value;
    A linear motor system characterized in that a voltage corresponding to the third voltage command value is applied to the winding.
  6. 　請求項４に記載のリニアモータシステムにおいて、
    　前記リニアモータ駆動装置は、
    前記基本波振幅又は基本波余弦成分の大きさの低域通過フィルタ処理後の値である負荷電流を出力する負荷電流検出器を備え、
    　前記電圧降下成分作成器は、前記負荷電流に基づき前記第１の電圧指令値を出力することを特徴とするリニアモータシステム。     In the linear motor system according to claim 4,
    The linear motor drive device
    A load current detector for outputting a load current which is a low-pass filtered value of the magnitude of the fundamental wave amplitude or the magnitude of the fundamental cosine component;
    The linear motor system, wherein the voltage drop component creator outputs the first voltage command value based on the load current.
  7. 　請求項５に記載のリニアモータシステムにおいて、
    　前記リニアモータ駆動装置は、
    前記基本波振幅又は基本波余弦成分の大きさの低域通過フィルタ処理後の値である負荷電流を出力する負荷電流検出器を備え、
    　前記電圧降下成分作成器は、前記負荷電流に基づき前記第１の電圧指令値を出力することを特徴とするリニアモータシステム。     In the linear motor system according to claim 5,
    The linear motor drive device
    A load current detector for outputting a load current which is a low-pass filtered value of the magnitude of the fundamental wave amplitude or the magnitude of the fundamental cosine component;
    The linear motor system, wherein the voltage drop component creator outputs the first voltage command value based on the load current.
  8. 　請求項３に記載のリニアモータシステムにおいて、
    　前記リニアモータ駆動装置は、
    　前記基本波振幅を用いて前記巻線の抵抗及びインダクタンスの電圧降下分に相当する第１の電圧指令値の正弦成分及び前記第１の電圧指令値の余弦成分を出力する電圧降下成分作成器と、
    　前記可動子の速度に応じて生じる誘起電圧に相当する第２の電圧指令値を出力する誘起電圧成分作成器と、
    　前記第１の電圧指令値の余弦成分及び第２の電圧指令値をベクトル加算し第３の電圧指令値を出力すると共に、前記第１の電圧指令値の正弦成分に基づき電圧指令位相を出力する電圧指令値作成器と、を備え、
    　前記第３の電圧指令値及び前記電圧指令位相に相当する電圧を前記巻線に印加することを特徴とするリニアモータシステム。     In the linear motor system according to claim 3,
    The linear motor drive device
    A voltage drop component generator for outputting a sine component of a first voltage command value corresponding to a voltage drop of the resistance and inductance of the winding using the fundamental wave amplitude and a cosine component of the first voltage command value; ,
    An induced voltage component generator that outputs a second voltage command value corresponding to the induced voltage generated according to the speed of the mover;
    The vector addition of the cosine component of the first voltage command value and the second voltage command value is performed to output a third voltage command value, and the voltage command phase is output based on the sine component of the first voltage command value. And a voltage command value generator,
    A linear motor system characterized in that a voltage corresponding to the third voltage command value and the voltage command phase is applied to the winding.
  9. 　請求項８に記載のリニアモータシステムにおいて、
    　前記リニアモータ駆動装置は、
    　前記基本波振幅又は基本波余弦成分の大きさの低域通過フィルタ処理後の値である負荷電流の正弦成分及び前記負荷電流の余弦成分を出力する負荷電流検出器を備え、
    　前記電圧降下成分作成器は、前記負荷電流の正弦成分及び前記負荷電流の余弦成分に基づき前記第１の電圧指令値の正弦成分及び前記第１の電圧指令値の余弦成分を出力することを特徴とするリニアモータシステム。     In the linear motor system according to claim 8,
    The linear motor drive device
    A load current detector for outputting a sine component of a load current which is a value after low-pass filtering of the amplitude of the fundamental wave or the magnitude of the cosine component of the fundamental wave and a cosine component of the load current;
    The voltage drop component creator may output a sine component of the first voltage command value and a cosine component of the first voltage command value based on a sine component of the load current and a cosine component of the load current. Linear motor system.
  10. 　請求項９に記載のリニアモータシステムにおいて、
    　前記可動子にピストンが接続され、前記ピストンが負荷として流体又は空気を圧縮する圧縮機であることを特徴とするリニアモータシステム。     In the linear motor system according to claim 9,
    A linear motor system characterized in that a piston is connected to the mover, and the piston is a compressor that compresses fluid or air as a load.
  11. 　請求項１０に記載のリニアモータシステムにおいて、
    　前記低域通過フィルタ処理の遮断周波数を前記圧縮機の吸入または吐出圧力に応じて変更することを特徴とするリニアモータシステム。     The linear motor system according to claim 10,
    A linear motor system, wherein the cutoff frequency of the low pass filtering is changed according to the suction or discharge pressure of the compressor.
  12. 　請求項４に記載のリニアモータシステムにおいて、
    　車体側と車輪側との間に介装され空気の給排に応じて車高調整を行う複数のエア室に、　前記可動子にピストンが接続され、前記ピストンが負荷として流体又は空気を圧縮する圧縮機にて圧縮した空気を供給するエアサスペンションシステムであることを特徴とするリニアモータシステム。     In the linear motor system according to claim 4,
    A piston is connected to the movable element to a plurality of air chambers interposed between the vehicle body side and the wheel side to adjust the vehicle height according to the supply and discharge of air, and the piston compresses fluid or air as a load A linear motor system characterized in that it is an air suspension system for supplying air compressed by a compressor.
  13. 　請求項５に記載のリニアモータシステムにおいて、
    　車体側と車輪側との間に介装され空気の給排に応じて車高調整を行う複数のエア室に、　前記可動子にピストンが接続され、前記ピストンが負荷として流体又は空気を圧縮する圧縮機にて圧縮した空気を供給するエアサスペンションシステムであることを特徴とするリニアモータシステム。     In the linear motor system according to claim 5,
    A piston is connected to the movable element to a plurality of air chambers interposed between the vehicle body side and the wheel side to adjust the vehicle height according to the supply and discharge of air, and the piston compresses fluid or air as a load A linear motor system characterized in that it is an air suspension system for supplying air compressed by a compressor.
  14. 　請求項８に記載のリニアモータシステムにおいて、
    　車体側と車輪側との間に介装され空気の給排に応じて車高調整を行う複数のエア室に、　前記可動子にピストンが接続され、前記ピストンが負荷として流体又は空気を圧縮する圧縮機にて圧縮した空気を供給するエアサスペンションシステムであることを特徴とするリニアモータシステム。     In the linear motor system according to claim 8,
    A piston is connected to the movable element to a plurality of air chambers interposed between the vehicle body side and the wheel side to adjust the vehicle height according to the supply and discharge of air, and the piston compresses fluid or air as a load A linear motor system characterized in that it is an air suspension system for supplying air compressed by a compressor.
PCT/JP2018/018667 2017-07-06 2018-05-15 Linear motor system WO2019008907A1 (en)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
JP2017132634A JP6965048B2 (en) 2017-07-06 2017-07-06 Linear motor system
JP2017-132634 2017-07-06

Publications (1)

Publication Number Publication Date
WO2019008907A1 true WO2019008907A1 (en) 2019-01-10

Family

ID=64949923

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/JP2018/018667 WO2019008907A1 (en) 2017-07-06 2018-05-15 Linear motor system

Country Status (2)

Country Link
JP (1) JP6965048B2 (en)
WO (1) WO2019008907A1 (en)

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110401474A (en) * 2019-07-26 2019-11-01 成都天锐星通科技有限公司 A kind of phased antenna vector modulator control voltage determines method and system

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN111564995B (en) * 2020-05-25 2021-11-19 华中科技大学 Linear oscillation motor control method based on self-adaptive full-order displacement observer

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH09112438A (en) * 1995-10-20 1997-05-02 Sanyo Electric Co Ltd Driver of linear compressor
JPH11351143A (en) * 1998-06-10 1999-12-21 Matsushita Electric Ind Co Ltd Driving device for linear compressor
JP2014527593A (en) * 2011-08-19 2014-10-16 ワールプール・エシ・ア System and method for controlling stroke and operation at the resonant frequency of a resonant linear motor

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH09112438A (en) * 1995-10-20 1997-05-02 Sanyo Electric Co Ltd Driver of linear compressor
JPH11351143A (en) * 1998-06-10 1999-12-21 Matsushita Electric Ind Co Ltd Driving device for linear compressor
JP2014527593A (en) * 2011-08-19 2014-10-16 ワールプール・エシ・ア System and method for controlling stroke and operation at the resonant frequency of a resonant linear motor

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110401474A (en) * 2019-07-26 2019-11-01 成都天锐星通科技有限公司 A kind of phased antenna vector modulator control voltage determines method and system

Also Published As

Publication number Publication date
JP6965048B2 (en) 2021-11-10
JP2019017176A (en) 2019-01-31

Similar Documents

Publication Publication Date Title
WO2018070339A1 (en) Linear motor control device and compressor equipped with same
CA2874955C (en) Balancing vibrations at harmonic frequencies by injecting harmonic balancing signals into the armature of a linear motor/alternator coupled to a stirling machine
US8560129B2 (en) Vibration control device and vehicle
US8800302B2 (en) Driving an active vibration balancer to minimize vibrations at the fundamental and harmonic frequencies
JP6965048B2 (en) Linear motor system
JP6220967B2 (en) Linear motor and equipment equipped with linear motor
JP6899720B2 (en) Linear motor system and compressor with it
Suzuki et al. Position sensor-less resonant frequency estimation method for linear compressor with assist springs
KR102189035B1 (en) Linear motor system and compressor
JP6916053B2 (en) Linear motor system
WO2019176471A1 (en) Linear compressor and linear compressor control system
WO2023013230A1 (en) Linear motor control device, suspension system provided with same, and linear motor control method
Suzuki et al. Resonant frequency tracking control for a linear compressor with assist springs
Benecke et al. Design and control of a linear reluctance motor for a vacuum diaphragm pump
JP2009017755A (en) Linear-motor controller and stirling engine
Kimpara Mitigation of Torque Ripple and Vibration in Switched Reluctance Motor Drives: A Switching Optimization
KR20050096318A (en) Approximaion of estimated parameters in the sensorless stroke control for linear compressors

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 18828149

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 18828149

Country of ref document: EP

Kind code of ref document: A1