WO2018157797A1 - 一种全桥谐振变换器 - Google Patents

一种全桥谐振变换器 Download PDF

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Publication number
WO2018157797A1
WO2018157797A1 PCT/CN2018/077437 CN2018077437W WO2018157797A1 WO 2018157797 A1 WO2018157797 A1 WO 2018157797A1 CN 2018077437 W CN2018077437 W CN 2018077437W WO 2018157797 A1 WO2018157797 A1 WO 2018157797A1
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Prior art keywords
bridge
full
resonant
switching
rectifying
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PCT/CN2018/077437
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English (en)
French (fr)
Inventor
王跃斌
崔荣明
宋栋梁
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深圳市皓文电子有限公司
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Publication of WO2018157797A1 publication Critical patent/WO2018157797A1/zh

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Definitions

  • This invention relates to the field of power supplies and, more particularly, to a full bridge resonant converter.
  • the constant frequency phase shift control has become the first choice in the industry due to the simple control mode and parameter design.
  • the problem of constant frequency phase shift control is that the soft switching of the switching tube cannot be realized in a wide input voltage range and a wide load variation range, which affects the efficiency of the converter and causes serious EMI interference. Therefore, in the prior art, the resonant converter generally has the defects that the design method is complicated, the magnetic component is difficult to design, and the soft switch cannot be realized in a large range.
  • the technical problem to be solved by the present invention is that the above-mentioned design method is complicated for the prior art, the magnetic component is difficult to design, and the defect of the soft switch can not be realized in a large range, and the design method is simple and the magnetic component design is relatively easy.
  • the technical solution adopted by the present invention to solve the technical problem thereof is: constructing a full-bridge resonant converter, comprising a full-bridge inverter unit, wherein the full-bridge inverter unit converts an input DC voltage into a square wave, and the square wave sequentially Obtaining an output DC voltage through a resonant network, a high frequency transformer, and a rectifying and filtering unit, and further comprising: a passive switch disposed in the full bridge inverting unit for switching the switching tube of the full bridge inverting unit to zero crossing Auxiliary network.
  • the full bridge inverter unit includes a first half bridge and a second half bridge connected to two DC voltage input ends, and the first half bridge and the second half bridge respectively include two through a switch end serially connected to the switch tube on the DC input voltage terminal;
  • the auxiliary network includes two end points, and the end points are respectively connected with the two switch tubes of the first half bridge and the second half bridge connection.
  • the auxiliary network includes an auxiliary inductor and a DC blocking capacitor; one end of the auxiliary inductor is connected to one end of the DC blocking capacitor, and the other end of the auxiliary inductor and two switches of the first half bridge The connection points of the tubes are connected, and the other end of the DC blocking capacitor is connected to the connection point of the two switching tubes of the second half bridge.
  • the resonant network includes a resonant inductor, a first resonant capacitor, and a second resonant capacitor; the resonant inductor and the first resonant capacitor are serially connected in series to the full bridge inverter unit and the On the signal loop connected to the high frequency transformer winding, the second resonant capacitor is connected in parallel to the primary or secondary winding of the high frequency transformer.
  • one end of the first resonant capacitor is connected to a connection point of two switching tubes of the first half bridge, and the other end thereof is connected to one end of the resonant inductor; the other end of the resonant inductor and the high frequency
  • One end of the primary side of the transformer is connected, and the other end of the primary side is connected to a connection point of two switching tubes of the second half bridge; the second resonant capacitor is connected to the primary side or the secondary side of the high frequency transformer on.
  • the primary side of the high frequency transformer is a winding
  • the secondary side is one or more windings.
  • two switching tubes in a half bridge circuit are respectively controlled by pulse width modulation signals input by their control terminals and each having a 50% duty ratio and having a phase difference of 180 degrees, and two of the one half bridges
  • the control signal is advanced or delayed by a set width at its adjacent high-low level switching timing to form a dead zone of a set width to prevent the two switching tubes from being simultaneously turned on; the two half-bridge circuits are located therein
  • the control signals of the two switching tubes at the top diagonal position have a set phase difference or phase shift angle, and the set phase difference determines the pulse width of the output square wave of the inverter unit.
  • the rectifying and filtering circuit includes a rectifying portion and a filtering portion; the rectifying portion is constituted by a rectifying device, and a connection topology of the rectifying device includes a double current rectification, a full wave rectification form or a full bridge rectification form.
  • the rectifying device is a diode using a common anode or a common cathode current doubler rectifying circuit or a MOSFET using a synchronous rectifying circuit; and the filtering portion is an LC combined filter circuit.
  • the rectifying device is a diode using a common anode or a common cathode rectifying circuit; and the filtering portion thereof is an LC combined filter circuit.
  • a full-bridge resonant converter embodying the present invention has the following beneficial effects: since an auxiliary network is provided in the inverter unit (ie, the switching unit), there is a current in the auxiliary network that varies with the state of the switching device. These currents provide a beneficial supplement to the switching device when it changes state, and the resonant network connected to the output of the switching device cooperates to make the magnetic component of the converter more efficient and simpler in design, while ensuring a wide range of Switching device zero-crossing switching.
  • FIG. 1 is a schematic structural view of an embodiment of a full bridge resonant converter of the present invention
  • Figure 2 is a circuit diagram of a case in the embodiment
  • Figure 3 is a schematic diagram of waveforms in the embodiment
  • Fig. 4 is a schematic structural view of a converter in another case in the embodiment.
  • the full-bridge resonant converter includes a full-bridge inverter unit, and the full-bridge inverter unit inputs an input DC voltage (usually The input power supply is converted into a square wave, and the square wave sequentially passes through the resonant network, the high frequency transformer and the rectifying and filtering unit to obtain an output DC voltage, and further includes being disposed in the full bridge inverter unit (ie, connected in the a passive auxiliary network for switching the switching tube of the full-bridge inverter unit to zero-crossing.
  • the inverter unit of the converter includes four switch tubes, and the four switch tubes are connected together in a full bridge topology to form an inverter unit.
  • the above-described full-bridge inverter unit includes a first half bridge and a second half bridge connected to two DC voltage input ends, and the first half bridge and the second half bridge respectively include two through a switch end serially connected to the switch tube on the DC input voltage terminal;
  • the auxiliary network includes two end points, and the end points are respectively connected with the two switch tubes of the first half bridge and the second half bridge connection. That is to say, in the embodiment, the auxiliary network is connected between the other half bridges in the full bridge inverter unit, and the connection point of each half bridge is the switch end of the two switch tubes of the half bridge.
  • the auxiliary network includes an auxiliary inductor and a DC blocking capacitor; one end of the auxiliary inductor is connected to one end of the DC blocking capacitor, and the other end of the auxiliary inductor is opposite to the first half bridge
  • the connection points of the switch tubes are connected, and the other end of the DC blocking capacitor is connected to the connection point of the two switch tubes of the second half bridge.
  • the resonant network includes a resonant inductor, a first resonant capacitor, and a second resonant capacitor; the resonant inductor and the first resonant capacitor are connected in series and then connected in series to the full-bridge inverter unit The second resonant capacitor is connected in parallel to the winding of the high frequency transformer on the signal loop connected to the high frequency transformer winding.
  • one end of the first resonant capacitor is connected to a connection point of two switching tubes of the first half bridge, and the other end thereof is connected to one end of the resonant inductor; the other end of the resonant inductor and the high frequency
  • One end of the primary side of the transformer is connected, and the other end of the primary side is connected to a connection point of two switching tubes of the second half bridge; the second resonant capacitor is connected to the primary side or the secondary side of the high frequency transformer on.
  • two switching tubes in one half-bridge circuit are respectively controlled by pulse width modulation signals input by their control terminals and each having a 50% duty ratio and having a phase difference of 180 degrees, and
  • the two control signals of the one half bridge are respectively advanced or delayed by a set width at their adjacent high and low level switching timings to form a dead zone of a set width to prevent the two switch tubes from being simultaneously turned on; a set phase difference or phase shift angle between control signals of two switching tubes located at their topological diagonal positions in the two half bridge circuits, the set phase difference determining the square wave output of the inverter unit Pulse width.
  • the primary side of the high frequency transformer is a winding
  • the secondary side is one or more windings.
  • the second resonant capacitor can be connected to the primary winding of the high frequency transformer or to the secondary winding of the high frequency transformer.
  • the second resonant capacitor is connected to the primary winding of the high-frequency transformer. Referring to FIG. 2, in FIG. 2, the second resonant capacitor is connected to the primary winding of the high-frequency transformer. of.
  • FIG. 4 is a schematic structural view of the converter in one embodiment in the present embodiment.
  • the second resonant capacitor of the resonant network is connected to the secondary winding of the high frequency transformer.
  • Fig. 2 shows a specific circuit diagram of the full bridge resonant converter in one case in this embodiment.
  • the second resonant capacitor in Figure 2 is connected in parallel to the primary winding of the high frequency transformer.
  • the inverter unit includes switch tubes S1, S2, S3, and S4 and their accessory components
  • the auxiliary network includes an auxiliary inductor L a and a DC blocking capacitor C g
  • the resonant network includes a resonant inductor L r
  • the rectifying and filtering unit comprises a diode D1, a diode D2, an inductor L f1 , an inductor L f2 , and a capacitor C f
  • the T1 is a high frequency transformer.
  • the inverter unit includes a first half bridge circuit composed of a first switch tube S1 and a second switch tube S2, and a second half bridge circuit composed of a third switch tube S3 and a fourth switch tube S4;
  • One end of the input is sequentially connected to the two switch ends of the first switch tube S1 and the third switch tube S3 and the two switch ends of the second switch tube S2 and the fourth switch tube S4 to the other end of the DC input; in other words
  • the two half bridges are connected in parallel at both ends of the DC input; in the half bridge circuit, the two switch tubes are respectively controlled by pulse width modulation control signals input by the control terminals thereof and each having a 50% duty ratio and a phase difference of 180 degrees
  • the two half-bridge circuits are combined to form a full-bridge circuit having a set phase difference or phase shift angle between control signals of the two switching tubes at the diagonal position of the topology of the full-bridge circuit.
  • the phase difference determines a pulse width of the output square wave of the inverter unit; adjusting the phase difference or the phase shift angle can adjust a DC level of the output of the DC/DC full-bridge resonant converter; the two arms of the inverter unit Switching tube drive signal PWM1, PWM2, PWM3, PWM4 and the output inverter voltage waveform V AB and the current waveforms i r , i La flowing through the resonance network and the auxiliary network are as shown in FIG. 3 .
  • the auxiliary inductance L a of the auxiliary network unit and the DC blocking capacitor C g are obtained after the series connection, and the two ends are respectively connected with the common connection end of the first switching tube S1 and the second switching tube S2 of the first inverter half bridge. It is connected to the common connection end of the third switch tube S3 and the fourth switch tube S4 of the second inverter half bridge. As shown in FIG.
  • the auxiliary network When the second switch S2 and the third switch S3 are turned on, that is, during the [ t 3 - t 4 ] time period, the auxiliary network is connected in parallel with the input source, and the current i La in the auxiliary network is linearly reduced; When the second switch S2 and the fourth switch S4 are turned on, that is, during the [ t 4 - t 5 ] time period, the auxiliary network is disconnected from the input source, and the current i La in the auxiliary network is constant at this time.
  • an auxiliary network added between the first inverter half bridge and the second inverter half bridge interacts with the resonant network during operation, so that the resonant network
  • the performance improvement of the converter in which it is located is further improved.
  • the inductive component in the impedance network of the rear stage of the full bridge circuit is increased, so that the current phase flowing through the primary switching tube lags behind in the actual working process.
  • the full bridge circuit output voltage phase ensures zero-crossing switching of the switching device over a larger range of the converter. In this way, not only the consistency of the performance of the converter over a large range is ensured, but also the use range is wide, and at the same time, the complexity of the converter circuit or the magnetic circuit parameter design is also reduced to some extent. .
  • the resonant network obtains the square wave voltage output from the inverter unit from the connection point of the first switch tube S1 and the second switch tube S2, and the connection point between the third switch tube S3 and the fourth switch tube S4.
  • Figure 3 shows the v AB voltage waveform.
  • the resonant network extracts the fundamental component of the square wave voltage, transmits it to the high-frequency transformer for electrical isolation, and performs rectification and filtering by the rectifying and filtering unit to transmit to the load.
  • the rectifying and filtering circuit includes a rectifying portion and a filtering portion; the rectifying portion is composed of rectifying devices, and the rectifying devices may be connected together by using various topologies to complete rectification, for example, current doubler rectification, Topological connection form for full wave rectification or full bridge rectification.
  • the rectifying device may be a diode using a common anode or a common cathode current doubler rectifier circuit or a MOSFET using a synchronous rectification circuit; the filtering portion is an LC combined filter circuit.
  • the rectifying device may be a diode using a common anode or a common cathode rectifying circuit; the filtering portion is an LC combined filter circuit.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Rectifiers (AREA)

Abstract

一种全桥谐振变换器，包括全桥逆变单元、谐振网络、高频变压器和整流滤波单元。全桥逆变单元将输入直流电压转换为方波，该方波依次经过谐振网络、高频变压器和整流滤波单元而得到输出直流电压。该变换器还包括设置在全桥逆变单元内的、用于使全桥逆变单元的开关管过零切换的无源辅助网络。该变换器的磁性元件的利用率较高、设计较简单，同时保证了大范围内的开关器件过零切换。

Description

一种全桥谐振变换器 技术领域
本发明涉及电源领域，更具体地说，涉及一种全桥谐振变换器。
 背景技术
随着电力电子技术的高速发展，功率变换器的高功率密度和小型化就越受到重视，而变换器开关频率的提高对实现高功率密度和小型化有着非常重要的作用。然而随着开关频率的提高，器件的开关损耗极大的降低了变换器效率，也限制了变换器的高功率密度和小型化。为了提高变换器的转换效率，通常采用在变换器上增加谐振网络的办法，使得变换器成为谐振变换器。一般来讲，谐振变换器有自然的软开关特性，因此被广泛应用。在采用变频控制的谐振变换器中，由于开关频率的变化使得控制方法设计复杂，磁性元件较难设计；而恒频移相控制由于控制方式和参数设计简单便成了业界的首选。然而恒频移相控制的问题就是不能够在宽输入电压范围和宽负载变化范围内实现开关管的软开关，影响变换器效率的提升，还会带来严重的EMI干扰。因此，在现有技术中，谐振变换器通常存在设计方法复杂、磁性元件较难设计、不能再较大范围内实现软开关的缺陷。
 技术问题
在此处键入技术问题描述段落。
 技术解决方案
本发明要解决的技术问题在于，针对现有技术的上述设计方法复杂、磁性元件较难设计、不能再较大范围内实现软开关的缺陷，提供一种设计方法简单、磁性元件设计较为容易、能在较大范围内实现开关管的软开关的一种全桥谐振变换器。
本发明解决其技术问题所采用的技术方案是：构造一种全桥谐振变换器，包括全桥逆变单元，所述全桥逆变单元将输入直流电压转换为方波，所述方波依次经过谐振网络、高频变压器和整流滤波单元而得到输出直流电压，还包括设置在所述全桥逆变单元内的、用于使所述全桥逆变单元的开关管过零切换的无源辅助网络。
更进一步地，所述全桥逆变单元包括并接在两个直流电压输入端上的第一半桥和第二半桥，所述第一半桥和第二半桥分别包括两个通过其开关端依次串接在所述直流输入电压端上的开关管；所述辅助网络包括两个端点，所述端点分别与所述第一半桥和第二半桥的两个开关管的连接点连接。
更进一步地，所述辅助网络包括辅助电感和隔直电容；所述辅助电感的一端和所述隔直电容的一端相连，所述辅助电感的另一端与所述第一半桥的两个开关管的连接点相连，所述隔直电容的另一端与所述第二半桥的两个开关管的连接点相连。
更进一步地，所述谐振网络包括谐振电感、第一谐振电容和第二谐振电容；所述谐振电感和所述第一谐振电容串接后再串接在所述全桥逆变单元与所述高频变压器绕组连接的信号回路上，所述第二谐振电容并接在所述高频变压器的原边或副边绕组上。
更进一步地，所述第一谐振电容一端与所述第一半桥的两个开关管的连接点连接，其另一端与所述谐振电感一端连接；所述谐振电感另一端与所述高频变压器的原边一端连接，所述原边另一端与所述第二半桥的两个开关管的连接点连接；所述第二谐振电容并接在所述高频变压器的原边或副边上。
更进一步地，所述高频变压器的原边为一个绕组，其副边为一个或多个绕组。
更进一步地，一个半桥电路中两个开关管分别由其控制端输入的、各自具有50%占空比且相位相差180度的脉冲调宽调制信号控制，且所述一个半桥的两个控制信号在其相邻的高低电平转换时刻分别提前或延迟一个设定宽度，形成设定宽度的死区以防止所述两个开关管同时导通；所述两个半桥电路中位于其拓扑对角线位置的两个开关管的控制信号之间具有设定的相位差或移相角，所述设定的相位差决定所述逆变单元输出方波的脉冲宽度。
更进一步地，所述整流滤波电路包括整流部分和滤波部分；所述整流部分由整流器件构成，所述整流器件的连接拓扑包括倍流整流、全波整流形式或全桥整流形式。
更进一步地，在倍流整流时，所述整流器件为采用共阳极或共阴极倍流整流电路的二极管或为采用同步整流电路的MOSFET；其滤波部分为LC组合滤波电路。
更进一步地，在全波整流时，所述整流器件为采用共阳极或共阴极整流电路的二极管；其滤波部分为LC组合滤波电路。
 有益效果
实施本发明的一种全桥谐振变换器，具有以下有益效果：由于在逆变单元（即开关单元）内设置有辅助网络，使得该辅助网络中存在随开关器件的状态变化而变化的电流，这些电流在开关器件状态变化时，为其提供有益的补充，和连接在开关器件输出的谐振网络配合，使得变换器的磁性元件的利用率较高、设计较简单，同时保证了大范围内的开关器件过零切换。
 附图说明
图1是本发明一种全桥谐振变换器实施例的结构示意图；
图2是所述实施例中一种情况下的电路图；
图3是所述实施例中的波形示意图；
图4是所述实施例中另一种情况下的变换器的结构示意图。
 本发明的最佳实施方式
在此处键入本发明的最佳实施方式描述段落。
 本发明的实施方式
下面将结合附图对本发明实施例作进一步说明。
如图1所示，在本发明的一种全桥谐振变换器实施例中，该全桥谐振变换器，包括全桥逆变单元，该全桥逆变单元将输入的直流电压（通常是由输入电源提供的）转换为方波，所述方波依次经过谐振网络、高频变压器和整流滤波单元而得到输出直流电压，还包括设置在所述全桥逆变单元内的（即连接在所述逆变单元上的）、用于使所述全桥逆变单元的开关管过零切换的无源辅助网络。
在本实施例中，变换器的逆变单元为包括4个开关管，这4个开关管按照全桥拓扑连接在一起，构成逆变单元。更具体而言，上述全桥逆变单元包括并接在两个直流电压输入端上的第一半桥和第二半桥，所述第一半桥和第二半桥分别包括两个通过其开关端依次串接在所述直流输入电压端上的开关管；所述辅助网络包括两个端点，所述端点分别与所述第一半桥和第二半桥的两个开关管的连接点连接。也就是说，在本实施例中，上述辅助网络是连接在全桥逆变单元中的另个半桥之间的，和每个半桥的连接点就是该半桥两个开关管的开关端之间的连接点。在本实施例中，所述辅助网络包括辅助电感和隔直电容；所述辅助电感的一端和所述隔直电容的一端相连，所述辅助电感的另一端与所述第一半桥的两个开关管的连接点相连，所述隔直电容的另一端与所述第二半桥的两个开关管的连接点相连。
在本实施例中，所述谐振网络包括谐振电感、第一谐振电容和第二谐振电容；所述谐振电感和所述第一谐振电容串接后再串接在所述全桥逆变单元与所述高频变压器绕组连接的信号回路上，所述第二谐振电容并接在所述高频变压器的绕组上。更具体地，所述第一谐振电容一端与所述第一半桥的两个开关管的连接点连接，其另一端与所述谐振电感一端连接；所述谐振电感另一端与所述高频变压器的原边一端连接，所述原边另一端与所述第二半桥的两个开关管的连接点连接；所述第二谐振电容并接在所述高频变压器的原边或副边上。
对于驱动信号而言，在本实施例中，一个半桥电路中两个开关管分别由其控制端输入的、各自具有50%占空比且相位相差180度的脉冲调宽调制信号控制，且所述一个半桥的两个控制信号在其相邻的高低电平转换时刻分别提前或延迟一个设定宽度，形成设定宽度的死区以防止所述两个开关管同时导通；所述两个半桥电路中位于其拓扑对角线位置的两个开关管的控制信号之间具有设定的相位差或移相角，所述设定的相位差决定所述逆变单元输出方波的脉冲宽度。
此外，在本实施例中，上述高频变压器的原边为一个绕组，其副边为一个或多个绕组。而第二谐振电容在本实施例中既可以连接在高频变压器的原边绕组上，也可以连接在高频变压器的副边绕组上。在本实施例中，上述第二谐振电容连接在高频变压器的原边绕组上的情况请参见图2，在图2中，该第二谐振电容就是并接在高频变压器的原边绕组上述的。而在第二谐振电容连接在上述高频变压器副边，且该副边具有多个绕组的情况下，第二谐振电容可以连接在副边的一个绕组上或等效为多个较小电容分别连接在所述副边的多个绕组或多个绕组中间的部分绕组上。请参见图4，图4给出了本实施例中，一种情况下变换器的结构示意图，在图4中，谐振网络的第二谐振电容就是连接在高频变压器的副边绕组上的。
图2示出了在本实施例中一种情况下，该全桥谐振变换器的具体电路图。图2中的第二谐振电容并接在高频变压器的原边绕组上。具体来讲，在图2中，逆变单元包括开关管S1、S2、S3、S4及其附属元件，辅助网络包括辅助电感 L a和隔直电容C g，谐振网络包括谐振电感 L r、第一电容C s和第二电容C p，整流滤波单元包括二极管D1、二极管D2，电感L f1、电感L f2以及电容C f，T1为高频变压器。
在图2中，逆变单元包括由第一开关管S1和第二开关管S2组成的第一半桥电路以及由第三开关管S3和第四开关管S4组成的第二半桥电路；直流输入的一端分别依次通过第一开关管S1和第三开关管S3的两个开关端以及第二开关管S2和第四开关管S4的两个开关端连接到直流输入的另一端；换句话说，上述两个半桥并联在直流输入的两端；一个半桥电路中两个开关管分别由其控制端输入的、各自具有50%占空比且相位相差180度的脉冲宽度调制控制信号控制；两个半桥电路组合在一起形成一个全桥电路，位于该全桥电路的拓扑对角线位置的两个开关管的控制信号之间具有设定的相位差或移相角，该设定的相位差决定所述逆变单元输出方波的脉冲宽度；调节该相位差或移相角就能够调节所述DC/DC全桥谐振变换器输出的直流电平；所述逆变单元两桥臂上的开关管驱动信号PWM1、PWM2、PWM3、PWM4与输出逆变电压波形 V AB以及流过谐振网络与辅助网络的电流波形 i r、 i La如图3所示。
辅助网络单元的辅助电感 L a和隔直电容C g，在其串联之后得到两端，该两端分别与第一逆变半桥的第一开关管S1、第二开关管S2的公共连接端和第二逆变半桥的第三开关管S3、第四开关管S4的公共连接端连接。如图3所示，当第一开关管S1和第四开关管S4导通时，即[ t 1- t 2]时间段，辅助网络与输入源正向并联，此时辅助网络中电流 i La线性增大；当第一开关管S1和第三开关管S3导通时，即[ t 2- t 3]时间段，辅助网络与输入源断开，此时辅助网络中电流 i La恒定不变；当第二开关管S2和第三开关管S3导通时，即[ t 3- t 4]时间段，辅助网络与输入源反向并联，此时辅助网络中电流 i La线性减小；当第二开关管S2和第四开关管S4导通时，即[ t 4- t 5]时间段，辅助网络与输入源断开，此时辅助网络中电流 i La恒定不变。
换句话说，本实施例中，在所述第一逆变半桥与所述第二逆变半桥之间加入的辅助网络，在工作时与所述谐振网络的相互作用，使得该谐振网络对于其所在的变换器的性能改进得到进一步的提高。具体来讲，在本实施例中，通过引入辅助网络，使得全桥电路后级的阻抗网络中感性成分得以增加，于是在实际的工作过程中，流过原边开关管的电流相位更加滞后于全桥电路输出电压相位，保证了所述变换器更大范围内的开关器件过零切换。这样，不仅保证了该变换器在较大的范围内的性能的一致性，使得其使用范围较为宽广，同时，在某种程度上，还降低了该变换器电路或磁路参数设计的复杂程度。
在本实施例中，谐振网络从第一开关管S1与第二开关管S2的连接点、第三开关管S3与第四开关管S4的连接点中取得逆变单元输出的方波电压，参见图3中 v AB电压波形。谐振网络提取该方波电压中的基波分量，传递至高频变压器进行电气隔离，并经整流滤波单元进行整流滤波后传递至负载。
此外，在本实施例中，所述整流滤波电路包括整流部分和滤波部分；所述整流部分由整流器件构成，这些整流器件可以采用多种拓扑连接在一起而完成整流，例如，倍流整流、全波整流或全桥整流的拓扑连接形式。
在倍流整流时，所述整流器件可以为采用共阳极或共阴极倍流整流电路的二极管或为采用同步整流电路的MOSFET；其滤波部分为LC组合滤波电路。在全波整流时，所述整流器件可以为采用共阳极或共阴极整流电路的二极管；其滤波部分为LC组合滤波电路。
值得一提的是，在本实施例中，除了上述涉及的倍流整流、全波整流或全桥整流的具体电路结构外，倍流整流、全波整流或全桥整流还可以是现有技术中的符合上述整流拓扑结构的任何具体的电路结构。
以上所述实施例仅表达了本发明的几种实施方式，其描述较为具体和详细，但并不能因此而理解为对本发明专利范围的限制。应当指出的是，对于本领域的普通技术人员来说，在不脱离本发明构思的前提下，还可以做出若干变形和改进，这些都属于本发明的保护范围。因此，本发明专利的保护范围应以所附权利要求为准。  
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1、一种全桥谐振变换器，包括全桥逆变单元，所述全桥逆变单元将输入直流电压转换为方波，所述方波依次经过谐振网络、高频变压器和整流滤波单元而得到输出直流电压，其特征在于，还包括设置在所述全桥逆变单元内的、用于使所述全桥逆变单元的开关管过零切换的无源辅助网络。
根据权利要求1所述的全桥开关谐振变换器，其特征在于，所述全桥逆变单元包括并接在两个直流电压输入端上的第一半桥和第二半桥，所述第一半桥和第二半桥分别包括两个通过其开关端依次串接在所述直流输入电压端上的开关管；所述辅助网络包括两个端点，所述端点分别与所述第一半桥和第二半桥的两个开关管的连接点连接。
根据权利要求2所述的全桥谐振变换器，其特征在于，所述辅助网络包括辅助电感和隔直电容；所述辅助电感的一端和所述隔直电容的一端相连，所述辅助电感的另一端与所述第一半桥的两个开关管的连接点相连，所述隔直电容的另一端与所述第二半桥的两个开关管的连接点相连。
根据权利要求3所述的全桥谐振变换器，其特征在于，所述谐振网络包括谐振电感、第一谐振电容和第二谐振电容；所述谐振电感和所述第一谐振电容串接后再串接在所述全桥逆变单元与所述高频变压器绕组连接的信号回路上，所述第二谐振电容并接在所述高频变压器的绕组上。
根据权利要求4所述的全桥谐振变换器，其特征在于，所述第一谐振电容一端与所述第一半桥的两个开关管的连接点连接，其另一端与所述谐振电感一端连接；所述谐振电感另一端与所述高频变压器的原边一端连接，所述原边另一端与所述第二半桥的两个开关管的连接点连接；所述第二谐振电容并接在所述高频变压器的原边或副边上。
根据权利要求5所述的全桥谐振变换器，其特征在于，所述高频变压器的原边为一个绕组，其副边为一个或多个绕组。
根据权利要求2-6任何一项所述的全桥谐振变换器，其特征在于，一个半桥电路中两个开关管分别由其控制端输入的、各自具有50%占空比且相位相差180度的脉冲调宽调制信号控制，且所述一个半桥的两个控制信号在其相邻的高低电平转换时刻分别提前或延迟一个设定宽度，形成设定宽度的死区以防止所述两个开关管同时导通；所述两个半桥电路中位于其拓扑对角线位置的两个开关管的控制信号之间具有设定的相位差或移相角，所述设定的相位差决定所述逆变单元输出方波的脉冲宽度。
根据权利要求7所述的全桥谐振变换器，其特征在于，所述整流滤波电路包括整流部分和滤波部分；所述整流部分由整流器件构成，所述整流器件的连接拓扑包括倍流整流、全波整流或全桥整流形式。
根据权利要求8所述的全桥谐振变换器，其特征在于，在倍流整流时，所述整流器件为采用共阳极或共阴极倍流整流电路的二极管或为采用同步整流电路的MOSFET；其滤波部分为LC组合滤波电路。
根据权利要求8所述的全桥谐振变换器，其特征在于，在全波整流时，所述整流器件为采用共阳极或共阴极整流电路的二极管；其滤波部分为LC组合滤波电路。
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