WO2017193744A1 - 一种指纹传感器及应用其的智能设备 - Google Patents

一种指纹传感器及应用其的智能设备 Download PDF

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Publication number
WO2017193744A1
WO2017193744A1 PCT/CN2017/079931 CN2017079931W WO2017193744A1 WO 2017193744 A1 WO2017193744 A1 WO 2017193744A1 CN 2017079931 W CN2017079931 W CN 2017079931W WO 2017193744 A1 WO2017193744 A1 WO 2017193744A1
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Prior art keywords
sensing
feedback
output
signal
operational amplifier
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PCT/CN2017/079931
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English (en)
French (fr)
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戴孟均
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戴孟均
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Publication of WO2017193744A1 publication Critical patent/WO2017193744A1/zh

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    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06V—IMAGE OR VIDEO RECOGNITION OR UNDERSTANDING
    • G06V40/00—Recognition of biometric, human-related or animal-related patterns in image or video data
    • G06V40/10—Human or animal bodies, e.g. vehicle occupants or pedestrians; Body parts, e.g. hands
    • G06V40/12—Fingerprints or palmprints
    • G06V40/13—Sensors therefor
    • G06V40/1306—Sensors therefor non-optical, e.g. ultrasonic or capacitive sensing
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F21/00—Security arrangements for protecting computers, components thereof, programs or data against unauthorised activity
    • G06F21/30—Authentication, i.e. establishing the identity or authorisation of security principals
    • G06F21/31—User authentication
    • G06F21/32—User authentication using biometric data, e.g. fingerprints, iris scans or voiceprints
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06V—IMAGE OR VIDEO RECOGNITION OR UNDERSTANDING
    • G06V40/00—Recognition of biometric, human-related or animal-related patterns in image or video data
    • G06V40/10—Human or animal bodies, e.g. vehicle occupants or pedestrians; Body parts, e.g. hands
    • G06V40/12—Fingerprints or palmprints
    • G06V40/1347—Preprocessing; Feature extraction
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F2218/00—Aspects of pattern recognition specially adapted for signal processing
    • G06F2218/02—Preprocessing
    • G06F2218/04—Denoising

Definitions

  • the present invention relates to the field of sensor technologies, and in particular, to a capacitive fingerprint sensor and a smart device using the same.
  • Optical fingerprint sensor refers to traditional technology, has been industrialized for many years; Disadvantages: volume can not be miniaturized, can not be used for mobile phones); Ultrasonic fingerprint sensor (just emerging, need MEMS process Manufacturing sensors, special manufacturing process, high threshold, 3D digital image processing, floating point digital signal processing chip, high power consumption, high complexity of signal processing chip design, and therefore high cost); pressure sensitive fingerprint Sensor (no industrialization); thermal sensing fingerprint sensor (integrated on the chip, once industrialized, but has been eliminated); capacitive sensing fingerprint sensor.
  • the capacitive sensing fingerprint sensor cannot be directly installed under the screen protection glass of the mobile phone. It is necessary to install a capacitive sensing fingerprint sensor on the screen protection glass of the mobile phone. The process is complicated and the degree of freedom of design is also affected.
  • the typical thickness of the screen protector glass is 0.7mm. The current capacitive sensing fingerprint sensor cannot capture fingerprint images through 0.7mm insulating media.
  • the working principle of the capacitive sensing fingerprint sensor is as follows:
  • the fingerprint path of the human hand has a depth difference, the convex place is called “ridge”, and the concave place is called “valley”.
  • the distance between the "peak” and “valley” on the fingerprint and the sensing electrode Different, the corresponding sensing capacitors are different in size; the capacitance measuring circuit connected to the sensing electrodes converts the sensing capacitances on the corresponding electrodes of "peak” and “valley” into corresponding voltage signal outputs, thereby obtaining fingerprint "peak” and "valley” image.
  • a typical capacitive sensing fingerprint sensor is composed of a plurality of capacitive sensing electrodes and corresponding capacitance measuring circuits, each capacitive sensing electrode corresponding to one pixel of a fingerprint image.
  • the capacitance measuring circuit obtains the corresponding "peak” and “valley” information by measuring the capacitance between the capacitive sensing electrode and the finger.
  • the sensing electrodes of a typical capacitive sensing fingerprint sensor are arranged in an equidistant array of rows and columns. The row spacing is equal to the column spacing, the dimensions of the electrodes are the same, and the area of each electrode is the same. Therefore, it can be considered that the electrical characteristic parameter of each electrode is identical.
  • the finger does not have "peak” and "valley", the finger is very flat, and the distance between the finger and each electrode is exactly the same, then the capacitance of each electrode will be the same.
  • the actual finger has the existence of "peak” and “valley”.
  • the distance between "peak” and “valley” is different from that of each electrode, so the capacitance between "peak” and “valley” and the corresponding position of the sensing electrode is It will be different.
  • the fingerprint information image of the "peak” and "valley” of the finger can be obtained.
  • the difference in sensing capacitance between different sensing electrodes is the final useful fingerprint information, and the average capacitance of the sensing electrodes is not useful for fingerprint identification.
  • the closest distance is much smaller than the height difference between the "peak” and the “valley”.
  • the difference between the peak capacitance and the valley is larger than the maximum capacitance.
  • the closest distance is much larger than the height difference between the "peak” and the "valley”.
  • the difference between the "peak” and the "valley” corresponds to the difference between the sensing capacitance and the maximum capacitance. small.
  • the capacitance measuring circuit converts the capacitance value of the sensing capacitor into a corresponding voltage and amplifies the appropriate multiple. The amplified output voltage, the larger the better, but the saturation distortion.
  • the capacitance measuring circuit converts the inductive capacitance into a voltage and amplifies the process, and the difference between the capacitance values of the "peak” and the “valley” is constant with respect to the maximum capacitance value, so when the insulating medium between the finger and the sensing electrode is thick,
  • the output voltage difference corresponding to "peak” and “valley” accounts for only a small portion of the maximum output voltage.
  • the maximum value of the output voltage is limited by the operating supply voltage of the measuring circuit and is therefore limited.
  • the typical capacitor measurement circuit power supply voltage is 3 V.
  • the voltage output from the capacitance measurement circuit needs to be converted into a digital signal by an analog-to-digital converter and sent to the fingerprint identification algorithm module for digitization.
  • the analog-to-digital conversion has a certain number of resolution bits.
  • the analog-to-digital conversion is digitized, if the difference portion of the useful signal is a percentage Small, the digital amplitude of the difference after analog-to-digital conversion will be too small, and the signal quality of the effective part of the fingerprint signal will be poor.
  • the number of resolution bits of analog-to-digital conversion can be improved, due to various noises, when the resolution is smaller than the noise, it is useless to increase the resolution.
  • the number of bits of the high-modulus conversion resolution will also increase. The cost and power consumption of the circuit.
  • the method of subtracting the reference capacitance, subtracting a part of the average capacitance, or subtracting a part of the output average voltage is not easy to output the signal. Saturation distortion, which increases the amplification of the signal and ultimately increases the proportion of the effective signal in the output voltage signal.
  • the size of the offset must be set in advance, and the circuit can't automatically balance the offset capacitor average.
  • the average value drifts due to various reasons such as temperature, power supply, thickness variation of the dielectric, wear, etc., or it needs to be re-measured and then re- Set the offset value, or you can only leave a large enough tolerance to allow a certain range of changes.
  • the average value cannot be offset perfectly.
  • the finger body the finger and the human body are the same conductor, it can be regarded as the same object.
  • the capacitance between the ground and the ground of the sensing circuit is an indeterminate capacitance value. It may change greatly depending on the application. The capacitance between the human body and the circuit ground will also affect the final level.
  • the average capacitance value Therefore, the solution in this patent does not automatically maximize the proportion of useful signals, so that the best fingerprint image cannot be acquired. technical problem
  • the main object of the present invention is to provide a sensitivity that can improve the sensitivity of a capacitive sensing fingerprint sensor, and can reduce noise, thereby enabling a capacitive sensing fingerprint sensor to be obtained through a thicker insulating medium.
  • a fingerprint sensor includes: a plurality of sensing electrodes, a sensing conversion circuit and a feedback generating circuit, wherein the plurality of sensing electrodes are connected to an input end of the sensing conversion circuit, and an output of the sensing conversion circuit The end is connected to the input end of the feedback generating circuit, and the output end of the feedback generating circuit is connected to the input end of the sensing conversion circuit, wherein the plurality of sensing electrodes are used to collect multiple fingerprints generated by the finger a sensing capacitor circuit, configured to convert a plurality of sensing capacitors and feedback signals generated by the plurality of sensing electrodes into a plurality of output signals and output the signals to the feedback generating circuit;
  • the feedback generating circuit is configured to accumulate and amplify a plurality of output signals output by the sensing conversion circuit to obtain a feedback signal, and feed back the feedback signal to an input end of the sensing conversion circuit;
  • the sensing conversion circuit is further configured to obtain a plurality of fingerprint peak-to-valley height difference signals according to the plurality of sensing capacitors and the feedback signal.
  • the sensing conversion circuit is specifically configured to generate the output signal V8 by using the following formula, and output the output signal V8 through an output terminal:
  • V8 Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8*V12*C6.
  • C6 is a sensing capacitor connected to the input end of the sensing conversion circuit
  • V12 is a finger-induced noise
  • VI 0 is a feedback signal
  • Kc6_v8 is a conversion coefficient of the sensing capacitance converted into an output signal
  • Kvl0_v8 is a feedback signal converted into an output
  • the conversion factor of the signal, Kvl2_v8 is the conversion factor of the finger induced noise converted into the output signal.
  • the sensing conversion circuit is specifically configured to generate the fingerprint peak-to-valley height difference signal by using the following formula, and output the fingerprint peak-to-valley height difference through the output end.
  • Signal V 8[i]: V8[i] Kc6_v8*C6i/(lK)+Kc6_v8*Cdet[i].
  • C6i is the average sense capacitance
  • Cdet[i] is the difference between the sense capacitance of each measurement sensing electrode and the average inductive capacitance
  • Kc6_v8 is the conversion coefficient of the induction capacitance converted into the output signal
  • K is the amplification factor of the chirp ring.
  • a feedback component is connected between the output end of the feedback generating circuit and the input end of the sensing conversion circuit, and is configured to send a feedback signal output by the feedback generating circuit to the sensing Conversion circuit.
  • a feedback cancellation electrode is provided in one-to-one correspondence with the sensing electrodes, and the feedback element is disposed as a feedback coupling capacitance formed between the feedback cancellation electrode and the corresponding sensing electrode.
  • the feedback element comprises a resistor or capacitor or a feedback network or a shorted conductor.
  • the feedback generating circuit includes an accumulator U5 and a feedback amplifier U6, the accumulator U5 is provided with a plurality of input terminals, and each input end of the accumulator U5 and one of the sensing conversion circuits respectively The output end of the accumulator U5 is connected to the input end of the feedback amplifier U6, and the output end of the feedback amplifier U6 is connected to the input end of the sensing conversion circuit. .
  • the sensing conversion circuit includes a first sensing conversion unit 22 and a signal source VI
  • the first sensing conversion unit 22 includes a first operational amplifier U1, a second operational amplifier U2, and an inverter.
  • U4 a first capacitor C1, a first resistor R1, a second resistor R2, and a third resistor R3; an inverting input end of the first operational amplifier U1 is connected to one end of the sensing electrode and the feedback component, the first The non-inverting input terminal of the operational amplifier U1 is connected to the signal source VI.
  • the output end of the first operational amplifier U1 is connected to one end of the first capacitor C1 and one end of the first resistor R1.
  • the other end of the first capacitor C1 is connected to the first operation.
  • the other end of the third resistor R3 is connected to the inverting input end of the second operational amplifier U2, and the output end of the second operational amplifier U2 is connected to the input end of the feedback amplifying circuit, and the first resistor R1 is The second resistor R2 is equal.
  • the sensing conversion circuit includes a second sensing conversion unit 23 and a signal source V2; the second sensing conversion unit 23 includes a third operational amplifier U3 and a second capacitor C2, the third Operational amplifier U3 The inverting input terminal is connected to one end of the sensing electrode and the feedback component, and the signal source V2 is connected to the inverting input terminal of the third operational amplifier U3 through the sensing capacitor; the non-inverting input terminal of the third operational amplifier U3 is grounded.
  • the output end of the third operational amplifier U3 is connected to one end of the second capacitor C2, and the other end of the second capacitor C2 is connected to the inverting input end of the third operational amplifier U3, the third operational amplifier U3 The output is connected to the input of the feedback amplifier circuit.
  • a smart device comprising the fingerprint sensor described above.
  • the fingerprint sensor of the present invention is provided with a feedback generating circuit, and the feedback generating circuit accumulates and amplifies the output signal of the sensing conversion circuit and feeds it back to the input end of the sensing conversion circuit to make feedback.
  • the signal converted by the signal is added to the converted signal of the sensing capacitor, and after the converted average signal of the sensing capacitor is eliminated, the peak-to-valley height difference signal of the plurality of finger fingerprints is finally obtained, so that the capacitance of the finger fingerprint peak-to-valley height difference is obtained.
  • the ratio of the value of the useful signal to the maximum peak signal is maximized.
  • the same kind of finger and the sensing electrode generate the sensing capacitance, and the finger-induced noise is also generated.
  • the feedback signal is added to the finger-induced noise signal to eliminate the average value of the finger-induced noise, which improves the sensitivity of the fingerprint sensor of the present invention and reduces
  • the noise enables the fingerprint sensor of the present invention to obtain a clear fingerprint image through a thicker insulating medium, or to provide a higher and clearer fingerprint image with the same signal thickness, thereby improving the fingerprint identification comparison.
  • the correct rate reduces the false positive rate of fingerprint recognition.
  • FIG. 1 is a block diagram of a sensor circuit of the present invention
  • FIG. 2 is a schematic diagram of a sensor circuit of the present invention
  • FIG. 3 is a schematic diagram of a sensor circuit according to Embodiment 1 of the present invention.
  • FIG. 4 is a schematic diagram of a sensing conversion circuit according to Embodiment 1 of the present invention.
  • FIG. 5 is a schematic structural diagram of a sensing electrode and a feedback canceling electrode according to Embodiment 1 of the present invention.
  • FIG. 6 is a schematic structural diagram of a sensing electrode and a feedback canceling electrode according to another embodiment of the present invention.
  • FIG. 7 is a schematic plan view showing a structure of a sensing electrode and a feedback canceling electrode according to Embodiment 1 of the present invention.
  • Embodiment 8 is a schematic plan view showing a sensing electrode, a feedback canceling electrode, and a transmitting electrode according to Embodiment 2 of the present invention
  • 9 is a schematic diagram of a sensor circuit according to Embodiment 2 of the present invention.
  • FIG. 10 is a schematic diagram of a sensing conversion circuit according to Embodiment 2 of the present invention.
  • the capacitive sensing fingerprint sensor comprises a plurality of capacitive sensing electrodes, wherein the sensing electrodes are distributed on the same plane, and a layer of insulating medium is interposed between the sensed fingers, and the thickness of the insulating medium is uniform.
  • the peak of the fingerprint line contacts the insulating medium, and the distance between the peak of the fingerprint trace contacting the insulating medium and the sensing electrode is limited by a uniform thickness of the insulating medium, and the sensing electrode is used for sensing and finger fingerprint peak or The distance of the valley, thus obtaining a fingerprint image.
  • the sensing electrode and the fingerprint path are different in distance, and the sensing capacitance sensed by the sensing electrode is different.
  • the sensing capacitance sensed by the sensing electrode is converted into a sensor output voltage signal, and the sensor output voltage signal can be correlated with the fingerprint grain peak and valley after subsequent processing. Fingerprint image signal.
  • the capacitive sensing fingerprint sensor of the present invention includes an inductive electrode 1, a sensing conversion circuit 2, a feedback generating circuit 3, and a feedback element 4.
  • the input end of the sensing conversion circuit 2 is connected to the sensing electrode 1
  • the input end of the feedback generating circuit 3 is connected to the output end of the sensing conversion circuit 2
  • the output end of the feedback generating circuit 3 passes through the feedback element 4 and the sensing conversion circuit 2.
  • the input terminal is connected; the sensing electrode 1 is used for acquiring the sensing capacitance generated by the fingerprint of the finger, and the sensing conversion circuit 2 is configured to convert the sensing capacitance and the feedback signal into a plurality of output signals, and the feedback generating circuit converts the sensing circuit 2
  • the output signals are accumulated and amplified by a suitable multiple to obtain a feedback signal, and the feedback signal is sent to the input end of the sensing conversion circuit 2 via the feedback component 4, so that the converted signal and the converted capacitance are converted.
  • the signals are canceled, so that the signal finally outputted by the sensing conversion circuit 2 only has a signal of a peak-to-valley height difference of the fingerprint, that is, a signal having only a difference in peak-to-valley height between the sensor output signals, so that the useful signal of the peak-to-valley difference of the fingerprint is relatively largest. Peak signal The proportion of the proportion is maximized.
  • the peer feedback signal can also cancel the finger-induced average noise signal received on the sensing electrode. Therefore, the sensor of the present invention also has strong noise immunity.
  • a plurality of sensing electrodes 1 and sensing conversion circuits 2 respectively connected to the sensing electrodes 1 in one-to-one correspondence are provided, and image data of the sensing capacitance values of all the sensing electrodes corresponding to the peaks and valleys of the fingerprints are passed.
  • One measurement is obtained; the induction electrode involved in measuring fingerprint data is called a measurement induction electrode.
  • the output ends of each of the sensing conversion circuits 2 are respectively connected to one input terminal of the feedback generating circuit 3, and the output terminals of the feedback generating circuit 3 are respectively connected to the feedback elements 4 corresponding to the input ends of each of the sensing conversion circuits 2, respectively.
  • the inputs of the sensing conversion circuits 2 are connected.
  • the sensing capacitance between the finger and the measuring sensing electrode 1 is set to C6, and there is no direct connection between the finger and the circuit ground, but there is a sensing capacitor C11 between about lpF and 100pF or larger; between the finger and the circuit ground
  • the sensing capacitor C11 is much larger than the sensing capacitor C6 between the finger and the measuring sensing electrode 1. According to the circuit theory, it can be inferred that the equivalent capacitance between the sensing electrode 1 and the circuit ground is approximately equal to the sensing capacitor C6, so that the simplified equivalent circuit can be considered
  • the finger-to-circuit ground capacitance C11 is shorted.
  • the sensing conversion circuit 2 converts the sensing capacitor C6 into a part of the output voltage signal of the output terminal thereof.
  • the sensing conversion circuit 2 also converts the feedback signal fed back from the feedback generating circuit 3 into the output end of the sensing conversion circuit 2. Outputting another part of the voltage signal; if there is a finger-induced noise signal, the sensing conversion circuit 2 also converts the finger-induced noise signal into a part of the output voltage signal at the output of the sensing conversion circuit 2;
  • the voltage signal is at least a linear superposition of the three signals of the sensing capacitor C6, the feedback signal generated by the feedback generating circuit 3, and the finger-induced noise signal, which are converted by the sensing voltage conversion circuit 2 and the voltage electrical signals output by the sensing conversion circuit 2.
  • V8 Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8* V12*C6 (Formula 1)
  • Kc6_v8 is the influence coefficient of the sensing capacitor C6 on the output voltage signal V8 of the output of the sensing conversion circuit 2
  • Kvl0_v8 is the influence coefficient of the feedback signal V10 on the output voltage signal V8 of the output terminal of the sensing conversion circuit 2
  • Kvl2_v8 finger The influence coefficient of the induced noise V12 on the output voltage signal V8 of the output terminal of the sensing conversion circuit 2, since the influence of the finger induced noise V12 on the sensing output voltage signal V8 is also proportional to the sensing capacitor C6, Kvl2_v8*V12*C6 is The influence component of the finger induced noise V12 on the output voltage signal V8 of the sensing conversion circuit 2.
  • a sensing conversion circuit can be connected to a plurality of sensing electrodes at the same time.
  • Each of the plurality of sensing electrodes connected to the sensing circuit is involved in the measurement.
  • the sensing electrode participating in the measurement of the fingerprint data is referred to as a measuring sensing electrode, and the sensing capacitance of all the sensing electrodes corresponding to the peaks and valleys of the fingerprint
  • the image data of the value is obtained by multiple measurements.
  • the feedback generating circuit 3 includes an accumulator U5 and a feedback amplifier U6.
  • the accumulator U5 is provided with a plurality of input terminals, each of which is respectively connected to an output end of the sensing conversion circuit 2, and the output end of the accumulator U5 is The input terminals of the feedback amplifier U6 are connected, and the output terminals of the feedback amplifier U6 are connected to the input terminals of each of the sensing conversion circuits 2 via feedback elements 4 respectively connected to the input terminals of each of the sensing conversion circuits 2.
  • the accumulator U5 transmits the accumulated signal obtained by accumulating the voltage signal outputted by the sensing conversion circuit 2 to the feedback amplifier U6, and the feedback amplifier U6 amplifies the accumulated signal by a suitable multiple and outputs a feedback signal and feeds back the sensing signal through the feedback component 4.
  • the voltage signal converted by the feedback signal through the sensing conversion circuit 2 is offset by the average voltage signal converted from the sensing capacitor and the finger-induced average noise signal received on the sensing electrode.
  • the proper amplification factor of the feedback amplifier is that the amplification factor is as large as possible, but to ensure that the circuit after closed-loop feedback does not oscillate by itself, and there must be sufficient closed-loop loop stability.
  • the positive and negative polarities of the amplification factor should be correct. If the feedback signal is a negative feedback signal, it cannot be a positive feedback signal. The positive feedback signal will make the circuit oscillate. The larger the magnification of the ⁇ ring, the more the average sense capacitance is offset, but the more unstable the circuit after closed loop operation.
  • the specific amplification of the feedback amplifier and the design of the stabilization circuit can be realized by using the existing amplifier and feedback loop design techniques, which will not be described here.
  • the relationship between the feedback amplifier amplification factor and the average sense capacitance value that can be cancelled can be derived from the following derivation.
  • the feedback signal applied to the feedback element 4 is V10
  • the feedback signal V10 is coupled through the feedback element 4.
  • the voltage signal outputted by the sensing conversion circuit 2 is V8_l
  • a total of N 999 sensing electrodes participate in the fingerprint measurement, and the signal output from the accumulator U5 is V9, then the signal output by the accumulator U5 is:
  • Equation 1 Assuming that the finger-induced noise V12 is 0V, then after the closed loop, Equation 1 has:
  • V8 Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8*V12*C6
  • VIO AMP*N*Kc6_v8*C6/ (1- AMP*N*KvlO_v8)
  • V10 AMP*N*V8
  • V10/(AMP*N) is substituted, and Equation 7 is substituted, and Bay lj:
  • V8 Kc6_v8*C6/ (1- K) (Equation 8)
  • the resulting stable feedback signal V10 is associated with the sum of all C6 sums.
  • the amount of change in the average sense capacitance C6i of all the sensing electrodes, the effect on V10 is the change in the amount of C6 measured by a single sensing electrode. N times louder. N is the number of sensing electrodes. Because the effect of the sensing capacitor C6 of a single sensing electrode on the sum of all the sensing capacitors is only one effect, the average sensing capacitance has a N-fold effect on the sum of all the final sensing capacitors.
  • sensing capacitance C6 of a certain two sensing electrodes is relative to the average sensing capacitance C6i, one increases Cdet[i], one decreases Cdet[i], then the sum of the sensing capacitances C6 of all measuring sensing electrodes is not Change, so the V10 of this turn is the same as the original V10. Therefore, the output of the sensing circuit of the two other sensing electrodes other than the sensing electrode remains unchanged, and is still 1/(1) of the output of the ring. -K). 8 * variation Cdet [i] is changed Kc6_ V only two measurements of the output signal of the sensing electrode occurs.
  • the sensing capacitance C6 of all the measuring sensing electrodes is not equal to the average measuring sensing capacitance ⁇ , and the output of the sensing conversion circuit also conforms to the above formula, that is, the output of the sensing conversion circuit is equal to Kc6_v8*Cd et[ i]+(Kc6_v8* C6i/(1-K)), where Cdet[i] is the difference between the sensing capacitance of the sensing electrode and the average sensing capacitance.
  • the feedback signal V10 is determined by the sum of all measured sensing capacitances. 2.
  • the feedback signal V10 has a feedback loop, and the part that will be canceled in the output signal of each sensor conversion circuit is only related to the sum of all measured sensing capacitors, although the sensing capacitance of a single measurement sensing affects all the measured sensing capacitances. And, but in the end it is reflected in the sum of the sensing capacitors. 3.
  • the so-called average capacitance is one-ninth of the sum of the sensing capacitances. Therefore, the sum of the average capacitance and the sense capacitance is equivalent. 4. What affects V10 is the sum of the sense capacitors (ie, the average capacitance).
  • the average capacitance becomes one of the (1-K) fractions without feedback ⁇ in the output of the sense converter circuit.
  • the contribution portion of the output voltage signal of the sensing conversion circuit is not divided by (1-K).
  • the feedback causes the output component of each sensing conversion circuit to be offset by the feedback, and the affected portion of the sensing capacitance, if the average capacitance changes to Cdet[i], due to each sensing
  • the change Cdet[i], which only affects the sum of the final sense capacitance, Cdet[i] shows the change in capacitance of a single sense electrode, which basically does not affect the output of the feedback cancellation part of the output of the sensor conversion circuit. I.e., change in capacitance of a single sensing electrode 8 * still change amount Cdet [i] is reflected in Kc6_ V conversion output of this sensing circuit.
  • the capacitance change Cdet[i] of a single sensing electrode has an influence on the sum of the sensing capacitances, the influence portion thereof is equivalent to the influence of the variation of the average capacitance Cdet[i]/N on the sum of the sensing capacitances, and a single This effect of the change in capacitance of the sensing electrode has actually been reflected as an average capacitance, so it has been considered in the average capacitance.
  • the average noise voltage induced by the finger sensing output from the output of the sensing conversion circuit 2 is only one of the (1-K) fractions in the case of the ring, that is, the average noise is reduced (1-K ) times, only one (1-K) is left.
  • V8[i] Kc6_v8*C6i/(l-K)+Kc6_v8*Cdet[i].
  • the N[j] sensing capacitance is called the measuring sensing capacitance.
  • the finger-induced noise V12 is input to the sensor conversion circuit 2 through the sensing capacitor C6, which is reflected in the sensor output voltage signal, but due to the closed loop
  • the existence of a negative feedback loop will offset the average feedback of this noise.
  • the larger the amplification factor of the feedback amplifier the more the average value of the noise that will be cancelled out. If the difference between the sensing capacitance of the peak-to-valley sensing electrode is smaller, The more noise that is cancelled out, the more the noise that cannot be cancelled is proportional to the difference in induced capacitance.
  • the signal-to-noise ratio caused by the finger-induced noise V12 does not become smaller because the capacitance difference becomes smaller, and the signal-to-noise ratio does not deteriorate, so that a clearer fingerprint image can be obtained even through a thick insulating medium. Only the common mode portion (average sense capacitance) of the noise and the sense capacitor can be cancelled by the feedback signal, and the peak-to-valley difference of the sense capacitor will not be offset, and the difference portion will still be reflected in the sensor output voltage signal.
  • the terminal is assumed to be the inverting input of the operational amplifier, which is the virtual ground.
  • the noise voltage output from the sensing conversion circuit 2 is Voutnoise
  • the input noise current Inoise of the sensing conversion circuit 2 and the noise conversion voltage of the noise voltage Voutnoise outputted by the sensing conversion circuit 2 are Knoise
  • the two measuring sensing electrodes 1 The useful signal voltages of the chirped ring output are: Kc6_v8*C+ Kc6_v8* cdet and Kc6_v8*C—Kc6_v8* cdet, and the noise voltages of the two measuring electrodes are: Knoise* V12* + Knoise* V12* and Knoise* V12* - Knoise* V12*.
  • the amplification factor of the feedback amplifier is sufficiently large, and the average portion of the output voltage signal is substantially cancelled out, and the portion that is not cancelled is much smaller than the output component of the induced capacitance difference, that is, the average sensing
  • the voltage signal portion of the capacitor output is cancelled out by the feedback, and the average output noise signal portion caused by the finger induced noise is also substantially offset by the feedback.
  • the closed loop feedback, sensing signals output conversion circuit, useful in sensing capacitor C + output voltage signal is approximately CDET Kc6_ V 8 * C det
  • output noise voltage is approximately Knoise * V12 *
  • the useful voltage signal output is approximately - Kc6_v8*cdet
  • the output noise is about -Knoi Se * V12*
  • their signal-to-noise ratio is both. It can be seen that the signal-to-noise ratio caused by finger-induced noise is independent of the average induced capacitance value, and is independent of the capacitance difference between the peaks and valleys of the fingerprint.
  • the signal-to-noise ratio caused by finger-induced noise is independent of the difference between the capacitance of the peak and the valley and the average value of the induced capacitance, that is, the signal-to-noise ratio is independent of the thickness of the insulating medium, even if the distance between the finger and the measuring electrode is large.
  • the height difference between the fingerprint peaks and valleys is small, and the signal-to-noise ratio caused by finger-induced noise does not change.
  • the thickness of the insulating medium does not change, if the height difference of the peaks and valleys of the fingerprint is reduced, although the output useful signal amplitude will decrease, the output noise caused by the finger induced noise will also decrease.
  • the feedback component 4 is configured as a feedback coupling capacitor, and the feedback coupling capacitor is a spatial coupling capacitor formed between the at least one feedback canceling electrode 41 and the measuring sensing electrode.
  • the feedback cancellation electrode 41 forms a feedback coupling capacitor C7 between each of the measurement sensing electrodes 1 , and one end of the feedback coupling capacitor C7 is the measurement sensing electrode 1 , and the measuring sensing electrode 1 is connected to the input end of the sensing conversion circuit 2 , so the feedback coupling The other end of the capacitor C7 is connected to the input end of the sensing conversion circuit 2.
  • the other end of the feedback coupling capacitor C7 is a feedback canceling electrode 41.
  • the feedback canceling electrode 41 is connected to the output end of the feedback amplifier U6, so the other end of the feedback coupling capacitor C7 Connected to the output of feedback amplifier U6.
  • the feedback component 4 can also be configured as a resistor or a capacitor or a feedback network or other module. If the sensing conversion circuit 2 has a dedicated feedback signal input, the feedback element 4 can also be provided as a shorted conductor, and the function of the feedback element 4 can be implemented by other components inside the sensing conversion circuit 2.
  • the feedback canceling electrode 41 is designed to form a coupling capacitance with the finger, and the feedback canceling electrode 41 also forms a feedback coupling capacitor C7 with the sensing electrode 1.
  • the feedback canceling electrode 41 and the sensing electrode 1 are arranged in the same plane with an insulating medium between the fingers.
  • the feedback signal V10 is coupled to the finger through the finger coupling capacitor to cancel a portion of the noise signal sensed by the finger.
  • the feedback signal V10 applied to the feedback canceling electrode 41 is also coupled to the sensing electrode 1 via a feedback coupling capacitor C7.
  • the feedback canceling electrode 41 is designed to form a feedback coupling capacitor C7 with the sensing electrode 1.
  • one side of the sensing electrode 1 is an insulating dielectric material insulated from the finger, and the feedback canceling electrode 41 is disposed on the other side of the sensing electrode 1.
  • the sensing electrode 1 shields the feedback to cancel the signal coupling between the electrode 41 and the finger.
  • the feedback signal V10 applied to the feedback canceling electrode 41 is coupled to the sensing electrode 1 through a feedback coupling capacitor C7.
  • the sensing capacitor C6 is the same, if the feedback coupling capacitor C7 Similarly, the feedback signal V10 can cancel out the same average capacitance value on the sensing electrode 1 having the same distance from the finger, and can also cancel out the same finger-induced noise signal.
  • the sensing electrodes are arranged in a matrix of equal rows and columns, that is, the row distances of the adjacent sensing electrodes 1 are the same, and the columns of the adjacent sensing electrodes 1 are adjacent. The distance is the same.
  • Each of the sensing electrodes 1 has the same shape and size. All of the sensing electrodes 1 are arranged on the same plane. Between the sensing electrode 1 and the finger, one side of the sensing electrode 1 is covered with a uniform thickness of the insulating dielectric material, so that the finger pressing is performed to maintain the uniformity of the distance between the finger and the sensing electrode 1. As shown in FIG.
  • the sensing electrode 1 and the feedback canceling electrode 41 are arranged on the same plane, and the feedback canceling electrode 41 is connected by a grid-like metal into a single electrode, and the sensing electrode 1 is a square electrode of equal area, which is offset by feedback.
  • the electrode 41 separates the crucible.
  • the finger induced noise V12 can be regarded as the voltage source V12 whose output voltage is zero. Since the capacitance between the finger and the circuit ground C11 is much larger than the sensing capacitor C6, the influence of the capacitance between the finger and the circuit ground C11 on the sensing capacitor C6 can be neglected, and the capacitance between the finger and the circuit ground C11 can be regarded as a short-circuited component, which is simplified.
  • the equivalent circuit is shown in Figure 4.
  • the sensing conversion circuit 2 includes a first sensing conversion unit 22 and a signal source VI, wherein the first sensing conversion unit 22 includes a first operational amplifier U1, a second operational amplifier U2, an inverter U4, a first capacitor C1, The first resistor R1, the second resistor R2, and the third resistor R3.
  • the inverting input end of the first operational amplifier U1 is connected to the sensing electrode 1, and is equivalently connected to one end of the sensing capacitor C6.
  • the other end of the sensing capacitor C6 is connected to the finger induced noise voltage source V12, and the other end of the finger sensing noise voltage source V12
  • the other end of the feedback coupling capacitor C7 is the feedback cancellation capacitor 41, that is, the other end of the feedback coupling capacitor C7 and the feedback amplifier U6.
  • the outputs are connected.
  • the non-inverting input terminal of the first operational amplifier U1 is connected to one end of the signal source VI, and the other end of the source source VI is grounded; the output end of the first operational amplifier U1 is connected to one end of the first capacitor C1; the other end of the first capacitor C1 is An inverting input terminal of an operational amplifier U1 is connected; an output terminal of the first operational amplifier U1 is coupled to one end of the first resistor R1; and the other end of the first resistor R1 is connected to an inverting input terminal of the second operational amplifier U2;
  • the input terminal of the phaser U4 is connected to the signal source VI, the output end of the inverter U4 is connected to one end of the second resistor R2; the other end of the second resistor R2 is connected to the inverting input terminal of the second operational amplifier U2; the second operational amplifier U2
  • the non-inverting input is grounded, and the output of the second operational amplifier U2 outputs a sensor output voltage signal;
  • the resistances of the first resistor R1 and the second resistor R2 are equal. According to the theoretical principle of the circuit, it can be deduced.
  • the feedback coupling capacitor C7 is equivalent to the loop, that is, Equivalent to the ankle ring.
  • the finger-induced noise V12 is 0, the loop is not connected, the C7 loop is not connected, the signal source VI has a signal frequency of f, and the voltage value is VI.
  • the output terminal of the second operational amplifier U2 is connected to an input terminal of the accumulator U5, and the accumulator U5 accumulates the input signal V8 input from all the input terminals, and then outputs the accumulated signal V9 from the output terminal, and the output terminal of the accumulator U5 and the feedback amplifier
  • the input terminal of U6 is connected, and the feedback amplifier U6 amplifies the input accumulated signal V9 and outputs a feedback signal V10 from the output terminal of the feedback amplifier U6.
  • the output of the feedback amplifier U6 is connected to the feedback canceling electrode 41 to form a negative feedback loop amplifier, thereby forming a circuit device that can automatically cancel the average value of the sensing capacitor and the average noise output value caused by the finger induced noise.
  • Accumulator U5 and feedback amplifier U6 can be combined into one feedback signal generator, and the feedback signal generator can realize the functions of accumulation and feedback amplification.
  • the sensing conversion circuit of this embodiment includes a signal source V2 and a sensing conversion unit 23.
  • the signal source V2 is generated by the emitter 5.
  • the feedback canceling electrode 41 and the sensing electrode 1 are arranged in the same plane, and the feedback canceling electrode 41, the sensing electrode 1 and the finger have a layer of insulating medium having a uniform thickness.
  • the emitter electrode 5 is a ring-shaped electrode, which is disposed around the sensing electrode 1 and the feedback canceling electrode 41, and surrounds the sensing electrode 1 and the feedback canceling electrode 41 inside the ring.
  • a feedback coupling capacitor C7 is formed between the feedback canceling electrode 41 and the sensing electrode 1.
  • the feedback signal V10 is coupled to the finger by the feedback cancellation electrode 41 and the finger coupling capacitance between the fingers, and can cancel a part of the noise signal induced by the finger, but since the finger-to-circuit capacitance C11 is much larger than the finger coupling capacitance, and the capacitance C15 is much larger than the finger.
  • the coupling capacitor, a part of the noise signal canceled by the finger coupling capacitor, can be ignored.
  • FIG. 9 is an overall structural composition diagram of the present embodiment.
  • the analysis sensing conversion circuit 2 converts the sensing capacitor C6 into a voltage signal V8 output from the output end of the sensing conversion circuit 2, the finger induced noise V12 can be regarded as an output. Voltage source V12 with voltage 0.
  • the capacitance between the finger and the circuit ground Cl1 is much larger than the measurement sensing capacitance C6, and the capacitance C15 is much larger than the measurement sensing capacitance C6, the influence of the capacitance between the finger and the circuit ground Cl1 on the equivalent sensing capacitance C6 is negligible, and the capacitance C15 is opposite. The effect of the equivalent sense capacitor C6 can also be ignored.
  • the simplified equivalent circuit of the sensing conversion circuit 2 can be obtained as shown in FIG. 10.
  • the equivalent finger induced noise voltage source V12 and the equivalent emission signal source V2 in FIG. 10 are equivalent voltage values, and the values thereof are Compared with the values of the finger-induced noise voltage source V12 and the transmission signal source V2 in FIG. 9, the mutual conversion relationship between them can be determined according to the basic circuit theory, and the derivation of the conversion relationship and the calculation formula of the result are not repeated here.
  • the second sensing conversion unit 23 includes a third operational amplifier U3 and a second capacitor C2.
  • the inverting input terminal of the third operational amplifier U3 is connected to the sensing electrode 1, and is equivalently connected to one end of the sensing capacitor C6.
  • the other end of the capacitor C6 is connected in series with two equivalent voltage sources (ie, the equivalent finger induced noise voltage source V12 and the equivalent emission signal source V2), and the inverting input of the third operational amplifier U3 is also connected to the feedback.
  • One end of the coupling capacitor C7, the other end of the feedback coupling capacitor C7 is the feedback cancellation electrode 41; that is, the other end of the feedback coupling capacitor C7 is connected to the output terminal of the feedback amplifier U6; the non-inverting input terminal of the third operational amplifier U3 is grounded, the third operational amplifier The output of U3 is connected to one end of the second capacitor C2, the other end of the second capacitor C2 is connected to the inverting input of the third operational amplifier U3, and the output of the third operational amplifier U3 outputs a sensor output voltage signal.
  • the feedback coupling capacitor C7 is equivalent to the loop, that is equivalent to the loop, which is not difficult to derive according to the circuit theory.
  • the sensor output voltage signal V8 output from the third operational amplifier U3 is -V12* - V2*. That is, the pair of sensing conversion circuits 2
  • the output terminal of the third operational amplifier U3 is connected to one of the input terminals of the accumulator U5, and the accumulator U5 accumulates the sensed output voltage signal V8 from all the input terminals, and then outputs the accumulated signal V9 from the output terminal, and the output of the accumulator U5
  • the terminal is connected to the input terminal of the feedback amplifier U6, and the feedback amplifier U6 amplifies the input accumulated signal V9 and outputs a feedback signal V10 from the output terminal of the feedback amplifier U6.
  • the output of the feedback amplifier U6 is connected to the feedback canceling electrode 41 to form a negative feedback loop amplifier, thereby forming a circuit device that can automatically cancel the average value of the measured sensing capacitance and the average noise output value caused by the equivalent finger induced noise.
  • Accumulator U5 and feedback amplifier U6 can be combined into one feedback signal generator, and the feedback signal generator can realize the functions of accumulation and feedback amplification.
  • the results of the above derivation analysis are obtained after simplifying some conditions. If these conditions are not simplified, the analysis conclusions are the same.
  • the complicated analysis process that is not simplified is not described here, but only for the present invention. It is a preferred embodiment, but the scope of the present invention is not limited thereto, and any variations or alternatives that can be easily conceived by those skilled in the art within the technical scope of the present invention should be covered by the present invention. Within the scope of protection. Therefore, the scope of protection of the present invention should be determined by the scope of the claims.

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Abstract

一种指纹传感器，包括：多个感应电极(1)，传感转换电路(2)和反馈发生电路(3)，多个感应电极(1)与传感转换电路(2)的输入端连接，传感转换电路(2)的输出端与反馈发生电路(3)的输入端连接，反馈发生电路(3)的输出端与传感转换电路(2)的输入端连接。其中，多个感应电极(1)用于获取多个感应电容；传感转换电路(2)用于将多个感应电容(1)及反馈信号转换为多个输出信号并最终获得多个指纹峰谷高度差信号，反馈发生电路(3)根据多个输出信号获得反馈信号并反馈。提升了指纹传感器的灵敏度，降低了噪声，使指纹传感器可以透过更厚的绝缘介质获取清晰的指纹图像，或者在相同的介质厚度下，能够提供信噪比更高、更清晰的指纹图。

Description

说明书 发明名称：一种指纹传感器及应用其的智能设备 技术领域
[0001] 本发明涉及传感器技术领域， 特别涉及一种电容式指纹传感器及应用该传感器 的智能设备。
背景技术
[0002] 人体指纹识别已经是比较成熟的技术， 指纹识别主要由两项技术组成： 指纹图 像的采集和指纹的识别算法。 其中指纹图像的采集常用的技术有： 光学指纹传 感器 （是指传统技术， 已经产业化应用多年； 缺点： 体积无法小型化， 不能用 于手机） ； 超声波指纹传感器 （刚刚幵发出来， 需要 MEMS工艺制造传感器， 制 造工艺特殊， 门槛高， 还需要进行 3维数字图像处理， 需要浮点数数字信号处理 芯片， 芯片功耗高， 信号处理芯片设计复杂度也高， 因此成本也高） ； 压力感 应指纹传感器 （没有产业化） ； 热感应指纹传感器 （集成在芯片上， 曾经产业 化， 但目前已经淘汰） ； 电容感应指纹传感器。
[0003] 近几年随着智能手机的普及应用， 由于电容感应指纹传感器的体积小、 成本低 、 功耗小、 可靠性也较好， 在智能手机上得到了广泛应用。 但是目前在智能手 机上广泛使用的电容感应指纹传感器的灵敏度和信噪比有限， 手指和感应电极 间的距离不能太大， 已经量产使用的电容感应指纹传感器的电极与手指间的绝 缘介质厚度可以做到 200um,据说已有能做到绝缘介质厚度 400um的技术， 但目前 没有量产。 而且高灵敏度、 厚绝缘介质的产品需要升压电路， 成本和功耗都有 所增加。 绝缘介质厚度 400um以下的情况下， 不能直接将电容感应指纹传感器安 装在手机屏幕保护玻璃下， 需要在手机屏幕保护玻璃上幵孔安装电容感应指纹 传感器， 工艺复杂， 也影响外观设计的自由度。 手机屏幕保护玻璃的典型厚度 是 0.7mm， 目前的电容感应指纹传感器无法透过 0.7mm的绝缘介质采集指纹图像
[0004] 电容感应指纹传感器的工作原理如下： 人的手指纹路有深度差异， 凸的地方称 为"峰 (ridge)"， 凹的地方称为"谷 (valley)"。 指纹上 "峰"和"谷"与感应电极的距离 不同， 对应的感应电容大小不同;与感应电极连接的电容测量电路将 "峰"和"谷" 对应电极上的感应电容转化为相应的电压信号输出， 从而得到指纹 "峰"和"谷"的 图像。 典型的电容感应指纹传感器由多个电容感应电极和相应的电容测量电路 组成， 每个电容感应电极对应一个指纹图像的一个像素点。 电容测量电路通过 测量电容感应电极与手指间的电容量得到相应的 "峰"和"谷"的信息。 典型的电容 感应指纹传感器的感应电极是等间距的行列阵列排列， 行间距与列间距相等， 电极的尺寸形状也相同， 每个电极的面积也相同， 因此可以认为每个电极的电 特性参数是相同的。 根据平板电容原理， 如果手指没有 "峰"和"谷"， 手指非常平 整， 手指与每个电极的距离完全相同， 那么每个电极的电容也会相同。 但是， 实际的手指有 "峰"和"谷"的存在， "峰"和"谷"与每个电极的距离不同， 因而"峰" 和"谷"与对应位置的感应电极之间的电容就会不同， 只要测量感应电极与手指间 的电容的差别变化， 就能获得手指的"峰"和"谷"的指纹信息图像。 因此不同感应 电极之间的感应电容差， 才是最终的有用指纹信息， 而感应电极的平均电容对 指纹的识别没有用处。 当感应电极与手指的距离很近吋， 最近距离远小于"峰"和 "谷"的高度差吋， "峰"和"谷"对应感应电容的差值相对于最大电容的百分比就较 大， 而当感应电极与手指的距离很远吋， 最近距离远远大于"峰"和"谷"的高度差 吋， "峰"和"谷"对应感应电容的差值相对于最大电容的百分比就很小。 比如， 当 最近距离接近 0吋， "峰"和"谷"对应电容的差值相对于最大电容的变化百分比接 近 100%。 而当距离超过 400um吋， "峰"和"谷"对应电容的差值相对于最大电容的 百分比只有百分之几， 甚至小于百分之一。 电容测量电路将感应电容容值转换 为对应的电压并放大合适的倍数， 放大后的输出电压， 希望越大越好， 但也不 能饱和失真。 电容测量电路将感应电容量转换为电压并放大过程中， "峰"和"谷" 的容值差值相对最大电容值的百分比不变， 因此当手指和感应电极间的绝缘介 质较厚吋， "峰"和"谷"对应的输出电压差值只占最大输出电压的很少部分。 而输 出电压的最大值受限于测量电路的工作电源电压， 因此是有限的。 典型的电容 测量电路电源电压是 3 V， 电容测量电路输出的电压还需要经过模数转换器转换 为数字信号， 送给指纹识别算法模块进行数字化处理。 而模数转换是有一定的 分辨率位数的， 在模数转换数字化吋， 如果有用信号的差值部分所占百分比过 小， 模数转换后的差值部分数字幅度就会过小， 指纹信号有效部分的信号质量 就会很差。 虽然可以提高模数转换的分辨率位数， 但是由于存在各种噪声， 当 分辨率小于噪声吋， 再提高分辨率也就没有用处了， 过高的模数转换分辨率位 数， 也会增加电路的成本和功耗。
为了提高手指峰谷的差值部分在输出电压信号中的比例， 现有技术中， 采取减 基准电容的方法， 减去平均电容的一部分， 或者减去输出平均电压的一部分， 输出信号就不容易饱和失真， 从而可以将信号的放大倍数提高， 最终提高有效 信号在输出电压信号中的比例。 中国发明专利， 专利号 "201410105847.6"， 名称 为"电容指纹感应电路和感应器"的专利文献， 公幵一种电容指纹感应电路和感应 器， 感应电路包括抵消模块和反馈放大器， 抵消模块的第一输入端与第一外接 驱动源连接， 抵消模块的第二输入端通过按压在该输入端的手指与第二外接驱 动源连接， 抵消模块的输出端与反馈放大器的输入端连接， 第一外接驱动源和 第二外接驱动源输出的信号互为反相， 第一外接驱动源接入到抵消模块的信号 与第二外接驱动源耦合到抵消模块的信号产生抵消， 得到与指纹深度相关的指 纹感应信号， 从而实现增加有用信号的比例， 进而可以提高有用信号的放大倍 数。 但这种方案有两个缺点： 第一、 只能抵消一部分平均电容， 不能抵消手指 上感应的噪声信号， 当抵消平均电容后， 可以提高放大倍数， 但是， 提高放大 倍数吋， 手指上感应的噪声信号也被放大了， 特别是放大倍数较大吋， 手指上 感应的噪声信号就不能忽略， 因此有用的"峰"和"谷"差值电容占比很小吋， 手指 上感应的噪声信号严重影响指纹图像的质量。 第二、 抵消的大小必须预先设置 好， 电路不能自动平衡抵消电容平均值， 这样， 由于温度、 电源、 绝缘介质厚 度变化、 磨损等等各种原因造成平均值漂移吋， 要么需要重新测量后重新设置 抵消值， 要么只能留有足够大的容差， 允许一定范围的变化， 总之不能完美的 抵消平均值， 另外由于手指人体 （手指和人体是同一个导体， 可看成是同一个 物体） 与感应电路的地之间的电容是一个不确定的电容值， 随着不同的应用情 况， 可能会发生很大的变化， 而手指人体与电路地之间的电容大小， 也会影响 最终的等效平均电容值。 所以， 该专利中的方案做不到自动最大化有用信号的 占比， 从而不能采集到最好的指纹图像。 技术问题
[0006] 针对上述现有技术的不足， 本发明的主要目的在于提供一种能够提升电容感应 指纹传感器的灵敏度、 同吋能够降低噪声， 从而能够使得电容感应指纹传感器 透过更厚的绝缘介质获取清晰的指纹图像， 或者在相同的介质厚度下， 能够提 供信噪比更高、 更清晰的指纹图像的自动消除平均电容的电容感应指纹传感器
问题的解决方案
技术解决方案
[0007] 为了实现上述目的， 本发明采用以下技术方案：
[0008] 一种指纹传感器， 包括： 多个感应电极， 传感转换电路和反馈发生电路， 所述 多个感应电极与所述传感转换电路的输入端连接， 所述传感转换电路的输出端 与所述反馈发生电路的输入端连接， 所述反馈发生电路的输出端与所述传感转 换电路的输入端连接， 其中， 所述多个感应电极， 用于采集手指指纹生成的多 个感应电容； 所述传感转换电路， 用于将多个感应电极生成的多个感应电容以 及反馈信号转换为多个输出信号并输出至所述反馈发生电路；
[0009] 所述反馈发生电路， 用于将所述传感转换电路输出的多个输出信号进行累加放 大处理得到反馈信号， 并将所述反馈信号反馈至所述传感转换电路的输入端； 所述传感转换电路， 还用于根据所述多个感应电容和所述反馈信号得到多个指 纹峰谷高度差信号。
[0010] 优选地， 所述传感转换电路， 具体用于通过以下公式生成所述输出信号 V8， 并 通过输出端输出所述输出信号 V8：
[0011] V8= Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8*V12*C6。
[0012] 其中， C6为与传感转换电路输入端连接的感应电容， V12为手指感应噪声， VI 0为反馈信号; Kc6_v8为感应电容转换为输出信号的转换系数， Kvl0_v8为反馈信 号转换为输出信号的转换系数， Kvl2_v8为手指感应噪声转换为输出信号的转换 系数。
[0013] 优选地， 当手指感应噪声 V12为 0吋， 所述传感转换电路， 具体用于通过以下公 式生成所述指纹峰谷高度差信号， 并通过输出端输出所述指纹峰谷高度差信号 V 8[i]: V8[i]=Kc6_v8*C6i/(l-K)+Kc6_v8*Cdet[i]。
[0014] 其中， C6i是平均感应电容， Cdet[i]是各个测量感应电极的感应电容与平均感 应电容的差值， Kc6_v8为感应电容转换为输出信号的转换系数， K为幵环放大倍 数。
[0015] 优选地， 反馈元件， 连接在所述反馈发生电路的输出端和所述传感转换电路的 输入端之间， 用于将所述反馈发生电路输出的反馈信号发送至所述传感转换电 路。
[0016] 优选地， 设置有与所述感应电极一一对应的反馈抵消电极， 所述反馈元件设置 为由反馈抵消电极与对应感应电极间形成的反馈耦合电容。
[0017] 优选地， 所述反馈元件包括电阻或电容或反馈网路或短接的导线。
[0018] 优选地， 所述反馈发生电路包括累加器 U5和反馈放大器 U6， 所述累加器 U5设 置有多个输入端， 所述累加器 U5的各个输入端分别与一个所述传感转换电路的 输出端对应相连， 所述累加器 U5的输出端与所述反馈放大器 U6的输入端相连， 所述反馈放大器 U6的输出端与传感转换电路的输入端相连。 。
[0019] 优选地， 所述传感转换电路包括第一传感转换单元 22和信号源 VI， 所述第一传 感转换单元 22包括第一运算放大器 Ul、 第二运算放大器 U2、 反相器 U4、 第一电 容 Cl、 第一电阻 Rl、 第二电阻 R2和第三电阻 R3; 所述第一运算放大器 U1的反相 输入端与所述感应电极、 反馈元件的一端相连， 所述第一运算放大器 U1的同相 输入端接信号源 VI， 所述第一运算放大器 U1的输出端接第一电容 C1的一端、 第 一电阻 R1的一端； 所述第一电容 C1的另一端接第一运算放大器 U1的反相输入端 ； 所述反相器 U4的输入端接信号源 VI， 所述反相器 U4的输出端接第二电阻 R2的 一端； 所述第一电阻 R1的另一端、 所述第二电阻 R2的另一端均接所述第二运算 放大器 U2的反相输入端； 所述第二运算放大器 U2同相输入端接地， 所述第二运 算放大器 U2的输出端接第三电阻 R3的一端， 所述第三电阻 R3的另一端接第二运 算放大器 U2的反相输入端， 所述第二运算放大器 U2的输出端同吋接反馈放大电 路的输入端， 所述第一电阻 R1与第二电阻 R2相等。
[0020] 优选地， 所述传感转换电路包括第二传感转换单元 23和信号源 V2; 所述第二传 感转换单元 23包括第三运算放大器 U3和第二电容 C2， 所述第三运算放大器 U3的 反相输入端与感应电极、 反馈元件的一端相连， 同吋所述信号源 V2通过感应电 容与第三运算放大器 U3的反相输入端相连； 所述第三运算放大器 U3的同相输入 端接地， 所述第三运算放大器 U3的输出端与所述第二电容 C2的一端相连， 所述 第二电容 C2的另一端与第三运算放大器 U3的反相输入端相连， 所述第三运算放 大器 U3的输出端同吋接反馈放大电路的输入端。
[0021] 一种智能设备， 包括以上所述的指纹传感器。
发明的有益效果
有益效果
[0022] 相比于现有技术， 本发明的指纹传感器通过设置有反馈发生电路， 反馈发生电 路将传感转换电路的输出信号经累加、 放大后反馈至传感转换电路的输入端， 使反馈信号经转换后的信号与感应电容经转换后的信号相加， 消除感应电容经 转换后的平均信号后， 最终获得多个手指指纹的峰谷高度差信号， 使得手指指 纹峰谷高度差的电容值有用信号相对于最大峰值信号的占比比例最大化。 同吋 手指与感应电极产生感应电容的同吋也会产生手指感应噪声， 反馈信号同吋会 和手指感应噪声信号相加消除手指感应噪声的平均值信号， 提升了本发明指纹 传感器的灵敏度， 降低了噪声， 使本发明指纹传感器可以透过更厚的绝缘介质 获取清晰的指纹图像， 或者在相同的介质厚度下， 能够提供信噪比更高、 更清 晰的指纹图像， 提高了指纹识别比对的正确率， 降低指纹识别比对的认假率。 对附图的简要说明
附图说明
[0023] 图 1为本发明传感器电路框图；
[0024] 图 2为本发明传感器电路原理图；
[0025] 图 3为本发明实施例 1传感器电路原理图；
[0026] 图 4为本发明实施例 1的传感转换电路原理图；
[0027] 图 5为本发明实施例 1的感应电极和反馈抵消电极的结构示意图；
[0028] 图 6为本发明另一种实施例的感应电极和反馈抵消电极的结构示意图；
[0029] 图 7为本发明实施例 1的感应电极和反馈抵消电极的平面结构示意图；
[0030] 图 8为本发明实施例 2的感应电极、 反馈抵消电极和发射电极的平面示意图； [0031] 图 9为本发明实施例 2传感器电路原理图；
[0032] 图 10为本发明实施例 2的传感转换电路原理图；
[0033] 图中， 1、 感应电极； 2、 传感转换电路、 3、 反馈发生电路； 4、 反馈元件； 41 、 反馈抵消电极； 22、 第一传感转换单元； 23、 第二传感转换单元。
实施该发明的最佳实施例
本发明的最佳实施方式
[0034] 为了使本发明的目的、 技术方案及优点更加清楚明白， 以下结合附图及实施例
， 对本发明进行进一步详细说明。 应当理解， 此处所描述的具体实施例仅仅用 以解释本发明， 并不用于限定本发明。
[0035] 实施例 1
[0036] 电容感应指纹传感器包括有多个电容感应电极， 感应电极分布于同一个平面上 ， 与被感应手指间有一层绝缘介质， 绝缘介质厚度均匀。 测量感应指纹吋， 指 纹纹路的峰接触绝缘介质， 接触绝缘介质的指纹纹路的峰与感应电极的距离被 厚度均匀的绝缘介质限定于一个恒定的距离值， 感应电极用于感应与手指指纹 峰或谷的距离， 从而获得指纹图像。 感应电极与手指纹路距离不同， 感应电极 感应到的感应电容就不同， 将感应电极感应到的感应电容转换为传感器输出电 压信号， 传感器输出电压信号经过后续的处理就可以得到与指纹纹路峰谷相关 的指纹图像信号。
[0037] 如图 1所示， 本发明的电容感应指纹传感器包括感应电极 1、 传感转换电路 2、 反馈发生电路 3和反馈元件 4。 传感转换电路 2的输入端与感应电极 1连接， 反馈 发生电路 3的输入端与传感转换电路 2的输出端相连， 反馈发生电路 3的输出端通 过反馈元件 4与传感转换电路 2的输入端相连； 感应电极 1用于获取与手指指纹感 应产生的感应电容， 传感转换电路 2用于将感应电容及反馈信号转换为多个输出 信号， 反馈发生电路将传感转换电路 2的多个输出信号经累加、 放大合适的倍数 后得到反馈信号并将该反馈信号经反馈元件 4发送至传感转换电路 2的输入端， 使反馈信号经转换后的信号与感应电容经转换后的平均信号相抵消， 使传感转 换电路 2最终输出的信号中只有指纹峰谷高度差的信号， 即传感器输出信号中只 有指纹峰谷高度差的信号， 使得指纹峰谷差值的有用信号相对于最大峰值信号 的占比比例最大化。 同吋反馈信号还能够抵消感应电极上接收到的手指感应平 均噪声信号， 因此， 本发明的传感器还具有较强的抗噪能力。
[0038] 如图 2所示， 设置有多个感应电极 1和分别与各感应电极 1一一对应连接的传感 转换电路 2， 指纹峰谷对应的所有感应电极的感应电容值的图像数据通过一次测 量获得； 参与测量指纹数据的感应电极称为测量感应电极。 每个传感转换电路 2 的输出端分别与反馈发生电路 3的一个输入端相连， 反馈发生电路 3的输出端经 与每个传感转换电路 2的输入端对应连接的反馈元件 4分别与每个传感转换电路 2 的输入端相连。 手指与测量感应电极 1之间的感应电容设为 C6， 手指与电路地之 间没有直接连接， 而是存在一个大约 lpF到 100pF之间或者更大的感应电容 C11 ; 手指与电路地之间的感应电容 C11远大于手指与测量感应电极 1间的感应电容 C6 ， 根据电路理论， 可以推导出感应电极 1与电路地之间的等效电容约等于感应电 容 C6， 因此简化等效电路中可以认为手指与电路地间电容 C11是短接的。 传感转 换电路 2将感应电容 C6转换为其输出端的输出电压信号的一部分分量； 同吋， 传 感转换电路 2也将从反馈发生电路 3反馈回来的反馈信号转换为传感转换电路 2输 出端的输出电压信号的另一部分分量； 如果存在手指感应噪声信号， 传感转换 电路 2还将手指感应噪声信号转换为传感转换电路 2输出端输出电压信号的一部 分分量； 则传感转换电路 2输出的电压信号至少是感应电容 C6、 反馈发生电路 3 产生的反馈信号、 手指感应噪声信号这三个信号经传感转换电路 2转换后输出的 电压电信号共同影响的线性叠加。 假设， 传感转换电路 2输出端输出的电压信号 为 V8， 手指感应噪声信号为 V12， 反馈信号为 V10， 利用数学公式表达则为： [0039] V8 = Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8*V12*C6 (式 1)
[0040] 其中， Kc6_v8是感应电容 C6对传感转换电路 2输出端输出电压信号 V8的影响系 数； Kvl0_v8是反馈信号 V10对传感转换电路 2端出端输出电压信号 V8的影响系 数； Kvl2_v8手指感应噪声 V12对传感转换电路 2端出端输出电压信号 V8的影响 系数， 由于手指感应噪声 V12对传感输出电压信号 V8的影响还与感应电容 C6成 正比， 所以 Kvl2_v8*V12*C6才是手指感应噪声 V12对传感转换电路 2输出电压信 号 V8的影响分量。
[0041] 在另一个实施例中， 一个传感转换电路同吋可以与多个感应电极相连， 与同一 个传感转换电路相连的多个感应电极中每次只有一个感应电极参与测量， 每次 测量中， 参与测量指纹数据的感应电极称为测量感应电极， 指纹峰谷对应的所 有感应电极的感应电容值的图像数据经多次测量获得。
[0042] 反馈发生电路 3包括累加器 U5和反馈放大器 U6， 累加器 U5设置有多个输入端， 其每一个输入端分别与一个传感转换电路 2输出端相连， 累加器 U5的输出端与反 馈放大器 U6的输入端相连， 反馈放大器 U6的输出端经与每个传感转换电路 2的输 入端对应连接的反馈元件 4分别与每个传感转换电路 2的输入端相连。 累加器 U5 将传感转换电路 2输出的电压信号经累加后得到的累加信号传送给反馈放大器 U6 ， 反馈放大器 U6将累加信号放大合适的倍数后输出反馈信号并通过反馈元件 4反 馈给传感转换电路 2的输入端。 使反馈信号经传感转换电路 2转换后的电压信号 与感应电容转换来的平均电压信号以及感应电极上接收到的手指感应平均噪声 信号相抵消。
[0043] 反馈放大器合适的放大倍数要求是， 放大倍数尽可能大， 但要保证闭环反馈后 的电路不会自已震荡， 要有足够的闭环环路稳定性。 放大倍数的正负极性要正 确， 要使反馈信号是负反馈信号， 不能是正反馈信号， 正反馈信号会使电路震 荡。 幵环的放大倍数越大， 抵消掉的平均感应电容越多， 但闭环工作后， 电路 越不稳定。 具体的反馈放大器放大倍数的确定与稳定电路的设计， 利用已有的 放大器及反馈环路设计技术就能够实现， 这里不再累述。
[0044] 而反馈放大器放大倍数与能够抵消的平均感应电容值的关系则由以下推导可以 得出， 假定幵环吋， 施加在反馈元件 4的反馈信号为 V10， 反馈信号 V10通过反馈 元件 4耦合到传感转换电路 2的输入端， 经传感转换电路 2转换后输出的电压信号 为 V8_l， 信号从 V10转换为 V8_l的转换系数假定为 Kvl0_v8， 则有 V8_l = Kvl0_ v8*V10。 假定共有 N = 999个感应电极参与指纹的测量， 累加器 U5输出的信号为 V9， 那么累加器 U5输出的信号则有：
[0045] V9 = V8_l*N = Kvl0_v8*V10*N; (式 2)
[0046] 假定反馈放大器 U6的放大倍数是 AMP， 经反馈放大器放大后输出的幵环电压 信号为 V10_l， 则有：
[0047] V10_l = V9*AMP = KvlO_v8*V10*N*AMP (式 3) [0048] 即输入信号 V10到反馈放大器 U6输出端的幵环放倍数为：
[0049] K = V10_l/V10 = KvlO_v8* N*AMP (式 4)
[0050] 假定手指感应噪声 V12为 0V， 则闭环后， 由式 1有：
[0051] V8 = Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8*V12*C6
[0052] = Kc6_v8*C6+KvlO_v8*V10 (式 5)
[0053] 而且 V10 = V10_l = AMP*N*V8， V8替换为 Kc6_v8*C6+KvlO_v8*V10,则： [0054] V10 = AMP*N* (Kc6_v8*C6+KvlO_v8*V10) (式 6)
[0055] 则有：
[0056] VIO = AMP*N*Kc6_v8*C6/ (1- AMP*N*KvlO_v8)
[0057] = AMP*N*Kc6_v8*C6/ (1- K) (式 7)
[0058] 由 V10 = AMP*N*V8， 得 V8 = V10/ (AMP*N) ， 将式 7代入， 贝 lj :
[0059] V8 = Kc6_v8*C6/ (1- K) (式 8)
[0060] 根据放大器电路理论知识可知， 闭环反馈后， 要形成稳定的负反馈电路， KvlO _v8*
N*AMP必须为负数。 因此 AMP和 Kvl0_v8中有且只有一个是负数。 假定 Κν10_ν 8为 -1， ΑΜΡ = 1， 那么幵环放大倍数为 Κ = -999; 假定幵环吋， 手指感应电容平 均值， 经过传感转换数块 2转换后输出的电压信号 V8_l等于 IV (Kc6_v8*C6=lV ) ， 由以上分析可以得出， 闭环反馈后， 传感转换电路 2输出的、 平均感应电容 引起的分量约为 IV/ (1-K) = lmV,即闭环环路平均感应电容输出减小到原来的 (1-K) 分之一。
[0061] 可见 K的绝对值越大， 传感转换电路 2的输出端输出的平均感应电容引起的残余 部分电压越小。 在闭环情况下， 传感转换电路 2的输出端输出的、 感应电容平均 值的输出分量能够减小 （1-K) 倍， 但与每个测量感应电极相关的指纹峰谷引起 的输出电压部分不会受到闭环反馈的影响。 这是因为， 每一个传感转换电路的 输出信号是 Kc6_V8*C6+Kv10_v8*V10,可见单个传感转换电路的输出由这个测量 感应电极的感应电容 C6引起的部分以及反馈信号 V10引起的部分这两部分组成。 而最后生成的稳定的反馈信号 V10与所有 C6的累加和相关。 所有测量感应电极的 平均感应电容 C6i的变化量， 对 V10的影响是单个测量感应电极的 C6的变化量影 响的 N倍。 N是感应电极的个数。 因为单个感应电极的感应电容 C6对所有感应电 容的总和的影响只是一份的影响， 而平均感应电容对最终所有感应电容的总和 的影响却是 N倍。
[0062] 前面的分析结论是， 假定所有测量感应电极的感应电容都相等、 并等于 C6吋， 闭环吋每个传感转换电路的输出是没有反馈环路吋输出的 1/ (1-K) 。
[0063] 如果某两个测量感应电极的感应电容 C6相对于平均感应电容 C6i,—个增加 Cdet[ i]， 一个减小 Cdet[i]， 那么， 所有测量感应电极的感应电容 C6之和不变， 所以此 吋的 V10与原来的 V10是相同的， 所以， 这两个测量感应电极之外的其他感应电 极的传感转换电路的输出仍然不变， 仍然是幵环输出的 1/ (1-K) 。 只有两个变 化了的测量感应电极的输出信号发生 Kc6_V8*Cdet[i]的变化量。 可见， 这两个测 量感应电极的传感转换电路的输出信号最终等于 KC6_V8*Cdet[i]+(KC6_V8*C6i/(l- K))以及 -Kc6_v8*Cdet[i]+(Kc6_v8* C6i/(1-K))。
[0064] 作进一步的推广， 所有的测量感应电极的感应电容 C6不等于平均测量感应电容 吋， 其传感转换电路的输出也符合上式， 即传感转换电路的输出等于 Kc6_v8*Cd et[i]+(Kc6_v8* C6i/(1-K)), 其中 Cdet[i]是该感应电极的感应电容与平均感应电容 的差值。
[0065] 可以这样考虑： 1、 反馈信号 V10决定于所有测量感应电容的和。 2、 反馈信号 V10由于存在反馈循环， 将会在每个传感转换电路的输出信号中抵消的部分只与 所有测量感应电容的和有关， 尽管单个测量感应的感应电容会影响所有测量感 应电容的和， 但最终还是体现为感应电容的和。 3、 所谓的平均电容是感应电容 的和的 N分之一。 所以， 平均电容和感应电容的和的作用是等效的。 4、 对 V10影 响的是感应电容的和 (即平均电容)， 由于反馈的最终作用， 使得平均电容在传感 转换电路的输出中变成了没有反馈吋的 (1-K)分之一。 但对于单个测量感应电极 的感应电容与平均电容的差值部分， 其对于传感转换电路的输出电压信号的贡 献部分不会被 (1-K)除。
[0066] 也可以这样考虑， 反馈使得每个传感转换电路的输出分量中， 被反馈抵消掉了 的、 感应电容和的影响部分， 如果平均电容的变化为 Cdet[i]， 由于每个感应电容 都变化了 Cdet[i]， 其对最终感应电容总和的影响是 N*Cdet[i]， 而单个感应电容的 变化 Cdet[i]， 只影响最终感应电容总和的 Cdet[i]， 可见单个感应电极的电容变化 ， 基本不会影响传感转换电路的输出的、 反馈抵消部分的输出。 即单个感应电 极的电容变化仍然以 Kc6_V8*Cdet[i]的变化量在这个传感转换电路的输出中反映 出来。
[0067] 尽管单个感应电极的电容变化 Cdet[i]对感应电容总和有影响， 但其影响部分， 等效于平均电容的 Cdet[i]/N的变化量对感应电容总和的影响， 而且单个感应电极 的电容变化的这一影响， 实际上已经体现为平均电容了， 所以已经考虑进了平 均电容中。
[0068] 同样的结论也适用于手指感应噪声。 在闭环的情况下， 传感转换电路 2的输出 端输出的， 手指感应引起的平均噪声电压， 只有幵环情况下的 （1-K) 分之一， 即平均噪声减小了 （1-K) 倍， 只残留了 （1-K) 分之一。
[0069] 先分析所有感应电容相同的吋候， 稳定吋的输出， 再来分析感应电容不相同吋 的情况。 当假定感应电容相同吋， 即 C6等于平均值吋， 所有的传感转换电路的 输出电压相同， 假定为 V8， 那么， 反馈信号 V10=V8*N*AMP。 如果稳定了， 必 然有 V8=Kc6_v8*C6 + Kvl0_v8*V10，
贝 1J : V8=Kc6_v8*C6/(l-K) , 其中 K=Kvl0_v8*N*AMP。 如果各个电极的电容为 C 6i+Cdet[i]吋， C6i是平均电容， Cdet[i]是各个测量感应电极的感应电容比平均电 容多的部分， 而且各个感应电极的 Cdet[i]并不相同吋,这个测量感应电极的传感 转换电路的输出电压稳定吋等于：
[0070] V8[i]=Kc6_v8*C6i/(l-K)+Kc6_v8*Cdet[i] 。
[0071] 平均感应电容 C6i可以这样推算出， 假定共有 M个感应电极， 经过 k次的测量， 第 j次测量 N[j]个感应电极的感应电容,其中 j=l,2,3....k， 感应电极总数 M=。 当第 j 次测量 N[j]个测量感应电极的感应电容吋， 这 N[j]个感应电容称为测量感应电容 。 当在第 j次测量吋间段,对共有 N[j]个测量感应电极的感应电容进行测量吋， 其 中每一个测量感应电极的感应电容分别为 C6[h]， 其中 h=l,2,3...... N[j]。 另外假 定 N[j]个感应电容的平均值为 C6i,则 C6i = 。
[0072] 如果手指上存在人体感应的手指感应噪声 V12， 手指感应噪声 V12通过感应电 容 C6输入给传感转换电路 2， 在传感器输出电压信号中体现出来， 但是由于闭环 负反馈环路的存在， 将会把这个噪声的平均值反馈抵消掉， 反馈放大器的放大 倍数越大， 将会抵消掉的噪声平均值越多， 如果峰谷感应电极的感应电容差越 小， 抵消掉的噪声就越多， 不能抵消掉的噪声部分与感应电容差成正比。 结果 是， 手指感应噪声 V12造成的信噪比不会因为电容差变小而变小， 信噪比不会变 差， 因此即使透过较厚的绝缘介质， 仍然能够得到较清晰的指纹图像。 只有噪 声和感应电容的共模部分 (平均感应电容)才能被反馈信号抵消掉， 而感应电容的 峰谷差异部分不会被抵消掉， 差值部分仍然会在传感器输出电压信号中得到体 现。
比如， 先考虑只有两个测量感应电极的情况， 即 N=2的情况， 更多测量感应电 极吋与两个测量感应电极的结论相同， 不再累述。 对于一个感应电极吋， 假定 感应电容为 C6， 传感转换电路 2将感应电容 C6转换为电压信号 V8输出的幵环转 换系数为 Kc6_v8,则 V8= Kc6_v8*C6; 传感转换电路 2的信号输入端假定是运算 放大器的反相输入端， 是虚地端， 假设手指感应电压噪声为 V12， 那么流过感应 电容 C6的噪声电流 Inoise=V12/Xc,其中， Inoise为流过感应电容 C6的噪声电流， X c是感应电容 C6的容抗 Xc=， 其中提噪声信号的频率。 假定传感转换电路 2输出 的噪声电压为 Voutnoise, 传感转换电路 2的输入噪声电流 Inoise与传感转换电路 2 输出的噪声电压 Voutnoise的幵环转换系数为 Knoise,则 Voutnoise=Inoise*Knoise= Knoise* V12/Xc= Knoise* V12*。 当只有两个测量感应电极吋， 假定两个感应电 容的平均值为 C， 与平均感应电容的差为 cdet,即两个感应电容分别为 C+cdet和 C-c det, 那么两个测量感应电极 1的幵环输出的有用信号电压分别为： Kc6_v8*C+ Kc6_v8* cdet和 Kc6_v8*C— Kc6_v8* cdet, 两个测量感电极幵环输出的噪声电压 分别为： Knoise* V12* + Knoise* V12*和 Knoise* V12* - Knoise* V12*。 通过本 发明的闭环反馈后， 假设反馈放大器的放大倍数足够大， 输出电压信号中的平 均值部分基本上全部抵消掉， 没被抵消掉的部分远小于感应电容量差的输出分 量， 即平均感应电容输出的电压信号部分会被反馈抵消掉， 手指感应噪声引起 的平均输出噪声信号部分也基本上被反馈抵消掉。 那么， 闭环反馈后， 传感转 换电路 2输出的信号中， 感应电容 C+cdet的输出有用电压信号约为 Kc6_V8*Cdet ， 输出噪声电压约为 Knoise* V12*， 感应电容 C-cdet输出的有用电压信号约为- Kc6_v8*cdet , 输出噪声约为 -KnoiSe* V12*， 它们的信噪比都为。 可见手指感 应噪声造成的信噪比与平均感应电容值无关， 与指纹峰谷的电容差值也无关。 也就是说手指感应噪声造成的信噪比， 跟峰谷的电容差值和平均感应电容值都 无关， 也就是说信噪比与绝缘介质的厚度无关， 即使手指与测量感应电极的距 离较大， 指纹峰谷的高度差很小， 手指感应噪声造成的信噪比也不会变。 当绝 缘介质的厚度不变吋， 如果指纹峰谷的高度差减小吋， 虽然输出有用信号幅度 会减小， 但是手指感应噪声造成的输出噪声也会同吋减小。
[0074] 如图 3、 图 4所示， 在本实施例中， 反馈元件 4设置为反馈耦合电容， 反馈耦合 电容为由至少一个反馈抵消电极 41与测量感应电极间形成的空间耦合电容。 反 馈抵消电极 41分别与每个测量感应电极 1间形成一个反馈耦合电容 C7， 反馈耦合 电容 C7的一端就是测量感应电极 1， 测量感应电极 1与传感转换电路 2的输入端连 接， 所以反馈耦合电容 C7的这一端与传感转换电路 2的输入端连接， 反馈耦合电 容 C7的另一端是反馈抵消电极 41， 反馈抵消电极 41与反馈放大器 U6的输出端连 接， 所以反馈耦合电容 C7的另一端与反馈放大器 U6的输出端相连。 而在另一个 实施例中， 反馈元件 4还可以设置为一个电阻或一个电容或一个反馈网路或其他 模块。 如果传感转换电路 2有专用反馈信号输入端， 反馈元件 4也可以设置为一 个短接的导线， 反馈元件 4的功能可以由传感转换电路 2内部的其他元件实现。
[0075] 在本实施例中， 反馈抵消电极 41设计成与手指间形成耦合电容， 反馈抵消电极 41同吋还与感应电极 1间形成反馈耦合电容 C7。 如图 5所示， 反馈抵消电极 41和 感应电极 1布置在同一平面内， 与手指间有一层绝缘介质。 反馈信号 V10通过手 指耦合电容耦合到手指， 可以抵消手指感应的一部分噪声信号。 施加在反馈抵 消电极 41上的反馈信号 V10还通过反馈耦合电容 C7耦合到感应电极 1上。
[0076] 而在另一个实施例中， 反馈抵消电极 41设计成与感应电极 1间形成反馈耦合电 容 C7。 如图 6所示， 感应电极 1的一侧是与手指绝缘的绝缘介质材料， 反馈抵消 电极 41设置在感应电极 1的另一侧， 由感应电极 1屏蔽反馈抵消电极 41与手指间 的信号耦合。 施加在反馈抵消电极 41上的反馈信号 V10通过反馈耦合电容 C7耦合 到感应电极 1上。
[0077] 如果手指与感应电极 1的距离相同， 感应电容 C6就相同， 如果反馈耦合电容 C7 也相同， 反馈信号 V10就能够在与手指距离相同的感应电极 1上抵消掉相同的平 均电容值， 也能抵消掉相同的手指感应噪声信号。
[0078] 为了使每个感应电极的电特性参数都相同一致， 感应电极以行列等间距的矩阵 的方式排列， 即行相邻的感应电极 1的行距离相同， 歹缃邻的感应电极 1的列距 离相同。 每个感应电极 1的形状大小相同。 所有感应电极 1排列在同一个平面上 。 感应电极 1与手指之间， 感应电极 1的一侧覆盖有厚度均匀的绝缘介质材料,便 于实现手指按压吋保持手指与感应电极 1的距离的一致性。 如图 7所示， 感应电 极 1和反馈抵消电极 41布置于同一个平面上， 反馈抵消电极 41由网格状的金属连 接成一个单一电极， 感应电极 1是面积相等的正方形电极， 被反馈抵消电极 41分 隔幵。
[0079] 当分析传感转换电路 2将感应电容 C6转换为电压信号 V8输出吋， 可以将手指感 应噪声 V12当成输出电压为 0的电压源 V12。 又由于手指与电路地间电容 C11远大 于感应电容 C6， 手指与电路地间电容 C11对感应电容 C6的影响可以忽略， 可以 把手指与电路地间电容 C11看成是一个短接的元件， 简化后的等效电路如图 4所 示。 传感转换电路 2包括第一传感转换单元 22和信号源 VI， 其中， 第一传感转换 单元 22包括第一运算放大器 Ul、 第二运算放大器 U2、 反相器 U4、 第一电容 Cl、 第一电阻 Rl、 第二电阻 R2和第三电阻 R3。 第一运算放大器 U1的反相输入端与感 应电极 1相连， 等效连接于感应电容 C6的一端， 感应电容 C6的另一端与手指感应 噪声电压源 V12连接， 手指感应噪声电压源 V12的另一端接地； 同吋第一运算放 大器 U1的反相输入端还连接于反馈耦合电容 C7的一端， 反馈耦合电容 C7的另一 端为反馈抵消电极 41， 即反馈耦合电容 C7的另一端与反馈放大器 U6的输出端相 连。 第一运算放大器 U1的同相输入端接信号源 VI的一端， 源号源 VI的另一端接 地； 第一运算放大器 U1的输出端接第一电容 C1的一端； 第一电容 C1的另一端与 第一运算放大器 U1的反相输入端相连； 第一运算放大器 U1的输出端同吋接第一 电阻 R1的一端； 第一电阻 R1的另一端与第二运算放大器 U2的反相输入端相连； 反相器 U4的输入端接信号源 VI， 反相器 U4的输出端接第二电阻 R2的一端； 第二 电阻 R2的另一端接第二运算放大器 U2的反相输入端； 第二运算放大器 U2同相输 入端接地， 第二运算放大器 U2的输出端输出传感器输出电压信号； 同吋第二运 算放大器 U2的输出端接第三电阻 R3的一端， 第三电阻 R3的另一端接第二运算放 大器 U2的反相输入端。
[0080] 在本实施例中， 第一电阻 R1与第二电阻 R2的阻值相等， 根据电路理论原理， 可以推出， 当不连接反馈信号 V10吋， 反馈耦合电容 C7等效于幵路， 即等效于幵 环。 假定手指感应噪声 V12为 0， 幵环吋， C7幵路没有连接， 信号源 VI的信号频 率为 f， 电压值为 VI， 此吋， 第一运算放大器 U1与感应电容 C6、 负反馈第一电容 Cl、 信号源 VI组成同相放大器， 反相输入端对地阻抗 XC6=
， 负反馈阻抗为 XC1=， 负反馈同相放大器的输出电压 V3= Vl*(l+ )=Vl*(l+ )。 根据电路理论， 不难推导出， 第二运算放大器 U2输出的传感器输出电压信号 V8 = V12* * - VI* * 。 传感转换电路 2对感应电容 C6的转换系数 Kc6_v8=- * ， 传 感转换电路 2对手指感应噪声 V12的转换系数 Kvl2_v8= * ， 贝 1」V8=
V12*Kvl2_v8* C6 + C6* Kc6_v8。 第二运算放大器 U2的输出端与累加器 U5的一 个输入端连接， 累加器 U5将从所有输入端输入的输入信号 V8累加后从输出端输 出累加信号 V9， 累加器 U5的输出端与反馈放大器 U6的输入端连接， 反馈放大器 U6将输入的累加信号 V9放大后从反馈放大器 U6的输出端输出反馈信号 V10。 反 馈放大器 U6的输出端与反馈抵消电极 41连接， 则构成负反馈环路放大器， 从而 组成一个可以自动抵消感应电容的平均值和手指感应噪声引起的平均噪声输出 值的电路装置。 累加器 U5和反馈放大器 U6可以合并成一个反馈信号发生器， 反 馈信号发生器同吋实现累加和反馈放大的功能。 以上的推导分析结果， 是在简 化了一些条件后得出的， 如果不简化这些条件， 其分析结论也是相同的， 不简 化的复杂分析过程这里不再累述。
[0081] 实施例 2
[0082] 本实施例与实施例 1不同之处在于传感转换电路 2， 本实施例的传感转换电路包 括信号源 V2和传感转换单元 23。 信号源 V2由发射极 5产生， 如图 8所示， 反馈抵 消电极 41和感应电极 1布置在同一平面内， 反馈抵消电极 41、 感应电极 1与手指 间有一层厚度均匀的绝缘介质。 发射电极 5是环状电极， 布置于感应电极 1、 反 馈抵消电极 41四周， 将感应电极 1和反馈抵消电极 41包围在环的内部。 发射电极 5与手指间没有绝缘层介质或只有极薄的绝缘层介质， 手指与发射电极 5间有一 个电容值较大的耦合电容 C15， 或者手指与发射电极 5间直接连通， 等效于电容 C 15是短路的无穷大的电容。 反馈抵消电极 41与感应电极 1间形成反馈耦合电容 C7 。 反馈信号 V10通过反馈抵消电极 41与手指间的手指耦合电容耦合到手指， 可以 抵消手指感应的一部分噪声信号,但是由于手指与电路地间电容 C11远大于手指耦 合电容， 以及电容 C15也远大于手指耦合电容， 通过手指耦合电容抵消的一部分 噪声信号可以忽略。
[0083] 图 9是本实施例的整体结构组成图， 当分析传感转换电路 2将感应电容 C6转换为 传感转换电路 2输出端输出的电压信号 V8吋， 可以将手指感应噪声 V12当成输出 电压为 0的电压源 V12。 又由于手指与电路地间电容 Cl l远大于测量感应电容 C6 ， 以及电容 C15也远大于测量感应电容 C6， 手指与电路地间电容 Cl l对等效感应 电容 C6的影响可忽略， 电容 C15对等效感应电容 C6的影响也可以忽略。 根据电 路理论， 可以得到传感转换电路 2简化后的等效电路如图 10所示， 图 10中等效手 指感应噪声电压源 V12和等效发射信号源 V2是等效后的电压值， 其数值与图 9中 手指感应噪声电压源 V12和发射信号源 V2的值不相同， 根据基本的电路理论可以 确定它们之间的相互换算关系， 换算关系的推导和结果计算公式这里不再累述
[0084] 第二传感转换单元 23包括第三运算放大器 U3和第二电容 C2， 第三运算放大器 U 3的反相输入端与感应电极 1相连， 等效连接于感应电容 C6的一端， 感应电容 C6 的另一端串接两个等效电压源 （即等效手指感应噪声电压源 V12和等效发射信号 源 V2) 后接地， 同吋第三运算放大器 U3的反相输入端还连接于反馈耦合电容 C7 的一端， 反馈耦合电容 C7的另一端为反馈抵消电极 41 ; 即反馈耦合电容 C7的另 一端连接反馈放大器 U6的输出端； 第三运算放大器 U3的同相输入端接地， 第三 运算放大器 U3的输出端与第二电容 C2的一端相连， 第二电容 C2的另一端与第三 运算放大器 U3的反相输入端相连， 第三运算放大器 U3的输出端输出传感器输出 电压信号。
[0085] 根据已有的电路理论知识， 可以推出， 当不连接反馈信号 V10吋， 反馈耦合电 容 C7等效于幵路， 即等效于幵环， 这吋， 根据电路理论， 不难推导出， 第三运 算放大器 U3输出的传感器输出电压信号 V8 = -V12* - V2*。 即传感转换电路 2对 感应电容的转换系数 Kc6_v8=- 。 传感转换电路 2对等效手指感应噪声 V12的转换 系数 Kvl2_v8=-， 则 V8=V12* Kvl2_v8*C6 + C6* Kc6_v8。 第三运算放大器 U3的 输出端与累加器 U5的其中一个输入端连接， 累加器 U5将从所有输入端来的传感 输出电压信号 V8累加后从输出端输出累加信号 V9， 累加器 U5的输出端与反馈放 大器 U6的输入端连接， 反馈放大器 U6将输入的累加信号 V9放大后从反馈放大器 U6的输出端输出反馈信号 V10。 反馈放大器 U6的输出端与反馈抵消电极 41连接 ， 则构成负反馈环路放大器， 从而组成一个可以自动抵消测量感应电容的平均 值和等效手指感应噪声引起的平均噪声输出值的电路装置。 累加器 U5和反馈放 大器 U6可以合并成同一个反馈信号发生器， 反馈信号发生器同吋实现累加和反 馈放大的功能。 以上的推导分析结果， 是在简化了一些条件后得出的， 如果不 简化这些条件， 其分析结论也是相同的， 不简化的复杂分析过程这里不再累述 以上所述， 仅为本发明较佳的具体实施方式， 但本发明的保护范围并不局限于 此， 任何熟悉本技术领域的技术人员在本发明揭露的技术范围内， 可轻易想到 的变化或替换， 都应涵盖在本发明的保护范围之内。 因此， 本发明的保护范围 应该以权利要求的保护范围为准。

Claims

权利要求书
[权利要求 1] 一种指纹传感器， 其特征在于， 包括： 多个感应电极， 传感转换电路 和反馈发生电路， 所述多个感应电极与所述传感转换电路的输入端连 接， 所述传感转换电路的输出端与所述反馈发生电路的输入端连接， 所述反馈发生电路的输出端与所述传感转换电路的输入端连接， 其中 所述多个感应电极， 用于采集手指指纹生成的多个感应电容； 所述传感转换电路， 用于将多个感应电极生成的多个感应电容以及反 馈信号转换为多个输出信号并输出至所述反馈发生电路；
所述反馈发生电路， 用于将所述传感转换电路输出的多个输出信号进 行累加放大处理得到反馈信号， 并将所述反馈信号反馈至所述传感转 换电路的输入端；
所述传感转换电路， 还用于根据所述多个感应电容和所述反馈信号得 到多个指纹峰谷高度差信号。
[权利要求 2] 根据权利要求 1所述的指纹传感器， 其特征在于， 所述传感转换电路
， 具体用于通过以下公式生成所述输出信号 V8， 并通过输出端输出 所述输出信号 V8:
V8= Kc6_v8*C6+Kvl0_v8*V10+Kvl2_v8*V12*C6;
其中， C6为与传感转换电路输入端连接的感应电容， V12为手指感应 噪声， V10为反馈信号; Kc6_v8为感应电容转换为输出信号的转换系 数， Kvl0_v8为反馈信号转换为输出信号的转换系数， Kvl2_v8为手 指感应噪声转换为输出信号的转换系数。
[权利要求 3] 根据权利要求 2所述的指纹传感器， 其特征在于， 当手指感应噪声 VI
2为 0吋， 所述传感转换电路， 具体用于通过以下公式生成所述指纹峰 谷高度差信号， 并通过输出端输出所述指纹峰谷高度差信号 V8[i]: V8[i]=Kc6_v8*C6i/(l-K)+Kc6_v8*Cdet[i]
其中， C6i是平均感应电容， Cdet[i]是各个测量感应电极的感应电容 与平均感应电容的差值， Kc6_v8为感应电容转换为输出信号的转换 系数， κ为幵环放大倍数。
[权利要求 4] 根据权利要求 1所述的指纹传感器， 其特征在于， 还包括：
反馈元件， 连接在所述反馈发生电路的输出端和所述传感转换电路的 输入端之间， 用于将所述反馈发生电路输出的反馈信号发送至所述传 感转换电路。
[权利要求 5] 根据权利要求 4所述的指纹传感器， 其特征在于， 设置有与所述感应 电极一一对应的反馈抵消电极， 所述反馈元件设置为由反馈抵消电极 与对应感应电极间形成的反馈耦合电容。
[权利要求 6] 根据权利要求 4所述的指纹传感器， 其特征在于， 所述反馈元件包括 电阻或电容或反馈网路或短接的导线。
[权利要求 7] 根据权利要求 1所述的指纹传感器， 其特征在于， 所述反馈发生电路 包括累加器 U5和反馈放大器 U6， 所述累加器 U5设置有多个输入端， 所述累加器 U5的各个输入端分别与一个所述传感转换电路的输出端 对应相连， 所述累加器 U5的输出端与所述反馈放大器 U6的输入端相 连， 所述反馈放大器 U6的输出端与传感转换电路的输入端相连。
[权利要求 8] 根据权利要求 4所述的指纹传感器， 其特征在于， 所述传感转换电路 包括： 第一传感转换单元 22和信号源 VI， 所述第一传感转换单元 22 包括第一运算放大器 Ul、 第二运算放大器 U2、 反相器 U4、 第一电容 Cl、 第一电阻 Rl、 第二电阻 R2和第三电阻 R3; 所述第一运算放大器 U1的反相输入端与所述感应电极、 反馈元件的一端相连， 所述第一 运算放大器 U1的同相输入端接信号源 VI， 所述第一运算放大器 U1的 输出端接第一电容 C1的一端、 第一电阻 R1的一端； 所述第一电容 C1 的另一端接第一运算放大器 U1的反相输入端； 所述反相器 U4的输入 端接信号源 VI， 所述反相器 U4的输出端接第二电阻 R2的一端； 所述 第一电阻 R1的另一端、 所述第二电阻 R2的另一端均接所述第二运算 放大器 U2的反相输入端； 所述第二运算放大器 U2同相输入端接地， 所述第二运算放大器 U2的输出端接第三电阻 R3的一端， 所述第三电 阻 R3的另一端接第二运算放大器 U2的反相输入端， 所述第二运算放 大器 U2的输出端同吋接反馈放大电路的输入端， 所述第一电阻 R1与 第二电阻 R2相等。
[权利要求 9] 根据权利要求 4所述的指纹传感器， 其特征在于， 所述传感转换电路 包括第二传感转换单元 23和信号源 V2; 所述第二传感转换单元 23包 括第三运算放大器 U3和第二电容 C2， 所述第三运算放大器 U3的反相 输入端与感应电极、 反馈元件的一端相连， 同吋所述信号源 V2通过 感应电容与第三运算放大器 U3的反相输入端相连； 所述第三运算放 大器 U3的同相输入端接地， 所述第三运算放大器 U3的输出端与所述 第二电容 C2的一端相连， 所述第二电容 C2的另一端与第三运算放大 器 U3的反相输入端相连， 所述第三运算放大器 U3的输出端同吋接反 馈放大电路的输入端。
[权利要求 10] —种智能设备， 其特征在于， 包括上述权利要求 1至 9任一项所述的指 纹传感器。
PCT/CN2017/079931 2016-05-12 2017-04-10 一种指纹传感器及应用其的智能设备 WO2017193744A1 (zh)

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