WO2017174004A1 - 载波同步方法和装置 - Google Patents

载波同步方法和装置 Download PDF

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WO2017174004A1
WO2017174004A1 PCT/CN2017/079581 CN2017079581W WO2017174004A1 WO 2017174004 A1 WO2017174004 A1 WO 2017174004A1 CN 2017079581 W CN2017079581 W CN 2017079581W WO 2017174004 A1 WO2017174004 A1 WO 2017174004A1
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Prior art keywords
las
frequency offset
received signal
carrier
codes
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PCT/CN2017/079581
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English (en)
French (fr)
Inventor
刘若鹏
季春霖
吕长伟
张莎莎
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深圳超级数据链技术有限公司
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Priority claimed from CN201610216716.4A external-priority patent/CN107276948B/zh
Priority claimed from CN201610219188.8A external-priority patent/CN107294891A/zh
Priority claimed from CN201610219112.5A external-priority patent/CN107276956B/zh
Priority claimed from CN201610214970.0A external-priority patent/CN107294889B/zh
Priority claimed from CN201610216953.0A external-priority patent/CN107276951B/zh
Priority claimed from CN201610216455.6A external-priority patent/CN107294890B/zh
Priority claimed from CN201610216433.XA external-priority patent/CN107276946B/zh
Priority claimed from CN201610217038.3A external-priority patent/CN107276952B/zh
Priority claimed from CN201610216879.2A external-priority patent/CN107276950A/zh
Application filed by 深圳超级数据链技术有限公司 filed Critical 深圳超级数据链技术有限公司
Publication of WO2017174004A1 publication Critical patent/WO2017174004A1/zh

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes

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  • the present invention relates to a wireless communication system, and more particularly to a carrier synchronization method and apparatus.
  • Wireless communication networks are widely deployed to provide various communication services such as voice, video, packet data, messaging, broadcast, and the like. These wireless networks may be multiple-access networks capable of supporting multiple users by sharing available network resources. Examples of such multiple access networks include Code Division Multiple Access (CDMA) networks, Time Division Multiple Access (TDMA) networks, Frequency Division Multiple Access (FDMA) networks, Orthogonal FDMA (OFDMA) networks, and Single Carrier FDMA (SC-FDMA). )The internet.
  • CDMA Code Division Multiple Access
  • TDMA Time Division Multiple Access
  • FDMA Frequency Division Multiple Access
  • OFDMA Orthogonal FDMA
  • SC-FDMA Single Carrier FDMA
  • the Overlapped Time Division Multiplexing (OvTDM) system is such a solution to improve the spectrum efficiency of the system.
  • OvTDM Time Division Multiplexing
  • the OvTDM system not only do not need to be isolated from each other, but also have strong mutual overlap.
  • the OvTDM system significantly increases the spectrum utilization by artificially introducing overlaps between symbols and using multiple symbols to transmit data sequences in parallel in the time domain.
  • Overlapped Frequency Division Multiplexing (OvFDM) system is another solution to improve the spectrum efficiency of the system.
  • OvFDM Orthogonal Frequency Division Multiplexing
  • the spectrum utilization is further improved on the basis of the OFDM system by the higher degree of overlap between the sub-bands in the frequency domain.
  • a carrier synchronization method including:
  • Frequency offset correction is performed on the received signal based on the frequency offset.
  • Another aspect of the present invention provides a carrier synchronization apparatus, including:
  • a cross-correlation calculation unit configured to perform a cross-correlation operation on two training codes extracted from the received signal to obtain a frequency offset of a carrier between the receiving end and the transmitting end, wherein the received signal is from the transmitting end and includes a training code based Training sequence;
  • a frequency correction unit configured to perform frequency offset correction on the received signal based on the frequency offset.
  • the present invention has the following beneficial effects: the present invention designs a training sequence in a communication system, and utilizes a training code autocorrelation function to be an ideal impact function at the origin, where the origin is zero and the cross-correlation function is zero.
  • This training code is used for signal synchronization in signal reception processing.
  • the training code is used for carrier synchronization, the frequency offset accuracy of the system is improved by two synchronization processes, which lays a foundation for the subsequent channel estimation process and decoding process, and reduces the bit error rate of the whole system.
  • Figure 1 shows a block diagram of a transmitter modulation module of an OvTDM system
  • Figure 2 shows a block diagram of a signal pre-processing module at the receiving end of the OvTDM system
  • Figure 3 is a block diagram showing the receiving end sequence detecting module of the OvTDM system
  • Figure 4 shows a block diagram of a modulation module at the transmitting end of the OvFDM system
  • Figure 5 is a block diagram showing a signal pre-processing module at the receiving end of the OvFDM system
  • Figure 6 is a block diagram showing a signal detecting module at the receiving end of the OvFDM system
  • Figure 7 shows the autocorrelation properties of the M sequence
  • Figure 8 shows the autocorrelation properties of the LAS code
  • Figure 9 shows a distribution diagram of autocorrelation results of timing synchronization
  • Figure 10 shows a schematic diagram of a training sequence in the case where two peaks are detected
  • FIG. 11 is a block diagram showing a timing synchronization unit of a receiving end according to an aspect of the present invention.
  • FIG. 12 shows a flow chart of a timing synchronization method in accordance with an aspect of the present invention
  • Figure 13 shows a block diagram of a carrier synchronization unit in accordance with an aspect of the present invention
  • FIG. 14 shows a flow chart of a carrier synchronization method in accordance with an aspect of the present invention
  • Figure 16 is a diagram showing the arrangement of multipath channels
  • Figure 17 is a diagram showing the relationship between bandwidth and power spectral density of training sequences and data in accordance with an aspect of the present invention.
  • FIG. 18 is a schematic diagram showing a frequency spectrum when two carrier signals simultaneously transmit data according to an aspect of the present invention.
  • FIG. 19 is a functional block diagram of the OvCDM system
  • 20 is a schematic diagram of an encoder of an OvCDM
  • Figure 21 is a schematic diagram of the system of OvHDM.
  • the techniques described in this paper can also be widely used in practical mobile communication systems, such as TD-LTE, TD-SCDMA, etc., and can also be widely used in satellite communications.
  • network and “system” are often used interchangeably.
  • OvTDM OvTDM
  • OvFDM OvFDM
  • OvCDM OvHDM
  • OvHDM OvHDM
  • the OvTDM system uses multiple symbols to transmit data sequences in parallel in the time domain.
  • a transmitting signal in which a plurality of symbols overlap each other in the time domain is formed at the transmitting end, and the received signal is detected by the data sequence in the time domain according to a one-to-one correspondence between the transmitted data sequence and the time waveform of the transmitted data sequence.
  • the OvTDM system actively exploits these overlaps to create coding constraints, which greatly increases the spectral efficiency of the system.
  • FIG. 1 shows a block diagram of the transmit modulation module of the OvTDM system.
  • the transmitting end modulation module 100 may include a digital waveform generating unit 110, a shift register unit 120, a multiplying unit 130, and an adding unit 140.
  • the first modulated signal envelope waveform h(t) of the generated signal is digitally designed by the digital waveform generating unit 110, and the shift register unit 120 shifts the envelope waveform h(t) for a specific time to form The envelope waveform h (ti ⁇ ⁇ T) of the modulated signal at each other time, the multiplication unit 130 multiplies the parallel symbol x i to be transmitted and the envelope waveform h (ti ⁇ ⁇ T) at the corresponding time, and obtains the modulated time at each time.
  • the waveform of the signal to be transmitted x i h(ti ⁇ ⁇ T).
  • the addition unit 140 superimposes the formed waveforms to be transmitted to form a transmission signal waveform.
  • the receiving end of the OvTDM system is mainly divided into a signal pre-processing module 200 and a sequence detecting module 300.
  • Figure 2 shows a block diagram of the signal pre-processing module 200 at the receiving end of the OvTDM system.
  • the signal pre-processing module is operative to assist in forming a sequence of synchronized received digital signals within each frame.
  • the signal pre-processing module can include a synchronization unit 210, a channel estimation unit 220, and a digitization processing unit 230.
  • the synchronization unit 210 is configured to form symbol synchronization in the time domain for the received signal to maintain a synchronization state with the system, mainly including timing synchronization and carrier synchronization.
  • the channel estimation unit 220 performs channel estimation on the received signal. Calculated to estimate the parameters of the actual transmission channel.
  • the digitization processing unit 230 is configured to digitize the received signal in each frame to form a received digital signal sequence suitable for sequence detection by the sequence detecting portion.
  • the received signal may be sequence-detected in the sequence detecting module 300, the received waveform is cut according to the waveform transmission time interval, and the cut waveform is decoded according to a certain decoding algorithm.
  • Figure 3 shows a block diagram of the receiver sequencing module of the OvTDM system.
  • the sequence detection module 300 can include an analysis storage unit 310, a comparison unit 320, and a retention path storage unit and an Euclidean distance storage unit 330.
  • the analysis storage unit makes a complex convolutional coding model and a trellis diagram of the OvTDM system, and lists all states of the OvTDM system and stores them.
  • the comparison unit searches for a path with a minimum Euclidean distance of the received digital signal according to the trellis diagram in the analysis storage unit, and the reserved path storage unit and the Euclidean distance storage unit are respectively used to store the reserved path and the Euclidean output of the comparison unit. Distance or weighted Euclidean distance.
  • the reserved path storage unit and the Euclidean distance storage unit need to be prepared for each stable state.
  • the reserved path storage unit length may preferably be 4K to 5K.
  • the Euclidean distance storage unit preferably stores only relative distances.
  • FIG. 4 shows a block diagram of the modulation module at the transmitting end of the OvFDM system.
  • the OvFDM modulation module at the transmitting end may include a modulated carrier spectrum generating unit 410, a carrier spectrum shifting unit 420, a multiplying unit 430, an adding unit 440, and an inverse Fourier transform unit 450.
  • the modulated carrier spectrum generating unit 410 is designed to generate an envelope spectrum signal H(f) of one subcarrier, and the carrier spectrum shifting unit 420 sequentially shifts the envelope spectral signal H(f) by a specific carrier spectral interval ⁇ B.
  • the envelope spectrum signal of the next subcarrier is output, and the envelope spectrum signal of the next subcarrier is frequency-shifted by ⁇ B, and the spectrum waveform H (fi ⁇ B) of all subcarriers with the spectral interval ⁇ B is sequentially obtained.
  • the multiplying unit 430 multiplies the multiplexed symbols X i to be transmitted respectively with the generated corresponding subcarrier spectral waveforms H (fi ⁇ B) to obtain a modulated signal spectrum X i H multiplexed by the corresponding subcarriers ( Fi ⁇ ⁇ B).
  • the receiving end of the OvFDM system is mainly divided into a signal pre-processing module 500 and a signal detecting module 600.
  • Figure 5 shows a block diagram of the signal pre-processing module at the receiving end of the OvFDM system.
  • the pre-processing module can include a synchronization unit 510, a channel estimation unit 520, and a digitization processing unit 530.
  • the synchronization unit 510 is configured to form symbol synchronization in the time domain for the received signal to maintain a synchronization state with the system, mainly including timing synchronization and carrier synchronization.
  • the channel estimation unit 520 performs channel estimation on the received signal for estimating the parameters of the actual transmission channel.
  • the digitization processing unit 530 is configured to time intervals of each symbol The received signal is sampled and quantized to become a digital signal sequence.
  • the received signal can be detected in the signal detection module 600.
  • Figure 6 shows a block diagram of the signal detection module 600 at the receiving end of the OvFDM system.
  • the signal detection module 600 can include a Fourier transform unit 610, a frequency segmentation unit 620, a convolutional encoding unit 630, and a data detection unit 640.
  • the Fourier transform unit 610 is configured to convert the preprocessed time domain signal into a frequency domain signal, that is, perform a Fourier transform on the received digital signal sequence for each time symbol interval to form an actual received signal spectrum for each time symbol interval.
  • the frequency segmentation unit 620 is configured to segment the actual received signal spectrum of each time symbol interval in the frequency domain by a spectral interval ⁇ B to form an actual received signal segmentation spectrum.
  • the convolutional coding unit 630 is configured to form a one-to-one correspondence between the received signal spectrum and the transmitted data symbol sequence.
  • the data detecting unit 640 is configured to detect a sequence of data symbols according to a one-to-one correspondence formed by the convolutional coding unit.
  • the above describes the processing of the transmitting and receiving ends of the OvTDM system and the OvFDM system.
  • the above OvTDM system and OvFDM system have corresponding receiving and demodulating schemes to eliminate the interference caused by the overlapping of signals in the time domain or the frequency domain, the substantial increase in spectrum utilization still imposes higher requirements on signal reception.
  • the present invention further provides an OvCDM system (Overlapping Code Division Multiplexing).
  • the OvCDM can be regarded as a parallel complex domain convolutional code.
  • the system function diagram is as shown in FIG.
  • the encoder structure is as shown in FIG.
  • the key of the OvCDM system is the coding matrix, that is, the convolutional expansion coefficient.
  • the coding matrix is arranged as follows:
  • the OvCDM encoding process is as follows:
  • u i u i,0 u i,1 u i,2 ...
  • u 0 u 0,0 u 0,2 u 0,4 ...
  • u 1 u 1,1 u 1,3 u 1,5 ...
  • Each channel of data is sent to a shift register for weighted superposition.
  • OvCDM code rate Where n is the length of the substream. When n is long, the bit rate loss caused by the tailing of the shift register is negligible, so there is r OvCDM ⁇ k.
  • the code rate is generally less than 1, resulting in loss of spectral efficiency.
  • the complex coded convolutional code rate is equal to 1, a one-way convolutional coding extension does not result in loss of spectral efficiency and additional coding gain.
  • the OvHDM system (Overlapped Hybrid Division Multiplexing) and the real-time two-dimensional overlapping multiplexing system are further evolved. It not only overlaps the frame symbols in the time domain, but also overlaps the subcarriers in the frequency domain, achieving simultaneous overlapping of the time domain and the frequency domain.
  • OvHDM The system block diagram of OvHDM is shown in Figure 21. It can be regarded as the frequency domain overlap of subcarriers after passing through the OvTDM system, that is, the combination of OvTDM, OvFDM, OvCDM and OvHDM systems: the bit sequence to be processed is subjected to Gray mapping and modulation. Then, serial-to-parallel conversion is performed to generate a multi-path signal, and the time domain processing is performed by the OvTDM system in the foregoing, and then the frequency domain superposition processing of the subcarriers in the foregoing is performed, and the specific method is to add a signal processed by each OvTDM.
  • the complex baseband signal can be expressed as:
  • w(t) is the impulse response of the pulse shaping filter
  • u(l) is the lth symbol transmitted by the system
  • N is the number of subcarriers
  • ⁇ B is the subcarrier spacing
  • D is the number of frequency domain overlap multiplexing
  • the main lobe zero bandwidth B D * ⁇ B for each subcarrier.
  • the spectral efficiency of the system is
  • Q is the number of modulation levels
  • Timing synchronization, carrier synchronization and channel estimation are the three most important steps for the receiver to receive correctly. Therefore, the design of training symbols is critical, especially for ultra-high spectral efficiency communication systems such as OvTDM, OvFDM, OvCDM and OvHDM systems. If the error of any of these three steps is large, the impact on the whole system will be great, and the subsequent decoding process will be meaningless.
  • the M-series is often used as the training sequence in the communication system. Due to the poor autocorrelation and cross-correlation properties of the M-sequence, the success rate of the system synchronization process is low and the network access is slow.
  • Fig. 7 shows the autocorrelation property of the M-sequence. It can be seen from the figure that the autocorrelation property has a pulse at a certain time interval, and its autocorrelation property is not very good. Therefore, in the signal processing process, the synchronization accuracy of time and frequency is poor, and the success rate and access speed of the user accessing the network are reduced, and the user experience is deteriorated.
  • the training sequence is designed using the LAS code in the OvTDM system and the OvFDM system. It has been found through research that the LAS code has an autocorrelation function that is ideal for the impulse function at the origin. Zero, and the cross-correlation function is zero at all. This is an advantageous attribute for the training sequence.
  • LAS Large Area Synchronized code
  • N the number of pulses and K is between pulses.
  • L the code length.
  • the pulse is generated by a complete complementary orthogonal code, which is characterized by an autocorrelation function being an ideal impact function at the origin, zero everywhere except the origin, and the cross-correlation function is everywhere zero.
  • This feature of the LAS code is applied to the OvTDM system and the OvFDM system, and has better performance improvement for the synchronization success rate and access speed of the entire system.
  • the complete complementary orthogonal code has a dual relationship, and the generation method is to solve another pair of shortest basic complementary codes which are completely orthogonally complementary according to the shortest basic complementary code.
  • the complete complementary orthogonal code is generated by the basic short code +++-, and the generation process is as follows:
  • C 1 is obtained by inverting S 0
  • S 1 is inversed by C 0 and obtained by non-obtaining.
  • the code in matlab is expressed as:
  • fliplr is a function that flips the matrix to the left and right along the vertical axis
  • conj is a function that solves the complex conjugate.
  • C 1 [-1 1]
  • S 1 [-1 -1]
  • C 0 C 1 is combined to generate a new complementary code
  • C 0 ' [1 1 -1 1]
  • S 0 ' [1 -1 -1 -1]
  • the length of each complementary code is expanded from 2 to 4.
  • the length L N of the complementary code (L N is a power of 2) can be designed, that is, the lengths of C n and S n are respectively L N /2.
  • the generated LAS code is iterated, and its length is expanded to L N , the number of iterations is log 2 L N -2, and the resulting complementary code is C n , S n .
  • Figure 8 shows the autocorrelation properties of the LAS code.
  • a LAS code is employed to design a training sequence.
  • a zero sequence of the same length as the LAS short code may be included before the LAS short code, denoted by [0] SN .
  • the training sequence may include two identical LAS short codes such that in the case where one LAS short code is available for timing synchronization, another LAS short code may also be combined to form a LAS short code pair for carrier use. Synchronize.
  • the training sequence may include at least one pair of identical LAS codes. Since the LAS short code still has a good synchronization effect in the case of a large frequency offset, preferably, the training sequence includes at least one pair of identical LAS short codes.
  • carrier synchronization can be divided into two phases, namely carrier coarse synchronization and carrier fine synchronization.
  • the training sequence can include at least two pairs of LAS codes.
  • a pair of LAS codes can be the same LAS short code for carrier coarse synchronization
  • another pair of LAS codes can be the same LAS long code for carrier fine synchronization.
  • a zero sequence of the same length as the LAS short code may be included before each LAS short code, denoted by [0] SN .
  • the training sequence may include at least one LAS code, such as a LAS long code, or may also include two LAS long codes, and perform two-pass channel estimation for the two long LAS codes, thereby improving channel estimation. Success rate.
  • the lengths of the LAS long code and the LAS short code here are shown by way of example only, and may be designed to other lengths.
  • a LAS code training sequence that satisfies timing synchronization, carrier synchronization, and channel estimation can be designed as: [0] SN , [X las ] SN , [0] SN , [X las ] SN , [X las ] LN , [X las ] LN .
  • the first LAS code is a short code, which can realize timing synchronization, and the LAS short code still has a good synchronization effect when the frequency offset is large.
  • the first and second LAS short codes can be used for carrier coarse synchronization, and the short code has the advantage of being able to handle larger frequency offsets.
  • the last two LAS codes are long codes and can be used for fine frequency offset correction and channel estimation.
  • the receiver receives the signal and needs to synchronize with the communication system first, including timing synchronization and carrier synchronization.
  • the principle of timing synchronization is to directly correlate the received signal with the local LAS code by the matched filtering method to obtain the autocorrelation peak.
  • the position of the training symbol is found from the correlation peak according to a certain method.
  • the position of the training symbol is also determined to determine the starting position of the current frame, that is, the time synchronization of the received signal and the system is completed, and the timing synchronization process ends.
  • the LAS code is used for the design training. Practice symbols. Therefore, when calculating the correlation operation between the received signal and the LAS code, the peak size distribution is greatly different. By setting the threshold reasonably, the starting position of the LAS code can be accurately found, and the timing precision is high.
  • the signal reception length here is guaranteed to cover at least the LAS code to ensure that peaks can be detected.
  • the so-called sliding window method autocorrelation operation is to take a window processing on the received signal with the length of the LAS code as the window length, and correlate the signal in the current window with the local LAS code to obtain an autocorrelation result. Then, the window is slid backward, and then the received signal is windowed, and the signal in the current window is correlated with the local LAS code, thereby obtaining a related result. In this way, the window is continuously slid until all the received signals are correlated. From the calculated all autocorrelation results, the position of the LAS code is found by setting a threshold, that is, an autocorrelation result exceeding the threshold as a peak.
  • the training sequence includes only one LAS code, such as a LAS short code, because the short code still has a good synchronization effect in the case of a large frequency offset.
  • the length of the LAS short code can be used as a window length to process the received signal, and the signal in the current window is correlated with the local LAS short code to obtain an autocorrelation result. Then, the window is slid backward, and then the received signal is windowed, and the signal in the current window is correlated with the local LAS code, thereby obtaining a related result. In this way, the window is continuously slid until all the received signals are correlated. From the calculated all autocorrelation results, the position of the LAS code is found by setting a threshold, that is, an autocorrelation result exceeding the threshold as a peak.
  • Figure 9 shows a profile of the autocorrelation results of timing synchronization. Assuming that the threshold is 100, as shown in FIG. 9, there are two autocorrelation results exceeding the threshold 100, but the autocorrelation result at the 25 position is selected as the peak value of the current operation, so that the position at 25 is taken as the found LAS code. s position.
  • Figure 10 shows a schematic diagram of a training sequence in the case where two peaks are detected. Two training sequences for repeated cyclic transmissions are shown in FIG. The length of the received signal spans two training sequences, so look for One of the two peaks may be due to the first LAS short code of the next training sequence. Therefore, it is necessary to determine which LAS short code corresponds to each peak.
  • the first peak exceeding the threshold is selected as the starting position of the first short LAS code, and if the interval length between the two is greater than 2*SN, then the second The peak value exceeding the threshold is the starting position of the first short LAS code.
  • the matched filtering is a smoother peak rather than an independent point, so the peak point needs to be selected according to the actual band limiting filter.
  • FIG 11 shows a block diagram of a timing synchronization unit at the receiving end in accordance with an aspect of the present invention.
  • the timing synchronization unit may be part of the synchronization unit discussed above in connection with Figures 2 and 5.
  • the timing synchronization unit 1100 may include an autocorrelation calculation unit 1110 for performing an autocorrelation calculation.
  • the autocorrelation calculation unit 1110 may perform windowing on the received signal to perform autocorrelation calculation on the signal in the window using the local LAS code, and slide the window for the next autocorrelation calculation until the signal reception length is reached.
  • the timing synchronization unit 1100 may further include a peak determination unit 1120 for determining the position of the peak based on the obtained correlation result set to find the starting position of the LAS code.
  • the peak judging unit 1120 may select an appropriate threshold and use an autocorrelation result exceeding the threshold as a peak.
  • Figure 12 shows a flow chart of a timing synchronization method in accordance with an aspect of the present invention. As shown, the method can include:
  • Step 1201 Perform windowing on the received signal to perform autocorrelation calculation on the signal in the window by using the local LAS code, and slide the window to perform the next autocorrelation calculation until the signal receiving length is reached;
  • Step 1202 Determine the position of the peak according to the obtained correlation result set to find the starting position of the LAS code.
  • the first peak exceeding the threshold is selected as the starting position of the first short LAS code, if both The interval length is greater than 2*SN, and the second peak exceeding the threshold is the starting position of the first short LAS code.
  • the received signal and the system After receiving the signal, it needs to synchronize with the communication system first, including timing synchronization and carrier synchronization.
  • the received signal and the system first maintain time synchronization, and the starting position of the LAS code is obtained through timing synchronization, and then the frequency is performed. Synchronization.
  • a cross-correlation operation is performed on two training codes extracted from a received signal to obtain a frequency offset of a carrier between a receiving end and a transmitting end, wherein the received signal is from a transmitting end and includes a training code-based training sequence; as well as
  • Frequency offset correction is performed on the received signal based on the frequency offset.
  • the training sequence information portion of the received signal includes at least one pair of identical LAS codes.
  • a cross-correlation operation is performed on the repeated LAS codes to obtain a frequency deviation ⁇ f.
  • the received signal is expressed as:
  • the correlation coefficients of the two LAS codes before and after are:
  • L represents the interval between LAS codes.
  • the carrier frequency offset is:
  • the training sequence information portion may include two pairs of LAS codes, wherein a pair of identical LAS codes is a LAS short code, whereby carrier coarse synchronization may be performed first; and a pair of identical LAS long codes are further included. Carrier fine synchronization is possible.
  • the corresponding two-part short LAS code can be extracted according to the training symbol index returned by the timing synchronization, and the short LAS code is subjected to carrier coarse synchronization, and the short code can process a large frequency offset, which is calculated according to the above formula.
  • the estimated frequency offset value is ⁇ f 1 .
  • the starting position of the LAS code calculated by the timing synchronization is the starting position of the first short LAS code, which is 25.
  • the corresponding codes are extracted from the corresponding ones of the received signals.
  • the received signal is subjected to coarse frequency offset correction to obtain a received signal y n '.
  • the fine frequency offset process extracts two partial long LAS codes from y n ' according to the formula
  • the conjugate is multiplied to obtain a correlation coefficient R.
  • the corresponding fine frequency offset ⁇ f 2 is obtained , and L represents the interval between two long LAS codes.
  • the signal after the fine frequency offset correction of the received signal is obtained according to the formula y n ′ y n 'e j2 ⁇ ( ⁇ f)nT .
  • the two frequency offset corrected signals y n " are used as input signals to the channel estimation process, and the carrier synchronization process ends.
  • FIG. 13 shows a block diagram of the carrier synchronization unit 1300.
  • the carrier synchronization unit 1300 can be part of the synchronization unit discussed above in connection with Figures 2 and 5.
  • the carrier synchronization unit 1300 can include a cross correlation calculation unit 1310 and a frequency correction unit 1320.
  • the cross correlation calculation unit 1310 may perform a cross correlation calculation on a pair of LAS codes to obtain a frequency offset of a carrier between the receiving end and the transmitting end.
  • the frequency correcting unit 1320 can perform frequency offset correction on the received signal according to the frequency offset of the carrier.
  • the cross-correlation calculation unit 1310 may first perform a cross-correlation calculation of a pair of LAS short codes to obtain a coarse frequency offset of the carrier between the receiving end and the transmitting end.
  • the frequency correcting unit 1320 may first perform initial frequency offset correction on the received signal according to the coarse frequency offset.
  • the cross-correlation calculation unit 1310 performs cross-correlation calculation on a pair of LAS long codes extracted from the received signal subjected to the initial frequency offset correction to obtain a fine frequency offset of the carrier between the receiving end and the transmitting end.
  • the frequency correcting unit 1320 may further perform secondary frequency offset correction on the first frequency offset corrected received signal according to the fine frequency offset and the coarse frequency offset to obtain a final frequency offset corrected signal.
  • Figure 14 shows a flow chart of a carrier synchronization method in accordance with an embodiment.
  • the carrier synchronization method can include the following steps:
  • Step 1401 Perform cross-correlation on two LAS codes extracted from the received signal to obtain a frequency offset of a carrier between the receiving end and the transmitting end;
  • Step 1402 Perform frequency offset correction on the received signal based on the frequency offset.
  • FIG. 15 shows a flow chart of a carrier synchronization method according to another embodiment.
  • the carrier synchronization method can include the following steps:
  • Step 1501 Perform cross-correlation on two LAS short codes extracted from the received signal to obtain a coarse frequency offset of the carrier between the receiving end and the transmitting end;
  • Step 1502 Perform initial frequency offset correction on the received signal according to the coarse frequency offset.
  • Step 1503 Perform cross-correlation calculation on a pair of LAS long codes extracted from the initial frequency offset corrected received signal to obtain a fine frequency offset of a carrier between the receiving end and the transmitting end;
  • Step 1504 Perform secondary frequency offset correction on the received signal with the initial frequency offset correction according to the fine frequency offset and the coarse frequency offset.
  • Channel estimation is used to estimate the transmission characteristics of the channel, ie the effect of the channel on the transmitted signal.
  • the receiving end can perform channel estimation based on the known training symbols and the received training symbols. For example, the receiving end may perform correlation on the known training symbols and the received training symbols to determine the transmission characteristics of the channel. After performing channel estimation, the receiving end can use the determined channel estimate to demodulate the received unknown data signal to determine the actual data signal transmitted by the transmitting end.
  • the received signal is time synchronized and synchronized with the system hold time. Then, the carrier is synchronized with the received signal.
  • the carrier synchronization includes coarse synchronization and fine synchronization.
  • the carrier frequency offset ⁇ f of the receiver and the transmitter is obtained by synchronization, and the received signal is corrected by the carrier frequency offset to obtain the corrected received signal.
  • y fix do channel estimation for y fix .
  • the present invention utilizes the LAS code as a training sequence, e.g., the long LAS code L-LAS in the training symbol format can be used for channel estimation.
  • the channel estimate can be expressed as:
  • y n represents the received signal after carrier synchronization correction, ie y fix .
  • N represents the length of the LAS code.
  • x n represents the local LAS code, ie x n is represented as one of the last two long LAS codes in the training symbol.
  • R 0 represents the sum of squares of the LAS codes, and P represents the number of multipath channels.
  • the channel estimator estimates the impulse response h(t) of the channel from the received signal y fix of the training symbol, and then constructs an inverse channel system according to the estimated h(t), after the received data signal passes through the inverse channel system Reverting to an estimate of the signal fed to the channel by the transmitting end.
  • the received signal y n can be expressed as e n represents noise. Substituting it into the above formula leads to the following formula:
  • the autocorrelation coefficient By reasonably designing the autocorrelation coefficient to be zero, the estimated channel height is close to the real channel, which greatly improves the accuracy of channel estimation. According to the present invention, since the probability of occurrence of 0 in the LAS code autocorrelation is extremely high, the success rate of channel estimation is greatly improved when channel estimation is performed.
  • the M sequence is generally used in the art for channel estimation.
  • the autocorrelation property of the M sequence is shown in Fig. 7. It can be seen from the figure that the autocorrelation property has a pulse at a certain time interval, and its autocorrelation property is not very good.
  • the channel model and ideal model are actually estimated when doing channel estimation.
  • the deviation is small, which reduces the bit error rate of the system and improves the system performance.
  • the channel estimation process can be implemented by using any one of the long LAS codes, or the channel estimation can be performed for the two long LAS codes, thereby improving channel estimation. Success rate.
  • the receiver may determine whether there is a multipath channel according to the environment.
  • the channel estimate h can be directly calculated according to the above equation.
  • the channel estimation value h p of each multipath path can be separately calculated according to the above formula, wherein the local LAS code x n is offset for each multipath path, and the deviation of each path Can be 1.
  • the actual multipath channel can be, for example, six.
  • the local LAS codes are arranged into 6 columns according to the number of multipaths, and the deviation of each column path is 1, and the arrangement is as shown in FIG. 16.
  • the corresponding LAS code position is found from the correction signal y fix And extract it as y fix-las , a total of two parts.
  • the extracted y fix-las is respectively subjected to the formula of the local LAS code of the six multipath channels after rearrangement.
  • the channel estimation value h p of each multipath path is obtained. Since there are two parts of LAS can perform channel estimation code, after treatment of each part will obtain a channel estimation value h p, of the two parts can be obtained by averaging the last channel of each path of multipath estimation value h p.
  • the received data signal can be demodulated based on the channel estimation value h p of each multipath path, so that the transmitter signal of each multipath path can be recovered.
  • the design symbol structure in the system includes a training sequence TSC (traning sequence code) and data (data).
  • TSC transcription sequence code
  • data data
  • the design of training symbols is very important, which affects the three most important aspects of timing, synchronization and channel estimation of the whole system. If any of these three steps has large errors, the impact on the whole system will be great.
  • the decoding process is meaningless.
  • the design process of the training sequence bandwidth is more complicated.
  • the bandwidth is short, the corresponding power spectral density is large.
  • the data reception and transmission will be affected.
  • the bandwidth is too large, the corresponding power spectral density will be affected. Too small, the sensitivity of the transmitter and receiver of the system is extremely high.
  • the method in which the training sequence and the data have the same bandwidth is generally adopted, and the corresponding power spectral density is the same, and since the bandwidth is short in the general system, the transmission time corresponding to the time domain is long, and the signal is affected.
  • the synchronization, channel estimation processing time process, the waiting time of the subsequent decoding process also becomes longer, and the transmission rate of the system is lowered.
  • the training sequence since the training sequence has a long transmission time, when the signal is sampled, the sampling rate is low, and the temporal resolution is not fine enough, which affects the deviation of the channel estimation.
  • the invention makes the training sequence bandwidth much larger than the data bandwidth (for example, 5 times, 10 times, 15 times or more), so that the power spectral density of the training sequence is lower than the power spectral density of the data, and the training sequence and the bandwidth of the data
  • the relationship between power and spectral density is shown in FIG. Since the transmission power of the training sequence and the data need to be consistent, it can be seen from the figure that when the bandwidth of the training sequence is widened, the corresponding power spectral density is also greatly reduced, relative to the data power spectral density. The words are very low.
  • the system can use all available spreading codes, including m-sequence, Golomb code, CAN (Cyclic Algorithm New), and LAS code.
  • LAS code with complete complementary orthogonal characteristics as an example to introduce the processing of timing synchronization, carrier synchronization and channel estimation. Therefore, all the methods and apparatuses for performing timing synchronization, carrier synchronization, and channel estimation using the LAS code as a training code are also applicable to all combinations.
  • the appropriate spreading code is used as the training code for timing synchronization, carrier and training estimation. Therefore, the above-described algorithms for timing synchronization, carrier synchronization, and channel estimation, exemplified by the LAS code, are shown by way of example only, and the above-described contents of the present invention are applicable to all suitable training codes.
  • the characteristic of the LAS code is that the autocorrelation function is an ideal impact function at the origin, zero everywhere except the origin, and the cross-correlation function is everywhere zero.
  • the autocorrelation property of the LAS code is shown in Fig. 8. Therefore, when the training sequences overlap, they do not cause interference with each other. This design can improve the spectrum utilization and transmission rate of the system.
  • the training sequence and data can be superimposed and transmitted at the same time.
  • the training sequence and data are transmitted at least partially overlapping in frequency and/or time.
  • the structure diagram is as shown in Fig. 18. It can be seen from the figure that the actual data carried by the two carriers has guard bands in the middle, which will not overlap or interfere with each other. The bandwidth of the training sequence overlaps with the actual data. Since the power spectrum density of the training sequence is very low, it does not cause interference to the actual data. Furthermore, different training sequences can be distinguished by different spreading codes, which will not cause Confused. The training sequence does not monopolize specific frequency and time resources, improving the spectrum utilization and transmission rate of the system.
  • the LAS code with complete complementary orthogonal characteristics can be used as the training sequence in the system, which is characterized in that the autocorrelation function is an ideal impact function at the origin, zero everywhere except the origin, and the cross-correlation function is everywhere. Zero, the autocorrelation and cross-correlation properties of the LAS code are shown in Figure 5. Therefore, when the training sequences overlap, they do not cause interference with each other. This design can improve the spectrum utilization and transmission rate of the system.
  • information, signals, and data may be represented using any of a variety of different technologies and techniques.
  • the data, instructions, commands, information, signals, bits (bits), symbols, and chips recited throughout the above description may be by voltage, current, electromagnetic wave, magnetic field or magnetic particle, light field or optical particle, or any combination thereof. To represent.
  • DSPs digital signal processors
  • ASICs application specific integrated circuits
  • FPGAs field programmable gate arrays
  • a general purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine.
  • the processor may also be implemented as a combination of computing devices, such as a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.
  • a software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art.
  • An exemplary storage medium is coupled to the processor to enable the processor to read and write information to/from the storage medium.
  • the storage medium can be integrated into the processor.
  • the processor and the storage medium can reside in an ASIC.
  • the ASIC can reside in the user terminal.
  • the processor and the storage medium may reside as a discrete component in the user terminal.
  • the functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented as a computer program product in software, the functions may be stored on or transmitted as one or more instructions or code on a computer readable medium.
  • Computer readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another.
  • a storage medium may be any available media that can be accessed by a computer.
  • such computer readable media may comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, disk storage or other magnetic storage device, or can be used to carry or store instructions or data structures. Any other medium that is desirable for program code and that can be accessed by a computer.
  • any connection is also properly referred to as a computer readable medium.
  • the software is transmitted from a web site, server, or other remote source using coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave.
  • the coaxial cable, fiber optic cable, twisted pair cable, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of the medium.
  • Disks and discs as used herein include compact discs (CDs), laser discs, optical discs, digital versatile discs (DVDs), floppy discs, and Blu-ray discs, where the disks are often magnetic.
  • the data is reproduced, and the disc optically reproduces the data with a laser. Combinations of the above should also be included within the scope of computer readable media.

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Abstract

本发明提供一种载波同步方法，包括对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中所述接收信号是来自发送端且包括基于训练码的训练序列；以及基于所述频偏对所述接收信号执行频偏校正。本发明还提供一种载波同步装置，包括：互相关计算单元，用于对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中所述接收信号是来自发送端且包括基于训练码的训练序列；以及频率校正单元，用于基于所述频偏对所述接收信号执行频偏校正。本发明提高***频偏精度，降低整个***的误码率。

Description

载波同步方法和装置 技术领域
本发明涉及无线通信***，尤其涉及一种载波同步方法和装置。
背景技术
无线通信网络被广泛部署以提供诸如语音、视频、分组数据、消息接发、广播等各种通信服务。这些无线网络可以是能够通过共享可用的网络资源来支持多个用户的多址网络。这类多址网络的示例包括码分多址(CDMA)网络、时分多址(TDMA)网络、频分多址(FDMA)网络、正交FDMA(OFDMA)网络、以及单载波FDMA(SC-FDMA)网络。
随着全球移动通信不断增强的需求，无线通信的频率资源愈趋紧张。因此，除了基于TDM(时分复用)、FDM(频分复用)的上述传统高频谱利用率的无线通信***之外，还提出了对于频谱具有更高利用率的更激进的通信方案。
重叠时分复用(Overlapped Time Division Multiplexing，OvTDM)***正是这样一种提高***频谱效率的方案。在OvTDM***中，符号之间不但不需要相互隔离，而且可以有很强的相互重迭。换言之，OvTDM***通过人为地引入符号之间的重迭，利用多个符号在时域并行传输数据序列，大幅提高了频谱利用率。
重叠频分复用(Overlapped Frequency Division Multiplexing，OvFDM)***是另外一种提高***频谱效率的方案。在OvFDM***中，子载波频带之间可以有比正交频分复用OFDM更强的重叠。通过频域内各子频带之间更高的重叠程度，在OFDM***的基础上进一步提高了频谱利用率。
除了以上的两种复用***，还进一步演进出重叠码分复用(Overlapping Code Division Multiplexing，OvCDM)和时频二维重叠复用(Overlapped Hybrid Division Multiplexing，OvHDM)两种***。
尽管上述***具有相应的接收解调方案来排除信号在时域或频域的重叠所带来的干扰，但是频谱利用率的大幅提高仍然对信号的接收提出了更高要求。
因此，需要更高性能的网络接入方案。而现有的通信***所采用的m序列进行载波同步的方式并不能满足需求。
发明内容
根据本发明的一个方面，提供一种载波同步方法，包括：
对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中所述接收信号是来自发送端且包括基于训练码的训练序列；以及
基于所述频偏对所述接收信号执行频偏校正。
本发明的另一个方面，提供一种载波同步装置，包括：
互相关计算单元，用于对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中所述接收信号是来自发送端且包括基于训练码的训练序列；以及
频率校正单元，用于基于所述频偏对所述接收信号执行频偏校正。
本发明具有以下的有益效果：本发明通过在通信***中设计训练序列，利用训练码自相关函数在原点是理想的冲击函数，原点以外处处为零，而互相关函数处处为零的特性，在信号接收处理中使用该训练码做载波同步。采用训练码做载波同步时，通过两次同步过程，提高***频偏精度，为后续的信道估计过程和译码过程奠定了基础，降低整个***的误码率。
附图说明
图1示出了OvTDM***的发射端调制模块的框图；
图2示出了OvTDM***的接收端的信号预处理模块的框图；
图3示出了OvTDM***的接收端序列检测模块的框图；
图4示出了OvFDM***的发射端的调制模块框图；
图5示出了OvFDM***的接收端的信号预处理模块的框图；
图6示出了OvFDM***的接收端的信号检测模块的框图；
图7示出了M序列的自相关特性；
图8示出了LAS码的自相关特性；
图9示出了定时同步的自相关结果的分布图；
图10示出了检测到两个峰值情形下的训练序列的示意图；
图11示出了根据本发明的一方面的接收端的定时同步单元的框图；
图12示出了根据本发明的一方面的定时同步方法的流程图；
图13示出了根据本发明的一方面的载波同步单元的框图；
图14示出了根据本发明的一方面的载波同步方法的流程图；
图15示出了根据本发明的一方面的载波同步方法的流程图；
图16示出了多径信道的排列示意图；
图17示出了根据本发明的一方面的训练序列和数据的频宽及功率谱密度关系图；
图18示出了根据本发明的一方面的两个载波信号同时发送数据时的频谱示意图；
图19为OvCDM***功能框图；
图20为OvCDM的编码器示意图；
图21为OvHDM的***示意图。
具体实施方式
以下结合附图和具体实施例对本发明作详细描述。注意，以下结合附图和具体实施例描述的诸方面仅是示例性的，而不应被理解为对本发明的保护范围进行任何限制。
除了应用在OvTDM、OvFDM、OvCDM和OvHDM***中，本文中所描述的诸技术也可广泛应用于实际移动通信***中，如TD-LTE、TD-SCDMA等***，也可广泛应用于卫星通信、微波视距通信、散射通信、大气层光通信、红外通信与水生通信等任何无线通信***中。术语“网络”和“***”常被可互换地使用。
移动通信的不断发展以及新业务的层出不穷对数据传输速率提出了越来越高的要求，而移动通信的频率资源却十分有限，如何利用有限的频率资源实现数据的高速传输成为当今移动通信技术面临的一个重要问题
上述OvTDM、OvFDM、OvCDM和OvHDM***正是这种可以大幅提高频谱利用率的解决方案。下面简要介绍OvTDM***的发送和接收过程。
OvTDM***利用多个符号在时间域并行传输数据序列。在发射端形成多个符号在时间域上相互重叠的发射信号，在接收端根据传输数据序列与传输数据序列时间波形之间的一一对应关系，对接收信号进行时间域内的按数据序列检测。OvTDM***积极利用这些重叠使之产生编码约束关系，从而大幅度提高了***的频谱效率。
图1示出了OvTDM***的发射端调制模块的框图。发送端调制模块100可包括数字波形发生单元110、移位寄存单元120、乘法单元130及加法单元140。
首先，由数字波形发生单元110以数字方式设计生成发送信号的第一个调制信号包络波形h(t)，移位寄存单元120将该包络波形h(t)进行特定时间移位，形成其它各个时刻调制信号的包络波形h(t-i×ΔT)，乘法单元130将所要发送的并行的符号xi与相应时刻的包络波形h(t-i×ΔT)相乘，得到各个时刻经调制后的待发送信号波形xih(t-i×ΔT)。加法单元140将所形成的各个待发送波形进行叠加，形成发射信号波形。
OvTDM***的接收端主要分为信号预处理模块200和序列检测模块300。图2示出了OvTDM***的接收端的信号预处理模块200的框图。信号预处理模块用于辅助形成每一帧内的同步接收数字信号序列，如图所示，该信号预处理模块可包括同步单元210、信道估计单元220、和数字化处理单元230。
同步单元210用于对接收信号在时域形成符号同步，以与***保持同步状态，主要包括定时同步和载波同步。同步完成后信道估计单元220对接收信号做信道估 计，以用于估计实际传输信道的参数。数字化处理单元230用于对每一帧内的接收信号进行数字化处理，从而形成适合序列检测部分进行序列检测的接收数字信号序列。
在预处理之后，可在序列检测模块300内对接收信号进行序列检测，对接收到的波形按照波形发送时间间隔切割并按照一定的译码算法对切割后的波形进行译码。图3示出了OvTDM***的接收端序列检测模块的框图。如图所示，序列检测模块300可包括分析存储单元310、比较单元320、以及保留路径存储单元和欧氏距离存储单元330。在检测过程中，分析存储单元作出OvTDM***的复数卷积编码模型及格状图，并列出OvTDM***的全部状态，并存储。比较单元根据分析存储单元中的格状图，搜索出与接收数字信号最小欧氏距离的路径，而保留路径存储单元和欧氏距离存储单元则分别用于存储比较单元输出的保留路径和欧氏距离或加权欧氏距离。保留路径存储单元和欧氏距离存储单元需要为每一个稳定状态各准备一个。保留路径存储单元长度可以优选为4K～5K。欧氏距离存储单元优选为只存储相对距离。
图4示出了OvFDM***的发射端的调制模块框图。发射端的OvFDM调制模块可包括调制载波频谱产生单元410、载波频谱移位单元420、乘法单元430、加法单元440、以及傅立叶逆变换单元450。
首先，由调制载波频谱产生单元410设计生成一个子载波的包络频谱信号H(f)，载波频谱移位单元420将该包络频谱信号H(f)依次频移特定载波频谱间隔ΔB，得出下一个子载波的包络频谱信号，并将该下一个子载波的包络频谱信号频移ΔB，依次下去得到频谱间隔为ΔB的所有子载波的频谱波形H(f-i×ΔB)。
乘法单元430将所要发送的多路并行的符号Xi分别与生成的对应的各个子载波频谱波形H(f-i×ΔB)相乘，得到多路经过相应子载波调制的调制信号频谱XiH(f-i×ΔB)。
加法单元440将所形成的多路调制信号频谱进行叠加，形成复调制信号的频谱
Figure PCTCN2017079581-appb-000001
最后，由傅立叶逆变换单元450将生成的复调制信号的频谱进行离散付氏反变换，最终形成时域的复调制信号Signal(t)TX＝ifft(S(f))。
OvFDM***的接收端主要分为信号预处理模块500和信号检测模块600。图5示出了OvFDM***的接收端的信号预处理模块的框图。如图所示，预处理模块可包括同步单元510、信道估计单元520、和数字化处理单元530。
同步单元510用于对接收信号在时域形成符号同步，以与***保持同步状态，主要包括定时同步和载波同步。同步完成后信道估计单元520对接收信号做信道估计，以用于估计实际传输信道的参数。数字化处理单元530用于对各个符号时间区间 的接收信号进行取样和量化，使之变为数字信号序列。
在预处理之后，可在信号检测模块600中对接收信号进行检测。图6示出了OvFDM***的接收端的信号检测模块600的框图。如图所示，信号检测模块600可包括傅立叶变换单元610、频率分段单元620、卷积编码单元630、以及数据检测单元640。傅立叶变换单元610用于将经过预处理的时域信号转换成频率域信号，即对每个时间符号区间的接收数字信号序列进行傅立叶变换以形成每个时间符号区间的实际接收信号频谱。频率分段单元620用于对每个时间符号区间的实际接收信号频谱在频域以频谱间隔ΔB分段，形成实际接收信号分段频谱。卷积编码单元630用于形成接收信号频谱与发送的数据符号序列之间的一一对应关系。数据检测单元640用于根据卷积编码单元形成的一一对应关系，检测数据符号序列。
以上介绍了OvTDM***和OvFDM***的发送和接收端的处理过程。尽管上述OvTDM***和OvFDM***具有相应的接收解调方案来排除信号在时域或频域的重叠所带来的干扰，但是频谱利用率的大幅提高仍然对信号的接收提出了更高要求。
以上的两种重叠***并非仅有的重叠***，从并行编码的观点看，各种复用技术如时分复用TDM、频分复用FDM及正交频分复用OFDM、物理空分复用SDM、统计空分复用MIMO等，也是属于编码约束长度L＝1，码率高于1，编码矩阵仅仅是列矩阵，编码元素仅仅是对应的时域、频域、空域等，输入数据可以使任何调制信号的简单线性编码复用的特例。
本发明进一步给出一种OvCDM***(Overlapping Code Division Multiplexing,重叠码分复用)，OvCDM可以看做是一种并联的复数域卷积码，其***功能图如附图19所示，对应的编码器结构如附图20所示。OvCDM***的关键是编码矩阵，即卷积扩展系数，一般通过计算机搜索所有欧氏距离较大的矩阵作为编码矩阵，其编码矩阵排列如下：
Figure PCTCN2017079581-appb-000002
利用以上的***，以及相应的编码矩阵，OvCDM编码过程如下所述：
将待发送数据经过串并转换成为K路子数据流，第i路上的数据流记为 ui＝ui,0ui,1ui,2...。比如K＝2时，u0＝u0,0u0，2u0,4...，u1＝u1,1u1,3u1,5...
将每一路数据送入一个移位寄存器进行加权叠加，第i路的加权系数为bi＝bi,0bi,1bi,2...，其为一复向量。
把各路信号相加输出。
最终OvCDM编码器的输出为c＝c0c1c2...，
Figure PCTCN2017079581-appb-000003
OvCDM的码率为
Figure PCTCN2017079581-appb-000004
其中n为子数据流长度。当n很长时，由移位寄存器拖尾所带来的码率损失可以忽略不计，于是有rOvCDM≈k。
相较传统的二元域卷积编码模型码率一般小于1，导致频谱效率损失。而复数域的卷积编码码率等于1，单路的卷积编码扩展不会导致频谱效率损失，还会增加额外的编码增益。
另一方面，在上述的OvTDM***和OvFDM***的基础上，进一步演进出OvHDM***(Overlapped Hybrid Division Multiplexing)，即时频二维重叠复用***。其不仅在时域中帧符号间相互重叠，而且频域中子载波之间也相互重叠，实现了时域和频域同时重叠。
OvHDM的***框图如图21所示，可以视为在经过OvTDM***后的子载波的频域重叠，即OvTDM、OvFDM、OvCDM和OvHDM两个***的结合：待处理的比特序列经过格雷映射和调制之后进行串并转换，生成多路信号，并通过前述中的OvTDM***进行时域的处理，然后进行前述中的子载波的频域叠加处理，具体方法是，对每一路经过OvTDM处理的信号加入ej2π(N-1)ΔBt的调制，N＝1,2,3……，再进行叠加输出。经过传输后(此处以叠加白噪声作为简单的数学表述)，先经过N路子载波匹配滤波器处理(即子载波0匹配滤波器、子载波1匹配滤波器……子载波(N-1)匹配滤波器)，以实现对前面经过频域重叠处理的子载波的逆处理，然后再进行译码，此处以多用户最大似然值算法(Multi-User Maximum Likelihood Sequence Detection,MU-MLSD)为例子，最终确定各个路径，得到N路子载波输出，最后进行并串转换得到串行数据，然后进行解调和格雷逆映射等操作。
具体的，基于图21的OvHDM***，其复基带信号可以表示为：
Figure PCTCN2017079581-appb-000005
其中，时域参数：
w(t)是脉冲成型滤波器的冲击响应
u(l)是***发射的第l个符号
T是每个符号的周期
ΔT是发射符号的间隔，ΔT＝T/K，K为时域重叠复用次数
L是每帧发射的符号总数
Ta是每帧的帧长，且Ta＝(L+K-1)*ΔT。
频域参数：
N是子载波数
ΔB是子载波间隔
D为频域重叠复用次数
主瓣零点带宽Ba＝(N+D-1)*ΔB
每个子载波的主瓣零点带宽B＝D*ΔB。
而基于该OvHDM***，***的频谱效率为
Figure PCTCN2017079581-appb-000006
其中Q是调制电平数，λ是脉冲成型滤波器的时间带宽积，即λ＝BT。
如果L趋向无穷，
Figure PCTCN2017079581-appb-000007
如果子载波数也趋向无穷，得
Figure PCTCN2017079581-appb-000008
即为OvHDM***可达的极限频谱效率。
一般的通信***中都需要设计训练序列，其作用主要是在收到信号后经过处理，可同时实现定时同步、载波同步和信道估计。定时同步、载波同步和信道估计是接收端正确接收的三个最重要环节。因此，训练符号的设计至关重要，特别是对于OvTDM、OvFDM、OvCDM和OvHDM***这种超高频谱效率的通信***尤其如此。如果这三个步骤中任一步骤误差较大，对整个***的影响将会很大，后续的译码过程也就没有意义了。
目前通信***常采用M序列为训练序列，由于M序列自相关和互相关特性较差，导致***同步过程成功率低，网络接入慢。图7示出了M序列的自相关特性，从图中可以看到其自相关特性间隔一定时间都会出现脉冲，其自相关特性不是很好。因此在信号处理过程中，对时间和频率的同步精度较差，降低用户接入网络的成功率和接入速度，使用户体验变差。
根据本发明的一方面，在OvTDM***和OvFDM***中利用LAS码设计训练序列。经研究发现，LAS码具有自相关函数在原点是理想的冲激函数，原点以外处处 为零，而互相关函数处处为零的特性。这对于训练序列而言是及其有利的属性。
LAS(Large Area Synchronized，大区域同步)码是由一系列脉冲和不等长的0值脉冲间隔组成，可以表示为(N,K,L)，其中N表示脉冲个数，K表示脉冲之间的最短间隔长度，L表示码长。脉冲由完备互补正交码生成，其特点为自相关函数在原点是理想的冲击函数，原点以外处处为零，而互相关函数处处为零。利用LAS码的这个特点应用于OvTDM***和OvFDM***中，对于整个***的同步成功率和接入速度有较好的性能改善。
以下简要介绍LAS码的生成方法。
完备互补正交码具有对偶关系，生成方法是根据最短基本互补码求解出与之完全正交互补的另一对最短基本互补码。本案例中以基本短码+++-来生成完备互补正交码，生成过程如下：
C0＝[1 1]，对应为++，S0＝[1 -1],对应为+-，根据C0和S0分别求出其互补码C1和S1。C1为对S0取反得到，S1为对C0取反并求非得到，matlab中代码表示为：
C1＝fliplr(S0)，S1＝-1*conj(fliplr(C0))。其中fliplr为对矩阵进行沿垂直轴左右翻转的函数，conj为求复共轭的函数。
据此求得C1＝[-1 1],S1＝[-1 -1]，将C0C1组合生成新的互补码为C0'＝[1 1 -1 1]，S0'＝[1 -1 -1 -1]，此时每个互补码的长度由2扩充到4。
这里可以设计互补码的长度LN(LN为2的幂次方)，即Cn和Sn的长度分别为LN/2。采用上述方法，对生成的LAS码进行迭代，将其长度扩充为LN，迭代次数为log2LN-2，最终生成的互补码为Cn、Sn。
将这对互补码和零序列组合生成LAS码，表示形式为：Las＝[Cn L0 Sn]，其中L0表示0的个数，即Cn和Sn之间的最短间隔长度，最终生成的LAS码长度表示为L＝LN+L0。
图8示出了LAS码的自相关特性。
根据本发明的一方面，采用了LAS码来设计训练序列。
对于定时同步的用途，训练序列包括至少一个LAS码。由于LAS短码在频偏较大的情况下仍有较好的同步效果，因此，较优地，训练序列包括至少一个LAS短码，以[Xlas]SN表示，其中该LAS短码的长度记为SN，其互补码长和零序列长度分别表示为L短-N、L短-0，SN＝L短-N+L短-0。
为了进一步优化LAS码的自相关特性，在该LAS短码之前还可包括与该LAS短码相同长度的一个零序列，以[0]SN表示。
特定实施例中，训练序列可包括两个相同的LAS短码，这样在其中一个LAS短码可用于定时同步的情况下，还可以与另一LAS短码组成LAS短码对，以用于载波同步。
对于载波同步的用途，训练序列可包括至少一对相同的LAS码。由于LAS短码在频偏较大的情况下仍有较好的同步效果，因此，较优地，训练序列包括至少一对相同的LAS短码。
较优地，载波同步可以分为两个阶段，即载波粗同步和载波细同步。因此，训练序列可包括至少两对LAS码。较优地，一对LAS码可为相同的LAS短码以用于载波粗同步，另外一对LAS码可为相同的LAS长码，以用于载波细同步。LAS长码可用[Xlas]LN表示，其中该LAS长码的长度记为LN，其互补码长和零序列长度分别表示为L长-N、L长-0，LN＝L长-N+L长-0。
为了进一步优化LAS码的互相关特性，在每个LAS短码之前还可包括与LAS短码相同长度的一个零序列，以[0]SN表示。
对于信道估计的用途，训练序列可包括至少一个LAS码，例如一个LAS长码，或者，也可包括两个LAS长码，针对这两个长LAS码做两遍信道估计，从而提高信道估计的成功率。
作为特定示例，可设计L长-N＝256，L长-0＝16；L短-N＝16，L短-0＝8。当然，这里的LAS长码和LAS短码的长度仅作为示例示出，也可设计成其他的长度。
作为较优的实施例，一种同时满足定时同步、载波同步和信道估计的LAS码训练序列可设计为：[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN。在此实施例中，第一个LAS码为短码，可实现定时同步,LAS短码在频偏较大仍有好的同步效果。第一个和第二个LAS短码可用于载波粗同步，短码的好处是可以处理较大的频偏。最后两个LAS码为长码，可用于细频偏纠正和信道估计。
定时同步过程
接收机收到信号，需要先跟通信***保持同步，包括定时同步和载波同步。定时同步的原理是通过匹配滤波方法,直接将接收信号与本地LAS码求自相关运算，得到自相关峰值。从相关峰值中根据一定的方法找到训练符号的位置。找到训练符号的位置也就确定了当前帧的起始位置，即完成了接收信号和***的时间同步，定时同步过程结束。
如前所述，由于LAS码的自相关和互相关特性都比较好，将LAS码用于设计训 练符号。由此，在计算接收信号和LAS码的相关运算时，峰值大小分布差异较大，通过合理的设置阈值，可以很精确的找到LAS码的起始位置，定时精度较高。
具体在寻找LAS码的相关峰值时，根据训练符号结构，采取合适的信号接收长度，使用滑窗法自相关运算方式，将接收信号与本地LAS码求相关运算寻找自相关峰值来确定LAS码的位置。例如，这里的信号接收长度可保证至少涵盖有LAS码，以确保能检测到峰值。
所谓的滑窗法自相关运算，是以LAS码的长度为窗口长度对接收信号作取窗处理，将当前窗口内的这段信号与本地的LAS码作相关运算，从而得到一个自相关结果。然后，将窗口向后滑动，再对接收信号进行取窗，将当前窗口内的这段信号与本地的LAS码再作相关运算，从而再得到一个相关结果。以此方式，不断滑动窗口，直至对接收到的信号全部进行了相关运算。从计算得出的全部自相关结果，通过设置阈值，即超过阈值的自相关结果作为峰值，找到LAS码的位置。
在一实例中，训练序列中仅包括一个LAS码，例如一个LAS短码，因为短码在频偏较大的情况下仍有较好的同步效果。在此情况下，可以将该LAS短码的长度作为窗口长度对接收信号作取窗处理，将当前窗口内的这段信号与本地的LAS短码作相关运算，从而得到一个自相关结果。然后，将窗口向后滑动，再对接收信号进行取窗，将当前窗口内的这段信号与本地的LAS码再作相关运算，从而再得到一个相关结果。以此方式，不断滑动窗口，直至对接收到的信号全部进行了相关运算。从计算得出的全部自相关结果，通过设置阈值，即超过阈值的自相关结果作为峰值，找到LAS码的位置。
在多径信道的情况下，有可能出现后面几个径的幅值高过第一条径的幅值，应该选超过阈值的第一个峰值点，而不一定是全局最大值。图9示出了定时同步的自相关结果的分布图。假设阈值为100，如图9所示，超过阈值100的自相关结果有两个，但是选取在25位置的自相关结果作为本次运算的峰值，从而将此在25的位置作为找到的LAS码的位置。
在先前的较优的训练符号格式[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN的情况下，训练序列中存在两个LAS短码。此时，通过上述滑窗自相关计算法可以找出两个超过阈值的峰值。图9示出了存在两个峰值的自相关结果的分布图。此时，需要判断哪一个是在前短码的峰值，哪一个是在后短码的峰值。
图10示出了检测到两个峰值情形下的训练序列的示意图。在图10中示出了重复循环发送的两条训练序列。接收信号的长度跨越了两条训练序列，因此，找 出的两个峰值可能其中一个是由于下一个训练序列的第一个LAS短码所引起的。所以需要判断每一个峰值所对应的是哪一个LAS短码。
具体而言，如果两个峰值间隔长度为2*SN，那么选取第一个超过阈值的峰值为第一个短LAS码的起始位置，如果两者间隔长度为大于2*SN，则第二个超过阈值的峰值为第一个短LAS码的起始位置。
如果存在多径信道，那么滑窗后会出现两个部分集中分布相关峰，对每部分的相关峰分别和阈值进行比较，选取过阈值的第一个峰值点，两部分比较完后将得到两个超过阈值的点，再根据如上的方法确定对应LAS码的位置。
另外，如果发射信号经过了其他带限滤波器，则匹配滤波后是一个个较光滑的峰，而不是独立的点，所以需要根据实际带限滤波器选取峰值点。
图11示出了根据本发明的一方面的接收端的定时同步单元的框图。该定时同步单元可以是上文结合图2和图5所讨论的同步单元的一部分。
如图11所示，定时同步单元1100可包括自相关计算单元1110以用于执行自相关计算。该自相关计算单元1110可对接收到的信号进行取窗，以采用本地的LAS码对窗口内的信号作自相关计算，并滑动该窗口以进行下一次自相关计算，直至达到信号接收长度。定时同步单元1100还可包括峰值判断单元1120，以用于根据获得的相关结果集合来判断峰值的位置，以寻找LAS码的起始位置。峰值判断单元1120可选取合适的阈值，将超过阈值的自相关结果作为峰值。
图12示出了根据本发明的一方面的定时同步方法的流程图。如图所示，该方法可包括：
步骤1201：对接收到的信号进行取窗，以采用本地的LAS码对窗口内的信号作自相关计算，并滑动该窗口以进行下一次自相关计算，直至达到信号接收长度；以及
步骤1202：根据获得的相关结果集合来判断峰值的位置，以寻找LAS码的起始位置。
如上所述，在存在两个LAS短码的情况下，如果两个峰值间隔长度为2*SN，那么选取第一个超过阈值的峰值为第一个短LAS码的起始位置，如果两者间隔长度为大于2*SN，则第二个超过阈值的峰值为第一个短LAS码的起始位置。
载波同步过程
接收到信号后，需要先跟通信***保持同步，包括定时同步和载波同步，接收信号和***先保持时间上的同步，通过定时同步获取LAS码的起始位置，再进行频率 的同步。
在本申请中，对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中接收信号是来自发送端且包括基于训练码的训练序列；以及
基于频偏对接收信号执行频偏校正。
对于载波同步，接收信号的训练序列信息部分包括至少一对相同的LAS码。对重复的LAS码进行互相关运算,得到频率偏差Δf。
假设接收机与发射机之间的载波偏差为Δf，AD采样间隔为T,那么接收端忽略噪声信号影响时，收到的信号表示为：
yn＝xnej2πΔfnT
前后两个LAS码的相关系数为：
Figure PCTCN2017079581-appb-000009
其中L表示LAS码之间的间隔。
由上式可知，载波频偏为：
Figure PCTCN2017079581-appb-000010
较优地，训练序列信息部分可包括两对LAS码，其中，一对相同的LAS码为LAS短码，由此可以先进行载波粗同步；另外再包括一对相同的LAS长码，由此可以进行载波细同步。
由于已经完成了定时同步，可根据定时同步返回的训练符号索引提取出对应的两部分短LAS码，对短LAS码进行载波粗同步，短码可以处理较大的频偏，根据上述公式计算得到估计的频偏值为Δf1。然后再提取出两部分长LAS码，对长LAS码进行载波细频偏纠正，得到估计的频偏值为Δf2，参考粗同步的频偏，则最终输出的频偏为Δf＝Δf1+Δf2。
以先前的较优的训练符号格式[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN为例。令LN＝272，SN＝24，训练符号总长度为640。两个短LAS分别在(25:48)和(73:96)两个位置，长LAS码分别在(97:368)和(369：640)两个位置。
理想状态下，定时同步计算得到的LAS码的起始位置为第一个短LAS码的起始位置，即为25。根据此索引和长短码的码长LN和SN，从接收信号中对应的提取出相应的码。
载波粗同步
从接收信号中提取出两部分短LAS码，根据公式
Figure PCTCN2017079581-appb-000011
对其求共轭相乘，得到相关系数R。再根据公式
Figure PCTCN2017079581-appb-000012
求出对应的粗频偏Δf1，其中L表示两个短LAS码之间的间隔，由训练符号结构可以看出，L＝2*SN＝48。
根据计算出的粗频偏通过公式
Figure PCTCN2017079581-appb-000013
对接收信号进行频偏校正，得到第一次频偏校正后的信号。
载波细频偏校正
载波粗同步中对接收信号进行了粗频偏校正，得到接收信号yn'。细频偏过程为从yn'中提取出两部分长LAS码，根据公式
Figure PCTCN2017079581-appb-000014
对其求共轭相乘，得到相关系数R。再根据公式
Figure PCTCN2017079581-appb-000015
求出对应的细频偏Δf2，L表示两个长LAS码之间的间隔，由训练符号结构可以看出，L＝LN＝272。
参考粗同步的频偏，则最终输出的频偏为Δf＝Δf1+Δf2。并根据公式yn”＝yn'ej2π(-Δf)nT求出对接收信号细频偏纠正后的信号。
将两次频偏校正后的信号yn”作为输入信号给信道估计过程，载波同步过程结束。
图13示出了载波同步单元1300的框图。该载波同步单元1300可以是上文结合图2和图5所讨论的同步单元的一部分。
如图所示，载波同步单元1300可包括互相关计算单元1310和频率校正单元1320。互相关计算单元1310可对一对LAS码执行互相关计算以获得接收端和发射端之间载波的的频偏。频率校正单元1320可根据该载波的频偏，对接收信号执行频偏校正。
在一实施例中，互相关计算单元1310可首先执行一对LAS短码的互相关计算，以获得接收端和发射端之间载波的粗频偏。频率校正单元1320可先根据该粗频偏，对接收信号执行初次频偏校正。互相关计算单元1310再对从经过初次频偏校正的接收信号所提取的一对LAS长码执行互相关计算，以获得接收端和发射端之间载波的细频偏。频率校正单元1320可再根据该细频偏和该粗频偏，对经初次频偏校正的接收信号执行二次频偏校正，以得到最终频偏校正后的信号。
图14示出了根据一实施例的载波同步方法的流程图。如图所示，载波同步方法可包括以下步骤：
步骤1401：对从接收信号提取的两个LAS码执行互相关，以获得接收端和发射端之间载波的频偏；以及
步骤1402：基于该频偏对接收信号执行频偏校正。
图15示出了根据另一实施例的载波同步方法的流程图。如图所示，载波同步方法可包括以下步骤：
步骤1501：对从接收信号提取的两个LAS短码执行互相关，以获得接收端和发射端之间载波的粗频偏；
步骤1502：根据该粗频偏，对接收信号执行初次频偏校正；
步骤1503：对从经初次频偏校正的接收信号所提取的一对LAS长码执行互相关计算，以获得接收端和发射端之间载波的细频偏；以及
步骤1504：根据该细频偏和该粗频偏，对经初次频偏校正的接收信号执行二次频偏校正。
尽管为使解释简单化将上述方法图示并描述为一系列动作，但是应理解并领会，这些方法不受动作的次序所限，因为根据一个或多个实施例，一些动作可按不同次序发生和/或与来自本文中图示和描述或本文中未图示和描述但本领域技术人员可以理解的其他动作并发地发生。
信道估计过程
信道估计用于估计信道的传输特性，即信道对所传输的信号的影响。通过利用发送端和接收端双方已知的训练符号，接收端能够根据该已知的训练符号以及接收到的训练符号来执行信道估计。举例而言，接收端可以对已知的训练符号以及接收到的训练符号执行相关，从而确定信道的传输特性。在进行信道估计之后，接收端能够利用所确定的信道估计来解调接收到的未知数据信号，以确定发送端发送的实际数据信号。
接收信号经过定时同步，和***保持时间同步。然后再和接收信号做载波同步，载波同步包括粗同步和细同步，通过同步获取了接收机和发送机的载波频偏Δf，通过载波频偏对接收的信号做修正，得到修正后的接收信号yfix，对yfix做信道估计。
本发明利用LAS码作为训练序列，例如训练符号格式中的长LAS码L-LAS可用于信道估计。
信道估计可表示为：
Figure PCTCN2017079581-appb-000016
其中yn表示经过载波同步修正后的接收信号，即yfix。N表示LAS码长度。xn表示本地LAS码，即xn表示为训练符号中的最后两个长LAS码之一。R0表示LAS码的平方和，P表示多径信道个数。
信道估计器从训练符号的接收信号yfix中估计信道的冲激响应h(t)，然后根据估计出的h(t)构造一个逆信道***，接收到的数据信号经过该逆信道***之后被还原成对发送端馈送到信道的信号的估计。
一般接收信号yn可表达为
Figure PCTCN2017079581-appb-000017
en表示噪声。将其代入上式展开后得到如下公式：
Figure PCTCN2017079581-appb-000018
Figure PCTCN2017079581-appb-000019
表示训练序列的自相关，通过合理设计自相关系数为零，估计信道高度接近真实信道，从而极大地提高了信道估计的精度。根据本发明，由于LAS码自相关出现0的概率极高，因此在进行信道估计时大大提高了信道估计的成功率。
本领域一般采用M序列进行信道估计。M序列的自相关特性如附图7所示，从图中可以看到其自相关特性间隔一定时间都会出现脉冲，其自相关特性不是很好，对应信道估计公式
Figure PCTCN2017079581-appb-000020
中的
Figure PCTCN2017079581-appb-000021
值不为0的概率很大，因此估计出的信道模型和理想信道模型偏差较大，对于后续的译码处理影响很大，提高了***的误码率。
对比LAS码序列，其具有自相关函数在原点是理想的冲击函数，原点以外处处为零，而互相关函数处处为零的特点，因此在做信道估计时，实际估计出的信道模型和理想模型偏差很小，降低了***的误码率，对***性能得到了很好的改善。
根据本发明，由于训练符号中长LAS码共有两个，因此信道估计过程可以采用其中任一个长LAS码来实现，或者也可以针对这两个长LAS码做两遍信道估计，从而提高信道估计的成功率。
在通信环境中可存在一条信道或多径信道，接收机可根据环境来确定是否存在多径信道。在没有多径信道的情况下，即p＝0，根据上式可以直接计算出信道估计h。而在有多径信道的情况下，可以根据上式分别计算每条多径路径的信道估计值hp，其中针对每条多径路径将本地LAS码xn进行偏移，每一条路径的偏差可以为1。
举例而言，实际的多径信道可为例如6条。首先将本地LAS码按照多径个数排列成6列，每一列路径的偏差为1，排列方式如附图16所示。
根据训练符号格式[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN，从修正信号yfix中找到对应的LAS码位置，并提取出来为yfix-las，共两部分。
将提取出来的yfix-las分别与重新排列后的6条多径信道的本地LAS码经过公式
Figure PCTCN2017079581-appb-000022
处理后，得到每条多径路径的信道估计值hp。由于共有两部分LAS码可以进行信道估计，经过处理后每部分都会得到信道估计值hp，对两部分求平均值则可得到最后的每条多径路径的信道估计值hp。
然后，可基于每条多径路径的信道估计值hp来解调接收到的数据信号，从而得恢复出每条多径路径的发送端信号。
设计训练序列频宽
本***中设计符号结构包括训练序列TSC(traning sequence code)和数据(data)。训练符号的设计至关重要，影响了整个***的定时、同步、信道估计三个最重要的环节，如果这三个步骤中任一步骤误差较大，对整个***的影响将会很大，后续的译码过程也就没有意义了。
训练序列频宽的设计过程较为复杂，频宽较短时其对应的功率谱密度较大，当***中存在多个载波时会影响数据的接收和发送，频宽过大时对应的功率谱密度太小，对***的发送机和接收机的灵敏度要求极高。
在现有通信***中，一般采用训练序列和数据的频宽相同的方法，其对应的功率谱密度相同，且由于一般***中频宽都较短，因此对应于时域发送时间较长，影响信号同步、信道估计处理时间过程，后续译码过程等待时间也变长，降低了***的传输速率。另外，由于训练序列发送时间较长，因此在对信号进行采样时，其采样率较低，时间分辨率不够精细，影响信道估计的偏差。
本发明使得训练序列频宽远大于数据频宽(例如，5倍、10倍、15倍或以上)，从而训练序列的功率谱密度低于数据的功率谱密度，其训练序列、数据的频宽和功率谱密度关系图如附图17所示。由于训练序列和数据的发送功率需保持一致，由图中可以看出，当训练序列的频宽变宽后，其对应的功率谱密度随之也会大幅度降低，相对于数据功率谱密度而言是很低的。
本***可以使用所有的可用扩频码，包括m序列、Golomb码、CAN(Cyclic Algorithm New)、以及LAS码等。本***中我们以具有完备互补正交特性的LAS码为例，介绍定时同步、载波同步和信道估计的处理过程。因此，前文所述的利用LAS码作为训练码进行定时同步、载波同步、信道估计的所有方法及装置也适用于所有合 适的扩频码作为训练码进行定时同步、载波和训练估计。因此，上文以LAS码为例示出的定时同步、载波同步和信道估计的算法仅仅是作为示例示出的，本发明的上述内容适用于所有合适的训练码。
LAS码的特点是自相关函数在原点是理想的冲击函数，原点以外处处为零，而互相关函数处处为零，LAS码的自相关特性如附图8所示。因此当训练序列重叠时也不会相互造成干扰。这样设计可以提高***的频谱利用率和传输速率。
由公式
Figure PCTCN2017079581-appb-000023
可知，当频域频宽越大时，其对应在时域的时间越小，即在较短的时间内就可以完成训练序列的发送和接收过程。在信号接收过程，对于同样长度的数据，当接收时间变短，可以将信号的采样率提高，使得时间分辨率更精细。在信道估计过程提高时间分辨率的精确度，使得信道估计结果更精确。
在一方面，由于训练序列的功率谱密度极低，几乎不会对数据信号产生影响，因此训练序列和数据可在同一时间叠加发送。换言之，训练序列和数据是在频率和/或时间上至少部分重叠地发送的。当有两个载波信号同时发送数据时，其构造图如附图18所示，从图中可以看出，两个载波所承载的实际数据中间有保护带，不会重叠也不会相互造成干扰；而训练序列的频宽和实际数据有重叠，由于训练序列功率谱密度非常低，因此不会对实际数据造成干扰；再有，不同的训练序列可用不同的扩频码加以区分，不会造成混淆。训练序列不独占特定的频率和时间资源，提高了***的频谱利用率和传输速率。
在一个实施例中，本***中可以采用具有完备互补正交特性的LAS码为训练序列，其特点为自相关函数在原点是理想的冲击函数，原点以外处处为零，而互相关函数处处为零，LAS码的自相关和互相关特性如附图5所示。因此当训练序列重叠时也不会相互造成干扰。这样设计可以提高***的频谱利用率和传输速率。
本案例中我们设计训练序列的格式为：[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN。
本领域技术人员将可理解，信息、信号和数据可使用各种不同技术和技艺中的任何技术和技艺来表示。例如，以上描述通篇引述的数据、指令、命令、信息、信号、位(比特)、符号、和码片可由电压、电流、电磁波、磁场或磁粒子、光场或光学粒子、或其任何组合来表示。
本领域技术人员将进一步领会，结合本文中所公开的实施例来描述的各种解说性逻辑板块、模块、电路、和算法步骤可实现为电子硬件、计算机软件、或这两者的组合。为清楚地解说硬件与软件的这一可互换性，各种解说性组件、框、模块、电路、 和步骤在上面是以其功能性的形式作一般化描述的。此类功能性是被实现为硬件还是软件取决于具体应用和施加于整体***的设计约束。技术人员对于每种特定应用可用不同的方式来实现所描述的功能性，但这样的实现决策不应被解读成导致脱离了本发明的范围。
结合本文所公开的实施例描述的各种解说性逻辑模块、和电路可用通用处理器、数字信号处理器(DSP)、专用集成电路(ASIC)、现场可编程门阵列(FPGA)或其它可编程逻辑器件、分立的门或晶体管逻辑、分立的硬件组件、或其设计成执行本文所描述功能的任何组合来实现或执行。通用处理器可以是微处理器，但在替换方案中，该处理器可以是任何常规的处理器、控制器、微控制器、或状态机。处理器还可以被实现为计算设备的组合，例如DSP与微处理器的组合、多个微处理器、与DSP核心协作的一个或多个微处理器、或任何其他此类配置。
结合本文中公开的实施例描述的方法或算法的步骤可直接在硬件中、在由处理器执行的软件模块中、或在这两者的组合中体现。软件模块可驻留在RAM存储器、闪存、ROM存储器、EPROM存储器、EEPROM存储器、寄存器、硬盘、可移动盘、CD-ROM、或本领域中所知的任何其他形式的存储介质中。示例性存储介质耦合到处理器以使得该处理器能从/向该存储介质读取和写入信息。在替换方案中，存储介质可以被整合到处理器。处理器和存储介质可驻留在ASIC中。ASIC可驻留在用户终端中。在替换方案中，处理器和存储介质可作为分立组件驻留在用户终端中。
在一个或多个示例性实施例中，所描述的功能可在硬件、软件、固件或其任何组合中实现。如果在软件中实现为计算机程序产品，则各功能可以作为一条或更多条指令或代码存储在计算机可读介质上或藉其进行传送。计算机可读介质包括计算机存储介质和通信介质两者，其包括促成计算机程序从一地向另一地转移的任何介质。存储介质可以是能被计算机访问的任何可用介质。作为示例而非限定，这样的计算机可读介质可包括RAM、ROM、EEPROM、CD-ROM或其它光盘存储、磁盘存储或其它磁存储设备、或能被用来携带或存储指令或数据结构形式的合意程序代码且能被计算机访问的任何其它介质。任何连接也被正当地称为计算机可读介质。例如，如果软件是使用同轴电缆、光纤电缆、双绞线、数字订户线(DSL)、或诸如红外、无线电、以及微波之类的无线技术从web网站、服务器、或其它远程源传送而来，则该同轴电缆、光纤电缆、双绞线、DSL、或诸如红外、无线电、以及微波之类的无线技术就被包括在介质的定义之中。如本文中所使用的盘(disk)和碟(disc)包括压缩碟(CD)、激光碟、光碟、数字多用碟(DVD)、软盘和蓝光碟，其中盘(disk)往往以磁的方 式再现数据，而碟(disc)用激光以光学方式再现数据。上述的组合也应被包括在计算机可读介质的范围内。
提供对本公开的先前描述是为使得本领域任何技术人员皆能够制作或使用本公开。对本公开的各种修改对本领域技术人员来说都将是显而易见的，且本文中所定义的普适原理可被应用到其他变体而不会脱离本公开的精神或范围。由此，本公开并非旨在被限定于本文中所描述的示例和设计，而是应被授予与本文中所公开的原理和新颖性特征相一致的最广范围。

Claims (21)

  1. 一种载波同步方法，包括：
    对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中所述接收信号是来自发送端且包括基于训练码的训练序列；以及
    基于所述频偏对所述接收信号执行频偏校正。
  2. 如权利要求1所述的载波同步方法，其特征在于，所述发送端向所述接收端发送时域信号、频域信号、码分叠加信号或时频二维重叠信号。
  3. 如权利要求1所述的载波同步方法，其特征在于，所述提取自接收信号的两个训练码为LAS码。
  4. 如权利要求1所述的载波同步方法，其特征在于，所述接收信号包括训练序列和数据，其中训练序列频宽大于数据频宽且训练序列的功率谱密度低于数据的功率谱密度。
  5. 如权利要求4所述的载波同步方法，其特征在于，所述训练序列和所述数据是在频率和/或时间上至少部分重叠地发送的。
  6. 如权利要求3所述的的载波同步方法，其特征在于，所述训练序列包括两个LAS短码[Xlas]SN，SN为所述LAS短码的长度，其中，所述对提取自接收信号的两个LAS码执行互相关运算以获得接收端和发射端之间载波的频偏包括：
    对提取自所述接收信号的所述两个LAS短码执行互相关运算，以获得接收端和发射端之间载波的粗频偏，以及
    所述基于所述频偏对所述接收信号执行频偏校正包括：
    基于所述粗频偏对所述接收信号执行初次频偏校正。
  7. 如权利要求6所述的载波同步方法，其特征在于，所述训练序列包括[0]SN,[Xlas]SN,[0]SN,[Xlas]SN，其中[0]SN为长度为SN的0序列。
  8. 如权利要求6所述的载波同步方法，其特征在于，所述训练序列还包括两个 LAS长码[Xlas]LN，LN为所述LAS长码的长度，其中，所述对提取自接收信号的两个LAS码执行互相关运算以获得接收端和发射端之间载波的频偏还包括：
    对提取自经过所述初次频偏校正的接收信号的两个LAS长码执行互相关运算，以获得接收端和发射端之间载波的细频偏，以及
    所述基于所述频偏对所述接收信号执行频偏校正还包括：
    基于所述粗频偏和所述细频偏对经过所述初次频偏校正的接收信号执行二次频偏校正。
  9. 如权利要求6所述的载波同步方法，其特征在于，所述基于所述粗频偏和所述细频偏对经过所述初次频偏校正的接收信号执行二次频偏校正包括：
    基于所述粗频偏和所述细频偏的和来对经过所述初次频偏校正的接收信号执行所述二次频偏校正。
  10. 如权利要求6所述的载波同步方法，其特征在于，所述训练序列包括[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN，其中[0]SN为长度为SN的0序列。
  11. 如权利要求6所述的载波同步方法，其特征在于，所述两个LAS短码还被用于定时同步，所述两个LAS短码以及所述两个LAS长码的提取位置是基于定时同步的结果来确定的。
  12. 一种载波同步装置，包括：
    互相关计算单元，用于对提取自接收信号的两个训练码执行互相关运算，以获得接收端和发射端之间载波的频偏，其中所述接收信号是来自发送端且包括基于训练码的训练序列；以及
    频率校正单元，用于基于所述频偏对所述接收信号执行频偏校正。
  13. 如权利要求12所述的载波同步装置，其特征在于，所述发送端向所述接收端发送时域信号、频域信号、码分叠加信号或时频二维重叠信号。
  14. 如权利要求12所述的载波同步装置，其特征在于，所述提取自接收信号的两个训练码为LAS码。
  15. 如权利要求12所述的载波同步装置，其特征在于，所述接收信号包括训练序列和数据，其中训练序列频宽大于数据频宽且训练序列的功率谱密度低于数据的功率谱密度。
  16. 如权利要求15所述的载波同步装置，其特征在于，所述训练序列和所述数据是在频率和/或时间上至少部分重叠地发送的。
  17. 如权利要求12所述的载波同步装置，其特征在于，所述训练序列包括两个LAS短码[Xlas]SN，SN为所述LAS短码的长度，其中，所述相关计算单元进一步用于对提取自所述接收信号的所述两个LAS短码执行互相关运算，以获得接收端和发射端之间载波的粗频偏，以及
    所述频率校正单元进一步用于基于所述粗频偏对所述接收信号执行初次频偏校正。
  18. 如权利要求17所述的载波同步装置，其特征在于，所述训练序列包括[0]SN,[Xlas]SN,[0]SN,[Xlas]SN，其中[0]SN为长度为SN的0序列。
  19. 如权利要求17所述的载波同步装置，其特征在于，所述训练序列还包括两个LAS长码[Xlas]LN，LN为所述LAS长码的长度，其中，所述相关计算单元进一步用于对提取自经过所述初次频偏校正的接收信号的两个LAS长码执行互相关运算，以获得接收端和发射端之间载波的细频偏，以及
    所述频率校正单元进一步用于基于所述粗频偏和所述细频偏对经过所述初次频偏校正的接收信号执行二次频偏校正。
  20. 如权利要求19所述的载波同步装置，其特征在于，所述所述训练序列包括[0]SN,[Xlas]SN,[0]SN,[Xlas]SN,[Xlas]LN,[Xlas]LN，其中[0]SN为长度为SN的0序列。
  21. 如权利要求19所述的载波同步装置，其特征在于，所述两个LAS短码还被用于定时同步，所述两个LAS短码以及所述两个LAS长码的提取位置是基于定时同步的结果来确定的。
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