WO2014136485A1 - Motor control device and motor control method - Google Patents

Motor control device and motor control method Download PDF

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Publication number
WO2014136485A1
WO2014136485A1 PCT/JP2014/051284 JP2014051284W WO2014136485A1 WO 2014136485 A1 WO2014136485 A1 WO 2014136485A1 JP 2014051284 W JP2014051284 W JP 2014051284W WO 2014136485 A1 WO2014136485 A1 WO 2014136485A1
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Prior art keywords
inverter
waveform
pwm
motor
phase
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PCT/JP2014/051284
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French (fr)
Japanese (ja)
Inventor
浩二 柏瀬
月元　誠士
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カルソニックカンセイ株式会社
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Publication of WO2014136485A1 publication Critical patent/WO2014136485A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation

Definitions

  • the present invention relates to a motor control device and a motor control method for driving a multiphase AC motor.
  • the polyphase AC motor driven by polyphase AC is widely used in various industrial fields.
  • a car it is used as a power source for driving a compressor of an air conditioner, a power source of an electric car, and the like.
  • the current flowing through the shunt resistor is detected, the position of the rotor of the motor is estimated based on the detected current value, and the estimated position of the rotor is used. Thus, the timing of the drive signal input to the inverter is adjusted.
  • a predetermined time is required to reliably detect the current flowing through the shunt resistor.
  • the present invention has been made in view of the above circumstances.
  • the current flowing between the DC power supply and the inverter is accurately detected regardless of the time difference between the voltage signals of the phases constituting the multiphase AC.
  • the purpose is to do.
  • a motor control device and a motor control method according to an embodiment of the present invention accurately detect motor drive current and control motor rotation with high accuracy.
  • a motor control device includes a multiphase AC motor, an inverter composed of a plurality of switching elements that drive the multiphase AC motor, a DC power source that supplies a DC voltage to the inverter, Based on the current value detected by the current detection unit that detects the current flowing between the inverter and the DC power source, and a plurality of drive pulses that control conduction and non-conduction of the inverter, A PWM waveform generation unit that generates a pulse voltage by modulating a target voltage waveform such that the sum of instantaneous voltage values applied to each phase of the multiphase AC motor is 0, and the PWM waveform generation unit A PWM waveform correction unit that corrects the pulse widths of the plurality of PWM waveforms so that the time during which the inverter is conducted becomes equal to or greater than a predetermined value; Characterized in that it has a.
  • the PWM waveform generation unit has a sum of instantaneous voltage values applied to each phase of the multiphase AC motor based on the current value detected by the current detection unit as 0.
  • a plurality of PWM waveforms obtained by pulse-width-modulating each of the target voltage waveforms as described above, and the PWM waveform correcting unit corrects the pulse widths of the plurality of PWM waveforms thus generated to be equal to or longer than a predetermined time, Since the multiphase AC motor is driven by controlling the inverter connected to the DC power supply, the current flowing between the DC power supply and the inverter can be accurately detected by increasing the pulse width of the PWM waveform. Thus, the rotation of the multiphase AC motor can be controlled with high accuracy.
  • the motor control device and the motor control method when the motor rotates, the voltage between the DC power supply and the inverter is not affected by the time difference between the voltage signals of the phases constituting the multiphase AC. This makes it possible to accurately detect the current flowing through the motor, so that the rotation of the motor can be controlled with high accuracy.
  • FIG. 1 It is a block diagram which shows schematic structure of the Example of the motor control apparatus which concerns on one Embodiment of this invention. It is a figure explaining the three-phase alternating current voltage waveform for driving a three-phase alternating current motor. It is a figure explaining the principle of operation of a three-phase AC motor. Three-phase AC voltage waveforms E U , E V , E W for driving the three-phase AC motor, a correction voltage Ec generated to correct the three-phase AC voltage waveform, and correction voltage waveforms E ′ U , E ′ V , E ′ W is a diagram showing an example.
  • FIG. 6 is a diagram illustrating a pulse waveform for generating a current waveform and a current waveform flowing between a DC power supply and an inverter.
  • the motor control device 10 to which the present invention is applied has a multi-phase AC motor 20 that rotates by inputting a multi-phase AC voltage, and a multi-phase AC motor 20 that has a switching function.
  • An inverter 30 that supplies a drive voltage signal, a drive pulse generator 40 that generates a drive pulse signal supplied to switch the inverter 30, and a plurality of drive pulses generated by the drive pulse generator 40
  • a shunt resistor 70 for detection is provided.
  • the polyphase AC motor 20 further includes a stator 22 wound around an iron core and a permanent magnet having a predetermined magnetic pole, and the inside of the stator 22 can be rotated around a predetermined axis.
  • the rotor 24 is held by the rotor 24. The rotational force of the rotor 24 gives torque to the load connected to the multiphase AC motor 20.
  • the multiphase AC motor 20 is driven by a three-phase AC composed of a U phase, a V phase, and a W phase.
  • the stator 22 further includes a winding 22U, a winding 22V, and a winding 22W. One end of these windings is connected to each other at a neutral point 22N, and the other end is connected to a terminal 25U, respectively. It is connected to 25V and 25W.
  • the inverter 30 further includes switching elements such as IGBT (Insulated Gate Bipolar Transistor) and FET (Field Effect Transistor), and includes a switching element for U phase, a switching element for V phase, and a switching element for W phase.
  • the switching element for the U phase is composed of a switching element 32U on the high voltage side (hereinafter referred to as U + phase) and a switching element 34U on the low voltage side (hereinafter referred to as U-phase).
  • a switching element 32V on the side hereeinafter referred to as V + phase
  • V-phase switching element 34V on the low voltage side
  • the switching element for the W phase is on the high voltage side (hereinafter referred to as W + phase).
  • Switching element 32W and a switching element 34W on the low voltage side hereinafter referred to as W-phase).
  • the switching elements 32U, 34U, 32V, 34V, 32W, and 34W include diodes 32D U , 34D U , 32D V , and 34D having forward directions from the low voltage side to the high voltage side of the DC power supply 60, respectively. V , 32D W and 34D W are connected. These diodes function as so-called reflux diodes.
  • connection point between the switching element 32U and the switching element 34U is connected to the terminal 25U
  • connection point between the switching element 32V and the switching element 34V is connected to the terminal 25V
  • connection point between the switching element 32W and the switching element 34W is It is connected to the terminal 25W.
  • the driving pulse generation unit 40 based on a microcomputer, a current detecting unit 42 for detecting the shunt current i R flowing to the shunt resistor 70, the target voltage for generating a 3-phase AC voltage waveform applied to the multiphase AC motor 20
  • Target voltage waveform correction that corrects the voltage waveform generated by the target voltage waveform generation unit 44 to increase the fundamental wave component in order to reduce torque fluctuations when the waveform generation unit 44 and the multiphase AC motor 20 are driven.
  • a PWM waveform generation unit 46 that generates a pulse waveform PWM (pulse width modulated) from the voltage waveform generated by the target voltage waveform correction unit 45, and a pulse waveform generated by the PWM waveform generation unit 46 as a current the detection unit 42 corrects the pulse waveform can be reliably detected shunt current, also in accordance with the shunt current i R detected by the current detector 42,
  • the WM waveform consisting PWM waveform correction unit 48 to adjust the timing for supplying the inverter 30.
  • FIG. 2B shows the positional relationship between the direction of the current flowing through the windings of the multiphase AC motor 20 and the rotor 24 for each phase of the three-phase AC voltage waveform.
  • the switching element 34U constituting the U-phase is turned on, and the multi-phase AC motor 20 is The driving current flows from the neutral point 22N through the winding 22U and flows into the switching element 34U through the terminal 25U.
  • this current flows from the winding terminal 22Ub on the neutral point 22N side, It is expressed as flowing in the direction of flowing out from the winding terminal 22Ua on the terminal 25U side.
  • the winding terminals 22Va and 22Wb represent the winding terminals on the neutral point 22N side
  • the winding terminal 22Vb is the terminal 25V side
  • the winding terminal 22Wa is the winding on the terminal 25W side. Represents a terminal.
  • the magnitude and direction of the current flowing through each winding changes according to the change in the phase of the three-phase AC voltage waveform, so that a rotating magnetic field is generated in the stator 22 by electromagnetic induction, The magnetic poles of the rotor 24 are attracted and repelled by the rotating magnetic field generated in the stator 22, and the multi-phase AC motor 20 is rotated by changing its direction.
  • one cycle of the sine wave is divided into six equal parts based on the magnitude relationship of the voltages of the three-phase AC voltage waveforms E U , E V , and E W.
  • the sections thus divided into six are referred to as mode0, mode1, mode2, mode3, mode4, and mode5, respectively.
  • the section of mode 0 (t C ⁇ t ⁇ t D ) has a relationship of E U > E V > E W and the section of mode 1 (t D ⁇ t ⁇ t E ) has a relationship of E V > E U > E W
  • a mode2 interval (t E ⁇ t ⁇ t F ) has a relationship of E V > E W > E U
  • a mode 3 interval (0 ⁇ t ⁇ t A ) has a relationship of E W > E V > E U
  • a section of mode 4 (t A ⁇ t ⁇ t B ) has a relationship of E W > E U > E V.
  • mode 5 of the section (t B ⁇ t ⁇ t C ) is, E U> assumed to have a relation of E W> E V. That is, the change point of each mode is a point where the phase order of the three-phase AC voltage waveform changes.
  • a value that is half the intermediate value of the three voltage values of the three-phase AC voltage waveform is defined as a correction voltage Ec.
  • Ec E V / 2 in mode 0 and mode 3
  • Ec E U / 2 in mode 1 and mode 4
  • Ec E W / 2 in mode 2 and mode 5.
  • the correction voltage Ec thus calculated is shown in the middle graph of FIG.
  • the three-phase AC voltage waveforms E U , E V , E W are corrected to be corrected voltage waveforms E ′ U , E ′ V , E ′ W , so
  • the maximum value of the fundamental wave component of the applied voltage can be increased without changing the voltage. That is, since a line voltage equivalent to the conventional one can be obtained with a small voltage waveform, the voltage control range can be expanded.
  • the multiphase AC motor 20 is not actually driven directly by the voltage waveform described above. That is, actually, the multiphase AC motor 20 is driven by generating a pulse waveform having a pulse width corresponding to the voltage value of the voltage waveform and controlling the conduction and non-conduction of the inverter by this pulse wave.
  • generating a pulse waveform having a pulse width corresponding to the voltage value of the voltage waveform is referred to as PWM (Pulse Width Modulation).
  • PWM Pulse Width Modulation
  • the uppermost graph in FIG. 4 shows the waveform over one period Tc of the carrier wave C (t) used for pulse width modulation and the corrected voltage waveforms E ′ U , E ′ V , E ′ W in the vicinity of time ta.
  • the voltage value is shown.
  • the horizontal axis of FIG. 4 indicates the time axis, and a very short time range in the vicinity of the time ta shown in FIG. 3 is shown in FIG. Since the time ta is in the range of mode 0, the correction voltage waveform has a relationship of E ′ U > E ′ V > E ′ W.
  • a PWM waveform U ⁇ obtained by inverting the phase of the PWM waveform U + is generated as a PWM waveform U ⁇ for controlling the switching element 34U. If the PWM waveform U ⁇ is generated simply by inverting the phase of the PWM waveform U +, the timing at which the switching element 32U is turned on coincides with the timing at which the switching element 34U is turned off. Then, there is a possibility that the switching element 32U and the switching element 34U become conductive at the same time. In such a state, a short-circuit current flows between the switching element 32U and the switching element 34U, which causes excessive heat generation and, in the worst case, damage to the circuit.
  • a time difference is provided between the rising time of the high-voltage side pulse (for example, PWM waveform U +) and the falling time of the low-voltage side pulse (for example, PWM waveform U ⁇ ) to Both low voltage side switching elements (for example, 34U) ensure the time when it becomes non-conductive.
  • This predetermined time difference is called dead time DT.
  • a time of about several ⁇ sec is set as the dead time DT.
  • a time corresponding to the dead time DT is given to the PWM waveform U ⁇ that is a low-pressure side pulse.
  • a PWM waveform V + for controlling the switching element 32V, a PWM waveform V ⁇ for controlling the switching element 34V, a PWM waveform W + for controlling the switching element 32W, and a switching element 34W A PWM waveform W ⁇ to be controlled is generated.
  • the PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, W ⁇ generated in this way are applied to the corresponding switching elements, and each switching element repeats conduction and non-conduction sequentially.
  • the shunt resistor 70 flows shunt current i R shown at the bottom of the graph of FIG.
  • the line voltage of E ′ U and E ′ V (corresponding to the line voltage ⁇ a shown in FIG. 3) and the line voltage of E ′ V and E ′ W (line voltage ⁇ b shown in FIG.
  • the line voltage between E ′ U and the maximum value of the correction voltage waveform (corresponding to the line voltage ⁇ c shown in FIG. 3), and the line voltage between E ′ W and the minimum value of the correction voltage waveform (
  • the ratio of the magnitude of the line voltage ⁇ d shown in FIG. 3 is as follows: time t1 and time t2, time t2 and time t3, time t0 and time t1, time t3 and time t4, respectively. Correspond to each of the intervals.
  • FIGS. 5A to 5E show an example of generating a PWM waveform in a time range other than mode 3 and an outline of the shunt current i R flowing through the shunt resistor 70.
  • FIG. As shown in FIGS. 5A to 5E, since the voltage values of the correction voltage waveforms E ′ U , E ′ V , E ′ W change with time, the pulse width of the PWM waveform changes accordingly, and the shunt current i R The waveform of changes.
  • the waveform of the shunt current i R is uniquely determined in accordance with the state of the correction voltage waveform E 'U, E' V, E 'W. That is, if it is possible to measure the magnitude of the shunt current i R, can be estimated at this time, a position of the rotator 24 of the multi-phase AC motor 20.
  • the phases of the correction voltage waveforms E ′ U , E ′ V , E ′ W for controlling the switching elements are controlled, and a PWM waveform is generated accordingly. By re-doing, predetermined motor control can be realized.
  • the voltage values of two different phases approach each other at the switching position of adjacent modes. For example, at the position where the mode 0 is switched to the mode 1, the voltage value of the correction voltage E ′ U matches the voltage value of the correction voltage E ′ V. Further, at the position where the mode 1 is switched to the mode 2, the voltage value of the correction voltage E ′ U matches the voltage value of the correction voltage E ′ W.
  • the PWM waveform U + and the PWM waveform V + coincide with each other during the time when the mode 0 is switched to the mode 1. Therefore, the interval between times t1 and t2 and the interval between times t5 and t6 shown in the lowermost graph of FIG. 4 are gradually shortened.
  • the interval between time t1 and time t2 and the interval between time t5 and time t6 are both minimum. since falls below the time T min, it becomes impossible to detect the shunt current i R over the intervening time.
  • FIG. 7A is a diagram for explaining a change in the magnitude of the current i flowing through the U-phase winding 22U.
  • the state in which the current i flows from the switching element 32U toward the winding 22U (the current i at this time)
  • the orientation is i> 0).
  • the switching element 34U is in a non-conductive state and the switching element 32U is in a conductive state
  • the current i flows through the current path r1 through the switching element 32U.
  • the switching element 32U is nonconducting, be left 34U switching element is in a non-conducting state
  • the current i passes through the diode 34D U, flows through the current path r2.
  • FIG. 7B shows a state in which the current i flows from the winding 22U toward the switching element 32U (the direction of the current i at this time is i ⁇ 0).
  • 34U switching element becomes non-conducting state
  • the switching element 32U is made conductive
  • current i passes through the diode 32D U
  • the switching element 32U is nonconducting, be left 34U switching element is in a non-conducting state
  • the current i passes through the diode 32D U, flows through the current path r3.
  • shunt current i R flowing to the shunt resistor 70 is not changed.
  • timing the magnitude of the shunt current i R is small is different. As shown in FIGS. 7A, 7B, ⁇ If 0, the timing at which the magnitude of the shunt current i R becomes smaller, i> direction of the current i is i slower than when zero. This applies not only to the U-phase winding 22U but also to the V-phase winding 22V and the W-phase winding 22W.
  • the direction of the current i flowing through the windings 22U, 22V, and 22W is determined by the phase difference between the voltage applied to each phase and the current flowing through that phase.
  • the direction of the current i becomes i ⁇ 0.
  • the time for which the switching element is turned on at the time when the phase sequence changes is shunt current i R. since falls below a minimum time T min needed to detect, it becomes impossible to detect the shunt current i R.
  • FIG. 8 is a graph showing the operation range of the multiphase AC motor 20.
  • the horizontal axis of the graph represents the rotational speed P of the multiphase AC motor 20, and the vertical axis represents the torque T generated by the multiphase AC motor 20.
  • the multiphase AC motor 20 operates normally within the operation region R1 surrounded by the operation limit line C shown in FIG.
  • the unstable region R2 can be used as the operation region R1, for example, by using a multiphase AC motor having a larger capacity.
  • the cost increases and the motor size increases, so that the installation property deteriorates.
  • FIG. 9 shows a PWM waveform at the moment of transition from mode 0 to mode 1 described above.
  • a carrier wave C (t) used for pulse width modulation PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, W ⁇ for driving each switching element, and envelope of shunt current i R to be detected is shown.
  • the dead time DT or more is between the rise time of the PWM waveform V ⁇ and the fall time of the PWM waveform V +, and between the fall time of the PWM waveform V ⁇ and the rise time of the PWM waveform V +. I need time. Further, in order to reliably detect the shunt current i R needs time interval PWM waveform V- is in Hi level is minimum time T min or more.
  • tm 1 + tm 2 which is the time (OFF time) that the PWM waveform V + on the high-voltage side of the V phase is at the Lo level in the time to shift from mode 0 to mode 1. Since the time (referred to as time tm) is short, the time during which the PWM waveform V- is at the Hi level (ON time) is less than the minimum time Tmin . That is, the relationship of (Expression 14) is established.
  • tm ( tm 1 + tm 2 ) ⁇ 2DT + T min (Formula 14)
  • the time tm 2 of (Equation 14) is the time from when the PWM waveform V + becomes Lo level until the mode changes, as shown in FIG. 9, and the time tm 1 changes the mode. This is the time until the PWM waveform V + subsequently changes from the Lo level to the Hi level.
  • the time interval of the section in which the PWM waveform V- is at the Hi level is extended to at least the minimum time T min . This is done by shifting the position of point P 3 in FIG. 9 to the position of point P 3 ′.
  • the amount of time shifted at this time is represented by a correction amount k shown in FIG.
  • the PWM correction waveform V ⁇ ′ is a waveform in which the time interval of the Hi level section is the minimum time T min and has a Lo level section corresponding to the dead time DT on both sides thereof. That is, (Equation 15) is established in the PWM correction waveform V ⁇ ′.
  • the rise time of the pulse from point P 1 points P 1' PWM correction waveform V + due to moved to the position of, in response thereto, the pulse fall time of the PWM correction waveform V + ', PWM waveform V + standing of moves from point P 2 is a fall time to the position of the point P 2 '.
  • the movement amount at this time is equal to the correction amount k.
  • the duty ratio of the waveform changes. Therefore, the duty ratios of the other phases are also V-phase duty ratios. To match.
  • narrow PWM waveform U + is by the correction amount k also influences the time interval of the section (section between the point P 4 and the point P 5) in the Hi level, the point P 4 'and the point P 5 It is corrected to the position of “to obtain a PWM correction waveform U +”.
  • the time interval of the section at the Lo level is narrowed by the correction amount k on both the left and right sides, so that the points P 6 ′ and P 7 ′ The position is corrected to a PWM correction waveform U- '.
  • narrow by the correction amount k also influences the time interval (the interval between the points P 8 and the point P 9) PWM waveform W + is at the Hi level section, the point and the point P 8 'P 9 It is corrected to the position of “to obtain a PWM correction waveform W +”. Also for the PWM waveform W ⁇ , the time interval of the section at the Lo level (section between the points P 10 and P 11 ) is narrowed by the correction amount k on both the left and right sides, and the points P 10 ′ and P 11 ′ The position is corrected to a PWM correction waveform W ⁇ ′.
  • a similar correction is performed in the same manner for all scenes to which the mode shifts.
  • the PWM waveform V ⁇ is first corrected.
  • the waveform to be corrected first is any one of the PWM waveforms U ⁇ , V ⁇ , and W ⁇ according to the scene in which the mode shifts. become.
  • the PWM waveform in which the times tm 1 and tm 2 are defined is also changed each time.
  • the target voltage waveform generation unit 44 generates three-phase AC voltage waveforms E U , E V , and E W that are target voltages for driving the multiphase AC motor 20.
  • the waveform is generated by software stored in advance in the target voltage waveform generator 44, for example.
  • Step S102 the target voltage waveform correction unit 45 corrects the previously generated target voltage waveform.
  • the purpose of the correction and the correction method are as described above.
  • correction voltage waveforms E ′ U , E ′ V , E ′ W are obtained.
  • Step S104 the PWM waveform generator 46 performs pulse width modulation on the previously generated correction voltage waveforms E ′ U , E ′ V , E ′ W , and PWM waveforms U +, U ⁇ , V +, V ⁇ . , W +, W ⁇ .
  • the method of pulse width modulation is as described above.
  • the PWM waveform correction unit 48 corrects the PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, and W ⁇ so that the shunt current i R can be reliably detected, and the PWM correction waveform.
  • U + ', U-', V + ', V-', W + ', W-' are generated.
  • the method for correcting the PWM waveform is as described above.
  • correction of the PWM waveform is not necessary, correction is not performed.
  • the determination as to whether correction is necessary is made based on the above-described (Equation 14).
  • Step S106 the PWM correction waveforms U + ', U-', V + ', V-', W + ', W-' are input to the drive circuit 50 and further input to the corresponding switching elements, The conduction and non-conduction of the switching element corresponding to each PWM correction waveform is controlled.
  • the drive circuit 50 also serves as an interface between the drive pulse generation unit 40 configured by a microcomputer and the inverter 30 configured by a switching element such as an IGBT.
  • Step S108 The voltage output from the DC power supply 60 is three-phase AC by the inverter 30 that is intermittent at the timing determined by the PWM correction waveforms U + ', U-', V + ', V-', W + ', W-'.
  • the three-phase AC voltage waveform converted into a voltage waveform and thus generated is applied to the multi-phase AC motor 20, and the multi-phase AC motor 20 rotates.
  • Step S110 The voltage value applied to both ends of the shunt resistor 70 having the resistance value R is read into the current detection unit 42 and A / D converted inside the current detection unit 42, and thus the A / D converted voltage. by dividing the value by the resistance value R of the shunt resistor 70, the value of the shunt current i R flowing through the shunt resistor 70 is read.
  • Step S112 current detector 42, when the shunt current i R has been read, it is possible to estimate the position of the rotor 24 of the polyphase alternating current motor 20, as it continues the control of the motor. On the other hand, the current detector 42, when the unreadable shunt current i R proceeds to step S116.
  • Step S114 It is determined whether or not there is a stop command for the multiphase AC motor 20. If there is a stop command for the multiphase AC motor 20, the supply of the PWM correction pulse is stopped and the rotation of the motor is stopped. On the other hand, when there is no stop command for the multiphase AC motor 20, the process returns to step S106 and the motor control is continued.
  • Step S116 When the position of the rotor 24 of the multiphase AC motor 20 cannot be estimated, the PWM waveform correction unit 48 supplies the PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, W ⁇ to the inverter 30. Adjust timing.
  • Step S118 It is determined whether or not there is a stop command for the multiphase AC motor 20. If there is a stop command for the multiphase AC motor 20, the supply of the PWM correction pulse is stopped and the rotation of the motor is stopped. On the other hand, when there is no stop command for the multiphase AC motor 20, the process returns to step S102 to continue the motor control.
  • the PWM waveform generation unit 46 is applied to each phase of the multiphase AC motor 20 based on the current value detected by the current detection unit 42.
  • correction is made so that the pulse widths of the plurality of PWM waveforms generated in this way are longer than a predetermined time to obtain PWM correction waveforms U + ', U-', V + ', V-', W + ', W-', and direct current
  • the pulse width of the PWM waveform U +, U ⁇ , V +, V ⁇ , W +, W ⁇ is corrected to be long, and the DC A flow between
  • the PWM waveform correction unit 48 generates a plurality of PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, W ⁇ generated by the PWM waveform generation unit 46.
  • the time for which the inverter 30 is conducted to be longer than the minimum time T min required to detect the shunt current i R flowing between the inverter 30 and the DC power source 60 by the current detection unit 42 power source 60 and can be reliably detected shunt currents i R flowing between the inverter 30, thereby, it can be controlled with high precision the rotation of the multi-phase AC motor 20.
  • the PWM waveform correction unit 48 generates a plurality of PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, W ⁇ generated by the PWM waveform generation unit 46.
  • the time during which the inverter 30 is turned on is the OFF time of the PWM waveform on the high-voltage side of a phase applied to the multiphase AC motor 20, and the high-voltage side inverter and the low-voltage side inverter of the inverter 30 are not turned on simultaneously.
  • the PWM waveform correction unit 48 includes a plurality of PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, supplied from the inverter 30 to the multiphase AC motor 20. At the time when the phase order of W ⁇ changes, a plurality of PWM waveforms U +, U ⁇ , V +, V ⁇ , W +, W ⁇ generated by the PWM waveform generator 46 are corrected to generate PWM corrected waveforms U + ′, U ⁇ .
  • the current detection unit 42 detects the shunt current i R flowing through the shunt resistor 70 inserted in series between the inverter 30 and the DC power source 60. It is possible to detect the current i R easy.
  • the target voltage waveform correction unit 45 applies the three-phase AC voltage waveforms E U , E V , and E W applied to the respective phases of the multiphase AC motor 20 to the respective phases. Since the waveform obtained by adding half the median value of the instantaneous voltage value of the phase is the corrected voltage waveform E ′ U , E ′ V , E ′ W , the basics of the applied voltage without changing the line voltage between different phases The maximum value of the wave component can be further increased. That is, since a line voltage equivalent to the conventional one can be obtained with a small voltage waveform, the voltage control range can be expanded.
  • the DC power supply 60 and the inverter 30 are the shunt current i R flows in can be accurately detected, thereby, it is possible to control the rotation of the multi-phase AC motor 20 with high accuracy.

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  • Engineering & Computer Science (AREA)
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  • Control Of Motors That Do Not Use Commutators (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The present invention reliably detects a shunt current and stably controls a multi-phase AC motor. A PWM waveform generator (46) generates, on the basis of a current value detected by a current detector (42), a plurality of PWM waveforms by modulating the pulse widths of target voltage waveforms for which the total of the instantaneous voltages applied to each of the phases of a multi-phase motor (20) is zero. A PWM waveform correction unit (48) corrects the multiple PWM waveforms generated in the aforementioned manner so that the pulse widths thereof exceed a predetermined time, controls an inverter (30) connected to a DC power supply (60), and drives the multi-phase motor (20).

Description

モータ制御装置およびモータ制御方法Motor control device and motor control method
　この発明は、多相交流モータを駆動するモータ制御装置およびモータ制御方法に関するものである。 The present invention relates to a motor control device and a motor control method for driving a multiphase AC motor.
　多相交流によって駆動される多相交流モータは、様々な産業分野で広く利用されている。例えば自動車にあっては、空調装置のコンプレッサを駆動するための動力源や、電気自動車の動力源等に用いられている。 The polyphase AC motor driven by polyphase AC is widely used in various industrial fields. For example, in a car, it is used as a power source for driving a compressor of an air conditioner, a power source of an electric car, and the like.
　そして、このような多相交流モータの回転を制御するモータ制御装置にあっては、インバータと直流電源の間を流れる電流を検出して、検出された電流値に基づいて、モータの固定子を構成する各相に流れる電流を推定し、さらに、モータの回転子の位置を推定して、こうして推定された回転子の位置に対応するインバータの駆動信号を生成して、この駆動信号を用いてモータを回転させるモータ制御方法が採られている（例えば、特許文献１参照）。 In such a motor control device that controls the rotation of the multiphase AC motor, the current flowing between the inverter and the DC power source is detected, and the stator of the motor is adjusted based on the detected current value. Estimate the current that flows in each phase that constitutes, further estimate the position of the rotor of the motor, generate an inverter drive signal corresponding to the estimated rotor position, and use this drive signal A motor control method for rotating the motor is employed (see, for example, Patent Document 1).
特開２００７－６４５５２号公報JP 2007-64552 A
　特許文献１に記載されたモータ駆動装置では、シャント抵抗に流れる電流を検出して、検出された電流値に基づいてモータの回転子の位置を推定して、推定された回転子の位置に応じて、インバータに入力する駆動信号のタイミングを調整している。 In the motor driving device described in Patent Document 1, the current flowing through the shunt resistor is detected, the position of the rotor of the motor is estimated based on the detected current value, and the estimated position of the rotor is used. Thus, the timing of the drive signal input to the inverter is adjusted.
　このようなシャント抵抗を流れる電流に基づいてモータの回転を制御するモータ制御方法によると、シャント抵抗を流れる電流を確実に検出するためには、所定の時間が必要になる。 According to the motor control method for controlling the rotation of the motor based on the current flowing through the shunt resistor, a predetermined time is required to reliably detect the current flowing through the shunt resistor.
　しかしながら、モータを駆動する多相交流信号のうちいずれか２相の電圧値が接近したときには、その２相の電圧信号に基づいて生成されたパルス幅変調（ＰＷＭ）された２つのパルス波形の時間差が小さくなるため、その２相の時間差に相当する時間に亘って、シャント抵抗を流れる電流を検出することができなくなる。そして、当該２相の時間ずれの間、多相交流モータの各相の巻線を流れる電流を推定することができなくなるため回転子の位置を推定することができず、これによって、モータを滑らかに回転させる制御を行うことができなくなってしまうという問題があった。 However, when the voltage value of any two phases of the multiphase AC signal that drives the motor approaches, the time difference between the two pulse waveforms that are pulse-width modulated (PWM) generated based on the two-phase voltage signals Therefore, the current flowing through the shunt resistor cannot be detected over a time corresponding to the time difference between the two phases. During the time difference between the two phases, the current flowing through the windings of each phase of the multiphase AC motor cannot be estimated, so that the position of the rotor cannot be estimated. There is a problem that it becomes impossible to perform control to rotate the motor.
　本発明は上記事情に鑑みなされたもので、モータが回転したときに、多相交流を構成する各相の電圧信号の時間差によらずに、直流電源とインバータの間に流れる電流を正確に検出することを目的とする。 The present invention has been made in view of the above circumstances. When the motor rotates, the current flowing between the DC power supply and the inverter is accurately detected regardless of the time difference between the voltage signals of the phases constituting the multiphase AC. The purpose is to do.
　本発明の一実施形態に係るモータ制御装置およびモータ制御方法は、モータの駆動電流を正確に検出して、モータの回転を高い精度で制御するものである。 A motor control device and a motor control method according to an embodiment of the present invention accurately detect motor drive current and control motor rotation with high accuracy.
　すなわち、本発明の一実施形態に係るモータ制御装置は、多相交流モータと、前記多相交流モータを駆動する複数のスイッチング素子からなるインバータと、前記インバータに直流電圧を供給する直流電源と、前記インバータと前記直流電源の間に流れる電流を検出する電流検出部と、前記電流検出部によって検出された電流値に基づいて、前記インバータの導通、および非導通を制御する複数の駆動パルスを、前記多相交流モータの各相に印加される瞬時電圧値の総和が０となるような目標電圧波形をそれぞれパルス幅変調することによって生成するＰＷＭ波形生成部と、前記ＰＷＭ波形生成部によって生成された複数のＰＷＭ波形のパルス幅を、前記インバータを導通させる時間が所定値以上になるように補正するＰＷＭ波形補正部と、を有することを特徴とする。 That is, a motor control device according to an embodiment of the present invention includes a multiphase AC motor, an inverter composed of a plurality of switching elements that drive the multiphase AC motor, a DC power source that supplies a DC voltage to the inverter, Based on the current value detected by the current detection unit that detects the current flowing between the inverter and the DC power source, and a plurality of drive pulses that control conduction and non-conduction of the inverter, A PWM waveform generation unit that generates a pulse voltage by modulating a target voltage waveform such that the sum of instantaneous voltage values applied to each phase of the multiphase AC motor is 0, and the PWM waveform generation unit A PWM waveform correction unit that corrects the pulse widths of the plurality of PWM waveforms so that the time during which the inverter is conducted becomes equal to or greater than a predetermined value; Characterized in that it has a.
　このように構成されたモータ制御装置によれば、ＰＷＭ波形生成部が、電流検出部が検出した電流値に基づいて、多相交流モータの各相に印加される瞬時電圧値の総和が０となるような目標電圧波形をそれぞれパルス幅変調した複数のＰＷＭ波形を生成して、ＰＷＭ波形補正部が、こうして生成された複数のＰＷＭ波形のパルス幅が所定時間以上になるように補正して、直流電源が接続されたインバータを制御して多相交流モータを駆動するため、ＰＷＭ波形のパルス幅が長くなることによって、直流電源とインバータの間に流れる電流を正確に検出することができ、これによって、多相交流モータの回転を高い精度で制御することができる。 According to the motor control device configured as described above, the PWM waveform generation unit has a sum of instantaneous voltage values applied to each phase of the multiphase AC motor based on the current value detected by the current detection unit as 0. A plurality of PWM waveforms obtained by pulse-width-modulating each of the target voltage waveforms as described above, and the PWM waveform correcting unit corrects the pulse widths of the plurality of PWM waveforms thus generated to be equal to or longer than a predetermined time, Since the multiphase AC motor is driven by controlling the inverter connected to the DC power supply, the current flowing between the DC power supply and the inverter can be accurately detected by increasing the pulse width of the PWM waveform. Thus, the rotation of the multiphase AC motor can be controlled with high accuracy.
　本発明の一実施形態に係るモータ制御装置およびモータ制御方法によれば、モータが回転したときに、多相交流を構成する各相の電圧信号の時間差によらずに、直流電源とインバータの間に流れる電流を正確に検出することができるため、モータの回転を高い精度で制御することができるという効果が得られる。 According to the motor control device and the motor control method according to an embodiment of the present invention, when the motor rotates, the voltage between the DC power supply and the inverter is not affected by the time difference between the voltage signals of the phases constituting the multiphase AC. This makes it possible to accurately detect the current flowing through the motor, so that the rotation of the motor can be controlled with high accuracy.
本発明の一実施形態に係るモータ制御装置の実施例の概略構成を示すブロック図である。It is a block diagram which shows schematic structure of the Example of the motor control apparatus which concerns on one Embodiment of this invention. ３相交流モータを駆動するための３相交流電圧波形を説明する図である。It is a figure explaining the three-phase alternating current voltage waveform for driving a three-phase alternating current motor. ３相交流モータの動作原理を説明する図である。It is a figure explaining the principle of operation of a three-phase AC motor. ３相交流モータを駆動する３相交流電圧波形ＥＵ，ＥＶ，ＥＷと、３相交流電圧波形を補正するために生成される補正電圧Ｅｃと、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの一例を示す図である。Three-phase AC voltage waveforms E U , E V , E W for driving the three-phase AC motor, a correction voltage Ec generated to correct the three-phase AC voltage waveform, and correction voltage waveforms E ′ U , E ′ V , E ′ W is a diagram showing an example. ３相交流電圧波形の相順がＵＶＷであるときの、インバータの導通、非導通を制御するパルスの生成方法について説明する図であり、上部から順に、パルス幅変調を行う搬送波の波形と、Ｕ相の巻線に印加される電圧波形を生成するためのパルス波形と、Ｖ相の巻線に印加される電圧波形を生成するためのパルス波形と、Ｗ相の巻線に印加される電圧波形を生成するためのパルス波形と、直流電源とインバータの間に流れる電流波形と、をそれぞれ示す図である。It is a figure explaining the production | generation method of the pulse which controls conduction | electrical_connection and non-conduction of an inverter when the phase sequence of a three-phase alternating current voltage waveform is UVW, The waveform of the carrier wave which performs a pulse width modulation, and U A pulse waveform for generating a voltage waveform applied to a phase winding, a pulse waveform for generating a voltage waveform applied to a V phase winding, and a voltage waveform applied to a W phase winding FIG. 6 is a diagram illustrating a pulse waveform for generating a current waveform and a current waveform flowing between a DC power supply and an inverter. ３相交流電圧波形の相順がＶＵＷであるときに、巻線に印加される電圧波形を生成するためのパルス波形と、電流波形を示す図である。It is a figure which shows the pulse waveform for producing | generating the voltage waveform applied to a coil | winding, and an electric current waveform when the phase sequence of a three-phase alternating current voltage waveform is VUW. ３相交流電圧波形の相順がＶＷＵであるときに、巻線に印加される電圧波形を生成するためのパルス波形と、電流波形を示す図である。It is a figure which shows the pulse waveform and electric current waveform for producing | generating the voltage waveform applied to a coil | winding, when the phase order of a three-phase alternating current voltage waveform is VWU. ３相交流電圧波形の相順がＷＶＵであるときに、巻線に印加される電圧波形を生成するためのパルス波形と、電流波形を示す図である。It is a figure which shows the pulse waveform for producing | generating the voltage waveform applied to a coil | winding, and an electric current waveform when the phase sequence of a three-phase alternating current voltage waveform is WVU. ３相交流電圧波形の相順がＷＵＶであるときに、巻線に印加される電圧波形を生成するためのパルス波形と、電流波形を示す図である。It is a figure which shows the pulse waveform and electric current waveform for producing | generating the voltage waveform applied to a coil | winding when the phase sequence of a three-phase alternating current voltage waveform is WUV. ３相交流電圧波形の相順がＵＷＶであるときに、巻線に印加される電圧波形を生成するためのパルス波形と、電流波形を示す図である。It is a figure which shows the pulse waveform for producing | generating the voltage waveform applied to a coil | winding, and an electric current waveform when the phase order of a three-phase alternating current voltage waveform is UWV. Ｕ相とＶ相の電圧が一致したときに、巻線に印加される電圧波形を生成するためのパルス波形と電流波形を示す図である。It is a figure which shows the pulse waveform and electric current waveform for producing | generating the voltage waveform applied to a coil | winding, when the voltage of U phase and V phase correspond. モータに流れる電流の向きについて説明する第１の図である。It is a 1st figure explaining the direction of the electric current which flows into a motor. モータに流れる電流の向きについて説明する第２の図である。It is a 2nd figure explaining the direction of the electric current which flows into a motor. 多相交流モータの運転条件について説明する図である。It is a figure explaining the driving | running conditions of a polyphase alternating current motor. インバータの導通、非導通を制御するパルス波形の補正方法について説明する図である。It is a figure explaining the correction method of the pulse waveform which controls conduction and non-conduction of an inverter. 本発明の一実施形態に係るモータ制御装置におけるモータ制御の手順を示すフローチャートである。It is a flowchart which shows the procedure of the motor control in the motor control apparatus which concerns on one Embodiment of this invention.
　以下、本発明の一実施形態に係るモータ制御装置およびモータ制御方法の実施例について、図面を参照して説明する。 Hereinafter, examples of a motor control device and a motor control method according to an embodiment of the present invention will be described with reference to the drawings.
　以下、本発明の第１の実施例を、図面を用いて説明する。本発明を適用したモータ制御装置１０は、図１に示すように、多相交流電圧を入力することによって回転する多相交流モータ２０と、スイッチング機能を有して、多相交流モータ２０に対して駆動電圧信号を供給するインバータ３０と、インバータ３０をスイッチングするために供給される駆動パルス信号を生成する駆動パルス生成部４０と、駆動パルス生成部４０で生成された複数の駆動パルスをインバータ３０に印加するドライブ回路５０と、インバータ３０に所定の直流電圧を供給する直流電源６０と、インバータ３０と直流電源６０の間に直列に挿入されて、インバータ３０と直流電源６０の間を流れる電流を検出するシャント抵抗７０を備えている。 Hereinafter, a first embodiment of the present invention will be described with reference to the drawings. As shown in FIG. 1, the motor control device 10 to which the present invention is applied has a multi-phase AC motor 20 that rotates by inputting a multi-phase AC voltage, and a multi-phase AC motor 20 that has a switching function. An inverter 30 that supplies a drive voltage signal, a drive pulse generator 40 that generates a drive pulse signal supplied to switch the inverter 30, and a plurality of drive pulses generated by the drive pulse generator 40 A drive circuit 50 applied to the inverter 30, a DC power supply 60 for supplying a predetermined DC voltage to the inverter 30, and a current flowing between the inverter 30 and the DC power supply 60 inserted in series between the inverter 30 and the DC power supply 60. A shunt resistor 70 for detection is provided.
　前記多相交流モータ２０は、さらに、鉄心の周りに巻線が巻かれた固定子２２と、所定の磁極を有する永久磁石からなり、固定子２２の内部を、所定の軸の周りに回転可能に保持された回転子２４からなる。そして、回転子２４の回転力が、多相交流モータ２０に接続された負荷に対してトルクを与える。 The polyphase AC motor 20 further includes a stator 22 wound around an iron core and a permanent magnet having a predetermined magnetic pole, and the inside of the stator 22 can be rotated around a predetermined axis. The rotor 24 is held by the rotor 24. The rotational force of the rotor 24 gives torque to the load connected to the multiphase AC motor 20.
　なお、多相交流の相数として、一般的には３相交流が用いられているため、以下、相数を３として説明を行う。すなわち、多相交流モータ２０は、Ｕ相，Ｖ相，Ｗ相からなる３相交流によって駆動されるものとする。 In addition, since the three-phase alternating current is generally used as the number of phases of the multi-phase alternating current, the number of phases will be described below as 3. That is, the multiphase AC motor 20 is driven by a three-phase AC composed of a U phase, a V phase, and a W phase.
　前記固定子２２は、さらに、巻線２２Ｕ，巻線２２Ｖ，巻線２２Ｗを有し、これらの巻線の１端は中性点２２Ｎで互いに接続されており、他端はそれぞれ、端子２５Ｕ，２５Ｖ，２５Ｗに接続されている。 The stator 22 further includes a winding 22U, a winding 22V, and a winding 22W. One end of these windings is connected to each other at a neutral point 22N, and the other end is connected to a terminal 25U, respectively. It is connected to 25V and 25W.
　前記インバータ３０は、さらに、ＩＧＢＴ（絶縁ゲートバイポーラトランジスタ）やＦＥＴ（電界効果トランジスタ）等のスイッチング素子からなり、Ｕ相用のスイッチング素子，Ｖ相用のスイッチング素子，Ｗ相用のスイッチング素子からなる。Ｕ相用のスイッチング素子は、高圧側（以下、Ｕ＋相と呼ぶ）のスイッチング素子３２Ｕと低圧側（以下、Ｕ－相と呼ぶ）のスイッチング素子３４Ｕからなり、Ｖ相用のスイッチング素子は、高圧側（以下、Ｖ＋相と呼ぶ）のスイッチング素子３２Ｖと低圧側（以下、Ｖ－相と呼ぶ）のスイッチング素子３４Ｖからなり、Ｗ相用のスイッチング素子は、高圧側（以下、Ｗ＋相と呼ぶ）のスイッチング素子３２Ｗと低圧側（以下、Ｗ－相と呼ぶ）のスイッチング素子３４Ｗからなる。 The inverter 30 further includes switching elements such as IGBT (Insulated Gate Bipolar Transistor) and FET (Field Effect Transistor), and includes a switching element for U phase, a switching element for V phase, and a switching element for W phase. . The switching element for the U phase is composed of a switching element 32U on the high voltage side (hereinafter referred to as U + phase) and a switching element 34U on the low voltage side (hereinafter referred to as U-phase). A switching element 32V on the side (hereinafter referred to as V + phase) and a switching element 34V on the low voltage side (hereinafter referred to as V-phase). The switching element for the W phase is on the high voltage side (hereinafter referred to as W + phase). Switching element 32W and a switching element 34W on the low voltage side (hereinafter referred to as W-phase).
　また、スイッチング素子３２Ｕ，３４Ｕ，３２Ｖ，３４Ｖ，３２Ｗ，３４Ｗには、それぞれ、直流電源６０の低電圧側から高電圧側に向かう方向を順方向とするダイオード３２ＤＵ，３４ＤＵ，３２ＤＶ，３４ＤＶ，３２ＤＷ，３４ＤＷが接続されている。これらのダイオードは、所謂還流ダイオードとして機能する。 In addition, the switching elements 32U, 34U, 32V, 34V, 32W, and 34W include diodes 32D U , 34D U , 32D V , and 34D having forward directions from the low voltage side to the high voltage side of the DC power supply 60, respectively. V , 32D W and 34D W are connected. These diodes function as so-called reflux diodes.
　そして、スイッチング素子３２Ｕとスイッチング素子３４Ｕの接続点は、端子２５Ｕと接続され、スイッチング素子３２Ｖとスイッチング素子３４Ｖの接続点は、端子２５Ｖと接続され、スイッチング素子３２Ｗとスイッチング素子３４Ｗの接続点は、端子２５Ｗと接続されている。 The connection point between the switching element 32U and the switching element 34U is connected to the terminal 25U, the connection point between the switching element 32V and the switching element 34V is connected to the terminal 25V, and the connection point between the switching element 32W and the switching element 34W is It is connected to the terminal 25W.
　したがって、Ｕ＋相のスイッチング素子３２Ｕをスイッチングして導通させることによって、巻線２２Ｕには端子２５Ｕから中性点２２Ｎに向かう電流が流れ、Ｕ－相のスイッチング素子３４Ｕをスイッチングして導通させることによって、巻線２２Ｕには中性点２２Ｎから端子２５Ｕに向かう電流が流れる。 Therefore, by switching the U + phase switching element 32U to conduct, a current flows from the terminal 25U to the neutral point 22N through the winding 22U, and by switching the U− phase switching element 34U to conduct. The current flowing from the neutral point 22N to the terminal 25U flows through the winding 22U.
　また、Ｖ＋相のスイッチング素子３２Ｖをスイッチングして導通させることによって、巻線２２Ｖには端子２５Ｖから中性点２２Ｎに向かう電流が流れ、Ｖ－相のスイッチング素子３４Ｖをスイッチングして導通させることによって、巻線２２Ｖには中性点２２Ｎから端子２５Ｖに向かう電流が流れる。 In addition, by switching the V + phase switching element 32V to conduct, a current flows from the terminal 25V to the neutral point 22N through the winding 22V, and by switching the V− phase switching element 34V to conduct. The current flowing from the neutral point 22N to the terminal 25V flows through the winding 22V.
　そして、Ｗ＋相のスイッチング素子３２Ｗをスイッチングして導通させることによって、巻線２２Ｗには端子２５Ｗから中性点２２Ｎに向かう電流が流れ、Ｗ－相のスイッチング素子３４Ｗをスイッチングして導通させることによって、巻線２２Ｗには中性点２２Ｎから端子２５Ｗに向かう電流が流れる。 Then, by switching the W + phase switching element 32W to be conducted, a current flows from the terminal 25W to the neutral point 22N through the winding 22W, and by switching the W− phase switching element 34W to be conducted. The current flowing from the neutral point 22N to the terminal 25W flows through the winding 22W.
　前記駆動パルス生成部４０はマイクロコンピュータで構成されて、シャント抵抗７０に流れるシャント電流ｉＲを検出する電流検出部４２と、多相交流モータ２０に印加する３相交流電圧波形を生成する目標電圧波形生成部４４と、多相交流モータ２０駆動時のトルク変動を小さくするために、目標電圧波形生成部４４で生成された電圧波形を、基本波成分が大きくなるように補正する目標電圧波形補正部４５と、目標電圧波形補正部４５で生成した電圧波形からＰＷＭ（パルス幅変調）されたパルス波形を生成するＰＷＭ波形生成部４６と、ＰＷＭ波形生成部４６で生成されたパルス波形を、電流検出部４２においてシャント電流を確実に検出できるパルス波形に補正し、また、電流検出部４２で検出されたシャント電流ｉＲに応じて、ＰＷＭ波形をインバータ３０に供給するタイミングを調整するＰＷＭ波形補正部４８からなる。 The driving pulse generation unit 40 based on a microcomputer, a current detecting unit 42 for detecting the shunt current i R flowing to the shunt resistor 70, the target voltage for generating a 3-phase AC voltage waveform applied to the multiphase AC motor 20 Target voltage waveform correction that corrects the voltage waveform generated by the target voltage waveform generation unit 44 to increase the fundamental wave component in order to reduce torque fluctuations when the waveform generation unit 44 and the multiphase AC motor 20 are driven. 45, a PWM waveform generation unit 46 that generates a pulse waveform PWM (pulse width modulated) from the voltage waveform generated by the target voltage waveform correction unit 45, and a pulse waveform generated by the PWM waveform generation unit 46 as a current the detection unit 42 corrects the pulse waveform can be reliably detected shunt current, also in accordance with the shunt current i R detected by the current detector 42, The WM waveform consisting PWM waveform correction unit 48 to adjust the timing for supplying the inverter 30.
　［モータ回転動作の説明］
次に、本実施例で行われる、多相交流モータ２０が回転する原理について、図２Ａと図２Ｂを用いて説明する。 [Explanation of motor rotation]
Next, the principle of rotation of the multiphase AC motor 20 performed in this embodiment will be described with reference to FIGS. 2A and 2B.
　図２Ａに示す、位相が互いに１２０°ずれた３相交流電圧波形ＥＵ，ＥＶ，ＥＷを、多相交流モータ２０に印加した場合を考える。 Consider a case where the three-phase AC voltage waveforms E U , E V , and E W shown in FIG.
　図２Ｂは、３相交流電圧波形の位相毎に、多相交流モータ２０の巻線に流れる電流の方向と回転子２４の位置関係を示している。図２Ｂと図１を対照して説明すると、例えば、３相交流電圧波形の位相がθ＝０°のときは、Ｕ－相を構成するスイッチング素子３４Ｕが導通して、多相交流モータ２０を駆動する電流が、中性点２２Ｎから巻線２２Ｕを通って、端子２５Ｕを経てスイッチング素子３４Ｕに流れ込むが、これを図２Ｂでは、電流が中性点２２Ｎ側の巻線端子２２Ｕｂから流れ込んで、端子２５Ｕ側の巻線端子２２Ｕａから流れ出す方向に流れるものとして表現している。Ｖ相、Ｗ相についても同様であり、巻線端子２２Ｖａ，２２Ｗｂが中性点２２Ｎ側の巻線端子を表し、巻線端子２２Ｖｂが端子２５Ｖ側、巻線端子２２Ｗａが端子２５Ｗ側の巻線端子を表している。 FIG. 2B shows the positional relationship between the direction of the current flowing through the windings of the multiphase AC motor 20 and the rotor 24 for each phase of the three-phase AC voltage waveform. Referring to FIG. 2B and FIG. 1, for example, when the phase of the three-phase AC voltage waveform is θ = 0 °, the switching element 34U constituting the U-phase is turned on, and the multi-phase AC motor 20 is The driving current flows from the neutral point 22N through the winding 22U and flows into the switching element 34U through the terminal 25U. In FIG. 2B, this current flows from the winding terminal 22Ub on the neutral point 22N side, It is expressed as flowing in the direction of flowing out from the winding terminal 22Ua on the terminal 25U side. The same applies to the V phase and the W phase, wherein the winding terminals 22Va and 22Wb represent the winding terminals on the neutral point 22N side, the winding terminal 22Vb is the terminal 25V side, and the winding terminal 22Wa is the winding on the terminal 25W side. Represents a terminal.
　そして、図２Ｂに示すように、３相交流電圧波形の位相の変化に応じて、各巻線を流れる電流の大きさと方向が変化するため、電磁誘導によって固定子２２に回転磁界が発生して、回転子２４の磁極が、固定子２２に発生した回転磁界に吸引、反発されて、その方向を変化させることにより、多相交流モータ２０が回転する。 Then, as shown in FIG. 2B, the magnitude and direction of the current flowing through each winding changes according to the change in the phase of the three-phase AC voltage waveform, so that a rotating magnetic field is generated in the stator 22 by electromagnetic induction, The magnetic poles of the rotor 24 are attracted and repelled by the rotating magnetic field generated in the stator 22, and the multi-phase AC motor 20 is rotated by changing its direction.
　［電圧波形の生成および補正方法の説明］
次に、多相交流モータ２０に印加する目標電圧波形の生成方法およびその補正方法について、図３を用いて説明する。 [Description of voltage waveform generation and correction]
Next, a method for generating a target voltage waveform applied to the polyphase AC motor 20 and a method for correcting the target voltage waveform will be described with reference to FIG.
　図３の上段のグラフは、正弦波からなる３相交流電圧波形ＥＵ，ＥＶ，ＥＷを示している。ここで、３相交流電圧波形ＥＵ，ＥＶ，ＥＷは（式１）の関係を満たしている。
ＥＵ＋ＥＶ＋ＥＷ＝０　　　　　　　　　　　（式１） The upper graph in FIG. 3, three-phase AC voltage waveform E U consisting of sine wave shows E V, the E W. Here, 3-phase AC voltage waveform E U, E V, E W satisfies the relationship of Equation (1).
E U + E V + E W = 0 (Formula 1)
　ここで、３相交流電圧波形ＥＵ，ＥＶ，ＥＷの電圧の大小関係に基づいて、正弦波の１周期を６等分する。こうして６等分された区間をそれぞれ、ｍｏｄｅ０，ｍｏｄｅ１，ｍｏｄｅ２，ｍｏｄｅ３，ｍｏｄｅ４，ｍｏｄｅ５とする。 Here, one cycle of the sine wave is divided into six equal parts based on the magnitude relationship of the voltages of the three-phase AC voltage waveforms E U , E V , and E W. The sections thus divided into six are referred to as mode0, mode1, mode2, mode3, mode4, and mode5, respectively.
　すなわち、図３の上段のグラフに示すように、ｍｏｄｅ０の区間（ｔＣ≦ｔ＜ｔＤ）は、ＥＵ＞ＥＶ＞ＥＷの関係を有し、ｍｏｄｅ１の区間（ｔＤ≦ｔ＜ｔＥ）は、ＥＶ＞ＥＵ＞ＥＷの関係を有し、ｍｏｄｅ２の区間（ｔＥ≦ｔ＜ｔＦ）は、ＥＶ＞ＥＷ＞ＥＵの関係を有し、ｍｏｄｅ３の区間（０≦ｔ＜ｔＡ）は、ＥＷ＞ＥＶ＞ＥＵの関係を有し、ｍｏｄｅ４の区間（ｔＡ≦ｔ＜ｔＢ）は、ＥＷ＞ＥＵ＞ＥＶの関係を有し、ｍｏｄｅ５の区間（ｔＢ≦ｔ＜ｔＣ）は、ＥＵ＞ＥＷ＞ＥＶの関係を有しているものとする。すなわち、各ｍｏｄｅの変化点は、３相交流電圧波形の相順が変化する点とする。 That is, as shown in the upper graph of FIG. 3, the section of mode 0 (t C ≦ t <t D ) has a relationship of E U > E V > E W and the section of mode 1 (t D ≦ t < t E ) has a relationship of E V > E U > E W , a mode2 interval (t E ≦ t <t F ) has a relationship of E V > E W > E U , and a mode 3 interval (0 ≦ t <t A ) has a relationship of E W > E V > E U , and a section of mode 4 (t A ≦ t <t B ) has a relationship of E W > E U > E V. and, mode 5 of the section (t B ≦ t <t C ) is, E U> assumed to have a relation of E W> E V. That is, the change point of each mode is a point where the phase order of the three-phase AC voltage waveform changes.
　ここで、ｍｏｄｅ０，ｍｏｄｅ１，ｍｏｄｅ２，ｍｏｄｅ３，ｍｏｄｅ４，ｍｏｄｅ５の各区間において、３相交流電圧波形の３つの電圧値の中間値の半分の値を補正電圧Ｅｃと定義する。 Here, in each section of mode0, mode1, mode2, mode3, mode4, and mode5, a value that is half the intermediate value of the three voltage values of the three-phase AC voltage waveform is defined as a correction voltage Ec.
　すなわち、ｍｏｄｅ０とｍｏｄｅ３では、Ｅｃ＝ＥＶ／２、ｍｏｄｅ１とｍｏｄｅ４では、Ｅｃ＝ＥＵ／２、ｍｏｄｅ２とｍｏｄｅ５では、Ｅｃ＝ＥＷ／２となる。こうして算出された補正電圧Ｅｃを図示すると、図３の中段のグラフのようになる。 That is, Ec = E V / 2 in mode 0 and mode 3, Ec = E U / 2 in mode 1 and mode 4, and Ec = E W / 2 in mode 2 and mode 5. The correction voltage Ec thus calculated is shown in the middle graph of FIG.
　そして、補正電圧Ｅｃを元の３相交流電圧波形ＥＵ，ＥＶ，ＥＷに加算することによって、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗを生成すると、下記（式２）～（式４）のようになる。
Ｅ’Ｕ＝ＥＵ＋Ｅｃ＝（ＥＵ－ＥＷ）／２　　　　　（式２）
Ｅ’Ｖ＝ＥＶ＋Ｅｃ＝３ＥＶ／２　　　　　　　　　（式３）
Ｅ’Ｗ＝ＥＷ＋Ｅｃ＝（ＥＷ－ＥＵ）／２　　　　　（式４） Then, by adding the correction voltage Ec to the original three-phase AC voltage waveforms E U , E V , E W to generate the correction voltage waveforms E ′ U , E ′ V , E ′ W , the following (formula 2) (Formula 4)
E ′ U = E U + Ec = (E U −E W ) / 2 (Formula 2)
E ′ V = E V + Ec = 3 E V / 2 (Formula 3)
E ′ W = E W + Ec = (E W −E U ) / 2 (Formula 4)
　ここで、異なる相の間の電圧（線間電圧）を求めると、（式１）～（式３）より、
Ｅ’Ｕ－Ｅ’Ｖ＝ＥＵ－ＥＶ　　　　　　　　　　　（式５）
となる。同様にして、（式６），（式７）が成り立つ。
Ｅ’Ｕ－Ｅ’Ｗ＝ＥＵ－ＥＷ　　　　　　　　　　　（式６）
Ｅ’Ｖ－Ｅ’Ｗ＝ＥＶ－ＥＷ　　　　　　　　　　　（式７） Here, when the voltage between the different phases (line voltage) is obtained, from (Equation 1) to (Equation 3),
E ′ U −E ′ V = E U −E V (Formula 5)
It becomes. Similarly, (Expression 6) and (Expression 7) hold.
E ′ U −E ′ W = E U −E W (Formula 6)
E ′ V −E ′ W = E V −E W (Formula 7)
　すなわち、電圧波形の補正の前後で、各相の線間電圧は等しいことがわかる。 That is, it can be seen that the line voltage of each phase is equal before and after the correction of the voltage waveform.
　ここで、補正前の３相交流電圧波形ＥＵ，ＥＶ，ＥＷが、波高値Ｅ０を有し、それぞれ（式８）～（式１０）で記述されるものとする。
ＥＵ＝Ｅ０ｓｉｎθ　　　　　　　　　　　　　　　　（式８）
ＥＶ＝Ｅ０ｓｉｎ（θ＋２π／３）　　　　　　　　　（式９）
ＥＷ＝Ｅ０ｓｉｎ（θ＋４π／３）　　　　　　　　　（式１０） Here, it is assumed that the three-phase AC voltage waveforms E U , E V , and E W before correction have peak values E 0 and are described by (Equation 8) to (Equation 10), respectively.
E U = E 0 sin θ (Equation 8)
E V = E 0 sin (θ + 2π / 3) (Formula 9)
E W = E 0 sin (θ + 4π / 3) (Formula 10)
　すると、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗは、それぞれ、例えばｍｏｄｅ３の範囲内では、（式１１）～（式１３）のようになる。
Ｅ’Ｕ＝Ｅ０（３ｓｉｎθ＋３１／２ｃｏｓθ）／４　　（式１１）
Ｅ’Ｖ＝３Ｅ０｛ｓｉｎ（θ＋２π／３）｝／２　　　（式１２）
Ｅ’Ｗ＝Ｅ０（３１／２ｃｏｓθ－３ｓｉｎθ）／４　　（式１３） Then, the correction voltage waveforms E ′ U , E ′ V , and E ′ W are expressed as (Equation 11) to (Equation 13), respectively, within the range of mode 3, for example.
E ′ U = E 0 (3 sin θ + 3 1/2 cos θ) / 4 (formula 11)
E ′ V = 3E 0 {sin (θ + 2π / 3)} / 2 (Formula 12)
E ′ W = E 0 (3 1/2 cos θ-3 sin θ) / 4 (Formula 13)
　各ｍｏｄｅの範囲について（式１１）～（式１３）に対応する補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗを算出してグラフ化すると、図３の下段のグラフのようになる。このグラフからわかるように、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの振幅は、３相交流電圧波形ＥＵ，ＥＶ，ＥＷの振幅よりも小さくなる。 When the correction voltage waveforms E ′ U , E ′ V , and E ′ W corresponding to (Equation 11) to (Equation 13) are calculated and graphed for each mode range, a graph in the lower part of FIG. 3 is obtained. As can be seen from this graph, the amplitudes of the correction voltage waveforms E ′ U , E ′ V , E ′ W are smaller than the amplitudes of the three-phase AC voltage waveforms E U , E V , E W.
　すなわち、上記で説明したように３相交流電圧波形ＥＵ，ＥＶ，ＥＷを補正して、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗとすることによって、異なる相間の線間電圧を変えずに、印加電圧の基本波成分の最大値をより大きくすることができる。すなわち、小さな電圧波形で従来と同等の線間電圧を得ることができるため、電圧の制御範囲を拡大することができる。 That is, as described above, the three-phase AC voltage waveforms E U , E V , E W are corrected to be corrected voltage waveforms E ′ U , E ′ V , E ′ W , so The maximum value of the fundamental wave component of the applied voltage can be increased without changing the voltage. That is, since a line voltage equivalent to the conventional one can be obtained with a small voltage waveform, the voltage control range can be expanded.
　［ＰＷＭ波形の生成方法の説明］
次に、多相交流モータ２０に印加するＰＷＭ波形の生成方法について、図４、および図５Ａ～図５Ｅを用いて説明する。 [Description of PWM waveform generation method]
Next, a method for generating a PWM waveform to be applied to the multiphase AC motor 20 will be described with reference to FIGS. 4 and 5A to 5E.
　多相交流モータ２０は、実際には、先に説明した電圧波形によって直接駆動するのではない。すなわち、実際は、電圧波形の電圧値に応じたパルス幅を有するパルス波形を生成して、このパルス波によってインバータの導通、非導通を制御することによって、多相交流モータ２０を駆動する。ここで、電圧波形の電圧値に応じたパルス幅を有するパルス波形を生成することをＰＷＭ（Ｐｕｌｓｅ　Ｗｉｄｔｈ　Ｍｏｄｕｌａｔｉｏｎ：パルス幅変調）という。そして、ＰＷＭによって生成されたパルスをＰＷＭ波形と呼ぶ。 The multiphase AC motor 20 is not actually driven directly by the voltage waveform described above. That is, actually, the multiphase AC motor 20 is driven by generating a pulse waveform having a pulse width corresponding to the voltage value of the voltage waveform and controlling the conduction and non-conduction of the inverter by this pulse wave. Here, generating a pulse waveform having a pulse width corresponding to the voltage value of the voltage waveform is referred to as PWM (Pulse Width Modulation). A pulse generated by PWM is called a PWM waveform.
　ＰＷＭ波形を生成して、生成されたＰＷＭ波形によってインバータの導通、非導通を制御して多相交流モータ２０を駆動することは、モータ制御において一般的に行われていることであるため、説明は簡単に留める。 Since the PWM waveform is generated and the multi-phase AC motor 20 is driven by controlling the conduction and non-conduction of the inverter by the generated PWM waveform is generally performed in the motor control. Fasten easily.
　図４の最上段のグラフは、パルス幅変調を行うために用いる搬送波Ｃ（ｔ）の１周期Ｔｃに亘る波形と、時刻ｔａの近傍における補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの電圧値を示している。ここで、図４の横軸は時間軸を示しており、図３に示した時刻ｔａの近傍のごく短い時間範囲が図４に示されている。時刻ｔａはｍｏｄｅ０の範囲にあるため、補正電圧波形はＥ’Ｕ＞Ｅ’Ｖ＞Ｅ’Ｗの関係を有している。 The uppermost graph in FIG. 4 shows the waveform over one period Tc of the carrier wave C (t) used for pulse width modulation and the corrected voltage waveforms E ′ U , E ′ V , E ′ W in the vicinity of time ta. The voltage value is shown. Here, the horizontal axis of FIG. 4 indicates the time axis, and a very short time range in the vicinity of the time ta shown in FIG. 3 is shown in FIG. Since the time ta is in the range of mode 0, the correction voltage waveform has a relationship of E ′ U > E ′ V > E ′ W.
　次に、図４を用いて、スイッチング素子３２Ｕに印加するＰＷＭ波形Ｕ＋と、スイッチング素子３４Ｕに印加するＰＷＭ波形Ｕ－の生成方法について説明する。 Next, a method of generating the PWM waveform U + applied to the switching element 32U and the PWM waveform U− applied to the switching element 34U will be described with reference to FIG.
　搬送波Ｃ（ｔ）と電圧波形ＥＵの大小関係を比較すると、図４の最下段のグラフに記載した時刻ｔ１から時刻ｔ６の範囲に亘って、ＥＵ＞Ｃ（ｔ）となっている。このとき、スイッチング素子３２Ｕを制御するＰＷＭ波形Ｕ＋として、時刻ｔ１から時刻ｔ６の範囲に亘ってＨｉレベルを有し、それ以外の時間範囲ではＬｏレベルを有するＰＷＭ波形を生成する。 When the carrier C (t) and compares the magnitude relation between the voltage waveform E U, over a range from the time t1 as described at the bottom of the graph of the time t6 in FIG. 4, it has a E U> C (t). At this time, as a PWM waveform U + for controlling the switching element 32U, a PWM waveform having a Hi level over a range from time t1 to time t6 and having a Lo level in other time ranges is generated.
　さらに、スイッチング素子３４Ｕを制御するＰＷＭ波形Ｕ－として、ＰＷＭ波形Ｕ＋の位相を反転させたＰＷＭ波形Ｕ－を生成する。なお、単にＰＷＭ波形Ｕ＋の位相を反転させてＰＷＭ波形Ｕ－を生成すると、スイッチング素子３２Ｕが導通するタイミングとスイッチング素子３４Ｕが非導通するタイミングが一致してしまう。すると、スイッチング素子３２Ｕとスイッチング素子３４Ｕが同時に導通してしまう可能性がある。このような状態になると、スイッチング素子３２Ｕとスイッチング素子３４Ｕの間に短絡電流が流れるため、過剰な発熱や、最悪の場合は回路の破損を招く。 Further, a PWM waveform U− obtained by inverting the phase of the PWM waveform U + is generated as a PWM waveform U− for controlling the switching element 34U. If the PWM waveform U− is generated simply by inverting the phase of the PWM waveform U +, the timing at which the switching element 32U is turned on coincides with the timing at which the switching element 34U is turned off. Then, there is a possibility that the switching element 32U and the switching element 34U become conductive at the same time. In such a state, a short-circuit current flows between the switching element 32U and the switching element 34U, which causes excessive heat generation and, in the worst case, damage to the circuit.
　そのため、通常は、高圧側のパルス（例えばＰＷＭ波形Ｕ＋）の立ち上がりと低圧側のパルス（例えばＰＷＭ波形Ｕ－）の立ち下がりの時間に時間差を設けて、高圧側のスイッチング素子（例えば３２Ｕ）と低圧側のスイッチング素子（例えば３４Ｕ）がともに非導通となる時間を確保している。この所定の時間差のことをデッドタイムＤＴと呼ぶ。デッドタイムＤＴとして、例えば数μｓｅｃ程度の時間が設定される。本実施例では、図４のＰＷＭ波形Ｕ＋，Ｕ－に示すように、低圧側のパルスであるＰＷＭ波形Ｕ－にデッドタイムＤＴに相当する時間を付与している。 Therefore, normally, a time difference is provided between the rising time of the high-voltage side pulse (for example, PWM waveform U +) and the falling time of the low-voltage side pulse (for example, PWM waveform U−) to Both low voltage side switching elements (for example, 34U) ensure the time when it becomes non-conductive. This predetermined time difference is called dead time DT. For example, a time of about several μsec is set as the dead time DT. In this embodiment, as shown by PWM waveforms U + and U− in FIG. 4, a time corresponding to the dead time DT is given to the PWM waveform U− that is a low-pressure side pulse.
　そして、同様にして、図４に示すように、スイッチング素子３２Ｖを制御するＰＷＭ波形Ｖ＋，スイッチング素子３４Ｖを制御するＰＷＭ波形Ｖ－、および、スイッチング素子３２Ｗを制御するＰＷＭ波形Ｗ＋，スイッチング素子３４Ｗを制御するＰＷＭ波形Ｗ－が生成される。 Similarly, as shown in FIG. 4, a PWM waveform V + for controlling the switching element 32V, a PWM waveform V− for controlling the switching element 34V, a PWM waveform W + for controlling the switching element 32W, and a switching element 34W A PWM waveform W− to be controlled is generated.
　こうして生成されたＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－が、それぞれ対応するスイッチング素子に印加されて、各スイッチング素子が、順次、導通と非導通を繰り返す。その結果、シャント抵抗７０には、図４の最下段のグラフに示すシャント電流ｉＲが流れる。 The PWM waveforms U +, U−, V +, V−, W +, W− generated in this way are applied to the corresponding switching elements, and each switching element repeats conduction and non-conduction sequentially. As a result, the shunt resistor 70, flows shunt current i R shown at the bottom of the graph of FIG.
　図４の最下段のグラフにおいて、シャント電流ｉＲの量が変化する時刻を、左から順にｔ１，ｔ２，ｔ３，ｔ４とし、搬送波Ｃ（ｔ）の１周期の先頭の時刻をｔ０とすると、各時刻の間隔の比率は、それぞれ、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの線間電圧の比率に対応している。 In the lowermost graph in FIG. 4, the time to change the amount of shunt current i R, and t1, t2, t3, t4 from the left, when the time of the head of one cycle of the carrier wave C (t) and t0, The ratio of the intervals at each time corresponds to the ratio of the line voltages of the correction voltage waveforms E ′ U , E ′ V , E ′ W , respectively.
　すなわち、Ｅ’ＵとＥ’Ｖの線間電圧（図３に示す線間電圧τａに対応）と、Ｅ’ＶとＥ’Ｗの線間電圧（図３に示す線間電圧τｂに対応）と、Ｅ’Ｕと補正電圧波形の最大値との線間電圧（図３に示す線間電圧τｃに対応）と、Ｅ’Ｗと補正電圧波形の最小値との線間電圧（図３に示す線間電圧τｄに対応）の大きさの比率は、それぞれ、時刻ｔ１と時刻ｔ２の間隔、時刻ｔ２と時刻ｔ３の間隔、時刻ｔ０と時刻ｔ１の間隔、時刻ｔ３と時刻ｔ４の間隔に、それぞれ対応している。 That is, the line voltage of E ′ U and E ′ V (corresponding to the line voltage τ a shown in FIG. 3) and the line voltage of E ′ V and E ′ W (line voltage τ b shown in FIG. The line voltage between E ′ U and the maximum value of the correction voltage waveform (corresponding to the line voltage τ c shown in FIG. 3), and the line voltage between E ′ W and the minimum value of the correction voltage waveform ( The ratio of the magnitude of the line voltage τ d shown in FIG. 3 is as follows: time t1 and time t2, time t2 and time t3, time t0 and time t1, time t3 and time t4, respectively. Correspond to each of the intervals.
　なお、図５Ａ～図５Ｅに、ｍｏｄｅ３以外の時間範囲におけるＰＷＭ波形の生成例、およびシャント抵抗７０に流れるシャント電流ｉＲの概略を示す。図５Ａ～図５Ｅに示すように、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの電圧値が時間とともに変化するため、それに応じてＰＷＭ波形のパルス幅が変化し、シャント電流ｉＲの波形が変化する。 5A to 5E show an example of generating a PWM waveform in a time range other than mode 3 and an outline of the shunt current i R flowing through the shunt resistor 70. FIG. As shown in FIGS. 5A to 5E, since the voltage values of the correction voltage waveforms E ′ U , E ′ V , E ′ W change with time, the pulse width of the PWM waveform changes accordingly, and the shunt current i R The waveform of changes.
　［シャント電流ｉＲに基づくモータ制御の説明］
前記したように、シャント電流ｉＲの波形は、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの状態に応じて一意に決定する。すなわち、シャント電流ｉＲの大きさを測定することができれば、そのときの、多相交流モータ２０の回転子２４の位置を推定することができる。そして、回転子２４が所定の位置にないときには、スイッチング素子を制御するための補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗの位相を制御して、これに応じて、ＰＷＭ波形を生成し直すことによって、所定のモータ制御を実現することができる。 Description of motor control based on the shunt current i R]
As described above, the waveform of the shunt current i R, is uniquely determined in accordance with the state of the correction voltage waveform E 'U, E' V, E 'W. That is, if it is possible to measure the magnitude of the shunt current i R, can be estimated at this time, a position of the rotator 24 of the multi-phase AC motor 20. When the rotor 24 is not in a predetermined position, the phases of the correction voltage waveforms E ′ U , E ′ V , E ′ W for controlling the switching elements are controlled, and a PWM waveform is generated accordingly. By re-doing, predetermined motor control can be realized.
　［ＰＷＭ波形の補正方法の説明］
次に、多相交流モータ２０に印加するＰＷＭ波形の補正の必要性、および補正方法について、図６から図９を用いて説明する。 [Description of PWM waveform correction method]
Next, the necessity of correcting the PWM waveform applied to the polyphase AC motor 20 and the correction method will be described with reference to FIGS.
　図３に示したｍｏｄｅ０からｍｏｄｅ５のうち、隣り合うｍｏｄｅの切り替わり位置においては、２つの異なる相の電圧値が接近する。例えば、ｍｏｄｅ０からｍｏｄｅ１に切り替わる位置においては、補正電圧Ｅ’Ｕの電圧値と補正電圧Ｅ’Ｖの電圧値が一致する。また、ｍｏｄｅ１からｍｏｄｅ２に切り替わる位置においては、補正電圧Ｅ’Ｕの電圧値と補正電圧Ｅ’Ｗの電圧値が一致する。 Among the modes 0 to mode 5 shown in FIG. 3, the voltage values of two different phases approach each other at the switching position of adjacent modes. For example, at the position where the mode 0 is switched to the mode 1, the voltage value of the correction voltage E ′ U matches the voltage value of the correction voltage E ′ V. Further, at the position where the mode 1 is switched to the mode 2, the voltage value of the correction voltage E ′ U matches the voltage value of the correction voltage E ′ W.
　図６は、ｍｏｄｅ０からｍｏｄｅ１に切り替わる位置において、補正電圧波形Ｅ’Ｕの電圧値と補正電圧波形Ｅ’Ｖの電圧値が一致したときのＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－とシャント電流ｉＲの状態を示す。 6, at a position where switching to mode1 from mode0, PWM waveform when the voltage value of the correction voltage waveform E 'voltage values of the U and the correction voltage waveform E' V matches U +, U-, V +, V-, W +, The state of W− and the shunt current i R is shown.
　図６に示すように、ｍｏｄｅ０からｍｏｄｅ１に切り替わる時間においては、ＰＷＭ波形Ｕ＋とＰＷＭ波形Ｖ＋が一致する。したがって、図４の最下段のグラフに示す時刻ｔ１とｔ２の間隔、および時刻ｔ５とｔ６の間隔が徐々に短くなる。 As shown in FIG. 6, the PWM waveform U + and the PWM waveform V + coincide with each other during the time when the mode 0 is switched to the mode 1. Therefore, the interval between times t1 and t2 and the interval between times t5 and t6 shown in the lowermost graph of FIG. 4 are gradually shortened.
　ここで、シャント電流ｉＲを検出するためには、検出した電流をサンプリングしてＡＤ変換し、値を読み取る必要があるため、最小時間Ｔｍｉｎ以上の時間が必要となる。 Here, in order to detect the shunt current i R is to AD conversion by sampling the detected current, it is necessary to read the value, it is necessary minimum time T min or longer.
　したがって、ｍｏｄｅ０からｍｏｄｅ１に移行する前後、すなわち、３相交流の相順がＵＶＷからＶＵＷに変化する点の前後では、時刻ｔ１と時刻ｔ２の間隔、および時刻ｔ５と時刻ｔ６の間隔が、ともに最小時間Ｔｍｉｎを下回ってしまうため、その間の時間に亘ってシャント電流ｉＲを検出することができなくなってしまう。 Therefore, before and after the transition from mode 0 to mode 1, that is, before and after the point where the phase sequence of the three-phase alternating current changes from UVW to VUW, the interval between time t1 and time t2, and the interval between time t5 and time t6 are both minimum. since falls below the time T min, it becomes impossible to detect the shunt current i R over the intervening time.
　シャント電流ｉＲを検出することができなくなる状況は、ｍｏｄｅ０からｍｏｄｅ１に切り替わる時間以外にも、各ｍｏｄｅの切り替わり点、すなわち、３相交流の相順が変化する点の前後において発生する。そして、このような、相順が変化する点の前後においては、シャント電流ｉＲを検出できなくなるため、回転子２４の位置の推定ができず、モータ制御が不安定な状態となる。 The situation where the shunt current i R cannot be detected occurs before and after the switching point of each mode, that is, the point where the phase sequence of the three-phase alternating current changes, in addition to the time for switching from mode 0 to mode 1. Then, such, in the front and rear of the point that the phase sequence is changed, it becomes impossible to detect the shunt current i R, can not estimate the position of the rotor 24, motor control becomes unstable.
　なお、シャント電流ｉＲの流れる方向によっても、シャント電流ｉＲが検出しにくくなる状態が生じる。以下、図７Ａ，図７Ｂを用いて、これを説明する。 Even the direction of the flow of shunt current i R, occurs state hardly detected shunt current i R. Hereinafter, this will be described with reference to FIGS. 7A and 7B.
　図７Ａは、Ｕ相の巻線２２Ｕに流れる電流ｉの大きさの変化について説明する図であり、スイッチング素子３２Ｕから巻線２２Ｕに向けて電流ｉが流れている状態（このときの電流ｉの向きをｉ＞０とする）を示している。このとき、スイッチング素子３４Ｕが非導通状態になって、スイッチング素子３２Ｕが導通状態になっていると、電流ｉはスイッチング素子３２Ｕを通り、電流経路ｒ１を流れる。そして、スイッチング素子３２Ｕが非導通状態になると、スイッチング素子３４Ｕが非導通状態のままであっても、電流ｉはダイオード３４ＤＵを通り、電流経路ｒ２を流れる。このとき、シャント抵抗７０に流れるシャント電流ｉＲは小さくなる。そして、スイッチング素子３４Ｕが導通状態になっても、電流経路ｒ２は変化せず、シャント電流ｉＲも変化しない。このときのシャント電流ｉＲの変化の様子を、図７Ａ下部のタイムチャートに示す。 FIG. 7A is a diagram for explaining a change in the magnitude of the current i flowing through the U-phase winding 22U. The state in which the current i flows from the switching element 32U toward the winding 22U (the current i at this time) The orientation is i> 0). At this time, when the switching element 34U is in a non-conductive state and the switching element 32U is in a conductive state, the current i flows through the current path r1 through the switching element 32U. When the switching element 32U is nonconducting, be left 34U switching element is in a non-conducting state, the current i passes through the diode 34D U, flows through the current path r2. In this case, the smaller the shunt current i R flowing to the shunt resistor 70. Even if 34U switching element in a conductive state, a current path r2 is not changed, no change shunt current i R. The manner of change of the shunt current i R of this case, shown in the time chart in the lower Figure 7A.
　一方、図７Ｂは、スイッチング素子３２Ｕに向かって巻線２２Ｕから電流ｉが流れ込んでいる状態（このときの電流ｉの向きをｉ＜０とする）を示している。このとき、スイッチング素子３４Ｕが非導通状態になって、スイッチング素子３２Ｕが導通状態になっていると、電流ｉはダイオード３２ＤＵを通り、電流経路ｒ３を流れる。そして、スイッチング素子３２Ｕが非導通状態になると、スイッチング素子３４Ｕが非導通状態のままであっても、電流ｉはダイオード３２ＤＵを通り、電流経路ｒ３を流れる。このとき、シャント抵抗７０に流れるシャント電流ｉＲは変化しない。そして、スイッチング素子３４Ｕが導通状態になると、電流ｉがスイッチング素子３４Ｕを通り、電流経路ｒ４を流れる。このとき、シャント電流ｉＲは小さくなる。このときのシャント電流ｉＲの変化の様子を、図７Ｂ下部のタイムチャートに示す。 On the other hand, FIG. 7B shows a state in which the current i flows from the winding 22U toward the switching element 32U (the direction of the current i at this time is i <0). At this time, 34U switching element becomes non-conducting state, the switching element 32U is made conductive, current i passes through the diode 32D U, flows through the current path r3. When the switching element 32U is nonconducting, be left 34U switching element is in a non-conducting state, the current i passes through the diode 32D U, flows through the current path r3. In this case, shunt current i R flowing to the shunt resistor 70 is not changed. When switching element 34U becomes conductive, current i passes through switching element 34U and flows through current path r4. In this case, shunt current i R is small. The manner of change of the shunt current i R of this case, shown in the time chart in the lower Figure 7B.
　このように、巻線２２Ｕに流れる電流ｉの方向によって、シャント電流ｉＲの大きさが小さくなるタイミングが異なる。図７Ａ，図７Ｂに示すように、電流ｉの方向がｉ＜０のときは、シャント電流ｉＲの大きさが小さくなるタイミングが、ｉ＞０のときと比べて遅くなる。なお、これは、Ｕ相の巻線２２Ｕのみならず、Ｖ相の巻線２２Ｖ、Ｗ相の巻線２２Ｗについても同様である。 Thus, by the direction of the current i flowing through the windings 22U, timing the magnitude of the shunt current i R is small is different. As shown in FIGS. 7A, 7B, <If 0, the timing at which the magnitude of the shunt current i R becomes smaller, i> direction of the current i is i slower than when zero. This applies not only to the U-phase winding 22U but also to the V-phase winding 22V and the W-phase winding 22W.
　そして、シャント電流ｉＲの大きさが小さい区間が短くなることによって、その時間間隔が最小時間Ｔｍｉｎを下回ってしまうと、前述した通り、シャント電流ｉＲの検出ができなくなる。 Then, by the section size of the shunt current i R is small is shortened, if the time interval falls below a minimum time T min, as described above, can not be detected in the shunt current i R.
　なお、巻線２２Ｕ，２２Ｖ，２２Ｗを流れる電流ｉの方向は、各相に印加される電圧とその相を流れる電流の位相差によって決定する。電圧の位相に対して電流の位相が所定値以上遅れたときに、電流ｉの方向がｉ＜０となる。 The direction of the current i flowing through the windings 22U, 22V, and 22W is determined by the phase difference between the voltage applied to each phase and the current flowing through that phase. When the current phase is delayed by a predetermined value or more with respect to the voltage phase, the direction of the current i becomes i <0.
　以上をまとめると、巻線に印加される電圧に対して巻線に流れる電流の位相差が所定値以上遅れたときには、相順が変化する時刻において、スイッチング素子を導通させる時間がシャント電流ｉＲを検出するのに要する最小時間Ｔｍｉｎを下回ってしまうため、シャント電流ｉＲを検出することができなくなる。 In summary, when the phase difference of the current flowing through the winding is delayed by a predetermined value or more with respect to the voltage applied to the winding, the time for which the switching element is turned on at the time when the phase sequence changes is shunt current i R. since falls below a minimum time T min needed to detect, it becomes impossible to detect the shunt current i R.
　図８は、多相交流モータ２０の運転範囲を示すグラフである。グラフの横軸は多相交流モータ２０の回転数Ｐを示し、縦軸は多相交流モータ２０が発生するトルクＴを示している。そして、多相交流モータ２０は、図８に示す運転限界線Ｃに囲まれた運転領域Ｒ１の内部で正常に動作する。 FIG. 8 is a graph showing the operation range of the multiphase AC motor 20. The horizontal axis of the graph represents the rotational speed P of the multiphase AC motor 20, and the vertical axis represents the torque T generated by the multiphase AC motor 20. The multiphase AC motor 20 operates normally within the operation region R1 surrounded by the operation limit line C shown in FIG.
　そして、シャント電流ｉＲが検出できないときは、モータの回転数を高くして大きなトルクを発生させようとすると、多相交流モータ２０の動作領域が、図８に示す不安定領域Ｒ２に入ってしまい、モータ制御が不安定となる。このような不安定領域Ｒ２の発生を防ぐためには、例えば、容量がより大きな多相交流モータを用いることによって、前記不安定領域Ｒ２を運転領域Ｒ１として使用することができる。しかし、必要以上に容量が大きなモータを使用すると、コストが増大し、モータサイズが大きくなるため、設置性が悪化してしまう。 When the shunt current i R cannot be detected, the operating region of the multiphase AC motor 20 enters the unstable region R2 shown in FIG. As a result, motor control becomes unstable. In order to prevent the occurrence of such an unstable region R2, the unstable region R2 can be used as the operation region R1, for example, by using a multiphase AC motor having a larger capacity. However, if a motor having a larger capacity than necessary is used, the cost increases and the motor size increases, so that the installation property deteriorates.
　本実施例は、図９に示すように、ＰＷＭ波形の補正を行って、シャント電流ｉＲを確実に検出してモータ制御を行うことにより、容量がより大きな多相交流モータを用いることなしに、前記不安定領域Ｒ２をなくすことができるものである。以下、図９を用いて、ＰＷＭ波形の補正方法について説明する。 This embodiment, as shown in FIG. 9, by performing the correction of the PWM waveform, by performing reliably detected to motor control shunt current i R, without the capacity used larger multiphase AC motor The unstable region R2 can be eliminated. Hereinafter, a PWM waveform correction method will be described with reference to FIG.
　図９は、前記したｍｏｄｅ０からｍｏｄｅ１に移行する瞬間のＰＷＭ波形を示すものである。図９の上部には、パルス幅変調を行うために使用する搬送波Ｃ（ｔ）と、各スイッチング素子を駆動するためのＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－、および、検出されるシャント電流ｉＲの概形が示されている。 FIG. 9 shows a PWM waveform at the moment of transition from mode 0 to mode 1 described above. In the upper part of FIG. 9, a carrier wave C (t) used for pulse width modulation, PWM waveforms U +, U−, V +, V−, W +, W− for driving each switching element, and envelope of shunt current i R to be detected is shown.
　ここで、ＰＷＭ波形Ｖ－に着目する。前述したように、ＰＷＭ波形Ｖ－の立ち上がり時間とＰＷＭ波形Ｖ＋の立ち下がり時間の間、およびＰＷＭ波形Ｖ－の立ち下がり時間とＰＷＭ波形Ｖ＋の立ち上がり時間の間には、それぞれデッドタイムＤＴ以上の時間が必要である。また、シャント電流ｉＲを確実に検出するためには、ＰＷＭ波形Ｖ－がＨｉレベルになっている時間間隔が最小時間Ｔｍｉｎ以上である必要がある。 Here, attention is focused on the PWM waveform V−. As described above, the dead time DT or more is between the rise time of the PWM waveform V− and the fall time of the PWM waveform V +, and between the fall time of the PWM waveform V− and the rise time of the PWM waveform V +. I need time. Further, in order to reliably detect the shunt current i R needs time interval PWM waveform V- is in Hi level is minimum time T min or more.
　しかしながら、図９にあっては、ｍｏｄｅ０からｍｏｄｅ１に移行する時間において、Ｖ相の高圧側であるＰＷＭ波形Ｖ＋がＬｏレベルになっている時間（ＯＦＦ時間）であるｔｍ１＋ｔｍ２（以下、ＯＦＦ時間ｔｍと呼ぶ）が短いために、ＰＷＭ波形Ｖ－がＨｉレベルになっている時間（ＯＮ時間）が、最小時間Ｔｍｉｎに満たない。すなわち、（式１４）の関係になっている。
ｔｍ（＝ｔｍ１＋ｔｍ２）＜２ＤＴ＋Ｔｍｉｎ　　　　　（式１４） However, in FIG. 9, tm 1 + tm 2 (hereinafter referred to as OFF), which is the time (OFF time) that the PWM waveform V + on the high-voltage side of the V phase is at the Lo level in the time to shift from mode 0 to mode 1. Since the time (referred to as time tm) is short, the time during which the PWM waveform V- is at the Hi level (ON time) is less than the minimum time Tmin . That is, the relationship of (Expression 14) is established.
tm (= tm 1 + tm 2 ) <2DT + T min (Formula 14)
　ここで、（式１４）の時間ｔｍ２は、図９に図示するように、ＰＷＭ波形Ｖ＋がＬｏレベルになってからｍｏｄｅが変化するまでの時間であり、時間ｔｍ１は、ｍｏｄｅが変化した後でＰＷＭ波形Ｖ＋がＬｏレベルからＨｉレベルに変化するまでの時間である。 Here, the time tm 2 of (Equation 14) is the time from when the PWM waveform V + becomes Lo level until the mode changes, as shown in FIG. 9, and the time tm 1 changes the mode. This is the time until the PWM waveform V + subsequently changes from the Lo level to the Hi level.
　したがって、図９の点Ｓで示すように、ＰＷＭ波形Ｖ－がＨｉレベルになっている区間において、シャント電流ｉＲを確実に検出することができない。 Accordingly, as indicated by point S in FIG. 9, in a section where PWM waveform V- is in the Hi level, it is impossible to reliably detect the shunt current i R.
　なお、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－を生成した際に、ｍｏｄｅの変化点において（式１４）が成立するか否かを判断することができるため、実際にモータ制御を行う前に、前もってシャント電流ｉＲを確実に検出できるか否かを判断することができる。 Note that when the PWM waveforms U +, U−, V +, V−, W +, W− are generated, it can be determined whether or not (Equation 14) is satisfied at the mode change point. before performing the control, it can be determined whether the advance can be reliably detected shunt current i R.
　そして、シャント電流ｉＲを確実に検出できないときには、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－のタイミングを補正して、図９に示すＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’を生成する。以下、その補正方法について説明する。 When the shunt current i R cannot be reliably detected, the timings of the PWM waveforms U +, U−, V +, V−, W +, W− are corrected, and the PWM correction waveforms U + ′, U− ′, V + ', V-', W + ', W-' are generated. Hereinafter, the correction method will be described.
　まず、ｍｏｄｅ０からｍｏｄｅ１に移行する時間において、ＰＷＭ波形Ｖ－がＨｉレベルになっている区間の時間間隔を少なくとも最小時間Ｔｍｉｎまで延長する。これは、図９の点Ｐ３の位置を点Ｐ３’の位置までずらすことによって行う。このときにずらす時間の量は図９に示す補正量ｋで表される。 First, in the time to shift from mode 0 to mode 1, the time interval of the section in which the PWM waveform V- is at the Hi level is extended to at least the minimum time T min . This is done by shifting the position of point P 3 in FIG. 9 to the position of point P 3 ′. The amount of time shifted at this time is represented by a correction amount k shown in FIG.
　次に、ＰＷＭ波形Ｖ＋の立ち上がり時間を点Ｐ１から点Ｐ１’の位置まで移動する。このときの補正量は先に説明した補正量ｋと等しい。 Next, move the PWM waveform V + rise time from the point P 1 to the position of the point P 1 '. The correction amount at this time is equal to the correction amount k described above.
　これによって、ＰＷＭ補正波形Ｖ－’は、Ｈｉレベルの区間の時間間隔が最小時間Ｔｍｉｎであり、その両側にデッドタイムＤＴ分のＬｏレベル区間を有する波形となる。すなわち、ＰＷＭ補正波形Ｖ－’においては、（式１５）が成り立つ。
ｔｍ＝ｔｍ１＋ｔｍ２＝２ＤＴ＋Ｔｍｉｎ　　　　　（式１５） As a result, the PWM correction waveform V− ′ is a waveform in which the time interval of the Hi level section is the minimum time T min and has a Lo level section corresponding to the dead time DT on both sides thereof. That is, (Equation 15) is established in the PWM correction waveform V− ′.
tm = tm 1 + tm 2 = 2DT + T min (Formula 15)
　そして、ＰＷＭ補正波形Ｖ＋’では、パルスの立ち上がり時間を点Ｐ１から点Ｐ１’の位置まで移動したため、それに応じて、ＰＷＭ補正波形Ｖ＋’のパルスの立ち下がり時間を、ＰＷＭ波形Ｖ＋の立ち下がり時間である点Ｐ２から点Ｐ２’の位置まで移動する。このときの移動量は、前記補正量ｋと等しい。 Then, 'the, the rise time of the pulse from point P 1 points P 1' PWM correction waveform V + due to moved to the position of, in response thereto, the pulse fall time of the PWM correction waveform V + ', PWM waveform V + standing of moves from point P 2 is a fall time to the position of the point P 2 '. The movement amount at this time is equal to the correction amount k.
　このように、ＰＷＭ波形Ｖ＋，Ｖ－を補正してＰＷＭ補正波形Ｖ＋’，Ｖ－’とすることによって、波形のデューティ比が変化するため、他の相のデューティ比も、Ｖ相のデューティ比と一致するように変更する。 In this way, by correcting the PWM waveforms V + and V− to obtain the PWM corrected waveforms V + ′ and V− ′, the duty ratio of the waveform changes. Therefore, the duty ratios of the other phases are also V-phase duty ratios. To match.
　すなわち、Ｕ相については、ＰＷＭ波形Ｕ＋がＨｉレベルにある区間（点Ｐ４と点Ｐ５の間の区間）の時間間隔を左右とも補正量ｋずつ狭めて、点Ｐ４’と点Ｐ５’の位置に補正して、ＰＷＭ補正波形Ｕ＋’とする。そして、ＰＷＭ波形Ｕ－についても、Ｌｏレベルにある区間（点Ｐ６と点Ｐ７の間の区間）の時間間隔を左右とも補正量ｋずつ狭めて、点Ｐ６’と点Ｐ７’の位置に補正して、ＰＷＭ補正波形Ｕ－’とする。 That is, for the U-phase, narrow PWM waveform U + is by the correction amount k also influences the time interval of the section (section between the point P 4 and the point P 5) in the Hi level, the point P 4 'and the point P 5 It is corrected to the position of “to obtain a PWM correction waveform U +”. Also for the PWM waveform U−, the time interval of the section at the Lo level (the section between the points P 6 and P 7 ) is narrowed by the correction amount k on both the left and right sides, so that the points P 6 ′ and P 7 ′ The position is corrected to a PWM correction waveform U- '.
　また、Ｗ相については、ＰＷＭ波形Ｗ＋がＨｉレベルにある区間（点Ｐ８と点Ｐ９の間の区間）の時間間隔を左右とも補正量ｋずつ狭めて、点Ｐ８’と点Ｐ９’の位置に補正して、ＰＷＭ補正波形Ｗ＋’とする。そして、ＰＷＭ波形Ｗ－についても、Ｌｏレベルにある区間（点Ｐ１０と点Ｐ１１の間の区間）の時間間隔を左右とも補正量ｋずつ狭めて、点Ｐ１０’と点Ｐ１１’の位置に補正して、ＰＷＭ補正波形Ｗ－’とする。 As for the W-phase, narrow by the correction amount k also influences the time interval (the interval between the points P 8 and the point P 9) PWM waveform W + is at the Hi level section, the point and the point P 8 'P 9 It is corrected to the position of “to obtain a PWM correction waveform W +”. Also for the PWM waveform W−, the time interval of the section at the Lo level (section between the points P 10 and P 11 ) is narrowed by the correction amount k on both the left and right sides, and the points P 10 ′ and P 11 ′ The position is corrected to a PWM correction waveform W− ′.
　このようにして、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－をＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’に補正することによって、検出されるシャント電流ｉＲ’の波形は、図９の最下部に示すような形状になる。そして、ＰＷＭ波形の補正前には点Ｓにおいて検出できなかったシャント電流ｉＲが、ＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’を用いることによって、点Ｓ’を含む区間においてシャント電流ｉＲ’として確実に検出できるようになる。 In this way, by correcting the PWM waveforms U +, U−, V +, V−, W +, W− to PWM corrected waveforms U + ′, U− ′, V + ′, V− ′, W + ′, W− ′. The waveform of the detected shunt current i R ′ has a shape as shown at the bottom of FIG. Then, the shunt current i R that could not be detected at the point S before the correction of the PWM waveform is obtained by using the PWM correction waveforms U + ′, U− ′, V + ′, V− ′, W + ′, and W− ′. In the section including the point S ′, the shunt current i R ′ can be reliably detected.
　なお、ここで説明したのは、ｍｏｄｅ０からｍｏｄｅ１に移行する際のＰＷＭ波形の補正方法である。同様の補正が、ｍｏｄｅが移行する全てのシーンに対して同様に行われる。そして、前述した説明では、最初にＰＷＭ波形Ｖ－の補正を行ったが、ｍｏｄｅが移行するシーンに応じて、最初に補正する波形は、ＰＷＭ波形Ｕ－，Ｖ－，Ｗ－のうちいずれかになる。そして、時間ｔｍ１，ｔｍ２が定義されるＰＷＭ波形も、その都度変更される。 In addition, what was demonstrated here is the correction method of the PWM waveform at the time of shifting from mode0 to mode1. A similar correction is performed in the same manner for all scenes to which the mode shifts. In the above description, the PWM waveform V− is first corrected. However, the waveform to be corrected first is any one of the PWM waveforms U−, V−, and W− according to the scene in which the mode shifts. become. The PWM waveform in which the times tm 1 and tm 2 are defined is also changed each time.
　［作用説明］
次に、本実施例の一連の流れについて、図１０のフローチャートを用いて説明する。 [Description of operation]
Next, a series of flows of the present embodiment will be described with reference to the flowchart of FIG.
　（ステップＳ１００）目標電圧波形生成部４４において、多相交流モータ２０を駆動するための目標電圧となる３相交流電圧波形ＥＵ，ＥＶ，ＥＷを生成する。その波形は、例えば目標電圧波形生成部４４に予め格納されたソフトウェアによって生成される。 (Step S <b> 100) The target voltage waveform generation unit 44 generates three-phase AC voltage waveforms E U , E V , and E W that are target voltages for driving the multiphase AC motor 20. The waveform is generated by software stored in advance in the target voltage waveform generator 44, for example.
　（ステップＳ１０２）次に、目標電圧波形補正部４５において、先に生成された目標電圧波形の補正を行う。補正の目的、および補正方法は、先に説明した通りである。この補正によって、補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗが得られる。 (Step S102) Next, the target voltage waveform correction unit 45 corrects the previously generated target voltage waveform. The purpose of the correction and the correction method are as described above. By this correction, correction voltage waveforms E ′ U , E ′ V , E ′ W are obtained.
　（ステップＳ１０４）次に、ＰＷＭ波形生成部４６において、先に生成した補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗをパルス幅変調して、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－を生成する。パルス幅変調の方法は、先に説明した通りである。 (Step S104) Next, the PWM waveform generator 46 performs pulse width modulation on the previously generated correction voltage waveforms E ′ U , E ′ V , E ′ W , and PWM waveforms U +, U−, V +, V−. , W +, W−. The method of pulse width modulation is as described above.
　（ステップＳ１０５）次に、ＰＷＭ波形補正部４８において、シャント電流ｉＲが確実に検出できるように、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－を補正して、ＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’を生成する。ＰＷＭ波形の補正方法は、先に説明した通りである。なお、ここで、ＰＷＭ波形の補正が必要ないときには、補正は行わない。補正が必要か否かの判断は、前述した（１４式）に基づいて行う。 (Step S105) Next, the PWM waveform correction unit 48 corrects the PWM waveforms U +, U−, V +, V−, W +, and W− so that the shunt current i R can be reliably detected, and the PWM correction waveform. U + ', U-', V + ', V-', W + ', W-' are generated. The method for correcting the PWM waveform is as described above. Here, when correction of the PWM waveform is not necessary, correction is not performed. The determination as to whether correction is necessary is made based on the above-described (Equation 14).
　（ステップＳ１０６）次に、ＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’がドライブ回路５０に入力されて、さらに、対応するスイッチング素子に入力されて、各ＰＷＭ補正波形に対応するスイッチング素子の導通、非導通を制御する。なお、ドライブ回路５０は、マイクロコンピュータで構成された駆動パルス生成部４０と、ＩＧＢＴ等のスイッチング素子で構成されたインバータ３０とのインタフェースを兼ねている。 (Step S106) Next, the PWM correction waveforms U + ', U-', V + ', V-', W + ', W-' are input to the drive circuit 50 and further input to the corresponding switching elements, The conduction and non-conduction of the switching element corresponding to each PWM correction waveform is controlled. The drive circuit 50 also serves as an interface between the drive pulse generation unit 40 configured by a microcomputer and the inverter 30 configured by a switching element such as an IGBT.
　（ステップＳ１０８）直流電源６０が出力する電圧が、ＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’によって定められたタイミングで断続するインバータ３０によって３相交流電圧波形に変換されて、こうして生成された３相交流電圧波形が多相交流モータ２０に印加されて、多相交流モータ２０が回転する。 (Step S108) The voltage output from the DC power supply 60 is three-phase AC by the inverter 30 that is intermittent at the timing determined by the PWM correction waveforms U + ', U-', V + ', V-', W + ', W-'. The three-phase AC voltage waveform converted into a voltage waveform and thus generated is applied to the multi-phase AC motor 20, and the multi-phase AC motor 20 rotates.
　（ステップＳ１１０）抵抗値Ｒを有するシャント抵抗７０の両端にかかる電圧値が、電流検出部４２に読み込まれて、電流検出部４２の内部でＡ／Ｄ変換され、こうしてＡ／Ｄ変換された電圧値をシャント抵抗７０の抵抗値Ｒで除して、シャント抵抗７０を流れるシャント電流ｉＲの値が読み取られる。 (Step S110) The voltage value applied to both ends of the shunt resistor 70 having the resistance value R is read into the current detection unit 42 and A / D converted inside the current detection unit 42, and thus the A / D converted voltage. by dividing the value by the resistance value R of the shunt resistor 70, the value of the shunt current i R flowing through the shunt resistor 70 is read.
　（ステップＳ１１２）電流検出部４２において、シャント電流ｉＲが読み取られたときには、多相交流モータ２０の回転子２４の位置が推定できるため、そのままモータの制御を継続する。一方、電流検出部４２において、シャント電流ｉＲが読み取れないときには、ステップＳ１１６に進む。 (Step S112) current detector 42, when the shunt current i R has been read, it is possible to estimate the position of the rotor 24 of the polyphase alternating current motor 20, as it continues the control of the motor. On the other hand, the current detector 42, when the unreadable shunt current i R proceeds to step S116.
　（ステップＳ１１４）多相交流モータ２０の停止指令があるか否かを判断する。もし、多相交流モータ２０の停止指令があったときは、ＰＷＭ補正パルスの供給を中止してモータの回転を停止する。一方、多相交流モータ２０の停止指令がないときは、ステップＳ１０６に戻って、モータの制御を続行する。 (Step S114) It is determined whether or not there is a stop command for the multiphase AC motor 20. If there is a stop command for the multiphase AC motor 20, the supply of the PWM correction pulse is stopped and the rotation of the motor is stopped. On the other hand, when there is no stop command for the multiphase AC motor 20, the process returns to step S106 and the motor control is continued.
　（ステップＳ１１６）多相交流モータ２０の回転子２４の位置が推定できないときは、ＰＷＭ波形補正部４８において、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－をインバータ３０に供給するタイミングの調整を行う。 (Step S116) When the position of the rotor 24 of the multiphase AC motor 20 cannot be estimated, the PWM waveform correction unit 48 supplies the PWM waveforms U +, U−, V +, V−, W +, W− to the inverter 30. Adjust timing.
　（ステップＳ１１８）多相交流モータ２０の停止指令があるか否かを判断する。もし、多相交流モータ２０の停止指令があったときは、ＰＷＭ補正パルスの供給を中止してモータの回転を停止する。一方、多相交流モータ２０の停止指令がないときは、ステップＳ１０２に戻って、モータの制御を続行する。 (Step S118) It is determined whether or not there is a stop command for the multiphase AC motor 20. If there is a stop command for the multiphase AC motor 20, the supply of the PWM correction pulse is stopped and the rotation of the motor is stopped. On the other hand, when there is no stop command for the multiphase AC motor 20, the process returns to step S102 to continue the motor control.
　以上、説明したように、実施例１に係るモータ制御装置１０によれば、ＰＷＭ波形生成部４６が、電流検出部４２が検出した電流値に基づいて、多相交流モータ２０の各相に印加される瞬時電圧値の総和が０となるような目標電圧波形をそれぞれパルス幅変調した複数のＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－を生成して、ＰＷＭ波形補正部４８が、こうして生成された複数のＰＷＭ波形のパルス幅が所定時間以上になるように補正してＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’とし、直流電源６０が接続されたインバータ３０を制御して多相交流モータ２０を駆動するため、ＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－のパルス幅が長く補正されることによって、直流電源６０とインバータ３０の間に流れるシャント電流ｉＲを正確に検出することができ、これによって、多相交流モータ２０の回転を高い精度で制御することができる。 As described above, according to the motor control device 10 according to the first embodiment, the PWM waveform generation unit 46 is applied to each phase of the multiphase AC motor 20 based on the current value detected by the current detection unit 42. A plurality of PWM waveforms U +, U−, V +, V−, W +, W−, each of which is a pulse width modulation of the target voltage waveform such that the sum of instantaneous voltage values to be generated becomes 0, and a PWM waveform correction unit 48 However, correction is made so that the pulse widths of the plurality of PWM waveforms generated in this way are longer than a predetermined time to obtain PWM correction waveforms U + ', U-', V + ', V-', W + ', W-', and direct current In order to drive the multiphase AC motor 20 by controlling the inverter 30 to which the power supply 60 is connected, the pulse width of the PWM waveform U +, U−, V +, V−, W +, W− is corrected to be long, and the DC A flow between the power source 60 and the inverter 30 The cement current i R can be accurately detected, thereby, can be controlled with high precision the rotation of the multi-phase AC motor 20.
　また、実施例１に係るモータ制御装置１０によれば、ＰＷＭ波形補正部４８が、ＰＷＭ波形生成部４６によって生成された複数のＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－がインバータ３０を導通させる時間を、電流検出部４２によって、インバータ３０と直流電源６０との間に流れるシャント電流ｉＲを検出するのに要する最小時間Ｔｍｉｎよりも長くなるように補正するため、直流電源６０とインバータ３０の間に流れるシャント電流ｉＲを確実に検出することができ、これによって、多相交流モータ２０の回転を高い精度で制御することができる。 Further, according to the motor control apparatus 10 according to the first embodiment, the PWM waveform correction unit 48 generates a plurality of PWM waveforms U +, U−, V +, V−, W +, W− generated by the PWM waveform generation unit 46. In order to correct the time for which the inverter 30 is conducted to be longer than the minimum time T min required to detect the shunt current i R flowing between the inverter 30 and the DC power source 60 by the current detection unit 42, power source 60 and can be reliably detected shunt currents i R flowing between the inverter 30, thereby, it can be controlled with high precision the rotation of the multi-phase AC motor 20.
　また、実施例１に係るモータ制御装置１０によれば、ＰＷＭ波形補正部４８が、ＰＷＭ波形生成部４６によって生成された複数のＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－がインバータ３０を導通させる時間を、多相交流モータ２０に印加されるある相の高圧側のＰＷＭ波形のＯＦＦ時間をｔｍ、インバータ３０のうち、高圧側のインバータと低圧側のインバータが同時に導通しないようにするために設定するデットタイムをＤＴとしたとき、全ての相について、ｔｍを２ＤＴ＋Ｔｍｉｎよりも長くなるように補正して、ＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’とするため、直流電源６０とインバータ３０の間に流れるシャント電流ｉＲを確実に検出することができるようになり、これによって、多相交流モータ２０の回転を高い精度で制御することができる。 Further, according to the motor control apparatus 10 according to the first embodiment, the PWM waveform correction unit 48 generates a plurality of PWM waveforms U +, U−, V +, V−, W +, W− generated by the PWM waveform generation unit 46. The time during which the inverter 30 is turned on is the OFF time of the PWM waveform on the high-voltage side of a phase applied to the multiphase AC motor 20, and the high-voltage side inverter and the low-voltage side inverter of the inverter 30 are not turned on simultaneously. when the dead time to set was DT in order to, for all phases, by correcting the tm to be longer than 2DT + T min, PWM correction waveform U + ', U -', V + ', V-', W + ', W-' order to, will be able to reliably detect the shunt current i R flows between the DC power supply 60 and the inverter 30, thereby, multiphase AC motor 2 It is possible to control the rotational with high accuracy.
　また、実施例１に係るモータ制御装置１０によれば、ＰＷＭ波形補正部４８が、インバータ３０から多相交流モータ２０に供給される複数のＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－の相順が変化する時刻において、ＰＷＭ波形生成部４６によって生成された複数のＰＷＭ波形Ｕ＋，Ｕ－，Ｖ＋，Ｖ－，Ｗ＋，Ｗ－を補正してＰＷＭ補正波形Ｕ＋’，Ｕ－’，Ｖ＋’，Ｖ－’，Ｗ＋’，Ｗ－’とするため、複数のＰＷＭ波形の相順が変化する時刻において、複数のＰＷＭ波形の形状が一致したときであっても、ＰＷＭ波形がＰＷＭ波形補正部４８によって補正されるため、ＰＷＭ波形のパルス幅が長く補正されることによって、直流電源６０とインバータ３０の間に流れるシャント電流ｉＲを確実に検出することができるようになり、これによって、多相交流モータ２０の回転を高い精度で制御することができる。 Further, according to the motor control apparatus 10 according to the first embodiment, the PWM waveform correction unit 48 includes a plurality of PWM waveforms U +, U−, V +, V−, W +, supplied from the inverter 30 to the multiphase AC motor 20. At the time when the phase order of W− changes, a plurality of PWM waveforms U +, U−, V +, V−, W +, W− generated by the PWM waveform generator 46 are corrected to generate PWM corrected waveforms U + ′, U−. Since ', V +', V- ', W +', and W- ', even when the shapes of the plurality of PWM waveforms match at the time when the phase order of the plurality of PWM waveforms changes, Since the correction is performed by the PWM waveform correction unit 48, the shunt current i R flowing between the DC power supply 60 and the inverter 30 can be reliably detected by correcting the pulse width of the PWM waveform to be long. This Thus, the rotation of the multiphase AC motor 20 can be controlled with high accuracy.
　また、実施例１に係るモータ制御装置１０によれば、電流検出部４２が、インバータ３０と直流電源６０の間に直列に挿入されたシャント抵抗７０を流れるシャント電流ｉＲを検出するため、シャント電流ｉＲを簡単に検出することができる。 In addition, according to the motor control device 10 according to the first embodiment, the current detection unit 42 detects the shunt current i R flowing through the shunt resistor 70 inserted in series between the inverter 30 and the DC power source 60. it is possible to detect the current i R easy.
　また、実施例１に係るモータ制御装置１０によれば、目標電圧波形補正部４５が、多相交流モータ２０の各相に印加する３相交流電圧波形ＥＵ，ＥＶ，ＥＷに、各相の瞬時電圧値の中央値の半分の値を加えた波形を補正電圧波形Ｅ’Ｕ，Ｅ’Ｖ，Ｅ’Ｗとするため、異なる相間の線間電圧を変えずに、印加電圧の基本波成分の最大値をより大きくすることができる。すなわち、小さな電圧波形で従来と同等の線間電圧を得ることができるため、電圧の制御範囲を拡大することができる。 In addition, according to the motor control device 10 according to the first embodiment, the target voltage waveform correction unit 45 applies the three-phase AC voltage waveforms E U , E V , and E W applied to the respective phases of the multiphase AC motor 20 to the respective phases. Since the waveform obtained by adding half the median value of the instantaneous voltage value of the phase is the corrected voltage waveform E ′ U , E ′ V , E ′ W , the basics of the applied voltage without changing the line voltage between different phases The maximum value of the wave component can be further increased. That is, since a line voltage equivalent to the conventional one can be obtained with a small voltage waveform, the voltage control range can be expanded.
　また、実施例１で説明したモータ制御方法によって多相交流モータ２０を制御することにより、先に実施例１に係るモータ制御装置１０の効果として説明したように、直流電源６０とインバータ３０の間に流れるシャント電流ｉＲを正確に検出することができ、これによって、多相交流モータ２０の回転を高い精度で制御することができる。 Further, by controlling the multiphase AC motor 20 by the motor control method described in the first embodiment, as described as the effect of the motor control device 10 according to the first embodiment, the DC power supply 60 and the inverter 30 are the shunt current i R flows in can be accurately detected, thereby, it is possible to control the rotation of the multi-phase AC motor 20 with high accuracy.
　以上、この発明の実施例を図面により詳述してきたが、実施例はこの発明の例示にしか過ぎないものであるため、この発明は実施例の構成にのみ限定されるものではなく、この発明の要旨を逸脱しない範囲の設計の変更等があってもこの発明に含まれることは勿論である。 Although the embodiments of the present invention have been described in detail with reference to the drawings, the embodiments are only examples of the present invention, and the present invention is not limited to the configurations of the embodiments. Needless to say, design changes and the like within a range not departing from the gist of the invention are included in the present invention.
関連出願への相互参照Cross-reference to related applications
　本出願は、２０１３年３月５日に日本国特許庁に出願された特願２０１３－０４３４０９に基づいて優先権を主張し、その全ての開示は完全に本明細書で参照により組み込まれる。 This application claims priority based on Japanese Patent Application No. 2013-043409 filed with the Japan Patent Office on March 5, 2013, the entire disclosure of which is fully incorporated herein by reference.

Claims (12)

  1. 　多相交流モータと、
    　前記多相交流モータを駆動する複数のスイッチング素子からなるインバータと、
    　前記インバータに直流電圧を供給する直流電源と、
    　前記インバータと前記直流電源の間に流れる電流を検出する電流検出部と、
    　前記電流検出部によって検出された電流値に基づいて、前記インバータの導通、および非導通を制御する複数の駆動パルスを、前記多相交流モータの各相に印加される瞬時電圧値の総和が０となるような目標電圧波形をそれぞれパルス幅変調することによって生成するＰＷＭ波形生成部と、
    　前記ＰＷＭ波形生成部によって生成された複数のＰＷＭ波形のパルス幅を、前記インバータを導通させる時間が所定値以上になるように補正するＰＷＭ波形補正部と、を有することを特徴とするモータ制御装置。     A polyphase AC motor,
    An inverter composed of a plurality of switching elements for driving the polyphase AC motor;
    A DC power supply for supplying a DC voltage to the inverter;
    A current detection unit for detecting a current flowing between the inverter and the DC power supply;
    Based on the current value detected by the current detection unit, the sum of the instantaneous voltage values applied to each phase of the multiphase AC motor is 0 as a plurality of drive pulses for controlling conduction and non-conduction of the inverter. A PWM waveform generator that generates a target voltage waveform such that
    A motor control apparatus comprising: a PWM waveform correction unit that corrects the pulse widths of the plurality of PWM waveforms generated by the PWM waveform generation unit so that the time during which the inverter is conducted becomes equal to or greater than a predetermined value; .
  2. 　前記ＰＷＭ波形補正部は、前記ＰＷＭ波形生成部によって生成された複数のＰＷＭ波形が前記インバータを導通させる時間を、前記電流検出部によって、前記インバータと前記直流電源との間に流れる電流を検出するのに要する最小時間Ｔｍｉｎよりも長くなるように補正することを特徴とする請求項１に記載のモータ制御装置。 The PWM waveform correction unit detects a time during which the plurality of PWM waveforms generated by the PWM waveform generation unit conduct the inverter, and detects a current flowing between the inverter and the DC power source by the current detection unit. The motor control device according to claim 1, wherein the motor control device is corrected so as to be longer than a minimum time T min required for the operation.
  3. 　前記ＰＷＭ波形補正部は、前記ＰＷＭ波形生成部によって生成された複数のＰＷＭ波形が前記インバータを導通させる時間を、前記多相交流モータに印加されるある相の高圧側のＰＷＭ波形のＯＦＦ時間をｔｍ、前記インバータのうち、高圧側のインバータと低圧側のインバータが同時に導通しないようにするために設定するデットタイムをＤＴとしたとき、全ての相について、前記ｔｍを２ＤＴ＋Ｔｍｉｎよりも長くなるように補正することを特徴とする請求項１に記載のモータ制御装置。 The PWM waveform correction unit sets a time during which a plurality of PWM waveforms generated by the PWM waveform generation unit conduct the inverter, and an OFF time of a high-voltage side PWM waveform applied to the multiphase AC motor. tm, of the inverter, when the dead time the high pressure side of the inverter and the low-voltage side of the inverter is set in order to avoid simultaneous conduction was DT, for all phases, so that is longer than the tm 2DT + T min The motor control apparatus according to claim 1, wherein the motor control apparatus corrects to
  4. 　前記ＰＷＭ波形補正部は、前記インバータから前記多相交流モータに供給される複数のＰＷＭ波形の相順が変化する前後の時刻において、前記ＰＷＭ波形生成部によって生成されたＰＷＭ波形を補正することを特徴とする請求項１に記載のモータ制御装置。 The PWM waveform correction unit corrects the PWM waveform generated by the PWM waveform generation unit at a time before and after the phase sequence of a plurality of PWM waveforms supplied from the inverter to the multiphase AC motor changes. The motor control device according to claim 1, wherein
  5. 　前記電流検出部は、前記インバータと前記直流電源の間に直列に挿入された抵抗を流れる電流を検出するものであることを特徴とする請求項１に記載のモータ制御装置。 The motor control device according to claim 1, wherein the current detection unit detects a current flowing through a resistor inserted in series between the inverter and the DC power supply.
  6. 　前記多相交流モータの各相に印加する前記目標電圧波形に、前記目標電圧波形の各相の瞬時電圧値の中央値の半分の値を加えた波形を新たな目標電圧波形とする目標電圧波形補正部を有することを特徴とする請求項１に記載のモータ制御装置。 A target voltage waveform having a new target voltage waveform obtained by adding a half of the median value of the instantaneous voltage value of each phase of the target voltage waveform to the target voltage waveform applied to each phase of the multiphase AC motor The motor control device according to claim 1, further comprising a correction unit.
  7. 　複数のスイッチング素子からなるインバータによって多相交流モータを駆動するモータ制御方法であって、
    　前記インバータと前記インバータに直流電圧を供給する直流電源との間に流れる電流を検出して、
    　検出された電流値に基づいて、前記インバータの導通、および非導通を制御する複数の駆動パルスを、前記多相交流モータの各相に印加される瞬時電圧値の総和が０となる目標電圧波形をそれぞれパルス幅変調することによって生成して、
    　こうして生成された複数のＰＷＭ波形のパルス幅を、前記インバータを導通させる時間が所定値以上になるように補正して前記インバータを制御することによって前記多相交流モータを駆動することを特徴とするモータ制御方法。     A motor control method for driving a polyphase AC motor by an inverter composed of a plurality of switching elements,
    Detecting a current flowing between the inverter and a DC power source that supplies a DC voltage to the inverter,
    Based on the detected current value, a plurality of drive pulses for controlling conduction and non-conduction of the inverter, a target voltage waveform in which the sum of instantaneous voltage values applied to each phase of the multiphase AC motor is zero Are each generated by pulse width modulation,
    The multi-phase AC motor is driven by controlling the inverter after correcting the pulse widths of the plurality of PWM waveforms generated in this way so that the time for which the inverter is conducted becomes a predetermined value or more. Motor control method.
  8. 　前記複数のＰＷＭ波形のパルス幅は、前記複数のＰＷＭ波形が前記インバータを導通させる時間が、前記インバータと前記直流電源との間に流れる電流を検出できる最小時間Ｔｍｉｎよりも長くなるように補正されることを特徴とする請求項７に記載のモータ制御方法。 The pulse widths of the plurality of PWM waveforms are corrected so that the time during which the plurality of PWM waveforms make the inverter conductive is longer than the minimum time T min during which the current flowing between the inverter and the DC power source can be detected. The motor control method according to claim 7, wherein:
  9. 　前記ＰＷＭ波形が前記インバータを導通させる時間は、前記多相交流モータに印加されるある相の高圧側のＰＷＭ波形のＯＦＦ時間をｔｍ、前記インバータのうち、高圧側のインバータと低圧側のインバータが同時に導通しないようにするために設定するデットタイムをＤＴとしたとき、全ての相について、前記ｔｍが２ＤＴ＋Ｔｍｉｎよりも長くなるように補正されることを特徴とする請求項７に記載のモータ制御方法。 The time during which the PWM waveform makes the inverter conductive is the OFF time of the PWM waveform on the high-voltage side of a phase applied to the multiphase AC motor, and among the inverters, the high-voltage side inverter and the low-voltage side inverter are 8. The motor control according to claim 7, wherein when the dead time set to prevent conduction at the same time is DT, the tm is corrected to be longer than 2DT + T min for all phases. Method.
  10. 　前記複数のＰＷＭ波形のパルス幅は、前記インバータから前記多相交流モータに供給される複数のＰＷＭ波形の相順が変化する前後の時刻において補正されることを特徴とする請求項７に記載のモータ制御方法。 The pulse width of the plurality of PWM waveforms is corrected at a time before and after the phase sequence of the plurality of PWM waveforms supplied from the inverter to the multiphase AC motor is changed. Motor control method.
  11. 　前記インバータと前記直流電源との間に流れる電流は、前記インバータと前記直流電源の間に直列に挿入された抵抗を流れる電流として検出することを特徴とする請求項７に記載のモータ制御方法。 The motor control method according to claim 7, wherein a current flowing between the inverter and the DC power supply is detected as a current flowing through a resistor inserted in series between the inverter and the DC power supply.
  12. 　前記多相交流モータの各相に印加する前記目標電圧波形に、前記目標電圧波形の各相の瞬時電圧値の中央値の半分の値を加えた波形を新たな目標電圧波形とすることを特徴とする請求項７に記載のモータ制御方法。 A waveform obtained by adding a half of the median value of the instantaneous voltage value of each phase of the target voltage waveform to the target voltage waveform applied to each phase of the multiphase AC motor is used as a new target voltage waveform. The motor control method according to claim 7.
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