WO2013185734A2 - 相干光通信***中色散和非线性补偿方法及*** - Google Patents

相干光通信***中色散和非线性补偿方法及*** Download PDF

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WO2013185734A2
WO2013185734A2 PCT/CN2013/081517 CN2013081517W WO2013185734A2 WO 2013185734 A2 WO2013185734 A2 WO 2013185734A2 CN 2013081517 W CN2013081517 W CN 2013081517W WO 2013185734 A2 WO2013185734 A2 WO 2013185734A2
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sample point
compensation
nonlinear
dispersion
phase angle
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PCT/CN2013/081517
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English (en)
French (fr)
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WO2013185734A3 (zh
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沈百林
喻松
杨杰
李敏良
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中兴通讯股份有限公司
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Priority to KR1020157012750A priority Critical patent/KR101686927B1/ko
Priority to EP13805082.8A priority patent/EP2922219B1/en
Publication of WO2013185734A2 publication Critical patent/WO2013185734A2/zh
Publication of WO2013185734A3 publication Critical patent/WO2013185734A3/zh

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/25—Arrangements specific to fibre transmission
    • H04B10/2507—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion
    • H04B10/2513—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion due to chromatic dispersion
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/25—Arrangements specific to fibre transmission
    • H04B10/2507—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion
    • H04B10/2543—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion due to fibre non-linearities, e.g. Kerr effect
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/29—Repeaters
    • H04B10/291—Repeaters in which processing or amplification is carried out without conversion of the main signal from optical form
    • H04B10/299—Signal waveform processing, e.g. reshaping or retiming
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/60—Receivers
    • H04B10/61—Coherent receivers
    • H04B10/616—Details of the electronic signal processing in coherent optical receivers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B2210/00—Indexing scheme relating to optical transmission systems
    • H04B2210/25—Distortion or dispersion compensation
    • H04B2210/252—Distortion or dispersion compensation after the transmission line, i.e. post-compensation

Definitions

  • the present invention relates to the field of communications, and more particularly to a method and system for dispersion and nonlinear compensation in a coherent optical communication system.
  • IM/DD Intensity Modulation/Direct Detection
  • SNR Signal to Noise Ratio
  • the principle of the polarization multiplexed wavelength division multiplexing coherent receiving system is as shown in FIG. 1.
  • the received signals are optically demultiplexed and multiplexed to obtain signal light of each channel, and then the polarization multiplexer is used to receive polarization multiplexing.
  • the signal light of the X and Y polarization states is obtained separately, and the local oscillator is also combined with the polarization beam splitter to obtain the local oscillator of the X and Y polarization states of the synchronous intensity.
  • the signal light of the X and Y polarization states and the local oscillator light will respectively enter the coherent receiver of the corresponding polarization state for coherent demodulation, and after coherent demodulation, four signals of XI, XQ, YI and YQ of the channel are obtained, four paths
  • DSP Digital Signal Processing
  • the 16QAM (Quadature Amplitude Modulation) format of polarization multiplexed wavelength division multiplexing is due to its high modulation order and extremely high Spectral efficiency has attracted extensive research.
  • digital signal processing technology in the system of polarization mode dispersion, polarization crosstalk and frequency phase noise have a more mature processing scheme, and the effects of simultaneous dispersion and nonlinear effects have become the constraints of high-speed digital coherence.
  • the main factors in the development of optical communication systems Considering the cost of the actual commercial system, the dispersion and nonlinear compensation algorithms can not be too high in complexity while ensuring compensation accuracy. In recent research progress, relevant researchers have proposed dispersion and nonlinear compensation methods, which can be roughly divided into three categories.
  • the lumped nonlinear compensation method mainly compensates the phase noise caused by the SPM (Self-Phase Modulation) effect by the phase modulator, and the phase added by the phase modulation is directly related to the strength of the signal.
  • This method can compensate for the effects of SPM effect in the electrical or optical domain with low complexity, but this method ignores the influence of dispersion in the transmission process, and is only applicable to systems with low dispersion or compensated dispersion, and the compensation effect Poor.
  • the filter compensation method based on Volterra series, firstly construct a filter that conforms to the Volterra series structure, obtain the tap coefficients that can compensate the nonlinear effects of the current system through the training sequence, and then compensate the actual data with the filter after training. While maintaining the dynamic update of the filter coefficients to ensure that the filter can always maintain the effectiveness of the compensation.
  • the update filter coefficients are uniformly combined with the LMS (Minimum Mean Square Error) algorithm.
  • LMS Minimum Mean Square Error
  • This method makes full use of the modeling ability and compensation ability of Volterra series for nonlinear systems.
  • the algorithm has low complexity and good compensation effect, but it needs to compensate the dispersion of the fiber compensation system by using dispersion compensation, which increases the hardware complexity of the system. At the same time, the compensation capability of the WDM (Wavelength Division Multiplexing) system is poor.
  • SSFM that is, divides the fiber into N small segments, assuming that the light field is separated by dispersion and nonlinearity through each fiber micro-element, and an approximate result is obtained.
  • the traditional dispersion compensation is performed on all the sample points by combining the dispersion compensation operator, and then the nonlinear phase angle is obtained by calculating the power of each point, and then the reverse compensation is performed.
  • the reverse transmission compensation method can simultaneously process dispersion and nonlinearity, eliminates the need for dispersion compensation fiber, reduces hardware cost, and has high compensation accuracy.
  • the compensation method can be applied to deal with polarization multiplexing and wavelength division multiplexing after proper deformation.
  • the system is very versatile. However, the complexity of the algorithm is very high. Correlation analysis shows that in a single-carrier system, for a sequence of N-point samples, the complexity of processing in a single fiber micro-element is 14N + 3[(N + P)l. g 2 (N + P) + (N + P)] ; where p is the additional number of sample overheads. Combined with the total number of compensation steps and the number of cycles, the reverse transmission compensation method will bring high computational overhead.
  • Embodiments of the present invention provide a dispersion and nonlinear compensation method and system in a coherent optical communication system, which overcomes the defects of high computational complexity, dispersion compensation fiber compensation dispersion, and the like.
  • Embodiments of the present invention disclose a dispersion and nonlinear compensation method in a coherent optical communication system, including:
  • the total length of the fiber is divided into N steps of equal length, and each data point in the step is firstly combined with dispersion compensation operator in each step to perform dispersion compensation, and then nonlinearly compensated;
  • the step of performing nonlinear compensation in each step includes:
  • the 2k+l sample points of the X and Y polarization states are sampled, and the power of the 2 (2k+l) sample points is calculated, and the X and Y are calculated.
  • the powers of the same points in the polarization state are added to obtain 2k+1 power values, and the 2k+1 power values are weighted and summed, and the weighted summation values are multiplied by the set coefficients.
  • N and k are positive integers.
  • the step of performing weighted summation on the 2k+1 power values, and multiplying the weighted summation value by the set coefficient W to obtain the nonlinear phase angle of the current sample point includes: Calculate the nonlinear phase angle of the current sample point as described in the following formula: I 2 + 1 y- k ⁇ 2 )+ ⁇ ( *- ⁇ )( ⁇ - ⁇ ) I 2 + 1 y ⁇ 2 )+ ⁇ + ⁇ 0 (
  • 2 )] , in the above formula, ( ..] ⁇ ) indicates the weighting factor of the power value of the sample point, ⁇ - ⁇ 3 ⁇ 4:- 7 people.
  • the weighting coefficient of the power value of each sample point is inversely proportional to the distance between the sample point and the current sample point.
  • N is an integer, 2 N 5 .
  • k is an integer, l k 10.
  • the weighting coefficient of the power value of each sample point is a positive number greater than 0.
  • the embodiment of the invention also discloses a dispersion and nonlinear compensation system in a coherent optical communication system, comprising:
  • a third module which is configured to: perform nonlinear compensation within each step
  • the third module is configured to perform nonlinear compensation in each step in the following manner: centering on the position of the current sample point, performing 2k+l sample points of the X and Y polarization states. For example, calculating the power of the 2 ( 2k + l ) sample points, adding the powers of the same sample points on the X and Y polarization states to obtain 2k + 1 power values, for the 2k +l power values are weighted and summed, and the weighted summed values are multiplied by the set coefficient W to obtain a nonlinear phase angle of the current sample point, and the nonlinear phase angle is compensated according to the nonlinear phase angle;
  • N and k are positive integers.
  • the third module is configured to calculate a nonlinear phase angle of the current sample point according to the following calculation formula:
  • the weighting coefficient of the power value of each sample point is inversely proportional to the distance between the sample point and the current sample point.
  • N is an integer, 2 N 5 .
  • k is an integer, l k 10.
  • the weighting coefficient of the power value of each sample point is a positive number greater than zero.
  • the dispersion compensation fiber is not needed to compensate the dispersion, the calculation method of the nonlinear phase angle is improved, the step size is improved, and the computational complexity is reduced, and the polarization multiplexing wave is applied to the 16QAM format.
  • Sub-multiplexing system BRIEF abstract
  • 1 is a schematic diagram of a principle of a polarization multiplexed wavelength division multiplexing coherent receiving system
  • FIG. 2 is a schematic diagram of a principle of a dispersion and nonlinear compensation method for a polarization multiplexed wavelength division multiplexing system of a 16QAM format according to an embodiment of the present invention
  • Figure 3 is a schematic diagram showing the effect of different OSNR on dispersion and nonlinear compensation effects
  • Figure 4 is a star-seat diagram of the X and Y polarization states after the compensation method of Figure 2 for a signal-to-noise ratio of 25.16 dB.
  • This embodiment introduces a dispersion and nonlinear compensation method in a coherent optical communication system, and the principle of the scheme is as shown in FIG. 2.
  • the method uses the low complexity dispersion and nonlinear compensation algorithm to compensate the dispersion and nonlinearity in the polarization multiplexed wavelength division multiplexing system of 16QAM format.
  • the compensation process is explained below. First, the total length of the fiber is divided into N parts of equal length (N is much smaller than the requirements of the traditional inverse compensation algorithm), that is, N length steps are equal, and the dispersion and nonlinear effects are compensated separately in each step. First, the dispersion compensation operator is used to perform dispersion compensation on all data points in the step size; then nonlinear compensation is performed.
  • the nonlinear phase angle is no longer calculated by using the power of the current sample point as in the conventional reverse transmission compensation method, but is centered on the position of the current sample point, and X and The Y polarization states each 2k+l sample points are sampled, and the power of the 2 (2k+l) sample points is calculated, and the powers of the same sample points in the X and Y polarization states are added to obtain 2k. +l power values, then weighting and summing the 2k+l power values, multiplying the weighted summation value by a set coefficient W to obtain the nonlinear phase angle of the current sample point, according to the nonlinearity
  • the phase angle is compensated, wherein N and k are positive integers.
  • the nonlinear phase angle of the current sample point can be calculated as follows:
  • the weighting coefficient of the power value of each sample point is a positive number greater than 0, which can be determined empirically.
  • the weighting coefficient of the power value of each sample point is inversely proportional to the distance between the sample point and the current sample point. That is, for a sample point farther from the current sample point position, the weighting coefficient configured for its power value is smaller, and for the sample point closer to the current sample point position, the weighting coefficient of its power value is configured. The bigger it is.
  • the complexity of the dispersion and the nonlinear compensation is proportional to the number of steps N to be divided, and the conventional step compensation algorithm has a smaller compensation step size and a larger total number of steps. Therefore, the amount of calculation is large.
  • the priority scheme of this embodiment limits the value of N, and preferentially recommends N as an integer greater than or equal to 2 and less than or equal to 5, thereby greatly reducing computational complexity, reducing computation, and improving compensation efficiency.
  • the single-step nonlinear phase angle compensation amplitude is improved by finding the power weighted summation, and the total number of steps is also greatly reduced, and the complexity is reduced.
  • the 2k+l sample points centered on the position of the current sample point are also limited, that is, the sample point The number cannot be too much or too small, and lk 10 is preferred, and k is an integer.
  • setting coefficient W also called adjustment factor
  • the values are set to different values depending on the application system, that is, according to different system settings (for example, the fiber input power of each polarization state, the transmission length, and the system signal-to-noise ratio, etc.), the appropriate W is set as long as Ensure that the value of W is set to optimize system performance, which greatly increases the flexibility and scope of the system.
  • the different optical signal to noise ratio (OSNR) of the 16QAM polarization multiplexed wavelength division multiplexing system affects the dispersion and nonlinear compensation effects.
  • the laser line width shown in Fig. 3 is 100 kHz
  • the transmission length is 1000 km
  • the optical fiber power per polarization state is ldBm
  • the symbol length is 8192
  • the symbol rate is 180 Gbaud.
  • the system can maintain the bit error rate on the order of 10 - 3 within a certain signal to noise ratio range.
  • Figure 4 shows the constellation of the X and Y polarization states after the SNR and nonlinear compensation methods shown in Figure 2 above when the signal-to-noise ratio is 25.16 dB.
  • the laser line width shown in the figure is 100 kHz
  • the transmission length is 1000 km
  • the optical fiber power per polarization state is ldBm
  • the symbol length is 8192
  • the symbol rate is 180 Gbaud.
  • This embodiment describes a dispersion and nonlinear compensation system in a coherent optical communication system, the system including at least:
  • the first module is configured to: divide the total length of the optical fiber into N steps of equal length; the second module is configured to: combine each of the data points in the step by combining a dispersion compensation operator in each step Dispersion compensation;
  • a third module which is configured to: perform nonlinear compensation within each step
  • the third module is configured to perform nonlinear compensation in each step in the following manner: centering on the position of the current sample point, and performing 2k+l sample points on the X and Y polarization states, Calculating the power of the 2 ( 2k + l ) sample points, adding the powers of the same sample points in the X and Y polarization states to obtain 2k + 1 power values, and performing the 2k + 1 power values Weighted summation, then The value of the weighted sum is multiplied by the set coefficient W to obtain a nonlinear phase angle of the current sample point, and is compensated according to the nonlinear phase angle; wherein N and k are positive integers.
  • the third module can calculate the nonlinear phase angle of the current sample point according to the following formula:
  • the weighting coefficient of the power value of each sample point is a positive number greater than 0, which can be determined empirically.
  • the weighting coefficient of the power value of each sample point is inversely proportional to the distance between the sample point and the current sample point. That is, for a sample point farther from the current sample point position, the weighting coefficient configured for its power value is smaller, and for the sample point closer to the current sample point position, the weighting coefficient of its power value is configured. The bigger it is.
  • the complexity of dispersion and nonlinear compensation is proportional to the number of steps N that is divided.
  • the traditional step compensation algorithm has a smaller compensation step size and a larger total number of steps, so the calculation is large. .
  • the priority scheme of this embodiment limits the value of N, and preferentially recommends N as an integer greater than or equal to 2 and less than or equal to 5, thereby greatly reducing computational complexity, reducing computation, and improving compensation efficiency.
  • the 2k+l sample points centered on the position of the current sample point are also limited, that is, the sample is The number of points should not be too much or too little, and lk 10 is preferred, and k is an integer.
  • the coefficient W also called the adjustment factor
  • the value of the W can be set differently depending on the application system.
  • the value that is, according to different system settings (for example, the fiber input power of each polarization state, the transmission length and the system signal-to-noise ratio, etc.), set the appropriate W, as long as the value of W is set to maximize the system performance. Excellent, this can greatly increase the flexibility and scope of the system.
  • the dispersion compensation fiber is not required to compensate the dispersion, and is particularly suitable for the 16QAM polarization multiplexed wavelength division multiplexing coherent optical communication system, thereby avoiding the traditional reverse transmission compensation method.
  • the large amount of calculation caused by the small compensation of the fiber subdivision Take the X and Y polarization states centered on the current sample point 2k+l (the k value needs to be adjusted according to the specific system, the k value in this system is 10), and calculate the 2 ( 2k+ l) the power of a sample point, add the power of the same point on the X and Y polarization states to obtain 2k+l power values, and then find the weighted sum of the 2k+l power values, this weighting And multiply by an adjustment factor W to obtain the nonlinear phase angle of the current sample point.
  • the adjustment factor W can also be adjusted according to different systems.
  • the dispersion compensation fiber is not needed to compensate the dispersion, the calculation method of the nonlinear phase angle is improved, the step size is improved, and the calculation complexity is reduced.

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  • Electromagnetism (AREA)
  • Engineering & Computer Science (AREA)
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Abstract

一种相干光通信***中色散和非线性补偿方法及***。所述方法包括：将光纤总长划分为等长的N个步长，在各步长内先进行色散补偿，再进行非线性补偿；其中，在各步长内进行非线性补偿的步骤包括：以当前采样点为中心，对X和Y偏振态各2k+1个采样点采样，计算2（2k+1）个采样点的功率，将X和Y偏振态上位置相同的采样点的功率相加，得到2k+1个功率值，对2k+1个功率值加权求和，再将加权求和的值乘以设定系数W后得到当前采样点的非线性相角，根据非线性相角进行补偿，其中，所述N和k为正整数。本发明实施例的技术方案中，改进了非线性相角的计算方式，降低了计算复杂度。

Description

相干光通信***中色散和非线性补偿方法及***
技术领域
本发明涉及通信领域， 尤其涉及相干光通信***中的色散和非线性补偿 方法及***。
背景技术
光纤通信***始于 20世纪 70年代初， 釆用强度调制的半导体激光器， 接收到的光强度信号通过光电二极管转换成电信号。 这种方式被称为强度调 制 /直接检测 (Intensity Modulation/Direct Detection , IM/DD) , 并在当前的商用 光纤通信***中得到广泛的应用。 但是 IM/DD有着其固有的不足， 比如， 接 收及灵敏度受到噪声的限制， 频谱效率（Spectral Efficiency )不高， 不能够充 分利用带宽等。 相干检测受到人们的广泛关注， 一方面是因为相干检测允许 ***釆用更高阶的调制格式， 提高频谱利用率， 另外一方面相干检测能够提 高信噪比 (Signal to Noise Ratio, SNR)。
偏振复用波分复用相干接收***的原理如图 1所示， 首先对接收到的信 号进行光解波分复用， 得到各个信道的信号光， 然后利用偏振分束器接收偏 振复用， 分别得到 X和 Y偏振态的信号光， 同时本振光也要结合偏振分束器 得到同步调同强度的 X和 Y偏振态的本振光。 然后 X和 Y偏振态的信号光 和本振光将分别进入对应偏振态的相干接收机进行相干解调， 经过相干解调 后得到该信道的 X I、 X Q 、 Y I和 Y Q四路信号， 四路信号接下来进行 DSP ( Digital Signal Processing, 数字信号处理） ， 包括色散非线性补偿、 解偏振 串扰和频率相位噪声补偿。 在频率相位噪声以及偏振串扰已经有较为成熟的 解决方案的今天， 色散和非线性效应对光信号的影响已成为制约高速相干光 通信***发展的主要因素。
非线性效应的影响可由简化的非线性薛定谔方程（ Non Linear Schrodinger Equation, NLSE )—― ^ / " / 2 - l4 /23r2 + |2 = 0来描述，其中的 是信号光的复包络， 是色散相关的系数， 是归一化时间。 可见非线性效 应会使信号光的相位发生与幅度相关的变化， 同时色散也将对信号光的幅度 有影响。 色散和非线性的同时作用增加了问题的复杂度， 同时使得对二者的 补偿比较难以处理。
作为新一代高速数字相干光通信***中极具前景的技术之一， 偏振复用 波分复用的 16QAM ( Quadrature Amplitude Modulation, 正交振幅调制）格式 因其较高的调制阶数和极高的频谱效率吸引了广泛的研究。 随着数字信号处 理技术的发展， 在略去偏振模色散的***中， 偏振串扰和频率相位噪声已经 有较为成熟的处理方案， 而色散和非线性效应同时作用产生的影响已成为制 约高速数字相干光通信***发展的主要因素。 考虑到实际商用***的成本， 色散和非线性的补偿算法在保证补偿精度的同时复杂度不能太高。 最近的研 究进展中， 相关的研究学者纷纷提出了色散和非线性的补偿方法， 这些方法 大体上可以分为三类。
集总非线性补偿法主要通过相位调制器来补偿 SPM ( Self-Phase Modulation, 自相位调制）效应造成的相位噪声， 其相位调制附加的相位是与 信号的强度直接相关的。 这个方法可以在电域或光域以较低的复杂度补偿 SPM效应造成的影响， 但是该方法忽略色散在传输过程中的影响， 只适用于 低色散或已补偿了色散的***， 且补偿效果欠佳。
基于 Volterra级数的滤波器补偿法中，首先构建符合 Volterra级数结构的 滤波器， 通过训练序列获取能够补偿当前***非线性效应的抽头系数， 然后 用训练完成后的滤波器对实际数据进行补偿， 同时保持滤波器系数的动态更 新以保证滤波器能始终保持补偿的有效性。 更新滤波器系数统一釆用 LMS ( Minimum Mean Square Error, 最小均方误差）算法。 这个方法充分利用了 Volterra级数对非线性***的建模能力和补偿能力， 算法复杂度较低， 补偿效 果较好， 但是需要用色散补偿光纤补偿***的色散， 增加了***的硬件复杂 度， 同时对 WDM ( Wavelength Division Multiplexing, 波分多路复用）***的 补偿能力欠佳。
基于分步傅里叶算法（ Split-Step Fourier Method, SSFM ) 的反向传输非 线性补偿法的基本原理可以用推导 NLSE 的反向传输方程—— dE / dz = (D-l + N~l )E 来 表 示 ， 其 中 D~l = α / 2 + ίβ2 2 / 2 \ dt2 - β 3 / 3 \ dt3 , = ~ίγ^2 , 分别是补偿色散和非线性的逆向算子。 实际补偿时釆用的是 SSFM, 即， 将光纤分为 N个小段， 假定光场通过每个光纤微元时色散和非 线性分开作用， 得到近似结果。 就是在每个光纤微元内， 先结合色散补偿算 子对所有釆样点进行传统的色散补偿， 然后通过计算每个点的功率获得非线 性相角， 再进行逆向补偿。
反向传输补偿法能够同时处理色散和非线性， 无需色散补偿光纤， 降低 了硬件成本， 补偿精度较高， 该补偿法做适当的变形后还能适用于处理偏振 复用和波分复用的***， 适用性很广。 但是算法复杂度很高， 相关分析表明， 在单载波***中， 对于一个 N点的釆样点序列， 单个光纤微元中处理的复杂 度是 14N + 3[(N + P)l。g2(N + P) + (N + P)] ; 其中， p是额外的样点开销个数。 结合 总的补偿步数和循环数， 反向传输补偿法将带来很高的计算开销。
发明内容
本发明实施例提供一种相干光通信***中色散和非线性补偿方法及系 统， 以克服现有算法计算复杂度高、 需要色散补偿光纤补偿色散等缺陷。
本发明实施例公开了一种相干光通信***中色散和非线性补偿方法， 包 括：
将光纤总长划分为长度相等的 N个步长， 在每个步长内先结合色散补偿 算子对所述步长内的各数据点进行色散补偿， 再进行非线性补偿；
其中， 所述在每个步长内进行非线性补偿的步骤包括：
以当前釆样点的位置为中心， 对 X和 Y偏振态各 2k+l个釆样点进行釆 样， 计算所述 2 ( 2k+l )个釆样点的功率， 将所述 X和 Y偏振态上位置相同 的釆样点的功率相加， 得到 2k+l个功率值， 对所述 2k+l个功率值进行加权 求和，再将所述加权求和的值乘以设定系数 W后得到所述当前釆样点的非线 性相角， 根据所述非线性相角进行补偿；
其中， 所述 N和 k为正整数。
可选地， 上述对 2k+l个功率值进行加权求和，再将所述加权求和的值乘 以设定系数 W后得到所述当前釆样点的非线性相角的步骤包括：根据如下计 算公式所述当前釆样点的非线性相角： I2 + 1 y-k Ι2)+^(*-ΐ)(Ι^-ΐ) I2 + 1 y Ι2)+···+^0(|χ012 + 1 0 |2)+—+^(|χλ |2 + 1 ¾ |2)] , 上式中 ，
Figure imgf000006_0001
( ..]ί)表示釆样点的功率值的加权系数， χ =-Κ¾:-7人. ».. 表示 X偏振态的釆样值，以及 _y =-K¾:-7人. ».. 表示 Υ偏 振态的釆样值；
其中， 各釆样点的功率值的加权系数与所述釆样点与当前釆样点的距离 成反比。
可选地， 上述方法中， N为整数， 2 N 5。
可选地， 上述方法中， k为整数， l k 10。
可选地， 上述方法中， 各釆样点的功率值的加权系数为大于 0的正数。 本发明实施例还公开了一种相干光通信***中色散和非线性补偿***， 包括：
第一模块， 其设置成： 将光纤总长划分为长度相等的 N个步长； 第二模块， 其设置成： 在每个步长内结合色散补偿算子对所述步长内的 各数据点进行色散补偿； 以及
第三模块， 其设置成： 在每个步长内进行非线性补偿；
其中，所述第三模块是设置成以如下方式在每个步长内进行非线性补偿： 以当前釆样点的位置为中心， 对 X和 Y偏振态各 2k+l个釆样点进行釆 样， 计算所述 2 ( 2k+l )个釆样点的功率， 将所述 X和 Y偏振态上位置相同 的釆样点的功率相加， 得到 2k+l个功率值， 对所述 2k+l个功率值进行加权 求和，再将所述加权求和的值乘以设定系数 W后得到当前釆样点的非线性相 角， 根据所述非线性相角进行补偿；
其中， 所述 N和 k为正整数。
可选地， 上述***中， 所述第三模块是设置成根据如下计算公式计算所 述当前釆样点的非线性相角：
w[^-ki I2 + 1 y Ι2)+^(*-ΐ)(Ι^-ΐ) I2 + 1 y Ι2)+···+^0(|χ012 + 1 0 |2)+—+^(|χλ \2 + \yk |2)] , 上式中 ，
Figure imgf000006_0002
( ..]ί)表示釆样点的功率值的加权系数， χ =-Κ¾:-7人. ».. 表示 X偏振态的釆样值，以及 _y =-K¾:-7人. ».. 表示 Υ偏 振态的釆样值；
其中， 各釆样点的功率值的加权系数与所述釆样点与当前釆样点的距离 成反比。
可选地， 上述***中， N为整数， 2 N 5。
可选地， 上述***中， k为整数， l k 10。
可选地， 上述***中， 各釆样点的功率值的加权系数为大于 0的正数。
本发明实施例的技术方案中， 不需要用色散补偿光纤来补偿色散， 改进 了非线性相角的计算方式， 提高了步长， 从而降低了计算复杂度， 适用于 16QAM格式的偏振复用波分复用***。 附图概述
图 1为偏振复用波分复用相干接收***的原理示意图；
图 2为本发明实施例提供的 16QAM格式的偏振复用波分复用***的色 散和非线性补偿方法的原理示意图；
图 3为不同的 OSNR对色散和非线性补偿效果的影响的示意图； 图 4为信噪比为 25.16dB时经过图 2的补偿方法后的 X和 Y偏振态的星 座图。 本发明的较佳实施方式
下文将结合附图对本发明实施例的技术方案作详细说明。需要说明的是， 在不冲突的情况下， 本申请的实施例和实施例中的特征可以任意相互组合。
实施例 1
本实施例介绍一种相干光通信***中色散和非线性补偿方法， 该方案的 原理如图 2所示。 通过该方法在 16QAM格式的偏振复用波分复用***中釆 用低复杂度色散和非线性补偿算法对色散和非线性进行补偿， 补偿过程说明 ^口下。 首先， 将光纤总长划分为等长的 N个部分（N远小于传统反向补偿算法 的要求） ， 即， N个长度相等的步长， 在每个步长内将色散和非线性效应分 开补偿， 先结合色散补偿算子对步长内的全部数据点进行色散补偿； 然后再 进行非线性补偿。 其中， 进行非线性补偿时， 不再像传统的反向传输补偿法 那样仅仅是利用当前釆样点的功率来计算非线性相角， 而是以当前釆样点的 位置为中心， 对 X和 Y偏振态各 2k+l个釆样点进行釆样， 计算这 2 ( 2k+l ) 个釆样点的功率， 将 X和 Y偏振态上位置相同的釆样点的功率相加， 得到 2k+l个功率值， 然后对这 2k+l个功率值进行加权求和， 将加权求和的值乘 以一个设定系数 W后就得到当前釆样点的非线性相角，根据此非线性相角进 行补偿即可， 其中， N和 k为正整数。
可以按照如下公式来计算当前釆样点的非线性相角：
W [^-ki I2 + 1 y-k Ι2) + ^ (*-ΐ)(Ι ^-ΐ) I2 + 1 y Ι2)+···+ ^0(| χ0 12 + 1 0 |2)+—+ ^(| χλ \2 + \ yk |2)] , 上式中， w z=-K¾:-7人 表示相应位置上的加权系数、
Figure imgf000008_0001
-(1ί-1)... (λ ..1ί) 表示 X偏振态的釆样值， 以及 _y z=- :，-f¾:-7人 表示 Υ偏振态的釆样值。需 要说明的是， 各釆样点的功率值的加权系数为大于 0 的正数， 可根据经验来 确定。 但本实施例的优选方案中， 提出各釆样点的功率值的加权系数与该釆 样点与当前釆样点的距离成反比。 即， 对于距离当前釆样点位置越远的釆样 点， 为其功率值配置的加权系数越小， 而对于距离当前釆样点位置越近的釆 样点， 为其功率值的配置加权系数则越大。
另外， 由图 2可知， 色散和非线性补偿的复杂度与所划分的步长数 N是 成正比的， 传统反向补偿算法的补偿步长取值较小， 总的步长数较多， 因而 计算量很大。 而本实施例的优先方案对 N的取值进行限定， 优先推荐 N为大 于等于 2小于等于 5的整数， 从而大大降低计算复杂度， 减小计算量， 提高 补偿效率。
除了通过对 N的取值进行限制， 以降低计算的复杂度外。 本实施例通过 求功率加权求和的方式提高单步的非线性相角补偿幅度， 也大大减少了总步 数， 减低了复杂度。 且在一些优选方案中， 计算非线性相角的过程中， 还对 以当前釆样点的位置为中心的 X和 Y偏振态各 2k+l个釆样点也进行限制， 即， 釆样点个数不能过多也不能过少， 优先推荐 l k 10 , k为整数。 同时， 在计算非线性相角的过程中， 除了各釆样点的功率值的加权系数 夕卜， 还有一个重要的参数， 即， 设定系数 W (也可称为调整因子）， 该 W的 取值随着应用***的不同而设置为不同的值， 即， 根据不同的***设置（如， 每个偏振态的入纤功率、 传输长度和***信噪比等）设置适当的 W, 只要保 证设置的 W的取值可使***性能达到最优即可，这样可大大增加***的灵活 性和适用范围。
下面参见图 3 ,可以看出 16QAM偏振复用波分复用***不同的光信噪比 ( Optical Signal to Noise Ratio, OSNR )对色散和非线性补偿效果的影响。 图 3中所呈现的激光器线宽为 100kHz,传输长度为 1000km,每个偏振态的入纤 光功率为 ldBm, 符号长度为 8192, 符号率为 180Gbaud。 ***能在一定的信 噪比范围之内将误码率保持在 10— 3的数量级。
图 4所示为***在信噪比为 25.16dB时经过上述图 2所示的色散和非线 性补偿方法后 X 和 Y偏振态的星座图情况。 图中所呈现的激光器线宽为 100kHz, 传输长度为 1000km, 每个偏振态的入纤光功率为 ldBm, 符号长度 为 8192,符号率为 180Gbaud。可以看见经过补偿算法处理后的星座图分散程 度良好， 同时***的误码率也处在 10— 3的数量级。
实施例 2
本实施例介绍一种相干光通信***中色散和非线性补偿***， 该***至 少包括：
第一模块， 其设置成： 将光纤总长划分为长度相等的 N个步长； 第二模块， 其设置成： 在每个步长内结合色散补偿算子对该步长内的各 数据点进行色散补偿； 以及
第三模块， 其设置成： 在每个步长内进行非线性补偿；
其中， 第三模块是设置成以如下方式在每个步长内进行非线性补偿： 以 当前釆样点的位置为中心， 对 X和 Y偏振态各 2k+l个釆样点进行釆样， 计 算该 2 ( 2k+l )个釆样点的功率， 将 X和 Y偏振态上位置相同的釆样点的功 率相加， 得到 2k+l个功率值， 对该 2k+l个功率值进行加权求和， 再将所述 加权求和的值乘以设定系数 W后得到当前釆样点的非线性相角，根据所述非 线性相角进行补偿； 其中， N和 k为正整数。
第三模块可以按照如下公式来计算当前釆样点的非线性相角：
W[^-ki I2 + 1 y-k Ι2)+ ^ (*-ΐ)(Ι ^-ΐ) I2 + 1 y Ι2)+···+ ^0(| χ0 12 + 1 0 |2)+—+ ^(| χλ \2 + \ yk |2)] , 上式中， w z=-K¾:-7人 表示相应位置上的加权系数，
Figure imgf000010_0001
-(1ί-1)... (λ..1ί) 表示 X偏振态的釆样值， 以及 _y z=- :，-f¾:-7人 表示 Υ偏振态的釆样值。需 要说明的是， 各釆样点的功率值的加权系数为大于 0 的正数， 可根据经验来 确定。 但本实施例的优选方案中， 提出各釆样点的功率值的加权系数与该釆 样点与当前釆样点的距离成反比。 即， 对于距离当前釆样点位置越远的釆样 点， 为其功率值配置的加权系数越小， 而对于距离当前釆样点位置越近的釆 样点， 为其功率值的配置加权系数则越大。
另外， 色散和非线性补偿的复杂度与所划分的步长数 N是成正比的， 传 统反向补偿算法的补偿步长取值较小， 总的步长数较多， 因而计算量很大。 而本实施例的优先方案对 N的取值进行限定，优先推荐 N为大于等于 2小于 等于 5的整数， 从而大大降低计算复杂度， 减小计算量， 提高补偿效率。
除了通过对 N的取值进行限制， 以降低计算的复杂度外。 本实施例的优 选方案中， 计算非线性相角的过程中， 还对以当前釆样点的位置为中心的 X 和 Y偏振态各 2k+l个釆样点也进行限制， 即， 釆样点个数不能过多也不能 过少， 优先推荐 l k 10 , k为整数。
另外， 在计算非线性相角的过程中， 还有一个重要的参数， 即， 设定系 数 W (也可称为调整因子）， 该 W的取值可随着应用***的不同而设置为不 同的值， 即， 根据不同的***设置 （如， 每个偏振态的入纤功率、 传输长度 和***信噪比等）设置适当的 W, 只要保证设置的 W的取值可使***性能达 到最优即可， 这样可大大增加***的灵活性和适用范围。
从上述实施例可以看出， 本发明实施例的技术方案中， 无需色散补偿光 纤补偿色散， 特别适用于 16QAM偏振复用波分复用相干光通信***， 避免 了传统反向传输补偿法因对光纤细分时补偿较小而带来的庞大计算量， 通过 取以当前釆样点的位置为中心的 X和 Y偏振态各 2k+l ( k值需要根据具体的 ***进行调节， 本***中 k值取 10 )个釆样点， 计算这 2 ( 2k+l )个釆样点 的功率， 将 X和 Y偏振态上位置相同的釆样点的功率相加， 得到 2k+l个功 率值， 然后求这 2k+l个功率值的加权和， 这个加权和乘以一个调整因子 W 后就得到当前釆样点的非线性相角， 通过计算这样的计算方法大大提高了每 一次循环补偿中的非线性相角补偿幅度， 减少了总的补偿步数和计算量。 调 整因子 W也可以根据不同***进行调节。
本领域普通技术人员可以理解上述方法中的全部或部分步骤可通过程序 来指令相关硬件完成， 所述程序可以存储于计算机可读存储介质中， 如， 只 读存储器、 磁盘或光盘等。 可选地， 上述实施例的全部或部分步骤也可以使 用一个或多个集成电路来实现。 相应地， 上述实施例中的各模块 /单元可以釆 用硬件的形式实现， 也可以釆用软件功能模块的形式实现。 本发明实施例不 限制于任何特定形式的硬件和软件的结合。
以上所述， 仅为本发明的较佳实例而已， 并非用于限定本发明的保护范 围。 凡在本发明的精神和原则之内， 所做的任何修改、 等同替换、 改进等， 均应包含在本发明的保护范围之内。
工业实用性
本发明实施例的技术方案中， 不需要用色散补偿光纤来补偿色散， 改进 了非线性相角的计算方式， 提高了步长， 从而降低了计算复杂度。

Claims

权 利 要 求 书
1、 一种相干光通信***中色散和非线性补偿方法， 包括：
将光纤总长划分为长度相等的 N个步长， 在每个步长内先结合色散补偿 算子对所述步长内的各数据点进行色散补偿， 再进行非线性补偿；
其中， 所述在每个步长内进行非线性补偿的步骤包括：
以当前釆样点的位置为中心， 对 X和 Y偏振态各 2k+l个釆样点进行釆 样， 计算所述 2 ( 2k+l )个釆样点的功率， 将 所述 X和 Y偏振态上位置相 同的釆样点的功率相加， 得到 2k+l个功率值， 对所述 2k+l个功率值进行加 权求和，再将所述加权求和的值乘以设定系数 W后得到所述当前釆样点的非 线性相角， 根据所述非线性相角进行补偿；
其中， 所述 N和 k为正整数。
2、 如权利要求 1所述的方法， 其中， 所述对 2k+l个功率值进行加权求 和，再将所述加权求和的值乘以设定系数 W后得到所述当前釆样点的非线性 相角的步骤包括： 根据如下计算公式计算所述当前釆样点的非线性相角： W* [w_k{\ x—k I2 + 1 y_k \2) + w_(k_ (\ x_(k_i} I2 + 1 y_(k_i} |2) + ... + w。(| x。 |2 + 1 y。 |2) +— + w xj2 + 1 ¼ |2)] , 上式中 ，
Figure imgf000012_0001
-(Ιί-1)... ( ..]ί)表示釆样点的功率值的加权系数， χ =-Κ¾:-7人. ».. 表示 X偏振态的釆样值，以及 _y =-K¾:-7人. ».. 表示 Υ偏 振态的釆样值；
其中， 各釆样点的功率值的加权系数与所述釆样点与当前釆样点的距离 成反比。
3、 如权利要求 1或 2所述的方法， 其中，
N为整数， 2 N 5。
4、 如权利要求 1或 2所述的方法， 其中，
k为整数， l k 10。
5、 如权利要求 2所述的方法， 其中，
各釆样点的功率值的加权系数为大于 0的正数。
6、 一种相干光通信***中色散和非线性补偿***， 包括： 第一模块， 其设置成： 将光纤总长划分为长度相等的 N个步长； 第二模块， 其设置成： 在每个步长内结合色散补偿算子对所述步长内的 各数据点进行色散补偿； 以及
第三模块， 其设置成： 在每个步长内进行非线性补偿；
其中，所述第三模块是设置成以如下方式在每个步长内进行非线性补偿： 以当前釆样点的位置为中心， 对 X和 Y偏振态各 2k+l个釆样点进行釆 样， 计算所述 2 ( 2k+l )个釆样点的功率， 将所述 X和 Y偏振态上位置相同 的釆样点的功率相加， 得到 2k+l个功率值， 对所述 2k+l个功率值进行加权 求和，再将所述加权求和的值乘以设定系数 W后得到当前釆样点的非线性相 角， 根据所述非线性相角进行补偿；
其中， 所述 N和 k为正整数。
7、 如权利要求 6所述的***， 其中， 所述第三模块是设置成根据如下计 算公式计算所述当前釆样点的非线性相角：
W[^-ki I2 + 1 y-k Ι2)+^(*-ΐ)(Ι^-ΐ) I2 + 1 y Ι2)+···+^0(|χ012 + 1 0 |2)+—+^(|χλ \2 + \yk |2)] , 上式中 ，
Figure imgf000013_0001
( ..]ί)表示釆样点的功率值的加权系数， χ =-Κ¾:-7人. ».. 表示 X偏振态的釆样值，以及 _y =-K¾:-7人. ».. 表示 Υ偏 振态的釆样值；
其中， 各釆样点的功率值的加权系数与所述釆样点与当前釆样点的距离 成反比。
8、 如权利要求 6或 7所述的***， 其中，
N为整数， 2 N 5。
9、 如权利要求 6或 7所述的***， 其中，
k为整数， l k 10。
10、 如权利要求 7所述的***， 其中，
各釆样点的功率值的加权系数为大于 0的正数。
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