WO2012106862A1 - 功率因数校正转换器以及功率因数校正转换设备 - Google Patents

功率因数校正转换器以及功率因数校正转换设备 Download PDF

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Publication number
WO2012106862A1
WO2012106862A1 PCT/CN2011/076781 CN2011076781W WO2012106862A1 WO 2012106862 A1 WO2012106862 A1 WO 2012106862A1 CN 2011076781 W CN2011076781 W CN 2011076781W WO 2012106862 A1 WO2012106862 A1 WO 2012106862A1
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WO
WIPO (PCT)
Prior art keywords
factor correction
power factor
input
autotransformer
boost inductor
Prior art date
Application number
PCT/CN2011/076781
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English (en)
French (fr)
Inventor
潘灯海
Original Assignee
华为技术有限公司
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Filing date
Publication date
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to EP11847892.4A priority Critical patent/EP2667499A4/en
Priority to CN201180001243.1A priority patent/CN102301574B/zh
Priority to PCT/CN2011/076781 priority patent/WO2012106862A1/zh
Publication of WO2012106862A1 publication Critical patent/WO2012106862A1/zh
Priority to US13/590,938 priority patent/US8531854B2/en

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0064—Magnetic structures combining different functions, e.g. storage, filtering or transformation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0083—Converters characterised by their input or output configuration
    • H02M1/0085—Partially controlled bridges
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Definitions

  • the invention belongs to the technical field of power rectification, and particularly relates to a power factor correction converter and a power factor correction conversion device.
  • the communication power rectifier module usually adopts a two-stage circuit topology.
  • the main function of the power factor correction (PFC) circuit is to realize power factor correction and provide the DC-DC conversion circuit of the latter stage.
  • a stable DC high voltage input typically 400V
  • the downstream DC-DC converter circuit converts the non-isolated DC high voltage isolation into the 43 ⁇ 58V DC voltage required by the communication equipment.
  • Embodiments of the present invention provide a power factor correction converter and a power factor correction conversion device, which solve the technical problems of low conversion efficiency and low power density in the prior art.
  • the power factor correction converter includes at least two sets of bidirectional switches, an autotransformer, a boost inductor, a bus filter capacitor, at least two front legs, and a rear arm; each set of front ends of the bidirectional switches are a coil of the autotransformer, correspondingly connected, a rear end of each set of the bidirectional switch is connected to one end of the AC input power grid; a center tap of the autotransformer is connected to an output end of the boost inductor The input end of the boost inductor is for connecting to the other end of the AC input power grid, or the center tap of the autotransformer is connected to the other end of the AC input power grid, the boost inductor set Formed in the autotransformer; the front ends of the at least two sets of bidirectional switches are each connected to a middle node of one of the front arms, and the rear ends of the at least two sets of bidirectional switches are connected in the rear arm
  • the front bridge arm includes two diodes connected in series in the same direction, and the middle node is located
  • the power factor correction conversion device includes the above power factor correction converter, an AC input power grid, and a load; one end of the AC input power grid is connected to a rear end of each group of the bidirectional switches, and the other end of the AC input power grid is The input ends of the boost inductors are connected; the two ends of the load are respectively connected to the two ends of the bus filter capacitor.
  • the boost inductor, the autotransformer and the two sets of bidirectional switches constitute a three-state switch unit circuit, including two sets of bidirectional switches simultaneously conducting. At the same time, one group is disconnected, one group is turned on, and one group is disconnected. In order to achieve better technical results, an autotransformer with the same number of turns of two coils can be selected.
  • the above technical solution provided by the present invention has the following advantages: With a three-state switching unit circuit, the boost inductor undergoes two charging and discharging in one switching cycle, thereby reducing the boosting inductance. And the ripple on the bus filter capacitor, the current value of the current flowing through the bidirectional switch transistor is smaller, and the conversion efficiency is improved; and the frequency of the ripple on the boost inductor is twice the switching frequency, so the boost inductor The inductance can be reduced by half, making the circuit smaller and achieving higher power density, thus solving the technical problems of the prior art conversion efficiency and low power density.
  • the multi-state switching unit circuit composed of the boost inductor, the autotransformer, and the plurality of sets of bidirectional switches has four or more operating states. Each of them includes two sets of bidirectional switches that are simultaneously turned on and off at the same time. When some bidirectional switches are turned on and some bidirectional switches are turned off, more kinds of working states appear.
  • the boost inductor experiences the same number of charge and discharge cycles as the number of bidirectional switches, so more sets of bidirectional switches are used to make the ripple on the boost inductor smaller. Therefore, the inductance of the boost inductor can be further reduced, thereby further improving the conversion efficiency and power density.
  • FIG. 1 is a schematic diagram showing the connection of a power factor correction converter according to Embodiment 1 of the present invention
  • FIG. 2 is a schematic diagram of an application mode of a three-state switching unit circuit in a power factor correction converter according to Embodiment 1 of the present invention
  • FIG. 3 is an equivalent diagram of a state 1 of a power factor correction converter according to Embodiment 1 of the present invention when an AC input power grid is in a positive half cycle;
  • FIG. 4a and FIG. 4b are schematic diagrams showing the equivalent state of the power factor correction converter provided in Embodiment 1 of the present invention in the positive half cycle of the AC input power grid;
  • FIG. 5 is an equivalent diagram of a state III of a power factor correction converter according to Embodiment 1 of the present invention when an AC input power grid is in a positive half cycle;
  • FIG. 6 is an equivalent diagram of a state 1 of a power factor correction converter according to Embodiment 1 of the present invention when an AC input power grid is in a negative half cycle;
  • FIG. 7a and 7b are equivalent diagrams showing the state 2 of the power factor correction converter provided in Embodiment 1 of the present invention when the AC input power grid is in a negative half cycle;
  • Embodiment 8 is an equivalent diagram of a state III of the power factor correction converter provided in Embodiment 1 of the present invention when the AC input power grid is in a negative half cycle;
  • 9a to 9f are schematic diagrams showing an embodiment of a bidirectional switch of a power factor correction converter according to Embodiment 1 of the present invention.
  • Figure 10 is a schematic diagram showing the connection of a power factor correction converter according to Embodiment 2 of the present invention.
  • FIG. 11 is a connection of a power factor correction converter according to Embodiment 3 of the present invention. Schematic.
  • Embodiments of the present invention provide a power factor correction converter including at least two sets of bidirectional switches, an autotransformer, a boost inductor, a bus filter capacitor, at least two front legs, and a rear arm; each set of bidirectional switches The front ends are respectively connected to one coil of the autotransformer, and the rear end of each set of bidirectional switches is connected to one end of the AC input power grid; the center tap of the autotransformer is connected to the output end of the boost inductor, and the input of the boost inductor The end is for connecting to the other end of the AC input power grid, or the center tap of the autotransformer is for connecting to the other end of the AC input power grid, the boost inductor is integrated in the autotransformer; the front end of at least two sets of bidirectional switches Each is connected to a middle node of a front leg, and the rear ends of at least two sets of bidirectional switches are connected to a middle node of the rear leg; the front arms each include two diodes in the same direction, and
  • the multi-state switching unit circuit composed of the boosting inductor, the autotransformer and the plurality of sets of bidirectional switches has various working states. It includes two states of simultaneous turn-on and simultaneous disconnection of each set of bidirectional switches, and various working states that occur when some bidirectional switches are turned on and some bidirectional switches are turned off.
  • the boost inductor experiences the same number of charge and discharge cycles as the number of bidirectional switches, thus reducing the ripple on the boost inductor and the bus filter capacitor, and the current flowing through the transistors of the bidirectional switch is effective.
  • the value is smaller, the conversion efficiency is improved; and the frequency of the ripple on the boost inductor is several times the switching frequency, and this multiple is also equal to the number of sets of the bidirectional switch, so the inductance of the boost inductor can be less, so that the circuit Smaller size A higher power density results in a technical problem of lower conversion efficiency and lower power density in the prior art.
  • the power factor correction converter provided by the embodiment of the present invention has two sets of bidirectional switches S1-S2 and S3-S4, because the coil in the autotransformer T1 and the front arm and the bidirectional switch are corresponding to each other. Therefore, in the embodiment of the present invention, the coil of the autotransformer T1 and the front arm are both, and in order to achieve better technical effects, the autotransformer T1 having the same number of turns of the two coils is selected.
  • the power factor correction converter provided in this embodiment further includes a guard arm, and the guard arm includes two guard diodes D7, D8, D8, and D8 in the same direction and a boost inductor L1.
  • the input ends are connected, and the two ends of the guard arm are respectively connected with the two ends of the bus filter capacitor C3.
  • the protective diodes of the protective diodes D7 and D8 are used to prevent lightning surge energy from damaging the main topology.
  • the boosting inductor L1, the autotransformer T1 and the two sets of bidirectional switches S1-S2, S3-S4 form a three-state switching unit circuit, including two sets of bidirectional switches S1- S2, S3-S4 are simultaneously turned on, simultaneously turned off, and one set is turned on and off in three states, and tri-state is realized by conventional pulse width modulation (PWM) control.
  • PWM pulse width modulation
  • 2 is an application mode of the three-state switching unit circuit, wherein D1, D2, Sl, and S2 respectively correspond to D1, D3, Sl, and S3 in FIG. 1, and Tl and T2 represent two of the autotransformer ⁇ 1 in FIG. Coils. The following describes in detail how the three states work.
  • the pulse width modulation (PWM) carrier signals of the bidirectional switches K1, ⁇ 2 are sawtooth waves with a phase difference of 180 degrees, and the duty cycle command of the PWM is given by the PFC control circuit, when the duty ratio of the bidirectional switches K1, ⁇ 2 is greater than At 50%, the bidirectional switches K1 and ⁇ 2 are simultaneously turned on. At this time, the two coils of the autotransformer T1 are short-circuited by the two-way switches K1 and ⁇ 2, respectively, and the AC input grid V charges the boost inductor L1, and the voltage on the load R is supplied by the bus filter capacitor C3.
  • PWM pulse width modulation
  • the bus voltage at the load terminal is 400V
  • the two coils of the self-twisting transformer T1 are connected to the positive and negative terminals of the bus through the conducting diodes D3 and D6, respectively, since the current flowing through the two coils is the boosting inductor L1 current.
  • Half of the voltage, so the voltage on the center tap of the autotransformer T1, that is, the voltage at the output of the boost inductor L1 to the negative terminal of the bus is 200V of the bus voltage.
  • the coil connected to the bidirectional switch K2 in the autotransformer T1 is directly connected to the AC input grid V, and the current on the coil is returned to the AC input grid V;
  • the input grid V and the boost inductor L1 are connected in series through the coil connected to the bidirectional switch K1 on the autotransformer T1, the diode D1 on the front arm and the diode D6 on the rear arm to supply the load R, and charge the bus filter capacitor C3.
  • the boost inductor L1 may be in a state of charge or may be in a discharged state depending on the instantaneous voltage value of the AC input power grid V.
  • the voltage at the center tap of the autotransformer T1, that is, the voltage at the output of the boost inductor L1 to the negative terminal of the bus is 200V of the bus voltage.
  • the bidirectional switches K1 and K2 are simultaneously disconnected.
  • the AC input power grid V and the boost inductor L1 are connected in series through the two coils of the autotransformer T1, the diodes D1 and D3 on the two front arms, and the diode D6 on the rear arm to supply the load R, and
  • the bus filter capacitor C3 is charged, the self-turning transformer is short-circuited, and the boost inductor L1 is in a discharged state.
  • the three-state switching unit circuit In the negative half cycle of the AC input grid V voltage, the three-state switching unit circuit also has the above three working states, the principle is the same as the working state in the positive half cycle, only in The currents in the three-state switching unit circuit are in opposite directions, and the current flows through the diodes on the legs.
  • the bus voltage of the load terminal is 400V
  • the two coils of the self-twisting transformer T 1 are connected to the positive and negative ends of the bus bar through the conducting diodes D5 and D4 respectively, since the current flowing through the two coils is the boosting inductor L 1 half of the current. Therefore, the voltage on the center tap of the autotransformer T 1 , that is, the voltage at the output of the boost inductor L 1 to the negative terminal of the bus is 200V of the bus voltage.
  • the AC input power grid V and the boost inductor L 1 are connected in series
  • the load R is supplied with voltage through the two coils in the autotransformer T l , the diodes D2 and D4 on the front arm and the diode D5 on the rear arm, and the bus filter capacitor C3 is charged, and the self-transformer is short-circuited.
  • the piezoelectric inductor L 1 is in a discharged state.
  • the boost inductor L 1 undergoes two charges and discharges during one PWM switching cycle, thus reducing the ripple on the boost inductor L 1 and the bus filter capacitor C3, then flowing through the two-way
  • the current value of the transistors on the switches S 1 -S2 and S3-S4 is smaller, which improves the conversion efficiency; and the frequency of the ripple on the boost inductor L 1 is twice the PWM switching frequency, so the boost inductor L 1
  • the inductance can be reduced by half, making the circuit smaller and achieving higher power density, thus solving the technical problems of the prior art conversion efficiency and low power density.
  • the power factor correction converter provided by the embodiment of the invention has lower bidirectional switching loss; and the input voltage is closer to a sine wave, so the total harmonic distortion (THD) and the power factor are better.
  • the output of the boost inductor will have more values than the AC input grid V voltage, and the input voltage will be closer to the sine wave, achieving higher specifications.
  • the two front bridge arms are formed by two fast switching diodes connected in series in the same direction. Because the voltage of the front-arm diode is changed from 0 to 400V according to the PWM switching frequency (for example, 50kHz), it is necessary to reverse-recover fast fast switching diode, and the rear-arm diode withstand voltage from 0 to 400V is doubled.
  • the frequency for example, 100 Hz
  • MOS tubes or fast diodes can be used instead of slow rectifier diodes to further improve efficiency.
  • the bidirectional switch includes a diode, a MOS transistor, a transistor, and one of three devices One or more of them may be combined in a combined form, and of course, a combination of other transistors may be used.
  • this embodiment adopts a combination of a diode and a MOS tube to realize a bidirectional switch, or a transistor instead of a MOS transistor, as shown in FIG. 9b.
  • Figures 9c to 9f show the implementation of a bidirectional switch in which the diode, MOS transistor, and transistor are combined in other combinations.
  • a boost inductor is integrated in the autotransformer.
  • the boost inductor can be integrated into the autotransformer to adjust the coupling coefficient M of the autotransformer to obtain the desired boost inductor value, further reducing the size and cost of the converter.
  • This embodiment is basically the same as the first embodiment, and the difference is as follows: as shown in FIG. 11 , in this embodiment, two clamping capacitors C l and C2 for suppressing common mode noise interference are also included, and two clamps are provided. One ends of the bit capacitors C l and C2 are respectively connected to two ends of the AC input network, and the other ends of the two clamp capacitors C l and C2 are connected, and are connected to one end of the bus filter capacitor C3.
  • the two clamp capacitors C l, C2 are capable of suppressing electromagnetic interference, especially common mode noise interference.
  • the embodiment of the invention further provides a power factor correction conversion device, comprising the above power factor correction converter, an AC input power grid and a load; one end of the AC input power grid is connected to the back end of each set of bidirectional switches, and the other end of the AC input power grid is The input ends of the boost inductors are connected; the two ends of the load are respectively connected to the two ends of the bus filter capacitor.
  • the power factor correction conversion device provided by the embodiment of the present invention has two sets of bidirectional switches, because the coils in the autotransformer and the front bridge arm and the bidirectional switch are correspondingly, so in the embodiment of the present invention, the coil of the autotransformer As well as two front axle arms, and for better technical results, two autotransformers with the same number of turns of the two coils are selected.
  • the boost inductor has experienced a switching cycle Two charges and discharges, thus reducing the ripple on the boost inductor and the bus filter capacitor, the current value of the transistor flowing through the bidirectional switch is smaller, improving the conversion efficiency; and the ripple on the boost inductor
  • the frequency is twice the switching frequency, so the inductance of the boost inductor can be reduced by half, making the circuit smaller and achieving higher power density, thus solving the technical problems of lower conversion efficiency and lower power density in the prior art.
  • the power factor correction converter provided by the embodiment of the invention has lower bidirectional switching loss; and the input voltage is closer to a sine wave, so the total harmonic distortion (THD) and the power factor are better.
  • the output of the boost inductor will have more values than the AC input grid voltage, and the input voltage will be closer to the sine wave, achieving higher specifications.
  • the embodiment of the present invention has the same technical features as the above-described carrying device provided by the embodiment of the present invention, the same technical effect can be produced and the same technical problem can be solved.
  • a pulse width modulation (PWM) controller is further included; the bidirectional switch comprises a MOS transistor or a transistor; the PWM controller is connected to the gate of the MOS transistor or to the base of the transistor.
  • the PWM carrier signal is a sawtooth wave with a difference of 180 degrees, and the duty cycle command of the PWM is given by the PFC control circuit to realize various working states of the bidirectional switch.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)
  • Dc-Dc Converters (AREA)

Description

功率因数校正转换器以及功率因数校正转换设备 技术领域
本发明属于电源整流技术领域， 具体涉及一种功率因数校正转 换器以及一种功率因数校正转换设备。
背景技术
通信电源整流器模块通常采用两级电路拓朴构成， 其中前级功 率因数校正 （ Power Factor Correction; PFC ) 电路主要功能是实现 功率因数校正， 同时给后级直流 -直流 （ DC-DC ) 变换电路提供一个 稳定的与电网非隔离的直流高压输入 （一般为 400V ) , 后级 DC-DC 变换电路再将非隔离的直流高压隔离转换成通信设备所需的 43 ~ 58V直流电压。
目前是采用交错并联（ interleaving )技术与双向开关型无桥 PFC 相结合的通信电源整流器模块。 但是交错并联技术中的升压电感两 端的峰值电压为 400V , 电感纹波电流大， 流过开关晶体管电流的有 效值也较大， 并且体积较大， 所以转换效率和功率密度较低。
发明内容
本发明实施例提供了一种功率因数校正转换器以及一种功率因 数校正转换设备， 解决了现有技术的转换效率和功率密度较低的技 术问题。
为达到上述目 的， 本发明的实施例采用如下技术方案：
该功率因数校正转换器， 包括至少两组双向开关、 自耦变压器、 升压电感、 母线滤波电容、 至少两个前桥臂和一个后桥臂； 每组所 述双向开关的前端各与所述自耦变压器的一个线圈——对应相连， 每组所述双向开关的后端用于连接至交流输入电网的一端； 所述自 耦变压器的中心抽头与所述升压电感的输出端相连， 所述升压电感 的输入端用于连接至交流输入电网的另一端， 或者， 所述自耦变压 器的中心抽头用于连接至交流输入电网的另一端， 所述升压电感集 成在所述自耦变压器中； 所述至少两组双向开关的前端各与一个所 述前桥臂的中节点相连， 所述至少两组双向开关的后端连接在所述 后桥臂的中节点上； 所述前桥臂均包括两个同向串联的二极管， 且 所述中节点位于所述两个二极管之间， 所述前桥臂的两端分别与所 述母线滤波电容的两端对应相连； 所述后桥臂包括两个同向串联的 二极管， 且所述中节点位于所述两个二极管之间， 所述后桥臂的两 端分别与所述母线滤波电容的两端对应相连。
该功率因数校正转换设备， 包括上述功率因数校正转换器、 交 流输入电网和负载； 所述交流输入电网的一端与每组所述双向开关 的后端相连， 所述交流输入电网的另一端与所述升压电感的输入端 相连； 所述负载的两端分别与所述母线滤波电容的两端对应相连。
以两组双向开关为例， 本发明实施例提供的功率因数校正转换 器中， 由升压电感、 自耦变压器和两组双向开关构成了三态开关单 元电路， 包括两组双向开关同时导通、 同时断开、 一组导通一组断 开三种工作状态。 为达到更好的技术效果， 可以选取两个线圈的匝 数相等的自耦变压器。
与现有技术相比，本发明所提供的上述技术方案具有如下优点： 采用三态开关单元电路， 在一个开关周期内， 升压电感经历了两次 充电和放电， 因此减小了升压电感和母线滤波电容上的纹波， 则流 经双向开关的晶体管上的电流有效值更小， 提高了转换效率； 并且 升压电感上纹波的频率是开关频率的两倍， 所以升压电感的电感量 可以减少一半， 使电路的体积更小， 获得更高的功率密度， 故而解 决了现有技术的转换效率和功率密度较低的技术问题。
如果采用三组或者更多的双向开关， 则由升压电感、 自耦变压 器和多组双向开关构成的多态开关单元电路， 就有四种或者更多种 工作状态。 其中都包括每组双向开关同时导通和同时断开两种状态， 在有的双向开关导通、 有的双向开关断开时， 会出现更多种工作状 态。 在一个开关周期内， 升压电感经历的充电和放电次数与双向开 关的组数相等， 所以采用更多组双向开关使升压电感上的纹波更小， 所以可进一步减少升压电感的电感量， 从而进一步提高转换效率和 功率密度。
附图说明
为了更清楚地说明本发明实施例或现有技术中的技术方案， 下 面将对实施例或现有技术描述中所需要使用的附图作简单地介绍， 显而易见地， 下面描述中的附图仅仅是本发明的一些实施例， 对于 本领域普通技术人员来讲， 在不付出创造性劳动的前提下， 还可以 根据这些附图获得其他的附图。
图 1 为本发明的实施例 1 所提供的功率因数校正转换器的连接 示意图；
图 2为本发明的实施例 1 所提供的功率因数校正转换器中三态 开关单元电路的一种应用方式示意图；
图 3 为本发明的实施例 1 所提供的功率因数校正转换器在交流 输入电网正半周期时状态一的等效示意图；
图 4a和图 4b为本发明的实施例 1 所提供的功率因数校正转换 器在交流输入电网正半周期时状态二的等效示意图；
图 5为本发明的实施例 1 所提供的功率因数校正转换器在交流 输入电网正半周期时状态三的等效示意图；
图 6为本发明的实施例 1 所提供的功率因数校正转换器在交流 输入电网负半周期时状态一的等效示意图；
图 7a和图 7b为本发明的实施例 1 所提供的功率因数校正转换 器在交流输入电网负半周期时状态二的等效示意图；
图 8为本发明的实施例 1 所提供的功率因数校正转换器在交流 输入电网负半周期时状态三的等效示意图；
图 9a〜图 9f 为本发明的实施例 1所提供的功率因数校正转换器 的双向开关的实施方式示意图；
图 10为本发明的实施例 2所提供的功率因数校正转换器的连接 示意图；
图 1 1为本发明的实施例 3所提供的功率因数校正转换器的连接 示意图。
具体实施方式
下面将结合本发明实施例中的附图， 对本发明实施例中的技术 方案进行清楚、 完整地描述， 显然， 所描述的实施例仅仅是本发明 一部分实施例， 而不是全部的实施例。 基于本发明中的实施例， 本 领域普通技术人员在没有付出创造性劳动的前提下所获得的所有其 他实施例， 都属于本发明保护的范围。
本发明实施例提供了一种功率因数校正转换器， 包括至少两组 双向开关、 自耦变压器、 升压电感、 母线滤波电容、 至少两个前桥 臂和一个后桥臂；， 每组双向开关的前端各与 自耦变压器的一个线圈 对应相连， 每组双向开关的后端用于连接至交流输入电网的一 端； 自耦变压器的中心抽头与升压电感的输出端相连， 升压电感的 输入端用于连接至交流输入电网的另一端， 或者， 自耦变压器的中 心抽头用于连接至交流输入电网的另一端， 升压电感集成在所述自 耦变压器中； 至少两组双向开关的前端各与一个前桥臂的中节点相 连， 至少两组双向开关的后端连接在后桥臂的中节点上； 前桥臂均 包括两个同向串联的二极管， 且中节点位于两个二极管之间， 前桥 臂的两端分别与母线滤波电容的两端对应相连； 后桥臂包括两个同 向串联的二极管， 且中节点位于两个二极管之间， 后桥臂的两端分 别与母线滤波电容的两端对应相连。
本发明实施例提供的功率因数校正转换器中， 由升压电感、 自 耦变压器和多组双向开关构成的多态开关单元电路， 有多种工作状 态。 其中包括每组双向开关同时导通和同时断开两种状态， 以及有 的双向开关导通、 有的双向开关断开时， 出现的多种工作状态。 在 一个开关周期内， 升压电感经历的充电和放电次数与双向开关的组 数相等， 因此减小了升压电感和母线滤波电容上的纹波， 则流经双 向开关的晶体管上的电流有效值更小， 提高了转换效率； 并且升压 电感上纹波的频率是开关频率的数倍， 这一倍数也与双向开关的组 数相等， 所以升压电感的电感量可以更少， 使电路的体积更小， 获 得更高的功率密度， 故而解决了现有技术的转换效率和功率密度较 低的技术问题。
实施例 1:
如图 1 所示， 本发明实施例所提供的功率因数校正转换器具有 两组双向开关 Sl-S2、 S3-S4, 因为 自耦变压器 T1 中的线圈以及前 桥臂与双向开关是——对应的， 所以本发明实施例中， 自耦变压器 T1 的线圈以及前桥臂均为两个， 并且为达到更好的技术效果， 选取 两个线圈的匝数相等的自耦变压器 Tl。
作为一个优选方案， 本实施例提供的功率因数校正转换器还包 括防护桥臂， 防护桥臂包括两个同向串联的防护二极管 D7、 D8, D7、 D8之间的中节点与升压电感 L1 的输入端相连， 防护桥臂的两 端分别与母线滤波电容 C3 的两端对应相连。 防护二极管 D7、 D8构 成的防护桥臂用于防止雷击浪涌能量对主拓朴电路造成损坏。
本发明实施例提供的功率因数校正转换器中， 由升压电感 Ll、 自耦变压器 T1 和两组双向开关 Sl-S2、 S3-S4构成了三态开关单元 电路， 包括两组双向开关 Sl-S2、 S3-S4 同时导通、 同时断开、 一组 导通一组断开三种状态， 并且通过惯用的脉冲宽度调制 （ PWM) 控 制实现三态。 图 2为三态开关单元电路的一种应用方式，其中的 Dl、 D2、 Sl、 S2分别对应图 1 中的 Dl、 D3、 Sl、 S3, Tl、 T2表示图 1 中 自耦变压器 Τ1 的两个线圈。 以下详细说明三种状态的工作原理。
如图 3所示， 两组双向开关 S 1 -S2、 S3-S4可以简化的看 #丈两个 开关 Kl、 Κ2。 在交流输入电网 V电压的正半周期中， 状态一：
双向开关 Kl、 Κ2的脉冲宽度调制 （ PWM) 载波信号为相位差 180度的锯齿波， 并且该 PWM的占空比命令由 PFC控制电路给出， 当双向开关 Kl、 Κ2 的工作占空比大于 50%时， 双向开关 Kl、 Κ2 就存在同时导通的情况。 此时， 自耦变压器 T1 的两个线圈分别被双 向开关 Kl、 Κ2短路， 交流输入电网 V给升压电感 L1充电， 而负载 R上的电压由母线滤波电容 C3提供。
状态二： 如图 4a所示， 当双向开关 Kl 导通、 K2 断开时， 自耦变压器 T1 中与双向开关 K1相连的线圈直接与交流输入电网 V连通， 该线 圈上的电流返回交流输入电网 V； 交流输入电网 V 与升压电感 L1 串联通过自耦变压器 T1上与双向开关 Κ2相连的线圈、 前桥臂上的 二极管 D3和后桥臂上的二极管 D6给负载 R供电， 并给母线滤波电 容 C3充电， 升压电感 L1可能处于充电状态， 也可能处于放电状态， 这取决于交流输入电网 V的瞬时电压值。 此时， 负载端的母线电压 为 400V, 自藕变压器 T1 的两个线圈分别通过导通的二极管 D3 和 D6 与母线的正负端相连， 由于两个线圈流过的电流均为升压电感 L1 电流的一半， 所以 自耦变压器 T1 的中心抽头上的电压， 即升压 电感 L1 的输出端对母线负端的电压为母线电压的一半 200V。
如图 4b 所示， 当双向开关 K2 导通、 K1 断开时， 自耦变压器 T1 中与双向开关 K2相连的线圈直接与交流输入电网 V连通， 该线 圈上的电流返回交流输入电网 V； 交流输入电网 V 与升压电感 L1 串联通过自耦变压器 T1上与双向开关 K1相连的线圈、 前桥臂上的 二极管 D1和后桥臂上的二极管 D6给负载 R供电， 并给母线滤波电 容 C3充电， 升压电感 L1可能处于充电状态， 也可能处于放电状态， 这取决于交流输入电网 V的瞬时电压值。 同样的， 此时自耦变压器 T1 的中心抽头上的电压， 即升压电感 L1 的输出端对母线负端的电 压为母线电压的一半 200V。
状态三：
如图 5所示， 当双向开关 Kl、 K2的工作占空比小于 50%时， 双向开关 Kl、 K2 就存在同时断开的情况。 此时， 交流输入电网 V 与升压电感 L1 串联分别通过自耦变压器 T1 的两个线圈、 两个前桥 臂上的二极管 Dl、 D3和后桥臂上的二极管 D6给负载 R提供电压， 并给母线滤波电容 C3充电， 自藕变压器被短路， 升压电感 L1处于 放电状态。
在交流输入电网 V电压的负半周期中， 三态开关单元电路也有 上述三种工作状态， 其原理与正半周期中的工作状态相同， 只是在 三态开关单元电路中的电流方向相反， 电流在各桥臂上流经的二极 管不同。
如图 6所示， 在交流输入电网 V电压的负半周期中， 状态一： 自耦变压器 T 1 的两个线圈分别被双向开关 Kl 、 K2短路， 交流 输入电网 V给升压电感 L 1 充电， 而负载 R上的电压由母线滤波电 容 C3提供。
状态二：
如图 7a所示， 当双向开关 K1 导通、 K2 断开时， 自耦变压器 T 1 中与双向开关 K1相连的线圈直接与交流输入电网 V连通， 该线 圈上的电流返回交流输入电网 V； 交流输入电网 V 与升压电感 L 1 串联通过自耦变压器 T 1上与双向开关 K2相连的线圈、 前桥臂上的 二极管 D4和后桥臂上的二极管 D5给负载 R供电， 并给母线滤波电 容 C3充电， 升压电感 L 1可能处于充电状态， 也可能处于放电状态， 这取决于交流输入电网 V的瞬时电压值。 此时， 负载端的母线电压 为 400V , 自藕变压器 T 1 的两个线圈分别通过导通的二极管 D5 和 D4 与母线的正负端相连， 由于两个线圈流过的电流均为升压电感 L 1 电流的一半。 所以 自耦变压器 T 1 的中心抽头上的电压， 即升压 电感 L 1 的输出端对母线负端的电压为母线电压的一半 200V。
如图 7b 所示， 当双向开关 K2 导通、 K1 断开时， 自耦变压器 T 1 中与双向开关 K2相连的线圈直接与交流输入电网 V连通， 该线 圈上的电流返回交流输入电网 V； 交流输入电网 V 与升压电感 L 1 串联通过自耦变压器 T 1上与双向开关 K1相连的线圈、 前桥臂上的 二极管 D2和后桥臂上的二极管 D5给负载 R供电， 并给母线滤波电 容 C3充电， 升压电感 L 1可能处于充电状态， 也可能处于放电状态， 这取决于交流输入电网 V的瞬时电压值。 同样的， 此时自耦变压器 T 1 的中心抽头上的电压， 即升压电感 L 1 的输出端对母线负端的电 压为母线电压的一半 200V。
状态三：
如图 8所示， 此时， 交流输入电网 V与升压电感 L 1 串联分别 通过自耦变压器 T l 中的两个线圈、 前桥臂上的二极管 D2、 D4 以及 后桥臂上的二极管 D5给负载 R提供电压， 并给母线滤波电容 C3充 电， 自藕变压器被短路， 升压电感 L 1处于放电状态。
采用三态开关单元电路， 在一个 PWM 开关周期内， 升压电感 L 1 经历了两次充电和放电， 因此减小了升压电感 L 1 和母线滤波电 容 C3 上的纹波， 则流经双向开关 S 1 -S2、 S3-S4 的晶体管上的电流 有效值更小， 提高了转换效率； 并且升压电感 L 1 上纹波的频率是 PWM开关频率的两倍， 所以升压电感 L 1 的电感量可以减少一半， 使电路的体积更小， 获得更高的功率密度， 故而解决了现有技术的 转换效率和功率密度较低的技术问题。
除此之外， 由于自耦变压器 T 1 的两个线圈是对称的， 所以从 两个线圈流出的电流也更加均衡， 克服了交错并联方案中由于器件 参数的差异导致的电流不能均匀分配的问题。 由于采用 了三态开关 单元电路， 升压电感 L 1 的输出端相对于交流输入电网 V共有 5种 电压： 0V , ±200V , ±400V; 而传统交错并联的技术方案中只有三种 电压： 0V , ±400V。 因此本发明实施例提供的功率因数校正转换器 的双向开关损耗更低； 并且输入电压更接近正弦波， 因此总谐波失 真 （ THD )、 功率因数的指标也更好。 如果采用更多组双向开关， 升 压电感的输出端相对于交流输入电网 V电压就会出现更多种数值， 输入电压就更加接近正弦波， 实现更高的技术指标。 采用的双向开 关的越多， 输入电压就越接近正弦波， 有益效果越明显。
本发明实施例中， 两个前桥臂由两个快速开关二极管同向串联 构成。 因为前桥臂二极管所承受的电压从 0到 400V按 PWM开关频 率 （例如 50kHz ) 变化， 因此需要反向恢复快的快速开关二极管， 而后桥臂二极管承受电压从 0到 400V 的是按 2倍电网频率 （例如 100Hz ) 变化， 因此采用成本较低的慢速整流二极管， 就能够满足需 求。 当然也可以采用 MOS管或者快速二极管代替慢速整流二极管， 进一步提高效率。
双向开关包括二极管、 MOS管、 晶体三极管， 三种器件中的一 种或多种， 以组合形式构成， 当然也可以采用其他晶体管的组合形 式构成。 如图 1 和图 9a所示， 本实施例采用了二极管与 MOS管组 合的形式实现双向开关， 或者用晶体三极管代替 MOS 管， 如图 9b 所示。
图 9c〜9f 为二极管、 MOS管、 晶体三极管以其他组合构成双向 开关的实现方式。
实施例 2 :
本实施例与实施例 1基本相同， 其不同点在于： 如图 10所示， 本实施例中， 升压电感集成在所述自耦变压器中。 对于小功率的应 用， 可以将升压电感集成到 自耦变压器中， 通过调整自耦变压器的 耦合系数 M来获取所需的升压电感值， 进一步减小变换器的体积和 成本。
实施例 3 :
本实施例与实施例 1基本相同， 其不同点在于： 如图 1 1所示， 本实施例中，还包括两个用于抑制共模噪声干扰的箝位电容 C l、 C2 , 两个箝位电容 C l、 C2 的一端分别连接至交流输入电网的两端， 两 个箝位电容 C l、 C2的另一端相连， 并连接至母线滤波电容 C3的一 端。 两个箝位电容 C l、 C2 能够抑制电磁干扰， 尤其是共模噪声干 扰。
本发明实施例还提供一种功率因数校正转换设备， 包括上述功 率因数校正转换器、 交流输入电网和负载； 交流输入电网的一端与 每组双向开关的后端相连， 交流输入电网的另一端与升压电感的输 入端相连； 负载的两端分别与母线滤波电容的两端对应相连。
实施例 4 :
本发明实施例所提供的功率因数校正转换设备具有两组双向开 关， 因为 自耦变压器中的线圈以及前桥臂与双向开关是——对应的， 所以本发明实施例中， 自耦变压器的线圈以及前桥臂均为两个， 并 且为达到更好的技术效果， 选取两个线圈的匝数相等的自耦变压器。
采用三态开关单元电路， 在一个开关周期内， 升压电感经历了 两次充电和放电， 因此减小了升压电感和母线滤波电容上的纹波， 则流经双向开关的晶体管上的电流有效值更小， 提高了转换效率； 并且升压电感上纹波的频率是开关频率的两倍， 所以升压电感的电 感量可以减少一半， 使电路的体积更小， 获得更高的功率密度， 故 而解决了现有技术的转换效率和功率密度较低的技术问题。
除此之外， 由于自耦变压器的两个线圈是对称的， 所以从两个 线圈流出的电流也更加均衡， 克服了交错并联方案中由于器件参数 的差异导致的电流不能均匀分配的问题。 由于采用了三态开关单元 电路， 升压电感的输出端相对于交流输入电网共有 5种电压： 0V , 士 200V , ±400V; 而传统交错并联的技术方案中只有三种电压： 0V , ±400V。因此本发明实施例提供的功率因数校正转换器的双向开关损 耗更低； 并且输入电压更接近正弦波， 因此总谐波失真 （ THD )、 功 率因数的指标也更好。 如果采用更多组双向开关， 升压电感的输出 端相对于交流输入电网电压就会出现更多种数值， 输入电压就更加 接近正弦波， 实现更高的技术指标。 采用的双向开关的越多， 输入 电压就越接近正弦波， 有益效果越明显。
由于本发明实施例与上述本发明实施例所提供的搬运装置具有 相同的技术特征， 所以也能产生相同的技术效果， 解决相同的技术 问题。
本发明实施例中， 还包括脉冲宽度调制 （ PWM ) 控制器； 双向 开关包括 MOS管或晶体三极管； PWM控制器与 MOS管的栅极相连 , 或者与晶体三极管的基极相连。 PWM载波信号为相差 180度的锯齿 波， 并且该 PWM的占空比命令由 PFC控制电路给出， 实现双向开 关的多种工作状态。
以上所述， 仅为本发明的具体实施方式， 但本发明的保护范围 并不局限于此， 任何熟悉本技术领域的技术人员在本发明揭露的技 术范围内， 可轻易想到的变化或替换， 都应涵盖在本发明的保护范 围之内。 因此， 本发明的保护范围应以权利要求的保护范围为准。

Claims

权 利 要 求 书
1、 一种功率因数校正转换器， 其特征在于： 包括至少两组双向 开关、 自耦变压器、 升压电感、 母线滤波电容、 至少两个前桥臂和一 个后桥臂；
每组所述双向开关的前端各与所述自耦变压器的一个线圈—— 对应相连， 每组所述双向开关的后端用于连接至交流输入电网的一 端；
所述自耦变压器的中心抽头与所述升压电感的输出端相连，所述 升压电感的输入端用于连接至交流输入电网的另一端， 或者， 所述自 耦变压器的中心抽头用于连接至交流输入电网的另一端， 所述升压电 感集成在所述自耦变压器中；
所述至少两组双向开关的前端各与一个所述前桥臂的中节点相 连， 所述至少两组双向开关的后端连接在所述后桥臂的中节点上； 所述前桥臂均包括两个同向串联的二极管，且所述中节点位于所 述两个二极管之间， 所述前桥臂的两端分别与所述母线滤波电容的两 端对应相连；
所述后桥臂包括两个同向串联的二极管，且所述中节点位于所述 两个二极管之间， 所述后桥臂的两端分别与所述母线滤波电容的两端 对应相连。
2、 根据权利要求 1 所述的功率因数校正转换器， 其特征在于： 所述至少两组双向开关为两组。
3、 根据权利要求 1 所述的功率因数校正转换器， 其特征在于： 所述自耦变压器的每个线圈的匝数相等。
4、 根据权利要求 1 所述的功率因数校正转换器， 其特征在于： 所述至少两个前桥臂均由两个快速二极管同向串联构成。
5、 根据权利要求 1 所述的功率因数校正转换器， 其特征在于： 还包括防护桥臂， 所述防护桥臂包括两个同向串联的防护二极管， 所 述两个防护二极管之间的中节点与所述升压电感的输入端相连， 所述 防护桥臂的两端分别与所述母线滤波电容的两端对应相连。
6、 根据权利要求 1 所述的功率因数校正转换器， 其特征在于： 还包括两个用于抑制共模噪声干扰箝位电容， 所述两个箝位电容的一 端分别用于连接至交流输入电网的两端， 所述两个箝位电容的另一端 相连， 并连接至所述母线滤波电容的一端。
7、 一种功率因数校正转换设备， 其特征在于： 包括权利要求 1 至 6任一项所述的功率因数校正转换器、 交流输入电网和负载；
所述交流输入电网的一端与每组所述双向开关的后端相连，所述 交流输入电网的另一端与所述升压电感的输入端相连；
所述负载的两端分别与所述母线滤波电容的两端对应相连。
8、 根据权利要求 7所述的功率因数校正转换设备， 其特征在于： 还包括脉冲宽度调制控制器；
所述双向开关包括 MOS管或晶体三极管；
所述脉冲宽度调制控制器与所述 MOS管的栅极相连， 或者与所 述晶体三极管的基极相连。
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