WO2012101905A1 - Switching power supply device - Google Patents

Switching power supply device Download PDF

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Publication number
WO2012101905A1
WO2012101905A1 PCT/JP2011/078244 JP2011078244W WO2012101905A1 WO 2012101905 A1 WO2012101905 A1 WO 2012101905A1 JP 2011078244 W JP2011078244 W JP 2011078244W WO 2012101905 A1 WO2012101905 A1 WO 2012101905A1
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WO
WIPO (PCT)
Prior art keywords
circuit
switching
voltage
power supply
supply device
Prior art date
Application number
PCT/JP2011/078244
Other languages
French (fr)
Japanese (ja)
Inventor
細谷達也
Original Assignee
株式会社村田製作所
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Filing date
Publication date
Application filed by 株式会社村田製作所 filed Critical 株式会社村田製作所
Priority to EP11857040.7A priority Critical patent/EP2670037B1/en
Priority to JP2012554631A priority patent/JP5321758B2/en
Priority to CN201180066145.6A priority patent/CN103339843B/en
Priority to KR1020137019765A priority patent/KR101439495B1/en
Publication of WO2012101905A1 publication Critical patent/WO2012101905A1/en
Priority to US13/941,753 priority patent/US9106141B2/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/338—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement
    • H02M3/3381—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement using a single commutation path
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K3/00—Circuits for generating electric pulses; Monostable, bistable or multistable circuits
    • H03K3/02—Generators characterised by the type of circuit or by the means used for producing pulses
    • H03K3/26—Generators characterised by the type of circuit or by the means used for producing pulses by the use, as active elements, of bipolar transistors with internal or external positive feedback
    • H03K3/30—Generators characterised by the type of circuit or by the means used for producing pulses by the use, as active elements, of bipolar transistors with internal or external positive feedback using a transformer for feedback, e.g. blocking oscillator
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to a switching power supply device that includes a switching element on the primary side and a rectifier circuit on the secondary side, and transmits power using an electromagnetic resonance phenomenon.
  • a current resonance half bridge converter that operates by flowing a sinusoidal resonance current through a transformer using the LC resonance phenomenon is highly efficient in a market such as a flat-screen TV in which the characteristics of output current ripple are relatively relaxed. Utilizing the features, practical application is progressing.
  • Patent Document 1 is disclosed as an LC series resonance type DC-DC converter.
  • FIG. 1 is a basic circuit diagram of a switching power supply device disclosed in Patent Document 1.
  • This switching power supply device is a current resonance type half-bridge DC / DC converter, and an LC resonance circuit composed of an inductor Lr and a capacitor Cr and two switching elements Q1 and Q2 are connected to a primary winding np of a transformer T. ing.
  • a rectifying and smoothing circuit including diodes Ds1 and Ds2 and a capacitor Co is formed in the secondary windings ns1 and ns2 of the transformer T.
  • the switching elements Q1 and Q2 are alternately turned on and off with a dead time therebetween, and the current waveform flowing through the transformer T becomes a sinusoidal resonance waveform.
  • power is transmitted from the primary side to the secondary side in both the on-period / off-period of the two switching elements Q1, Q2.
  • the transformer is used as an insulating transformer using electromagnetic induction, and is only used as a transformer using magnetic coupling.
  • transformers that use electromagnetic induction it is important to efficiently convert from electricity to magnetism and electricity by linking the magnetic flux generated by the current flowing in the primary winding with the secondary winding to flow the current. ing.
  • the ratio of the magnetic flux interlinked with the secondary winding out of the magnetic flux generated by the current flowing through the primary winding is called (magnetic) coupling degree.
  • the power conversion efficiency is increased.
  • the switching power supply device of the present invention is configured as follows. (1) an electromagnetic field coupling circuit including a primary winding and a secondary winding; A primary side AC voltage that includes a switching circuit connected to the primary winding and that is configured by a parallel connection circuit of a switching element, a diode, and a capacitor, and generates an AC voltage from an input DC voltage Generating circuit; A secondary side rectifier circuit for rectifying the AC voltage into a DC voltage; A first resonant circuit configured on the primary side and including a first series resonant inductor and a first series resonant capacitor; A second resonant circuit configured on the secondary side and including a second series resonant inductor and a second series resonant capacitor; A switching control circuit for generating a substantially square wave or trapezoidal AC voltage by alternately turning on / off switching elements of the primary side AC voltage generation circuit with a dead time therebetween; In a switching power supply device comprising: In the switching control circuit, a current flowing into a multi-re
  • the switching operation of the switching element of the primary side AC voltage generating circuit is performed at a switching frequency higher than a specific resonance frequency at which the impedance is smallest with respect to the double resonance circuit
  • the electromagnetic field coupling circuit constitutes an electromagnetic field resonance circuit in which a magnetic field coupling via a mutual inductance and an electric field coupling via a mutual capacitance are mixed between the primary winding and the secondary winding,
  • the first resonance circuit and the second resonance circuit resonate to transmit power from the primary side to the secondary side of the electromagnetic field coupling circuit.
  • the switching control circuit has a switching frequency of the primary side AC voltage generating circuit constant, and a plurality of switching operations with a period in which current is conducted to the switching circuit being an on period and another period being an off period.
  • the output power obtained from the secondary side rectifier circuit is adjusted by controlling the on-period ratio of the circuit.
  • the switching control circuit makes the switching frequency of the primary side AC voltage generation circuit constant, and controls the on-period ratio of the switching element, whereby the output power obtained from the secondary side rectifier circuit Adjust.
  • the secondary side rectifier circuit stores the voltage generated in the secondary winding as electrostatic energy in the second resonant capacitor in either the on period or the off period, or both periods, It is preferable to add the voltage generated in the secondary winding during each of the on period and the off period and output the result as a DC voltage.
  • a parallel resonant capacitor is provided in parallel with the primary winding or the secondary winding.
  • the parallel resonant capacitor is composed of a stray capacitance of the primary winding or the secondary winding.
  • the mutual capacitance is composed of a stray capacitance formed between the primary winding and the secondary winding.
  • the first series resonant inductor or the second series resonant inductor is configured by a leakage inductance of the electromagnetic field coupling circuit.
  • the mutual inductance is constituted by an exciting inductance formed equivalently between the primary winding and the secondary winding.
  • the switching circuit is preferably a MOSFET.
  • the rectifying element that rectifies the AC voltage provided in the secondary side rectifier circuit into a DC voltage is a MOSFET.
  • the secondary rectifier circuit acts as the primary AC voltage generator, and the primary AC voltage generator is It is preferable that it acts as a secondary side rectifier circuit and can transmit power in both directions.
  • the primary winding is a winding provided on the primary side of a transformer having a magnetic core such as ferrite
  • the secondary winding is a winding provided on the secondary side of the transformer.
  • the primary winding is a power transmission coil provided in a power transmission device
  • the secondary winding is a power reception coil provided in a power reception device arranged close to the power transmission device.
  • the LC resonance circuit is provided on both the primary side and the secondary side, and the two LC resonance circuits are resonated to couple the magnetic field and the electric field between the primary winding and the secondary winding. Can be used to transmit power.
  • the resonance phenomenon only active power is transmitted from the primary side to the secondary side, and reactive power is circulated in both the primary and secondary LC resonance circuits, so power loss is extremely high. Small.
  • switching frequency switching is performed at a frequency higher than the natural resonance frequency at which the input impedance becomes the smallest in the entire double resonance circuit including the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit.
  • a zero voltage switching (ZVS) operation in the switching element can be performed by turning on and off the element.
  • FIG. 1 is a basic circuit diagram of a switching power supply device disclosed in Patent Document 1.
  • FIG. 2 is a circuit diagram of the switching power supply apparatus 101 according to the first embodiment.
  • FIG. 3 is a voltage-current waveform diagram of each part of the switching power supply apparatus 101 shown in FIG.
  • FIG. 4 is a circuit diagram of the switching power supply apparatus 102 according to the second embodiment.
  • FIG. 5A is a waveform diagram of voltage / current of each part of the switching power supply apparatus 102.
  • FIG. 5B is a waveform diagram of voltage and current in each part of the switching power supply device shown in FIG.
  • FIG. 6 is a circuit diagram of the switching power supply apparatus 103 according to the third embodiment.
  • FIG. 7 is a circuit diagram of the switching power supply device 104 of the fourth embodiment.
  • FIG. 8 is a circuit diagram of the switching power supply device 105 of the fifth embodiment.
  • FIG. 9 shows the waveform of the voltage input to the series resonant capacitor Cr shown in FIG.
  • FIG. 10 shows the output voltage Vo with respect to the on-time ratio D, which is the ratio of the conduction period of the switching circuit S1 to the switching cycle, and the on-period ratio Da, which is the ratio of the conduction period of the switching circuit S2 to the conduction period of the switching circuit S1.
  • D the on-time ratio
  • Da the on-period ratio
  • FIG. 11 is a circuit diagram of the switching power supply device 106 of the sixth embodiment.
  • FIG. 11 is a circuit diagram of the switching power supply device 106 of the sixth embodiment.
  • FIG. 12 is a circuit diagram of the switching power supply device 107 of the seventh embodiment.
  • FIG. 13 is a circuit diagram of the switching power supply device 108 of the eighth embodiment.
  • FIG. 14 is a circuit diagram of the switching power supply device 109 of the ninth embodiment.
  • FIG. 15 is a circuit diagram of the switching power supply device 110 of the tenth embodiment.
  • FIG. 16 is a circuit diagram of a switching power supply device 111 used as the power transmission system of the eleventh embodiment.
  • FIG. 17 is a circuit diagram of the switching power supply device 112 used as the power transmission system of the twelfth embodiment.
  • FIG. 18 is a circuit diagram of the switching power supply device 113 used as the power transmission system of the thirteenth embodiment.
  • FIG. 19 is a circuit diagram of the switching power supply device 114 used as the power transmission system of the fourteenth embodiment.
  • FIG. 20 is a circuit diagram of the switching power supply device 115 used as the power transmission system of the fifteenth embodiment.
  • FIG. 21 is a circuit diagram of the switching power supply 116 used as the power transmission system of the sixteenth embodiment.
  • FIG. 22 is a circuit diagram of a switching power supply device 117 used as the power transmission system of the seventeenth embodiment.
  • FIG. 23 is a circuit diagram of a switching power supply device 118 used as the power transmission system of the eighteenth embodiment.
  • FIG. 2 is a circuit diagram of the switching power supply apparatus 101 according to the first embodiment.
  • the switching power supply device 101 is a circuit in which an input power supply Vi is input to an input unit and supplies stable DC power from an output unit to a load Ro.
  • the switching power supply apparatus 101 includes the following units.
  • An electromagnetic field coupling circuit 90 using a transformer having a primary winding np and a secondary winding ns A switching circuit S1 including a switching element Q1 and a switching circuit S2 including a switching element Q2 connected to the primary winding np -Rectifier diodes D3 and D4 connected to the secondary winding ns and the smoothing capacitor Co
  • a double resonance circuit 40 including the electromagnetic coupling circuit 90 and including a first LC series resonance circuit and a second LC series resonance circuit.
  • the electromagnetic field coupling circuit constitutes an electromagnetic field coupling circuit (electromagnetic resonance circuit) in which magnetic field coupling and electric field coupling are combined. Both the series resonant capacitors Cr and Crs also serve as capacitors for holding a DC voltage.
  • the capacitor Cr On the primary side, the capacitor Cr is charged during the conduction period of the switching element Q1, and the capacitor Cr is discharged during the conduction period of the switching element Q2.
  • the capacitor Crs On the other hand, on the secondary side, the capacitor Crs is discharged during the conduction period of the switching element Q1, and the capacitor Crs is charged using the voltage generated in the secondary winding ns during the conduction period of the switching element Q2 as electrostatic energy.
  • Q2 add the voltages of the secondary windings ns generated during each conduction period and output the result.
  • the circuit including the rectifier diodes D3 and D4 and the capacitor Crs constitutes an addition rectifier circuit 80 that performs charge / discharge and rectification.
  • the primary-side inductor Lm may be an inductor as a component, or may represent the excitation inductance of the primary winding np of the transformer T.
  • the primary-secondary capacitance Cm may be a capacitance as a component or may represent a mutual capacitance which is a stray capacitance of the transformer T.
  • a portion surrounded by a thick broken line constitutes an electromagnetic field coupling circuit 90
  • a portion surrounded by a thin broken line constitutes a multiple resonance circuit 40.
  • the multi-resonance circuit 40 including the electromagnetic field coupling circuit 90 resonates with two LC resonance circuits on the primary side and the secondary side.
  • the specific action is as follows.
  • (1) The first resonance circuit made of Lr—Cr and the second resonance circuit made of Lrs—Crs resonate to resonate with each other, and between the primary winding np and the secondary winding ns.
  • power transmission is performed using two couplings of a magnetic field due to mutual inductance and an electric field due to mutual capacitance.
  • the exciting inductance of the transformer T is used as a mutual inductance (Lm), and illustration as a circuit element is omitted.
  • the capacitors Cp and Cs promote power transmission through electromagnetic coupling. That is, the capacitors Cp and Cs and the mutual capacitance Cm constitute a power transmission circuit by ⁇ -type electric field coupling to transmit power. Incidentally, the mutual capacitance Cm constitutes a power transmission circuit by electric field coupling with the resonance capacitors Cr and Crs.
  • the capacitors Cp and Cs are arranged such that, in the commutation period when the switching element is turned off, on the primary side, the resonance current ir flowing in the resonance capacitor Cr is parallel to the switching circuit parallel capacitor (switching elements Q1 and Q2).
  • the current flows to the capacitor (connected capacitor) and the capacitor Cp.
  • the larger the resonance current ir the larger the current flowing through the capacitor Cp.
  • the current flowing through the parallel capacitance of the switching circuit during the commutation period is substantially constant.
  • the dead time period can be reduced with respect to fluctuations in output power.
  • the difference between the period when the current path is switched between the diode D3 and the diode D4 can be corrected.
  • the switching elements Q1 and Q2 are alternately turned on and off with a dead time therebetween, thereby shaping the DC voltage Vi into a square wave or trapezoidal voltage waveform.
  • the rectifier diodes D3 and D4 are alternately turned on to shape a square wave or trapezoidal voltage waveform into a DC voltage.
  • the double resonance circuit 40 includes two resonance circuits including a primary side and a secondary side including the electromagnetic field coupling circuit 90.
  • the double resonance circuit 40 has a specific resonance frequency fr that minimizes the combined impedance of the double resonance circuit 40, and the switching frequency fs and the resonance frequency fr approach each other to resonate.
  • the flowing current increases and the output power increases. That is, the switching element is turned on / off at a switching frequency fs higher than the inherent resonance frequency fr of the entire multiple resonance circuit 90 that is a combination of the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit. As the switching frequency fs approaches the inherent resonance frequency fr and resonates, the current flowing into the multiple resonance circuit increases and the output power increases.
  • the primary-side and secondary-side capacitors Cr and Crs perform two actions, that is, an operation for holding a DC voltage and a resonance operation.
  • the transformer 2 includes a parasitic inductance of the transformer T such as the exciting inductance Lm of the primary winding np of the transformer T, the exciting inductance Lms of the secondary winding ns, the series resonant inductors Lr and Lrs, and the capacitors Cp and Cs. You may be comprised with the component.
  • the transformer can be referred to as a resonant composite transformer that integrates a function as a transformer that enables electrical insulation and electrical parameters such as a resonant inductor and a resonant capacitor, and is used as an electromagnetic coupling device. be able to.
  • FIG. 3 is a voltage-current waveform diagram of each part of the switching power supply apparatus 101 shown in FIG. The operation of the switching power supply apparatus 101 at each timing is as follows.
  • the exciting inductance of the primary winding np of the transformer T is represented by Lm, and the exciting current is represented by im.
  • the gate-source voltages of the switching elements Q1, Q2, Q3, and Q4 are represented by vgs1 and vgs2, the drain-source voltages are represented by vds1 and vds2, respectively, and the drain current of Q1 is represented by id1.
  • Q1 and Q2 are alternately turned on and off with a short dead time when both switching elements are turned off, and the current flowing in Q1 and Q2 is commutated in the dead time period, respectively, so that zero voltage switching (ZVS) operation is performed. I do.
  • ZVS zero voltage switching
  • the switching control circuit 10 performs the following control.
  • the switching frequency is set higher than the natural resonance frequency fr at which the input impedance is minimized for the entire multiple resonance circuit including the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit. .
  • ZVS zero voltage switching
  • the on-period ratio of the switching elements Q1 and Q2 is set to 1, and the switching elements Q1 and Q2 are controlled so as to change the switching frequency of the primary side AC voltage generation circuit, thereby secondary side rectification. Adjust the output power available from the circuit.
  • the output obtained from the secondary side rectifier circuit is controlled by combining the on period ratio control of (2) and the switching frequency control of (3) so as to obtain an optimum control characteristic. Adjust the power.
  • an electromagnetic field coupling circuit (electromagnetic resonance circuit) that combines magnetic field coupling and electric field coupling using resonance on the primary side and the secondary side, power transmission is performed only by magnetic field coupling.
  • the power transmission efficiency becomes high and high-efficiency operation becomes possible.
  • the switching frequency is set to the natural resonance frequency fr where the input impedance is minimized.
  • a converter By configuring an electromagnetic field coupling circuit using leakage inductance, excitation inductance, stray capacitance, mutual capacitance, etc. of the transformer, a converter can be configured with a small number of parts, and a reduction in size and weight can be achieved. .
  • the primary-side and secondary-side capacitors Cr and Crs perform the two roles of holding the DC voltage and the resonance operation, thereby converting the DC voltage into the AC voltage, while the double resonance circuit is Since the resonance operation is performed as the resonance capacitance to be configured, the number of components can be reduced. Further, ON period ratio control (PWM control) at a constant switching frequency is possible.
  • FIG. 4 is a circuit diagram of the switching power supply apparatus 102 according to the second embodiment.
  • switching elements Q3 and Q4 using FETs are provided instead of the rectifier diodes D3 and D4 on the secondary side. That is, the secondary side rectifier circuit is constituted by the switching elements Q3 and Q4.
  • the switching elements Q3 and Q4 each include a diode (parasitic diode) and a capacitor (parasitic capacitance) in parallel, and constitute switching circuits S3 and S4.
  • a capacitor Ci is provided in the power input section.
  • the switching control circuit 20 controls the secondary side switching elements Q3 and Q4.
  • the excitation inductances of the primary winding np and the secondary winding ns of the transformer T are Lm and Lms, the parasitic capacitances of the switching elements Q1, Q2, Q3, and Q4 and the parasitic diode are not shown.
  • the secondary side switching control circuit 20 turns on / off the switching element Q3 in synchronization with the primary side switching element Q1, and turns on / off the switching element Q4 in synchronization with the primary side switching element Q2. That is, synchronous rectification is performed.
  • the overall operation of the switching power supply 102 is the same as that of the switching power supply 101 shown in the first embodiment.
  • FIG. 5A is a waveform diagram of voltage and current of each part of the switching power supply device 102.
  • FIG. FIG. 5B is a waveform diagram of voltage and current in each part of the switching power supply device shown in FIG.
  • vds1 is the drain-source voltage of the switching element Q1
  • ir is the current flowing through the capacitor Cr
  • Vds3 is the drain-source voltage of the switching element Q3
  • id3 is the current flowing through the switching element Q3
  • id4 is flowing through the switching element Q4. Current.
  • a diode element having a low withstand voltage has a small forward voltage drop and a FET having a low withstand voltage. Since the on-resistance is small, the conduction loss due to the flowing current can be reduced, and high-efficiency operation is possible.
  • the switching power supply 102 of the second embodiment has a symmetrical topology between input and output. Therefore, when power is transmitted from the output part of the secondary side rectifier circuit, the secondary side rectifier circuit acts as a primary side AC voltage generation circuit, and the primary side AC voltage generation circuit by the switching elements Q1 and Q2 is 2 Acts as a secondary rectifier circuit. Therefore, power transmission is possible in both directions from the primary side to the secondary side of the transformer T or from the secondary side to the primary side.
  • the load Ro is a rechargeable battery, a storage capacitance, or a circuit including its charge / discharge control circuit
  • power is transmitted from the primary side to the secondary side of the transformer T, whereby the rechargeable battery Is charged.
  • a load circuit is connected to the portion to which the input power source Vi is connected in FIG. 4, the primary side from the secondary side of the transformer T with the rechargeable battery or the storage capacitance as the input power source and the direction of power transmission reversed. Power transmission to the side becomes possible.
  • FIG. 6 is a circuit diagram of the switching power supply apparatus 103 according to the third embodiment.
  • capacitors Ci1 and Ci2 that divide the voltage of the input power source Vi and capacitors Cis1 and Cis2 that divide the output voltage Vo are provided.
  • the primary winding np of the transformer T the exciting inductance of the secondary winding ns, or inductors Lm and Lms which are external inductances are illustrated. Others are the same as those of the second embodiment shown in FIG.
  • the input voltage Vi is divided by the capacitors Ci1 and Ci2, and the output voltage Vo is divided by the capacitors Cis1 and Cis2. Since the capacitors Ci1 and Cis1 divide the DC input voltage and perform the function of holding the DC voltage, the series resonance capacitors Cr and Crs act as resonance capacitors and hold the DC voltage, that is, DC voltage components. It does not function as a function of resonance operation by biasing.
  • the overall converter operation is as shown in the first embodiment.
  • the input power source Vi is divided as the respective voltages of the capacitors Ci1 and Ci2, and current flows from the input power source Vi to the capacitors Ci1 and Ci2 in both ON / OFF cycles of the switching elements Q1 and Q2, and the input power source Vi The effective value of the input current flowing out of the current becomes small, and the conduction loss in the current path is reduced.
  • the output voltage Vo is divided as the respective voltages of the capacitors Cis1 and Cis2, and the output voltage from the capacitors Cis1 and Cis2 in both the on / off cycles of the switching elements Q1 and Q2.
  • the effective value of the current flowing to Vo becomes small, and the conduction loss is reduced.
  • FIG. 7 is a circuit diagram of the switching power supply device 104 of the fourth embodiment.
  • capacitors Cr1 and Cr2 for dividing the voltage of the input power source Vi and capacitors Crs1 and Crs2 for dividing the output voltage Vo are provided. That is, the series resonance capacitor Cr in the switching power supply device shown in the second embodiment is divided into Cr1 and Cr2, and the series resonance capacitor Crs is divided into Crs1 and Crs2.
  • an equivalent mutual inductance Lm formed between the primary winding np and the secondary winding ns of the transformer T is illustrated, and the transformer T composed of the primary winding np and the secondary winding ns is , Illustrated as an ideal transformer.
  • the inductor Lr, the inductor Lrs, and the capacitors Cp and Cs can be configured by a single circuit element. It is also possible to configure the electromagnetic coupling circuit 90 itself by a single resonance composite transformer using parasitic elements of the transformer T. Others are the same as those of the second embodiment shown in FIG.
  • the loss due to the capacitor is dispersed, the overall loss is reduced, and the heat generation is dispersed.
  • the capacitors Cr1 and Cr2 and the capacitors Crs1 and Crs2 play both roles of holding a DC voltage and acting as a series resonance capacitor.
  • FIG. 8 is a circuit diagram of the switching power supply device 105 of the fifth embodiment.
  • a capacitor Cc is provided on the primary side to constitute a voltage clamp circuit.
  • Others are the same as those of the second embodiment shown in FIG.
  • the switching power supply device shown in FIG. 8 After the switching element Q1 is turned off, the voltage of the primary winding np is charged to the capacitor Cc through the parasitic diode of the switching element Q2 in the direction shown in FIG. When Q2 is on, the voltage (+ Vc) charged in the capacitor Cc is applied to the multiple resonance circuit. That is, the input voltage Vi is converted into a square wave voltage, and the square wave voltage has voltage amplitudes of + Vi and ⁇ Vc.
  • FIG. 9 is a waveform of a voltage applied to the multiple resonance circuit including the series resonance capacitor Cr, the electromagnetic field coupling circuit 90, and the series resonance capacitor Crs shown in FIG.
  • the solid line is the waveform in the case of the fifth embodiment
  • the broken line is the waveform in the case of the first to fourth embodiments.
  • the input power supply voltage to the resonance circuit changes between + Vi and 0 V
  • the voltage amplitude is Vi in the fifth embodiment, compared to + Vi.
  • the voltage Vc across the capacitor Cc constituting the voltage clamp circuit changes according to the ON period ratio D which is the ratio of the conduction period of the switching element Q1 to the switching period, and the output voltage Vo can be controlled over a wide range.
  • the present invention is excellent in application when the input power supply voltage varies over a wide range when the output voltage is constant.
  • control characteristics with respect to fluctuations in the input voltage are improved. That is, the output voltage can be stabilized even if the input voltage varies greatly.
  • FIG. 10 shows the output voltage Vo with respect to the on-time ratio D, which is the ratio of the conduction period of the switching circuit S1 to the switching cycle, and the on-period ratio Da, which is the ratio of the conduction period of the switching circuit S2 to the conduction period of the switching circuit S1.
  • the solid line is the characteristic curve of the on-period ratio Da
  • the broken line is the characteristic curve of the on-time ratio D.
  • FIG. 11 is a circuit diagram of the switching power supply device 106 of the sixth embodiment.
  • a capacitor Cc is provided on the primary side to constitute a voltage clamp circuit.
  • capacitors Ci1 and Ci2 that divide the voltage of the input power source Vi and capacitors Cis1 and Cis2 that divide the output voltage Vo are provided.
  • the exciting inductance of the primary winding np is shown as a circuit parameter.
  • an equivalent mutual inductance Lm formed between the primary winding np and the secondary winding ns of the transformer T is illustrated, and the transformer T composed of the primary winding np and the secondary winding ns is: It is shown as an ideal transformer.
  • the inductor Lr, the inductor Lrs, and the capacitors Cp and Cs can be configured by a single circuit element. It is also possible to configure the electromagnetic coupling circuit 90 itself by a single resonance composite transformer using parasitic elements of the transformer T. Others are the same as those of the second embodiment shown in FIG.
  • the control characteristic with respect to the fluctuation of the input voltage is improved.
  • the input power source Vi is divided by the capacitors Ci1 and Ci2
  • FIG. 12 is a circuit diagram of the switching power supply device 107 of the seventh embodiment.
  • a capacitor Cc is provided on the primary side to constitute a voltage clamp circuit on the primary side
  • a capacitor Ccs is provided on the secondary side to constitute a voltage clamp circuit on the secondary side.
  • Others are the same as those of the fifth embodiment shown in FIG.
  • the input voltage Vi is converted into a square wave voltage, and the square wave voltage has voltage amplitudes of + Vi and ⁇ Vc. Further, since the secondary capacitor Ccs is charged with a negative voltage (Vcs), the AC square wave voltage applied to the synchronous rectifier circuit by the switching elements Q3 and Q4 has a voltage amplitude of + Vo and ⁇ Vcs. Since the voltage amplitude is thus increased, the control characteristics with respect to fluctuations in the output voltage are also improved. That is, the output voltage can be easily adjusted over a wide range.
  • FIG. 13 is a circuit diagram of the switching power supply device 108 of the eighth embodiment.
  • a primary AC voltage generating circuit having a full bridge circuit configuration by four switching elements Q1, Q2, Q5, and Q6 is provided.
  • a secondary side rectifier circuit having a bridge rectification configuration by four switching elements Q3, Q4, Q7, and Q8 is provided.
  • the primary side switching elements Q1, Q2, Q5, Q6 and the secondary side switching elements Q3, Q4, Q7, Q8 Since the voltage applied to each becomes half, the loss in the switching element can be reduced.
  • FIG. 14 is a circuit diagram of the switching power supply device 109 of the ninth embodiment.
  • the primary-side resonant capacitor is divided into two capacitors Cr1 and Cr2
  • the secondary-side resonant capacitor is divided into two capacitors Crs1 and Crs2.
  • a primary side AC voltage generating circuit having a full bridge circuit configuration by four switching elements Q1, Q2, Q5, and Q6 is provided.
  • a secondary side rectifier circuit having a bridge rectification configuration by four switching elements Q3, Q4, Q7, and Q8 is provided.
  • the voltage applied to each of the resonant capacitors Cr and Crs shown in the first to third embodiments and the like is divided and applied to two capacitors. Loss can be distributed. Further, since the voltages applied to the primary side switching elements Q1, Q2, Q5, and Q6 and the secondary side switching elements Q3, Q4, Q7, and Q8 are each halved, the loss in the switching elements can be reduced. .
  • the capacitors Cr1 and Cr2 and the capacitors Crs1 and Crs2 play both roles of holding a DC voltage and acting as a series resonance capacitor.
  • FIG. 15 is a circuit diagram of the switching power supply device 110 of the tenth embodiment.
  • Lp is a power transmission coil on the power transmission device side
  • Ls is a power reception coil on the power reception device side.
  • the equivalent leakage inductor of the power transmission coil Lp is the inductor Lr shown in the first to ninth embodiments
  • the parallel capacitor between the equivalent windings of the power transmission coil Lp is the first to ninth embodiments.
  • the inductor Lrs shown in the first to ninth embodiments is an equivalent leakage inductor of the power receiving coil Ls
  • the parallel capacitor between the equivalent windings of the power receiving coil Ls is shown in the first to ninth embodiments.
  • the capacitor Cs is configured.
  • an equivalent inductance involved in magnetic field coupling in the power transmission coil Lp is configured as a mutual inductance Lm
  • an equivalent capacitance involved in electric field coupling between the power transmission coil Lp and the reception coil Ls is configured as a mutual capacitance Cm.
  • the mutual coefficient M as the electromagnetic field coupling is configured by combining the magnetic field coupling by the mutual inductance (mutual coefficient Ml) and the electric field coupling by the mutual capacitance (mutual coefficient Mc).
  • the switching power supply 110 used as the power transmission system has a symmetrical topology between input and output, as shown in FIG. Therefore, when power is transmitted from the output part of the secondary side rectifier circuit, the secondary side rectifier circuit acts as a primary side AC voltage generation circuit, and the primary side AC voltage generation circuit by the switching elements Q1 and Q2 is 2 Acts as a secondary rectifier circuit. Therefore, it is possible to transmit power by exchanging the relationship between power transmission and power reception.
  • the load Ro is a rechargeable battery, a storage capacitance, or a circuit including its charge / discharge control circuit
  • the rechargeable battery is charged by transmitting power from the power transmission coil Lp to the power reception coil Ls. .
  • a load circuit is connected to a portion to which the input power source Vi is connected in FIG. 15, power is transmitted from the power receiving coil Ls to the power transmitting coil Lp using the rechargeable battery or the storage capacitance as an input power source.
  • the tenth embodiment has the following effects.
  • FIG. 16 is a circuit diagram of a switching power supply device 111 used as the power transmission system of the eleventh embodiment.
  • a parallel capacitor Cp is provided on the power transmission device side, and a parallel capacitor Cs is provided on the power reception device side.
  • FIG. 17 is a circuit diagram of the switching power supply device 112 used as the power transmission system of the twelfth embodiment.
  • the resonance capacitors Cr and Crs are divided into resonance capacitors Cr1 and Cr2, Crs1 and Crs2, respectively.
  • the current flowing through each of the capacitors Cr and Crs is divided into two capacitors, so that losses in the capacitors can be dispersed and heat generation is also dispersed.
  • the capacitors Cr1 and Cr2 and the capacitors Crs1 and Crs2 play both roles of holding a DC voltage and acting as a series resonance capacitor.
  • FIG. 18 is a circuit diagram of the switching power supply device 113 used as the power transmission system of the thirteenth embodiment.
  • an AC voltage generation circuit on the power transmission device side is configured by a full bridge circuit including four switching elements Q1, Q2, Q5, and Q6.
  • a side rectifier circuit on the power receiving device side is configured by a bridge rectifier circuit including four switching elements Q3, Q4, Q7, and Q8.
  • the parameter M shown in FIG. 18 is shown as a mutual coefficient of electromagnetic field coupling by combining magnetic field coupling by mutual inductance and electric field coupling by mutual capacitance.
  • the switching elements Q1, Q2, Q5, Q6 on the power transmission device side, and the switching elements Q3, Q4, Q7, Q8 on the power reception device side Since the voltage applied to each becomes half, the loss in the switching element can be reduced.
  • FIG. 19 is a circuit diagram of the switching power supply device 114 used as the power transmission system of the fourteenth embodiment.
  • a capacitor Cc is provided on the power transmission device side to constitute a voltage clamp circuit.
  • Others are the same as those of the tenth embodiment shown in FIG.
  • the fourteenth embodiment if the negative voltage charged to the capacitor Cc is ⁇ Vc, the square wave voltage generated on the power transmission device side has voltage amplitudes of + Vi and ⁇ Vc.
  • the control characteristic with respect to the fluctuation of the is improved.
  • FIG. 20 is a circuit diagram of the switching power supply device 115 used as the power transmission system of the fifteenth embodiment.
  • a capacitor Ccs is provided on the power receiving device side, and a voltage clamp circuit is configured on the secondary side.
  • Others are the same as those of the fourteenth embodiment shown in FIG.
  • the input voltage Vi is converted into a square wave voltage on the power transmission device side, and the square wave voltage has voltage amplitudes of + Vi and ⁇ Vc.
  • the negative voltage (Vcs) is charged in the capacitor Ccs on the power receiving device side
  • the AC square wave voltage applied to the synchronous rectifier circuit by the switching elements Q3 and Q4 has voltage amplitudes of + Vo and ⁇ Vcs. Since the voltage amplitude is increased in this way, the control characteristics with respect to fluctuations in the output voltage are also improved. That is, the output voltage can be easily adjusted over a wide range.
  • FIG. 21 is a circuit diagram of the switching power supply 116 used as the power transmission system of the sixteenth embodiment.
  • the rectifier circuit on the power receiving device side is constituted by rectifier diodes D3 and D4. According to this configuration, the power receiving device can be used as a unidirectional power transmission system with a simple configuration.
  • FIG. 22 is a circuit diagram of a switching power supply device 117 used as the power transmission system of the seventeenth embodiment.
  • an AC voltage generation circuit on the power transmission device side is configured by a full bridge circuit including four switching elements Q1, Q2, Q5, and Q6.
  • the rectifier circuit on the power receiving device side is constituted by a diode bridge formed by rectifier diodes D3, D4, D7, and D8.
  • the seventeenth embodiment it can be used as a unidirectional power transmission system.
  • the withstand voltage of the rectifier diode can be halved.
  • FIG. 23 is a circuit diagram of a switching power supply device 118 used as the power transmission system of the eighteenth embodiment.
  • Ml is the mutual inductance that is the magnetic field coupling between the inductors Lp and Ls of the coil
  • Mc is the mutual capacitance that is the electric field coupling between the capacitors Cp and Cs
  • Mcr is the mutual capacitance that is the electric field coupling between the capacitors Cr and Crs.
  • the electromagnetic field coupling circuit 90 including the capacitors Cr and Crs is illustrated.
  • an electromagnetic resonance circuit by appropriately setting the mutual inductance Ml, the mutual capacitance Mc, and the mutual capacitance Mcr, and perform high-efficiency power transmission by electromagnetic coupling. it can.

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Abstract

The present invention causes a first resonant circuit, which comprises an inductor (Lr) and a capacitor (Cr), to resonate with a second resonant circuit, which comprises an inductor (Lrs) and a capacitor (Crs), resulting in sympathetic vibration. Thereby, power is transmitted between a first coil (np) and a second coil (ns) by using a combination of a magnetic field and an electric field. Furthermore, by raising the switching frequency of the specific resonance frequency of the entire multi-resonant circuit and actuating switching, zero voltage switching is carried out, enabling a significant reduction in switching loss and highly efficient operation. Thus configured is a switching power supply device which is reduced in size and which increases power conversion efficiency.

Description

スイッチング電源装置Switching power supply
　本発明は、１次側にスイッチング素子、２次側に整流回路を備え、電磁界共鳴現象を利用して電力伝送を行うスイッチング電源装置に関するものである。 The present invention relates to a switching power supply device that includes a switching element on the primary side and a rectifier circuit on the secondary side, and transmits power using an electromagnetic resonance phenomenon.
　近年、電子機器の小型軽量化は進み、スイッチング電源の高効率化、小型軽量化の市場要求はいっそう高まっている。例えばＬＣ共振現象を利用してトランスに正弦波状の共振電流を流して動作をさせる電流共振ハーフブリッジコンバータは、出力電流リップルの特性が比較的緩和される薄型テレビなどの市場において、高効率である特長を活かして実用化が進んでいる。 In recent years, electronic devices have become smaller and lighter, and the market demand for higher efficiency and smaller and lighter switching power supplies is increasing. For example, a current resonance half bridge converter that operates by flowing a sinusoidal resonance current through a transformer using the LC resonance phenomenon is highly efficient in a market such as a flat-screen TV in which the characteristics of output current ripple are relatively relaxed. Utilizing the features, practical application is progressing.
　例えばＬＣ直列共振型ＤＣ－ＤＣコンバータとして特許文献１が開示されている。図１は特許文献１のスイッチング電源装置の基本的な回路図である。このスイッチング電源装置は電流共振型のハーフブリッジＤＣ／ＤＣコンバータであり、トランスＴの１次巻線ｎｐに、インダクタＬｒとキャパシタＣｒとからなるＬＣ共振回路および二つのスイッチング素子Ｑ１，Ｑ２が接続されている。トランスＴの２次巻線ｎｓ１，ｎｓ２にはダイオードＤｓ１，Ｄｓ２およびキャパシタＣｏからなる整流平滑回路が構成されている。 For example, Patent Document 1 is disclosed as an LC series resonance type DC-DC converter. FIG. 1 is a basic circuit diagram of a switching power supply device disclosed in Patent Document 1. In FIG. This switching power supply device is a current resonance type half-bridge DC / DC converter, and an LC resonance circuit composed of an inductor Lr and a capacitor Cr and two switching elements Q1 and Q2 are connected to a primary winding np of a transformer T. ing. A rectifying and smoothing circuit including diodes Ds1 and Ds2 and a capacitor Co is formed in the secondary windings ns1 and ns2 of the transformer T.
　このような構成により、スイッチング素子Ｑ１，Ｑ２はデッドタイムを挟んで交互にオン、オフされて、トランスＴに流れる電流波形は正弦波状の共振波形となる。また、この二つのスイッチング素子Ｑ１，Ｑ２のオン期間／オフ期間の両期間ともに１次側から２次側に電力が伝送される。 With such a configuration, the switching elements Q1 and Q2 are alternately turned on and off with a dead time therebetween, and the current waveform flowing through the transformer T becomes a sinusoidal resonance waveform. In addition, power is transmitted from the primary side to the secondary side in both the on-period / off-period of the two switching elements Q1, Q2.
特開平９－３０８２４３号公報JP-A-9-308243
　しかしながら、特許文献１のスイッチング電源装置においては、トランスは電磁誘導を利用した絶縁トランスとして用いており、磁気結合を利用した変圧器として利用しているに過ぎない。電磁誘導を利用したトランスでは、１次巻線に流れる電流により発生した磁束を２次巻線と鎖交させて電流を流し、電気から磁気、そして電気へと効率よく変換することが重要となっている。一般に、１次巻線に流れる電流により発生した磁束のうち、２次巻線と鎖交する磁束の割合は（磁気）結合度と呼ばれ、電磁誘導を利用したトランスでは、電力変換効率を高めるためには磁気結合度を高めることが重要となっている。しかしながら、磁気飽和を防止するため、または物理的な制約によりトランスの磁気結合度を大きくすることが困難な場合も多く、結果的に電力変換効率を低下させるという結果となっている。 However, in the switching power supply of Patent Document 1, the transformer is used as an insulating transformer using electromagnetic induction, and is only used as a transformer using magnetic coupling. In transformers that use electromagnetic induction, it is important to efficiently convert from electricity to magnetism and electricity by linking the magnetic flux generated by the current flowing in the primary winding with the secondary winding to flow the current. ing. In general, the ratio of the magnetic flux interlinked with the secondary winding out of the magnetic flux generated by the current flowing through the primary winding is called (magnetic) coupling degree. In a transformer using electromagnetic induction, the power conversion efficiency is increased. For this purpose, it is important to increase the degree of magnetic coupling. However, in many cases, it is difficult to increase the magnetic coupling degree of the transformer in order to prevent magnetic saturation or due to physical limitations, resulting in a decrease in power conversion efficiency.
　また特許文献１のスイッチング電源装置においては、出力制御には周波数制御、PFM（Pulse Frequency Modulation）制御を用いるため、最低動作周波数に合わせて平滑回路を設計する必要があり、小型化の妨げとなっている。また、磁性部品の小型化を図るためにMHz級での動作を考えると、動作周波数が変化することは、出力の制御性、EMC（電磁両立性）などを考慮すると課題が大きい。 In the switching power supply of Patent Document 1, since frequency control and PFM (Pulse Frequency Modulation) control are used for output control, it is necessary to design a smoothing circuit in accordance with the minimum operating frequency, which hinders miniaturization. ing. Considering the operation at the MHz class in order to reduce the size of the magnetic component, the change of the operating frequency has a big problem in consideration of output controllability, EMC (electromagnetic compatibility) and the like.
　本発明は、小型化を図りつつ電力変換効率を高めたスイッチング電源装置を提供することを目的としている。 It is an object of the present invention to provide a switching power supply device that increases the power conversion efficiency while reducing the size.
　本発明のスイッチング電源装置は次のように構成される。
（１）１次巻線および２次巻線を備える電磁界結合回路と、
前記１次巻線に接続されたスイッチング回路を備え、該スイッチング回路をスイッチング素子、ダイオード、およびキャパシタの並列接続回路で構成して、入力される直流電圧から交流電圧を発生する１次側交流電圧発生回路と、
　前記交流電圧を直流電圧に整流する２次側整流回路と、
　１次側に構成され、第１の直列共振インダクタおよび第１の直列共振キャパシタを含む第１の共振回路と、
　２次側に構成され、第２の直列共振インダクタおよび第２の直列共振キャパシタを含む第２の共振回路と、
　前記１次側交流電圧発生回路のスイッチング素子をデッドタイムを挟んで交互にオン／オフすることによりほぼ方形波状または台形波状の交流電圧を発生させるスイッチング制御回路と、
　を備えたスイッチング電源装置において、
　前記スイッチング制御回路は、電磁界結合回路を含めた前記第１の共振回路と前記第２の共振回路とを合わせた全体となる複共振回路に流入する電流が、前記１次側交流電圧発生回路から発生する交流電圧よりも遅れる正弦波状の共振電流波形となって、前記スイッチング素子のオン期間およびオフ期間の両期間に前記電磁界結合回路を介して１次側から２次側へ電力が伝送されるように、前記複共振回路に対してインピーダンスが最も小さくなる固有の共振周波数よりも高いスイッチング周波数で前記１次側交流電圧発生回路のスイッチング素子をスイッチング動作し、
　前記電磁界結合回路は、前記１次巻線と前記２次巻線との間で相互インダクタンスを介した磁界結合および相互キャパシタンスを介した電界結合とが混合した電磁界共鳴回路を構成し、
　前記第１の共振回路と前記第２の共振回路とが共鳴して前記電磁界結合回路の１次側から２次側へ電力が伝送されることを特徴とする。 The switching power supply device of the present invention is configured as follows.
(1) an electromagnetic field coupling circuit including a primary winding and a secondary winding;
A primary side AC voltage that includes a switching circuit connected to the primary winding and that is configured by a parallel connection circuit of a switching element, a diode, and a capacitor, and generates an AC voltage from an input DC voltage Generating circuit;
A secondary side rectifier circuit for rectifying the AC voltage into a DC voltage;
A first resonant circuit configured on the primary side and including a first series resonant inductor and a first series resonant capacitor;
A second resonant circuit configured on the secondary side and including a second series resonant inductor and a second series resonant capacitor;
A switching control circuit for generating a substantially square wave or trapezoidal AC voltage by alternately turning on / off switching elements of the primary side AC voltage generation circuit with a dead time therebetween;
In a switching power supply device comprising:
In the switching control circuit, a current flowing into a multi-resonance circuit including the first resonance circuit including the electromagnetic field coupling circuit and the second resonance circuit is converted into the primary AC voltage generation circuit. A sinusoidal resonance current waveform that is delayed from the AC voltage generated from the power source, and power is transmitted from the primary side to the secondary side via the electromagnetic field coupling circuit during both the on period and the off period of the switching element. As described above, the switching operation of the switching element of the primary side AC voltage generating circuit is performed at a switching frequency higher than a specific resonance frequency at which the impedance is smallest with respect to the double resonance circuit,
The electromagnetic field coupling circuit constitutes an electromagnetic field resonance circuit in which a magnetic field coupling via a mutual inductance and an electric field coupling via a mutual capacitance are mixed between the primary winding and the secondary winding,
The first resonance circuit and the second resonance circuit resonate to transmit power from the primary side to the secondary side of the electromagnetic field coupling circuit.
（２）例えば、前記スイッチング制御回路は、前記１次側交流電圧発生回路のスイッチング周波数を一定にし、前記スイッチング回路に電流が導通する期間をオン期間、その他の期間をオフ期間として、複数のスイッチング回路のオン期間比率を制御することで、前記２次側整流回路から得られる出力電力を調整する。 (2) For example, the switching control circuit has a switching frequency of the primary side AC voltage generating circuit constant, and a plurality of switching operations with a period in which current is conducted to the switching circuit being an on period and another period being an off period. The output power obtained from the secondary side rectifier circuit is adjusted by controlling the on-period ratio of the circuit.
（３）例えば、前記スイッチング制御回路は、前記１次側交流電圧発生回路のスイッチング周波数を一定にし、前記スイッチング素子のオン期間比率を制御することで、前記２次側整流回路から得られる出力電力を調整する。 (3) For example, the switching control circuit makes the switching frequency of the primary side AC voltage generation circuit constant, and controls the on-period ratio of the switching element, whereby the output power obtained from the secondary side rectifier circuit Adjust.
（４）前記２次側整流回路は、オン期間またはオフ期間のいずれか、または両期間に、前記２次巻線に発生する電圧を静電エネルギーとして前記第２の共振キャパシタに蓄えて、前記オン期間とオフ期間のそれぞれの期間に前記２次巻線に発生する電圧を加算して直流電圧として出力することが好ましい。 (4) The secondary side rectifier circuit stores the voltage generated in the secondary winding as electrostatic energy in the second resonant capacitor in either the on period or the off period, or both periods, It is preferable to add the voltage generated in the secondary winding during each of the on period and the off period and output the result as a DC voltage.
（５）前記第１の直列共振キャパシタと前記第２の直列共振キャパシタのいずれかまたは両方は直流電圧を保持することが好ましい。 (5) It is preferable that either or both of the first series resonant capacitor and the second series resonant capacitor hold a DC voltage.
（６）前記１次巻線または前記２次巻線に対して並列に並列共振キャパシタが備えられていることが好ましい。 (6) It is preferable that a parallel resonant capacitor is provided in parallel with the primary winding or the secondary winding.
（７）前記並列共振キャパシタは前記１次巻線または前記２次巻線の浮遊容量で構成されていることが好ましい。 (7) It is preferable that the parallel resonant capacitor is composed of a stray capacitance of the primary winding or the secondary winding.
（８）前記相互キャパシタンスは前記１次巻線と前記２次巻線との間に形成される浮遊容量で構成されていることが好ましい。 (8) It is preferable that the mutual capacitance is composed of a stray capacitance formed between the primary winding and the secondary winding.
（９）前記第１の直列共振インダクタまたは前記第２の直列共振インダクタは前記電磁界結合回路の漏れインダクタンスで構成されていることが好ましい。 (9) It is preferable that the first series resonant inductor or the second series resonant inductor is configured by a leakage inductance of the electromagnetic field coupling circuit.
（１０）前記相互インダクタンスを前記１次巻線と前記２次巻線との間に等価的に形成される励磁インダクタンスで構成されていることが好ましい。 (10) It is preferable that the mutual inductance is constituted by an exciting inductance formed equivalently between the primary winding and the secondary winding.
（１１）前記スイッチング回路はMOSFETであることが好ましい。 (11) The switching circuit is preferably a MOSFET.
（１２）前記２次側整流回路に備えられる前記交流電圧を直流電圧に整流する整流素子はMOSFETであることが好ましい。 (12) It is preferable that the rectifying element that rectifies the AC voltage provided in the secondary side rectifier circuit into a DC voltage is a MOSFET.
（１３）前記２次側整流回路の出力部から電力が伝送されるとき、前記２次側整流回路は前記１次側交流電圧発生回路として作用するとともに、前記１次側交流電圧発生回路は前記２次側整流回路として作用し、双方向に電力伝送が可能であることが好ましい。 (13) When power is transmitted from the output of the secondary rectifier circuit, the secondary rectifier circuit acts as the primary AC voltage generator, and the primary AC voltage generator is It is preferable that it acts as a secondary side rectifier circuit and can transmit power in both directions.
（１４）例えば、前記１次巻線はフェライトなどの磁芯を有するトランスの１次側に設けられた巻線、前記２次巻線は前記トランスの２次側に設けられた巻線である。 (14) For example, the primary winding is a winding provided on the primary side of a transformer having a magnetic core such as ferrite, and the secondary winding is a winding provided on the secondary side of the transformer. .
（１５）例えば、前記１次巻線は送電装置に設けられた送電コイル、前記２次巻線は前記送電装置に近接配置される受電装置に設けられた受電コイルである。 (15) For example, the primary winding is a power transmission coil provided in a power transmission device, and the secondary winding is a power reception coil provided in a power reception device arranged close to the power transmission device.
　本発明によれば、１次側と２次側の双方にＬＣ共振回路を備え、２つのＬＣ共振回路を共鳴させて、１次巻線と２次巻線との間で磁界と電界の結合を利用して電力伝送を行うことができる。また、共振現象を利用することで有効電力のみを１次側から２次側へ伝送し、無効電力は１次側と２次側の双方のＬＣ共振回路において循環しているため電力損失は非常に小さい。さらにスイッチング周波数としては、電磁界結合回路を含めた１次側共振回路と２次側共振回路とを合わせた全体の複共振回路において入力インピーダンスが最も小さくなる固有共振周波数よりも高い周波数にてスイッチング素子をオンオフ動作させることにより、スイッチング素子におけるゼロ電圧スイッチング（ＺＶＳ）動作が可能となる。 According to the present invention, the LC resonance circuit is provided on both the primary side and the secondary side, and the two LC resonance circuits are resonated to couple the magnetic field and the electric field between the primary winding and the secondary winding. Can be used to transmit power. In addition, by utilizing the resonance phenomenon, only active power is transmitted from the primary side to the secondary side, and reactive power is circulated in both the primary and secondary LC resonance circuits, so power loss is extremely high. Small. Further, as a switching frequency, switching is performed at a frequency higher than the natural resonance frequency at which the input impedance becomes the smallest in the entire double resonance circuit including the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit. A zero voltage switching (ZVS) operation in the switching element can be performed by turning on and off the element.
図１は特許文献１のスイッチング電源装置の基本的な回路図である。FIG. 1 is a basic circuit diagram of a switching power supply device disclosed in Patent Document 1. In FIG. 図２は第１の実施形態のスイッチング電源装置１０１の回路図である。FIG. 2 is a circuit diagram of the switching power supply apparatus 101 according to the first embodiment. 図３は、図２に示したスイッチング電源装置１０１の各部の電圧電流波形図である。FIG. 3 is a voltage-current waveform diagram of each part of the switching power supply apparatus 101 shown in FIG. 図４は第２の実施形態のスイッチング電源装置１０２の回路図である。FIG. 4 is a circuit diagram of the switching power supply apparatus 102 according to the second embodiment. 図５（Ａ）はスイッチング電源装置１０２の各部の電圧電流の波形図である。図５（Ｂ）は、図１に示したスイッチング電源装置の各部の電圧電流の波形図である。FIG. 5A is a waveform diagram of voltage / current of each part of the switching power supply apparatus 102. FIG. 5B is a waveform diagram of voltage and current in each part of the switching power supply device shown in FIG. 図６は第３の実施形態のスイッチング電源装置１０３の回路図である。FIG. 6 is a circuit diagram of the switching power supply apparatus 103 according to the third embodiment. 図７は第４の実施形態のスイッチング電源装置１０４の回路図である。FIG. 7 is a circuit diagram of the switching power supply device 104 of the fourth embodiment. 図８は第５の実施形態のスイッチング電源装置１０５の回路図である。FIG. 8 is a circuit diagram of the switching power supply device 105 of the fifth embodiment. 図９は図８に示した直列共振キャパシタＣｒに入力される電圧の波形である。FIG. 9 shows the waveform of the voltage input to the series resonant capacitor Cr shown in FIG. 図１０はスイッチング周期に対するスイッチング回路Ｓ１の導通期間の比率であるオン時比率Ｄと、スイッチング回路Ｓ１の導通期間に対するスイッチング回路Ｓ２の導通期間の比率であるオン期間比率Ｄａとについて、出力電圧Ｖｏとの関係を示す図である。ここで実線はオン期間比率Ｄａの特性カーブ、破線はオン時比率Ｄの特性カーブである。FIG. 10 shows the output voltage Vo with respect to the on-time ratio D, which is the ratio of the conduction period of the switching circuit S1 to the switching cycle, and the on-period ratio Da, which is the ratio of the conduction period of the switching circuit S2 to the conduction period of the switching circuit S1. It is a figure which shows the relationship. Here, the solid line is the characteristic curve of the on-period ratio Da, and the broken line is the characteristic curve of the on-time ratio D. 図１１は第６の実施形態のスイッチング電源装置１０６の回路図である。FIG. 11 is a circuit diagram of the switching power supply device 106 of the sixth embodiment. 図１２は第７の実施形態のスイッチング電源装置１０７の回路図である。FIG. 12 is a circuit diagram of the switching power supply device 107 of the seventh embodiment. 図１３は第８の実施形態のスイッチング電源装置１０８の回路図である。FIG. 13 is a circuit diagram of the switching power supply device 108 of the eighth embodiment. 図１４は第９の実施形態のスイッチング電源装置１０９の回路図である。FIG. 14 is a circuit diagram of the switching power supply device 109 of the ninth embodiment. 図１５は第１０の実施形態のスイッチング電源装置１１０の回路図である。FIG. 15 is a circuit diagram of the switching power supply device 110 of the tenth embodiment. 図１６は第１１の実施形態の電力送電システムとして用いられるスイッチング電源装置１１１の回路図である。FIG. 16 is a circuit diagram of a switching power supply device 111 used as the power transmission system of the eleventh embodiment. 図１７は第１２の実施形態の電力送電システムとして用いられるスイッチング電源装置１１２の回路図である。FIG. 17 is a circuit diagram of the switching power supply device 112 used as the power transmission system of the twelfth embodiment. 図１８は第１３の実施形態の電力送電システムとして用いられるスイッチング電源装置１１３の回路図である。FIG. 18 is a circuit diagram of the switching power supply device 113 used as the power transmission system of the thirteenth embodiment. 図１９は第１４の実施形態の電力送電システムとして用いられるスイッチング電源装置１１４の回路図である。FIG. 19 is a circuit diagram of the switching power supply device 114 used as the power transmission system of the fourteenth embodiment. 図２０は第１５の実施形態の電力送電システムとして用いられるスイッチング電源装置１１５の回路図である。FIG. 20 is a circuit diagram of the switching power supply device 115 used as the power transmission system of the fifteenth embodiment. 図２１は第１６の実施形態の電力送電システムとして用いられるスイッチング電源装置１１６の回路図である。FIG. 21 is a circuit diagram of the switching power supply 116 used as the power transmission system of the sixteenth embodiment. 図２２は第１７の実施形態の電力送電システムとして用いられるスイッチング電源装置１１７の回路図である。FIG. 22 is a circuit diagram of a switching power supply device 117 used as the power transmission system of the seventeenth embodiment. 図２３は第１８の実施形態の電力送電システムとして用いられるスイッチング電源装置１１８の回路図である。FIG. 23 is a circuit diagram of a switching power supply device 118 used as the power transmission system of the eighteenth embodiment.
《第１の実施形態》
　図２は第１の実施形態のスイッチング電源装置１０１の回路図である。
　スイッチング電源装置１０１は、入力部に入力電源Ｖｉが入力され、出力部から負荷Ｒｏへ安定した直流電力を供給する回路である。スイッチング電源装置１０１は次の各部を備えている。
　・１次巻線ｎｐおよび２次巻線ｎｓを備えるトランスを用いた電磁界結合回路９０
　・１次巻線ｎｐに接続された、スイッチング素子Ｑ１を含むスイッチング回路Ｓ１、スイッチング素子Ｑ２を含むスイッチング回路Ｓ２
　・２次巻線ｎｓに接続された整流ダイオードＤ３，Ｄ４および平滑キャパシタＣｏ
　・１次巻線ｎｐに接続された直列共振インダクタＬｒおよび直列共振キャパシタＣｒによる第１のＬＣ直列共振回路
　・２次巻線ｎｓに接続された直列共振インダクタＬｒｓおよび直列共振キャパシタＣｒｓによる第２のＬＣ直列共振回路
　・電磁界結合回路９０を含めて、第１のＬＣ直列共振回路と第２のＬＣ直列共振回路により構成される複共振回路４０
　・スイッチング素子Ｑ１，Ｑ２に接続されたスイッチング制御回路１０
　・負荷Ｒｏへの出力電圧の検出信号をスイッチング制御回路１０へフィードバックする絶縁回路３０
　・１次巻線ｎｐに対して並列に接続された並列共振キャパシタＣｐ
　・１次巻線ｎｓに対して並列に接続された並列共振キャパシタＣｓ
　・１次巻線ｎｐと２次巻線ｎｓとの間に接続された相互キャパシタンスＣｍ
　前記電磁界結合回路は磁界結合と電界結合を融合した電磁界結合回路（電磁界共鳴回路）を構成している。前記直列共振キャパシタＣｒ，Ｃｒｓは共に直流電圧を保持するためのキャパシタを兼ねている。 << First Embodiment >>
FIG. 2 is a circuit diagram of the switching power supply apparatus 101 according to the first embodiment.
The switching power supply device 101 is a circuit in which an input power supply Vi is input to an input unit and supplies stable DC power from an output unit to a load Ro. The switching power supply apparatus 101 includes the following units.
An electromagnetic field coupling circuit 90 using a transformer having a primary winding np and a secondary winding ns
A switching circuit S1 including a switching element Q1 and a switching circuit S2 including a switching element Q2 connected to the primary winding np
-Rectifier diodes D3 and D4 connected to the secondary winding ns and the smoothing capacitor Co
A first LC series resonance circuit by a series resonance inductor Lr and a series resonance capacitor Cr connected to the primary winding np A second LC series resonance circuit by a series resonance inductor Lrs and a series resonance capacitor Crs connected to the secondary winding ns LC series resonance circuit A double resonance circuit 40 including the electromagnetic coupling circuit 90 and including a first LC series resonance circuit and a second LC series resonance circuit.
-Switching control circuit 10 connected to switching elements Q1, Q2
An isolation circuit 30 that feeds back a detection signal of an output voltage to the load Ro to the switching control circuit 10
A parallel resonant capacitor Cp connected in parallel to the primary winding np
A parallel resonant capacitor Cs connected in parallel to the primary winding ns
A mutual capacitance Cm connected between the primary winding np and the secondary winding ns
The electromagnetic field coupling circuit constitutes an electromagnetic field coupling circuit (electromagnetic resonance circuit) in which magnetic field coupling and electric field coupling are combined. Both the series resonant capacitors Cr and Crs also serve as capacitors for holding a DC voltage.
　１次側ではスイッチング素子Ｑ１の導通期間にキャパシタＣｒを充電し、スイッチング素子Ｑ２の導通期間にキャパシタＣｒを放電する。一方、２次側では、スイッチング素子Ｑ１の導通期間にキャパシタＣｒｓを放電し、スイッチング素子Ｑ２の導通期間に２次巻線ｎｓに発生する電圧を静電エネルギーとしてキャパシタＣｒｓを充電し、スイッチング素子Ｑ１，Ｑ２それぞれの導通期間に発生する２次巻線ｎｓの電圧を加算して出力する。前記整流ダイオードＤ３，Ｄ４およびキャパシタＣｒｓによる回路は、充放電と整流を行う加算整流回路８０を構成する。 On the primary side, the capacitor Cr is charged during the conduction period of the switching element Q1, and the capacitor Cr is discharged during the conduction period of the switching element Q2. On the other hand, on the secondary side, the capacitor Crs is discharged during the conduction period of the switching element Q1, and the capacitor Crs is charged using the voltage generated in the secondary winding ns during the conduction period of the switching element Q2 as electrostatic energy. , Q2 add the voltages of the secondary windings ns generated during each conduction period and output the result. The circuit including the rectifier diodes D3 and D4 and the capacitor Crs constitutes an addition rectifier circuit 80 that performs charge / discharge and rectification.
　なお、１次側のインダクタＬｍは部品としてのインダクタであっても、トランスＴの１次巻線ｎｐの励磁インダクタンスを表したものであっても良い。同様に、１次－２次間のキャパシタンスＣｍは部品としてのキャパシタンスであっても、トランスＴの浮遊容量である相互キャパシタンスを表したものであっても良い。 The primary-side inductor Lm may be an inductor as a component, or may represent the excitation inductance of the primary winding np of the transformer T. Similarly, the primary-secondary capacitance Cm may be a capacitance as a component or may represent a mutual capacitance which is a stray capacitance of the transformer T.
　図２において太い破線で囲んだ部分は電磁界結合回路９０、細い破線で囲んだ部分は複共振回路４０を構成している。この電磁界結合回路９０を含めた複共振回路４０は、１次側と２次側の２つのＬＣ共振回路で共鳴動作する。 In FIG. 2, a portion surrounded by a thick broken line constitutes an electromagnetic field coupling circuit 90, and a portion surrounded by a thin broken line constitutes a multiple resonance circuit 40. The multi-resonance circuit 40 including the electromagnetic field coupling circuit 90 resonates with two LC resonance circuits on the primary side and the secondary side.
　具体的な作用は次のとおりである。
（１）Ｌｒ－Ｃｒからなる第１の共振回路と、Ｌｒｓ－Ｃｒｓからなる第２の共振回路とが共鳴することによりそれぞれが共振し、１次巻線ｎｐと２次巻線ｎｓとの間で相互インダクタンスによる磁界と相互キャパシタンスによる電界の２つの結合を利用して電力伝送を行う。ただし、図２ではトランスＴの励磁インダクタンスを相互インダクタンス（Ｌｍ）として利用し、回路素子としての図示を略している。 The specific action is as follows.
(1) The first resonance circuit made of Lr—Cr and the second resonance circuit made of Lrs—Crs resonate to resonate with each other, and between the primary winding np and the secondary winding ns. Thus, power transmission is performed using two couplings of a magnetic field due to mutual inductance and an electric field due to mutual capacitance. However, in FIG. 2, the exciting inductance of the transformer T is used as a mutual inductance (Lm), and illustration as a circuit element is omitted.
　なお、キャパシタＣｐ、Ｃｓは電磁界結合での電力伝送を促進する。すなわち、キャパシタＣｐ、Ｃｓ、そして相互キャパシタンスＣｍとでπ型の電界結合による電力伝送回路を構成して電力を伝送する。因みに相互キャパシタンスＣｍは、共振キャパシタＣｒ、Ｃｒｓとも電界結合による電力伝送回路を構成している。 Note that the capacitors Cp and Cs promote power transmission through electromagnetic coupling. That is, the capacitors Cp and Cs and the mutual capacitance Cm constitute a power transmission circuit by π-type electric field coupling to transmit power. Incidentally, the mutual capacitance Cm constitutes a power transmission circuit by electric field coupling with the resonance capacitors Cr and Crs.
　また、キャパシタＣｐ、Ｃｓは、スイッチング素子がターンオフした際の転流期間において、１次側では、共振キャパシタＣｒに流れる共振電流ｉｒをスイッチング回路の並列キャパシタ（スイッチング素子Ｑ１，Ｑ２に対して並列に接続されているキャパシタ）へ流れる電流とキャパシタＣｐとに分流する。共振電流ｉｒが大きくなればなるほど、キャパシタＣｐに流れる電流は大きくなり、転流期間においてスイッチング回路の並列キャパシタンスへ流れる電流はほぼ一定となり、キャパシタＣｐのキャパシタンスを適宜設定することにより、出力電力の変動に対してデッドタイム期間と転流期間との差分を補正することができる。２次側でも同様に、2次側の共振電流が大きくなればなるほど、キャパシタＣｓに流れる電流は大きくなり、キャパシタＣｓのキャパシタンスを適宜設定することにより、出力電力の変動に対してデッドタイム期間と、ダイオードＤ３とダイオードＤ４とで電流経路が切り替わる際の期間との差分を補正することができる。 Further, the capacitors Cp and Cs are arranged such that, in the commutation period when the switching element is turned off, on the primary side, the resonance current ir flowing in the resonance capacitor Cr is parallel to the switching circuit parallel capacitor (switching elements Q1 and Q2). The current flows to the capacitor (connected capacitor) and the capacitor Cp. The larger the resonance current ir, the larger the current flowing through the capacitor Cp. The current flowing through the parallel capacitance of the switching circuit during the commutation period is substantially constant. By appropriately setting the capacitance of the capacitor Cp, the output power varies. In contrast, the difference between the dead time period and the commutation period can be corrected. Similarly, on the secondary side, the larger the resonance current on the secondary side, the larger the current flowing in the capacitor Cs. By appropriately setting the capacitance of the capacitor Cs, the dead time period can be reduced with respect to fluctuations in output power. The difference between the period when the current path is switched between the diode D3 and the diode D4 can be corrected.
（２）スイッチング素子Ｑ１とＱ２はデッドタイムを挟んで交互にオンオフすることにより、直流電圧Ｖｉを方形波状または台形波状の電圧波形に整形する。一方、整流ダイオードＤ３とＤ４は交互に導通することにより方形波状または台形波状の電圧波形を直流電圧に整形する。 (2) The switching elements Q1 and Q2 are alternately turned on and off with a dead time therebetween, thereby shaping the DC voltage Vi into a square wave or trapezoidal voltage waveform. On the other hand, the rectifier diodes D3 and D4 are alternately turned on to shape a square wave or trapezoidal voltage waveform into a DC voltage.
（３）スイッチング素子Ｑ１とＱ２によるスイッチング周波数ｆｓに対して１次側と２次側の２つの共振回路は共鳴する。電磁界結合回路９０を含めた１次側と２次側の２つの共振回路から複共振回路４０は構成される。複共振回路４０は、複共振回路４０の合成インピーダンスが最も小さくなる固有の共振周波数ｆｒを有しており、スイッチング周波数ｆｓと共振周波数ｆｒとが近づいて共振することにより、それぞれ２つの共振回路に流れる電流は大きくなり、出力電力は増加する。すなわち、電磁界結合回路を含めた１次側共振回路と２次側共振回路とを合成した全体の複共振回路９０が有する固有の共振周波数ｆｒよりも高いスイッチング周波数ｆｓでスイッチング素子をオンオフ動作させ、スイッチング周波数ｆｓが固有の共振周波数ｆｒに近づいて共振することにより、複共振回路に流入する電流は大きくなり、出力電力は増加する。 (3) The two resonance circuits on the primary side and the secondary side resonate with respect to the switching frequency fs by the switching elements Q1 and Q2. The double resonance circuit 40 includes two resonance circuits including a primary side and a secondary side including the electromagnetic field coupling circuit 90. The double resonance circuit 40 has a specific resonance frequency fr that minimizes the combined impedance of the double resonance circuit 40, and the switching frequency fs and the resonance frequency fr approach each other to resonate. The flowing current increases and the output power increases. That is, the switching element is turned on / off at a switching frequency fs higher than the inherent resonance frequency fr of the entire multiple resonance circuit 90 that is a combination of the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit. As the switching frequency fs approaches the inherent resonance frequency fr and resonates, the current flowing into the multiple resonance circuit increases and the output power increases.
　一方、スイッチング周波数ｆｓを一定にして動作をさせる場合においては、２つのスイッチング回路の導通期間の比率となるオン期間比ＤａがＤａ＝１、すなわちスイッチング周期に対する第１のスイッチング回路Ｓ１の導通期間の比率である、コンバータのオン期間比率ＤがＤ＝０．５に近づくほど出力電力は増加する。 On the other hand, when the operation is performed with the switching frequency fs constant, the ON period ratio Da, which is the ratio of the conduction periods of the two switching circuits, is Da = 1, that is, the conduction period of the first switching circuit S1 with respect to the switching period. The output power increases as the converter ON period ratio D, which is the ratio, approaches D = 0.5.
（４）１次側、２次側のキャパシタＣｒとＣｒｓは、直流電圧を保持する動作と共振動作の２つの作用を果たす。 (4) The primary-side and secondary-side capacitors Cr and Crs perform two actions, that is, an operation for holding a DC voltage and a resonance operation.
　図２における電磁界結合回路９０は、トランスＴの１次巻線ｎｐの励磁インダクタンスＬｍ、２次巻線ｎｓの励磁インダクタンスＬｍｓ、直列共振インダクタＬｒ，ＬｒｓおよびキャパシタＣｐ，Ｃｓなど、トランスＴの寄生成分で構成されていてもよい。この場合、トランスは、電気的な絶縁を可能とするトランスとしての機能と共振インダクタ、共振キャパシタなどの電気的パラメータとを一体化した共振複合トランスと称することができ、電磁界結合装置として利用することができる。 2 includes a parasitic inductance of the transformer T such as the exciting inductance Lm of the primary winding np of the transformer T, the exciting inductance Lms of the secondary winding ns, the series resonant inductors Lr and Lrs, and the capacitors Cp and Cs. You may be comprised with the component. In this case, the transformer can be referred to as a resonant composite transformer that integrates a function as a transformer that enables electrical insulation and electrical parameters such as a resonant inductor and a resonant capacitor, and is used as an electromagnetic coupling device. be able to.
　図３は、図２に示したスイッチング電源装置１０１の各部の電圧電流波形図である。スイッチング電源装置１０１の各タイミングでの動作は次のとおりである。 FIG. 3 is a voltage-current waveform diagram of each part of the switching power supply apparatus 101 shown in FIG. The operation of the switching power supply apparatus 101 at each timing is as follows.
　まず、トランスＴの１次巻線ｎｐの励磁インダクタンスをＬｍ、励磁電流をｉｍで表す。スイッチング素子Ｑ１、Ｑ２、Ｑ３、Ｑ４のゲート・ソース間電圧をｖｇｓ１、ｖｇｓ２で表し、ドレイン・ソース間電圧をそれぞれｖｄｓ１、ｖｄｓ２で表し、Ｑ１のドレイン電流をｉｄ１で表す。Ｑ１、Ｑ２は、両スイッチング素子がオフとなる短いデッドタイムを挟んで交互にオン、オフ動作を行い、デッドタイム期間にＱ１、Ｑ２に流れる電流をそれぞれ転流させてゼロ電圧スイッチング（ＺＶＳ）動作を行う。１スイッチング周期における各状態での動作を以下に示す。 First, the exciting inductance of the primary winding np of the transformer T is represented by Lm, and the exciting current is represented by im. The gate-source voltages of the switching elements Q1, Q2, Q3, and Q4 are represented by vgs1 and vgs2, the drain-source voltages are represented by vds1 and vds2, respectively, and the drain current of Q1 is represented by id1. Q1 and Q2 are alternately turned on and off with a short dead time when both switching elements are turned off, and the current flowing in Q1 and Q2 is commutated in the dead time period, respectively, so that zero voltage switching (ZVS) operation is performed. I do. The operation in each state in one switching cycle is shown below.
［１］State1　時刻ｔ０～ｔ１
　スイッチング素子Ｑ１は導通しており、巻線ｎｐに電流が流れ、キャパシタＣｒは充電される。ダイオードＤ３は導通しており、巻線ｎｐに印加された電圧により、巻線ｎｓに電圧が誘起され、巻線ｎｓに誘起された電圧とキャパシタＣｒｓの両端電圧が加算されて負荷に電圧が印加され、キャパシタＣｒｓは放電して電流が供給される。スイッチング素子Ｑ１がターンオフするとState2となる。 [1] State1 time t0 to t1
The switching element Q1 is conductive, a current flows through the winding np, and the capacitor Cr is charged. The diode D3 is conductive, and a voltage applied to the winding np induces a voltage in the winding ns. The voltage induced in the winding ns and the voltage across the capacitor Crs are added to apply a voltage to the load. The capacitor Crs is discharged and supplied with current. When the switching element Q1 is turned off, the state becomes State2.
［２］State2　時刻ｔ１～ｔ２
　インダクタＬｒに流れていた電流ｉｒにより、スイッチング素子Ｑ１の並列キャパシタ（寄生容量）は充電され、スイッチング素子Ｑ２の並列キャパシタ（寄生容量）は放電される。電圧ｖｄｓ１が電圧Ｖｉ、電圧ｖｄｓ２が０ＶになるとState3となる。 [2] State2 time t1 to t2
The parallel capacitor (parasitic capacitance) of the switching element Q1 is charged by the current ir flowing through the inductor Lr, and the parallel capacitor (parasitic capacitance) of the switching element Q2 is discharged. When the voltage vds1 becomes the voltage Vi and the voltage vds2 becomes 0V, the state 3 is obtained.
［３］State3　時刻ｔ２～ｔ３
　スイッチング素子Ｑ２の並列ダイオードは導通している。この期間においてスイッチング素子Ｑ２をターンオンすることでＺＶＳ動作が行われる。ダイオードＤ３に流れる電流が０ＡになるとState4となる。 [3] State3 time t2 to t3
The parallel diode of the switching element Q2 is conductive. In this period, the ZVS operation is performed by turning on the switching element Q2. When the current flowing through the diode D3 becomes 0A, the state becomes State4.
［４］State4　時刻ｔ３～ｔ４
　スイッチング素子Ｑ２は導通しており、巻線ｎｐには電流が流れ、キャパシタＣｒは放電される。ダイオードＤ４は導通しており、巻線ｎｐに印加された電圧により、巻線ｎｓに電圧が誘起され、キャパシタＣｒｓは充電される。負荷にはキャパシタＣｏの電圧が印加されて電流が供給される。このようにしてインダクタＬｒに流れる電流ｉｒは正弦波状の共振電流波形となる。ダイオードＤ４に流れる電流が０になるとState5となる。 [4] State4 time t3 to t4
The switching element Q2 is conductive, a current flows through the winding np, and the capacitor Cr is discharged. The diode D4 is conducting, and a voltage is induced in the winding ns by the voltage applied to the winding np, and the capacitor Crs is charged. The load is supplied with current by applying the voltage of the capacitor Co. In this way, the current ir flowing through the inductor Lr has a sinusoidal resonance current waveform. When the current flowing through the diode D4 becomes 0, State5 is obtained.
［５］State5　時刻ｔ４～ｔ５
　１次側ではトランスの励磁電流ｉｍが流れ、電流ｉｒと等しくなる。２次側では、負荷にはキャパシタＣｏの電圧が印加されて電流が供給される。Ｑ２がターンオフするとState6となる。 [5] State5 time t4 to t5
On the primary side, an exciting current im of the transformer flows and becomes equal to the current ir. On the secondary side, the voltage of the capacitor Co is applied to the load to supply current. When Q2 turns off, it becomes State6.
［６］State6　時刻ｔ５～ｔ０
　インダクタＬｒに流れていた電流ｉｒにより、スイッチング素子Ｑ１の並列キャパシタ（寄生容量）は放電され、スイッチング素子Ｑ２の並列キャパシタ（寄生容量）は充電される。電圧ｖｄｓ１が電圧０Ｖ、電圧ｖｄｓ２がＶｉになるとState1となる。 [6] State6 time t5 to t0
Due to the current ir flowing in the inductor Lr, the parallel capacitor (parasitic capacitance) of the switching element Q1 is discharged, and the parallel capacitor (parasitic capacitance) of the switching element Q2 is charged. When the voltage vds1 becomes 0V and the voltage vds2 becomes Vi, State1 is obtained.
　以後、State1～6を周期的に繰り返す。 Thereafter, State 1 to 6 are repeated periodically.
　スイッチング制御回路１０は次の制御を行う。
（１）電磁界結合回路を含めた１次側共振回路と２次側共振回路とを合わせた全体の複共振回路に対して、入力インピーダンスが最も小さくなる固有共振周波数ｆｒよりスイッチング周波数を高くする。このことにより、そのスイッチング周波数では複共振回路は誘導性となる。そのため、インダクタＬｒに流れる電流位相が、１次側交流電圧発生回路による方形波（台形波）状の交流電圧の電圧位相に対して遅れた状態にできるので、スイッチング素子Ｑ１の電圧Ｖｄｓ１が０の状態でスイッチング素子Ｑ１をターンオンできる。同様に、スイッチング素子Ｑ２の電圧ｖｄｓ２が０の状態でスイッチング素子Ｑ２をターンオンできる。すなわちＺＶＳ（ゼロ電圧スイッチング）動作を行うことになり、スイッチング損失を大幅に低減でき、高効率動作が可能となる。また、全負荷範囲において共振周波数ｆｒより高いスイッチング周波数にて動作をするため、全負荷範囲に亘ってゼロ電圧スイッチング（ＺＶＳ）動作が実現できる。 The switching control circuit 10 performs the following control.
(1) The switching frequency is set higher than the natural resonance frequency fr at which the input impedance is minimized for the entire multiple resonance circuit including the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit. . This makes the double resonant circuit inductive at that switching frequency. Therefore, since the phase of the current flowing through the inductor Lr can be delayed with respect to the voltage phase of the square-wave (trapezoidal) AC voltage generated by the primary AC voltage generation circuit, the voltage Vds1 of the switching element Q1 is 0. In this state, the switching element Q1 can be turned on. Similarly, the switching element Q2 can be turned on when the voltage vds2 of the switching element Q2 is zero. That is, ZVS (zero voltage switching) operation is performed, switching loss can be greatly reduced, and high-efficiency operation is possible. Further, since the operation is performed at a switching frequency higher than the resonance frequency fr in the entire load range, a zero voltage switching (ZVS) operation can be realized over the entire load range.
（２）１次側交流電圧発生回路のスイッチング周波数を一定にし、スイッチング素子Ｑ１を含むスイッチング回路と，スイッチング素子Ｑ２を含むスイッチング回路の導通期間の比率、すなわちオン期間比を制御することで、２次側整流回路から得られる出力電力を調整する。 (2) By making the switching frequency of the primary side AC voltage generating circuit constant and controlling the ratio of the conduction period of the switching circuit including the switching element Q1 and the switching circuit including the switching element Q2, that is, the ON period ratio, 2 The output power obtained from the secondary rectifier circuit is adjusted.
（３）またはスイッチング素子Ｑ１，Ｑ２のオン期間比を１に設定して、１次側交流電圧発生回路のスイッチング周波数を変化させるようにスイッチング素子Ｑ１，Ｑ２を制御することで、２次側整流回路から得られる出力電力を調整する。 (3) Alternatively, the on-period ratio of the switching elements Q1 and Q2 is set to 1, and the switching elements Q1 and Q2 are controlled so as to change the switching frequency of the primary side AC voltage generation circuit, thereby secondary side rectification. Adjust the output power available from the circuit.
（４）さらには、前記（２）のオン期間比制御と（３）のスイッチング周波数制御を組み合わせて、最適な制御特性が得られるように制御することで、２次側整流回路から得られる出力電力を調整する。 (4) Further, the output obtained from the secondary side rectifier circuit is controlled by combining the on period ratio control of (2) and the switching frequency control of (3) so as to obtain an optimum control characteristic. Adjust the power.
　第１の実施形態によれば次のような効果を奏する。
（ａ）１次側と２次側の共振を用いて磁界結合と電界結合を融合した電磁界結合回路（電磁界共鳴回路）を構成することにより、磁界結合だけで電力伝送を行う場合よりも電力伝送効率が高くなり、高効率動作が可能となる。 According to the first embodiment, there are the following effects.
(A) By constructing an electromagnetic field coupling circuit (electromagnetic resonance circuit) that combines magnetic field coupling and electric field coupling using resonance on the primary side and the secondary side, power transmission is performed only by magnetic field coupling. The power transmission efficiency becomes high and high-efficiency operation becomes possible.
（ｂ）電磁界結合回路を含めた１次側共振回路と２次側共振回路とを合わせた全体の複共振回路に対して、入力インピーダンスが最も小さくなる固有共振周波数ｆｒに対してスイッチング周波数を高くすることにより、前述のとおりＺＶＳ（ゼロ電圧スイッチング）動作を行うことになり、スイッチング損失を大幅に低減でき、高効率動作が可能となる。 (B) For the entire multiple resonance circuit including the primary side resonance circuit including the electromagnetic field coupling circuit and the secondary side resonance circuit, the switching frequency is set to the natural resonance frequency fr where the input impedance is minimized. By increasing the value, ZVS (zero voltage switching) operation is performed as described above, switching loss can be greatly reduced, and high-efficiency operation is possible.
（ｃ）１次側のスイッチング素子のオン期間とオフ期間のそれぞれの期間に発生する２次巻線電圧を加算して直流電圧として出力するように構成することで一定のスイッチング周波数でのオン期間比制御（ＰＷＭ制御）で出力電圧の安定化が可能となる。 (C) The on-period at a constant switching frequency by adding the secondary winding voltage generated in each of the on-period and off-period of the switching element on the primary side and outputting it as a DC voltage. The output voltage can be stabilized by the ratio control (PWM control).
（ｄ）１次側のスイッチング素子のオン期間とオフ期間のそれぞれの期間に発生する２次巻線電圧を加算して直流電圧として出力するように構成することで、整流器に印加される電圧をセンタータップ整流の場合と比較して半分にすることができ、損失を低減することができる。 (D) The secondary winding voltage generated in each of the ON period and the OFF period of the primary-side switching element is added and output as a DC voltage, whereby the voltage applied to the rectifier is Compared with the center tap rectification, it can be halved and the loss can be reduced.
（ｅ）トランスの漏れインダクタンス、励磁インダクタンス、浮遊容量、相互キャパシタンスなどを用いて電磁界結合回路を構成することにより、少ない部品数でコンバータを構成することができ、小型軽量化を図ることができる。 (E) By configuring an electromagnetic field coupling circuit using leakage inductance, excitation inductance, stray capacitance, mutual capacitance, etc. of the transformer, a converter can be configured with a small number of parts, and a reduction in size and weight can be achieved. .
（ｆ）１次側、２次側のキャパシタＣｒとＣｒｓが直流電圧を保持する動作と共振動作の２つの役割を行うことで、直流電圧を交流電圧に変換しながら、他方では複共振回路を構成する共振キャパシタンスとして共振動作を行うため、部品数を減らすことができる。また、一定のスイッチング周波数でのオン期間比制御（ＰＷＭ制御）が可能となる。 (F) The primary-side and secondary-side capacitors Cr and Crs perform the two roles of holding the DC voltage and the resonance operation, thereby converting the DC voltage into the AC voltage, while the double resonance circuit is Since the resonance operation is performed as the resonance capacitance to be configured, the number of components can be reduced. Further, ON period ratio control (PWM control) at a constant switching frequency is possible.
（ｇ）キャパシタＣｐ、Ｃｓにより相互キャパシタンスと合わせてπ型の電界結合による電力伝送回路を構成して電磁界結合での電力伝送を促進する。また、そのキャパシタンスＣｐ、Ｃｓを適宜設定することにより、出力電力の変動に対してデッドタイム期間と転流期間との差分を補正することができる。 (G) The capacitors Cp and Cs are combined with the mutual capacitance to constitute a power transmission circuit by π-type electric field coupling to promote power transmission by electromagnetic coupling. In addition, by appropriately setting the capacitances Cp and Cs, the difference between the dead time period and the commutation period can be corrected with respect to fluctuations in the output power.
《第２の実施形態》
　図４は第２の実施形態のスイッチング電源装置１０２の回路図である。この例では第１の実施形態のスイッチング電源装置１０１と異なり、２次側の整流ダイオードＤ３，Ｄ４に代えてＦＥＴによるスイッチング素子Ｑ３，Ｑ４を備えている。すなわちスイッチング素子Ｑ３，Ｑ４で２次側整流回路を構成している。スイッチング素子Ｑ３，Ｑ４は、それぞれ並列にダイオード（寄生ダイオード）、キャパシタ（寄生容量）を備えており、スイッチング回路Ｓ３、Ｓ４を構成している。また、電源入力部にキャパシタＣｉを設けている。スイッチング制御回路２０は２次側のスイッチング素子Ｑ３，Ｑ４の制御を行う。 << Second Embodiment >>
FIG. 4 is a circuit diagram of the switching power supply apparatus 102 according to the second embodiment. In this example, unlike the switching power supply device 101 of the first embodiment, switching elements Q3 and Q4 using FETs are provided instead of the rectifier diodes D3 and D4 on the secondary side. That is, the secondary side rectifier circuit is constituted by the switching elements Q3 and Q4. The switching elements Q3 and Q4 each include a diode (parasitic diode) and a capacitor (parasitic capacitance) in parallel, and constitute switching circuits S3 and S4. In addition, a capacitor Ci is provided in the power input section. The switching control circuit 20 controls the secondary side switching elements Q3 and Q4.
　なお、トランスＴの１次巻線ｎｐ、２次巻線ｎｓの励磁インダクタンスをＬｍ、Ｌｍｓ、スイッチング素子Ｑ１，Ｑ２，Ｑ３，Ｑ４の寄生容量および寄生ダイオードは図示を略している。 The excitation inductances of the primary winding np and the secondary winding ns of the transformer T are Lm and Lms, the parasitic capacitances of the switching elements Q1, Q2, Q3, and Q4 and the parasitic diode are not shown.
　２次側のスイッチング制御回路２０は、１次側のスイッチング素子Ｑ１と同期してスイッチング素子Ｑ３をオン／オフさせ、１次側のスイッチング素子Ｑ２と同期してスイッチング素子Ｑ４をオン／オフさせる。すなわち同期整流を行う。スイッチング電源装置１０２全体の動作は第１の実施形態で示したスイッチング電源装置１０１と同様である。 The secondary side switching control circuit 20 turns on / off the switching element Q3 in synchronization with the primary side switching element Q1, and turns on / off the switching element Q4 in synchronization with the primary side switching element Q2. That is, synchronous rectification is performed. The overall operation of the switching power supply 102 is the same as that of the switching power supply 101 shown in the first embodiment.
　図５（Ａ）は前記スイッチング電源装置１０２の各部の電圧電流の波形図である。また、図５（Ｂ）は、図１に示したスイッチング電源装置の各部の電圧電流の波形図である。ここでｖｄｓ１はスイッチング素子Ｑ１のドレイン・ソース間電圧、ｉｒはキャパシタＣｒに流れる電流、Ｖｄｓ３はスイッチング素子Ｑ３のドレイン・ソース間電圧、ｉｄ３はスイッチング素子Ｑ３に流れる電流、ｉｄ４はスイッチング素子Ｑ４に流れる電流である。 FIG. 5A is a waveform diagram of voltage and current of each part of the switching power supply device 102. FIG. FIG. 5B is a waveform diagram of voltage and current in each part of the switching power supply device shown in FIG. Here, vds1 is the drain-source voltage of the switching element Q1, ir is the current flowing through the capacitor Cr, Vds3 is the drain-source voltage of the switching element Q3, id3 is the current flowing through the switching element Q3, and id4 is flowing through the switching element Q4. Current.
　図５（Ａ）、図５（Ｂ）に違いが表れているように、従来のセンタータップ整流方式では、二つの整流ダイオードのうち一方のダイオードにしか電流が流れず、不均一な動作となり、整流ダイオードに流れる電流のピーク値および実効電流値は大きくなり、導通損が増加する。また、第２の実施形態のスイッチング電源装置では、２次側のスイッチング素子Ｑ３，Ｑ４に印加される電圧は、出力電圧とほぼ同じとなるが、従来のセンタータップ整流方式では出力電圧の２倍の電圧が印加されることが分かる。第２の実施形態によれば、電圧ストレスを低減することにより耐電圧の小さい整流素子を用いることができ、一般に、耐電圧の小さいダイオード素子は順方向電圧降下が小さく、また耐電圧の小さいFETはオン抵抗が小さいために流れる電流による導通損を低減することができ、高効率動作が可能となる。 As shown in FIG. 5 (A) and FIG. 5 (B), in the conventional center tap rectification method, current flows only to one of the two rectifier diodes, resulting in uneven operation. The peak value and effective current value of the current flowing through the rectifier diode increase, and the conduction loss increases. In the switching power supply device of the second embodiment, the voltage applied to the secondary side switching elements Q3 and Q4 is almost the same as the output voltage, but in the conventional center tap rectification method, it is twice the output voltage. It can be seen that the following voltage is applied. According to the second embodiment, it is possible to use a rectifying element having a low withstand voltage by reducing voltage stress. Generally, a diode element having a low withstand voltage has a small forward voltage drop and a FET having a low withstand voltage. Since the on-resistance is small, the conduction loss due to the flowing current can be reduced, and high-efficiency operation is possible.
　第２の実施形態のスイッチング電源装置１０２は、図４に表れているように、そのトポロジーは入出力間で対称性を有する。そのため、２次側整流回路の出力部から電力が送電されるとき、２次側整流回路は１次側交流電圧発生回路として作用し、スイッチング素子Ｑ１，Ｑ２による１次側交流電圧発生回路は２次側整流回路として作用する。したがって、トランスＴの１次側から２次側へ、または２次側から１次側への双方向について電力伝送が可能である。 As shown in FIG. 4, the switching power supply 102 of the second embodiment has a symmetrical topology between input and output. Therefore, when power is transmitted from the output part of the secondary side rectifier circuit, the secondary side rectifier circuit acts as a primary side AC voltage generation circuit, and the primary side AC voltage generation circuit by the switching elements Q1 and Q2 is 2 Acts as a secondary rectifier circuit. Therefore, power transmission is possible in both directions from the primary side to the secondary side of the transformer T or from the secondary side to the primary side.
　例えば、負荷Ｒｏが充電電池や蓄電キャパシタンスであったり、その充放電制御回路を含む回路であったりする場合、トランスＴの１次側から２次側へ電力が伝送されることにより、前記充電電池は充電される。そして、図４において入力電源Ｖｉが接続される部分に負荷回路が接続されれば、前記充電電池や蓄電キャパシタンスを入力電源とし、電力伝送の方向を逆にしたトランスＴの２次側から１次側への電力伝送が可能となる。 For example, when the load Ro is a rechargeable battery, a storage capacitance, or a circuit including its charge / discharge control circuit, power is transmitted from the primary side to the secondary side of the transformer T, whereby the rechargeable battery Is charged. Then, if a load circuit is connected to the portion to which the input power source Vi is connected in FIG. 4, the primary side from the secondary side of the transformer T with the rechargeable battery or the storage capacitance as the input power source and the direction of power transmission reversed. Power transmission to the side becomes possible.
　第２の実施形態によれば、第１の実施形態で述べた効果以外に次のような効果を奏する。
（ａ）ＦＥＴによるスイッチング素子Ｑ３，Ｑ４で同期整流動作を行うことにより、順方向降下電圧が小さくなり、整流回路での導通損が低減できる。 According to the second embodiment, there are the following effects in addition to the effects described in the first embodiment.
(A) By performing the synchronous rectification operation with the switching elements Q3 and Q4 using FETs, the forward voltage drop is reduced and the conduction loss in the rectifier circuit can be reduced.
（ｂ）１次側と２次側を入れ替えて逆方向に電力を伝送する双方向コンバータとしての動作が可能である。 (B) Operation as a bidirectional converter that transmits power in the reverse direction by switching the primary side and the secondary side is possible.
《第３の実施形態》
　図６は第３の実施形態のスイッチング電源装置１０３の回路図である。この例では入力電源Ｖｉの電圧を分圧するキャパシタＣｉ１，Ｃｉ２、および出力電圧Ｖｏを分圧するキャパシタＣｉｓ１，Ｃｉｓ２を備えている。ここでは、トランスＴの１次巻線ｎｐ、２次巻線ｎｓの励磁インダクタンス、または外付けのインダクタンスであるインダクタＬｍ、Ｌｍｓを図示している。その他は第２の実施形態で図４に示したものと同様である。 << Third Embodiment >>
FIG. 6 is a circuit diagram of the switching power supply apparatus 103 according to the third embodiment. In this example, capacitors Ci1 and Ci2 that divide the voltage of the input power source Vi and capacitors Cis1 and Cis2 that divide the output voltage Vo are provided. Here, the primary winding np of the transformer T, the exciting inductance of the secondary winding ns, or inductors Lm and Lms which are external inductances are illustrated. Others are the same as those of the second embodiment shown in FIG.
　第３の実施形態では、入力電圧ＶｉがキャパシタＣｉ１，Ｃｉ２で分圧され、出力電圧ＶｏがキャパシタＣｉｓ１，Ｃｉｓ２で分圧される。なお、キャパシタＣｉ１、Ｃｉｓ１が直流入力電圧を分圧するため、直流電圧を保持する機能を果たすため、直列共振キャパシタＣｒ，Ｃｒｓは共振用キャパシタとして作用し、直流電圧を保持する用、すなわち直流電圧成分をバイアスして共振動作する機能としては作用しない。全体のコンバータ動作は第１の実施形態で示したとおりである。 In the third embodiment, the input voltage Vi is divided by the capacitors Ci1 and Ci2, and the output voltage Vo is divided by the capacitors Cis1 and Cis2. Since the capacitors Ci1 and Cis1 divide the DC input voltage and perform the function of holding the DC voltage, the series resonance capacitors Cr and Crs act as resonance capacitors and hold the DC voltage, that is, DC voltage components. It does not function as a function of resonance operation by biasing. The overall converter operation is as shown in the first embodiment.
　第３の実施形態によれば、第１・第２の実施形態で述べた効果以外に次のような効果を奏する。
（ａ）入力電源ＶｉがキャパシタＣｉ１，Ｃｉ２のそれぞれの電圧として分圧され、スイッチング素子Ｑ１，Ｑ２のオン／オフの両サイクルで入力電源ＶｉからキャパシタＣｉ１，Ｃｉ２へと電流が流れ、入力電源Ｖｉから流出する入力電流の実効値が小さくなって、電流経路での導通損が低減される。 According to the third embodiment, there are the following effects in addition to the effects described in the first and second embodiments.
(A) The input power source Vi is divided as the respective voltages of the capacitors Ci1 and Ci2, and current flows from the input power source Vi to the capacitors Ci1 and Ci2 in both ON / OFF cycles of the switching elements Q1 and Q2, and the input power source Vi The effective value of the input current flowing out of the current becomes small, and the conduction loss in the current path is reduced.
（ｂ）上述（ａ）と同様に、出力電圧ＶｏはキャパシタＣｉｓ１，Ｃｉｓ２のそれぞれの電圧として分圧されており、スイッチング素子Ｑ１，Ｑ２のオン／オフの両サイクルでキャパシタＣｉｓ１，Ｃｉｓ２から出力電圧Ｖｏへ流れる電流の実効値は小さくなり、導通損失は低減される。 (B) Similar to the above (a), the output voltage Vo is divided as the respective voltages of the capacitors Cis1 and Cis2, and the output voltage from the capacitors Cis1 and Cis2 in both the on / off cycles of the switching elements Q1 and Q2. The effective value of the current flowing to Vo becomes small, and the conduction loss is reduced.
《第４の実施形態》
　図７は第４の実施形態のスイッチング電源装置１０４の回路図である。この例では入力電源Ｖｉの電圧を分圧するキャパシタＣｒ１，Ｃｒ２、および出力電圧Ｖｏを分圧するキャパシタＣｒｓ１，Ｃｒｓ２を備えている。すなわち、第２の実施形態で示したスイッチング電源装置における直列共振キャパシタＣｒをＣｒ１，Ｃｒ２に分割し、直列共振キャパシタＣｒｓをＣｒｓ１，Ｃｒｓ２に分割したものである。ここでは、トランスＴの１次巻線ｎｐと２次巻線ｎｓとの間に形成される等価的な相互インダクタンスＬｍを図示し、１次巻線ｎｐと２次巻線ｎｓからなるトランスＴは、理想トランスとして図示されている。トランスＴを理想的なトランスにて構成する場合、インダクタＬｒ、インダクタＬｒｓ、およびキャパシタＣｐ、Ｃｓを単体の回路素子にて構成できる。また、トランスＴの寄生要素を用いて電磁結合回路９０そのものを単体の共振複合トランスにて構成することも可能である。その他は第２の実施形態で図４に示したものと同様である。 << Fourth Embodiment >>
FIG. 7 is a circuit diagram of the switching power supply device 104 of the fourth embodiment. In this example, capacitors Cr1 and Cr2 for dividing the voltage of the input power source Vi and capacitors Crs1 and Crs2 for dividing the output voltage Vo are provided. That is, the series resonance capacitor Cr in the switching power supply device shown in the second embodiment is divided into Cr1 and Cr2, and the series resonance capacitor Crs is divided into Crs1 and Crs2. Here, an equivalent mutual inductance Lm formed between the primary winding np and the secondary winding ns of the transformer T is illustrated, and the transformer T composed of the primary winding np and the secondary winding ns is , Illustrated as an ideal transformer. When the transformer T is configured by an ideal transformer, the inductor Lr, the inductor Lrs, and the capacitors Cp and Cs can be configured by a single circuit element. It is also possible to configure the electromagnetic coupling circuit 90 itself by a single resonance composite transformer using parasitic elements of the transformer T. Others are the same as those of the second embodiment shown in FIG.
　第４の実施形態では、直列共振キャパシタに流れる電流が２つのキャパシタに分割されるので、キャパシタによる損失が分散され全体の損失が低減され、発熱が分散される。 In the fourth embodiment, since the current flowing through the series resonant capacitor is divided into two capacitors, the loss due to the capacitor is dispersed, the overall loss is reduced, and the heat generation is dispersed.
　なお、キャパシタＣｒ１，Ｃｒ２およびキャパシタＣｒｓ１，Ｃｒｓ２は、直流電圧を保持する作用と直列共振用キャパシタとしての作用の両方の役割を果たす。 The capacitors Cr1 and Cr2 and the capacitors Crs1 and Crs2 play both roles of holding a DC voltage and acting as a series resonance capacitor.
《第５の実施形態》
　図８は第５の実施形態のスイッチング電源装置１０５の回路図である。この例では１次側にキャパシタＣｃを設けて電圧クランプ回路を構成している。その他は第２の実施形態で図４に示したものと同様である。 << Fifth Embodiment >>
FIG. 8 is a circuit diagram of the switching power supply device 105 of the fifth embodiment. In this example, a capacitor Cc is provided on the primary side to constitute a voltage clamp circuit. Others are the same as those of the second embodiment shown in FIG.
　図８に示したスイッチング電源装置では、スイッチング素子Ｑ１のターンオフ後、１次巻線ｎｐの電圧がスイッチング素子Ｑ２の寄生ダイオードを介してキャパシタＣｃに図８に示す方向の電圧がチャージされ、スイッチング素子Ｑ２がオンのときにキャパシタＣｃにチャージされた電圧（+Ｖｃ）が複共振回路へ印加される。すなわち、入力電圧Ｖｉが方形波電圧に変換され、その方形波電圧は＋Ｖｉと－Ｖｃの電圧振幅となる。 In the switching power supply device shown in FIG. 8, after the switching element Q1 is turned off, the voltage of the primary winding np is charged to the capacitor Cc through the parasitic diode of the switching element Q2 in the direction shown in FIG. When Q2 is on, the voltage (+ Vc) charged in the capacitor Cc is applied to the multiple resonance circuit. That is, the input voltage Vi is converted into a square wave voltage, and the square wave voltage has voltage amplitudes of + Vi and −Vc.
　図９は図８に示した直列共振キャパシタＣｒと電磁界結合回路90と直列共振キャパシタＣｒｓからなる複共振回路に与えられる電圧の波形である。ここで実線は第５の実施形態の場合の波形、破線は第１～第４の実施形態の場合の波形である。このように、第１～第４の実施形態では共振回路への入力電源電圧が＋Ｖｉと０Ｖと変化し、電圧振幅は、Ｖｉであるのに比べ、第５の実施形態では入力電源電圧が＋Ｖｉから－Ｖｃへと大きく変化し、電圧振幅は、（Ｖｉ+Ｖｃ）で動作することになる。また、電圧クランプ回路を構成するキャパシタＣｃの両端電圧Ｖｃは、スイッチング周期に対するスイッチング素子Ｑ１の導通期間の比率であるオン期間比率Ｄによって変化し、出力電圧Ｖｏを広範囲に亘って制御できる。このことは出力電圧が一定である場合に入力電源電圧が広範囲に亘って変化する場合への適用に優れることを表している。このように電圧クランプ回路を構成することにより、入力電圧の変動に対する制御特性が改善される。すなわち入力電圧が大きく変動しても出力電圧の安定化が図れる。 FIG. 9 is a waveform of a voltage applied to the multiple resonance circuit including the series resonance capacitor Cr, the electromagnetic field coupling circuit 90, and the series resonance capacitor Crs shown in FIG. Here, the solid line is the waveform in the case of the fifth embodiment, and the broken line is the waveform in the case of the first to fourth embodiments. As described above, in the first to fourth embodiments, the input power supply voltage to the resonance circuit changes between + Vi and 0 V, and the voltage amplitude is Vi in the fifth embodiment, compared to + Vi. Greatly changes from −Vc to a voltage amplitude of (Vi + Vc). Further, the voltage Vc across the capacitor Cc constituting the voltage clamp circuit changes according to the ON period ratio D which is the ratio of the conduction period of the switching element Q1 to the switching period, and the output voltage Vo can be controlled over a wide range. This indicates that the present invention is excellent in application when the input power supply voltage varies over a wide range when the output voltage is constant. By configuring the voltage clamp circuit in this way, control characteristics with respect to fluctuations in the input voltage are improved. That is, the output voltage can be stabilized even if the input voltage varies greatly.
　図１０はスイッチング周期に対するスイッチング回路Ｓ１の導通期間の比率であるオン時比率Ｄと、スイッチング回路Ｓ１の導通期間に対するスイッチング回路Ｓ２の導通期間の比率であるオン期間比率Ｄａとについて、出力電圧Ｖｏとの関係を示す図である。ここで実線はオン期間比率Ｄａの特性カーブ、破線はオン時比率Ｄの特性カーブである。このように、オン期間比率ＤａではＤａ＝１のとき、オン時比率ＤではＤ＝０．５のときに出力電圧は最も大きくなる。 FIG. 10 shows the output voltage Vo with respect to the on-time ratio D, which is the ratio of the conduction period of the switching circuit S1 to the switching cycle, and the on-period ratio Da, which is the ratio of the conduction period of the switching circuit S2 to the conduction period of the switching circuit S1. It is a figure which shows the relationship. Here, the solid line is the characteristic curve of the on-period ratio Da, and the broken line is the characteristic curve of the on-time ratio D. As described above, the output voltage becomes the highest when Da = 1 at the on-period ratio Da and when D = 0.5 at the on-time ratio D.
《第６の実施形態》
　図１１は第６の実施形態のスイッチング電源装置１０６の回路図である。この例では１次側にキャパシタＣｃを設けて電圧クランプ回路を構成している。また、入力電源Ｖｉの電圧を分圧するキャパシタＣｉ１，Ｃｉ２、および出力電圧Ｖｏを分圧するキャパシタＣｉｓ１，Ｃｉｓ２を備えている。また１次巻線ｎｐの励磁インダクタンスを回路パラメータとして表記している。ここでは、トランスＴの１次巻線ｎｐ、２次巻線ｎｓの間に形成される等価的な相互インダクタンスＬｍを図示し、１次巻線ｎｐと２次巻線ｎｓからなるトランスＴは、理想トランスとして図示されている。トランスＴを理想的なトランスにて構成する場合、インダクタＬｒ、インダクタＬｒｓ、およびキャパシタＣｐ、Ｃｓを単体の回路素子にて構成できる。また、トランスＴの寄生要素を用いて電磁結合回路９０そのものを単体の共振複合トランスにて構成することも可能である。その他は第２の実施形態で図４に示したものと同様である。 << Sixth Embodiment >>
FIG. 11 is a circuit diagram of the switching power supply device 106 of the sixth embodiment. In this example, a capacitor Cc is provided on the primary side to constitute a voltage clamp circuit. Further, capacitors Ci1 and Ci2 that divide the voltage of the input power source Vi and capacitors Cis1 and Cis2 that divide the output voltage Vo are provided. The exciting inductance of the primary winding np is shown as a circuit parameter. Here, an equivalent mutual inductance Lm formed between the primary winding np and the secondary winding ns of the transformer T is illustrated, and the transformer T composed of the primary winding np and the secondary winding ns is: It is shown as an ideal transformer. When the transformer T is configured by an ideal transformer, the inductor Lr, the inductor Lrs, and the capacitors Cp and Cs can be configured by a single circuit element. It is also possible to configure the electromagnetic coupling circuit 90 itself by a single resonance composite transformer using parasitic elements of the transformer T. Others are the same as those of the second embodiment shown in FIG.
　この第６の実施形態によれば、入力電源電圧が＋Ｖｉから－Ｖｃの大きな電圧振幅で動作することになるので、入力電圧の変動に対する制御特性が改善される。また、入力電源ＶｉがキャパシタＣｉ１とＣｉ２とで分圧されるので、スイッチング素子Ｑ１，Ｑ２のオン／オフの両サイクルで入力電源ＶｉからキャパシタＣｉ１、Ｃｉ２へと電流が流れ、入力電流の実効値が小さくなって、電流経路での導通損が低減される。さらに、出力電圧ＶｏによりキャパシタＣｉｓ１，Ｃｉｓ２へ流れる電流においても電流実効値は小さくなり、導通損は低減される。 According to the sixth embodiment, since the input power supply voltage operates with a large voltage amplitude of + Vi to −Vc, the control characteristic with respect to the fluctuation of the input voltage is improved. In addition, since the input power source Vi is divided by the capacitors Ci1 and Ci2, current flows from the input power source Vi to the capacitors Ci1 and Ci2 in both ON / OFF cycles of the switching elements Q1 and Q2, and the effective value of the input current Is reduced, and the conduction loss in the current path is reduced. Further, the effective current value is reduced even in the current flowing to the capacitors Cis1 and Cis2 by the output voltage Vo, and the conduction loss is reduced.
《第７の実施形態》
　図１２は第７の実施形態のスイッチング電源装置１０７の回路図である。この例では１次側にキャパシタＣｃを設けて１次側に電圧クランプ回路を構成し、２次側にキャパシタＣｃｓを設けて２次側にも電圧クランプ回路を構成している。その他は第５の実施形態で図７に示したものと同様である。 << Seventh Embodiment >>
FIG. 12 is a circuit diagram of the switching power supply device 107 of the seventh embodiment. In this example, a capacitor Cc is provided on the primary side to constitute a voltage clamp circuit on the primary side, and a capacitor Ccs is provided on the secondary side to constitute a voltage clamp circuit on the secondary side. Others are the same as those of the fifth embodiment shown in FIG.
　図１２に示したスイッチング電源装置では、入力電圧Ｖｉが方形波電圧に変換され、その方形波電圧は＋Ｖｉと－Ｖｃの電圧振幅となる。また、２次側のキャパシタＣｃｓに負電圧（Ｖｃｓ）がチャージされるので、スイッチング素子Ｑ３，Ｑ４による同期整流回路に印加される交流方形波電圧は＋Ｖｏと－Ｖｃｓの電圧振幅となる。このように電圧振幅が大きくなるので、出力電圧の変動に対する制御特性も改善される。すなわち出力電圧の調整が広範囲に亘って容易となる。 In the switching power supply device shown in FIG. 12, the input voltage Vi is converted into a square wave voltage, and the square wave voltage has voltage amplitudes of + Vi and −Vc. Further, since the secondary capacitor Ccs is charged with a negative voltage (Vcs), the AC square wave voltage applied to the synchronous rectifier circuit by the switching elements Q3 and Q4 has a voltage amplitude of + Vo and −Vcs. Since the voltage amplitude is thus increased, the control characteristics with respect to fluctuations in the output voltage are also improved. That is, the output voltage can be easily adjusted over a wide range.
《第８の実施形態》
　図１３は第８の実施形態のスイッチング電源装置１０８の回路図である。この例では４つのスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６によるフルブリッジ回路構成の１次側交流電圧発生回路を設けている。また、４つのスイッチング素子Ｑ３，Ｑ４，Ｑ７，Ｑ８によるブリッジ整流構成の２次側整流回路を設けている。 << Eighth Embodiment >>
FIG. 13 is a circuit diagram of the switching power supply device 108 of the eighth embodiment. In this example, a primary AC voltage generating circuit having a full bridge circuit configuration by four switching elements Q1, Q2, Q5, and Q6 is provided. Further, a secondary side rectifier circuit having a bridge rectification configuration by four switching elements Q3, Q4, Q7, and Q8 is provided.
　この第８の実施形態によれば、第１～第７の実施形態に比べて、１次側のスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６、および２次側のスイッチング素子Ｑ３，Ｑ４，Ｑ７，Ｑ８に印加される電圧がそれぞれ半分となるため、スイッチング素子での損失を低減できる。 According to the eighth embodiment, compared to the first to seventh embodiments, the primary side switching elements Q1, Q2, Q5, Q6 and the secondary side switching elements Q3, Q4, Q7, Q8. Since the voltage applied to each becomes half, the loss in the switching element can be reduced.
《第９の実施形態》
　図１４は第９の実施形態のスイッチング電源装置１０９の回路図である。この例では１次側の共振キャパシタを二つのキャパシタＣｒ１，Ｃｒ２に分割配置し、２次側の共振キャパシタを二つのキャパシタＣｒｓ１，Ｃｒｓ２に分割配置している。また、４つのスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６によるフルブリッジ回路構成の１次側交流電圧発生回路を設けている。また、４つのスイッチング素子Ｑ３，Ｑ４，Ｑ７，Ｑ８によるブリッジ整流構成の２次側整流回路を設けている。 << Ninth embodiment >>
FIG. 14 is a circuit diagram of the switching power supply device 109 of the ninth embodiment. In this example, the primary-side resonant capacitor is divided into two capacitors Cr1 and Cr2, and the secondary-side resonant capacitor is divided into two capacitors Crs1 and Crs2. Further, a primary side AC voltage generating circuit having a full bridge circuit configuration by four switching elements Q1, Q2, Q5, and Q6 is provided. Further, a secondary side rectifier circuit having a bridge rectification configuration by four switching elements Q3, Q4, Q7, and Q8 is provided.
　この第９の実施形態によれば、第１～第３の実施形態等で示した共振キャパシタＣｒ，Ｃｒｓのそれぞれに印加される電圧が２つのキャパシタに分割されて印加されるので、キャパシタでの損失を分散することができる。また、１次側のスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６、および２次側のスイッチング素子Ｑ３，Ｑ４，Ｑ７，Ｑ８に印加される電圧がそれぞれ半分となるため、スイッチング素子での損失を低減できる。 According to the ninth embodiment, the voltage applied to each of the resonant capacitors Cr and Crs shown in the first to third embodiments and the like is divided and applied to two capacitors. Loss can be distributed. Further, since the voltages applied to the primary side switching elements Q1, Q2, Q5, and Q6 and the secondary side switching elements Q3, Q4, Q7, and Q8 are each halved, the loss in the switching elements can be reduced. .
　なお、キャパシタＣｒ１，Ｃｒ２およびキャパシタＣｒｓ１，Ｃｒｓ２は、直流電圧を保持する作用と直列共振用キャパシタとしての作用の両方の役割を果たす。 The capacitors Cr1 and Cr2 and the capacitors Crs1 and Crs2 play both roles of holding a DC voltage and acting as a series resonance capacitor.
《第１０の実施形態》
　これまでに示した各実施形態では、部品としてのトランスを備えて、ＤＣ－ＤＣコンバータとして用いるスイッチング電源装置を例に挙げたが、以降の各実施形態では対向する装置間で電気的には非接触で電力伝送を行う装置の例を示す。 << Tenth Embodiment >>
In each of the embodiments described so far, a switching power supply device provided with a transformer as a component and used as a DC-DC converter has been described as an example. However, in each of the following embodiments, there is no electrical connection between opposing devices. The example of the apparatus which transmits electric power by contact is shown.
　図１５は第１０の実施形態のスイッチング電源装置１１０の回路図である。図１５においてＬｐは送電装置側の送電コイル、Ｌｓは受電装置側の受電コイルである。 FIG. 15 is a circuit diagram of the switching power supply device 110 of the tenth embodiment. In FIG. 15, Lp is a power transmission coil on the power transmission device side, and Ls is a power reception coil on the power reception device side.
　この例では、送電コイルＬｐの等価的な漏れインダクタで第１～第９の実施形態に示したインダクタＬｒ、送電コイルＬｐの等価的な巻線間の並列キャパシタで第１～第９の実施形態に示したキャパシタＣｐを構成している。また、受電コイルＬｓの等価的な漏れインダクタで第１～第９の実施形態に示したインダクタＬｒｓ、受電コイルＬｓの等価的な巻線間の並列キャパシタで第１～第９の実施形態に示したキャパシタＣｓを構成している。また、送電コイルＬｐで磁界結合に関与する等価的なインダクタンスを相互インダクタンスＬｍ、送電コイルＬｐと受信コイルＬｓとの間で電界結合に関与する等価的なキャパシタンスを相互キャパシタンスＣｍとして構成する。 In this example, the equivalent leakage inductor of the power transmission coil Lp is the inductor Lr shown in the first to ninth embodiments, and the parallel capacitor between the equivalent windings of the power transmission coil Lp is the first to ninth embodiments. The capacitor Cp shown in FIG. Further, the inductor Lrs shown in the first to ninth embodiments is an equivalent leakage inductor of the power receiving coil Ls, and the parallel capacitor between the equivalent windings of the power receiving coil Ls is shown in the first to ninth embodiments. The capacitor Cs is configured. Further, an equivalent inductance involved in magnetic field coupling in the power transmission coil Lp is configured as a mutual inductance Lm, and an equivalent capacitance involved in electric field coupling between the power transmission coil Lp and the reception coil Ls is configured as a mutual capacitance Cm.
　図１５に示すパラメータＭｌは磁界結合の相互係数を示したものであり、Ｍｃは電界結合の相互係数を示したものである。相互インダクタンスによる磁界結合（相互係数Ｍｌ）と相互キャパシタンスによる電界結合（相互係数Ｍｃ）との合成により電磁界結合としての相互係数Ｍは構成される。 15 indicates the mutual coefficient of magnetic field coupling, and Mc indicates the mutual coefficient of electric field coupling. The mutual coefficient M as the electromagnetic field coupling is configured by combining the magnetic field coupling by the mutual inductance (mutual coefficient Ml) and the electric field coupling by the mutual capacitance (mutual coefficient Mc).
　この電力伝送システムとして用いられるスイッチング電源装置１１０は、図１５に表れているように、そのトポロジーは入出力間で対称性を有する。そのため、２次側整流回路の出力部から電力が伝送されるとき、２次側整流回路は１次側交流電圧発生回路として作用し、スイッチング素子Ｑ１，Ｑ２による１次側交流電圧発生回路は２次側整流回路として作用する。したがって、送電と受電の関係を交換して電力伝送することも可能である。 The switching power supply 110 used as the power transmission system has a symmetrical topology between input and output, as shown in FIG. Therefore, when power is transmitted from the output part of the secondary side rectifier circuit, the secondary side rectifier circuit acts as a primary side AC voltage generation circuit, and the primary side AC voltage generation circuit by the switching elements Q1 and Q2 is 2 Acts as a secondary rectifier circuit. Therefore, it is possible to transmit power by exchanging the relationship between power transmission and power reception.
　例えば、負荷Ｒｏが充電電池や蓄電キャパシタンスであったり、その充放電制御回路を含む回路であったりする場合、送電コイルＬｐから受電コイルＬｓへ電力伝送されることにより、前記充電電池は充電される。そして、図１５において入力電源Ｖｉが接続される部分に負荷回路が接続されれば、前記充電電池や蓄電キャパシタンスを入力電源とし、受電コイルＬｓから送電コイルＬｐへ電力が伝送される。 For example, when the load Ro is a rechargeable battery, a storage capacitance, or a circuit including its charge / discharge control circuit, the rechargeable battery is charged by transmitting power from the power transmission coil Lp to the power reception coil Ls. . Then, if a load circuit is connected to a portion to which the input power source Vi is connected in FIG. 15, power is transmitted from the power receiving coil Ls to the power transmitting coil Lp using the rechargeable battery or the storage capacitance as an input power source.
　第１０の実施形態によれば次のような効果を奏する。
（ａ）非常にシンプルな電力伝送システムとして利用できる。 The tenth embodiment has the following effects.
(A) It can be used as a very simple power transmission system.
（ｂ）送電コイルと受電コイルを離して設置することにより、ワイヤレス電力伝送回路システムとして利用できる。 (B) By installing the power transmission coil and the power reception coil apart, it can be used as a wireless power transmission circuit system.
（ｃ）送信側と受信側と入れ替えることにより、双方向の電力伝送回路システムとして利用することができる。 (C) By replacing the transmission side and the reception side, it can be used as a bidirectional power transmission circuit system.
《第１１の実施形態》
　図１６は第１１の実施形態の電力送電システムとして用いられるスイッチング電源装置１１１の回路図である。 << Eleventh Embodiment >>
FIG. 16 is a circuit diagram of a switching power supply device 111 used as the power transmission system of the eleventh embodiment.
　第１０の実施形態で図１５に示した回路と異なり、送電装置側に並列キャパシタＣｐ、受電装置側に並列キャパシタＣｓを設けている。このように部品としての並列キャパシタＣｐ，Ｃｓを設けることにより、送電装置側の共振周波数と受電装置側の共振周波数をそれぞれ任意に設定することができる。したがって、最適化が容易となる。 Unlike the circuit shown in FIG. 15 in the tenth embodiment, a parallel capacitor Cp is provided on the power transmission device side, and a parallel capacitor Cs is provided on the power reception device side. By providing the parallel capacitors Cp and Cs as components in this way, the resonance frequency on the power transmission device side and the resonance frequency on the power reception device side can be set arbitrarily. Therefore, optimization becomes easy.
《第１２の実施形態》
　図１７は第１２の実施形態の電力送電システムとして用いられるスイッチング電源装置１１２の回路図である。 << Twelfth Embodiment >>
FIG. 17 is a circuit diagram of the switching power supply device 112 used as the power transmission system of the twelfth embodiment.
　第１０の実施形態で図１５に示した回路と異なり、共振キャパシタＣｒ，Ｃｒｓを共振キャパシタＣｒ１とＣｒ２，Ｃｒｓ１とＣｒｓ２とにそれぞれ分割して構成した例である。この構成により、キャパシタＣｒ，Ｃｒｓそれぞれに流れる電流がそれぞれ２つのキャパシタに分割されるのでキャパシタでの損失を分散することができ、発熱も分散される。 Unlike the circuit shown in FIG. 15 in the tenth embodiment, the resonance capacitors Cr and Crs are divided into resonance capacitors Cr1 and Cr2, Crs1 and Crs2, respectively. With this configuration, the current flowing through each of the capacitors Cr and Crs is divided into two capacitors, so that losses in the capacitors can be dispersed and heat generation is also dispersed.
　なお、キャパシタＣｒ１，Ｃｒ２およびキャパシタＣｒｓ１，Ｃｒｓ２は、直流電圧を保持する作用と直列共振用キャパシタとしての作用の両方の役割を果たす。 The capacitors Cr1 and Cr2 and the capacitors Crs1 and Crs2 play both roles of holding a DC voltage and acting as a series resonance capacitor.
《第１３の実施形態》
　図１８は第１３の実施形態の電力送電システムとして用いられるスイッチング電源装置１１３の回路図である。この例では４つのスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６によるフルブリッジ回路で送電装置側の交流電圧発生回路を構成している。また、４つのスイッチング素子Ｑ３，Ｑ４，Ｑ７，Ｑ８によるブリッジ整流回路で受電装置側の側整流回路を構成している。図１８に示すパラメータＭは、相互インダクタンスによる磁界結合と相互キャパシタンスによる電界結合との合成による電磁界結合の相互係数として示している。 << Thirteenth embodiment >>
FIG. 18 is a circuit diagram of the switching power supply device 113 used as the power transmission system of the thirteenth embodiment. In this example, an AC voltage generation circuit on the power transmission device side is configured by a full bridge circuit including four switching elements Q1, Q2, Q5, and Q6. Further, a side rectifier circuit on the power receiving device side is configured by a bridge rectifier circuit including four switching elements Q3, Q4, Q7, and Q8. The parameter M shown in FIG. 18 is shown as a mutual coefficient of electromagnetic field coupling by combining magnetic field coupling by mutual inductance and electric field coupling by mutual capacitance.
　この第１３の実施形態によれば、第１０～第１２の実施形態に比べて、送電装置側のスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６、および受電装置側のスイッチング素子Ｑ３，Ｑ４，Ｑ７，Ｑ８に印加される電圧がそれぞれ半分となるため、スイッチング素子での損失を低減できる。 According to the thirteenth embodiment, compared to the tenth to twelfth embodiments, the switching elements Q1, Q2, Q5, Q6 on the power transmission device side, and the switching elements Q3, Q4, Q7, Q8 on the power reception device side. Since the voltage applied to each becomes half, the loss in the switching element can be reduced.
《第１４の実施形態》
　図１９は第１４の実施形態の電力送電システムとして用いられるスイッチング電源装置１１４の回路図である。この例では、送電装置側にキャパシタＣｃを設けて電圧クランプ回路を構成している。その他は第１０の実施形態で図１５に示したものと同様である。 << Fourteenth embodiment >>
FIG. 19 is a circuit diagram of the switching power supply device 114 used as the power transmission system of the fourteenth embodiment. In this example, a capacitor Cc is provided on the power transmission device side to constitute a voltage clamp circuit. Others are the same as those of the tenth embodiment shown in FIG.
　この第１４の実施形態によれば、キャパシタＣｃにチャージされる負電圧を－Ｖｃとすれば、送電装置側で発生される方形波電圧は＋Ｖｉと－Ｖｃの電圧振幅となるので、送信直流電圧の変動に対する制御特性が改善される。 According to the fourteenth embodiment, if the negative voltage charged to the capacitor Cc is −Vc, the square wave voltage generated on the power transmission device side has voltage amplitudes of + Vi and −Vc. The control characteristic with respect to the fluctuation of the is improved.
《第１５の実施形態》
　図２０は第１５の実施形態の電力送電システムとして用いられるスイッチング電源装置１１５の回路図である。この例では、受電装置側にキャパシタＣｃｓを設けて２次側にも電圧クランプ回路を構成している。その他は第１４の実施形態で図１９に示したものと同様である。 << 15th Embodiment >>
FIG. 20 is a circuit diagram of the switching power supply device 115 used as the power transmission system of the fifteenth embodiment. In this example, a capacitor Ccs is provided on the power receiving device side, and a voltage clamp circuit is configured on the secondary side. Others are the same as those of the fourteenth embodiment shown in FIG.
　この例では送電装置側で入力電圧Ｖｉが方形波電圧に変換され、その方形波電圧は＋Ｖｉと－Ｖｃの電圧振幅となる。また、受電装置側のキャパシタＣｃｓに負電圧（Ｖｃｓ）がチャージされるので、スイッチング素子Ｑ３，Ｑ４による同期整流回路に印加される交流方形波電圧は＋Ｖｏと－Ｖｃｓの電圧振幅となる。このように電圧振幅が大きくなるので、出力電圧の変動に対する制御特性も改善される。すなわち出力電圧の調整が広範囲に亘って容易となる。 In this example, the input voltage Vi is converted into a square wave voltage on the power transmission device side, and the square wave voltage has voltage amplitudes of + Vi and −Vc. Further, since the negative voltage (Vcs) is charged in the capacitor Ccs on the power receiving device side, the AC square wave voltage applied to the synchronous rectifier circuit by the switching elements Q3 and Q4 has voltage amplitudes of + Vo and −Vcs. Since the voltage amplitude is increased in this way, the control characteristics with respect to fluctuations in the output voltage are also improved. That is, the output voltage can be easily adjusted over a wide range.
《第１６の実施形態》
　図２１は第１６の実施形態の電力送電システムとして用いられるスイッチング電源装置１１６の回路図である。この例では、受電装置側の整流回路を整流ダイオードＤ３，Ｄ４で構成している。この構成によれば、受電装置は簡素な構成で単方向の電力伝送システムとして用いることができる。 << Sixteenth Embodiment >>
FIG. 21 is a circuit diagram of the switching power supply 116 used as the power transmission system of the sixteenth embodiment. In this example, the rectifier circuit on the power receiving device side is constituted by rectifier diodes D3 and D4. According to this configuration, the power receiving device can be used as a unidirectional power transmission system with a simple configuration.
《第１７の実施形態》
　図２２は第１７の実施形態の電力送電システムとして用いられるスイッチング電源装置１１７の回路図である。この例では、４つのスイッチング素子Ｑ１，Ｑ２，Ｑ５，Ｑ６によるフルブリッジ回路で送電装置側の交流電圧発生回路を構成している。また、受電装置側の整流回路を整流ダイオードＤ３，Ｄ４，Ｄ７，Ｄ８によるダイオードブリッジで構成している。 << Seventeenth Embodiment >>
FIG. 22 is a circuit diagram of a switching power supply device 117 used as the power transmission system of the seventeenth embodiment. In this example, an AC voltage generation circuit on the power transmission device side is configured by a full bridge circuit including four switching elements Q1, Q2, Q5, and Q6. Further, the rectifier circuit on the power receiving device side is constituted by a diode bridge formed by rectifier diodes D3, D4, D7, and D8.
　第１７の実施形態によれば、単方向の電力伝送システムとして用いることができる。また、整流ダイオードの耐電圧が半分で済むこととなる。 According to the seventeenth embodiment, it can be used as a unidirectional power transmission system. In addition, the withstand voltage of the rectifier diode can be halved.
《第１８の実施形態》
　図２３は第１８の実施形態の電力送電システムとして用いられるスイッチング電源装置１１８の回路図である。この例では、コイルのインダクタＬｐ，Ｌｓ間の磁界結合となる相互インダクタンスをＭｌ、キャパシタＣｐ，Ｃｓ間の電界結合となる相互キャパシタンスをＭｃ、キャパシタＣｒ，Ｃｒｓ間の電界結合となる相互キャパシタンスをＭｃｒとしてそれぞれ示している。ここでは、キャパシタＣｒ，Ｃｒｓを含めて電磁界結合回路９０を構成する実施例を図示している。 << Eighteenth embodiment >>
FIG. 23 is a circuit diagram of a switching power supply device 118 used as the power transmission system of the eighteenth embodiment. In this example, Ml is the mutual inductance that is the magnetic field coupling between the inductors Lp and Ls of the coil, Mc is the mutual capacitance that is the electric field coupling between the capacitors Cp and Cs, and Mcr is the mutual capacitance that is the electric field coupling between the capacitors Cr and Crs. As shown respectively. Here, an embodiment in which the electromagnetic field coupling circuit 90 including the capacitors Cr and Crs is illustrated.
　第１８の実施形態の構成によれば、相互インダクタンスＭｌ、相互キャパシタンスＭｃ、相互キャパシタンスＭｃｒを適切に設定することにより電磁界共鳴回路を構成し、電磁界結合による高効率な電力伝送を行なうことができる。 According to the configuration of the eighteenth embodiment, it is possible to configure an electromagnetic resonance circuit by appropriately setting the mutual inductance Ml, the mutual capacitance Mc, and the mutual capacitance Mcr, and perform high-efficiency power transmission by electromagnetic coupling. it can.
Ｃｏ…平滑キャパシタ
Ｃｐ，Ｃｓ…並列共振キャパシタ
Ｃｍ…相互キャパシタンス
Ｃｒ，Ｃｒｓ…直列共振キャパシタ
Ｃｒ１，Ｃｒ２…共振キャパシタ
Ｃｒｓ…直列共振キャパシタ
Ｃｒｓ１，Ｃｒｓ２…キャパシタ
Ｄ３，Ｄ４，Ｄ７，Ｄ８…整流ダイオード
Ｄｓ１，Ｄｓ２…ダイオード
ｉｍ…励磁電流
Ｌｐ…送電コイル
Ｌｍ、Ｌｍｓ…励磁インダクタンス、または相互インダクタンス
Ｌｓ…受電コイル
Ｌｒ，Ｌｒｓ…直列共振インダクタ
Ｍｃ…電界結合の相互係数
Ｍｃｒ…電界結合の相互係数
Ｍｌ…磁界結合の相互係数
ｎｐ…１次巻線
ｎｓ…２次巻線
Ｑ１，Ｑ２，Ｑ３，Ｑ４…スイッチング素子
Ｑ５，Ｑ６，Ｑ７，Ｑ８…スイッチング素子
Ｓ１，Ｓ２，Ｓ３，Ｓ４…スイッチング回路
Ｓ５，Ｓ６，Ｓ７，Ｓ８…スイッチング回路
１０，２０…スイッチング制御回路
３０…絶縁回路
４０…複共振回路
９０…電磁界結合回路
１０１～１１８…スイッチング電源装置 Co ... smoothing capacitors Cp, Cs ... parallel resonance capacitors Cm ... mutual capacitance Cr, Crs ... series resonance capacitors Cr1, Cr2 ... resonance capacitors Crs ... series resonance capacitors Crs1, Crs2 ... capacitors D3, D4, D7, D8 ... rectifier diodes Ds1, Ds2 ... Diode im ... Excitation current Lp ... Transmission coil Lm, Lms ... Excitation inductance or mutual inductance Ls ... Receiving coils Lr, Lrs ... Series resonant inductor Mc ... Electric coupling mutual coefficient Mcr ... Electric coupling mutual coefficient Ml ... Magnetic coupling Reciprocal coefficient np ... primary winding ns ... secondary windings Q1, Q2, Q3, Q4 ... switching elements Q5, Q6, Q7, Q8 ... switching elements S1, S2, S3, S4 ... switching circuits S5, S6, S7 , S8 ... switching circuits 10, 20 ... Switching control circuit 30 ... insulating circuit 40 ... multi-resonant circuit 90 ... electromagnetic coupling circuits 101-118 ... switching power supply device

Claims (15)

  1. 　１次巻線および２次巻線を備える電磁界結合回路と、
    　前記１次巻線に接続されたスイッチング回路を備え、該スイッチング回路をスイッチング素子、ダイオード、およびキャパシタの並列接続回路で構成して、入力される直流電圧から交流電圧を発生する１次側交流電圧発生回路と、
    　前記交流電圧を直流電圧に整流する２次側整流回路と、
    　１次側に構成され、第１の直列共振インダクタおよび第１の直列共振キャパシタを含む第１の共振回路と、
    　２次側に構成され、第２の直列共振インダクタおよび第２の直列共振キャパシタを含む第２の共振回路と、
    　前記１次側交流電圧発生回路のスイッチング素子をデッドタイムを挟んで交互にオン／オフすることによりほぼ方形波状または台形波状の交流電圧を発生させるスイッチング制御回路と、
    　を備えたスイッチング電源装置において、
    　前記スイッチング制御回路は、電磁界結合回路を含めた前記第１の共振回路と前記第２の共振回路とを合わせた全体となる複共振回路に流入する電流が、前記１次側交流電圧発生回路から発生する交流電圧よりも遅れる正弦波状の共振電流波形となって、前記スイッチング素子のオン期間およびオフ期間の両期間に前記電磁界結合回路を介して１次側から２次側へ電力が伝送されるように、前記複共振回路に対してインピーダンスが最も小さくなる固有の共振周波数よりも高いスイッチング周波数で前記１次側交流電圧発生回路のスイッチング素子をスイッチング動作し、
    　前記電磁界結合回路は、前記１次巻線と前記２次巻線との間で相互インダクタンスを介した磁界結合および相互キャパシタンスを介した電界結合とが混合した電磁界共鳴回路を構成し、
    　前記第１の共振回路と前記第２の共振回路とが共鳴して前記電磁界結合回路の１次側から２次側へ電力が伝送されることを特徴とするスイッチング電源装置。     An electromagnetic coupling circuit comprising a primary winding and a secondary winding;
    A primary side AC voltage that includes a switching circuit connected to the primary winding and that is configured by a parallel connection circuit of a switching element, a diode, and a capacitor, and generates an AC voltage from an input DC voltage Generating circuit;
    A secondary side rectifier circuit for rectifying the AC voltage into a DC voltage;
    A first resonant circuit configured on the primary side and including a first series resonant inductor and a first series resonant capacitor;
    A second resonant circuit configured on the secondary side and including a second series resonant inductor and a second series resonant capacitor;
    A switching control circuit for generating a substantially square wave or trapezoidal AC voltage by alternately turning on / off switching elements of the primary side AC voltage generation circuit with a dead time therebetween;
    In a switching power supply device comprising:
    In the switching control circuit, a current flowing into a multi-resonance circuit including the first resonance circuit including the electromagnetic field coupling circuit and the second resonance circuit is converted into the primary AC voltage generation circuit. A sinusoidal resonance current waveform that is delayed from the AC voltage generated from the power source, and power is transmitted from the primary side to the secondary side via the electromagnetic field coupling circuit during both the on period and the off period of the switching element. As described above, the switching operation of the switching element of the primary side AC voltage generating circuit is performed at a switching frequency higher than a specific resonance frequency at which the impedance is smallest with respect to the double resonance circuit,
    The electromagnetic field coupling circuit constitutes an electromagnetic field resonance circuit in which a magnetic field coupling via a mutual inductance and an electric field coupling via a mutual capacitance are mixed between the primary winding and the secondary winding,
    A switching power supply device, wherein the first resonance circuit and the second resonance circuit resonate to transmit power from a primary side to a secondary side of the electromagnetic field coupling circuit.
  2. 　前記スイッチング制御回路は、前記１次側交流電圧発生回路のスイッチング周波数を一定にし、前記スイッチング回路に電流が導通する期間をオン期間、その他の期間をオフ期間として、複数のスイッチング回路のオン期間比率を制御することで、前記２次側整流回路から得られる出力電力を調整する、請求項１に記載のスイッチング電源装置。 The switching control circuit is configured such that a switching frequency of the primary AC voltage generating circuit is constant, a period in which a current is conducted to the switching circuit is an on period, and another period is an off period. The switching power supply according to claim 1, wherein the output power obtained from the secondary side rectifier circuit is adjusted by controlling the output power.
  3. 　前記スイッチング制御回路は、前記１次側交流電圧発生回路のスイッチング周波数を変化させて前記スイッチング素子のオン期間比率を制御することで、前記２次側整流回路から得られる出力電力を調整する、請求項１に記載のスイッチング電源装置。 The switching control circuit adjusts output power obtained from the secondary side rectifier circuit by changing an on period ratio of the switching element by changing a switching frequency of the primary side AC voltage generation circuit. Item 4. The switching power supply device according to Item 1.
  4. 　前記２次側整流回路は、オン期間またはオフ期間のいずれか、または両期間に、前記２次巻線に発生する電圧を静電エネルギーとして前記第２の共振キャパシタに蓄えて、前記オン期間とオフ期間のそれぞれの期間に前記２次巻線に発生する電圧を加算して直流電圧として出力する、請求項１～３のいずれかに記載のスイッチング電源装置。 The secondary-side rectifier circuit stores the voltage generated in the secondary winding as electrostatic energy in the second resonance capacitor in either the on period or the off period, or in both periods, The switching power supply device according to any one of claims 1 to 3, wherein a voltage generated in the secondary winding is added during each of the off periods and output as a DC voltage.
  5. 　前記第１の直列共振キャパシタと前記第２の直列共振キャパシタのいずれかまたは両方は直流電圧を保持する、請求項１～４のいずれかに記載のスイッチング電源装置。 5. The switching power supply device according to claim 1, wherein either or both of the first series resonant capacitor and the second series resonant capacitor hold a DC voltage.
  6. 　前記１次巻線または前記２次巻線に対して並列に並列共振キャパシタを備えた、請求項１～５のいずれかに記載のスイッチング電源装置。 The switching power supply device according to any one of claims 1 to 5, further comprising a parallel resonant capacitor in parallel with the primary winding or the secondary winding.
  7. 　前記並列共振キャパシタを前記１次巻線または前記２次巻線の浮遊容量で構成した、請求項６に記載のスイッチング電源装置。 The switching power supply device according to claim 6, wherein the parallel resonant capacitor is configured by a stray capacitance of the primary winding or the secondary winding.
  8. 　前記相互キャパシタンスを前記１次巻線と前記２次巻線との間に形成される浮遊容量で構成した、請求項１～７のいずれかに記載のスイッチング電源装置。 The switching power supply device according to any one of claims 1 to 7, wherein the mutual capacitance is configured by a stray capacitance formed between the primary winding and the secondary winding.
  9. 　前記第１の直列共振インダクタまたは前記第２の直列共振インダクタを前記電磁界結合回路の漏れインダクタンスで構成した、請求項１～８のいずれかに記載のスイッチング電源装置。 9. The switching power supply device according to claim 1, wherein the first series resonant inductor or the second series resonant inductor is configured by a leakage inductance of the electromagnetic field coupling circuit.
  10. 　前記相互インダクタンスを前記１次巻線と前記２次巻線との間に等価的に形成される励磁インダクタンスで構成した、請求項１～９のいずれかに記載のスイッチング電源装置。 The switching power supply device according to any one of claims 1 to 9, wherein the mutual inductance is configured by an excitation inductance formed equivalently between the primary winding and the secondary winding.
  11. 　前記スイッチング回路はMOSFETである、請求項１～１０のいずれかに記載のスイッチング電源装置。 The switching power supply device according to any one of claims 1 to 10, wherein the switching circuit is a MOSFET.
  12. 　前記２次側整流回路に備えられる前記交流電圧を直流電圧に整流する整流素子はMOSFETである、請求項１～１１のいずれかに記載のスイッチング電源装置。 12. The switching power supply device according to claim 1, wherein the rectifying element that rectifies the AC voltage provided in the secondary side rectifier circuit into a DC voltage is a MOSFET.
  13. 　前記２次側整流回路の出力部から電力が伝送されるとき、前記２次側整流回路は前記１次側交流電圧発生回路として作用するとともに、前記１次側交流電圧発生回路は前記２次側整流回路として作用し、
    　双方向に電力伝送が可能な、請求項１２に記載のスイッチング電源装置。     When power is transmitted from the output of the secondary side rectifier circuit, the secondary side rectifier circuit acts as the primary side AC voltage generation circuit, and the primary side AC voltage generation circuit is the secondary side rectifier circuit. Acts as a rectifier circuit,
    The switching power supply device according to claim 12, capable of bidirectional power transmission.
  14. 　前記１次巻線は磁芯を有するトランスの１次側に設けられた巻線、前記２次巻線は前記トランスの２次側に設けられた巻線である、請求項１～１３のいずれかに記載のスイッチング電源装置。 The primary winding is a winding provided on the primary side of a transformer having a magnetic core, and the secondary winding is a winding provided on the secondary side of the transformer. A switching power supply device according to claim 1.
  15. 　前記１次巻線は送電装置に設けられた送電コイル、前記２次巻線は前記送電装置に向けて配置される受電装置に設けられた受電コイルである、請求項１～１３のいずれかに記載のスイッチング電源装置。 The primary winding is a power transmission coil provided in a power transmission device, and the secondary winding is a power reception coil provided in a power reception device arranged toward the power transmission device. The switching power supply device described.
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US9106141B2 (en) 2015-08-11
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KR20130114693A (en) 2013-10-17
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US20130301308A1 (en) 2013-11-14
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