WO2010097034A1

WO2010097034A1 - Interference suppression method and communication device

Info

Publication number: WO2010097034A1
Application number: PCT/CN2010/070717
Authority: WO
Inventors: 李晔; 王霞; 秦龙; 胡宏杰; 宋安东尼
Original assignee: 华为技术有限公司
Priority date: 2009-02-27
Filing date: 2010-02-23
Publication date: 2010-09-02
Also published as: CN101557377A; CN101557377B

Abstract

The present invention provides an interference suppression method and communication device, wherein the method includes: obtaining the total length of a channel response and a partial channel response; obtaining pre-filtering coefficients according to the total length of the channel response and the partial channel response; performing pre-filtering processing for time-domain signals according to the pre-filtering coefficients; wherein the total length of the channel response is greater than the length of the cyclic prefix; the partial channel response includes a first partial channel response and/or a second partial channel response, the first partial channel response is the part of the partial channel response the length of which does not exceed the length of the cyclic prefix, and the second partial channel response is the part of the partial channel response the length of which exceeds the length of the cyclic prefix. The present invention performs the pre-filtering processing similar to the channel shortening equalization for the time-domain signals awaiting transmitting, so that the energy of the equivalent channel response in the pre-filtered time-domain signals is concentrated in the range shorter than the length of the cyclic prefix as far as possible, thus enabling the prominent reduction of the complexity for implementing the interference suppression required by the receiving end, and leading to good interference suppression effect.

Description

干扰抑制方法和通信设备本申请要求于 2009年 2月 27 日提交中国专利局、申请号为 2009100786 50.7、发明名称为 "预滤波系数计算和干扰抑制方法、装置和***" 的中国专利申请的优先权，其全部内容通过引用结合在本申请中。技术领域 Interference suppression method and communication device The present application claims priority to Chinese patent application filed on February 27, 2009 by the Chinese Patent Office, application number 2009100786 50.7, and entitled "Pre-filter coefficient calculation and interference suppression method, device and system" The entire contents of which are incorporated herein by reference. Technical field

本发明及其实施例涉及无线宽带通信领域，特别是涉及一种干扰抑制方法和通信设备。背景技术 The present invention and its embodiments relate to the field of wireless broadband communications, and in particular, to an interference suppression method and a communication device. Background technique

正交频分复用 ( Orthogonal Frequency Division Multiplexing, 简称 OFDM ) ***或釆用频域均衡技术的单载波*** ( single carrier system wi th frequency domain equalization, 简称 SC-FDE) , 通常通过***循环前缀（Cyclic Prefix, 简称 CP) 来消除无线信道多径传播带来的符号间干扰 ( Inter Symbol Interference, 简称 ISI ) 。当 CP的长度大于或等于信道时延扩展长度时，理论上可以完全消除由于无线信道多径传播效应产生的符号间干 4尤（Inter Symbol Interference, 简称 ISI ) , 保持频域子载波间的正交性，这意味着在理想同步的前提下接收端将不存在块间干扰（ Inter B1 ock Interference, 简称 IBI )和载波间干扰 ( Inter Carrier Interf erenc e, 简称 ICI ) 。但是，当信道时延扩展长度大于 CP长度，导致通常所称的 "CP不足" 时，即便釆取了相应的符号同步技术以减少干扰，但 IBI和 ICI 还是难以避免，一般会通过在接收机中釆用干扰抑制技术来降低 IBI 和 ICI 的影响，保证接收性能。 Orthogonal Frequency Division Multiplexing (OFDM) system or single carrier system wi-frequency domain equalization (SC-FDE), usually by inserting a cyclic prefix (Cyclic) Prefix (CP for short) eliminates Inter Symbol Interference (ISI) caused by multipath propagation in wireless channels. When the length of the CP is greater than or equal to the channel delay spread length, it is theoretically possible to completely eliminate the Inter Symbol Interference (ISI) due to the multipath propagation effect of the wireless channel, and maintain the positive between the frequency domain subcarriers. Intercommunication, which means that there will be no Inter-Impact Interference (IBI) and Inter-Carrier Interference (ICI) at the receiving end under the premise of ideal synchronization. However, when the channel delay spread length is greater than the CP length, which is commonly referred to as "CP deficiency", even if the corresponding symbol synchronization technique is adopted to reduce interference, IBI and ICI are still difficult to avoid, and generally pass through the receiver. The medium uses interference suppression technology to reduce the impact of IBI and ICI to ensure reception performance.

现有一类典型的 IBI抑制技术是所谓的残留符号间干扰消除（Residual ISI cance l la t ion, 简称 RISIC ) 及其类似方法，其基本思路是通过基于解调块判决反馈或译码反馈，来重构前一个符号块对当前符号块的 IBI 以及当前符号块缺失的 CP部分，然后在 FFT之前在时域对这两者进行补偿。该技术需要基于迭代来进行干扰消除，而众所周知，迭代接收导致的额外处理时延和计算量将大大增加接收机的复杂度。 A typical class of IBI suppression techniques is the so-called residual intersymbol interference cancellation (Residual The basic idea of the ISI cance l ray, which is RISIC for short, is to reconstruct the IBI of the current symbol block and the missing of the current symbol block by demodulating block decision feedback or decoding feedback. The CP part then compensates for both in the time domain before the FFT. This technique requires interference cancellation based on iterations, and it is well known that the extra processing delay and computational complexity caused by iterative reception will greatly increase the complexity of the receiver.

现有另外一类典型的 IBI抑制技术是在信号接收端对时域接收信号进行信道缩短 ( Channel Shorten ) 均衡，使得等效信道 ( Equiva lent Channel ) 响应的能量集中到 CP长度范围内，从而来最小化 IBI和 ICI影响，保证频域子载波的正交性；等效信道为实际信道和信道缩短均衡器的级联***响应。 Another typical IBI suppression technique is to perform channel shorten equalization on the time domain received signal at the signal receiving end, so that the energy of the equivalent channel (Equivallent Channel) response is concentrated within the CP length range. Minimize the impact of IBI and ICI to ensure the orthogonality of the frequency domain subcarriers; the equivalent channel is the cascaded system response of the actual channel and channel shortening equalizer.

由于宽带***中时域信道响应抽头数 4艮多，相应地将导致时域信道缩短均衡系数求取中需要大矩阵求逆或分解，运算复杂度很高，基于当前应用较为广泛的接收机现有能力，实现难度大，不利于推广使用。发明内容 Since the number of time domain channel response taps in the wideband system is more than 4, correspondingly, the time domain channel shortening equalization coefficient is required to obtain large matrix inversion or decomposition, and the operation complexity is high. Based on the current widely used receivers, The ability to achieve difficulty is not conducive to promotion. Summary of the invention

本发明实施例提供了一种干扰抑制方法和通信设备，用以在循环前缀的长度小于信道响应总长度情形下，降低接收端抑制干扰的运算复杂度。 The embodiment of the present invention provides an interference suppression method and a communication device, which are used to reduce the computational complexity of suppressing interference at the receiving end when the length of the cyclic prefix is less than the total length of the channel response.

本发明实施例提供了一种干扰抑制方法，包括： An embodiment of the present invention provides an interference suppression method, including:

获取信道响应总长度和部分信道响应； Obtaining the total length of the channel response and part of the channel response;

根据所述信道响应总长度和所述部分信道响应获取预滤波系数；根据所述预滤波系数，对时域信号进行预滤波处理； Obtaining a pre-filtering coefficient according to the total length of the channel response and the partial channel response; performing pre-filtering processing on the time domain signal according to the pre-filtering coefficient;

其中，所述信道响应总长度大于循环前缀长度；所述部分信道响应包括：第一部分信道响应和 /或第二部分信道响应；所述第一部分信道响应为信道没有超出循环前缀长度部分的信道响应，所述第二部分信道响应为信道超出循环前缀长度部分的信道响应。 The channel response total length is greater than a cyclic prefix length; the partial channel response includes: a first partial channel response and/or a second partial channel response; and the first partial channel response is a channel response in which the channel does not exceed a cyclic prefix length portion The second partial channel response is a channel response in which the channel exceeds a cyclic prefix length portion.

本发明实施例还提供了一种通信设备，包括： The embodiment of the invention further provides a communication device, including:

信道信息获取模块，获取信道响应总长度和部分信道响应；预滤波系数计算模块，根据所述信道响应总长度和所述部分信道响应，获取预滤波系数； a channel information acquisition module, which acquires a total channel response length and a partial channel response; a pre-filter coefficient calculation module, which acquires a pre-filter coefficient according to the total length of the channel response and the partial channel response;

预滤波处理模块，根据所述预滤波系数，对时域信号进行预滤波处理；其中，所述信道响应总长度大于循环前缀长度；所述部分信道响应包括：第一部分信道响应和 /或第二部分信道响应；所述第一部分信道响应为信道没有超出循环前缀长度部分的信道响应，所述第二部分信道响应为信道超出循环前缀长度部分的信道响应。 The pre-filtering processing module performs pre-filtering processing on the time domain signal according to the pre-filtering coefficient; wherein, the total length of the channel response is greater than a cyclic prefix length; and the partial channel response includes: a first part of the channel response and/or the second Partial channel response; the first partial channel response is a channel response in which the channel does not exceed a cyclic prefix length portion, and the second partial channel response is a channel response in which the channel exceeds a cyclic prefix length portion.

本发明实施例可应用但不限于 OFDM或 SC-FDE ***中，用以获取预滤波系数，以便发送端或接收端根据预滤波系数对接收或发送的信号进行干扰消除；基于该预滤波系数对时域信号进行类似信道缩短均衡的预滤波处理，使得预滤波处理后的时域信号中，等效信道响应的能量尽量集中在短于循环前缀长度的范围内，相对于背景技术中的现有干扰抑制技术而言，能够显著降低接收端为了抑制干扰所需的实现复杂度，且具有很好的干扰抑制效果。基于本发明实施例的接收机实现原理非常简单，信号接收性能也达到或好于现有技术。附图说明 The embodiment of the present invention may be applied to, but not limited to, an OFDM or SC-FDE system, to obtain a pre-filtering coefficient, so that the transmitting end or the receiving end performs interference cancellation on the received or transmitted signal according to the pre-filtering coefficient; The time domain signal performs pre-filtering processing similar to channel shortening equalization, so that the energy of the equivalent channel response in the pre-filtered time domain signal is concentrated as much as possible within a range shorter than the cyclic prefix length, compared to existing ones in the background art. The interference suppression technology can significantly reduce the implementation complexity required for the receiver to suppress interference, and has a good interference suppression effect. The receiver implementation principle based on the embodiment of the present invention is very simple, and the signal receiving performance is also achieved or better than the prior art. DRAWINGS

为了更清楚地说明本发明实施例或现有技术中的技术方案，下面将对实施例或现有技术描述中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动性的前提下，还可以根据这些附图获得其他的附图。 In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings used in the embodiments or the description of the prior art will be briefly described below. Obviously, the drawings in the following description are only It is a certain embodiment of the present invention, and other drawings can be obtained from those skilled in the art without any inventive labor.

图 1为本发明一实施例提供的干扰抑制方法流程图； FIG. 1 is a flowchart of an interference suppression method according to an embodiment of the present invention;

图 2为本发明另一实施例提供的干扰抑制方法流程图； 2 is a flowchart of an interference suppression method according to another embodiment of the present invention;

图 3为本发明另一实施例提供的干扰抑制方法在 OF丽***应用场景的实现框图； 3 is a block diagram of an implementation scenario of an interference suppression method in an OF system according to another embodiment of the present invention;

图 4为本发明另一实施例提供的一种 OFDM***帧结构示意图；图 5为本发明另一实施例提供的预滤波系数获取方法流程图；图 6为本发明实施例干扰抑制方法和现有干扰抑制方法的性能比较示意图一； FIG. 4 is a schematic structural diagram of a frame of an OFDM system according to another embodiment of the present invention; FIG. FIG. 5 is a flowchart of a method for acquiring a pre-filtering coefficient according to another embodiment of the present invention; FIG. 6 is a schematic diagram 1 for comparing performance of an interference suppression method and an existing interference suppression method according to an embodiment of the present invention;

图 7为本发明实施例干扰抑制方法和其他干扰抑制方法的性能比较示意图二； FIG. 7 is a schematic diagram of performance comparison of an interference suppression method and other interference suppression methods according to an embodiment of the present invention; FIG.

图 8为本发明另一实施例提供的发射机结构图； FIG. 8 is a structural diagram of a transmitter according to another embodiment of the present invention; FIG.

图 9为本发明另一实施例提供的接收机结构图； FIG. 9 is a structural diagram of a receiver according to another embodiment of the present invention;

图 10为本发明另一实施例提供的干扰抑制***结构图； FIG. 10 is a structural diagram of an interference suppression system according to another embodiment of the present invention;

图 11为本发明另一实施例提供的预滤波系数计算装置结构图。具体实施方式 FIG. 11 is a structural diagram of a pre-filter coefficient calculation apparatus according to another embodiment of the present invention. detailed description

下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有作出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。 The technical solutions in the embodiments of the present invention are clearly and completely described in the following with reference to the accompanying drawings in the embodiments of the present invention. It is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments obtained by a person of ordinary skill in the art based on the embodiments of the present invention without creative efforts are within the scope of the present invention.

本发明实施例可应用于正交频分复用***、釆用频域均衡的单载波*** 或釆用符号块进行信号发送的其它宽带***中进行干扰抑制。对于实际信道的信道响应总长度大于循环前缀长度的情形，通过在发送端对待发送的时域信号进行类似信道缩短均衡的预滤波处理，使得接收端接收到时域信号中，等效信道响应的能量尽量集中在短于循环前缀长度的范围内，从而最小化干扰，保证频域子载波的正交性。本发明实施例使得接收端无需进行复杂的干扰抑制处理，仅需要釆用通常的简单的单抽头均衡来完成接收处理，因而能够显著降低接收端为了抑制干扰所需的实现复杂度，且具有很好的干扰抑制效果。 Embodiments of the present invention can be applied to an Orthogonal Frequency Division Multiplexing system, a single carrier system using frequency domain equalization, or interference suppression in other wideband systems that use a symbol block for signal transmission. For the case where the total channel response length of the actual channel is greater than the cyclic prefix length, the pre-filtering process similar to channel shortening equalization is performed on the time domain signal to be transmitted at the transmitting end, so that the receiving end receives the time domain signal, and the equivalent channel response The energy is concentrated as much as possible within the range of the cyclic prefix length, thereby minimizing interference and ensuring the orthogonality of the frequency domain subcarriers. In the embodiment of the present invention, the receiving end does not need to perform complex interference suppression processing, and only needs to use the usual simple one-tap equalization to complete the receiving process, thereby significantly reducing the implementation complexity required by the receiving end to suppress interference, and has a very high Good interference suppression effect.

图 1为本发明一实施例提供的干扰抑制方法流程图。本实施例从发送端对本发明实施例的技术方案进行说明。如图 1所示，本实施例包括：步骤 11、根据接收端的反馈获得部分信道信息； FIG. 1 is a flowchart of an interference suppression method according to an embodiment of the present invention. This embodiment describes the technical solution of the embodiment of the present invention from the transmitting end. As shown in FIG. 1, this embodiment includes: Step 11. Obtain partial channel information according to feedback of the receiving end.

其中，部分信道信息可以包括：部分信道响应；而部分信道响应可包括但不限于实际信道超出循环前缀（CP ) 长度部分的信道响应。 The partial channel information may include: a partial channel response; and the partial channel response may include, but is not limited to, a channel response in which the actual channel exceeds the cyclic prefix (CP) length portion.

本实施例中，信道响应总长度可大于循环前缀长度。发送端获取信道响应总长度的方式可包括：发送端预先估算获得信道响应总长度；或者，接收端获取信道响应总长度，并将信道响应总长度反馈给发送端，该情形下，接收端向发送端反馈的部分信道信息中还可包括信道响应总长度。 In this embodiment, the total length of the channel response may be greater than the cyclic prefix length. The manner in which the sending end obtains the total length of the channel response may include: the transmitting end pre-estimates the total length of the channel response; or the receiving end acquires the total length of the channel response, and feeds back the total length of the channel response to the transmitting end. In this case, the receiving end The total channel response length may also be included in the partial channel information fed back by the transmitting end.

由于本实施例在发送端进行类似信道压缩均衡 ( Channel Shortening Equa l izat ion, 简称 CSE ) 的预滤波（Pre-Proces s ing Fi l ter )处理，因此需要接收端向发送端反馈部分信道响应。接收端可将实际信道超出 CP长度部分的响应，和 /或实际信道没有超出 CP长度部分的响应，作为本发明实施例中的部分信道响应，反馈给发送端。 In this embodiment, the pre-filtering (Pre-Proces s Fier ter) processing is performed on the transmitting end, so that the receiving end needs to feed back a partial channel response to the transmitting end. The receiving end may forward the response of the actual channel beyond the length of the CP, and/or the response of the actual channel without exceeding the length of the CP, as part of the channel response in the embodiment of the present invention, and feed back to the transmitting end.

步骤 12、根据获得的部分信道信息，计算预滤波系数。 Step 12: Calculate a pre-filtering coefficient according to the obtained partial channel information.

其中，关于预滤波系数的计算方法可以参考后续图 5对应实施例的相关描述，此处不再赘述。 For the calculation method of the pre-filtering coefficient, reference may be made to the related description of the corresponding embodiment in FIG. 5, and details are not described herein again.

步骤 13、根据得到的预滤波系数，对时域信号（如：时域 OFDM调制信号）进行预滤波处理后发送。 Step 13. Perform pre-filtering processing on the time domain signal (such as a time domain OFDM modulation signal) according to the obtained pre-filtering coefficient.

所述时域信号为发送符号经过调制（如： OFDM调制）并添加了 CP后的时域信号。 The time domain signal is a time domain signal after the transmission symbol is modulated (e.g., OFDM modulation) and the CP is added.

本实施例针对信道响应总长度大于循环前缀（CP )长度的情形，通过发送端在对待发送的 OFDM时域信号进行类似信道缩短均衡的预滤波处理，使得接收端接收到时域信号中，等效信道响应的能量尽量集中在短于 CP长度的范围内，从而保证接收信号所受 IBI和 ICI得到有效抑制，相对于背景技术中的现有干扰抑制技术而言，本实施例降低了接收端为了抑制干扰所需的运算复杂度和处理时延，明显降低了实现的难度。 In this embodiment, in the case that the total channel response length is greater than the cyclic prefix (CP) length, the transmitting end performs pre-filtering processing similar to channel shortening equalization on the OFDM time domain signal to be transmitted, so that the receiving end receives the time domain signal, etc. The energy of the effective channel response is concentrated as much as possible within a range shorter than the length of the CP, thereby ensuring that the IBI and ICI of the received signal are effectively suppressed. Compared with the existing interference suppression technology in the background art, the embodiment reduces the receiving end. In order to suppress the computational complexity and processing delay required for interference, the difficulty of implementation is significantly reduced.

本发明实施例可应用于 OFDM***、或接收端基于频域均衡技术的单载波***、或釆用符号块进行信号发送的其它宽带***中，特别是应用在上述***的下行传输中，可明显降低考虑 CP长度不足导致的干扰抑制时接收端的实现复杂度。 The embodiments of the present invention can be applied to an OFDM system, or a single carrier based on frequency domain equalization technology at the receiving end. In other broadband systems in which the wave system or the signal block is used for signal transmission, especially in the downlink transmission of the above system, the implementation complexity of the receiving end when the interference suppression due to the insufficient CP length is considered can be significantly reduced.

图 2为本发明另一实施例提供的干扰抑制方法流程图。本实施例与图 1 对应实施例的区别在于，本实施例从接收端对本发明实施例的技术方案进行说明。如图 2所示，本实施例包括： FIG. 2 is a flowchart of an interference suppression method according to another embodiment of the present invention. The difference between this embodiment and the corresponding embodiment of FIG. 1 is that the technical solution of the embodiment of the present invention is described from the receiving end. As shown in FIG. 2, this embodiment includes:

步骤 21、进行信道估计以获得时域信道响应。 Step 21. Perform channel estimation to obtain a time domain channel response.

接收端通过信道估计，可获知信道响应和信道响应总长度。 The receiving end can obtain the channel response and the total length of the channel response through channel estimation.

步骤 22、根据所述时域信道响应，获取部分信道信息并向发送端反馈，以供所述发送端根据所述部分信道信息计算预滤波系数，并根据所述预滤波系数对待发送的时域信号进行预滤波处理。 Step 22: Acquire partial channel information according to the time domain channel response, and feed back to the transmitting end, so that the sending end calculates a pre-filtering coefficient according to the partial channel information, and sends a time domain according to the pre-filtering coefficient. The signal is pre-filtered.

其中，部分信道信息包括部分信道响应。而部分信道响应可包括但不限于实际信道超出循环前缀（CP )长度部分的信道响应。 Wherein, part of the channel information includes a partial channel response. The partial channel response may include, but is not limited to, the channel response of the actual channel beyond the cyclic prefix (CP) length portion.

本实施例中，信道响应总长度可大于循环前缀长度。接收端可获取信道响应总长度，并将信道响应总长度反馈给发送端，该情形下，接收端向发送端反馈的部分信道信息中还可包括信道响应总长度。 In this embodiment, the total length of the channel response may be greater than the cyclic prefix length. The receiving end can obtain the total length of the channel response, and feed back the total length of the channel response to the transmitting end. In this case, the partial channel information fed back by the receiving end to the transmitting end may further include the total length of the channel response.

本实施例针对实际信道的信道响应总长度大于循环前缀（CP ) 长度的情形，区别于现有技术的是：通过发送端在对待发送的时域信号进行信道缩短均衡的预滤波处理。由于发送端无法获知在通信过程中时域信号传输的信道信息，也无法对信道响应进行估计，因此，需要接收端将部分信道信息，反馈给发送端，以供发送端根据获取的部分信道信息进行信道缩短均衡的预滤波处理，使得接收端接收到时域信号中，等效信道响应的能量尽量集中在短于 CP长度的范围内，相对于背景技术中的现有干扰抑制技术而言，本实施例降低了接收端为了抑制干扰所需的运算复杂度和处理时延，明显降低了实现的难度。 In this embodiment, the case where the total length of the channel response of the actual channel is greater than the cyclic prefix (CP) length is different from the prior art: the pre-filtering process of the channel shortening equalization is performed by the transmitting end on the time domain signal to be transmitted. Since the transmitting end cannot know the channel information of the time domain signal transmission during the communication process, and can not estimate the channel response, the receiving end needs to feed part of the channel information to the transmitting end, so that the transmitting end can obtain the partial channel information according to the obtained channel information. Pre-filtering processing for channel shortening equalization, so that the energy of the equivalent channel response is concentrated in the range shorter than the CP length in the time domain signal received by the receiving end, compared with the existing interference suppression technology in the background art, This embodiment reduces the computational complexity and processing delay required by the receiving end to suppress interference, and significantly reduces the difficulty of implementation.

图 3为本发明另一实施例提供的干扰抑制方法在 OF丽***应用场景的实现框图。本实施例从发送端和接收端二侧 , 以本发明实施例在 OFDM***应用场景下的实现为例，说明本发明实施例的技术方案。本领域技术人员可以理解，以下技术方案同样适用于釆用频域均衡的单载波***或者其他的系统，图 3中示意的模块或装置稍作改动后，也完全可以用于釆用频域均衡的单载波***或者其他的***。 FIG. 3 is a schematic diagram of an interference suppression method according to another embodiment of the present invention in an application system of an OF system Now the block diagram. In this embodiment, the implementation of the OFDM system in the application scenario of the embodiment of the present invention is taken as an example to describe the technical solution of the embodiment of the present invention. Those skilled in the art can understand that the following technical solutions are equally applicable to a single carrier system or other system that uses frequency domain equalization. The module or device illustrated in FIG. 3 can be used for frequency domain equalization after a slight modification. Single carrier system or other system.

参照图 3的实现框图，本实施例进行干扰抑制的实现方法包括：步骤 301、接收端的信道估计模块 91进行信道估计以获得时域信道响应。接收端根据接收到的信号进行信道估计，获取时域信道响应。 Referring to the implementation block diagram of FIG. 3, the method for implementing interference suppression in this embodiment includes: Step 301: A channel estimation module 91 at a receiving end performs channel estimation to obtain a time domain channel response. The receiving end performs channel estimation according to the received signal to obtain a time domain channel response.

步骤 302、接收端中的反馈信道响应抽取单元 921从获得的时域信道响应中抽取部分信息。 Step 302: The feedback channel response extracting unit 921 in the receiving end extracts part of the information from the obtained time domain channel response.

步骤 303、接收端中的反馈信道响应量化单元 922对抽取的部分信息进行量化处理，并将量化处理后的信息反馈给发送端。 Step 303: The feedback channel response quantization unit 922 in the receiving end quantizes the extracted partial information, and feeds back the quantized information to the transmitting end.

量化处理后的信息可包括：部分信道响应及信道响应总长度。 The quantized information may include: a partial channel response and a total length of the channel response.

量化处理可包括：将抽取的部分信息中的浮点数转化成定点数，以方便接收端向发送端反馈。 The quantization process may include: converting the floating point number in the extracted partial information into a fixed point number, so as to facilitate the receiving end to feed back to the transmitting end.

步骤 304、发送端中的 OFDM调制模块 31对待发送符号进行 OFDM调制，得到时域 OFDM信号。 Step 304: The OFDM modulation module 31 in the transmitting end performs OFDM modulation on the symbol to be transmitted to obtain a time domain OFDM signal.

本步骤得到的时域 0FDM信号为发送符号 kW Jo 二经过 0FDM调制模块 31调制后获得的时域信号。其中， t__∞中的 n为 0FDM符号块序号， k 表示 OFDM符号块子载波符号序号， K表示总的子载波符号数。对待发送符号

0FDM调制，主要包括通常的串并变换、 IFFT (反傅里叶）变换、并串变换， *** CP , 得到时域 OFDM信号 x [n] 。 x [n]表示经过 OFDM调制并*** CP的时域信号，这里的 n仅表示时间意义，与 fcW Jo :二中的 n没有直接的对应关系。 The time domain 0FDM signal obtained in this step is a time domain signal obtained after the transmission symbol kW Jo is modulated by the 0FDM modulation module 31. Wherein, n in t_ _∞ is a 0FDM symbol block number, k represents an OFDM symbol block subcarrier symbol number, and K represents a total number of subcarrier symbols. Treating symbols

0FDM modulation mainly includes normal serial-to-parallel transform, IFFT (anti-Fourier) transform, parallel-to-serial transform, and insertion into CP to obtain time-domain OFDM signal x [n]. x [n] represents a time domain signal modulated by OFDM and inserted into the CP, where n only represents time meaning, and has no direct correspondence with n of fcW Jo : two.

步骤 305、发送端中的预滤波模块 32基于接收端反馈的部分信道信息，计算预滤波系数，根据预滤波系数对待发送的时域 OFDM信号进行预滤波。预¾ 波系数可基于最大信干噪 t匕准贝 ( Maximum S igna l to Interference plus Noi se Ra t io, 简称 MS INR )进行计算，具体实现方法详见图 5对应实施例的记载。 Step 305: The pre-filtering module 32 in the transmitting end is based on partial channel information fed back by the receiving end. The pre-filter coefficient is calculated, and the time-domain OFDM signal to be transmitted is pre-filtered according to the pre-filter coefficient. The pre-synthesis coefficient can be calculated based on the maximum singularity of the singularity of the singularity, and the specific implementation method is as shown in the corresponding embodiment of FIG.

发送端将预滤波处理后的信号通过信道向接收端发送，接收端接收的信号 y [n]中通常叠加有噪声 n [m] 。 The transmitting end sends the pre-filtered signal to the receiving end through the channel, and the signal y [n] received by the receiving end is usually superimposed with noise n [m].

本实施例可釆用的一种便于接收端获取时域信道响应的方法是：发送端和接收端基于某种特定设计的发送信号结构进行通信，接收端根据信号结构中***的前导信息进行信道估计。 A method for facilitating the receiving end to obtain the time domain channel response in the embodiment is: the transmitting end and the receiving end communicate according to a specific designed transmission signal structure, and the receiving end performs channel according to the preamble information inserted in the signal structure. estimate.

图 4为本发明另一实施例提供的一种 OFDM ***帧结构示意图。如图 4 所示的 OFDM***帧结构中，发送端在待发送的 OFDM符号 322之前***前导 321。该前导 321 可釆用 PN ( Pseudo Noi se , 伪随机）序列，且该前导 321 可在预滤波处理模块对 OFDM时域信号进行预滤波处理后***。发送端釆用如图 4所示的帧结构发送时域信号，使得接收端可基于信号中的该前导 321釆用业界所共知的方法进行信道估计，获得实际信道响应。 FIG. 4 is a schematic diagram of a frame structure of an OFDM system according to another embodiment of the present invention. In the frame structure of the OFDM system as shown in FIG. 4, the transmitting end inserts the preamble 321 before the OFDM symbol 322 to be transmitted. The preamble 321 can use a PN (Pseudo Noi se, pseudo-random) sequence, and the preamble 321 can be inserted after the pre-filtering processing module pre-filters the OFDM time domain signal. The transmitting end transmits the time domain signal by using the frame structure as shown in FIG. 4, so that the receiving end can perform channel estimation based on the preamble 321 in the signal by using a well-known method in the signal to obtain an actual channel response.

以上只是给出了一种实际信道估计获取方法，本领域技术人员可以理解，釆取其它方法进行信道估计，不影响本实施例需要获取实际信道时域响应的特征。 The above is only an actual channel estimation acquisition method. Those skilled in the art can understand that other methods are used for channel estimation, which does not affect the characteristics of the actual channel time domain response that need to be obtained in this embodiment.

接下来对本发明预滤波环节中涉及的预滤波系数的求取方法进行说明。为便于理解该预滤波系数求取方法的本质及其合理性，下面给出预滤波系数求取的详细推导过程： Next, the method for obtaining the pre-filter coefficient involved in the pre-filtering step of the present invention will be described. In order to facilitate the understanding of the nature and rationality of the pre-filtering coefficient method, the detailed derivation process of the pre-filtering coefficient is given below:

本发明实施例可根据通用的最大信干噪比准则，基于 CP不足时的接收信号模型以及反馈部分信道响应，来具体推导预滤波系数 4«]： The embodiment of the present invention can specifically derive the pre-filtering coefficient 4«] according to the universal maximum signal to interference and noise ratio criterion, based on the received signal model when the CP is insufficient and the feedback part channel response.

首先参照图 3给出的实施例实现框图，具体定义实际信道响应为 h [n] , 并不失一般性地假定实际信道响应是因果的，即：实际信道响应的起始位置为 0, 终止位置为 V , 对应着 w < 0和《> v时， /^] = 0。则信道响应的长度（即：信道时延扩展长度）为 V + l Referring first to the embodiment shown in FIG. 3, a block diagram is implemented to specifically define the actual channel response as h [n] without assuming that the actual channel response is causal, ie, the starting position of the actual channel response is 0, terminating The position is V, corresponding to w < 0 and >> v, /^] = 0. Then the length of the channel response (ie: Channel delay spread length) is V + l

接收信号中的等效信道。响应等于预滤波系数与实际信道响应的卷积： The equivalent channel in the received signal. The response is equal to the convolution of the pre-filter coefficient with the actual channel response:

( 1 ) ( 1 )

式（1 ) 中， 4«]为预滤波系数， g[n]为等效信道响应。设预滤波的起始响应位置为 -Μ,, 预滤波的终止响应位置为 Μ₂；

In equation (1), 4«] is the pre-filter coefficient, and g[n] is the equivalent channel response. Let the initial response position of the pre-filtering be -Μ, and the pre-filtered termination response position is Μ ₂ ;

预滤波系数为 0, 即！ «] = 0, 因此，预滤波系数长度为 Μ,+Μ₂+1。 Μ_λ ^Μ₂ 取值可根据实际需要预先设定。 The pre-filtering factor is 0, ie! «] = 0, therefore, the pre-filter coefficient length is Μ, +Μ ₂ +1. Μ _λ ^Μ _{2 The} value can be preset according to actual needs.

将式（ 1 )表达成矩阵形式，可得： Expressing the formula (1) into a matrix form, you can get:

为了进一步简化式（2) 的矩阵形式，可进行如下定义: To further simplify the matrix form of equation (2), the following definitions can be made:

由此可得: Therefore:

g = Hw (4 ) 其中， w为预滤波系数向量，为（Mi+M₂+1 )行的列向量，其中元素从 w [M₂]; H为总时域信道响应矩阵，为 (M!+M₂+v+l )行 X (Mi+M₂+1 ) 列的矩阵； g为等效信道响应向量，为（Mi+Mz+v+l )行的列向量，其元素由公式（2 ) 确定。由于本发明实施例是针对长时延信道导致的 CP不足下的 IBI和 ICI抑制技术，因此在以下分析中假定 CP长度 ^小于信道长度，即 N_g<v + 1。假设接收端反馈的部分信道响应为超过 CP长度的信道响应部分，为方便描述，将信道响应拆分为两部分，其中 ^ = 表示实际信道没有超出 CP长度部分（称为第一部分信道响应），以及实际信道超出 CP长度部分的信道响应 = + (称为第二部分信道响应），本实施例中将第二部分信道响应作为需要反馈的部分，相应地定义对应于第一部分信道响应和第二部分信道响应的信道矩阵： g = Hw (4 ) where w is the prefiltering coefficient vector, which is the column vector of the (Mi+M ₂ +1 ) row, where the element is from w [M ₂ ]; H is the total time domain channel response matrix, which is (M !+M ₂ +v+l ) The matrix of the row X (Mi+M ₂ +1 ); g is the equivalent channel response vector, which is the column vector of the (Mi+Mz+v+l) row, whose elements are determined by the formula (2) OK. Since the embodiment of the present invention is directed to the IBI and ICI suppression techniques for CP deficiency caused by long delay channels, it is assumed in the following analysis that the CP length ^ is smaller than the channel length, that is, N _g < v + 1. It is assumed that the partial channel response fed back by the receiving end is a channel response part exceeding the length of the CP. For convenience of description, the channel response is split into two parts, where ^ = indicates that the actual channel does not exceed the CP length part (referred to as the first part channel response). And the channel response of the actual channel beyond the length of the CP = + (referred to as the second part of the channel response), in this embodiment, the second part of the channel response is used as the part requiring feedback, correspondingly corresponding to the first part of the channel response and the second Channel matrix for partial channel response:

HH

其中，为第一部分信道响应的矩阵形式，称为第一部分信道响应矩阵， H₂为第二部分信道响应的矩阵形式，称为第二部分信道响应矩阵。 1^ 和11₂都是（Mi+Mz+v+l )行 X (Mi+M₂+1 ) 列的矩阵，其中，由第一部分信道响应 i = C^ _;'''^i D中元素构成，具体就是将总信道矩阵 H 中包括的所有第二部分信道响应 = ^ + ^'''^中元素置 0; H₂由第二部分信道响应 = + ιυψ 中元素构成，具体就是将总信道矩阵 Η 中包括的所有第一部分信道响应 =( ''.> Λ' ])中元素置 _0; 因此，式（4) 中，预滤波后的等效信道响应可表示为： The matrix form of the first partial channel response is called the first partial channel response matrix, and H ₂ is the matrix form of the second partial channel response, which is called the second partial channel response matrix. 1^ and 11 ₂ are both (Mi+Mz+v+l) rows of X (Mi+M ₂ +1 ) columns, where the first part of the channel response i = C^ _; '''^i D elements The composition is specifically to set all the second partial channel responses included in the total channel matrix H = ^ + ^'''^ to 0; H _{2 is} composed of the second partial channel response = + ι υψ elements, specifically The elements in the first part of the channel response = included in the channel matrix = = ( ''.> Λ ' ]) are set to _0; therefore, in equation (4), the pre-filtered equivalent channel response can be expressed as:

g二 ( + )W g two ( + )W

接收信号等于等效信道响应和时域信号的卷积与噪声的叠加： The received signal is equal to the superposition of the convolution and noise of the equivalent channel response and the time domain signal:

(6 ) 其中， n为 OFDM符号块的序数， m为样点的序数， x„[ ]为第"个 OFDM符号块的第∞个样点，每块 OFDM符号样点总数为 N = N_g+JS：， 0<m<N，为快速傅里叶变换（FFT) 的积分长度，

为经 OFDM调制后的第 "个 OFDM符号块发送时域信号， n(m)为叠加的噪声。 (6) Where n is the ordinal number of the OFDM symbol block, m is the ordinal number of the sample, x„[ ] is the second sample of the "first OFDM symbol block, and the total number of OFDM symbol samples per block is N = N _{g + J} S :, 0<m<N, is the integral length of the Fast Fourier Transform (FFT),

The time domain signal is transmitted for the OFDM modulated OFDM symbol block, and n(m) is superimposed noise.

由于块间干扰（ IBI )通常来自当前 OF丽符号块相邻二个 OFDM符号块，因此，仅考虑相邻的二个 OFDM符号块（第 n-1个 OFDM符号块和第 n+1个 OFDM 符号块）对当前 OFDM符号块（第 n个 OFDM符号块）的块间干扰，接收信号的模型（式（6) )可进一步表示为： Since the inter-block interference (IBI) usually comes from two adjacent OFDM symbol blocks of the current OF symbol block, only two adjacent OFDM symbol blocks (the n-1th OFDM symbol block and the n+1th OFDM) are considered. Symbol block) For inter-block interference of the current OFDM symbol block (nth OFDM symbol block), the model of the received signal (Equation (6)) can be further expressed as:

y_n[m] = g[m] * x„ [m] + g[m] * x„_, [m + N] + g[m] * x_n+l [m-N] + n[m] y _n [m] = g[m] * x„ [m] + g[m] * x„_, [m + N] + g[m] * x _n+l [mN] + n[m]

M₂ +v M₂ +v M ₂ +v M ₂ +v

1=-Μ_γ l=N_g l=-M_x _{_{1 = -Μ γ l = N g}} l = -M x

将式（ 7 )表示的时域接收信号进行离散傅里叶变换 (简称 DFT )

频域子载波符号： Performing a discrete Fourier transform (referred to as DFT) on the time domain received signal represented by equation (7)

Frequency domain subcarrier symbols:

其中， 0<m<K, 为离散傅里叶变换（DFT) 的积分长度。 Where 0 < m < K is the integral length of the discrete Fourier transform (DFT).

将式（7)代入上式，可得： Substituting equation (7) into the above equation, you can get:

= s_n [k] + e„ [k] + n[k] 式（8) 中，噪声部分为：

块间干扰（IBI)部分为: = s _n [k] + e„ [k] + n[k] In equation (8), the noise part is:

The Inter-Block Interference (IBI) section is:

[m-l + N] + (10)

[ml + N] + (10)

第 Α子载波符号和其所受载波间干扰（ ICI )部分为: The first subcarrier symbol and its inter-carrier interference (ICI) portion are:

K-1 2 n K-1 2 n

1=-Μ, ^Λ m=

1=-Μ, ^Λ m=

式（11A) 中:

In equation (11A):

其中， /为等效信道的抽头信号的序号， m和 k均为子载波的序号， k是目标子载波序号， m是对目标子载波造成 ICI的其它子载波序号。 — J]表示第 /抽头对应的接收信号对应的频域中，第个子载波对第 A个子载波的载波间干扰（ICI) 系数。 Where / is the number of the tap signal of the equivalent channel, m and k are the sequence numbers of the subcarriers, k is the target subcarrier number, and m is the other subcarrier number that causes the ICI to the target subcarrier. – J] indicates the inter-carrier interference (ICI) coefficient of the first subcarrier to the first subcarrier in the frequency domain corresponding to the received signal corresponding to the tap/tap.

根据公式（11A)和（12) , 可进一步将拆分成频域子载波符号部分 (即有用信号部分）和载波间干扰（ ICI )部分： s_n[k]= J g [l](a₀[l]s_n[k] + a_k__m[l]s_n[m])e s_n[k]∑ g[/ [/]e— 7 + ∑ g[l]a_k__m[l]e (11B) According to equations (11A) and (12), it can be further split into frequency domain subcarrier symbol parts (ie, useful signal parts) and inter-carrier interference (ICI) parts: s _n [k]= J g [l] (a ₀ [l]s _n [k] + a _k _ _m [l]s _n [m])es _n [k]∑ g[/ [/]e— 7 + ∑ g[l]a _k _ _m [l ]e (11B)

与通常的多普勒频偏或者扩展导致的 ICI不同的是，本实施例 ICI系数与等效信道中的抽头信号有关，不同的抽头信号对应的 ICI系数 J] 的值不同，具体的， ICI系数可表示为： Different from the ICI caused by the usual Doppler frequency offset or extension, the ICI coefficient of this embodiment is related to the tap signal in the equivalent channel, and the value of the ICI coefficient J] corresponding to different tap signals is different, specifically, ICI The coefficient can be expressed as:

1 ιι«Λ — 1 _ λ 1 ιι«Λ — 1 _ λ

e κ (13) e κ (13)

A=max(— /,0) A=max(— /,0)

根据等效信道的抽头序号 /在有效区间 ≤ /≤ M₂ + V中的不同位置，可分别获得相应的 ICI系数 [/]: According to the tap number of the equivalent channel / different positions in the effective interval ≤ / ≤ M ₂ + V, can be divided Do not get the corresponding ICI coefficient [/]:

N + l N + l

ifm = 0 and -M,≤l(-N_g Ifm = 0 and -M, ≤ l (-N _g

K K

π{-1-Ν _g +l)m π{-1-Ν _g +l)m

sm Sm

K ifm≠ 0 and -Μ ≤ /〈一 N, K ifm≠ 0 and -Μ ≤ /<一一,

K ππι K ππι

sm Sm

K K

if m Q and (14) a_m [I] If m Q and (14) a _m [I]

ifm≠ Q and Ifm≠ Q and

ifm =0 and 0</ < ₂ +v Ifm =0 and 0</ < ₂ +v

K K

πΐπι Πΐπι

_Λ sm _Λ sm

1 K ^J 1 K ^J

ifm≠ 0 and 0</ < ₂ +v Ifm≠ 0 and 0</ < ₂ +v

K sm . 7tm K sm . 7tm

K K

根据式（14)可知，对时延位置在 CP范围内的径，即- N_g≤/≤0时，信号经 FFT变换后将不存在 ICI。 According to equation (14), when the path of the delay position in the CP range, that is, - N _g ≤ / ≤ 0, the ICI is not present after the signal is FFT-converted.

下面对基于最大信干噪比准则进行预滤波系数计算的原理进行说明。总信干噪比即为所有子载波符号总功率与所有子载波符号上的干扰噪声总功率的比值，可釆用下式进行表示： The principle of pre-filtering coefficient calculation based on the maximum signal to interference and noise ratio criterion will be described below. The total signal to interference and noise ratio is the ratio of the total power of all subcarrier symbols to the total power of the interference noise on all subcarrier symbols, which can be expressed by the following equation:

SINR(w (15) SINR(w (15)

PlCl(^W + ^PIBl(^W)+ ^PAWGN PlCl( ^W + ^P IBl( ^W )+ ^P AWGN

式（I⁵)中， SINR(w)为总信干噪比； (w)为信号总功率，等于当前 OFDM 符号（下同）所有子载波符号总功率； _ew(_W)为噪声平均功率，等于所有子载波上噪声总功率； P_CT(w)为所有子载波所受 ICI总功率； P (w)为 IBI总功率。显然，总信干噪比和各功率均与预滤波系数 w有关。 In equation (I ⁵ ), SINR(w) is the total signal to interference and noise ratio; (w) is the total signal power, which is equal to the total power of all subcarrier symbols of the current OFDM symbol (the same below); _ew ( _W ) is the average power of noise, Equal to the total noise power on all subcarriers; P _CT (w) is the total ICI power of all subcarriers; P (w) is the total IBI power. Obviously, the total signal dry noise ratio and each power are related to the pre-filtering coefficient w.

接下来，基于上述推导过程中的信号部分， ICI部分、 IBI部分和噪声部分的信号模型，分别求取式（15) 中涉及的各部分功率。 Next, based on the signal portion of the above derivation process, the signal model of the ICI portion, the IBI portion, and the noise portion, the powers of the respective portions involved in the equation (15) are respectively obtained.

1、所有子载波信号总功率？的计算 1. What is the total power of all subcarrier signals? Calculation

根据式（8) 、（11B)和（14) , 信号总功率 P_s可表示为：

According to equations (8), (11B) and (14), the total signal power P _s can be expressed as:

根据维纳 -辛钦原理，可进一步得 According to the Wiener-Sinqin principle, it can be further

P_s{w) = Ka_s ² P _s {w) = Ka _s ²

= Κσ' (w^M H D_sH_xw + w^H H" D_sH₂w 其中为发送子载波符号功率，上标 H表示相应矩阵的共轭转置。 D_x(不妨称为信号功率计算矩阵）是为进一步将式（ 16B)表示为更为简洁的方式而定义的： = Κσ' (w ^M HD _s H _x w + w ^H H" D _s H ₂ w where is the transmitted subcarrier symbol power, and the superscript H indicates the conjugate transpose of the corresponding matrix. D _x (may be called signal power calculation Matrix) is defined to further express equation (16B) as a more concise way:

由公式（14)可得 Available from formula (14)

O_s为 (M, +M₂+v) 行 X ( , +M₂+v)列的对角矩阵，对角元素可表示为：为预设的预滤波

O _s is the diagonal matrix of the (M, +M ₂ +v) row of X ( , +M ₂ +v) columns, and the diagonal elements can be expressed as: Pre-filtered for the preset

的起始响应位置， M₂为预设的预滤波的终止响应位置；预设的实际信道响应的起始位置为 0, 终止位置 V为预设的实际信道响应的终止位置； ^：为预设的 FFT (快速傅里叶）变换长度。通过式（17B)可知，信号功率计算矩阵 1^只包含一个未知变量 V, ( V+1) 为信道响应总长度，因此，在获得了信道长度信息，根据式（17 ) 即可计算得到 D_x。 The initial response position, M ₂ is the preset pre-filtered termination response position; the initial position of the preset actual channel response is 0, and the termination position V is the termination position of the preset actual channel response; ^: Pre- Set the FFT (Fast Fourier) transform length. It can be known from equation (17B) that the signal power calculation matrix 1^ contains only one unknown variable V, and (V+1) is the total length of the channel response, and therefore, the channel length is obtained. Information, D _x can be calculated according to equation (17).

2、所有子载波所受 ICI总功率 P_ra(_w)的计算首先根据公式（ 8 ) 、（ 11B )和（ I⁴ )可获得其它子载波对第 k个子载波的 ICI为：

2, all the subcarrier suffered ICI calculated total power P _ra _(w) is first according to equation (8), (11B) is obtained and (I ⁴⁾ of the other sub-carriers ICI k-th subcarrier is:

对应的第 k个子载波所受 ICI功率为： P_ICI[k] |_g[/]"J/] ( 20)

The ICI power of the corresponding kth subcarrier is: P _ICI [k] | _g [/]"J/] ( 20)

为了将式（20 )表示为更为简洁的方式以便于计算，可定义载波间干扰功率计算矩阵（即： ICI功率计算矩阵） _D： In order to express equation (20) in a more concise manner for calculation, an inter-carrier interference power calculation matrix (ie: ICI power calculation matrix) _D can be defined:

-l-N„ —l_N。 -l-N„ —l_N.

-Μ≤ l(-N_i -Μ≤ l(-N _i

K (1―" K ^), K (1―" K ^),

0. -N <1≤0

ICI功率计算矩阵 /)₁为（^₁+ ^₂+ 行 X (M_{1 +}M₂+ 列的对角矩阵。 /为满足不等式 -Μ^ Λ^+ν的整数，为预设的预滤波的起始响应位置，预设的预滤波的终止响应位置；预设的信道响应总长度为 v + 1; f为预设的快速傅里叶变换长度； N_g为循环前缀的长度。通过式（21 )可知， CP长度 N_g和 The ICI power calculation matrix /) ₁ is (^ ₁ + ^ ₂ + row X (the diagonal matrix of M _{1 +} M ₂ + columns. / is an integer that satisfies the inequality - Μ^ Λ^+ν, is the preset pre-filtering The initial response position, the preset pre-filtered termination response position; the preset channel response total length is v + 1; f is the preset fast Fourier transform length; N _g is the length of the cyclic prefix. (21) It can be seen that the CP length N _g and

FFT变换长度^均为***已知参数，在获得了信道响应长度信息，即可计算得 The FFT transform length ^ is a known parameter of the system. After obtaining the channel response length information, it can be calculated.

从公式（20)可知，不同子载波上所受到的 ICI功率是相同的。进一步才艮据式（20)和（21 ) , 可得到所有子载波所受 ICI 总功率 P_ra(_w)釆用 ICI 功率计算矩阵 A表示的计算公式为： It can be seen from equation (20) that the ICI power received on different subcarriers is the same. Further According to equations (20) and (21), the ICI total power P _ra ( _w ) received by all subcarriers can be obtained. The calculation formula represented by the ICI power calculation matrix A is:

P_ICI(w) = Kow^HH^HD_xH_xw ( 22 ) P _ICI (w) = Kow ^H H ^H D _x H _x w ( 22 )

3、块间干扰总功率 (_w)的计算 3. Calculation of total power ( _w ) between blocks

根据公式（8 )和（10) , 块间干扰总功率 ^(w = Ka² H^HD₂Hw ( 23 )

According to formulas (8) and (10), the total power of inter-block interference ^(w = Ka ² H ^H D ₂ Hw ( 23 )

其中块间干扰功率计算矩阵 D₂ (即： IBI 功率计算矩阵）是为进一步将 P_IBI (w)表示为更为简洁的方式而定义的： The inter-block interference power calculation matrix D ₂ (ie: IBI power calculation matrix) is defined to further express P _IBI (w) as a more concise way:

ΙΒΙ功率计算矩阵/ )₂为（Μ,+Μ +ν) 行 X (M_{1 +}M₂+ 列的对角矩阵。 /为满足不等式 -Μ^ Λ^+ν的整数，为预设的预滤波的起始响应位置，预设的预滤波的终止响应位置；预设信道响应总长度为 v + 1; f为预设的快速傅里叶变换长度； N_g为循环前缀的长度。通过式（23 )可知， CP 长度 N_g和 ΙΒΙ Power calculation matrix / ) ₂ is (Μ, +Μ +ν) Row X (M _{1 +} M ₂ + diagonal matrix of the column. / Integer for satisfying the inequality -Μ^ Λ^+ν, is the preset pre- The initial response position of the filter, the preset pre-filtered termination response position; the total length of the preset channel response is v + 1; f is the preset fast Fourier transform length; N _g is the length of the cyclic prefix. (23) It can be seen that the CP length N _g and

FFT变换长度^均为***已知参数,在获得了信道长度信息，即可计算得到 D₂。 4、所有子载波上噪声总功率 _ew(_W)的计算： The FFT transform length ^ is a known parameter of the system. When the channel length information is obtained, D ₂ can be calculated. 4. Calculation of total noise power _ew ( _W ) on all subcarriers:

根据式（ 9 )可得：

According to formula (9):

其中 σ„²为子载波符号上的噪声功率。 Where σ „ ² is the noise power on the subcarrier symbol.

5、所有子载波符号总信干噪比 SINR(w)的表达式 5. Expression of the total signal to interference and noise ratio SINR(w) of all subcarrier symbols

根据公式（ 16B ) 、 ( 22 ) , ( 23 )和（ 25 )可得:

According to the formulas ( 16B ) , ( 22 ) , ( 23 ) and ( 25 ):

Ka_s ²(w^HH^ID_sH_lw + w^HH"D_sH₂w Ka _s ² (w ^H H ^I D _s H _l w + w ^H H"D _s H ₂ w

Ko_s ^z (w^HH^HD_lHw +

+ Κσ_η (27) Ko _s ^z (w ^H H ^H D _l Hw +

+ Κσ _η (27)

w^HH"D_IH_lw + W^HH"D_IH₂W) +丄/ w^HAw + -I w ^H H"D _I H _l w + W ^H H"D _I H ₂ W) +丄/ w ^H Aw + -I

P P P P

式（27) 中， (不妨称！^为干扰功率计算矩

In equation (27), (may wish to call! ^ is the interference power calculation moment

B = H"D_VH, +H"D_VH, (28) B = H"D _V H, +H"D _V H, (28)

A = H D_TH, +H"D_TH (29) 通过最大化总信干噪比 SINR(w) , 可以得到最大信干噪比准则下预滤波系数 w。由式（27 )可知，总信干噪比 SINR(w)取最大值的等价条件为：在 w^ffBw = l的约束条件下最小化其分母 w^ffAw+丄，又进一步等价于最小化 A = HD _T H, +H"D _T H (29) By maximizing the total signal to interference and noise ratio SINR(w), the pre-filtering coefficient w under the maximum signal to interference and noise ratio criterion can be obtained. From equation (27), the total The equivalent condition for taking the maximum value of the signal-to-noise ratio SINR(w) is: Minimizing the denominator w ^ff Aw+丄 under the constraint of w ^ff Bw = l, and further equivalent to minimizing

P P

w^ffAw。基于该等效求解模型，不妨称 B为约束矩阵，称 A为目标矩阵。很显然， A和 B矩阵将直接影响求解得到的预滤波系数。 w ^ff Aw. Based on the equivalent solution model, it may be said that B is a constraint matrix, and A is called a target matrix. Obviously, the A and B matrices will directly affect the prefilter coefficients obtained by the solution.

由于本实施例中接收端向发送端反馈的部分信道响应为实际信道超出 CP 长度部分的信道响应，因此第一部分信道响应矩阵对于发送端而言是未知的，第二部分信道响应矩阵则是可知的。因此，（28)和（29)定义的 B、 Since the partial channel response fed back by the receiving end to the transmitting end in this embodiment is the channel response of the actual channel exceeding the length of the CP, the first partial channel response matrix is unknown to the transmitting end, and the second partial channel response matrix is known. of. Therefore, B, as defined in (28) and (29),

A矩阵均是不可得的。对^ A矩阵中不可求的部分和 ^/]0 分别用其期望和/ ·' （对某些信道场景，可以根据标准信道模型通过预先计算得到这两部分）来代替，则可以解决上述问题。 A matrix is not available. For the unavoidable part of the ^ A matrix and ^/]0 respectively The above problem can be solved by the expectation and / (' (for some channel scenarios, which can be obtained by pre-calculating the two parts according to the standard channel model).

基于实验和理论分析获知，最大化严格意义上的总信干噪比并不一定增强 OFDM的解调性能，因为解调性能还决定于信号和干扰功率在各子载波上的分布特性。因此，本实施例进一步给出了关于 A和 B矩阵的以下替代方法（为区别起见，分别用 A和 έ来表示，仍分别称为目标矩阵和约束矩阵）：

Based on experimental and theoretical analysis, it is known that maximizing the total signal to interference and noise ratio in the strict sense does not necessarily enhance the demodulation performance of OFDM, because the demodulation performance is also determined by the distribution characteristics of the signal and interference power on each subcarrier. Therefore, the present embodiment further gives the following alternative methods for the A and B matrices (for the sake of distinction, denoted by A and έ, respectively, still referred to as the target matrix and the constraint matrix, respectively):

(2) .^ = H^_DIH₂ ( 30) (2) .^ = H^ _DI H ₂ ( 30)

(3) .λ = Ε{Η(^ΙΌ_ΙΗ_ι} (3) .λ = Ε{Η( ^Ι Ό _Ι Η _ι }

(a) , ft = E{H D_sH_x} + H"D_SH₂ (a) , ft = E{HD _s H _x } + H"D _S H ₂

(b) . ft = H^D_SH₂ ( 31 ) (b) . ft = H^D _S H ₂ ( 31 )

{c).% = E{H^D_sH_l} 式（30) 示出的目标矩阵 A的三种计算方法（1〜3) , 与式（31 ) 示出的约束矩阵 έ的三种计算方法（a〜c) , 可进行两两任意组合，因此，组合可包括： ft =

(1-a) ft = Ε{Η; Η:卜 Hf/¾H₂ S = H₂¾H₂ (1-b) ft = Ε{Η; Η: + Hf/¾H₂ = E{H D_sH_x} (1-c) ft = ^2 ^DI^H2 = E{H D_sH_x} + H^D_SH₂ (2-a) ft = ^2 ^DI^H2 δ = H"D_SH₂ (2-b) ft % = E{H^D_SH,} (2-c) ft = Ε{Η; Η: \ = E{H D_sH_x} + H^D_SH₂ (3-a) ft = Ε{Η; Η: \ ^=H"D_SH₂ (3-b) ft = Ε{Η; Η: (3-c) 基于上述分析过程可知，目标矩阵 A和约束矩阵 έ的计算，涉及到的信号功率计算矩阵! )_χ和干扰功率计算矩阵 _D ，均可以根据接收端反馈的信道长度预先计算好，由于无线信道长度变化相对緩慢，因此反馈更新周期可以较长，如此 D o E^可预先算好。基于 D o E^的该特性，将!^和亦可分别称为预置信号功率计算矩阵和预置干扰功率计算矩阵。 {c).% = E{H^D _s H _l } Three kinds of calculation methods (1 to 3) of the target matrix A shown by equation (30), and three types of constraint matrix 示出 shown by equation (31) The calculation method (a~c) can be used in any combination of two or two. Therefore, the combination can include: ft =

(1-a) ft = Ε{Η; Η: 卜Hf/3⁄4H ₂ S = H ₂ 3⁄4H ₂ (1-b) ft = Ε{Η; Η: + Hf/3⁄4H ₂ = E{HD _s H _x } (1-c) ft = ^2 ^D I ^H 2 = E{HD _s H _x } + H^D _S H ₂ (2-a) ft = ^2 ^D I ^H 2 δ = H"D _S H ₂ ( 2-b) ft % = E{H^D _S H,} (2-c) ft = Ε{Η; Η: \ = E{HD _s H _x } + H^D _S H ₂ (3-a) Ft = Ε{Η; Η: \ ^=H"D _S H ₂ (3-b) ft = Ε{Η; Η: (3-c) Based on the above analysis process, the calculation of the target matrix A and the constraint matrix έ , the letter involved No. Power calculation matrix!) _χ and interference power calculation matrix _D can be pre-calculated according to the channel length fed back by the receiving end. Since the length of the wireless channel changes relatively slowly, the feedback update period can be long, so D o E^ can be pre- Well done. Based on this characteristic of D o E^, !^ and may also be referred to as a preset signal power calculation matrix and a preset interference power calculation matrix, respectively.

通过公式（29 ) 下面一段的基于最大信干噪比准则下预滤波系数 w求解及相应的等效求解模型的分析，可进一步釆用现有矩阵数学方法求解获得预滤波系数 W: w = I ( 30 ) 其中， I为 έ的乔里斯基（Cholesky )分解， I和 έ满足关系：

By analyzing the pre-filter coefficient w and the corresponding equivalent solution model based on the maximum signal to interference and noise ratio criterion in the following paragraph (29), the existing matrix mathematical method can be further used to obtain the pre-filter coefficient W: w = I (30) Among them, I is the decomposition of Cholesky, and I and έ satisfy the relationship:

v_mm为矩阵 I"— 'A ^ff的最小特征值对应的特征向量。 v _mm is the eigenvector corresponding to the minimum eigenvalue of the matrix I" - 'A ^ff .

本实例中上述给出的 CP不足情形下的预滤波（本质上也可以认为是信道缩短均衡器）系数求取方法，尽管是针对在发射机进行的预滤波进行说明的，但本领域普通技术人员可以理解：所述预滤波器或信道缩短均衡系数求取方法，稍作改动，可以完全用于在接收机实施的信道缩短均衡器的系数求取，而所述改动不需要付出额外的创新性劳动，也并不使相应技术方案的本质脱离本发明实施例技术方案的精神和范围。 In the present example, the pre-filtering (which may also be considered as a channel shortening equalizer) coefficient obtaining method in the case of CP shortage given above is described, although it is described for pre-filtering performed at the transmitter, but the prior art It can be understood by the personnel that: the pre-filter or channel shortening equalization coefficient obtaining method can be completely used for the coefficient of the channel shortening equalizer implemented in the receiver, and the modification does not require additional innovation. Sexual labor does not deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

通过上述预滤波器系数求取原理的分析，可知预滤波系数计算方法可根据预先获取的信道响应总长度，计算信号功率计算矩阵! ^和干扰功率计算矩阵；根据得到的信号功率计算矩阵!^和干扰功率计算矩阵以及预先获取的部分信道响应，计算约束矩阵和目标矩阵；根据得到的约束矩阵和目标矩阵，计算当信干噪比为最大值时对应的预滤波系数，所述信干噪比等于信号功率和干扰噪声功率的比值。 Through the analysis of the pre-filter coefficient finding principle, it can be known that the pre-filter coefficient calculation method can calculate the signal power calculation matrix according to the total length of the channel response obtained in advance! ^ and the interference power calculation matrix; calculate the matrix according to the obtained signal power! And an interference power calculation matrix and a pre-acquired partial channel response, calculating a constraint matrix and a target matrix; and calculating, according to the obtained constraint matrix and the target matrix, a corresponding pre-filter coefficient when the signal to interference and noise ratio is a maximum value, the signal dry noise The ratio is equal to the ratio of signal power to interference noise power.

以下说明本实施例基于最大信干噪比准则进行预滤波系数计算的实现流程。图 5为本发明另一实施例提供的预滤波系数获取方法流程图。图 5所示的流程亦可作为图 3中步骤 305的一个实现流程。如图 5所示，预滤波系数获取方法包括： The implementation flow of the pre-filter coefficient calculation based on the maximum signal to interference and noise ratio criterion in the present embodiment will be described below. FIG. 5 is a flowchart of a method for acquiring pre-filter coefficients according to another embodiment of the present invention. The flow shown in FIG. 5 can also be used as an implementation flow of step 305 in FIG. As shown in FIG. 5, the pre-filter coefficient acquisition method includes:

步骤 51、根据信道响应总长度，分别获取信号功率计算矩阵 I ^和干扰功率计算矩阵。 Step 51: Obtain a signal power calculation matrix I^ and an interference power calculation matrix according to the total length of the channel response.

假设信道响应总长度为（ v + 1 ) , 可以分别根据公式（ 17B) 、（ 21 )和 (24)计算得到信号功率计算矩阵 D_x、 ICI功率计算矩阵 A和 IBI功率计算矩阵，并进一步由 =D₁₊D₂计算得到干扰功率计算矩阵 D Assuming that the total channel response length is (v + 1), the signal power calculation matrix D _x , the ICI power calculation matrix A and the IBI power calculation matrix can be calculated according to the formulas ( 17B), ( 21 ) and ( 24 ), respectively, and further =D ₁₊ D ₂ Calculate the interference power calculation matrix D

步骤 52、根据实际信道超过 CP长度的部分信道响应 (^; Step 52: According to the partial channel response of the actual channel exceeding the CP length ( ^;

构造第二部分信道响应矩阵 H₂。A second partial channel response matrix H _{2 is constructed} .

H₂中第 1行到第 ^行的各元素均为 0, 且 H₂中第 N行到第 Mi+Mz+v+l行的各元素与总信道矩阵 H中第 N_g行到第 Mi+Mz+v+l行的各元素相同。相关参数 N_g、、 M₂均是***预置参量，具体意义见上文，（v + 1 )为信道响应总长度。 Each element of H ₂ in the first row to the second ^ line are 0, and H ₂ N-th row to Mi + Mz + v + each element l rows and the overall channel matrix H of N _g-th row to Mi The elements of the +Mz+v+l line are the same. The relevant parameters N _g , , M ₂ are system preset parameters, the specific meaning is above, (v + 1 ) is the total length of the channel response.

步骤 53、根据步骤 51得到的信号功率计算矩阵 D_x和干扰功率计算矩阵以及步骤 52得到的第二部分信道响应矩阵 H₂, 分别获取目标矩阵 A和约束矩阵 B; Step 53: Calculate the matrix D _x and the interference power calculation matrix according to the signal power obtained in step 51 and the second partial channel response matrix H ₂ obtained in step 52, respectively acquiring the target matrix A and the constraint matrix B;

其中，具体的获取方式可以参考式（l_a)、（l_b)、（l_c)、（2_a)、 ( 2- b ) 、（ 2- c ) 、（ 3- a ) 、（ 3- b )或（ 3- c ) 。步骤 54、根据步骤 53得到的目标矩阵 A和约束矩阵 έ , 由公式（ 30 )获取预滤波系数 w。 Wherein, the specific acquisition manner can refer to the formula (l_a), (l_b), (l_c), (2_a), (2-b), (2-c), (3-a), (3-b) or ( 3- c ). Step 54: Obtain the pre-filter coefficient w from the formula (30) according to the target matrix A and the constraint matrix 得到 obtained in step 53.

通过图 5所示的预滤波系数的获取方法流程得到预滤波系数之后，就可以对信号进行预滤波处理。预滤波系数的获取方法可在发送端获取，或在接收端获取。基于上述预滤波系数的预滤波处理，可以在发送端实施，也可以在接收端实施，本发明实施例并不限制。如果获取预滤波系数以及预滤波处理在发送端实施的话，则发送端需要接收端反馈信道响应总长度和部分信道响应，根据发送端反馈的信道响应总长度和部分信道响应获取预滤波系数对时域信号进行预滤波处理。如果获取预滤波系数以及预滤波处理在接收端实施的话，接收端自身可通过信道估计等方法就可以获取到信道响应总长度和部分信道响应信息，并根据信道响应总长度和部分信道响应获取预滤波系数对时域信号进行预滤波处理。矩阵 A和约束矩阵 έ的不同取值得到的预滤波系数在发端进行预滤波处理，所获得的 ΙΒΙ抑制效果，与背景技术中所介绍的 OFDM***中典型的 IBI抑制技术，即残留符号间干扰消除 ( RI SIC算法）的 IBI抑制效果，基于仿真进行了比较。 After the pre-filtering coefficients are obtained by the acquisition method flow of the pre-filter coefficient shown in Fig. 5, the signal can be pre-filtered. The method of obtaining the pre-filter coefficient can be obtained at the transmitting end or acquired at the receiving end. The pre-filtering process based on the foregoing pre-filtering coefficients may be implemented on the transmitting end or in the receiving end, which is not limited in the embodiment of the present invention. If the pre-filtering coefficient and the pre-filtering process are implemented at the transmitting end, the transmitting end needs the receiving end to feedback the total length of the channel response and the partial channel response, and obtain the pre-filtering coefficient timing according to the total length of the channel response fed back by the transmitting end and the partial channel response. The domain signal is pre-filtered. If the pre-filtering coefficient is obtained and the pre-filtering process is implemented at the receiving end, the receiving end itself can obtain the total length of the channel response and part of the channel response information by means of channel estimation, etc., and obtain the pre-acquisition according to the total length of the channel response and part of the channel response. The filter coefficients pre-filter the time domain signals. The pre-filtering coefficients obtained by different values of the matrix A and the constraint matrix 在 are pre-filtered at the origin, and the obtained ΙΒΙ suppression effect is compared with the typical IBI suppression technique in the OFDM system introduced in the background art, that is, residual intersymbol interference. The IBI suppression effect of the (RI SIC algorithm) was eliminated and compared based on simulation.

图 6为本发明实施例干扰抑制方法和现有干扰抑制方法的性能比较示意图一；图 7为本发明实施例干扰抑制方法和其他干扰抑制方法的性能比较示意图二。图 6和图 7对应的仿真是基于微波存取全球互通技术（Wor ldwide Interoperabi l i ty for Microwave Acces s , 简称 WIMAX )标准中的 OFDM系统，每个 OFDM符号块子载波数为 914 , 总带宽为 MHz , CP长度为 1/8符号周期长度。仿真的场景为一种典型山区信道模型（HT信道），发送符号调制方式为 4QAM (正交幅度调制）。接收机釆用理想符号同步。图 6中横坐标为接收信号的信噪比（SNR ) , 单位为 dB; 纵坐标为误符号率（SER ) 。图 7中横坐标为接收信号的信噪比（SNR ) , 单位为 dB; 纵坐标为误符号块率（WER ) 。图 6和图 7中，被比曲线为在接收端釆用现有 RISIC算法进行接收信号预滤波后的性能曲线；优化曲线 1为基于式（2-b ) 的目标矩阵 A和约束矩阵 B求得的预滤波系数，进行发送信号预滤波后的性能曲线；同理，优化曲线 2-5 分别为基于式（ 3-b ) 、（ 1-b ) 、 ( 1-a )和（ 3-c ) 的目标矩阵 A和约束矩阵 έ求得的预滤波系数，进行发送信号预滤波后的性能曲线。通过比较图 6 和图 7的各性能曲线可知，优化曲线 1的性能最优，且要明显优于被比曲线，以在 10%的工作点为例，优化曲线 1在 SER和 WER性能指标意义上，分别具有 5dB和 10dB左右的性能增益。这意味着发送端釆用优化曲线 1对应的预滤波系数对发送信号进行预滤波处理后，接收端即使不再进行额外的干扰抑制处理，亦可获得相当甚至优于目前在接收端实施的复杂 IBI干扰抑制处理的性能。 FIG. 6 is a schematic diagram 1 of performance comparison between an interference suppression method and a prior art interference suppression method according to an embodiment of the present invention; FIG. 7 is a second schematic diagram of performance comparison of an interference suppression method and other interference suppression methods according to an embodiment of the present invention. The simulation corresponding to FIG. 6 and FIG. 7 is based on the OFDM system in the Wor ldwide Interoperabi ty for Microwave Acces s (WIMAX ) standard, and the number of subcarriers per OFDM symbol block is 914, and the total bandwidth is MHz, CP length is 1/8 symbol period length. The simulated scene is a typical mountain channel model (HT channel), and the transmitted symbol modulation method is 4QAM (Quadrature Amplitude Modulation). The receiver uses ideal symbol synchronization. In Figure 6, the abscissa is the signal-to-noise ratio (SNR) of the received signal in dB; the ordinate is the symbol error rate (SER). In Figure 7, the abscissa is the signal-to-noise ratio (SNR) of the received signal in dB; the ordinate is the symbolic block rate (WER). In Fig. 6 and Fig. 7, the ratio curve is the performance curve after the received signal is pre-filtered by the existing RISIC algorithm at the receiving end; the optimization curve 1 is based on the target matrix A and the constraint matrix B of the formula (2-b). The obtained pre-filtering coefficient is used to perform the pre-filtered performance curve of the transmitted signal; similarly, the optimization curves 2-5 are based on the equations (3-b), (1-b), (1-a) and (3-c), respectively. The target filter A and the pre-filter coefficient obtained by the constraint matrix are used to perform a pre-filtered performance curve of the transmitted signal. By comparing the performance curves of Fig. 6 and Fig. 7, it can be seen that the performance of the optimization curve 1 is optimal, and it is obviously better than the ratio curve, so as to take the 10% working point as an example, the significance of the optimization curve 1 in the SER and WER performance indexes Above, there are performance gains of about 5dB and 10dB, respectively. This means that after the transmitting end pre-filters the transmitted signal with the pre-filtering coefficient corresponding to the optimization curve 1, the receiving end can obtain a considerable or even better performance than that currently implemented at the receiving end even if no additional interference suppression processing is performed. IBI interference suppression processing performance.

综上分析，本实施例在多载波***中发送端根据接收端的反馈获得部分信道信息，基于实施例 5中所给出方法计算预滤波系数，并所述预滤波系数，对时域信号进行预滤波处理后发送，能够充分达到类似信道缩短的目的，使得接收端接收到时域信号中，等效信道响应的能量尽量集中在短于循环前缀长度的范围内，从而对由于信道时延扩展大于 CP长度导致的 IBI具有有效抑制效果，同时避免了接收机为了 IBI所需显著增加的实现复杂度。因此，相对于现有技术，本实施例明显降低了接收机抑制干扰所需的运算复杂度，具有实现简单、易推广应用等优点。 In summary, in the multi-carrier system, the transmitting end obtains partial channel information according to the feedback of the receiving end, calculates a pre-filtering coefficient based on the method given in Embodiment 5, and performs pre-filtering coefficients on the time domain signal. After filtering and transmitting, the channel shortening can be fully achieved, so that the energy received by the receiving end in the time domain signal is concentrated in a range shorter than the cyclic prefix length, so that the channel delay is larger than The IBI caused by the CP length has an effective suppression effect while avoiding the significantly increased implementation complexity of the receiver for IBI. Therefore, compared with the prior art, the embodiment significantly reduces the computational complexity required for the receiver to suppress interference, and has the advantages of simple implementation, easy application, and the like.

图 8为本发明另一实施例提供的发射机结构图。如图 8所示，本实施例发射机包括：信道信息获取模块 81、预滤波系数计算模块 82和预滤波处理模块 83。 FIG. 8 is a structural diagram of a transmitter according to another embodiment of the present invention. As shown in FIG. 8, the transmitter of this embodiment includes: a channel information acquisition module 81, a pre-filter coefficient calculation module 82, and a pre-filter processing module 83.

信道信息获取模块 81用于根据接收端的反馈获得部分信道信息，所述部分信道信息可包括部分信道响应。 The channel information obtaining module 81 is configured to obtain partial channel information according to feedback from the receiving end, and the partial channel information may include a partial channel response.

预滤波系数计算模块 82用于根据获得的部分信道信息，计算预滤波系数。预滤波处理模块 83用于根据得到的所述预滤波系数，对时域信号进行预滤波处理后发送。在上述技术方案的基础上中，所述部分信道信息还可包括信道响应总长度。预滤波系数计算模块 82可进一步包括：预置矩阵计算单元 821、目标矩阵和约束矩阵计算单元 822和预滤波系数求取单元 823。 The pre-filter coefficient calculation module 82 is configured to calculate a pre-filter coefficient according to the obtained partial channel information. The pre-filtering processing module 83 is configured to perform pre-filtering processing on the time domain signal according to the obtained pre-filtering coefficient and then transmitting. In the above technical solution, the partial channel information may further include a total length of the channel response. The pre-filter coefficient calculation module 82 may further include: a preset matrix calculation unit 821, a target matrix and constraint matrix calculation unit 822, and a pre-filter coefficient determination unit 823.

预置矩阵计算单元 821用于根据所述信道响应总长度，计算信号功率计算矩阵和干扰功率计算矩阵。 The preset matrix calculation unit 821 is configured to calculate a signal power calculation matrix and an interference power calculation matrix according to the total length of the channel response.

目标矩阵和约束矩阵计算单元 822用于根据得到的信号功率计算矩阵和干扰功率计算矩阵以及信道信息获取模块 81获取的部分信道响应，计算约束矩阵和目标矩阵。 The target matrix and constraint matrix calculation unit 822 is configured to calculate a constraint matrix and a target matrix according to the obtained signal power calculation matrix and the interference power calculation matrix and the partial channel response acquired by the channel information acquisition module 81.

部分信道响应可包括：实际信道超出循环前缀长度部分的信道响应（即：第二部分信道响应）。目标矩阵和约束矩阵计算单元 822可选用信道信息获耳4莫块 81获取的部分信道响应构造的第二部分信道响应矩阵，根据信号功率计算矩阵、干扰功率计算矩阵及第二部分信道响应矩阵计算约束矩阵和目标矩阵。 The partial channel response may include: the channel response of the actual channel beyond the cyclic prefix length portion (ie: the second partial channel response). The target matrix and constraint matrix calculation unit 822 can select the second partial channel response matrix of the partial channel response structure acquired by the channel information acquisition block 81, and calculate according to the signal power calculation matrix, the interference power calculation matrix, and the second partial channel response matrix. Constraint matrix and target matrix.

预滤波系数求取单元 823用于根据得到的约束矩阵和目标矩阵，基于最大信干噪比准则，计算最大化信干噪比时对应的预滤波系数。本实施例信道信息获取模块 81、预滤波系数计算模块 82和预滤波处理模块 83的功能可集成为一个功能模块，如集成为图 3对应实施例中所示的预滤波模块 32。基于本实施例发射机在信道响应总长度大于循环前缀（CP ) 长度的应用场景中，在发送端实现信号的预滤波处理的实现原理详见图 3-图 5 对应实施例的记载，不再赘述。 The pre-filtering coefficient obtaining unit 823 is configured to calculate a pre-filtering coefficient corresponding to the maximum signal to interference and noise ratio based on the maximum signal to interference and noise ratio criterion according to the obtained constraint matrix and the target matrix. The functions of the channel information obtaining module 81, the pre-filtering coefficient calculating module 82 and the pre-filtering processing module 83 of this embodiment can be integrated into one functional module, such as the pre-filtering module 32 shown in the corresponding embodiment of FIG. The implementation principle of the pre-filtering process for implementing the signal at the transmitting end in the application scenario in which the total channel response length is greater than the cyclic prefix (CP) length in this embodiment is shown in FIG. 3 to FIG. 5 corresponding to the description of the embodiment, no longer Narration.

本实施例针对信道响应总长度大于循环前缀（CP )长度的情形，通过发射机对待发送的 OFDM时域信号进行类似信道缩短均衡的预滤波处理，使得接收端接收信号中，等效信道响应的能量尽量集中在短于 CP长度的范围内，从而保证接收信号所受 IBI和 ICI得到有效预抑制。本实施例有利于降低接收端为了抑制干扰所需的运算复杂度和处理时延，明显降低了实现的难度。 In this embodiment, for the case where the total length of the channel response is greater than the length of the cyclic prefix (CP), the OFDM time domain signal to be transmitted by the transmitter is subjected to pre-filtering processing similar to channel shortening equalization, so that the receiving channel receives the equivalent channel response. The energy is concentrated as much as possible within a range shorter than the length of the CP, thereby ensuring that the IBI and ICI of the received signal are effectively pre-suppressed. This embodiment is advantageous for reducing the computational complexity and processing delay required by the receiving end to suppress interference, and significantly reducing the difficulty of implementation.

图 9为本发明另一实施例提供的接收机结构图。如图 9所示，本实施例接收机包括：信道估计模块 91和反馈模块 92。信道估计模块 91用于进行信道估计以获得时域信道响应。 FIG. 9 is a structural diagram of a receiver according to another embodiment of the present invention. As shown in FIG. 9, the receiver of this embodiment includes: a channel estimation module 91 and a feedback module 92. Channel estimation module 91 is operative to perform channel estimation to obtain a time domain channel response.

反馈模块 92 用于从所述时域信道响应抽取部分信道信息并向发送端反馈，以供所述发送端根据所述部分信道信息计算预滤波系数，并根据所述预滤波系数对时域信号进行预滤波处理；所述部分信道信息包括部分信道响应。 The feedback module 92 is configured to extract partial channel information from the time domain channel response and feed back to the transmitting end, so that the sending end calculates a pre-filtering coefficient according to the partial channel information, and pairs the time domain signal according to the pre-filtering coefficient. Pre-filtering processing is performed; the partial channel information includes a partial channel response.

在上述技术方案的基础上，所述部分信道信息还可包括信道响应总长度。反馈模块 92可进一步包括：反馈信道响应抽取单元 921和反馈信道响应量化单元 922。 Based on the foregoing technical solution, the partial channel information may further include a total length of the channel response. The feedback module 92 may further include a feedback channel response extraction unit 921 and a feedback channel response quantization unit 922.

反馈信道响应抽取单元 921用于在所述时域信道响应中抽取部分信息。反馈信道响应量化单元 922用于对抽取的所述部分信息进行量化处理得到所述部分信道响应及信道响应总长度，并向所述发送端反馈。 The feedback channel response extraction unit 921 is configured to extract partial information in the time domain channel response. The feedback channel response quantization unit 922 is configured to quantize the extracted partial information to obtain the partial channel response and the total length of the channel response, and feed back to the transmitting end.

基于本实施例接收机信道估计模块 91 和反馈模块 92 (包括反馈信道响应抽取单元 921 和反馈信道响应量化单元 922 ) , 在信道响应总长度大于循环前缀（CP ) 长度的应用场景中，在发送端实现信号的预滤波处理的实现原理详见图 3-图 5对应实施例的记载，不再赘述。 Based on the present embodiment, the receiver channel estimation module 91 and the feedback module 92 (including the feedback channel response extraction unit 921 and the feedback channel response quantization unit 922) are in the application scenario in which the total channel response length is greater than the cyclic prefix (CP) length. The implementation principle of the pre-filtering process of the terminal implementation signal is shown in the description of the corresponding embodiment in FIG. 3 to FIG. 5, and details are not described herein again.

本实施例在信道响应总长度大于循环前缀（CP ) 长度的应用场景中，接收机将部分信道响应等信道信息，反馈给发送端，以供发送端根据获取的部分信道信息进行预滤波处理，使得接收机接收到时域信号中，等效信道响应的能量尽量集中在短于 CP长度的范围内，本实施例降低了接收端为了抑制干扰所需的运算复杂度和处理时延，明显降低了实现的难度。 In an application scenario in which the total channel response length is greater than the cyclic prefix (CP) length, the receiver feeds back channel information such as part of the channel response to the transmitting end, so that the transmitting end performs pre-filtering processing according to the obtained partial channel information. In the time domain signal received by the receiver, the energy of the equivalent channel response is concentrated in a range shorter than the length of the CP. This embodiment reduces the computational complexity and processing delay required by the receiving end to suppress interference, and is significantly reduced. The difficulty of implementation.

图 10为本发明另一实施例提供的干扰抑制***结构图。如图 10所示，本实施例干扰抑制***包括：发射机 101和接收机 102。 FIG. 10 is a structural diagram of an interference suppression system according to another embodiment of the present invention. As shown in FIG. 10, the interference suppression system of this embodiment includes: a transmitter 101 and a receiver 102.

发射机 101用于根据接收机 102的反馈获得部分信道信息，根据获得的部分信道信息计算预滤波系数，根据得到的所述预滤波系数对时域信号进行预滤波处理后发送，所述部分信道信息包括部分信道响应。 The transmitter 101 is configured to obtain partial channel information according to the feedback of the receiver 102, calculate a pre-filtering coefficient according to the obtained partial channel information, perform pre-filtering processing on the time domain signal according to the obtained pre-filtering coefficient, and send the partial channel. The information includes a partial channel response.

接收机 102用于通过信道估计获得时域信道响应；从所述时域信道响应抽取所述部分信道信息并向发射机 101反馈。本实施例发射机的细化结构可参见图 8对应实施例的记载，接收机的细化结构可参见图 9对应实施例的记载，基于本实施例干扰抑制***在信道响应总长度大于循环前缀（CP )长度的应用场景中，进行干扰抑制的实现原理，详见图 1-图 7对应实施例的记载，不再赘述。 The receiver 102 is configured to obtain a time domain channel response by channel estimation; extract the partial channel information from the time domain channel response and feed back to the transmitter 101. For the refinement structure of the transmitter in this embodiment, refer to the description of the corresponding embodiment in FIG. 8. The refinement structure of the receiver can be referred to the description of the corresponding embodiment in FIG. 9. The interference suppression system according to the embodiment has a total channel response length greater than the cyclic prefix. The implementation principle of the interference suppression in the application scenario of the (CP) length is shown in the description of the corresponding embodiment in FIG. 1 to FIG. 7 and will not be described again.

本发明实施例干扰抑制***可应用于正交频分复用***或釆用频域均衡的单载波***中进行干扰抑制，对于实际信道的信道响应总长度大于循环前缀长度的情形，通过在发射机对待发送的时域信号进行类似信道缩短均衡的预滤波处理，使得接收机接收到时域信号中，等效信道响应的能量尽量集中在短于循环前缀长度的范围内，因而能够显著降低接收端为了抑制干扰所需的实现复杂度，且具有艮好的干扰抑制效果。 The interference suppression system in the embodiment of the present invention can be applied to an OFDM system or a single carrier system using frequency domain equalization for interference suppression. For a case where the total channel response length of the actual channel is greater than the cyclic prefix length, The pre-filtering process similar to the channel shortening equalization is performed on the time domain signal to be transmitted, so that the energy of the equivalent channel response in the receiver receives the time domain signal as much as possible within a range shorter than the cyclic prefix length, thereby significantly reducing the reception. In order to suppress the implementation complexity required for interference, the terminal has a good interference suppression effect.

图 11 为本发明另一实施例提供的预滤波系数计算装置结构图。如图 11 所示，本实施例预滤波系数计算装置包括：预置矩阵计算单元 111、目标矩阵和约束矩阵计算单元 112和预滤波系数求取单元 1 1 3。 FIG. 11 is a structural diagram of a pre-filter coefficient calculation apparatus according to another embodiment of the present invention. As shown in FIG. 11, the pre-filter coefficient calculation device of this embodiment includes: a preset matrix calculation unit 111, a target matrix and constraint matrix calculation unit 112, and a pre-filter coefficient obtaining unit 1 13 .

预置矩阵计算单元 1 11用于根据预先获取的信道响应总长度，计算信号功率计算矩阵和干扰功率计算矩阵。 The preset matrix calculation unit 1 11 is configured to calculate a signal power calculation matrix and an interference power calculation matrix according to the total length of the channel response acquired in advance.

目标矩阵和约束矩阵计算单元 112用于根据得到的信号功率计算矩阵和干扰功率计算矩阵以及预先获取的部分信道响应，计算约束矩阵和目标矩阵。 The target matrix and constraint matrix calculation unit 112 is configured to calculate a constraint matrix and a target matrix according to the obtained signal power calculation matrix and the interference power calculation matrix and the pre-acquired partial channel response.

预滤波系数求取单元 11 3根据得到的约束矩阵和目标矩阵，计算当信干噪比为最大值时对应的预滤波系数，所述信干噪比等于信号功率和干扰噪声功率的比值。 The pre-filtering coefficient obtaining unit 11 3 calculates a pre-filtering coefficient corresponding to the maximum value of the signal-to-noise ratio according to the obtained constraint matrix and the target matrix, and the signal-to-noise ratio is equal to the ratio of the signal power to the interference noise power.

本实施例提供的预滤波系数计算装置，可应用但不限于 OFDM ***或 SC-FDE***中，用以获取预滤波系数，以便发送端或接收端根据预滤波系数对接收或发送的信号进行干扰消除；基于该预滤波系数进行干扰消除，有利于降低发送端或接收端抑制干扰的运算复杂度。 The pre-filter coefficient calculation device provided in this embodiment may be applied to, but not limited to, an OFDM system or an SC-FDE system, to obtain a pre-filter coefficient, so that the transmitting end or the receiving end interferes with the received or transmitted signal according to the pre-filtering coefficient. Elimination; interference cancellation based on the pre-filtering coefficient, which is beneficial to reduce the computational complexity of suppressing interference at the transmitting end or the receiving end.

此外，本实施例还可作为一个用于计算预滤波系数的功能模块，集成在发送端或接收端中。图 8示出了本实施例集成在发送端中的一个应用场景，该情景下，需要接收端反馈部分信道响应，所述部分信道响应可包括超过 CP 长度的部分信道响应。此外，本实施例还可集成在接收端设备（如：接收机）中，该情形下，接收端自身就可以获取到信道响应总长度和部分信道响应信息，因此，可根据获取的信息进行预滤波系数的计算。有关本实施例预滤波系数的计算装置的工作原理，可参见图 3和图 5对应实施例的记载，不再赘述。 In addition, the embodiment can also be used as a function module for calculating pre-filter coefficients, which is integrated in the transmitting end or the receiving end. FIG. 8 shows an application scenario integrated in the sending end of the embodiment. In this scenario, the receiving end is required to feed back part of the channel response, and the partial channel response may include a partial channel response exceeding the CP length. In addition, the embodiment can also be integrated in a receiving end device (for example, a receiver). In this case, the receiving end itself can obtain the total length of the channel response and part of the channel response information, and therefore, can be pre-predicted according to the obtained information. Calculation of the filter coefficient. For the working principle of the computing device of the pre-filtering coefficient of this embodiment, reference may be made to the description of the corresponding embodiment in FIG. 3 and FIG. 5, and details are not described herein again.

需要说明的是，图 8所示发射机可以为一种通信设备，该通信设备包括：信道信息获取模块、预滤波系数计算模块和预滤波处理模块。 It should be noted that the transmitter shown in FIG. 8 may be a communication device, and the communication device includes: a channel information acquisition module, a pre-filter coefficient calculation module, and a pre-filter processing module.

信道信息获取模块用于获取信道响应总长度和部分信道响应。 The channel information acquisition module is configured to acquire a total channel response length and a partial channel response.

预滤波系数计算模块用于根据所述信道响应总长度和所述部分信道响应，获取预滤波系数。 The pre-filter coefficient calculation module is configured to obtain a pre-filter coefficient according to the total length of the channel response and the partial channel response.

预滤波处理模块用于根据所述预滤波系数，对时域信号进行预滤波处理。其中，所述信道响应总长度大于循环前缀长度；所述部分信道响应包括：第一部分信道响应和 /或第二部分信道响应；所述第一部分信道响应为信道没有超出循环前缀长度部分的信道响应，所述第二部分信道响应为信道超出循环前缀长度部分的信道响应。 The pre-filtering processing module is configured to perform pre-filtering processing on the time domain signal according to the pre-filtering coefficient. The channel response total length is greater than a cyclic prefix length; the partial channel response includes: a first partial channel response and/or a second partial channel response; and the first partial channel response is a channel response in which the channel does not exceed a cyclic prefix length portion The second partial channel response is a channel response in which the channel exceeds a cyclic prefix length portion.

可选的，预滤波系数计算模块可进一步包括：预置矩阵计算单元、目标矩阵和约束矩阵计算单元和预滤波系数求取单元。 Optionally, the pre-filter coefficient calculation module may further include: a preset matrix calculation unit, a target matrix and a constraint matrix calculation unit, and a pre-filter coefficient obtaining unit.

预置矩阵计算单元用于根据所述信道响应总长度，确定信号功率计算矩阵和干扰功率计算矩阵。 The preset matrix calculation unit is configured to determine a signal power calculation matrix and an interference power calculation matrix according to the total length of the channel response.

目标矩阵和约束矩阵计算单元用于根据所述信号功率计算矩阵和干扰功率计算矩阵以及所述部分信道响应，确定约束矩阵和目标矩阵。 The target matrix and the constraint matrix calculation unit are configured to determine the constraint matrix and the target matrix based on the signal power calculation matrix and the interference power calculation matrix and the partial channel response.

预滤波系数求取单元用于根据所述约束矩阵和目标矩阵，确定当信干噪比为最大值时对应的预滤波系数，所述信干噪比等于信号功率和干扰噪声功率的比值。 The pre-filtering coefficient obtaining unit is configured to determine, according to the constraint matrix and the target matrix, a pre-filtering coefficient corresponding to a maximum value of the signal-to-noise ratio, wherein the signal-to-noise ratio is equal to a ratio of the signal power to the interference noise power.

此外，该通信设备还可以为接收机。 Furthermore, the communication device can also be a receiver.

当通信设备为发射机时，信道信息获取模块可具体用于根据接收机的反馈，获取所述信道响应总长度和部分信道响应。当通信设备为接收机时，信道信息获取模块可具体用于通过信道估计获得时域信道响应，并根据所述时域信道响应，抽取所述信道响应总长度和所述部分信道响应。该情形下，可选的，通信设备还可包括：反馈模块。反馈模块用于向发射机反馈所述信道响应总长度和所述部分信道响应。 When the communication device is a transmitter, the channel information acquisition module may be specifically configured to obtain the total channel response length and part of the channel response according to the feedback of the receiver. When the communication device is a receiver, the channel information acquisition module may be specifically configured to obtain a time domain channel response by channel estimation, and extract the total channel response length and the partial channel response according to the time domain channel response. In this case, optionally, the communication device may further include: a feedback module. The feedback module is configured to feed back the total length of the channel response and the partial channel response to the transmitter.

该通信设备可应用但不限于 OFDM或 SC-FDE ***中，用以获取预滤波系数，以便发送端或接收端根据预滤波系数对接收或发送的信号进行干扰消除；基于该预滤波系数对时域信号进行类似信道缩短均衡的预滤波处理，使得预滤波处理后的时域信号中，等效信道响应的能量尽量集中在短于循环前缀长度的范围内，相对于背景技术中的现有干扰抑制技术而言，能够显著降低接收端为了抑制干扰所需的实现复杂度，且具有很好的干扰抑制效果。 The communication device is applicable to, but not limited to, an OFDM or SC-FDE system for acquiring pre-filtering coefficients, so that the transmitting end or the receiving end performs interference cancellation on the received or transmitted signal according to the pre-filtering coefficient; The domain signal performs pre-filtering processing similar to channel shortening equalization, so that the energy of the equivalent channel response in the pre-filtered time domain signal is concentrated as much as possible within the range of the cyclic prefix length, compared to the existing interference in the background art. In terms of suppression technology, the implementation complexity required for the receiver to suppress interference can be significantly reduced, and the interference suppression effect is excellent.

本领域普通技术人员可以理解：附图只是一个实施例的示意图，附图中的模块或流程并不一定是实施本发明所必须的。 It will be understood by those of ordinary skill in the art that the drawings are only a schematic representation of one embodiment, and the modules or processes in the drawings are not necessarily required to practice the invention.

本领域普通技术人员可以理解：实施例中的装置中的模块可以按照实施例描述分布于实施例的装置中，也可以进行相应变化位于不同于本实施例的 —个或多个装置中。上述实施例的模块可以合并为一个模块，也可以进一步拆分成多个子模块。 It will be understood by those skilled in the art that the modules in the apparatus in the embodiments may be distributed in the apparatus of the embodiment according to the embodiment, or may be correspondingly changed in one or more apparatuses different from the embodiment. The modules of the above embodiments may be combined into one module, or may be further split into a plurality of sub-modules.

上述本发明实施例序号仅仅为了描述，不代表实施例的优劣。 The serial numbers of the embodiments of the present invention are merely for the description, and do not represent the advantages and disadvantages of the embodiments.

本领域普通技术人员可以理解：实现上述方法实施例的全部或部分步骤可以通过程序指令相关的硬件来完成，前述的程序可以存储于一计算机可读取存储介质中，该程序在执行时，执行包括上述方法实施例的步骤；而前述的存储介质包括： R0M、 RAM, 磁碟或者光盘等各种可以存储程序代码的介质。 A person skilled in the art can understand that all or part of the steps of implementing the above method embodiments may be completed by using hardware related to program instructions, and the foregoing program may be stored in a computer readable storage medium, and the program is executed when executed. The steps of the foregoing method embodiments are included; and the foregoing storage medium includes: a medium that can store program codes, such as a ROM, a RAM, a magnetic disk, or an optical disk.

最后应说明的是：以上实施例仅用以说明本发明的技术方案，而非对其限制；尽管参照前述实施例对本发明进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述实施例所记载的技术方案进行 ^ί'爹改，或者对其中部分技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本发明实施例技术方案的精神和范围。 It should be noted that the above embodiments are only for explaining the technical solutions of the present invention, and are not intended to be limiting; although the present invention has been described in detail with reference to the foregoing embodiments, it will be understood by those skilled in the art that: The technical solutions described in the foregoing embodiments are modified, or some of the technical features are replaced by equivalents; and the modifications or substitutions do not deviate from the spirit of the technical solutions of the embodiments of the present invention. range.

Claims

权利要求 Rights request

1、一种干扰抑制方法，其特征在于，包括： A method for suppressing interference, characterized in that it comprises:

2、根据权利要求 1所述的方法，其特征在于，所述获取信道响应总长度和部分信道响应包括： 2. The method according to claim 1, wherein the acquiring the total channel response length and the partial channel response comprises:

接收端通过信道估计获得时域信道响应，并根据所述时域信道响应，抽取所述信道响应总长度和所述部分信道响应。 The receiving end obtains a time domain channel response by channel estimation, and extracts the total length of the channel response and the partial channel response according to the time domain channel response.

3、根据权利要求 1所述的方法，其特征在于，所述获取信道响应总长度和部分信道响应包括： The method according to claim 1, wherein the acquiring the total channel response length and the partial channel response comprises:

发送端根据接收端的反馈，获取所述信道响应总长度和部分信道响应。 The transmitting end acquires the total length of the channel response and part of the channel response according to the feedback of the receiving end.

4、根据权利要求 1-3任一所述的方法，其特征在于，所述根据所述信道响应总长度和所述部分信道响应获取预滤波系数包括： The method according to any one of claims 1-3, wherein the obtaining the pre-filtering coefficient according to the total length of the channel response and the partial channel response comprises:

根据所述信道响应总长度，确定信号功率计算矩阵和干扰功率计算矩阵；根据所述信号功率计算矩阵和干扰功率计算矩阵以及所述部分信道响应，确定约束矩阵和目标矩阵； Determining a signal power calculation matrix and an interference power calculation matrix according to the total length of the channel response; determining a constraint matrix and a target matrix according to the signal power calculation matrix and the interference power calculation matrix and the partial channel response;

才艮据所述约束矩阵和目标矩阵，确定当信干噪比为最大值时对应的预滤波系数，所述信干噪比等于信号功率和干扰噪声功率的比值。 According to the constraint matrix and the target matrix, a pre-filter coefficient corresponding to a signal to interference-to-noise ratio is determined, and the signal-to-noise ratio is equal to a ratio of signal power to interference noise power.

5、根据权利要求 4所述的方法，其特征在于，所述信号功率计算矩阵为： 5. The method of claim 4 wherein said signal power calculation matrix is:

其中，表示信号功率计算矩阵，是 (M,+M₂+v) 行 X (Μ,+Λ^+ν)列的对角矩阵，对角元素为: , 为满足不等式1≤ ≤^+ ₂+!的整数,

Wherein, the signal power calculation matrix is a diagonal matrix of (M, +M ₂ +v) rows of X (Μ, +Λ^+ν) columns, and the diagonal elements are: , in order to satisfy the inequality 1 ≤ ≤ ^ + ₂ The integer of +!,

预设的预滤波的起始响应位置， Μ₂为预设的预滤波的终止响应位置；（ν + 1) 为信道响应总长度； Κ为预设的快速傅里叶变换长度。 The preset pre-filtered initial response position, Μ ₂ is the preset pre-filtered termination response position; (ν + 1) is the total length of the channel response; Κ is the preset fast Fourier transform length.

6、根据权利要求 4所述的方法，其特征在于，所述干扰功率计算矩阵为：载波间干扰功率计算矩阵和块间干扰功率计算矩阵的叠加。 The method according to claim 4, wherein the interference power calculation matrix is: a superposition of an inter-carrier interference power calculation matrix and an inter-block interference power calculation matrix.

7、根据权利要求 6所述的方法，其特征在于，所述载波间干扰功率计算矩阵为： 7. The method according to claim 6, wherein the inter-carrier interference power calculation matrix is:

其中， Α表示载波间干扰功率计算矩阵，是 + ₂+v) 行 X ( , +M₂+v 的对角矩阵，对角元素为：

Where Α denotes the inter-carrier interference power calculation matrix, which is a diagonal matrix of + ₂ + v) rows X ( , +M ₂ +v, the diagonal elements are:

/为满足不等式 -Μ^ Λ^+ν的整数， -1^为预设的预滤波的起始响应位置， Μ₂为预设的预滤波的终止响应位置；（ ν + 1) 为信道响应总长度； Κ为预设的快速傅里叶变换长度； N_g为循环前缀的长度。 / To satisfy the integer of the inequality -Μ^ Λ^+ν, -1^ is the initial pre-filtered initial response position, Μ ₂ is the preset pre-filtered termination response position; ( ν + 1) is the channel response Total length; Κ is the preset fast Fourier transform length; N _g is the length of the cyclic prefix.

8、根据权利要求 6所述的方法，其特征在于，所述块间干扰功率计算矩阵为： 8. The method according to claim 6, wherein the inter-block interference power calculation matrix is:

其中， D₂表示块间干扰功率计算矩阵，是 (M₁₊M₂+v) 行 X(M₁₊M₂+v 的对角矩阵，对角元素为：

Where D ₂ represents the inter-block interference power calculation matrix, which is (M ₁₊ M ₂ +v) row X (M ₁₊ M ₂ +v The diagonal matrix, the diagonal elements are:

/为满足不等式 -M^ / A^+v的整数， - 1^为预设的预滤波的起始响应位置， M₂为预设的预滤波的终止响应位置；（ v + 1 ) 为信道响应总长度； K为预设的快速傅里叶变换长度； N_g为循环前缀的长度。 / To satisfy the integer of the inequality -M^ / A^+v, -1^ is the initial pre-filtered initial response position, M ₂ is the preset pre-filtered termination response position; (v + 1 ) is the channel The total length of the response; K is the preset fast Fourier transform length; N _g is the length of the cyclic prefix.

9、根据权利要求 4所述的方法，其特征在于，所述目标矩阵和约束矩阵分别为： 9. The method according to claim 4, wherein the target matrix and the constraint matrix are respectively:

k： H D H₂ k: HDH ₂

— H ) ϊϊ^ — H ) ϊϊ^

其中， Α表示目标矩阵； έ表示约束矩阵， Η₂为第二部分信道响应矩阵，Where Α denotes a target matrix; έ denotes a constraint matrix, Η ₂ denotes a second partial channel response matrix,

Η₂ ^Η为第二部分信道响应矩阵的共轭转置。 Η ₂ ^Η is the conjugate transpose of the second part of the channel response matrix.

10、根据权利要求 9所述的方法，其特征在于，所述第二部分信道响应矩阵为： 10. The method according to claim 9, wherein the second partial channel response matrix is:

H₂为第二部分信道响应矩阵，是 ( M!+M₂+v+l )行 X ( Mi+M₂+1 )列的矩阵，由第二部分信道响应中元素构成，（ v + l ) 为信道响应总长度； N_g为循环前缀的长度。 H ₂ is the second partial channel response matrix, which is a matrix of (M++M ₂ +v+l ) rows of X (Mi+M ₂ +1 ) columns, which is composed of elements in the second partial channel response, ( v + l ) is the total length of the channel response; N _g is the length of the cyclic prefix.

11、一种通信设备，其特征在于，包括： A communication device, comprising:

信道信息获取模块，获取信道响应总长度和部分信道响应； a channel information acquisition module, which acquires a total channel response length and a partial channel response;

预滤波系数计算模块，根据所述信道响应总长度和所述部分信道响应，获取预滤波系数；预滤波处理模块，根据所述预滤波系数，对时域信号进行预滤波处理；其中，所述信道响应总长度大于循环前缀长度；所述部分信道响应包括：第一部分信道响应和 /或第二部分信道响应；所述第一部分信道响应为信道没有超出循环前缀长度部分的信道响应，所述第二部分信道响应为信道超出循环前缀长度部分的信道响应。 a pre-filter coefficient calculation module, which acquires a pre-filter coefficient according to the total length of the channel response and the partial channel response; The pre-filtering processing module performs pre-filtering processing on the time domain signal according to the pre-filtering coefficient; wherein, the total length of the channel response is greater than a cyclic prefix length; and the partial channel response includes: a first part of the channel response and/or the second Partial channel response; the first partial channel response is a channel response in which the channel does not exceed a cyclic prefix length portion, and the second partial channel response is a channel response in which the channel exceeds a cyclic prefix length portion.

12、根据权利要求 11所述的通信设备，其特征在于，所述通信设备具体为发射机或接收机。 12. Communication device according to claim 11, characterized in that the communication device is in particular a transmitter or a receiver.

13、根据权利要求 12所述的通信设备，其特征在于，当所述通信设备为发射机时，所述信道信息获取模块具体用于根据接收机的反馈，获取所述信道响应总长度和部分信道响应。 The communication device according to claim 12, wherein when the communication device is a transmitter, the channel information acquisition module is specifically configured to acquire the total length and part of the channel response according to feedback of the receiver. Channel response.

14、根据权利要求 12所述的通信设备，其特征在于，当所述通信设备为接收机时，所述信道信息获取模块具体用于通过信道估计获得时域信道响应，并根据所述时域信道响应，抽取所述信道响应总长度和所述部分信道响应。 The communication device according to claim 12, wherein when the communication device is a receiver, the channel information acquisition module is specifically configured to obtain a time domain channel response by channel estimation, and according to the time domain Channel response, extracting the total length of the channel response and the partial channel response.

15、根据权利要求 14所述的通信设备，其特征在于，所述通信设备还包括： The communication device according to claim 14, wherein the communication device further comprises:

反馈模块，用于向发射机反馈所述信道响应总长度和所述部分信道响应。 And a feedback module, configured to feed back the total length of the channel response and the partial channel response to the transmitter.

16、根据权利要求 11-15任一所述的通信设备，其特征在于，所述预滤波系数计算模块包括： The communication device according to any one of claims 11-15, wherein the pre-filter coefficient calculation module comprises:

预置矩阵计算单元，用于根据所述信道响应总长度，确定信号功率计算矩阵和干扰功率计算矩阵； a preset matrix calculation unit, configured to determine a signal power calculation matrix and an interference power calculation matrix according to the total length of the channel response;

目标矩阵和约束矩阵计算单元，用于根据所述信号功率计算矩阵和干扰功率计算矩阵以及所述部分信道响应，确定约束矩阵和目标矩阵； a target matrix and a constraint matrix calculation unit, configured to determine a constraint matrix and a target matrix according to the signal power calculation matrix and the interference power calculation matrix and the partial channel response;

预滤波系数求取单元，用于根据所述约束矩阵和目标矩阵，确定当信干噪比为最大值时对应的预滤波系数，所述信干噪比等于信号功率和干扰噪声功率的比值。 And a pre-filtering coefficient obtaining unit, configured to determine, according to the constraint matrix and the target matrix, a pre-filtering coefficient corresponding to a maximum value of the signal-to-noise ratio, wherein the signal-to-noise ratio is equal to a ratio of the signal power to the interference noise power.