WO2009145458A2 - Parallel connected switching converter using charge sharing - Google Patents
Parallel connected switching converter using charge sharing Download PDFInfo
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- WO2009145458A2 WO2009145458A2 PCT/KR2009/001663 KR2009001663W WO2009145458A2 WO 2009145458 A2 WO2009145458 A2 WO 2009145458A2 KR 2009001663 W KR2009001663 W KR 2009001663W WO 2009145458 A2 WO2009145458 A2 WO 2009145458A2
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/06—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider
- H02M3/07—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider using capacitors charged and discharged alternately by semiconductor devices with control electrode, e.g. charge pumps
Definitions
- the present invention relates to a switching converter, and in particular, by using a charge sharing method, continuous conduction is controlled by controlling the paralleled converters to operate in a discontinuous conduction mode (DCM) while maintaining a constant phase difference between the paralleled converters.
- DCM discontinuous conduction mode
- the present invention relates to a parallel-connected switching converter suitable for solving a problem in which switching loss and switching noise increase due to a reverse recovery phenomenon when operating in a mode).
- Converters are generally known as step-up converters, step-down converters or inverse converters. These converters can be classified as Buck converters, Boost converters, Buck-Boost converters, etc., and are used for boost converters and single-ended primary inductance converters (SEPICs). Topologies are generally known in pulsed power supplies.
- Converters are often designed and used to connect large numbers of them in parallel to provide large power.
- the present invention provides a charge / discharge circuit as compared to the synchronization signal generator of US 5,905,369.
- FIG. 1 is a block diagram of a conventional interleaved switching converter, showing the configuration of US 5,905,369.
- the above-described US 5,905,369 invention corresponds to an interleaved switching converter having a variable switching frequency.
- the first driving signal having a predetermined duty ratio with the first switching means SW1 and 12 is provided.
- a first switching converter (10) having an output and a first control driver (14) for driving said first switching means (SW1, 12); Outputting a second driving signal having a predetermined on-period and second switching means (SW2, 22) connected in parallel with the first switching converter (10), and driving the second switching means (SW2, 22).
- a second switching converter 20 having a second control driver 24; First and second capacitors (not shown); The second capacitor is discharged when the first capacitor is charged and discharged while the first capacitor is charged and discharged when the first driving signal is started. The second capacitor is charged when the first capacitor is discharged. It is composed of a charge and discharge circuit (not shown in the drawing), the output of the second drive signal of the second control driver 24 compares the voltages of the first and second capacitors by the inversion of the positive voltage difference It is an interleaved switching converter provided at the same time and provided with means for discharging to near zero voltage or near zero voltage within a period from the detection of inversion of the positive voltage difference to the start of charging at the time of discharging the first and second capacitors.
- DC (1) is an input voltage
- Co (2) is an output capacitor
- Load (3) is a load
- Voltage detector 4 is a voltage detector.
- L1, L2 and L3 are inductors made of choke coils
- SW1 and SW2 (12 and 22) are switches made of transistors
- D1 and D2 (12 and 22) are diodes
- CS1 and CS2. 15 and 25 are current detection units.
- FIG. 2 is a detailed configuration diagram of the second control driver in FIG. 1, wherein reference numeral 24-1 is a D flip-flop, 24-2 is an inverter, 24-3, 24-4 is a transistor, and 24-5 is a Logical AND, 24-6 is a transistor, 24-7 is an inverter, 24-8 is a negative logic device, 24-9, 24-10, 24-11 is a transistor, 24-12 is a comparator, 24-13 is an inverter, 24-14 is an EX_NOR device, 24-15 is a comparator, and 24-16 is an RS flip-flop.
- 24-1 is a D flip-flop
- 24-2 is an inverter
- 24-3, 24-4 is a transistor
- 24-5 is a Logical AND
- 24-6 is a transistor
- 24-7 is an inverter
- 24-8 is a negative logic device
- 24-11 is a transistor
- 24-12 is a comparator
- 24-13 is an inverter
- 24-14 is an EX_NOR device
- Boost 1 the first switching converter 10, switches on when the L1 current becomes zero and detects the SW1 current.
- the control signal Vdr1 is turned off.
- Boost 2 which is the second switching converter 20, receives SW1 control signal Vdr1 of Boost 1 and turns on SW2 by 180 degrees out of phase with Vdr1 using a charge / discharge circuit. Generate a control signal Vdr2 to turn off.
- Figure 3 is a waveform diagram showing the operation of each part in Figures 1 and 2
- Figure 4 is an operational waveform diagram showing a problem of the conventional interleaved switching converter.
- Vdr1 is a waveform of the first drive signal output from the first control driver 14
- Vc1 and (3) Vc2 are waveforms showing charge and discharge
- Vdr2 is The waveform of the second drive signal output from the second control driver 24,
- Ii1 is the waveform of the input current input to the first control driver 14
- Ii2 is the waveform of the second control driver 24 This is the waveform of the input current.
- the control driver # 2 receives a Vdr1 signal and receives a current source.
- C1 and C2 are charged and discharged using I1, I2, I3, and I4 to generate a Vdr2 signal that turns on SW2 at the point where Vc1 and Vc2 are equal. Therefore, according to the aforementioned US 5,905,369, when two converters are connected in parallel, when I1, I2, I3, and I4 are the same and C1 and C2 are the same, Vdr1 and Vdr2 are 180 degrees as shown in the waveform diagram shown in FIG. The phase difference can be maintained.
- Boost2 does not detect the point where the current of L2 becomes zero unlike Boost1. If the device value of L2 is larger than L1, the slope of the current is lowered so that L2 SW2 is turned on before the current becomes zero, and it operates as CCM as shown in the waveform diagram of FIG. 4. In the case of operating as CCM, switching loss and switching noise of SW2 increases due to the reverse recovery of D2. Will be.
- boost2 also needs a circuit that detects the point where the L2 current becomes zero. If SW2 is turned on at the point where L2 current becomes zero even under conditions where L1 and L2 are not equal, the period of boost1 and boost2 will not match. Therefore, a problem arises in that the ripple of the input current and the output current increases, and a circuit for controlling the period and phase of Vdr1 and Vdr2 is required to be the same.
- the present invention has been proposed to solve the above-mentioned general problems.
- the present invention is a charge sharing method using both of the parallel connected converters while maintaining a constant phase difference of the parallel connected converter DCM (discontinuous conduction mode)
- the main technical problem is to provide a parallel-connected switching converter that can solve the problem that switching loss and switching noise increase due to reverse recovery phenomenon when operating in continuous conduction mode (CCM) by controlling to operate at the same time. .
- a parallel-connected switching converter that adjusts off to maintain a constant phase difference between converters, comprising: a first switching controller for generating a first switching control signal for controlling the turn-on and turn-off of the first converter; A first capacitor and a second capacitor, and conducting charge sharing between the first capacitor and the second capacitor by conducting the first capacitor and the second capacitor according to the first switching control signal of the first switching controller.
- a synchronization signal generator for adjusting a voltage of the first capacitor and generating a synchronization signal having a predetermined phase difference in synchronization with the first switching control signal using the voltage of the first capacitor; And a second switching controller for generating a second switching control signal according to the synchronization signal generated by the synchronization signal generator and controlling the turn-on and turn-off of the second converter.
- the synchronization signal generator is more specifically, a first capacitor; A second capacitor connected to the first capacitor through a switch element and connected to enable charge sharing with the first capacitor by being closed (ON) of the switch element; A comparator configured to generate a synchronization signal by comparing the voltage of the first capacitor with a reference voltage; And a charge / discharge unit connected to the first capacitor to charge or discharge the first capacitor by the synchronization signal and the first switching control signal. More preferably, the synchronization signal is generated.
- the unit may further include a short pulse generator for receiving the first switching control signal and converting the short switching pulse into a short pulse having a shorter HIGH period than the first switching control signal.
- the present invention as described above may further include a technical configuration for matching the period of the synchronization signal and the second switching control signal generated as described above, for this purpose, the second switching control unit, the second switching control A phase comparator for receiving a feedback signal and comparing a phase of the sync signal generated by the sync signal generator with a phase of the second switching control signal to output a phase error according to a phase difference; A control voltage adjusting unit which generates a driving control voltage by adjusting a reference control voltage according to the phase error output from the phase comparing unit; And a switching control signal generator configured to generate a second switching control signal by using the driving control voltage generated by the control voltage regulator.
- the reference control voltage used in the control voltage adjusting unit as described above it is preferable to use a voltage adjusted so that the detected voltage is equal to the reference voltage based on the voltage detected by the voltage detector installed in the output terminal.
- a current mode pulse width modulation (PWM) method for controlling the converter switching time by detecting a current flowing in the converter or using a reference voltage source A voltage mode PWM method for controlling the switching time point by comparing the driving control voltage may be used. That is, in the present invention, the switching control signal generation unit compares the driving control voltage generated by the control voltage generation unit with the voltage applied to the switching sensing resistor by the current passing through the converter switch included in the second converter.
- a current mode PWM method for generating a switching control signal, or further includes a ramp generator for generating a ramp waveform, and compares the driving control voltage generated by the control voltage generator with the ramp voltage to generate a second switching control signal. It is also possible to apply a voltage mode PWM method to generate.
- the first zero current detector for detecting whether the current flowing into the first converter is zero; And a second zero current detector for detecting whether the current flowing into the second converter becomes zero, wherein the first and second switching controllers respectively supply inflow currents from the first and second zero current detectors.
- the first and second switching controllers respectively supply inflow currents from the first and second zero current detectors.
- the parallel-connected converters are controlled to operate in a discontinuous conduction mode (CCM) while maintaining the phase difference of the parallel-connected converters constant. It is possible to solve the problem that the switching loss and the switching noise are increased by the reverse recovery phenomenon that occurs when operating in the).
- CCM discontinuous conduction mode
- FIG. 1 is a block diagram of a conventional interleaved switching converter.
- FIG. 2 is a detailed configuration diagram of the second control driver in FIG. 1.
- FIGS. 3 is a waveform diagram showing the operation of each part in FIGS.
- FIG. 4 is an operation waveform diagram showing a problem of a conventional switching converter.
- FIG. 5 is a block diagram of a switching converter according to an embodiment of the present invention.
- FIG. 6 is a detailed configuration diagram of the first switching control unit in FIG. 5.
- FIG. 7 is a detailed configuration diagram of the voltage detector in FIG. 5.
- FIG. 8 is a detailed configuration diagram of the second switching control unit in FIG. 5.
- FIG. 9 is a detailed configuration diagram of a synchronization signal generator in FIG. 8.
- FIG. 10 is a timing chart of the waveform of each signal in the second switching controller of the present invention.
- FIG. 11A is a detailed configuration diagram showing another configuration example of the synchronization signal generator in FIG. 8, and FIG. 11B includes a synchronization signal generator implemented as in FIG. A timing chart of the waveforms of the signals in the second switching controller.
- FIG. 12 is a block diagram of a switching converter according to another embodiment of the present invention.
- FIG. 13 is a detailed configuration diagram of the first switching control unit in FIG. 12.
- FIG. 14 is a detailed configuration diagram of the second switching control unit in FIG. 12.
- 15 is a block diagram illustrating a parallel-connected switching converter when two or more converters are connected in parallel according to another embodiment of the present invention.
- FIG. 16 is a timing chart for the waveform of each signal in the embodiment shown in FIG.
- 17 is a block diagram illustrating a parallel-connected switching converter when two or more converters are connected in parallel according to another embodiment of the present invention.
- FIG. 18 is a timing chart for the waveform of each signal in the embodiment shown in FIG.
- FIG. 19 is a block diagram showing a parallel-connected switching converter using an AC power supply and a current mode PWM according to another embodiment of the present invention.
- FIG. 20 is a block diagram illustrating a parallel-connected switching converter using an AC power supply and a voltage mode PWM according to another embodiment of the present invention.
- first switching control unit 200 second converter
- control voltage regulator 247 switching control signal generator
- 5 to 13 show a first embodiment of the parallel-connected switching converter of the present invention, which is supplied with DC power as the converter input voltage, uses current mode PWM, An example of using a converter is shown.
- FIG. 5 is a block diagram of a parallel-connected switching converter using a current mode PWM according to an embodiment of the present invention.
- the parallel-connected switching converter provided by the present invention is basically a power input 101. It can be seen that a plurality of converters including the first converter 100 and the second converter 200 are connected in parallel.
- the converter exemplifies a boost converter which is a kind of a switching mode power supply (SMPS) having a boost circuit so as to obtain a large output voltage compared to an input voltage.
- SMPS switching mode power supply
- Other types of converters can be applied without particular limitations.
- the first converter 100 and the second converter 200 are basically inductors L1 and L2, diodes D1 and D2, switches 120 and 220, and filter capacitors.
- C0 C0
- MOSFETs may be most preferably used as the switch 120 and 220 elements.
- other types of transistor elements such as BJT and IGBT may be used in some cases.
- the present invention relates to a technique for reducing the ripple of the input current and the output current by allowing a plurality of converters connected in parallel as described above to have a switching operation with a constant phase difference, the present invention as described above
- the most important technical feature is to apply a method of charge sharing to generate a control signal having a constant phase difference even when the switching frequency is varied.
- the present invention includes a synchronization signal generator 244 which generates a synchronization signal having a predetermined phase difference while synchronizing with the first switching control signal Vdr1 of the first converter by charge sharing.
- the generating unit 244 basically includes a first capacitor C1 and a second capacitor C2, and the first capacitor C1 and the first capacitor C1 according to the first switching signal Vdr1 of the first switching controller 140.
- the second capacitor C2 conducts to regulate the voltage of the first capacitor through charge sharing between the first capacitor and the second capacitor, and the voltage Vc1 of the adjusted first capacitor C1 is adjusted as described above. Is synchronized with the first switching control signal Vdr1 to generate a synchronization signal having a constant phase difference.
- the synchronization signal generator 244 is a first capacitor connected in parallel Charge and discharge unit (244-10) connected to (C1) and the second capacitor (C2), the first capacitor (C1) to charge / discharge the first capacitor (C1) and / or the second capacitor (C2), And a comparator 244-1 generating the synchronization signal Sync by comparing the voltage of the first capacitor C1 with a reference voltage.
- the first capacitor (C1) and the second capacitor (C2) is configured to be selectively connected by an external control signal by being connected through the switch element 244-6 as shown, wherein the first The switch element 244-6 connecting the first capacitor C1 and the second capacitor C2 is turned on / off by the first switching control signal Vdr1 of the first control driver 110 for a predetermined time. It is possible to control the voltage of the first capacitor C1 by sharing charge.
- the short pulse generator 244-2 adjusts the duty ratio of the first switching control signal Vdr1 to high. It is more preferable in terms of phase difference control because it is possible to arbitrarily control the charging time and the amount of charge of the second capacitor C2 when converting to a short pulse SP having a short duration of HIGH.
- the voltage Vc1 applied to the first capacitor C1 and the predetermined reference voltage Vref are applied. Is compared through the comparator 244-1 to generate a synchronization signal Sync when the signal level becomes HIGH when the two voltages are the same, and outputs it as a signal for adjusting the second switching control signal Vdr2.
- the sync signal Sync generated as described above may be input to the charge / discharge unit 244-10 to be used as a signal for controlling charge / discharge operations of the capacitors C1 and C2.
- the charge / discharge unit 244-10 is connected to one end of the first capacitor C1 in the synchronization signal generator 244 of the present invention as shown in FIG. It is a component to charge and discharge electric charges to the two capacitors (C2).
- the charging and discharging unit 244-10 may operate, for example, as the charging current power source I1 and the discharging current power source I2 and the charging and discharging current power sources I1 and I2 as in the illustrated embodiment.
- an RS flip-flop 244-3 for controlling the charge / discharge unit 244-10.
- the charge / discharge unit 244-10 is converted into a sync signal and a short pulse generated by the comparator 244-1.
- the charging / discharging operation can be controlled by the first switching control signal SP.
- the synchronization signal generator 244 is illustrated in a form included in a circuit of the second switching controller 240 without separately configuring a circuit for simplifying the circuit. It is also possible to implement as a circuit separate from the two switching control unit 240 separately.
- the second converter 200 when controlling the turn-on and turn-off of the second converter 200, the second converter 200 also detects a time when the current Ii2 flowing in the second inductor L2 becomes zero. And a second zero current detector 210 for turning on and then turning on after the inflow current becomes zero, so that the second converter 200 performs DCM operation similarly to the first converter 100.
- the first switching controller 140 included in the first converter 100 electromagnetically couples the first inductor L1 element generally provided at the power input terminal of the boost circuit.
- the first induction coil L3 is further provided to constitute a first zero current detector 110 capable of detecting an inflow current by the voltage Vzcd1 induced in the first induction coil L3.
- the first switching controller 140 receives the Vzcd1 signal, compares it with a predetermined reference voltage Vth, and at the moment when the Vzcd1 signal falls below the reference voltage Vth.
- the converter switch is turned on by determining that the inflow current is zero and making the first switching control signal (hereinafter, Vdr1 signal) high.
- the first switching controller 140 outputs the voltage Vi1 applied to the switching sensing resistor CS1 by the turn-on of the converter switch and the voltage detector 104 installed at the power output terminal.
- the converter switch is turned off by making the Vdr1 signal low when the Vi1 voltage rises above the Ve voltage in response to the reference control voltage Ve.
- the voltage detector 104 detects the output voltage V0 using the resistor dividers R1 and R2 as shown in FIG. 7, and the reference control voltage Ve so that the detected voltage becomes the reference voltage Vo_Ref.
- the feedback is supplied to the first switching controller 140 and the second switching controller 240 by adjusting.
- the second switching controller 240 the voltage is induced by the second induction coil L4 of the second zero current detector 210.
- the second switching control signal hereinafter referred to as the 'Vdr2 signal'
- the switch sensing voltage Vi2 caused by the current passing through the converter switch is higher than the drive control voltage Vc.
- the Vdr2 signal goes low.
- the present invention when turning on the second converter, unlike the conventional parallel-connected switching converter detects the time when the current becomes zero in the second converter 200 as well as the first converter 100. When the zero current is detected, the converter switch is turned on so that both the first converter and the second converter operate as DCM. Also, in the present invention, when the current of the second converter becomes zero, the first switching control signal Vdr1 controls the operation of the first converter 100 in the case where the periods between the converters are not matched by turning on the switch.
- an operation process of controlling the second converter by the synchronization signal generator 244 and the synchronization signal generated by the synchronization signal generator 244 is the most characteristic configuration of the present invention. Referring to 10 in more detail as follows.
- the synchronization signal generator 244 receives a Vdr1 signal and converts it into a short pulse SP having a short HIGH period through the short pulse generator 244-2.
- the Vdr1 signal (that is, the SP signal) converted into the short pulse as described above turns on the switch 244-6 connecting the first capacitor C1 and the second capacitor C2, and at the same time, the RS flip-flop S44-. 3) the charge switch 244-4 connected to the reset terminal of the charging current source I1 and the first capacitor C1 is turned on, and the switch 244-5 connected to the discharge current source I2 is turned off. .
- capacitor switch 244-6 goes OFF and switch 244-4 remains ON because the output of RS flip-flop 244-3 does not change. Therefore, in this state, the charging current source I1 continuously charges only the first capacitor C1, so that the voltage Vc1 applied to the first capacitor C1 continues to rise, and the second capacitor C2 stops charging by turning off the capacitor switch 244-6. The final voltage at the moment the switch is broken is maintained. (Section 'b' in the waveform diagram of Fig. 10)
- the comparator CMP1 signal becomes HIGH and the synchronization signal (hereinafter referred to as a 'sync signal') is made high.
- the Sync signal generated as described above is output to the phase comparator 245 to be described later and input to the SET terminal of the flip-flop 244-3. Accordingly, the charge switch 244-4 is turned off and the discharge switch 244-5 is turned off. Will light up. As described above, when the discharge switch 244-5 is turned on, the discharge current source I2 discharges C1. As a result, the Vc1 voltage continuously decreases and the Sync signal falls back to LOW. ('C' section in the waveform diagram of Figure 10)
- Vdr1 and Sync have a phase difference of 180 degrees, and by adjusting I1 and I2, C1 and C2, the phase difference can be freely adjusted to an arbitrary value. .
- FIG. 11 illustrates another implementation manner of the synchronization signal generator 244 of the present invention as described above.
- the terminal and the terminal of the RS flip-flop are switched so that the charge switch 244-4 is turned on by the sync signal to start charging to the first capacitor C1, and the discharge switch 244 is performed by the short pulse SP.
- the charge switch 244-4 is turned on by the sync signal to start charging to the first capacitor C1
- the discharge switch 244 is performed by the short pulse SP.
- -5 is turned on to discharge C1.
- FIG. 11B shows signal waveforms for the respective parts of the parallel-connected switching converter shown in FIG. 11A, and each waveform shown in FIG. 11B is shown in FIG. In comparison with, it can be seen that the waveforms of (1) Vc2, (2) Vc1 and (3) Vref are inverted only up and down in the waveform of FIG.
- the sync signal generated by the sync signal generator 244 has a phase difference of 180 degrees with the switching control signal Vdr1 of the first converter, and once such a sync signal (Sync) is secured, the second sync signal is used.
- Controlling the switching control signal Vdr2 of the converter to generate a signal having a constant phase difference from Vdr1 can be relatively easily implemented at the level of those skilled in the art.
- a circuit for synchronizing the period of Vdr2 to the Sync signal using phase comparison will be described as an example of the circuit for controlling the Sync signal and Vdr2 to be synchronized.
- the sync signal generated by the sync signal generator 244 is input to the phase comparator 245 to control the Vdr2 signal to be synchronized with the sync signal. That is, the phase comparator 245 receives the second switching control signal Vdr2 as a feedback input as shown in FIG. 8, and compares the phases of the sync signal and the Vdr2 signal input from the sync signal generator 244 to Vdr2. Phase error signal is generated and output depending on whether the signal is in phase lag or phase lead.
- the phase comparison unit functions similar to the 74HC4046 phase lock loop (PLL).
- phase error generated by the phase comparator 245 is input to the control voltage controller 246, and the control voltage controller 246 is supplied from the phase comparator 245 to match the phase of Vdr2 and Sync. Based on the input phase error, the predetermined reference control voltage Ve is adjusted to generate a new driving control voltage Vc.
- the control voltage controller 246 lowers the Vc voltage by adjusting the Ve voltage in response to the phase delay signal in order to increase the frequency.
- the Vc voltage is increased by adjusting the Ve voltage to decrease the frequency.
- the reference control voltage Ve used in the control voltage adjusting unit 246 may preferably be a voltage output from the voltage detecting unit 104 connected to the power output terminal. Referring to FIG. Since the above description has been made, repeated descriptions will be omitted.
- the switching control signal generation unit 247 of FIG. 8 the second switching control for driving the switch of the second converter 200 based on the driving control voltage Vc generated by the control voltage adjusting unit 246 as described above. Generate the signal Vdr2.
- the switching control signal generator 247 compares the driving control voltage Vc input from the control voltage adjusting unit 246 with a predetermined turn-off reference voltage Vi2 to generate an RS flip-flop 243. ) To turn off the second converter switch SW2 by making the Vdr2 signal low.
- Vdr2 in determining the time of turning off the second converter switch SW2 as described above, Vdr2 is compared to the Sync signal by phase comparison between the Sync signal following the Vdr1 signal and the second switching control signal Vdr2.
- the period of Vi2 rises and becomes the same as Vc, resulting in a shorter period of Vdr2.
- the point at which Vc becomes equal to Vc is delayed, and the period of Vdr2 becomes longer, thereby automatically synchronizing with Vdr1.
- the first switching controller 140 may provide a ramp signal. It is provided with a ramp generator 144 that generates, and compares the output voltage Vr signal of the ramp generator 144 and the voltage detector voltage Ve instead of the switching sensing voltage Vi1 to determine the time to make the Vdr1 signal from HIGH to LOW Able to know.
- the second switching controller 240 compares the output voltage Vr signal of the ramp generator 248 with the driving control voltage Vc instead of Vi2 to determine a time point at which the Vdr2 signal is made from HIGH to LOW. .
- the other parts operate as in the case of Figs. 5 to 11 using the current mode PWM, and the repetitive description thereof will be omitted.
- 15 to 18 show an example using three or more converters 100 to 300 as compared with the first embodiment described above.
- the second switching controller 240 is not configured as a separate circuit to simplify the circuit in configuring the second synchronization signal generator and the third synchronization signal generator.
- the second synchronous signal generator and the third synchronous signal generator are shown in the drawings. Not shown separately.
- the basic circuits of the second switching controller 240 and the second synchronous signal generator are two converters as described above. Is the same as when connected in parallel, and in the same circuit as shown in FIG. 10 in the synchronizing signal generation unit (included in the second switching control unit), if I1 is doubled to I2, the second converter as shown in the waveform of FIG. Is operated while maintaining a 120 degree phase difference with the first converter.
- the third switching controller 340 when the third switching controller 340 also receives the Vdr2 signal and operates in the same manner, the third converter 300 operates while maintaining a 120 degree phase difference from the second converter 200. As a result, the three converters The module operates while maintaining a 120 degree phase difference.
- a k-synchronous signal generation unit including a first capacitor and a second capacitor and generating a synchronization signal by charge sharing as described above for the control driving of the k-th converter, the k-synchronous signal generation
- the controller shares the charges of the first capacitor and the second capacitor with the k-1 switching control signal for a predetermined time to adjust the voltage of the first capacitor and the synchronization signal based on the voltage of the first capacitor. It is configured to generate.
- the k-1 th when the N converters are connected in parallel, the k-1 th based on the k-1 switching control signal of the adjacent k-1 switching control unit in generating the synchronization signal in each converter
- the technical feature is that the k-th synchronizing signal having a predetermined phase difference with the switching control signal is generated and the signal for synchronizing control of the k-th converter is generated using the k-th synchronizing signal.
- 17 and 18 show an example implemented in a somewhat different manner from the third embodiment shown in FIG. 15 in the case where three converters are connected in parallel.
- the circuit of the second switching controller 240 is basically the same as that of FIG. 15, but the third synchronization signal generator included in the third switching controller 340 is different from the case of FIG. 15.
- the signal of Vdr1 is input and generates a signal having synchronization with the first converter 100 while having a constant phase difference.
- the second synchronous signal generator makes I1 twice the I2
- the third synchronous signal generator makes I2 twice the I11
- the second converter makes a 120 degree phase difference from the first converter.
- the third converter operates while maintaining a phase difference of 240 degrees with the first converter. In this way, the three modules operate with a 120-degree phase shift.
- the above method may be applied even when three or more N converters are connected in parallel.
- a k-synchronous signal generation unit including a first capacitor and a second capacitor and generating a synchronization signal by charge sharing as described above for the control driving of the k-th converter, the k-synchronous signal generation The unit is configured to receive the first switching control signal Vdr1 of the first switching control unit to adjust the voltage of the first capacitor based on the Vdr1 signal and generate the synchronization signal based on the voltage of the first capacitor in generating the synchronization signal.
- the first switching control signal Vdr1 of the first switching control unit is commonly received in generating the synchronization signal in each converter, and the synchronization signal is based on the Vdr1 signal.
- the technical characteristics of the method are to generate the N, and according to this method, the N converter modules can operate while maintaining a constant phase difference from each other.
- the converter input voltage is a DC voltage, and the same may be controlled in the case of an AC power source.
- an input voltage is changed from a DC power source to an AC power source, and a BD (bridge diode) 105 is added as a DC power supply unit for converting an AC power source to a DC power source, that is, a power factor.
- BD bridge diode
- the input voltage described above is exactly the same as for direct current.
- a buck or other type of switching mode power supply (SMPS) topology generates a synchronization signal using charge sharing as described above and controls a plurality of converters using the same. By applying this, the switching phase difference of each paralleled converter can be kept constant.
- SMPS switching mode power supply
- the present invention can be suitably applied to a switching mode power supply using a plurality of converters connected in parallel to secure a large output.
- the switching mode power supply to which the present invention is applied has an advantage in that all of the parallel-connected converters operate in a discontinuous conduction mode (DCM), thereby effectively preventing the generation of switching loss and switching noise caused by reverse recovery.
- DCM discontinuous conduction mode
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Abstract
The present invention relates to a parallel connected switching converter in which the turn-on and turn-off operations of a plurality of converters connected in parallel with each other are controlled to keep the phase difference between the converters constant. The parallel connected switching converter of the present invention includes: a first switching control unit for generating a first switching control signal for controlling the turn-on and turn-off operations of a first converter; a synchronous signal generating unit provided with a first capacitor and a second capacitor, interconnecting the first capacitor and the second capacitor in accordance with a first switching control signal output from the first switching control unit to control the voltage of the first capacitor through charge sharing between the first capacitor and the second capacitor, and generating a synchronous signal which is synchronous to the first switching control signal and having a predetermined phase difference by using the voltage of the first capacitor; and a second switching control unit for generating a second switching control signal in accordance with the synchronous signal output from the synchronous signal generating unit to control the turn-on and turn-off operations of the second converter. The parallel connected switching converter of the present invention keeps the phase difference between parallel connected converters constant by using a charge sharing scheme, and controls all of the parallel connected converters to operate in a DCM (discontinuous conduction mode), thereby being capable of preventing an increase in switching loss and switching noise caused by reverse recovery resulting from their operation in a CCM (continuous conduction mode).
Description
본 발명은 스위칭 컨버터에 관한 것으로, 특히 전하 공유(charge sharing) 방법을 사용하여 병렬 연결된 컨버터들 간의 위상차를 일정하게 유지하면서 병렬 연결된 컨버터 모두 DCM(discontinuous conduction mode)으로 동작하게 제어함으로써 CCM(continuous conduction mode)으로 동작할 경우에 발생하는 역회복 현상에 의해 스위칭 손실과 스위칭 노이즈가 증가하는 문제점을 해결하기에 적당하도록 한 병렬연결 스위칭 컨버터에 관한 것이다.BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching converter, and in particular, by using a charge sharing method, continuous conduction is controlled by controlling the paralleled converters to operate in a discontinuous conduction mode (DCM) while maintaining a constant phase difference between the paralleled converters. The present invention relates to a parallel-connected switching converter suitable for solving a problem in which switching loss and switching noise increase due to a reverse recovery phenomenon when operating in a mode).
일반적으로 컨버터는 스텝-업(step-up) 컨버터, 스텝-다운(step-down) 컨버터 또는 인버스(inverse) 컨버터로서 알려져 있다. 이러한 컨버터는 Buck 컨버터(강압 변환기), Boost Converter, Buck-Boost 컨버터(승강압 변환기) 등의 종류가 있으며, 부스트(boost) 컨버터 및 SEPIC(single-ended primary inductance converter, 단일 단부 주 인덕턴스 컨버터)의 토폴로지(topology)들은 일반적으로 펄스 전력 공급기들에서 공지되었다.Converters are generally known as step-up converters, step-down converters or inverse converters. These converters can be classified as Buck converters, Boost converters, Buck-Boost converters, etc., and are used for boost converters and single-ended primary inductance converters (SEPICs). Topologies are generally known in pulsed power supplies.
이러한 토폴로지를 이용하여 큰 파워의 컨버터를 구성하는 경우, 컨버터를 구성하는 소자의 정격이 상승하여 적당한 소자를 손쉽게 구하기 어려우며, 손실의 증가와 설계의 어려움 등이 있는바, 이를 해결하기 위해 작은 파워의 컨버터를 설계하고 이들을 여러 개 병렬로 연결하여 큰 파워를 공급하도록 사용하는 경우가 많다. When a converter with a large power is constructed using such a topology, it is difficult to easily find a suitable device due to an increase in the rating of the devices constituting the converter, and there is an increase in losses and design difficulties. Converters are often designed and used to connect large numbers of them in parallel to provide large power.
한편, 상기와 같이 병렬 연결된 컨버터를 구동하는 경우, 입력 전류의 리플과 출력 전류의 리플을 줄임으로써 입력 필터와 출력 필터의 크기를 감소시킬 필요가 있으며, 이에 대한 방편으로서 병렬 연결된 컨버터의 스위치 제어 신호가 일정한 위상 차이를 가지도록 제어하는 방법이 널리 사용되고 있다. 그런데, 스위칭 주파수가 고정된 경우라면 일정한 위상 차이를 가지는 제어 신호를 생성하는 것에는 기술적으로 큰 어려움이 없지만 스위칭 주파수가 가변되는 경우에는 스위칭 주기가 변하므로 일정한 위상 차이를 가지는 신호를 생성하는 것이 용이하지 않다.On the other hand, when driving the parallel-connected converter as described above, it is necessary to reduce the size of the input filter and the output filter by reducing the ripple of the input current and the output current, as a way to switch control signal of the converter connected in parallel Has been widely used to control the phase difference. However, if the switching frequency is fixed, it is not technically difficult to generate a control signal having a constant phase difference, but when the switching frequency is variable, the switching period is changed, so it is easy to generate a signal having a constant phase difference. Not.
이와 관련하여, 본 발명 이전에 공지된 기술로는 미국등록특허 US 5,905,369호를 들 수 있는데, 상기 선등록 특허에 의하면, 충방전 회로를 사용하여 boost1 스위치 ON 이후 180도 지점에서 boost2의 스위치를 ON 함으로써 L1 값과 L2 값이 같을 경우 boost1과 boost2 모두 DCM(discontinuous conduction mode)으로 동작하면서 180도 위상 차이를 유지할 수 있도록 하고 있다.In this regard, known techniques prior to the present invention include US Patent US 5,905,369. According to the pre-registered patent, the switch of boost2 is turned on at 180 degrees after the boost1 switch is turned on by using the charge / discharge circuit. Thus, when L1 and L2 are the same, both boost1 and boost2 operate in discontinuous conduction mode (DCM) while maintaining a 180 degree phase difference.
하지만, 상기한 미국등록특허에 따르면, L2의 값이 L1의 값보다 클 경우에 L2의 전류가 영(0)이 되기 전에 다시 상승하여 CCM(continuous conduction mode)으로 들어갈 수 있으며, 이 경우 D2의 역회복(reverse recovery) 현상에 의해 스위치의 손실이 증가하고 노이즈가 심해진다는 문제가 있었다.However, according to the above-mentioned US patent, when the value of L2 is greater than the value of L1, the current of L2 may rise again before it becomes zero and enter the continuous conduction mode (CCM), in which case D2 There is a problem that the loss of the switch increases and the noise increases due to reverse recovery.
따라서, Boost1에 동기를 맞추면서 L2 전류가 영(zero)이 된 후에 스위치를 ON 하도록 제어하는 회로가 필요하다고 할 수 있는바, 이에 본 발명은 상기한 US 5,905,369호의 동기신호 발생회로에 비해 충방전 회로가 줄어서 간단하게 회로를 구현할 수 있음과 동시에, L1, L2 값이 서로 다른 경우에도 DCM을 유지하면서 위상 차이를 일정하게 유지할 수 있는 방법을 제공하고자 하는 것이다.Accordingly, it can be said that a circuit for controlling the switch to be turned on after the L2 current becomes zero while synchronizing with Boost1 is required. Accordingly, the present invention provides a charge / discharge circuit as compared to the synchronization signal generator of US 5,905,369. In addition, it is possible to simplify the circuit implementation by reducing the number and to provide a method of maintaining the phase difference while maintaining DCM even when L1 and L2 values are different.
이하, 상기한 US 5,905,369호에서 나타난 문제점 및 후술하는 본 발명에 대한 이해를 돕기 위해 상기한 US 5,905,369호의 구성 및 동작에 대하여 더욱 상세히 설명하면 다음과 같다.Hereinafter, the configuration and operation of the above-described US Pat. No. 5,905,369 will be described in more detail to help understand the problems shown in US Pat. No. 5,905,369 and the present invention described below.
도1은 종래 인터리브 방식 스위칭 컨버터의 블록구성도로서, US 5,905,369호의 구성을 보인 것이다. 도1에 도시된 바와 같이, 상기한 US 5,905,369호 발명은 스위칭 주파수가 변동하는 인터리브 방식 스위칭 컨버터에 해당하는 것으로서, 제 1 스위칭 수단(SW1, 12)과 소정의 듀티비를 가지는 제 1 구동 신호를 출력하고, 상기 제 1 스위칭 수단(SW1, 12)을 구동하는 제 1 제어 구동부(14)를 가지는 제 1 스위칭 컨버터(10)와; 상기 제 1 스위칭 컨버터(10)와 병렬로 접속되는 제 2 스위칭 수단(SW2, 22)과 소정의 온 기간을 가지는 제 2 구동 신호를 출력하고, 상기 제 2 스위칭 수단(SW2, 22)을 구동하는 제 2 제어 구동부(24)를 가지는 제 2 스위칭 컨버터(20)와; 제 1 및 제 2 콘덴서(도면상에 미도시)와; 상기 제 1 구동 신호의 기동에 의해 상기 제 1 콘덴서를 충방전하는 것과 동시에 상기 제 1 콘덴서를 충전하는 때에는 상기 제 2 콘덴서를 방전하고, 상기 제 1 콘덴서를 방전하는 때에는 상기 제 2 콘덴서를 충전하는 충방전 회로(도면상에 미도시)로 구성되어 있으며, 상기 제 2 제어 구동부(24)의 제 2 구동 신호의 출력은 상기 제 1 및 제 2 콘덴서의 전압을 비교하고 양전압차의 반전에 의해 동시에 일어나게 행해지고, 아울러 제 1, 제 2 콘덴서의 방전시에 있어서 양 전압차의 반전 검출로부터 충전 개시까지의 기간 내에 영 전압 또는 영 전압 근처까지 방전하는 수단을 설치한 인터리브 방식 스위칭 컨버터이다.Figure 1 is a block diagram of a conventional interleaved switching converter, showing the configuration of US 5,905,369. As shown in FIG. 1, the above-described US 5,905,369 invention corresponds to an interleaved switching converter having a variable switching frequency. The first driving signal having a predetermined duty ratio with the first switching means SW1 and 12 is provided. A first switching converter (10) having an output and a first control driver (14) for driving said first switching means (SW1, 12); Outputting a second driving signal having a predetermined on-period and second switching means (SW2, 22) connected in parallel with the first switching converter (10), and driving the second switching means (SW2, 22). A second switching converter 20 having a second control driver 24; First and second capacitors (not shown); The second capacitor is discharged when the first capacitor is charged and discharged while the first capacitor is charged and discharged when the first driving signal is started. The second capacitor is charged when the first capacitor is discharged. It is composed of a charge and discharge circuit (not shown in the drawing), the output of the second drive signal of the second control driver 24 compares the voltages of the first and second capacitors by the inversion of the positive voltage difference It is an interleaved switching converter provided at the same time and provided with means for discharging to near zero voltage or near zero voltage within a period from the detection of inversion of the positive voltage difference to the start of charging at the time of discharging the first and second capacitors.
여기서, DC(1)는 입력전압, Co(2)는 출력 커패시터이며, Load(3)는 부하이고, Voltage detector(4)는 전압 검출부이다. 또한, L1, L2, L3(11, 21)은 초크코일로 이루어진 인덕터이고, SW1, SW2(12, 22)는 트랜지스터로 이루어진 스위치이며, D1, D2(12, 22)는 다이오드이고, CS1, CS2(15, 25)는 전류 검출부이다.Here, DC (1) is an input voltage, Co (2) is an output capacitor, Load (3) is a load, and Voltage detector 4 is a voltage detector. In addition, L1, L2 and L3 (11 and 21) are inductors made of choke coils, SW1 and SW2 (12 and 22) are switches made of transistors, and D1 and D2 (12 and 22) are diodes and CS1 and CS2. 15 and 25 are current detection units.
또한, 도 2는 도 1에서 제 2 제어 구동부에 대한 상세 구성도로서, 여기서 참조번호 24-1은 D 플립플롭, 24-2는 인버터, 24-3, 24-4는 트랜지스터, 24-5는 논리곱소자, 24-6은 트랜지스터, 24-7은 인버터, 24-8은 부정논리합소자, 24-9, 24-10, 24-11은 트랜지스터, 24-12는 비교기, 24-13은 인버터, 24-14는 EX_NOR 소자, 24-15는 비교기, 24-16은 RS 플립플롭이다.2 is a detailed configuration diagram of the second control driver in FIG. 1, wherein reference numeral 24-1 is a D flip-flop, 24-2 is an inverter, 24-3, 24-4 is a transistor, and 24-5 is a Logical AND, 24-6 is a transistor, 24-7 is an inverter, 24-8 is a negative logic device, 24-9, 24-10, 24-11 is a transistor, 24-12 is a comparator, 24-13 is an inverter, 24-14 is an EX_NOR device, 24-15 is a comparator, and 24-16 is an RS flip-flop.
상기와 같은 구성으로 이루어진 US 5,905,369호의 스위칭 컨버터의 기본적인 스위칭 제어 방식을 보게 되면, 제 1 스위칭 컨버터(10)인 Boost 1은, L1 전류가 영(zero)이 되면 스위치를 켜고, SW1 전류를 검출한 전압 Vi1이 전압 검출부(4; Voltage detector)의 제어전압 Ve와 같아지면 스위치를 오프하는 제어 신호 Vdr1를 생성한다. 또한, 제 2 스위칭 컨버터(20)인 Boost 2는, 상기 Boost 1의 SW1 제어 신호 Vdr1을 입력받아 충방전 회로를 이용하여 Vdr1과 180도 위상차를 가지면서 SW2를 켜고 Ve와 Vi2가 같아지면 SW2를 끄는 제어신호 Vdr2를 생성한다.Referring to the basic switching control method of the switching converter of US 5,905,369 having the above configuration, Boost 1, the first switching converter 10, switches on when the L1 current becomes zero and detects the SW1 current. When the voltage Vi1 is equal to the control voltage Ve of the voltage detector 4, the control signal Vdr1 is turned off. In addition, Boost 2, which is the second switching converter 20, receives SW1 control signal Vdr1 of Boost 1 and turns on SW2 by 180 degrees out of phase with Vdr1 using a charge / discharge circuit. Generate a control signal Vdr2 to turn off.
도3은 도1 및 도2에서 각 부분의 동작을 보인 파형도이고, 도4는 종래 인터리브 방식 스위칭 컨버터의 문제점을 보인 동작 파형도이다. 도3 및 도4에서 (1)Vdr1은 제 1 제어 구동부(14)에서 출력되는 제 1 구동신호의 파형, (2)Vc1과 (3)Vc2는 충방전 모습을 나타내는 파형, (4)Vdr2는 제 2 제어 구동부(24)에서 출력되는 제 2 구동신호의 파형, (5)Ii1은 제 1 제어 구동부(14)에 입력되는 입력전류의 파형, (6)Ii2는 제 2 제어 구동부(24)에 입력되는 입력전류의 파형이다.Figure 3 is a waveform diagram showing the operation of each part in Figures 1 and 2, Figure 4 is an operational waveform diagram showing a problem of the conventional interleaved switching converter. 3 and 4, (1) Vdr1 is a waveform of the first drive signal output from the first control driver 14, (2) Vc1 and (3) Vc2 are waveforms showing charge and discharge, and (4) Vdr2 is The waveform of the second drive signal output from the second control driver 24, (5) Ii1 is the waveform of the input current input to the first control driver 14, (6) Ii2 is the waveform of the second control driver 24 This is the waveform of the input current.
상기 도1, 도2와 함께 도3 및 도4를 참조하여 US 5,905,369호 스위칭 컨버터의 제어 방식에 대해 좀더 자세히 설명하면, 제 2 제어 구동부(24)인 Control driver #2는 Vdr1 신호를 입력받아 전류원 I1, I2, I3, I4를 이용하여 C1, C2를 충방전하여 Vc1과 Vc2가 같아지는 지점에서 SW2를 켜는 Vdr2 신호를 생성한다. 따라서, 전술한 US 5,905,369호에 의하면 두 개의 컨버터가 병렬 연결된 경우에 있어서, I1, I2, I3, I4가 같고 C1, C2가 같을 경우 Vdr1과 Vdr2는 도3에 도시된 파형도에서와 같이 180도 위상차를 유지할 수 있게 된다.Referring to the control method of the US 5,905,369 switching converter with reference to FIGS. 3 and 4 together with FIGS. 1 and 2, the control driver # 2, the second control driver 24, receives a Vdr1 signal and receives a current source. C1 and C2 are charged and discharged using I1, I2, I3, and I4 to generate a Vdr2 signal that turns on SW2 at the point where Vc1 and Vc2 are equal. Therefore, according to the aforementioned US 5,905,369, when two converters are connected in parallel, when I1, I2, I3, and I4 are the same and C1 and C2 are the same, Vdr1 and Vdr2 are 180 degrees as shown in the waveform diagram shown in FIG. The phase difference can be maintained.
그러나, 상기와 같은 선등록 특허에 따를 경우, Boost2는 Boost1과 달리 L2의 전류가 영(zero)이 되는 지점을 검출하지 않는바, 만일 L2의 소자값이 L1보다 클 경우 전류의 기울기가 낮아져서 L2의 전류가 영이 되기 전에 SW2가 ON 되어 도4의 파형도에서와 같이 CCM으로 동작하게 되며, 이와 같이 CCM으로 동작할 경우 D2의 역회복 현상에 의해 SW2의 스위칭 손실과 스위칭 노이즈가 증가하는 문제점이 있게 된다.However, in accordance with the pre-registered patent as described above, Boost2 does not detect the point where the current of L2 becomes zero unlike Boost1. If the device value of L2 is larger than L1, the slope of the current is lowered so that L2 SW2 is turned on before the current becomes zero, and it operates as CCM as shown in the waveform diagram of FIG. 4. In the case of operating as CCM, switching loss and switching noise of SW2 increases due to the reverse recovery of D2. Will be.
따라서, boost2도 L2 전류가 영이 되는 지점을 검출하는 회로가 필요하다고 할 수 있는데, 이때 L1과 L2가 같이 않은 조건에서 boost2도 L2 전류가 영이 되는 지점에서 SW2를 켜게 되면 boost1과 boost2의 주기가 맞지 않아 입력전류와 출력전류의 리플이 상승하게 되는 문제가 발생하게 되는바, Vdr1과 Vdr2의 주기와 위상을 같아지게 제어하는 회로가 필요하다.Therefore, boost2 also needs a circuit that detects the point where the L2 current becomes zero. If SW2 is turned on at the point where L2 current becomes zero even under conditions where L1 and L2 are not equal, the period of boost1 and boost2 will not match. Therefore, a problem arises in that the ripple of the input current and the output current increases, and a circuit for controlling the period and phase of Vdr1 and Vdr2 is required to be the same.
이에 본 발명은 상기와 같은 종래의 제반 문제점을 해결하기 위해 제안된 것으로서, 구체적으로 본 발명은 전하 공유 방법을 사용하여 병렬 연결된 컨버터의 위상차를 일정하게 유지하면서 병렬 연결된 컨버터 모두 DCM(discontinuous conduction mode)으로 동작하게 제어함으로써 CCM(continuous conduction mode)으로 동작할 경우에 발생하는 역회복 현상에 의해 스위칭 손실과 스위칭 노이즈가 증가하는 문제점을 해결할 수 있는 병렬연결 스위칭 컨버터를 제공하는 것을 그 주요한 기술적 과제로 한다.Accordingly, the present invention has been proposed to solve the above-mentioned general problems. Specifically, the present invention is a charge sharing method using both of the parallel connected converters while maintaining a constant phase difference of the parallel connected converter DCM (discontinuous conduction mode) The main technical problem is to provide a parallel-connected switching converter that can solve the problem that switching loss and switching noise increase due to reverse recovery phenomenon when operating in continuous conduction mode (CCM) by controlling to operate at the same time. .
상기와 같은 기술적 과제를 달성하기 위하여 본 발명에서 제공하는 병렬 연결 스위칭 컨버터는, 전원 입력단에 제1컨버터와 제2컨버터를 포함하는 복수개의 컨버터가 병렬로 연결되고, 상기 복수개의 컨버터의 턴온 및 턴오프를 조절하여 컨버터들 간의 위상차를 일정하게 유지하는 병렬 연결 스위칭 컨버터에 있어서, 상기 제1컨버터의 턴온 및 턴오프를 제어하기 위한 제1스위칭제어신호를 생성하는 제1스위칭제어부와; 제1커패시터와 제2커패시터를 구비하고, 상기 제1스위칭제어부의 제1스위칭제어신호에 따라 상기 제1커패시터와 제2커패시터를 도통시켜 상기 1커패시터와 제2커패시터 간의 전하공유(charge sharing)를 통해 상기 제1커패시터의 전압을 조절하고, 상기 제1커패시터의 전압을 이용하여 상기 제1스위칭제어신호와 동기를 이루면서 일정한 위상차를 갖는 동기신호를 생성하는 동기신호발생부; 및, 상기 동기신호발생부에서 생성된 동기신호에 따라 제2스위칭제어신호를 생성하여 상기 제2컨버터의 턴온 및 턴오프를 제어하는 제2스위칭제어부;를 포함하여 구성된 것을 그 기본적인 구성상의 특징으로 한다.In order to achieve the above technical problem, in the parallel-connected switching converter provided in the present invention, a plurality of converters including a first converter and a second converter are connected in parallel to a power input terminal, and turn on and turn on the plurality of converters. 1. A parallel-connected switching converter that adjusts off to maintain a constant phase difference between converters, comprising: a first switching controller for generating a first switching control signal for controlling the turn-on and turn-off of the first converter; A first capacitor and a second capacitor, and conducting charge sharing between the first capacitor and the second capacitor by conducting the first capacitor and the second capacitor according to the first switching control signal of the first switching controller. A synchronization signal generator for adjusting a voltage of the first capacitor and generating a synchronization signal having a predetermined phase difference in synchronization with the first switching control signal using the voltage of the first capacitor; And a second switching controller for generating a second switching control signal according to the synchronization signal generated by the synchronization signal generator and controlling the turn-on and turn-off of the second converter. do.
상기와 같은 본 발명의 병렬 연결 스위칭 컨버터에 있어서, 상기 동기신호발생부는 더욱 구체적으로, 제1커패시터와; 상기 제1커패시터와 스위치 소자를 통해 연결되어 상기 스위치 소자의 닫힘(ON)에 의해 상기 제1커패시터와 전하 공유를 할 수 있도록 연결된 제2커패시터와; 상기 제1커패시터의 전압과 기준전압을 비교하여 동기신호를 생성하는 비교부; 및, 상기 제1커패시터와 연결되어 상기 동기신호 및 상기 제1스위칭제어신호에 의해 상기 제1커패시터를 충전 또는 방전시키는 충방전부;를 포함하여 구성될 수 있으며, 더욱 바람직하게는, 상기 동기신호발생부는 상기 제1스위칭제어신호를 입력받아 상기 제1스위칭제어신호에 비해 하이(HIGH) 기간이 짧은 단펄스로 변환하는 단펄스발생부를 더욱 포함하여 구성될 수 있다.In the parallel-connected switching converter of the present invention as described above, the synchronization signal generator is more specifically, a first capacitor; A second capacitor connected to the first capacitor through a switch element and connected to enable charge sharing with the first capacitor by being closed (ON) of the switch element; A comparator configured to generate a synchronization signal by comparing the voltage of the first capacitor with a reference voltage; And a charge / discharge unit connected to the first capacitor to charge or discharge the first capacitor by the synchronization signal and the first switching control signal. More preferably, the synchronization signal is generated. The unit may further include a short pulse generator for receiving the first switching control signal and converting the short switching pulse into a short pulse having a shorter HIGH period than the first switching control signal.
또한, 상기와 같은 본 발명에 있어서 상기와 같이 생성된 동기신호와 제2스위칭제어신호의 주기를 일치시키기 위한 기술적 구성이 더욱 포함될 수 있는데, 이를 위해 상기 제2스위칭제어부는, 상기 제2스위칭제어신호를 피드백 입력받고 상기 동기신호발생부에서 생성된 동기신호와 상기 제2스위칭제어신호의 위상을 비교하여 위상 차이에 따른 위상에러를 출력하는 위상비교부와; 상기 위상비교부에서 출력된 위상에러에 따라 기준제어전압을 조절하여 구동제어전압을 생성하는 제어전압조절부와; 상기 제어전압조절부에서 생성된 구동제어전압을 이용하여 제2스위칭제어신호를 발생하는 스위칭제어신호발생부;를 더욱 포함하여 구성될 수 있다. 이때, 상기와 같은 제어전압조절부에서 사용되는 기준제어전압은 출력단에 설치된 전압검출부에서 검출된 전압을 기초로, 검출된 전압이 기준전압과 같아지도록 조절한 전압을 사용하는 것이 바람직하다. In addition, the present invention as described above may further include a technical configuration for matching the period of the synchronization signal and the second switching control signal generated as described above, for this purpose, the second switching control unit, the second switching control A phase comparator for receiving a feedback signal and comparing a phase of the sync signal generated by the sync signal generator with a phase of the second switching control signal to output a phase error according to a phase difference; A control voltage adjusting unit which generates a driving control voltage by adjusting a reference control voltage according to the phase error output from the phase comparing unit; And a switching control signal generator configured to generate a second switching control signal by using the driving control voltage generated by the control voltage regulator. At this time, the reference control voltage used in the control voltage adjusting unit as described above, it is preferable to use a voltage adjusted so that the detected voltage is equal to the reference voltage based on the voltage detected by the voltage detector installed in the output terminal.
또한, 상기와 같은 스위칭제어신호발생부에서 제2스위칭제어신호를 생성함에 있어서는, 예컨대 컨버터에 흐르는 전류를 검출하여 컨버터 스위칭 시점을 제어하는 전류 모드 PWM(Pulse Width Modulation) 방식을 사용하거나 기준 전압원과 구동제어전압을 비교하여 스위칭 시점을 제어하는 전압 모드 PWM 방식을 사용할 수 있다. 즉, 본 발명에 있어서 상기 스위칭제어신호발생부는 상기한 제어전압발생부에서 생성된 구동제어전압과 제2컨버터에 포함된 컨버터 스위치를 통과하는 전류에 의해 스위칭 센싱 저항에 인가되는 전압을 비교하여 제2스위칭제어신호를 생성하는 전류 모드 PWM 방식을 적용하거나, 또는 램프 파형을 발생하는 램프발생기를 더욱 구비하고 상기 제어전압발생부에서 생성된 구동제어전압과 램프 전압을 비교하여 제2스위칭제어신호를 생성하는 전압 모드 PWM 방식을 적용할 수도 있다.In addition, in generating the second switching control signal in the switching control signal generator as described above, for example, a current mode pulse width modulation (PWM) method for controlling the converter switching time by detecting a current flowing in the converter or using a reference voltage source A voltage mode PWM method for controlling the switching time point by comparing the driving control voltage may be used. That is, in the present invention, the switching control signal generation unit compares the driving control voltage generated by the control voltage generation unit with the voltage applied to the switching sensing resistor by the current passing through the converter switch included in the second converter. 2 applies a current mode PWM method for generating a switching control signal, or further includes a ramp generator for generating a ramp waveform, and compares the driving control voltage generated by the control voltage generator with the ramp voltage to generate a second switching control signal. It is also possible to apply a voltage mode PWM method to generate.
나아가, 상기 복수개의 컨버터들을 턴온시키는 시점과 관련하여, 본 발명에 대한 더욱 바람직한 구성에 따르면, 상기 제1컨버터에 유입되는 전류가 영(0)이 되는지를 검출하는 제1영전류검출부와, 상기 제2컨버터에 유입되는 전류가 영(0)이 되는지를 검출하는 제2영전류검출부를 더욱 구비하고, 상기 제1, 제2스위칭제어부는 각각 상기 제1, 제2영전류검출부에서 유입 전류가 영으로 검출되면 턴온 제어신호를 생성하도록 구성할 수 있으며, 이와 같이 구성된 본 발명에 따르면 제1컨버터 뿐 아니라 제2컨버터 및 그 이상으로 병렬 연결된 제3컨버터 이하 모든 컨버터들이 임계 전류 모드로 동작하는 것이 가능하게 된다.Further, in relation to a time point for turning on the plurality of converters, according to a more preferred configuration of the present invention, the first zero current detector for detecting whether the current flowing into the first converter is zero; And a second zero current detector for detecting whether the current flowing into the second converter becomes zero, wherein the first and second switching controllers respectively supply inflow currents from the first and second zero current detectors. When it is detected as zero, it can be configured to generate a turn-on control signal. According to the present invention configured as described above, it is preferable that not only the first converter but also all the converters below the second converter and the third converter connected in parallel operate in the threshold current mode. It becomes possible.
상기와 같이 구성된 본 발명의 병렬 연결 스위칭 컨버터에 따르면, 전하 공유 방법을 사용하여 병렬 연결된 컨버터의 위상차를 일정하게 유지하면서 병렬 연결된 컨버터 모두 DCM(discontinuous conduction mode)으로 동작하게 제어함으로써 CCM(continuous conduction mode)으로 동작할 경우에 발생하는 역회복 현상에 의해 스위칭 손실과 스위칭 노이즈가 증가하는 문제점을 해결할 수 있게 된다.According to the parallel-connected switching converter of the present invention configured as described above, by using a charge sharing method, the parallel-connected converters are controlled to operate in a discontinuous conduction mode (CCM) while maintaining the phase difference of the parallel-connected converters constant. It is possible to solve the problem that the switching loss and the switching noise are increased by the reverse recovery phenomenon that occurs when operating in the).
도1은 종래 인터리브 방식 스위칭 컨버터의 블록구성도이다.1 is a block diagram of a conventional interleaved switching converter.
도2는 도1에서 제 2 제어 구동부의 상세구성도이다.FIG. 2 is a detailed configuration diagram of the second control driver in FIG. 1.
도3은 도1 및 도2에서 각 부분의 동작을 보인 파형도이다.3 is a waveform diagram showing the operation of each part in FIGS.
도4는 종래 스위칭 컨버터의 문제점을 보인 동작 파형도이다.4 is an operation waveform diagram showing a problem of a conventional switching converter.
도5는 본 발명의 일 실시예에 의한 스위칭 컨버터의 블록 구성도이다.5 is a block diagram of a switching converter according to an embodiment of the present invention.
도6은 도5에서의 제1스위칭제어부에 대한 상세 구성도이다.FIG. 6 is a detailed configuration diagram of the first switching control unit in FIG. 5.
도7은 도5에서의 전압 검출부에 대한 상세 구성도이다.FIG. 7 is a detailed configuration diagram of the voltage detector in FIG. 5.
도8은 도5에서의 제2스위칭제어부에 대한 상세 구성도이다.FIG. 8 is a detailed configuration diagram of the second switching control unit in FIG. 5.
도9는 도8에서의 동기신호발생부에 대한 상세 구성도이다.FIG. 9 is a detailed configuration diagram of a synchronization signal generator in FIG. 8.
도10은 본 발명의 제2스위칭제어부에서 각 신호들의 파형에 대한 타이밍 차트이다. 10 is a timing chart of the waveform of each signal in the second switching controller of the present invention.
도11의(a)는 도8에서의 동기신호발생부에 대한 또 다른 구성예를 보인 상세 구성도이고, 도11의(b)는 도11의(a)와 같이 구현된 동기신호발생부를 포함한 제2스위칭제어부에서 각 신호들의 파형에 대한 타이밍 차트이다. FIG. 11A is a detailed configuration diagram showing another configuration example of the synchronization signal generator in FIG. 8, and FIG. 11B includes a synchronization signal generator implemented as in FIG. A timing chart of the waveforms of the signals in the second switching controller.
도12는 본 발명의 다른 실시예에 의한 스위칭 컨버터의 블록 구성도이다.12 is a block diagram of a switching converter according to another embodiment of the present invention.
도13은 도12에서의 제1스위칭제어부에 대한 상세 구성도이다.FIG. 13 is a detailed configuration diagram of the first switching control unit in FIG. 12.
도14는 도12에서의 제2스위칭제어부에 대한 상세 구성도이다.FIG. 14 is a detailed configuration diagram of the second switching control unit in FIG. 12.
도15는 본 발명의 또 다른 실시예에 의한 두 개 이상의 컨버터가 병렬로 연결된 경우의 병렬 연결 스위칭 컨버터를 보인 블록구성도이다.15 is a block diagram illustrating a parallel-connected switching converter when two or more converters are connected in parallel according to another embodiment of the present invention.
도16은 도15에 도시된 실시예에서 각 신호들의 파형에 대한 타이밍차트이다.FIG. 16 is a timing chart for the waveform of each signal in the embodiment shown in FIG.
도17은 본 발명의 또 다른 실시예에 의한 두 개 이상의 컨버터가 병렬로 연결된 경우의 병렬 연결 스위칭 컨버터를 보인 블록 구성도이다.17 is a block diagram illustrating a parallel-connected switching converter when two or more converters are connected in parallel according to another embodiment of the present invention.
도18은 도17에 도시된 실시예에서 각 신호들의 파형에 대한 타이밍차트이다.FIG. 18 is a timing chart for the waveform of each signal in the embodiment shown in FIG.
도19는 본 발명의 또 다른 실시예에 의한 컨버터 입력이 교류 전원이고 전류 모드 PWM을 사용한 병렬 연결 스위칭 컨버터를 보인 블록구성도이다.19 is a block diagram showing a parallel-connected switching converter using an AC power supply and a current mode PWM according to another embodiment of the present invention.
도20은 본 발명의 또 다른 실시예에 의한 컨버터 입력이 교류 전원이고 전압 모드 PWM을 사용한 병렬 연결 스위칭 컨버터를 보인 블록구성도이다.FIG. 20 is a block diagram illustrating a parallel-connected switching converter using an AC power supply and a voltage mode PWM according to another embodiment of the present invention.
* 도면의 주요 부분에 대한 부호의 설명 *Explanation of symbols on the main parts of the drawings
100 : 제1컨버터(Boost1) 101 : 전원 입력부100: first converter (Boost1) 101: power input unit
102 : 출력 커패시터(Co) 103 : 부하(Load)102: output capacitor (Co) 103: load
104 : 전압 검출부 110 : 제1영전류검출부104: voltage detector 110: first zero current detector
140 : 제1스위칭제어부 200 : 제 2 컨버터140: first switching control unit 200: second converter
210 : 제2영전류검출부 240 : 제2스위칭제어부210: second zero current detection unit 240: second switching control unit
244 : 동기신호발생부 245 : 위상비교부244: synchronization signal generator 245: phase comparison unit
246 : 제어전압조절부 247 : 스위칭제어신호발생부246: control voltage regulator 247: switching control signal generator
이하, 첨부한 도면을 참조하여 본 발명을 더욱 상세하게 설명한다. 한편, 본 명세서에 있어서, 각 도면에 도시된 여러 실시예들에서 동일한 기능을 하는 동일한 구성 요소의 경우, 이해의 혼동을 방지하기 위해 별도의 참조부호나 명칭을 사용하지 않고 동일한 명칭을 사용하여 동일한 참조 부호로 표시하였다.Hereinafter, the present invention will be described in more detail with reference to the accompanying drawings. On the other hand, in the present specification, in the case of the same component having the same function in the various embodiments shown in each drawing, the same name using the same name without using separate reference numerals or names to avoid confusion of understanding. Reference numerals are indicated.
[실시예 1]Example 1
도5 내지 도13은 본 발명의 병렬 연결 스위칭 컨버터에 대한 제1실시예를 도시한 도면으로서, 상기 제1실시예는 컨버터 입력 전압으로서 직류 전원을 공급받고, 전류모드 PWM을 사용하며, 2개의 컨버터를 사용한 예를 보여주고 있다. 5 to 13 show a first embodiment of the parallel-connected switching converter of the present invention, which is supplied with DC power as the converter input voltage, uses current mode PWM, An example of using a converter is shown.
도5는 본 발명의 일 실시예에 의한 전류 모드 PWM을 사용한 병렬연결 스위칭 컨버터의 블록 구성도로서, 도5에 도시된 바와 같이, 본 발명에서 제공하는 병렬 연결 스위칭 컨버터는 기본적으로 전원 입력단(101)에 제1컨버터(100)와 제2컨버터(200)를 포함하는 복수개의 컨버터들이 병렬로 연결된 구성을 가지고 있음을 알 수 있다. FIG. 5 is a block diagram of a parallel-connected switching converter using a current mode PWM according to an embodiment of the present invention. As shown in FIG. 5, the parallel-connected switching converter provided by the present invention is basically a power input 101. It can be seen that a plurality of converters including the first converter 100 and the second converter 200 are connected in parallel.
여기서, 상기 컨버터는 일반적으로 입력 전압에 비해 큰 출력 전압을 얻을 수 있도록 부스트(Boost) 회로를 구비한 스위칭 모드 파워 서플라이(SMPS)의 일종인 부스트 컨버터를 예시하고 있으며, 이러한 부스트 컨버터 이외에도 본 발명은 다른 종류의 컨버터들에 대해서도 특별한 제한 없이 적용될 수 있다. Herein, the converter exemplifies a boost converter which is a kind of a switching mode power supply (SMPS) having a boost circuit so as to obtain a large output voltage compared to an input voltage. Other types of converters can be applied without particular limitations.
도5에 도시된 실시예에 따르면, 상기 제1컨버터(100)와 제2컨버터(200)는 기본적으로 인덕터(L1, L2), 다이오드(D1, D2), 스위치(120, 220), 필터 커패시터(C0; 102)를 구비하고 있으며, 여기에 상기 스위치(120, 220)의 개폐를 제어하기 위한 제1스위칭제어부(140)와 제2스위칭제어부(240)가 포함되어 구성된다. 이때, 상기 스위치(120, 220) 소자로서는 일반적으로 MOSFET가 가장 바람직하게 사용될 수 있으며, 이외에도 경우에 따라 BJT나 IGBT 등 다른 종류의 트랜지스터 소자를 사용하여도 무방하다.According to the embodiment shown in FIG. 5, the first converter 100 and the second converter 200 are basically inductors L1 and L2, diodes D1 and D2, switches 120 and 220, and filter capacitors. (C0) 102, which includes a first switching controller 140 and a second switching controller 240 for controlling the opening and closing of the switches 120 and 220. In this case, MOSFETs may be most preferably used as the switch 120 and 220 elements. In addition, other types of transistor elements such as BJT and IGBT may be used in some cases.
한편, 본 발명은 상기와 같이 병렬 연결된 다수개의 컨버터들이 서로 일정한 위상차를 가지고 스위칭 동작할 수 있도록 함으로써 입력 전류의 리플과 출력 전류의 리플을 줄일 수 있게 하는 기술에 관한 것으로서, 본 발명은 이와 같이 병렬 연결된 컨버터들 간의 위상차를 일정하게 유지함에 있어 전하 공유(charge sharing)의 방법을 응용하여 스위칭 주파수가 가변되는 경우에도 일정한 위상 차이를 갖는 제어신호를 생성할 수 있도록 한 것에 가장 주요한 기술적 특징이 있다. On the other hand, the present invention relates to a technique for reducing the ripple of the input current and the output current by allowing a plurality of converters connected in parallel as described above to have a switching operation with a constant phase difference, the present invention as described above In maintaining the phase difference between the connected converters, the most important technical feature is to apply a method of charge sharing to generate a control signal having a constant phase difference even when the switching frequency is varied.
이를 위해 본 발명에서는 전하 공유에 의해 제1컨버터의 제1스위칭제어신호(Vdr1)와 동기를 이루면서 일정한 위상 차이를 갖는 동기신호를 생성하는 동기신호발생부(244)를 구비하고 있는데, 상기 동기신호발생부(244)는 기본적으로 제1커패시터(C1)와 제2커패시터(C2)를 구비하고 상기 제1스위칭제어부(140)의 제1스위칭신호(Vdr1)에 따라 상기 제1커패시터(C1)와 제2커패시터(C2)를 도통시켜 상기 1커패시터와 제2커패시터 간의 전하공유(charge sharing)를 통해 상기 제1커패시터의 전압을 조절하고, 상기와 같이 조절된 제1커패시터(C1)의 전압(Vc1)에 기초하여 상기 제1스위칭제어신호(Vdr1)와 동기를 이루면서 일정한 위상차를 갖는 동기신호를 생성하도록 구성되어 있다. To this end, the present invention includes a synchronization signal generator 244 which generates a synchronization signal having a predetermined phase difference while synchronizing with the first switching control signal Vdr1 of the first converter by charge sharing. The generating unit 244 basically includes a first capacitor C1 and a second capacitor C2, and the first capacitor C1 and the first capacitor C1 according to the first switching signal Vdr1 of the first switching controller 140. The second capacitor C2 conducts to regulate the voltage of the first capacitor through charge sharing between the first capacitor and the second capacitor, and the voltage Vc1 of the adjusted first capacitor C1 is adjusted as described above. Is synchronized with the first switching control signal Vdr1 to generate a synchronization signal having a constant phase difference.
도9는 상기와 같은 본 발명의 동기신호발생부에 대한 바람직한 일 구성예를 도시한 도면으로서, 도9에 도시된 것과 같이 본 발명에 있어서 상기 동기신호발생부(244)는 병렬 연결된 제1커패시터(C1)와 제2커패시터(C2), 상기 제1커패시터(C1)와 연결되어 상기 제1커패시터(C1) 및/또는 제2커패시터(C2)를 충/방전시키는 충방전부(244-10), 그리고 상기 제1커패시터(C1)의 전압을 기준전압과 비교하여 동기신호(Sync)를 생성하는 비교부(244-1)를 포함하여 구성될 수 있다.9 is a view showing a preferred configuration of the synchronization signal generator of the present invention as described above, in the present invention, as shown in FIG. 9, the synchronization signal generator 244 is a first capacitor connected in parallel Charge and discharge unit (244-10) connected to (C1) and the second capacitor (C2), the first capacitor (C1) to charge / discharge the first capacitor (C1) and / or the second capacitor (C2), And a comparator 244-1 generating the synchronization signal Sync by comparing the voltage of the first capacitor C1 with a reference voltage.
여기서, 상기 제1커패시터(C1)와 제2커패시터(C2)는 도시된 바와 같이 스위치 소자(244-6)를 통해 연결됨으로써 외부 제어 신호에 의해 선택적으로 도통될 수 있도록 구성되는데, 이때, 상기 제1커패시터(C1)와 제2커패시터(C2)를 연결하는 스위치 소자(244-6)는 상기 제1제어구동부(110)의 제1스위칭제어신호(Vdr1)에 의해 온/오프가 제어됨으로써 일정 시간 동안 전하를 공유하여 상기 제1커패시터(C1)의 전압을 조절할 수 있도록 되어 있다. Here, the first capacitor (C1) and the second capacitor (C2) is configured to be selectively connected by an external control signal by being connected through the switch element 244-6 as shown, wherein the first The switch element 244-6 connecting the first capacitor C1 and the second capacitor C2 is turned on / off by the first switching control signal Vdr1 of the first control driver 110 for a predetermined time. It is possible to control the voltage of the first capacitor C1 by sharing charge.
또한, 본 발명에 대한 더욱 바람직한 구성에 따르면, 상기 스위치 소자(244-6)의 온/오프 동작을 제어하기 위한 제어 신호로는 상기한 바와 같이 제1제어구동부(110)의 제1스위칭제어신호(Vdr1)를 그대로 사용할 수도 있지만, 도9에 도시된 예에서 보는 것과 같이 단펄스발생부(244-2)를 통해 상기 제1스위칭제어신호(Vdr1)의 듀티비(duty ratio)를 조절하여 하이(HIGH) 기간이 짧은 단펄스(SP)로 변환해서 사용하게 되면 상기 제2커패시터(C2)의 충전 시간 및 전하량을 임의로 조절하는 것이 가능하므로 위상차 제어의 측면에서 더욱 바람직하다.Further, according to a more preferred configuration of the present invention, as a control signal for controlling the on / off operation of the switch element 244-6 as described above, the first switching control signal of the first control driver 110 as described above. Although Vdr1 may be used as it is, as shown in the example shown in FIG. 9, the short pulse generator 244-2 adjusts the duty ratio of the first switching control signal Vdr1 to high. It is more preferable in terms of phase difference control because it is possible to arbitrarily control the charging time and the amount of charge of the second capacitor C2 when converting to a short pulse SP having a short duration of HIGH.
또한, 상기 제1실시예에 따르면 상기 동기신호발생부(244)에서 동기신호(Sync)를 생성함에 있어서는 상기와 같은 제1커패시터(C1)에 걸리는 전압(Vc1)과 소정의 기준전압(Vref)을 비교기(244-1)를 통해 비교하여 양 전압이 같아졌을 때 신호 레벨이 HIGH로 되는 동기신호(Sync)를 생성하여 제2스위칭제어신호(Vdr2)의 조절을 위한 신호로서 출력하게 되며, 아울러 상기와 같이 생성된 동기신호(Sync)는 후술하는 바와 같이 충방전부(244-10)에 입력되어 커패시터(C1, C2)의 충/방전 동작을 제어하는 신호로서 이용될 수 있다.In addition, according to the first embodiment, in generating the synchronization signal Sync by the synchronization signal generator 244, the voltage Vc1 applied to the first capacitor C1 and the predetermined reference voltage Vref are applied. Is compared through the comparator 244-1 to generate a synchronization signal Sync when the signal level becomes HIGH when the two voltages are the same, and outputs it as a signal for adjusting the second switching control signal Vdr2. The sync signal Sync generated as described above may be input to the charge / discharge unit 244-10 to be used as a signal for controlling charge / discharge operations of the capacitors C1 and C2.
상기 충방전부(244-10)는 도9에서 보는 것과 같이 본 발명의 동기신호발생부(244)에 있어서 상기 제1커패시터(C1)의 일단에 연결되어 상기 제1커패시터(C1) 및/또는 제2커패시터(C2)에 전하를 충전 및 방전시킬 수 있도록 하는 구성요소이다. 여기서, 상기 충방전부(244-10)는 예컨대 도시된 실시예에서와 같이 충전용 전류전원(I1)과 방전용 전류전원(I2) 및, 상기 충, 방전용 전류전원(I1, I2)의 동작을 제어하기 위한 RS 플립플롭(244-3)을 포함하여 구성될 수 있으며, 이러한 충방전부(244-10)는 상기 비교기(244-1)에서 생성된 동기신호(Sync) 및 단펄스로 변환된 제1스위칭제어신호(SP)에 의해 그 충전/방전 동작을 제어할 수 있도록 구성되어 있다.9, the charge / discharge unit 244-10 is connected to one end of the first capacitor C1 in the synchronization signal generator 244 of the present invention as shown in FIG. It is a component to charge and discharge electric charges to the two capacitors (C2). Here, the charging and discharging unit 244-10 may operate, for example, as the charging current power source I1 and the discharging current power source I2 and the charging and discharging current power sources I1 and I2 as in the illustrated embodiment. And an RS flip-flop 244-3 for controlling the charge / discharge unit 244-10. The charge / discharge unit 244-10 is converted into a sync signal and a short pulse generated by the comparator 244-1. The charging / discharging operation can be controlled by the first switching control signal SP.
한편, 도시된 실시예에 있어, 상기 동기신호발생부(244)는 회로의 간략화를 위해 별도의 회로를 따로 구성하지 않고 제2스위칭제어부(240)의 회로 안에 포함된 형태로 예시되어 있으나, 제2스위칭제어부(240)와 별개로 분리된 회로로서 구현하는 것도 물론 가능하다. Meanwhile, in the illustrated embodiment, the synchronization signal generator 244 is illustrated in a form included in a circuit of the second switching controller 240 without separately configuring a circuit for simplifying the circuit. It is also possible to implement as a circuit separate from the two switching control unit 240 separately.
이하, 상기와 같이 구성된 본 발명의 병렬 연결 스위칭 컨버터의 동작 과정을 더욱 상세하게 설명한다.Hereinafter, an operation process of the parallel-connected switching converter of the present invention configured as described above will be described in more detail.
도5를 참조하여 보면, 본 발명에서는 제2컨버터(200)의 턴온 및 턴오프를 제어함에 있어 제2컨버터(200)에도 제2인덕터(L2)에 흐르는 전류(Ii2)가 영이 되는 시점을 검출하기 위한 제2영전류검출부(210)을 구비하고 유입 전류가 영이 된 이후에 턴온되도록 함으로써 제2컨버터(200)도 제1컨버터(100)과 마찬가지로 DCM 동작을 하도록 제어하며, 이러한 점에 본 발명의 일차적인 특징이 존재한다. Referring to FIG. 5, in the present invention, when controlling the turn-on and turn-off of the second converter 200, the second converter 200 also detects a time when the current Ii2 flowing in the second inductor L2 becomes zero. And a second zero current detector 210 for turning on and then turning on after the inflow current becomes zero, so that the second converter 200 performs DCM operation similarly to the first converter 100. There is a primary characteristic of.
이에 대하여 더욱 상세하게 설명하면, 먼저 제1컨버터(100)에 포함된 제1스위칭제어부(140)에서는 부스트 회로에서 전원 입력단에 일반적으로 구비되어 있는 제1인덕터(L1) 소자에 전자기적으로 커플링된 제1유도코일(L3)을 추가로 설치하여 상기 제1유도코일(L3)에 유도되는 전압(Vzcd1)에 의해 유입 전류를 검출할 수 있는 제1영전류검출부(110)를 구성하고 있다. 따라서 제1스위칭제어부(140)에서는 도6에 도시된 바로부터 알 수 있는 것과 같이 상기 Vzcd1 신호를 입력받아 소정의 기준 전압(Vth)과 비교하고 상기 Vzcd1 신호가 기준 전압(Vth) 이하로 내려가는 순간 유입 전류가 영인 것으로 판별하여 제1스위칭제어신호(이하, Vdr1 신호)를 하이(HIGH)로 만듦으로써 컨버터 스위치를 턴온시킨다.In more detail, first, the first switching controller 140 included in the first converter 100 electromagnetically couples the first inductor L1 element generally provided at the power input terminal of the boost circuit. The first induction coil L3 is further provided to constitute a first zero current detector 110 capable of detecting an inflow current by the voltage Vzcd1 induced in the first induction coil L3. Accordingly, as shown in FIG. 6, the first switching controller 140 receives the Vzcd1 signal, compares it with a predetermined reference voltage Vth, and at the moment when the Vzcd1 signal falls below the reference voltage Vth. The converter switch is turned on by determining that the inflow current is zero and making the first switching control signal (hereinafter, Vdr1 signal) high.
또한, 상기 제1스위칭제어부(140)에서는 컨버터 스위치의 턴온에 의해 컨버터 스위치를 통과하는 전류가 스위칭 센싱 저항(CS1)에 인가하는 전압(Vi1)과 전원 출력단에 설치된 전압검출부(104)에서 출력된 기준제어전압(Ve)을 받아 상기 Vi1 전압이 Ve 전압 이상으로 올라가는 순간 Vdr1 신호를 로우(LOW)로 만듦으로써 컨버터 스위치를 턴오프시킨다. 이때, 상기 전압검출부(104)는 도7에 도시된 바와 같이 출력전압 V0를 저항 분배기(R1, R2)를 사용하여 검출하고, 검출된 전압이 기준 전압(Vo_Ref)이 되도록 기준제어전압(Ve)을 조절하여 제1스위칭제어부(140) 및 제2스위칭제어부(240)에 피드백 공급한다.In addition, the first switching controller 140 outputs the voltage Vi1 applied to the switching sensing resistor CS1 by the turn-on of the converter switch and the voltage detector 104 installed at the power output terminal. The converter switch is turned off by making the Vdr1 signal low when the Vi1 voltage rises above the Ve voltage in response to the reference control voltage Ve. At this time, the voltage detector 104 detects the output voltage V0 using the resistor dividers R1 and R2 as shown in FIG. 7, and the reference control voltage Ve so that the detected voltage becomes the reference voltage Vo_Ref. The feedback is supplied to the first switching controller 140 and the second switching controller 240 by adjusting.
다음으로, 상기 제2스위칭제어부(240)에서는 도5 및 도8에 도시된 바로부터 알 수 있는 것과 같이, 제2영전류검출부(210)의 제2유도코일(L4)에 유도된 전압에 의한 Vzcd2 신호를 입력받아 Vzcd2 전압이 기준전압 Vth 이하로 내려가는 순간 제2스위칭제어신호(이하 'Vdr2 신호')를 HIGH로 만들고, 컨버터 스위치를 통과하는 전류에 의한 스위치 센싱 전압 Vi2가 구동제어전압 Vc 이상으로 올라가는 순간 Vdr2 신호를 LOW로 만든다.Next, as shown in FIGS. 5 and 8, in the second switching controller 240, the voltage is induced by the second induction coil L4 of the second zero current detector 210. As soon as the Vzcd2 signal is received and the Vzcd2 voltage drops below the reference voltage Vth, the second switching control signal (hereinafter referred to as the 'Vdr2 signal') is made HIGH and the switch sensing voltage Vi2 caused by the current passing through the converter switch is higher than the drive control voltage Vc. As soon as it goes up, the Vdr2 signal goes low.
즉, 상기에서와 같이 본 발명에 따르면, 기존의 병렬 연결 스위칭 컨버터와는 달리 제2컨버터를 턴온시킴에 있어 제1컨버터(100)뿐 아니라 제2컨버터(200)에도 전류가 영이 되는 시점을 검출하고 영 전류 검출시 컨버터 스위치가 턴온 되도록 제어함으로써 제1컨버터와 제2컨버터 모두 DCM으로 동작하도록 하고 있다. 또한, 본 발명에서는 이와 같이 제2컨버터의 전류가 영이 되는 시점에서 스위치를 켬으로써 컨버터들 간의 주기가 맞지 않는 점에 대해서는 상기 제1컨버터(100)의 동작을 제어하는 제1스위칭제어신호(Vdr1)와 일정한 위상차를 갖는 동기신호(Sync)를 생성하는 별도의 동기신호발생부(244)를 별도로 구비하고, 제2스위칭제어부(240)에서는 상기 동기신호(Sync)를 기초로 Vdr2 신호를 조절하여 제2컨버터 스위치(220)의 도통 시간을 조절함으로써 Vdr2 신호가 Vdr1 신호와 동기를 이루도록 제어한다. That is, according to the present invention as described above, when turning on the second converter, unlike the conventional parallel-connected switching converter detects the time when the current becomes zero in the second converter 200 as well as the first converter 100. When the zero current is detected, the converter switch is turned on so that both the first converter and the second converter operate as DCM. Also, in the present invention, when the current of the second converter becomes zero, the first switching control signal Vdr1 controls the operation of the first converter 100 in the case where the periods between the converters are not matched by turning on the switch. ) And a separate synchronization signal generator 244 for generating a synchronization signal (Sync) having a predetermined phase difference, and the second switching controller 240 adjusts the Vdr2 signal based on the synchronization signal (Sync) By controlling the conduction time of the second converter switch 220, the Vdr2 signal is controlled to be synchronized with the Vdr1 signal.
이와 같이 본 발명의 가장 특징적인 구성인 동기신호발생부(244) 및 상기 동기신호발생부(244)에서 생성된 동기신호(Sync)에 의해 제2컨버터를 제어하는 동작 과정에 대하여 도9 및 도10을 참조하여 더욱 상세하게 설명하면 다음과 같다.As described above, an operation process of controlling the second converter by the synchronization signal generator 244 and the synchronization signal generated by the synchronization signal generator 244 is the most characteristic configuration of the present invention. Referring to 10 in more detail as follows.
먼저, 도9를 보게 되면, 상기 동기신호발생부(244)는 Vdr1 신호를 입력받아 단펄스생성부(244-2)를 통해서 하이(HIGH) 기간이 상대적으로 짧은 단펄스 SP로 변환한다. 이와 같이 단펄스로 변환된 Vdr1 신호(즉, SP 신호)는 제1커패시터(C1)와 제2커패시터(C2)를 연결하는 스위치(244-6)를 ON 시킴과 동시에, RS 플립플롭(S44-3)의 리셋단에 연결되어 충전 전류원(I1)과 제1커패시터(C1)를 연결하는 충전 스위치(244-4)를 ON 시키고, 방전 전류원(I2)에 연결된 스위치(244-5)를 OFF 시킨다. First, referring to FIG. 9, the synchronization signal generator 244 receives a Vdr1 signal and converts it into a short pulse SP having a short HIGH period through the short pulse generator 244-2. The Vdr1 signal (that is, the SP signal) converted into the short pulse as described above turns on the switch 244-6 connecting the first capacitor C1 and the second capacitor C2, and at the same time, the RS flip-flop S44-. 3) the charge switch 244-4 connected to the reset terminal of the charging current source I1 and the first capacitor C1 is turned on, and the switch 244-5 connected to the discharge current source I2 is turned off. .
이때, 상기와 같이 SP 신호에 의해 스위치 244-6이 ON 되는 순간, C1과 C2의 소자값이 같을 경우 두 커패시터에 충전된 전하는 전하 공유 현상에 의해 균일하게 나누어지게 된다. 따라서, 상기와 같은 스위치 244-6의 ON에 의해 C1에 걸리는 전압 Vc1과 C2에 걸리는 전압 Vc2는 같아지게 된다. (도10의 파형도에서 'A'점)At this time, when the switch 244-6 is turned on by the SP signal as described above, when the device values of C1 and C2 are the same, the charges charged in the two capacitors are uniformly divided by the charge sharing phenomenon. Accordingly, the voltage Vc1 applied to C1 and the voltage Vc2 applied to C2 become equal by the ON of the switch 244-6 as described above. ('A' point in the waveform diagram of Fig. 10)
또한, 상기와 같이 SP가 HIGH 인 구간에서 충전 스위치 244-4와 커패시터 스위치 244-6이 ON 상태이므로 충전 전류원 I1은 C1과 C2를 동시에 충전하므로 제1커패시터 전압 Vc1과 제2커패시터 전압 Vc2는 일정한 기울기를 가지면서 함께 상승하게 된다. (도10의 파형도에서 'a'구간)In addition, as described above, since the charge switch 244-4 and the capacitor switch 244-6 are turned on in the period where the SP is HIGH, since the charging current source I1 charges C1 and C2 simultaneously, the first capacitor voltage Vc1 and the second capacitor voltage Vc2 are constant. Ascending with the slope. (Section 'a' in the waveform diagram of FIG. 10)
SP가 HIGH에서 LOW로 가면, 커패시터 스위치 244-6은 OFF로 되고 스위치 244-4는 RS 플립플롭 244-3의 출력이 바뀌지 않으므로 계속 ON 상태를 유지하게 된다. 따라서, 이 상태에서 충전 전류원 I1은 제1커패시터 C1만을 계속 충전하게 되어 제1커패시터 C1에 걸리는 전압 Vc1은 계속 상승하게 되며, 제2커패시터 C2는 커패시터 스위치 244-6의 OFF에 의해 충전이 중단되므로 스위치가 끊어진 순간의 최종 전압을 그대로 유지한다. (도10의 파형도에서 'b'구간)When SP goes from HIGH to LOW, capacitor switch 244-6 goes OFF and switch 244-4 remains ON because the output of RS flip-flop 244-3 does not change. Therefore, in this state, the charging current source I1 continuously charges only the first capacitor C1, so that the voltage Vc1 applied to the first capacitor C1 continues to rise, and the second capacitor C2 stops charging by turning off the capacitor switch 244-6. The final voltage at the moment the switch is broken is maintained. (Section 'b' in the waveform diagram of Fig. 10)
상기와 같이 제1커패시터 C1의 충전에 의해 Vc1 전압이 상승하다가 소정의 기준 전압 Vref 전압과 같아지게 되면 비교기 CMP1 신호가 HIGH로 되면서 동기신호(이하 'Sync 신호')를 HIGH로 만들게 된다. 이와 같이 생성된 Sync 신호는 후술하는 위상비교부(245)로 출력되는 것과 동시에, 플립플롭(244-3)의 SET 단자에 입력되며, 이에 따라 충전 스위치 244-4가 꺼지고, 방전 스위치 244-5가 켜지게 된다. 이와 같이 방전 스위치 244-5가 켜지면 방전 전류원 I2는 C1을 방전하게 되고 이에 따라 Vc1 전압은 지속적으로 감소하고 Sync 신호는 다시 LOW로 떨어진다. (도10의 파형도에서 'c'구간)As described above, when the voltage Vc1 increases due to the charging of the first capacitor C1 and becomes equal to the predetermined reference voltage Vref, the comparator CMP1 signal becomes HIGH and the synchronization signal (hereinafter referred to as a 'sync signal') is made high. The Sync signal generated as described above is output to the phase comparator 245 to be described later and input to the SET terminal of the flip-flop 244-3. Accordingly, the charge switch 244-4 is turned off and the discharge switch 244-5 is turned off. Will light up. As described above, when the discharge switch 244-5 is turned on, the discharge current source I2 discharges C1. As a result, the Vc1 voltage continuously decreases and the Sync signal falls back to LOW. ('C' section in the waveform diagram of Figure 10)
이와 같이 제1커패시터의 방전에 의해 Vc1 전압이 일정 수준 떨어진 후 다시 Vdr1 신호가 HIGH로 들어오면 이상에서 설명한 것과 같은 동작을 반복하게 된다. 즉, SP 신호에 의해 스위치 244-6이 ON되어 제1커패시터 C1과 제2커패시터 C2의 도통에 의해 전하 공유가 이루어지게 되며, 이로써 제1커패시터 전압 Vc1이 제2커패시터 전압 Vc2와의 평균값으로 즉시 회복(커패시터 간의 쇼트시 전하 공유시간은 극히 짧으므로 무시해도 좋다)되어 일정한 초기 전압치를 확보할 수 있게 된다. (도10의 파형도에서 'A'점)As described above, when the Vdr1 signal comes HIGH again after the Vc1 voltage drops to a certain level by the discharge of the first capacitor, the same operation as described above is repeated. That is, the switch 244-6 is turned on by the SP signal, and charge sharing is performed by the conduction of the first capacitor C1 and the second capacitor C2, thereby immediately recovering the average value of the first capacitor voltage Vc1 to the second capacitor voltage Vc2. (The charge sharing time during short between capacitors is extremely short and can be ignored.) It is possible to secure a constant initial voltage value. ('A' point in the waveform diagram of Fig. 10)
한편, 상기와 같은 회로에서 I1과 I2가 같고, C1과 C2가 같으면 Vdr1과 Sync는 180도의 위상차를 갖게 되며, I1과 I2, C1과 C2를 조절하면 위상차를 임의의 값으로 자유롭게 조절할 수 있게 된다.On the other hand, in the circuit as described above, if I1 and I2 are the same and C1 and C2 are the same, Vdr1 and Sync have a phase difference of 180 degrees, and by adjusting I1 and I2, C1 and C2, the phase difference can be freely adjusted to an arbitrary value. .
도11은 상기와 같은 본 발명의 동기신호발생부(244)에 대한 또 다른 구현 방식을 도시한 것으로서, 도11의(a)에 도시된 바와 같이 앞서 설명한 예와는 기본적으로 같은 구성을 가지면서, 다만 충/방전의 제어에 있어 RS 플립플롭의 단자와 단자가 바뀜으로써 Sync 신호에 의해 충전 스위치 244-4가 ON 되어 제1커패시터 C1에 대한 충전이 개시되고, 단펄스 SP에 의해서 방전 스위치 244-5가 ON 되어 C1에 대한 방전이 이루어진다는 점을 제외하고는 앞서 설명한 예와 동일하다.FIG. 11 illustrates another implementation manner of the synchronization signal generator 244 of the present invention as described above. As shown in FIG. However, in the charging / discharging control, the terminal and the terminal of the RS flip-flop are switched so that the charge switch 244-4 is turned on by the sync signal to start charging to the first capacitor C1, and the discharge switch 244 is performed by the short pulse SP. Same as the previous example except that -5 is turned on to discharge C1.
도11의(b)는 도11의(a)에 도시된 병렬 연결 스위칭 컨버터의 각 부분에 대한 신호 파형을 도시한 것으로서, 도11의(b)에 도시된 각 파형들을 도10에 도시된 파형과 비교하여 보면, (1)Vc2와 (2)Vc1 및 (3)Vref의 파형에 있어 도10의 파형과 상하만 전도되었을 뿐 근본적인 동작 원리는 같음을 확인할 수 있다. FIG. 11B shows signal waveforms for the respective parts of the parallel-connected switching converter shown in FIG. 11A, and each waveform shown in FIG. 11B is shown in FIG. In comparison with, it can be seen that the waveforms of (1) Vc2, (2) Vc1 and (3) Vref are inverted only up and down in the waveform of FIG.
상기와 같이 동기신호발생부(244)에서 생성된 Sync 신호는 제1컨버터의 스위칭제어신호 Vdr1과 180도의 위상 차이를 갖고 있는바, 일단 이와 같은 동기신호(Sync)가 확보되면 이를 이용하여 제2컨버터의 스위칭 제어신호 Vdr2를 조절하여 Vdr1과 일정한 위상차를 갖는 신호로 생성하는 것은 당업자의 수준에서 비교적 쉽게 구현할 수 있다. 이에 본 발명에서는 상기와 같은 Sync 신호와 Vdr2가 동기화되도록 제어하는 회로에 대한 가장 바람직한 일 예로서 위상 비교를 이용하여 Vdr2의 주기를 Sync 신호에 일치시키는 회로에 대하여 설명한다. As described above, the sync signal generated by the sync signal generator 244 has a phase difference of 180 degrees with the switching control signal Vdr1 of the first converter, and once such a sync signal (Sync) is secured, the second sync signal is used. Controlling the switching control signal Vdr2 of the converter to generate a signal having a constant phase difference from Vdr1 can be relatively easily implemented at the level of those skilled in the art. In the present invention, a circuit for synchronizing the period of Vdr2 to the Sync signal using phase comparison will be described as an example of the circuit for controlling the Sync signal and Vdr2 to be synchronized.
이상에서 설명한 것과 같이 동기신호발생부(244)에서 생성된 Sync 신호는 위상비교부(245)에 입력되어 Vdr2 신호가 Sync 신호와 동기되도록 제어된다. 즉, 상기 위상비교부(245)는 도8에 도시된 바와 같이 제2스위칭제어신호 Vdr2를 피드백 입력받고, 상기 동기신호발생부(244)에서 입력된 Sync 신호와 Vdr2 신호의 위상을 비교하여 Vdr2가 Sync 신호에 뒤지는지(phase lag) 또는 앞서는 지(phase lead)에 따라 위상에러(Phase error) 신호를 생성하여 출력한다. 여기서 상기 위상비교부는 74HC4046 PLL(Phase Lock Loop)와 유사한 기능을 한다.As described above, the sync signal generated by the sync signal generator 244 is input to the phase comparator 245 to control the Vdr2 signal to be synchronized with the sync signal. That is, the phase comparator 245 receives the second switching control signal Vdr2 as a feedback input as shown in FIG. 8, and compares the phases of the sync signal and the Vdr2 signal input from the sync signal generator 244 to Vdr2. Phase error signal is generated and output depending on whether the signal is in phase lag or phase lead. Here, the phase comparison unit functions similar to the 74HC4046 phase lock loop (PLL).
이렇게 위상비교부(245)에서 생성된 위상에러는 제어전압조절부(246)로 입력되고, 상기 제어전압조절부(246)는 Vdr2와 Sync의 위상을 일치시키기 위해서 상기 위상비교부(245)로부터 입력된 위상에러를 기초로 소정의 기준제어전압 Ve을 조절하여 새로운 구동제어전압 Vc를 생성한다. The phase error generated by the phase comparator 245 is input to the control voltage controller 246, and the control voltage controller 246 is supplied from the phase comparator 245 to match the phase of Vdr2 and Sync. Based on the input phase error, the predetermined reference control voltage Ve is adjusted to generate a new driving control voltage Vc.
즉, 상기 제어전압조절부(246)에서는 상기 위상에러(Phase error) 신호가 위상 지연(Phase lag)인 경우에는 주파수를 높이기 위해서 위상 지연 신호에 대응하게 Ve 전압을 조절하여 Vc 전압을 낮추고, 위상 진행(Phase lead)인 경우에는 반대로 주파수를 낮추기 위해서 Ve 전압을 조절하여 Vc 전압을 높인다. (도10의 파형도 참조)That is, when the phase error signal is a phase lag, the control voltage controller 246 lowers the Vc voltage by adjusting the Ve voltage in response to the phase delay signal in order to increase the frequency. In the case of a phase lead, the Vc voltage is increased by adjusting the Ve voltage to decrease the frequency. (See the waveform diagram in FIG. 10)
한편, 여기서, 상기 제어전압조절부(246)에서 사용되는 기준제어전압 Ve는 전원 출력단에 연결된 전압 검출부(104)에서 출력된 전압이 바람직하게 사용될 수 있으며, 이와 같은 전압 검출부에 대해서는 도7을 참조하여 앞서 설명이 이루어졌는바 반복되는 설명은 생략하기로 한다. Meanwhile, the reference control voltage Ve used in the control voltage adjusting unit 246 may preferably be a voltage output from the voltage detecting unit 104 connected to the power output terminal. Referring to FIG. Since the above description has been made, repeated descriptions will be omitted.
다음으로, 도8의 스위칭제어신호발생부(247)에서는 상기와 같이 제어전압조절부(246)에서 생성된 구동제어전압 Vc을 기초로 제2컨버터(200)의 스위치 구동을 위한 제2스위칭제어신호(Vdr2)를 생성한다. 이때, 상기 스위칭제어신호발생부(247)는 도8에 도시된 바와 같이 상기 제어전압조절부(246)에서 입력된 구동제어전압 Vc과 소정의 턴오프 기준전압 Vi2을 비교하여 RS 플립플롭(243)을 RESET 시켜서 Vdr2 신호를 LOW로 만듦으로써 제2컨버터 스위치 SW2를 턴오프시키게 된다.Next, in the switching control signal generation unit 247 of FIG. 8, the second switching control for driving the switch of the second converter 200 based on the driving control voltage Vc generated by the control voltage adjusting unit 246 as described above. Generate the signal Vdr2. In this case, as shown in FIG. 8, the switching control signal generator 247 compares the driving control voltage Vc input from the control voltage adjusting unit 246 with a predetermined turn-off reference voltage Vi2 to generate an RS flip-flop 243. ) To turn off the second converter switch SW2 by making the Vdr2 signal low.
즉, 본 발명에 따르면, 상기와 같이 제2컨버터 스위치(SW2)를 턴오프시키는 시점을 결정함에 있어, Vdr1 신호에 추종하는 Sync 신호와 제2스위칭제어신호 Vdr2를 위상 비교하여 Vdr2가 Sync 신호에 비해 위상이 늦어 Vc 전압을 낮춘 경우에는 그만큼 Vi2가 상승하다가 Vc와 같아지는 시점이 빨라짐으로써 결과적으로 Vdr2의 주기가 단축되게 되며, 반대로 Vdr2가 Sync 신호에 비해 위상이 빨라 Vc 전압이 높아진 경우에는 Vi2가 Vc와 같아지는 시점이 지연되어 Vdr2의 주기가 길어짐으로써 자동적으로 Vdr1에 동기를 맞출 수 있게 되는 것이다. That is, according to the present invention, in determining the time of turning off the second converter switch SW2 as described above, Vdr2 is compared to the Sync signal by phase comparison between the Sync signal following the Vdr1 signal and the second switching control signal Vdr2. In the case of lowering the Vc voltage due to the late phase, the period of Vi2 rises and becomes the same as Vc, resulting in a shorter period of Vdr2. The point at which Vc becomes equal to Vc is delayed, and the period of Vdr2 becomes longer, thereby automatically synchronizing with Vdr1.
[실시예 2]Example 2
한편, 지금까지의 구현 방식은 컨버터의 스위치 전류를 검출하여 전압 검출부(104)의 출력 전압 Ve와 비교하여 스위치를 끄는 시점을 결정하는 전류모드 PWM (pulse width modulation) 을 사용하는 경우에 대해서 설명하였고, 스위치 전류의 정보 대신에 램프 신호와 Ve와 비교하여 스위치를 끄는 시점을 결정하는 전압모드 PWM의 경우에도 스위치를 끄는 시점을 결정하는 방법 외에는 앞에서 설명한 방법과 동일한 방법을 사용하여 컨버터 간의 위상차를 일정하게 유지할 수 있다.On the other hand, the foregoing implementation method has been described in the case of using the current mode pulse width modulation (PWM) for detecting the switch current of the converter to determine when to turn off the switch compared to the output voltage Ve of the voltage detector 104 has been described. In the case of voltage mode PWM which determines when to turn off the switch by comparing the lamp signal and Ve, instead of the information of the switch current, the phase difference between the converters is constant by using the same method as described above. I can keep it.
도12 내지 도14는 전술한 제1실시예와 비교하여 전류 모드 PWM 대신 전압 모드 PWM을 사용한 예를 보인 것으로서, 도12 내지 도14를 참조할 때, 제1스위칭제어부(140)는 램프 신호를 발생하는 램프 발생기(144)를 구비하고, 스위칭 센싱 전압 Vi1 대신에 상기 램프 발생기(144)의 출력 전압 Vr 신호와 전압검출부 전압 Ve를 비교하여 Vdr1 신호를 HIGH에서 LOW로 만드는 시점을 결정하고 있음을 알 수 있다.12 to 14 show an example of using a voltage mode PWM instead of a current mode PWM as compared to the first embodiment described above. Referring to FIGS. 12 to 14, the first switching controller 140 may provide a ramp signal. It is provided with a ramp generator 144 that generates, and compares the output voltage Vr signal of the ramp generator 144 and the voltage detector voltage Ve instead of the switching sensing voltage Vi1 to determine the time to make the Vdr1 signal from HIGH to LOW Able to know.
또한, 제2스위칭제어부(240)에서도 도14에 도시된 것과 같이 Vi2 대신에 램프 발생기(248)의 출력전압 Vr신호와 구동제어전압 Vc를 비교하여 Vdr2 신호를 HIGH에서 LOW로 만드는 시점을 결정한다. 그 외의 부분은 전류모드 PWM을 사용하는 도5 내지 도11에서의 경우와 같이 동작하는바, 별도의 반복적인 설명은 생략한다.In addition, as shown in FIG. 14, the second switching controller 240 compares the output voltage Vr signal of the ramp generator 248 with the driving control voltage Vc instead of Vi2 to determine a time point at which the Vdr2 signal is made from HIGH to LOW. . The other parts operate as in the case of Figs. 5 to 11 using the current mode PWM, and the repetitive description thereof will be omitted.
[실시예 3]Example 3
이상에서는 두 개의 컨버터가 병렬로 연결된 경우를 예로 들어 설명하였으나, 본 발명은 이에 그치지 않고 3개 이상의 컨버터가 병렬로 연결된 경우도 비슷한 방법으로 각 컨버터 간 위상차를 일정하게 제어할 수 있다. 도15 내지 도18은 상기한 제1실시예와 비교하여 3개 이상의 컨버터(100 ~ 300)를 사용한 예를 보인 것이다. 한편, 도15 내지 도18에 도시된 실시예들의 경우 제2동기신호발생부와 제3동기신호발생부를 구성함에 있어 회로의 단순화를 위해 별도의 회로로 구성하지 않고 각각 제2스위칭제어부(240)와 제3스위칭제어부(340)의 회로 내에 포함되도록 구성된 예를 보여주고 있는바, 상기 도15 내지 도18에 도시된 실시예들에 있어서 제2동기신호발생부와 제3동기신호발생부는 도면상에 별도로 도시되지 않고 있다. In the above description, a case where two converters are connected in parallel has been described as an example. However, the present invention is not limited thereto, and in the case where three or more converters are connected in parallel, the phase difference between the converters can be controlled in a similar manner. 15 to 18 show an example using three or more converters 100 to 300 as compared with the first embodiment described above. Meanwhile, in the exemplary embodiments illustrated in FIGS. 15 to 18, the second switching controller 240 is not configured as a separate circuit to simplify the circuit in configuring the second synchronization signal generator and the third synchronization signal generator. And an example configured to be included in a circuit of the third switching controller 340. In the embodiments shown in FIGS. 15 to 18, the second synchronous signal generator and the third synchronous signal generator are shown in the drawings. Not shown separately.
도15 및 도16을 참조하여 보면, 세 개의 컨버터가 병렬로 연결된 경우의 예를 들고 있는데, 여기서 제2스위칭제어부(240) 및 제2동기신호발생부의 기본 회로는 지금까지 설명한 것과 같이 두 개의 컨버터가 병렬로 연결된 경우와 같고, 동기신호발생부(제2스위칭제어부에 포함)를 구성함에 있어서 도10에 도시된 것과 같은 회로에서 I1을 I2의 두 배로 만들어 주면 도16의 파형과 같이 제2컨버터는 제1컨버터와 120도 위상차를 유지하면서 동작하게 된다.15 and 16, an example of three converters connected in parallel is shown. Here, the basic circuits of the second switching controller 240 and the second synchronous signal generator are two converters as described above. Is the same as when connected in parallel, and in the same circuit as shown in FIG. 10 in the synchronizing signal generation unit (included in the second switching control unit), if I1 is doubled to I2, the second converter as shown in the waveform of FIG. Is operated while maintaining a 120 degree phase difference with the first converter.
또한, 제3스위칭제어부(340) 역시 Vdr2 신호를 입력받아 같은 방법으로 동작하게 되면 제3컨버터(300)는 제2컨버터(200)와 120도 위상차를 유지하면서 동작하게 되고, 결과적으로 세 개의 컨버터 모듈은 120도 위상차를 유지하면서 동작하게 된다.In addition, when the third switching controller 340 also receives the Vdr2 signal and operates in the same manner, the third converter 300 operates while maintaining a 120 degree phase difference from the second converter 200. As a result, the three converters The module operates while maintaining a 120 degree phase difference.
나아가, 3개 이상으로 N개의 컨버터가 병렬 연결된 경우에도 기본적으로 상기에서 설명한 것과 같이 구현할 수 있다. Furthermore, even when three or more N converters are connected in parallel, it can be basically implemented as described above.
구체적으로는 제k컨버터의 제어 구동을 위하여 상술한 것과 같이 제1커패시터와 제2커패시터를 구비하고 전하공유에 의해 동기신호를 생성하는 제k동기신호발생부를 더욱 포함하되, 상기 제k동기신호발생부는 동기신호를 생성함에 있어 제1커패시터와 제2커패시터의 전하를 제k-1스위칭제어신호에 의해 일정 시간동안 공유하여 제1커패시터의 전압을 조절하고 상기 제1커패시터의 전압을 기초로 동기신호를 생성하도록 구성된다. Specifically, a k-synchronous signal generation unit including a first capacitor and a second capacitor and generating a synchronization signal by charge sharing as described above for the control driving of the k-th converter, the k-synchronous signal generation In generating the synchronization signal, the controller shares the charges of the first capacitor and the second capacitor with the k-1 switching control signal for a predetermined time to adjust the voltage of the first capacitor and the synchronization signal based on the voltage of the first capacitor. It is configured to generate.
즉, 본 실시예에 따르면, N개의 컨버터가 병렬 연결된 경우, 각각의 컨버터에서 동기신호를 생성함에 있어 인접하고 있는 제k-1스위칭제어부의 제k-1스위칭제어신호에 기초하여 k-1번째 스위칭제어신호와 일정한 위상차를 갖는 k번째 동기신호를 생성하고, 이를 이용하여 k번째 컨버터의 동기제어를 위한 신호를 생성하도록 구성한 것에 그 기술적인 특징이 있다.That is, according to the present embodiment, when the N converters are connected in parallel, the k-1 th based on the k-1 switching control signal of the adjacent k-1 switching control unit in generating the synchronization signal in each converter The technical feature is that the k-th synchronizing signal having a predetermined phase difference with the switching control signal is generated and the signal for synchronizing control of the k-th converter is generated using the k-th synchronizing signal.
[실시예 4]Example 4
도17 및 도18은 세 개의 컨버터가 병렬로 연결된 경우에 있어 전술한 도15에 도시된 실시예 3과 다소 다른 방식으로 구현한 예를 도시한 것이다.17 and 18 show an example implemented in a somewhat different manner from the third embodiment shown in FIG. 15 in the case where three converters are connected in parallel.
본 실시예 4에 있어 제2스위칭제어부(240)의 회로는 기본적으로 도15의 경우와 동일하지만, 제3스위칭제어부(340)에 포함된 제3동기신호발생부는 도15의 경우와는 달리 Vdr2 신호를 입력받는 대신, 제2동기신호발생부에서와 같이 Vdr1의 신호를 입력받아 제1컨버터(100)와 일정한 위상차를 가지면서 동기를 이루는 신호를 생성한다. 이때, 제2동기신호발생부에서는 I1을 I2의 두 배로 만들어 주고 제3동기신호발생부에서 I2를 I1의 두 배로 만들어 주면 도18의 파형과 같이 제2컨버터는 제1컨버터와 120도 위상차를 유지하고 제3컨버터는 제1컨버터와 240도 위상차를 유지하면서 동작하게 된다. 이런 방법으로 세 개의 모듈은 각각 120도 위상차를 유지하면서 동작한다.In the fourth embodiment, the circuit of the second switching controller 240 is basically the same as that of FIG. 15, but the third synchronization signal generator included in the third switching controller 340 is different from the case of FIG. 15. Instead of receiving the signal, as in the second synchronous signal generator, the signal of Vdr1 is input and generates a signal having synchronization with the first converter 100 while having a constant phase difference. At this time, if the second synchronous signal generator makes I1 twice the I2 and the third synchronous signal generator makes I2 twice the I1, as shown in the waveform of FIG. 18, the second converter makes a 120 degree phase difference from the first converter. The third converter operates while maintaining a phase difference of 240 degrees with the first converter. In this way, the three modules operate with a 120-degree phase shift.
상기와 같은 방식은 3개 이상으로 N개의 컨버터가 병렬 연결된 경우에도 그대로 적용될 수 있다. The above method may be applied even when three or more N converters are connected in parallel.
구체적으로는 제k컨버터의 제어 구동을 위하여 상술한 것과 같이 제1커패시터와 제2커패시터를 구비하고 전하공유에 의해 동기신호를 생성하는 제k동기신호발생부를 더욱 포함하되, 상기 제k동기신호발생부는 동기신호를 생성함에 있어 제1스위칭제어부의 제1스위칭제어신호 Vdr1를 입력받아 상기 Vdr1 신호에 의해 제1커패시터의 전압을 조절하고 상기 제1커패시터의 전압을 기초로 동기신호를 생성하도록 구성된다. Specifically, a k-synchronous signal generation unit including a first capacitor and a second capacitor and generating a synchronization signal by charge sharing as described above for the control driving of the k-th converter, the k-synchronous signal generation The unit is configured to receive the first switching control signal Vdr1 of the first switching control unit to adjust the voltage of the first capacitor based on the Vdr1 signal and generate the synchronization signal based on the voltage of the first capacitor in generating the synchronization signal. .
즉, 본 실시예에 따르면, N개의 컨버터가 병렬 연결된 경우, 각각의 컨버터에서 동기신호를 생성함에 있어 공통적으로 제1스위칭제어부의 제1스위칭제어신호 Vdr1를 입력받아 상기 Vdr1 신호를 기초로 동기신호를 생성하도록 하는 것에 그 기술적인 특징이 있으며, 이러한 방식에 따르면 N개의 컨버터 모듈은 서로 일정한 위상차를 유지하면서 동작할 수 있게 된다.That is, according to the present embodiment, when N converters are connected in parallel, the first switching control signal Vdr1 of the first switching control unit is commonly received in generating the synchronization signal in each converter, and the synchronization signal is based on the Vdr1 signal. The technical characteristics of the method are to generate the N, and according to this method, the N converter modules can operate while maintaining a constant phase difference from each other.
[실시예 5]Example 5
한편 이상에서 설명한 경우들은 컨버터 입력 전압이 직류 전압인 경우이고, 교류 전원인 경우에도 같은 방법으로 제어할 수 있다.In the case described above, the converter input voltage is a DC voltage, and the same may be controlled in the case of an AC power source.
도19 및 도20을 참조하여 보면, 입력 전압이 직류 전원에서 교류 전원으로 바뀌고, 전원 입력단에 교류 전원을 직류 전원으로 바꾸기 위한 직류전원 공급부로서 BD(브릿지 다이오드; 105)가 추가된 것, 즉 역률 보상 컨버터를 사용한 것 외에는 앞서 설명한 입력 전압이 직류인 경우와 완전히 동일하다.19 and 20, an input voltage is changed from a DC power source to an AC power source, and a BD (bridge diode) 105 is added as a DC power supply unit for converting an AC power source to a DC power source, that is, a power factor. Except for the use of a compensating converter, the input voltage described above is exactly the same as for direct current.
이상의 실시 예에서는 boost 토폴로지를 기준으로 설명하였으나 buck이나 다른 종류의 SMPS(Switching Mode Power Supply) 토폴로지에서도 상기한 것과 같은 전하 공유를 이용하여 동기신호를 발생하고 이를 이용하여 다수의 컨버터를 제어하도록 하는 방식을 적용하면 병렬 연결된 각 컨버터의 스위칭 위상차를 일정하게 유지할 수 있게 된다. Although the above embodiments have been described based on a boost topology, a buck or other type of switching mode power supply (SMPS) topology generates a synchronization signal using charge sharing as described above and controls a plurality of converters using the same. By applying this, the switching phase difference of each paralleled converter can be kept constant.
이상에서 본 발명은 기재된 실시예를 기준으로 상세히 설명되었으나, 본 발명이 속하는 기술 분야에서 통상의 지식을 가진 자라면 상기에서 설명된 기술적 사상을 벗어나지 않는 범위 내에서 여러가지 치환, 부가 및 변형이 가능할 것임은 당연한 것으로, 이와 같은 변형된 실시 형태들 역시 아래에 첨부한 특허청구범위에 의하여 정하여지는 본 발명의 보호 범위에 속하는 것으로 이해되어야 할 것이다.Although the present invention has been described in detail with reference to the embodiments described, those skilled in the art to which the present invention pertains will be capable of various substitutions, additions and modifications without departing from the technical spirit described above. It is to be understood that such modified embodiments also fall within the protection scope of the present invention as defined by the appended claims below.
본 발명은 큰 출력을 확보하기 위해 다수개의 컨버터들을 병렬로 연결하여 사용하는 스위칭 모드 파워 서플라이에 적합하게 적용될 수 있다. 이러한 본 발명이 적용된 스위칭 모드 파워 서플라이는 병렬 연결된 컨버터들 모두가 DCM(discontinuous conduction mode)으로 동작함으로써 역회복 현상에 의해 발생할 수 있는 스위칭 손실과 스위칭 노이즈의 발생이 효과적으로 방지되는 장점이 있다. The present invention can be suitably applied to a switching mode power supply using a plurality of converters connected in parallel to secure a large output. The switching mode power supply to which the present invention is applied has an advantage in that all of the parallel-connected converters operate in a discontinuous conduction mode (DCM), thereby effectively preventing the generation of switching loss and switching noise caused by reverse recovery.
Claims (12)
- 전원 입력단에 제1컨버터와 제2컨버터를 포함하는 복수개의 컨버터가 병렬로 연결되고, 상기 복수개의 컨버터의 턴온 및 턴오프를 조절하여 컨버터들 간의 위상차를 일정하게 유지하는 병렬 연결 스위칭 컨버터에 있어서, In the parallel-connected switching converter that is connected in parallel with a plurality of converters including a first converter and a second converter in parallel, by adjusting the turn on and off of the plurality of converters to maintain a constant phase difference between the converters,상기 제1컨버터의 턴온 및 턴오프를 제어하기 위한 제1스위칭제어신호를 생성하는 제1스위칭제어부와;A first switching controller for generating a first switching control signal for controlling the turn-on and turn-off of the first converter;제1커패시터와 제2커패시터를 구비하고, 상기 제1스위칭제어부의 제1스위칭제어신호에 따라 상기 제1커패시터와 제2커패시터를 도통시켜 상기 1커패시터와 제2커패시터 간의 전하공유(charge sharing)를 통해 상기 제1커패시터의 전압을 조절하고, 상기 제1커패시터의 전압을 이용하여 상기 제1스위칭제어신호와 동기를 이루면서 일정한 위상차를 갖는 동기신호를 생성하는 동기신호발생부; 및,A first capacitor and a second capacitor, and conducting charge sharing between the first capacitor and the second capacitor by conducting the first capacitor and the second capacitor according to the first switching control signal of the first switching controller. A synchronization signal generator for adjusting a voltage of the first capacitor and generating a synchronization signal having a predetermined phase difference in synchronization with the first switching control signal using the voltage of the first capacitor; And,상기 동기신호발생부에서 생성된 동기신호에 따라 제2스위칭제어신호를 생성하여 상기 제2컨버터의 턴온 및 턴오프를 제어하는 제2스위칭제어부;A second switching controller configured to generate a second switching control signal according to the synchronization signal generated by the synchronization signal generator to control turn-on and turn-off of the second converter;를 포함하여 구성된 병렬 연결 스위칭 컨버터.Parallel-connected switching converter configured to include.
- 제1항에 있어서, 상기 동기신호발생부는, The method of claim 1, wherein the synchronization signal generator,제1커패시터;A first capacitor;상기 제1커패시터와 스위치 소자를 통해 연결되어 상기 스위치 소자의 닫힘(ON)에 의해 상기 제1커패시터와 전하 공유를 할 수 있도록 연결된 제2커패시터;A second capacitor connected to the first capacitor through a switch element and connected to enable charge sharing with the first capacitor by being closed (ON) of the switch element;상기 제1커패시터의 전압과 기준전압을 비교하여 동기신호를 생성하는 비교부; 및,A comparator configured to generate a synchronization signal by comparing the voltage of the first capacitor with a reference voltage; And,상기 제1커패시터와 연결되어 상기 동기신호 및 상기 제1스위칭제어신호에 의해 상기 제1커패시터를 충전 또는 방전시키는 충방전부;A charge / discharge unit connected to the first capacitor to charge or discharge the first capacitor by the synchronization signal and the first switching control signal;를 포함하여 구성된 것을 특징으로 하는 병렬 연결 스위칭 컨버터.Parallel-connected switching converter, characterized in that configured to include.
- 제2항에 있어서, 상기 동기신호발생부는, 상기 제1스위칭제어신호를 입력받아 상기 제1스위칭제어신호에 비해 하이(HIGH) 기간이 짧은 단펄스로 변환하는 단펄스발생부를 더욱 포함하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.3. The apparatus of claim 2, wherein the synchronization signal generator further includes a short pulse generator configured to receive the first switching control signal and convert the first switching control signal into a short pulse having a shorter duration than the first switching control signal. Paralleled switching converter.
- 제1항 내지 제3항 중 어느 한 항에 있어서, 상기 제2스위칭제어부는, The method according to any one of claims 1 to 3, wherein the second switching control unit,상기 제2스위칭제어신호를 피드백 입력받고, 상기 동기신호발생부에서 생성된 동기신호와 상기 제2스위칭제어신호의 위상을 비교하여 위상 차이에 따른 위상에러를 출력하는 위상비교부와; A phase comparing unit receiving a feedback input of the second switching control signal and comparing a phase of the synchronization signal generated by the synchronization signal generation unit with a phase of the second switching control signal and outputting a phase error according to a phase difference;상기 위상비교부에서 출력된 위상에러를 기초로 기준제어전압을 조절하여 구동제어전압을 생성하는 제어전압조절부; 및,A control voltage adjusting unit generating a driving control voltage by adjusting a reference control voltage based on the phase error output from the phase comparing unit; And,상기 제어전압조절부에서 생성된 구동제어전압을 이용하여 제2스위칭제어신호를 발생하는 스위칭제어신호발생부;A switching control signal generation unit generating a second switching control signal using the driving control voltage generated by the control voltage adjusting unit;를 더욱 포함하여 구성된 것을 특징으로 하는 병렬 연결 스위칭 컨버터.Parallel-connected switching converter, characterized in that further comprises a.
- 제4항에 있어서, 상기 제1컨버터와 제2컨버터의 출력 전압을 검출하는 전압검출부를 더욱 포함하며, 상기 제어전압조절부의 기준제어전압은 상기 전압검출부에서 검출된 전압에 대응하는 전압인 것을 특징으로 하는 병렬 연결 스위칭 컨버터.5. The apparatus of claim 4, further comprising a voltage detector configured to detect output voltages of the first converter and the second converter, wherein the reference control voltage of the control voltage controller is a voltage corresponding to the voltage detected by the voltage detector. Paralleled switching converter.
- 제4항에 있어서, 상기 스위칭제어신호발생부는 상기 제어전압발생부에서 생성된 구동제어전압과 제2컨버터에 포함된 컨버터 스위치를 통과하는 전류에 의해 스위칭 센싱 저항에 인가되는 전압을 비교하여 제2스위칭제어신호를 생성하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The method of claim 4, wherein the switching control signal generator is configured to compare the driving control voltage generated by the control voltage generator with a voltage applied to the switching sensing resistor by a current passing through the converter switch included in the second converter. A parallel connection switching converter, characterized in that for generating a switching control signal.
- 제4항에 있어서, 상기 제2구동부에는 램프 파형을 발생하는 램프발생기가 더욱 구비되며, 상기 스위칭제어신호발생부는 상기 제어전압발생부에서 생성된 구동제어전압과 상기 램프발생기에서 발생되는 램프 전압을 비교하여 제2스위칭제어신호를 생성하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The method of claim 4, wherein the second driver is further provided with a ramp generator for generating a ramp waveform, wherein the switching control signal generator is a driving control voltage generated by the control voltage generator and the lamp voltage generated by the lamp generator And comparing the second switching control signal with the second switching control signal.
- 제1항에 있어서, 상기 제1컨버터에 유입되는 전류가 영(0)이 되는지를 검출하는 제1영전류검출부와; 상기 제2컨버터에 유입되는 전류가 영(0)이 되는지를 검출하는 제2영전류검출부를 더욱 포함하여 구성되며,2. The apparatus of claim 1, further comprising: a first zero current detector for detecting whether a current flowing into the first converter is zero; It further comprises a second zero current detector for detecting whether the current flowing into the second converter is zero,상기 제1스위칭제어부는 상기 제1영전류검출부에서 전류가 영으로 검출되면 상기 제1컨버터를 턴온시키는 제1스위칭제어신호를 생성하고, 상기 제2스위칭제어부는 상기 제2영전류검출부에서 전류가 영으로 검출되면 상기 제2컨버터를 턴온시키는 제2스위칭제어신호를 생성하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The first switching controller generates a first switching control signal for turning on the first converter when a current is detected as zero by the first zero current detector, and the second switching controller generates a current by the second zero current detector. And if it is detected as zero, generating a second switching control signal for turning on the second converter.
- 제8항에 있어서, 상기 제1영전류검출부는 한 쌍의 코일에 의해 전원 입력단과 커플링되어 유도전압이 형성되는 제1인덕터를 포함하고, 상기 제1스위칭제어부는 상기 제1인덕터에 유도된 전압과 기준 전압을 비교하여 상기 제1컨버터를 턴온시키는 제1스위칭제어신호를 생성하며,The method of claim 8, wherein the first zero current detector includes a first inductor coupled to a power input terminal by a pair of coils to form an induced voltage, and the first switching controller is connected to the first inductor. Comparing the voltage with the reference voltage to generate a first switching control signal for turning on the first converter,상기 제2영전류검출부는 한 쌍의 코일에 의해 전원 입력단과 커플링되어 유도전압이 형성되는 제2인덕터를 포함하고, 상기 제2스위칭제어부는 상기 제2인덕터에 유도된 전압과 기준 전압을 비교하여 상기 제1컨버터를 턴온시키는 제2스위칭제어신호를 생성하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The second zero current detector includes a second inductor coupled to a power input terminal by a pair of coils to form an induced voltage, and the second switching controller compares the voltage induced in the second inductor with a reference voltage. And generating a second switching control signal for turning on the first converter.
- 제1항에 있어서, 상기 병렬 연결 스위칭 컨버터는 상기 제1컨버터와 제2컨버터에 병렬로 연결된 제3컨버터를 더욱 포함하고,The method of claim 1, wherein the parallel-connected switching converter further comprises a third converter connected in parallel to the first converter and the second converter,상기 제1스위칭제어부의 제1스위칭제어신호를 받아 상기 제1스위칭제어신호와 동기를 이루면서 일정한 위상차를 갖는 동기신호를 생성하는 제3동기신호발생부와;A third synchronous signal generator which receives the first switching control signal of the first switching control unit and generates a synchronous signal having a predetermined phase difference in synchronization with the first switching control signal;상기 제3동기신호발생부에서 생성된 동기신호에 따라 상기 제3컨버터의 턴온 및 턴오프를 제어하기 위한 제3스위칭제어신호를 생성하는 제3스위칭제어부;를 더욱 포함하여 구성되며,And a third switching controller for generating a third switching control signal for controlling the turn-on and turn-off of the third converter according to the synchronization signal generated by the third synchronization signal generator.상기 제3동기신호발생부는 제1커패시터와 제2커패시터를 구비하고, 상기 제1스위칭제어신호에 따라 상기 제1커패시터와 제2커패시터를 도통시켜 전하 공유를 통해 상기 제1커패시터의 전압을 조절하며 상기 제1커패시터의 전압을 이용하여 동기신호를 생성하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The third synchronous signal generator includes a first capacitor and a second capacitor, conducts the first capacitor and the second capacitor according to the first switching control signal, and adjusts the voltage of the first capacitor through charge sharing. And a synchronization signal is generated using the voltage of the first capacitor.
- 제1항에 있어서, 상기 병렬 연결 스위칭 컨버터는 상기 제1컨버터와 제2컨버터에 병렬로 연결된 제3컨버터를 더욱 포함하고,The method of claim 1, wherein the parallel-connected switching converter further comprises a third converter connected in parallel to the first converter and the second converter,상기 제2스위칭제어부의 제2스위칭제어신호를 받아 상기 제2스위칭제어신호와 동기를 이루면서 일정한 위상차를 갖는 동기신호를 생성하는 제3동기신호발생부와;A third synchronous signal generator which receives the second switching control signal of the second switching control unit and generates a synchronous signal having a predetermined phase difference in synchronization with the second switching control signal;상기 제3동기신호발생부에서 생성된 동기신호에 따라 상기 제3컨버터의 턴온 및 턴오프를 제어하기 위한 제3스위칭제어신호를 생성하는 제3스위칭제어부;를 더욱 포함하여 구성되며,And a third switching controller for generating a third switching control signal for controlling the turn-on and turn-off of the third converter according to the synchronization signal generated by the third synchronization signal generator.상기 제3동기신호발생부는 제1커패시터와 제2커패시터를 구비하고, 상기 제2스위칭제어신호에 따라 상기 제1커패시터와 제2커패시터를 도통시켜 전하 공유를 통해 상기 제1커패시터의 전압을 조절하고 상기 제1커패시터의 전압을 이용하여 동기신호를 생성하는 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The third synchronous signal generator includes a first capacitor and a second capacitor, and conducts the first capacitor and the second capacitor according to the second switching control signal to adjust the voltage of the first capacitor through charge sharing. And a synchronization signal is generated using the voltage of the first capacitor.
- 제1항에 있어서, 상기 병렬 연결 스위칭 컨버터는 전원 입력단에 교류 전원을 직류 전원으로 변화하여 공급하는 직류전원 공급부가 포함되어 있는 역률 보상 컨버터인 것을 특징으로 하는 병렬 연결 스위칭 컨버터.The parallel-connected switching converter according to claim 1, wherein the parallel-connected switching converter is a power factor correction converter including a DC power supply unit configured to supply AC power to DC power at a power input terminal.
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