Description
PRIMARY PARALLELED THREE-PHASE FULL-BRIDGE DC- DC CONVERTER
CROSS-REFERENCE TO RELATED APPLICATION
[1] The present invention claims the benefit of provisional U.S. Patent Application
Serial No. 60/608,070 entitled, "Primary Paralleled Three-Phase Full-Bridge DC-DC Converter", and filed September 8, 2004.
BACKGROUND OF THE INVENTION
[2] The present invention relates generally to voltage conversion and, more par¬ ticularly, a primary paralleled, three-phase full-bridge DC-DC voltage converter.
[3] Direct-current-to-direct-current voltage converters (DC-DC converters) are used frequently in electrical and electronic systems to convert one voltage potential to another voltage potential. High power DC-DC converters with galvanic isolation are needed, for example, in hybrid electric trucks, such as utility trucks, to interface DC energy storage to the auxiliary AC power generation devices. Generally, DC-DC converters are designed to accept a DC input voltage and produce a DC output voltage that is typically at a different voltage level than the input voltage. In addition to modifying voltage levels, DC-DC converters provide noise isolation, power bus regulation, and the like. In typical isolating DC-DC converters, an input DC voltage is converted to an AC voltage using high frequency switching of the input voltage. The AC voltage is then fed to a transformer that converts the input AC voltage to another AC voltage. The AC output of the transformer is then fed to a rectifier circuit that converts the AC voltage to a DC voltage. The DC output of the rectifier can then have a voltage level greater than, less than or equal to the original DC input. Not only does the transformer "buck" or "boost" the voltage level, but it also provides isolation between the DC input and the DC output.
[4] One challenge that is faced when designing DC-DC converters is the efficiency of such converters. Another challenge is to cover a wide range of input voltage variation with controlled output voltage. In transformer isolated DC-DC converters, duty-cycle loss due to the commutation overlap in diode rectifier may cause poor power throughput, especially for high load current operation. Conventional single-phase DC- DC converters typically produce a low ripple frequency resulting in a higher DC inductance.
[5] Another challenge faced when designing DC-DC converters is the transformer core volume. A three-phase transformer core typically has less volume compared to three single-phase transformer cores. In addition, splitting the three-phase high-frequency
transformer into two pieces may lead to even more volume savings because larger total core surface area can be obtained for the same total core volume.
[6] Therefore, it would be desirable to have an isolating DC-DC converter that suf¬ ficiently provides power for a high power loads without the need for relatively large transformer cores, and can control the output DC voltage.
BRIEF DESCRIPTION OF THE INVENTION
[7] The present invention is directed to an apparatus and method of DC to DC conversion that overcomes the aforementioned drawbacks.
[8] A DC-DC converter is presented having two active, three-phase full-bridge DC-AC inverters arranged in parallel on a primary side of a pair of high-frequency transformers. Modulation schemes of the three-phase full-bridges include phase-shifted operation between two six step controlled three-phase bridges. A second scheme includes a pulse width modulation (PWM) operation in each three-phase active bridge.
[9] In accordance with one aspect of the present invention, a voltage converter includes a first three-phase full-bridge inverter having an input and an output, and a second three-phase full-bridge inverter having an input and an output. The input of the first three-phase full-bridge inverter is connected in parallel with the input of the second three-phase full-bridge inverter.
[10] In accordance with another aspect of the present invention, a voltage converter includes a first transformer having a primary set of windings and a first voltage inverter connected to the primary set of windings of the first transformer. A second transformer is included that has a primary set of windings, and a second voltage inverter is connected to the primary set of windings of the second transformer. The primary set of windings of the first transformer and the first voltage inverter are in parallel with the primary set of windings of the second transformer and the second voltage inverter.
[11] In accordance with yet another aspect of the present invention, a method of supplying conditioned power to a load includes the step of receiving a DC input voltage. The method includes the steps of converting the DC input voltage into a first three-phased AC voltage and converting the DC input voltage into a second three- phased AC voltage. The first three-phased AC voltage is supplied to a first transformer and the second three-phased AC voltage is supplied to a second transformer in parallel with the first transformer.
[12] Various other features and advantages of the present invention will be made apparent from the following detailed description and the drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
[13] The drawings illustrate one preferred embodiment presently contemplated for
carrying out the invention. [14] In the drawings:
[15] Fig. 1 is a schematic diagram of a DC-DC converter in accordance with the present invention.
[16] Fig. 2 is a graph illustrating v out vs. phase-shift angle, φ , characteristics for the DC-
DC converter of Fig. 1 having a ΔΔ-Y transformer configuration.
[17] Fig. 3 is a graph illustrating v out vs. phase-shift angle, φ , characteristics for the DC-
DC converter of Fig. 1 having a YY- Δ transformer configuration.
[18] Fig. 4 is a graph illustrating v vs. phase-shift angle, φ , characteristics for the DC- out
DC converter of Fig. 1 having a ΔΔ-Δ or a YY-Y transformer configuration. [19] Fig. 5 is a graph illustrating v out vs. phase-shift angle, φ , characteristics for the DC-
DC converter of Fig. 1 having a ΔY-Y transformer configuration. [20] Fig. 6 is a graph illustrating v out vs. phase-shift angle, φ , characteristics for the DC-
DC converter of Fig. 1 having a ΔY- Δ transformer configuration.
[21] Figs. 7-9 are graphs showing typical transformer and rectifier output voltages of the
DC-DC converter of Fig. 1.
[22] Fig. 10 is a graph illustrating the relation between output voltage, v out , and phase- shift angle, φ of the DC-DC converter of Fig. 1.
[23] Figs. 11-14 show representative circuit simulation results using computer software for the DC-DC converter of Fig. 1.
[24] Fig. 15 is a schematic diagram of another DC-DC converter in accordance with the present invention.
[25] Fig. 16 is a graph showing an exemplary pulse width modulation (PWM) switching control sequence for DC-DC converter of Figs. 1 and 15.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
[26] Fig. 1 shows a schematic diagram of a DC-DC converter 10 in accordance with one exemplary embodiment of the present invention. Converter 10 has a pair of active three-phase full-bridge inverters 14, 16 that receive a DC input voltage 12. The inverters 14, 16 are connected to a primary winding 18, 19 of a pair of high-frequency transformers 20, 22, respectively. Each three-phase full-bridge inverter 14, 16 contains a plurality of high frequency power switches 24. In a preferred embodiment, the plurality of high frequency power switches 24 are Insulated Gate Bipolar Transistors (IGBTs); however, one skilled in the art would appreciate that other high frequency power switches may be used, such as Bipolar Junction Transistors (BJTs), Metal- Oxide-Semiconductor Field-Effect Transistors (MOSFETs), and the like. Three-phase full-bridge inverters 14, 16 convert the DC input voltage 12 into an AC voltage for input into the primary windings 18, 19 of transformers 20, 22. The AC voltage (v abl α and v abl β ) at the output of each three-phase full-bridge inverter 14, 16, are at the same
fundamental frequency and are phase shifted from each other by an angle φ.
[27] A secondary winding 26 of transformer 22 is serially connected to a secondary winding 28 of transformer 20. The AC voltage at the output of the series windings 26 and 28 is preferably a multi-level waveform with a maximum of seven levels. The AC voltage input into the primary windings 18, 19 of transformers 20, 22 is converted by transformers 20, 22, and the AC output is fed into a three-phase rectifier bridge 30 connected to secondary winding 28 of transformer 20. Preferably, three-phase rectifier bridge 30 includes a pair of diodes 32 for each phase. The three-phase rectifier bridge 30 is connected to an L-C filter circuit 34, and a DC output voltage 36 is fed to a load 38. Standard six-pack, three-phase full-bridge modules can be utilized for both the active bridges and the passive bridge with potential cost savings. In the exemplary embodiment, the transformers are designed to "boost" the voltage input thereto, but is contemplated that the transformers may also be designed to reduce or "buck" the input voltage.
[28] As set forth, the three-phase transformer core advantageously has less volume than three single-phase transformer cores designed to provide a similar voltage conversion. In addition, splitting the three-phase high-frequency transformer into two sections is believed to provide additional volume savings because larger total core surface area can be obtained for the same total core volume.
[29] As shown in Fig. 1, the transformer configuration of transformers 20, 22 is ΔΔ-Y, where primary winding 18 of transformer 20 is in a Δ configuration, the primary winding 19 of second transformer 22 is in a Δ configuration, and secondary windings 26, 28 of transformers 20 and 22 are connected in series to form a Y winding con¬ figuration. However, one skilled in the art will recognize that other transformer con¬ figurations, such as ΔΔ-Δ, YY-Δ, YY-Y, ΔY-Y and ΔY-Δ, are possible. Figs. 2-6 show output voltage, v , vs. phase-shift angle, φ, relations where output voltage is out normalized by nv , for the various transformer configurations. in
[30] Figs. 2 and 3 show v out vs. φ characteristics for transformer configurations ΔΔ-Y and YY-Δ, respectively, of the DC-DC converter 10 of Fig. 1. As shown, the v out vs. φ characteristics for transformer configurations ΔΔ-Y and YY-Δ are similar. The waveforms differ by a factor of three because of the voltage relation of delta windings and wye windings. Hence, its output voltage is also different by the factor of three at any given value of φ. This scaled relation suggests that ΔΔ-Y is preferred for higher output voltage design and YY-Δ is suitable for lower output voltage design. Further, the maximum number of available voltage levels in the transformer secondary multi¬ level waveforms for these transformer configurations is seven. [31] Fig. 4 shows that ΔΔ-Δ and YY-Y transformer configurations of the DC-DC converter 10 of Fig. 1 share the same v vs. φ characteristics. Since at least one of the
secondary line-to-line voltages is 2nv or -2nv over a full switching cycle up to φ = in in
60°, v out does not change for φ = 0 ~ 60°. Excluding this range of phase-shift angle, v out can be controlled over 0 ~ 2nv in without slope change. This facilitates easier controller tuning over the full-range of output voltage than the cases of ΔΔ-Y and YY-Δ. However, since only one of the delta or wye configurations is used on both primary and secondary windings, the maximum number of available voltage levels in the transformer secondary multi-level waveforms, five levels, is lower than for the ΔΔ-Y or YY-Δ configurations.
[32] The highest absolute secondary line-to-line voltage among three line-to-line voltages appears at the diode rectifier output per the operating principle of a diode rectifier. Table 1 summarizes the voltage levels that appear at the rectifier output, v rect , for different regions of phase-shift angle. Table 1 : Voltage levels at diode rectifier output
Φ ΔΔ-Y YY-Δ ΔΔ-Δ, YY-Y
0 - 60° 4nvin ~ 3nvin (4/3)nvin ~ - (3/3)nvin
60 - 120° 3nvin ~ 2nvin (3/3)nvin ~ 2nvin - nvin (2/3)nvin
120 - 180° 2nvin ~ 0 (2/3)nvin ~ 0 nvin ~ 0
[33] Figs. 5 and 6 show v out vs. φ characteristics of the DC-DC converter 10 of Fig. 1 for mixed primary winding configurations ΔY-Y and ΔY-Δ, respectively. In these cases, two voltage waveforms, which are summed at the series-connected secondary winding terminals, are different. Hence, complete voltage cancellation does not occur at φ = 180°, and v out does not reach zero a ,ccordingly.
[34] The modulation strategy of the primary paralleled, three-phase full-bridge DC-DC converter 10 is based on the combination of two schemes. A first scheme is preferably a three-phase, six-step operation in each active three-phase full-bridge 14, 16. A second scheme is a phase-shifted operation between the active three-phase full-bridges 14, 16. Although a duty ratio of each line-to-line voltage is fixed as two-thirds due to the three-phase six-step operation, changing the phase-shift angle φ between three- phase full-bridges 14, 16 can adjust an overlap period of two series-connected transformer secondary voltages whereby the output voltage can be controlled.
[35] Figs. 7-9 show typical transformer and rectifier input/output voltages of DC-DC converter 10 of Fig. 1. One of the three output line-to-line voltages of each three-phase full-bridge 14, 16 is shown along with one of the three output line-to-line voltages of transformers 20, 22 and the output voltage of three-phase rectifier bridge 30. The
voltages v abl α and v abl β are normalized by in rput voltag °e v in , and the voltag °e v ab2 is normalized by nv in , where n is the transformer turns-ratio normalized by the number of primary turns. A switching frequency, f , of 10 kHz is used. As shown in Figs. 7-9, the transformer secondary line-to-line voltages (only v ab2 is shown in the figures) have multi-level waveforms. The highest absolute secondary line-to-line voltage among the three secondary line-to-line voltages appears at the three-phase rectifier bridge 30 output. The three-phase rectifier bridge 30 output voltage, v rect , is (a) 3nv in + square wave with height nv in as shown in Fig. 7, (b) 2nv in + square wave with height nv in as shown in Fig. 8, or (c) 0 + square wave with height 2nv as shown in Fig. 9, depending on φ. The closed form input-output voltage relation is:
(Eqn. 1), for 0° < φ < 120°, and
(Eqn. 2), for l20° < φ < 180°.
[36] As can be seen from Figs. 7-9, rectifier output voltage ripple frequency is 6f . This is three times the rectifier output voltage ripple frequency in conventional single-phase full-bridge DC-DC converters. This higher ripple frequency results in a smaller DC inductance than those in the conventional single-phase full-bridge DC-DC converter.
[37] In addition, the multi-level voltage in rectifier output, v , provides an even smaller DC inductance than the conventional single-phase or three-phase full-bridge DC-DC converters for the same peak-to-peak DC inductor ripple current restriction. This is because of a lower driving voltage across the DC inductor, L , due to the multi- level voltage waveform in the rectifier output. [38] Fig. 10 shows a trace 40 that illustrates the relation between output voltage, v out , and phase-shift angle, φ, of the DC-DC converter 10 of Fig. 1. At a phase-shift angle of 120°, the slope of trace 40 changes because the height of the square wave portion in rectifier output voltage, v rect , changes from nv in to 2nv in at this angle.
[39] In the converter shown in Figure 1, a careful inspection of commutation processes shows that the difference between incoming and outgoing line-to-line voltages for a diode commutation is 2nv in for 0° < φ < 60° and nv in for 60° < φ < 180° . Complicated situations, in which one of the active three-phase full-bridges initiates a commutation in the diode rectifier and the other active three-phase full-bridge takes switching action before the commutation completes, may occur at about φ = 60° . Since the rectifier output voltage, v , is the average of the absolute values of incoming and outgoing
line-to-line voltages during a commutation period, the lost volt-second during this period is (nv in )t ov for 0 ° < φ < 60° and (0.5nv in )t ov for 60° < φ < 180°, where t ov is the commutation overlap time.
[40] The closed form input-output voltage relation with taking the duty-cycle loss into account is:
(Eqn. 3), for 0° < φ < 60°,
(Eqn. 4), for 60° < φ < 120°,
(Eqn. 5), for 120° < φ < 180°, where
(Eqn. 6), for 0° < φ < 60°, and
(Eqn. 7), for 60° < φ < 180°.
[41] In Eqns. 3, 4, and 5, their first terms are the same as Eqns. 1 and 2 for each cor¬ responding φ, and their second terms are the effects of duty cycle loss. In Eqns. 6 and 7, L Ltot2ph is the secondary J referred total leakag °e inductance * per r phase-cou Vpling ° and i Ldc is the DC current flowing through DC inductor L dc .
[42] Now, the closed form input-output voltage relation for the well-known single-phase full-bridge isolated DC-DC converter with taking the duty-cycle loss into account is:
^ w, - — n ' ' single - n ΓL>Λ ~ ^ 9f'sw n ' l single r Uv
(Eqn. 8), for 0° < φ < 180°, where
Llioi2singl& 2 J Ldc tov = n Xf i ' singte vin
(Eqn. 9), for O° < φ < 180°. [43] In Eqns. 8 and 9, n is the turns-ratio of single-phase high-frequency transformer single normalized by the number of primary turns and L is the secondary referred total
Uot2single
leakage inductance.
[44] Since duty cycle loss is a problem for high voltage output with high load current, a comparison of Eqn. 3 combined with Eqn. 6 and Eqn. 8 combined with Eqn. 9 is sufficient. Then, the duty cycle loss terms for the primary paralleled three-phase and single-phase DC-DC converters are:
(Eqn. 10), for 0° < φ < 60°, and
V lost single fsw J Ldc
— 4 L Ltot2 single Vin Vin
(Eqn. 11), for 0° < φ < 180°, respectively. Assuming that f sw , i Ldc and v in are the same for both DC-DC converters, the break-even can be shown by the ratio of L Ltot2ph : L
Ltot2single = 1 : 3. Making 3L Ltot2ph < L Ltot2single is not difficult if input-output voltage speci- fications are the same for two converters, because n : n single = 1 : 4 due to the ΔΔ-Y configuration, where a factor of two comes from Δ-Y configuration and another factor of two comes from series connection of secondary windings. Given that the number of turns weig bhs the inductance with sq ^uared manner, L Ltot2ph can be one-sixteenth of L
. This advantage may be utilized to increase power throughput of the converter
Ltot2single or to relax the tolerable limit of leakage inductance in the high-frequency transformers.
[45] Figs. 11-14 illustrate representative circuit simulation results obtained using Matlab® Simulink® of the DC-DC converter 10 of Fig. 1 . Matlab® and Simulink® are registered trademarks of The MathWorks, Inc. of Natick, Massachusetts. Table 2 ' shows specification summary of the DC-DC converter simulated. The assumed ap¬ plication is an auxiliary power conversion unit of a hybrid electric vehicle which is equipped with a high-voltage battery stack. The operating voltage of the battery stack is 240V-400V (340V nominal). In addition to the specifications in Table 2, 2.5mH of magnetizing inductance and 3 μH of secondary referred total leakage inductance are assumed for each phase of the high-frequ ency transformers. Table 2: Specification summary of simulated DC-DC converter
[46] Fig. 11 shows primary side voltage and current waveforms of the leading converter and Fig. 12 shows those of the lagging converter at the rated operating point, i.e. v in =
340V, v out = 400V and load resistance is 5.9 Ω, where φ = 85°. The line current values
42 overlaid with line-to-line voltages 44 are artificially doubled for the sake of clearer presentation. Traces 46, 48, and 50 represent delta winding currents a, b, and c, re¬ spectively.
[47] As described above, it can be seen from Figs. 11 and 12 that both the leading converter configuration and the lagging converter configuration are under six-step operation with switching frequency f = 10kHz. Their switching cycles are syn¬ chronized with controlled phase angle difference φ . Accordingly, power transfer frequency through each transformer coupling is the same as f . Since the transformer secondary windings are connected in series, the currents in the leading and lagging converters are in phase. The currents flowing through the transformer windings are trapezoidal and the line current waveforms are like six-step due to Δ -Y coupling.
[48] Fig. 13 shows diode rectifier voltage and current waveforms for the same operating point. Traces 52, 54, and 56 are secondary side AC currents a, b, and c, respectively. Trace 58 illustrates the rectified voltage, v rect . The DC inductor current, i Ldc , is il- lustrated in trace 60. The effect of commutation overlap can be seen, which was neglected in operating principle as described above in Figs. 7-10. For this particular operating point, the outgoing voltage is nv in = 170V and the incoming voltage is 2nv in
= 340V. Accordingly, the rectifier output voltage during a commutation overlap period is (3/2)nv in = 255V. This voltage is observed in the mid trace 58 of Fig. 13 during each commutation. The repetitive square wave frequency in the rectified voltage is 6f , as expected. This high frequency and smaller voltage swing between 3nv in and (3/2)nv in results in small peak-to-peak current ripples in the DC inductor, as can be seen in trace 60.
[49] Fig. 14 shows two AC side currents 62, 64 (phase a and b) of the diode rectifier and two anode-cathode voltages 66, 68 of the upper diodes for these two phases in order to show incoming/outgoing diode currents and voltages. Looking at the commutation at around 0.01384 sec for example, it can be seen that the driving voltage for the commutation is nv in = 170V. This low commutation driving voltage allows for mild turn-off processes in the switching diodes.
[50] An exemplary primary paralleled, three-phase full-bridge DC-DC converter 10 of
Fig. 1 was constructed and tested. Its specification summary was the same as in Table 2 except that C dc = 1650 μF. Tests of transformer voltages and currents with no-load
for φ = 33° and φ = 102°, respectively, where v i.n = 340V, illustrated that different multi-level voltage waveforms appear at the transformer secondary terminal depending on phase-shift angle φ, as expected from its operating principle. [51] Tests of transformer secondary line-to-line voltage, v ab2 , and a diode anode-cathode voltage with low input voltage, v in = 50V, under a lightly loaded condition, load = 24
Ω, illustrated the reverse bias voltage at diode turn-off can be low because of the multilevel voltage waveform, as expected from its operating principle. It was found that a voltage spike was induced by the parasitic inductance including the transformer leakage and di/dt of diode reverse recovery and that the voltage oscillation was due to the L-C resonance between the parasitic inductance and diode junction capacitance. Therefore, in an exemplary embodiment, a voltage clamp (not shown) or snubbing in the diode bridge may reduce voltage spikes and oscillations.
[52] Fig. 15 shows another embodiment of a DC-DC converter 70 according to the present invention. The DC-DC converter 70 has a pair of three-phase full-bridges 72, 74 and the primary windings 76, 78 of a pair of transformers 80, 82 connected in parallel to a DC input voltage 84. The secondary windings 86, 88 are each connected to a three-phase diode rectifier and L-C filter set 90, 92, respectively. The three-phase diode rectifier and L-C filter sets 90, 92 are connected in series allowing a voltage stress across each diode 94 to be one-half of the voltage stress across each diode 32 in the three-phase rectifier bridge 30 of the DC-DC converter 10 of Fig. 1. In this manner, a lossless voltage clamp scheme may be introduced. A DC output voltage 96 is fed to a load 98.
[53] As shown in Fig. 15, the transformer configuration of transformers 80, 82 is Δ-Y.
However, one skilled in the art will recognize that other transformer configurations, such as Δ-Δ, Y-Δ, Y-Y are possible.
[54] Fig. 16 shows an exemplary pulse width modulation (PWM) switching sequence
100 that may be used to control the output voltage 96 of the DC-DC converter 70 of Fig. 15 and the output voltage 36 of the DC-DC converter 10 of Fig. 1. Six gate pulses, gl-g6, are phase-shifted by 60 degrees with the order from gl to g6, and each gate pulse 102 has the same pulse width T on , which is between T sw /6 and T sw /2 while avoiding shoot-through. [55] The phase-shifted six-step or PWM switching schemes described above, or combination of both, may be used to control the output DC voltage 36 of the three- phase full-bridge DC-DC converter 10 of Fig. 1. The PWM switching schemes described above, or a combination of PWM and phase-shifted operation, may be used to control the output DC voltage 96 of the three-phase full-bridge DC-DC converter 70 of Fig. 15. Having a variety of different switching schemes allows a degree of freedom in DC-DC power conversion designs to optimize the design depending on the required
specifications.
[56] The DC-DC converter described herein advantageously allows for a smaller secondary side DC inductor, lower total volume of high-frequency isolation transformers, lower duty cycle loss, and utilization of standard six-pack power semi¬ conductor devices when compared to conventional single phase and/or single three- phase full bridge converters.
[57] Therefore, in accordance with one embodiment of the present invention, a voltage converter includes a first three-phase full-bridge inverter having an input and an output and a second three-phase full-bridge inverter having an input and an output. The input of the first three-phase full-bridge inverter is connected in parallel with the input of the second three-phase full-bridge inverter.
[58] In accordance with another embodiment of the present invention, a voltage converter includes a first transformer having a primary set of windings and a first voltage inverter connected to the primary set of windings of the first transformer. A second transformer is included that has a primary set of windings, and a second voltage inverter is connected to the primary set of windings of the second transformer. The primary set of windings of the first transformer and the first voltage inverter are in parallel with the primary set of windings of the second transformer and the second voltage inverter.
[59] In accordance with yet another embodiment of the present invention, a method of supplying conditional power to a load includes the step of receiving a DC input voltage. The method includes the steps of converting the DC input voltage into a first three-phased AC voltage and converting the DC input voltage into a second three- phased AC voltage. The first three-phased AC voltage is supplied to a first transformer and the second three-phased AC voltage is supplied to a second transformer in parallel with the first transformer.
[60] The present invention has been described in terms of the preferred embodiment, and it is recognized that equivalents, alternatives, and modifications, aside from those expressly stated, are possible and within the scope of the appending claims.