WO1998012817A1 - Commutateur haute tension a semi-conducteur et alimentation a decoupage - Google Patents

Commutateur haute tension a semi-conducteur et alimentation a decoupage Download PDF

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Publication number
WO1998012817A1
WO1998012817A1 PCT/US1996/015214 US9615214W WO9812817A1 WO 1998012817 A1 WO1998012817 A1 WO 1998012817A1 US 9615214 W US9615214 W US 9615214W WO 9812817 A1 WO9812817 A1 WO 9812817A1
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WO
WIPO (PCT)
Prior art keywords
current
stage
state
component
terminal
Prior art date
Application number
PCT/US1996/015214
Other languages
English (en)
Inventor
James L. Cooper
Mark Adams
Original Assignee
Eldec Corporation
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Eldec Corporation filed Critical Eldec Corporation
Priority to PCT/US1996/015214 priority Critical patent/WO1998012817A1/fr
Priority to AU71646/96A priority patent/AU7164696A/en
Priority to EP96933091A priority patent/EP0927463A4/fr
Publication of WO1998012817A1 publication Critical patent/WO1998012817A1/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/10—Modifications for increasing the maximum permissible switched voltage
    • H03K17/102—Modifications for increasing the maximum permissible switched voltage in field-effect transistor switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/16—Modifications for eliminating interference voltages or currents
    • H03K17/161—Modifications for eliminating interference voltages or currents in field-effect transistor switches
    • H03K17/165—Modifications for eliminating interference voltages or currents in field-effect transistor switches by feedback from the output circuit to the control circuit
    • H03K17/166—Soft switching

Definitions

  • the present invention relates generally to high voltage power supplies, and more particularly to a power supply having a solid-state switch for switching a high voltage or a floating high voltage in a controlled manner,
  • Bipolar electrostatic chucks are used to hold semiconductor wafers during the manufacturing process.
  • DC direct current
  • electrostatic charge is applied to the wafer.
  • the charges on the wafer and the chuck electrodes produce an electrostatic attractive force that presses the wafer against the upper face of the chuck.
  • the chucking voltages are set to a high enough value to produce an electrostatic force between the wafer and the chuck that is adequate to prevent wafer movement during subsequent process steps.
  • the chucking voltage may range between ⁇ 3 00 and ⁇ 3 000 volts.
  • a recognized problem with the above-described method of removing a wafer from an electrostatic chuck is that repetitively charging .and grounding the chuck electrodes causes the electrodes to retain some charge. If any electrostatic attractive force remains between the wafer and the chuck due to the retained charge, excessive force may be required to remove the wafer. The force can crack the wafer or cause the wafer to pop off the chuck into a position from which it is difficult to retrieve and align properly in an automatic assembly line.
  • Several different techniques have been suggested to solve the wafer sticking problem. In one technique, the polarity of the chucking voltages placed on the chucking electrodes is periodically reversed.
  • the polarity of each electrode is switched so that the first electrode is operated at -3000 volts and the second electrode operated at +3000 volts.
  • Periodically changing the polarity of the chucking voltage has been found to remove the tendency of the chuck electrodes to retain an electrostatic charge.
  • FIGURE 1 depicts a representative circuit of a switching power supply 8 that may be used to reverse the polarity of a chucking voltage applied to one of the chuck electrodes.
  • a first high voltage DC generator 10 produces a high voltage on an output line 12, for example 3000 volts.
  • a second high voltage DC generator 14 produces a high voltage of the opposite polarity on a load line 16, for example -3000 volts.
  • the load line is connected to one of the chuck electrodes for application of the chucking voltage.
  • two switches are provided in the power supply.
  • a first high voltage switch SWl is connected between the output of the first high voltage generator 10 and the load line 16.
  • a second high voltage switch SW2 is connected between the second high voltage generator 14 and ground.
  • switch SWl When switch SWl is closed, switch SW2 open, and the second high voltage generator turned off, the load line 16 is maintained at +3000 volts.
  • switch SWl When switch SWl is open, switch SW2 closed, and the second high voltage generator turned on, the load line 16 is maintained at -3000 volts.
  • Appropriately switching the two switches and the high voltage generators therefore allows the voltage on the load line to be switched between the two high voltage potentials, in this particular case causing a reversal in the output voltage polarity on one of the chuck electrodes.
  • a second power supply similar to that shown in FIGURE 1 is required to switch the polarity of the other chuck electrode.
  • the switching time of electromechanical relays is uncontrolled and often too fast to allow for compensation by other components within the switching power supply.
  • the high voltage provided on output line 12 must generally be maintained at an exact level for use in the electrostatic chuck.
  • switch SWl is closed very quickly, the first high voltage generator 10 is connected with the load line 16 causing the load placed on the generator to increase suddenly.
  • the high voltage generator cannot immediately compensate for the change in load, causing the output from the generator to droop.
  • the first high voltage generator will ultimately compensate for the added load, the droop in the output from the power supply is detrimental to the operation of the electrostatic chuck.
  • the droop in output voltage may result in dechucking and loss of a wafer or damage to the. chuck.
  • the present invention provides a solid-state switching circuit for switching high voltages in a controlled and linear manner in a high voltage power supply.
  • the switching circuit consists of multiple MOSFET transistor stages connected in cascade. One end of the switching circuit is connected to a first high voltage potential. The other end of the switching circuit is coupled to a second high voltage potential, which is lower than the first high voltage potential, (hereinafter referred to as the "low voltage potential").
  • a blocking diode is connected in parallel with each stage. There is therefore a continuous series of discrete blocking diodes between the high and low voltage potentials.
  • Each stage in the high voltage switching circuit can be biased on or off. When biased on, the stage provides a conductive path. When biased off, the stage acts as an open circuit up to the breakdown value of the blocking diode across each stage.
  • the stage coupled to the low voltage potential is a current gain stage, and includes a current sense resistor in the conductive path.
  • the stages coupled to the current gain stage do not contain a sense resistor, and will hereinafter be referred to as the component stages.
  • the current gain stage is coupled to a driver stage.
  • An enable signal provided by the driver stage biases the current gain stage so that the gain stage begins to conduct current.
  • the current flow through the gain stage causes the adjacent component stage to be biased on.
  • the current flow through the adjacent component stage then causes the next component stage to be biased on. The process continues with each component stage turning on after the stage below it turns on.
  • the blocking diodes in the component stages of the switching circuit will avalanche at a known voltage rating. As the component stages are biased on, the blocking diodes across the component stages that are still biased-off will enter into conduction when the total voltage drop across the diodes exceeds the sum of the diode avalanche voltage ratings. Current will therefore flow through the avalanching diodes of the biased-off stages, through the transistors of the biased-on stages, and through the sense resistor as the upper component stages continue to turn on one-by- one. Eventually, all of the component stages are turned on, creating a conductive path through the stages and completing the switching of the switching circuit.
  • a feedback circuit is provided in the current gain stage to closely control the current flowing through the stage and the sense resistor.
  • the feedback circuit changes the bias point of the transistor in the current gain stage to maintain the current flow at a desired rate
  • the rate with which the component stages are biased on depends on the current flow through the current gain stage. During the period it takes for each subsequent component stage to be biased on, the voltage across the switching circuit is reduced in a controlled manner. In this manner, the switching circuit ensures a linear and controlled switch between the high voltage potential and the low voltage potential.
  • the controlled switching between the high voltage potential and the low voltage potential allows the switching circuit to be incorporated in a high voltage switching power supply. Incorporating one or more switching circuits in the power supply allows the power supply to produce an output voltage that can be reversed in polarity, or smoothly transitioned between multiple voltage levels.
  • the component stages are biased by the enable signal that is applied to the current gain stage. Biasing all of the component stages with the enable signal minimizes the number of components within the switching circuit.
  • the current gain stage may be isolated from the driver stage by a transformer.
  • the transformer isolates the switching circuit from the system or power supply in which the switching circuit is incorporated, allowing the switching circuit to operate in a fully floating mode.
  • the number of component stages can be varied to change the voltage that is switched.
  • Each component stage can switch a voltage roughly equivalent to the avalanche voltage rating of the blocking diode across the stage.
  • the number of component stages may therefore be selected depending on the anticipated maximum voltage swing in the switching operation, allowing the switching circuit to be simply and easily configured to operate in different environments.
  • An advantage of the disclosed switching circuit is that it allows high voltages to be switched to a linear and controlled manner.
  • the controlled switching reduces the amount of stress placed on the components in the switching circuit, allowing the circuit to be constructed using MOSFET transistors.
  • the controlled switching also allows the circuit to be incorporated in certain applications, such as power supplies for electrostatic chucks, where a longer switching time allows for compensation by other system components. For example, voltage droop is eliminated in the power supply because the longer switching time provides the high voltage generators sufficient time to compensate for any additional load.
  • Another advantage of the switching circuit is that it uses solid-state components that are readily available and relatively inexpensive. Because solid-state components are used, the entire switching circuit can be constructed to displace a very small volume and be of minimal weight. Constructing the switching circuit with solid-state components also makes the circuit more reliable than using a traditional electromechanical relay.
  • FIGURE 1 is a block diagram of a representative prior art switching power supply having an output that is switched between two voltages;
  • FIGURE 2 is a schematic of a solid-state switching circuit of the present invention in a floating configuration, wherein a current gain stage containing a sense resistor is used to regulate the current flow through the circuit during switching;
  • FIGURE 3 is a graph of the voltage across the switching circuit and the current through the switching circuit versus time generated by a representative embodiment of the invention
  • FIGURE 4 is a schematic of a solid-state switching circuit of the present invention that is referenced to ground;
  • FIGURE 5 is a block diagram of a switching power supply in accordance with the present invention.
  • FIGURE 6 is a timing diagram of representative control signals applied to the switching power supply of FIGURE 5 to cause the polarity of the output voltage to switch.
  • FIGURE 2 depicts the preferred embodiment of a solid-state switching circuit 50 in accordance with the present invention.
  • Switching circuit 50 consists of a number of component stages 52a, 52b, . . . 52n connected in cascade with a current gain stage 54. As will be described in more detail below, the switching circuit switches between one of two states. In an "off state, the component stages 52a, 52b,
  • the current gain stage 54 is initially biased off so that there is no conductive path between a first high voltage terminal 62 (hereinafter referred to as the high voltage terminal) and a second high voltage terminal 64 (hereinafter referred to as the low voltage terminal).
  • a first high voltage terminal 62 hereinafter referred to as the high voltage terminal
  • a second high voltage terminal 64 hereinafter referred to as the low voltage terminal.
  • an enable signal is applied to the current gain stage 54.
  • the current gain stage 54 is biased on and provides a current path through the current gain stage. Current flow through the current gain stage biases on the adjacent component stage 52a, which then biases on component stage 52b, and so on.
  • a blocking diode DD is connected in parallel with each stage. Since the voltage drop across a biased-on stage is very low, the voltage applied across the string of blocking diodes DD for the biased-off stages is increased as the stages turn on sequentially starting with the stage adjacent to the current gain stage. Ultimately, the voltage drop across the string of blocking diodes of the stages that are not yet biased on causes the diodes lo operate in avalanche and provide a current path from the high voltage terminal, around the stages that remain biased off, and through the biased-on component stages and current gain stage to the low voltage terminal. The rate at which the component stages are biased on is controlled by the regulated current through the current gain stage as discussed in greater detail below.
  • each component stage 52a, 52b, . . . 52n is constructed with the same circuit elements.
  • a generic component stage 52n will therefore be discussed as representative of all the component stages.
  • Component stage 52n is constructed around a pair of transistors TR, which in the preferred embodiment of the circuit are a pair of MOSFETs connected in cascade.
  • a blocking diode DD is connected between the drain 66 of the first transistor TR and the source 68 of the second transistor TR.
  • a capacitor CD is connected in parallel with the blocking diode.
  • Each transistor TR is biased with an identical biasing circuit.
  • a resistor RG is connected across the gate and source of each transistor, and in parallel with a capacitor CG and a Zener diode ZG. Resistor RG and Zener diode ZG are selected to prevent the transistor from conducting due to leakage current during biased-off operation, to protect the transistor from gate-to-source stress during biased-on operation, and to provide the desired gate-to-source voltage to turn the transistor on when a conductive path is generated.
  • the gate of each transistor in the component stage is connected in series with a diode DB and a resistor RB. Diodes DB are selected to ensure that reverse current will not flow between the component stages. Resistors RB are sized to limit the current flow into the component stage when the switching circuit is biased on. In an actual embodiment of the switching circuit, where each component stage is used to switch approximately 1000 volts, the circuit elements for each component stage are as follows:
  • the current gain stage 54 is similar to the component stages in that it is constructed around a pair of transistors TRA and TRB, preferably both MOSFETs.
  • a sense impedance preferably a sense resistor RS, is connected between the source 70 of transistor TRA and the low voltage terminal 64 in the switching circuit conductive path.
  • the sense resistor is selected to have a peak power capability sufficient to conduct the desired current when the switching circuit is turned on.
  • a diode DD and a capacitor CD are connected between the source 70 of transistor TRA and the drain 72 of transistor
  • a diode DD and a capacitor CD are also connected in parallel with the sense resistor RS.
  • Diodes DD and capacitors CD serve the same functions as they do in the component stages, that is, they are selected to provide over-voltage protection for the circuit.
  • the circuit elements for the current gain stage are as follows: Component Part Number or Rating
  • a drive circuit 58 is connected to the switching circuit 50 to turn the switching circuit on.
  • the drive circuit 58 generates a square wave that is applied to the primary winding of a transformer 56.
  • the square wave has a fifty per cent duty cycle, although other periodic waveforms having different duty cycles may also be used.
  • the secondary winding of the transformer 56 is connected a full-bridge rectifier 60 comprised of diodes DI, D2, D3, and D4. The diodes rectify the output of the transformer 56 to create a DC voltage that acts as the enable signal to turn on the switching circuit.
  • the use of a transformer to connect the drive circuit to the switching circuit allows the isolated operation of the switching circuit without reference to a ground potential. While a transformer is preferred, those skilled in the art will recognize that other components could be used to isolate the switching circuit and allow for floating operation.
  • the enable signal performs two functions in the switching circuit.
  • the enable signal is applied to the diodes DB in the component stages as a biasing potential.
  • the enable signal provides sufficient potential to the gates of the component stage transistors TR so that they will become biased on when the gate-to-source turn-on voltage of each transistor is exceeded by a voltage across resistor RG. That is, each transistor TR will become biased on when the current flow through resistor RG causes a voltage drop across the resistor that exceeds the turn-on voltage of each transistor.
  • the enable signal is also applied to the current gain stage 54 to bias the stage on.
  • the enable signal is first applied to a capacitor Cl connected across the output of the rectifier 60, which filters the enable signal.
  • the enable signal is then applied to a voltage divider consisting of resistors RI and R2 connected in series across the output of the rectifier 60.
  • the connection between resistors RI and R2 is coupled to the gate of transistor TRA.
  • the rectified and filtered enable signal is stepped-down by the voltage divider and applied to the gate of transistor TRA to bias the transistor on.
  • the bias point of transistor TRA is governed by a feedback loop to ensure that a desired current flows through the sense resistor, controlling the switching rate of the switching circuit.
  • a Zener diode ZG sized to limit the voltage applied to transistor TRA is connected between the gate 74 and source 70 of the transistor.
  • a capacitor CG is also connected between the gate and source of the transistor in order to filter the biasing voltage applied to the transistor TRA.
  • the feedback loop to ensure constant current through the sense resistor is constructed with two transistors Ql and Q2, which are preferably bipolar junction (BJT) transistors.
  • Transistor Q2 is an npn transistor that is connected so that the bias point of the transistor is dependent on the voltage drop across the sense resistor RS.
  • the base 76 of transistor Q2 is connected to a first lead of the sense resistor through a resistor R4.
  • the emitter 78 of transistor Q2 is connected to the second lead of the sense resistor through a Zener diode Zl.
  • the collector 80 of transistor Q2 is connected to the gate 74 of transistor TRA through a resistor R3.
  • a capacitor C2 is connected between the base and collector of the transistor Q2. In this configuration, the current flow through transistor Q2 will change depending upon the voltage drop across the sense resistor RS.
  • Transistor Ql is a pnp transistor that is controlled by the current flowing through transistor Q2.
  • the base 82 of transistor Ql is connected to the collector 80 of transistor Q2, and also to the gate 74 of transistor TRA through a resistor R3.
  • the collector 84 of transistor Ql is connected to the first lead of the sense resistor RS.
  • the emitter 86 of the transistor Ql is connected to the gate 74 of transistor TRA. Based on the bias point, transistor Ql shunts current away from the gate of transistor TRA.
  • the circuit elements are as follows:
  • Zener Diode Zl 1N4625 The feedback circuit controls the current flowing through transistor TRA to maintain the current at a desired level. Load current flowing through the sense resistor RS generates a voltage drop that changes the bias points of transistors Q2 and Ql. If the current through the sense resistor increases, transistor Ql will shunt additional current away from the gate of transistor TRA to reduce the current flowing through the transistor. If the current through the sense resistor decreases, transistor Ql will shunt less current away from the gate of transistor TRA to increase current through the transistor. In this manner, the feedback circuit ensures that the current flow through the sense resistor RS is maintained at a desired level.
  • component stages 52a, 52b, . . . 52n are biased off so that the transistors are nonconducting.
  • the current gain stage 54 is biased off, ensuring that no current flows through the sense resistor RS.
  • an enable signal is generated by the drive circuit 58 and applied to the current gain stage 54 through the isolation transformer 56 and rectifier 60.
  • the enable signal biases transistor TRA on.
  • the enable signal causes a biasing potential to be applied to the gates of transistors TR through diodes DB and resistor RB.
  • the voltage differential between the high and low voltage terminals is applied across fewer and fewer component stages whose transistors are still biased off.
  • the applied voltage is sufficiently great to force the blocking diodes of the biased-off stages into avalanche, allowing current to flow from the high voltage terminal 62 through the avalanching diodes to the component stages that are conducting.
  • the current then flows through the remaining biased-on transistors in each component stage, through the current gain stage transistors TRA and TRB, sense resistor RS, and finally to the low voltage terminal 64.
  • the component stages continue to turn on sequentially.
  • the output voltage at the low voltage terminal 64 increases by an amount roughly equivalent to the avalanche voltage rating of the blocking diode across the component stage.
  • the rate that the voltage increases is limited, however, by the rate of current flow through the current gain stage.
  • the transistors in all of the component stages have been turned on and a conductive path is present from the high voltage terminal to the low voltage terminal. When this occurs, the potential at the low voltage terminal will have been raised to approximately the potential of the high voltage terminal.
  • the result is that over a short period of time, and in a controlled manner, the low voltage terminal is connected to the high voltage terminal.
  • the switching rate is dependent on the current flow through the sense resistor. The switching rate is also dependent on any capacitive component contained within a load connected to the switching circuit.
  • FIGURE 3 is a representative graph of the voltage drop between the high and low voltage terminals 62 and 64, and the current through the switching circuit during a typical switching operation.
  • Line 100 represents the voltage between the high and low voltage terminals
  • line 102 is the current through the sense resistor RS.
  • the enable signal is turned on, biasing the current gain stage on and causing current to begin to flow through the sense resistor.
  • the first component stage begins to enter conduction.
  • the voltage across the biased-on stage changes from a value close to the breakdown voltage of the diode to approximately no voltage when the stage is conducting. The voltage change is linear and controlled, due to the current regulation provided by the current gain stage.
  • each component stage contributes to switching a voltage potential equal to the maximum avalanche voltage of the blocking diode for that stage.
  • the blocking diodes DD are selected to avalanche before transistors TR. The diode rating of each component stage 52a, 52b, . .
  • the current gain stage 54 is therefore used to determine the number of component stages necessary to switch a particular voltage. For example, if the switching circuit were to switch 6,000 volts, and if blocking diodes rated at 1,000 volts were used in the switching circuit, a total of five component stages would be required in the switching circuit. The total avalanche voltage of the five blocking diodes in the component stages and the single blocking diode in the current gain stage would add to a number approximating the required switching voltage of 6,000 volts. It will be appreciated that a greater or lesser number of component stages could be used to select the switching voltage of a switching circuit. Moreover, diodes having different ratings may also be selected to change the switching voltage capability. In most cases, however, the sum of the diode avalanche voltages must exceed the total voltage to be switched to prevent the blocking diode chain from conducting prior to application of the enable signal.
  • the switching circuit 50 depicted in FIGURE 2 is designed so that it may operate in a floating mode, that is, without reference to ground. In other applications it may be desirable to connect the switching circuit to ground.
  • a switching circuit 150 that has been modified so that it is referenced to ground is depicted in FIGURE 4.
  • the component stages 152a, 152b, . . . 152n in switching circuit 150 are identical in construction to those shown in switching circuit 50. Moreover, the component stages operate in the same manner as in switching circuit 50, turning on sequentially starting with the stage closest to a current gain stage 154 and continuing until all of the component stages have entered conduction. The difference between switching circuit 50, which is floating, and switching circuit 150, which is referenced to ground, is therefore found in the construction of the current gain stage 154.
  • the current gain stage 154 in switching circuit 150 is constructed around a pair of transistors TRA and TRB, preferably both MOSFETs.
  • a sense impedance preferably a sense resistor RS, is connected between the source 70 of tr.ansistor TRA and an output of an operational amplifier Ul.
  • the sense resistor is selected to have a peak power capability sufficient to conduct the desired current when the switching circuit is turned on.
  • a Zener diode Z2 and a capacitor C4 are connected in parallel between the source 70 of transistor TRA and ground. The Zener diode and capacitor operate to prevent current surges from damaging the operational amplifier Ul.
  • Transistor TRA is biased on by an enable signal that is provided by an enable circuit 156 connected to the gate 74 of the transistor.
  • the enable circuit 156 consists of a transistor TD, preferably a MOSFET, having a source connected to ground, a drain connected to a DC power supply through a resistor R5, and a gate connected to an on/off input terminal.
  • a resistor R6 is connected between the drain and source of transistor TD.
  • the drain of transistor TD is also connected to the gate 74 of transistor TRA.
  • a signal received on the on off input terminal causes transistor TD to enter conduction, applying a biasing voltage to the gate of the transistor TRA.
  • the biasing voltage turns on the transistor TRA, allowing the transistor to conduct current through the sense resistor RS.
  • the amount of current flowing through sense resistor RS is determined by a feedback circuit containing the operational amplifier Ul.
  • Line 158 connects the diodes DB in the component stages 152a, 152b, . . . 152n to a power supply through a resistor R7, shown at the lower left of FIGURE 5.
  • the component stages are therefore floating, so that the transistors TR will enter into conduction when the gate- to-source potential across the resistor RG in each component stage exceeds the turn- on voltage of the each transistor.
  • the lead of resistor R7 that is connected to line 158 is also connected to the noninverting input of operational amplifier Ul through a voltage divider consisting of a resistor R8 and a resistor R9.
  • the inverting input of amplifier Ul is connected to the power supply through a resistor Rll.
  • the noninverting input of the operational amplifier is also connected to the output of the operational amplifier through the parallel combination of a resistor Rl l and a capacitor C3.
  • the bias current provided through resistor R7 will increase. Due to the voltage drop across the resistor R7, the noninverting input of operational amplifier Ul will be at a lower potential than the noninverting input. The difference in potentials on the inputs of the operational amplifier cause the output of the operational amplifier to generate a negative voltage.
  • the feedback circuit is designed so that when all of the component stages 152a, 152b, . . . 152n are biased on, the negative voltage produced at the output of operational amplifier Ul is sufficiently negative so that the flow of the bias current through the sense resistor RS generates zero volts at the source 70 of the transistor TRA.
  • the component of the voltage at the source 70 of the transistor TRA due to the bias current flowing through the sense resistor RS is zero when all of the component stages are turned on. Designing the feedback circuit in this manner ensures that any potential across the sense resistor RS during operation of the switching circuit is therefore directly attributable to the load current flowing through transistor TRA, and not due to the bias current flowing to bias the component stages.
  • the bias point of transistor TRA changes since the gate-to-source voltage of the transistor is changing with the load current.
  • the load current flowing through the sense resistor is referenced to ground through the operational amplifier Ul.
  • the current through the switching circuit is thereby regulated so that the change in potential across the switching circuit is made in a controlled and linear manner.
  • switching circuits 50 and 150 can be used in a switching power supply 200 as shown in FIGURE 5.
  • the switching power supply 200 is suitable for use in many applications, such as the electrostatic chuck application described with respect to FIGURE 1.
  • the switching circuit 50 depicted in FIGURE 2 connects a first high voltage generator 204 with a second high voltage generator 206.
  • the switching circuit 150 depicted in FIGURE 4 connects the second high voltage generator 206 to ground.
  • a control circuit 210 is connected to the first and second high voltages 204 and 206 to turn the generators on and off.
  • the control circuit is also connected to switching circuits 50 and 150 to turn the switches on and off.
  • the first high voltage generator 204 generates +3,000 volts and the second high voltage generator 206 generates -3,000 volts.
  • the output voltage on load line 208 is switched between the two voltages generated by the high voltage generators to change the polarity of the output voltage.
  • a timing diagram depicting the control signals generated by the control circuit 210 to cause a change in voltage polarity on the load line is provided in FIGURE 6.
  • the switching circuits are turned off (or nonconducting).
  • the first and second high voltage generators 204 and 206 are turned on.
  • switching circuit 150 is turned on so that it connects the second high voltage generator 206 to ground.
  • the output voltage on the load line 208 begins to approach the voltage generated by the second high voltage generator.
  • the voltage on the load line drops linearly until -3,000 volts is reached at time t 2 .
  • switching circuit 50 isolates the load line from the output line, allowing the output line to be maintained at +3,000 volts.
  • the second high voltage generator 206 and the switching circuit 150 are turned off. Simultaneously, switching circuit 50 is turned on, causing a conductive path to be created in a controlled manner between the first high voltage generator 204 and the load line 208. Because of the controlled switching of the switching circuit, the voltage on the load line rises linearly from -3,000 volts until +3,000 volts is reached at time t4. The polarity of the output voltage from the switching power supply is thereby reversed.
  • the high voltage generator 204 is able to compensate for a change in load and ensure that the potential on an output line 202 does not droop.
  • the voltage reversal is performed in approximately 100 milliseconds. Those skilled in the art will appreciate that other switching times may be selected by appropriate selection of the components within the switching circuits. To turn off the switching power supply 200, at a time t 5 the first high voltage generator 204 and the switching circuit 50 are turned off.
  • the switching circuit of the present invention is advantageous in that the switching occurs in a controlled and linear manner, improving the operation of power supplies that incorporate the switching circuit. Moreover, because the switching circuit is reliant upon solid-state transistors for switching, the overall size of the device is minimized. Constructing the switching circuit from solid-state devices also greatly improves the reliability of the device.

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  • Dc-Dc Converters (AREA)

Abstract

Circuit de commutation (50) composé d'étages de transistors (52a, 52b...52n) montés en cascade, connecté entre une borne haute tension (62) et une résistance de détection (RS) couplée à une seconde borne basse tension (64). L'étage (54) du transistor le plus proche de la résistance de détection (RS) est désigné comme étage de gain en courant (54), et tous les autres étages (52a, 52b, ...52n) sont désignés comme étages composants (52a, 52b, ..52n). Des diodes anti-retour (DD) sont connectées aux bornes de tous les étages (52a, 52b, ...52n et 54), de sorte qu'un chemin continu soit formé entre la borne haute tension (62) et la borne basse tension (64). Lorsqu'un signal de validation est reçu par l'étage de gain en courant (54), les étages de transistor (52a, 52b, ...52n et 54) deviennent conducteurs de manière séquentielle, à commencer par l'étage de gain en courant (54) et se poursuivant par les étages composants (52a, 52b, ...52n). Chaque fois qu'un étage de transistor (52a, 52b, ...52n) devient conducteur, la chute de tension résultante provoque un effet d'avalanche sur chaque diode anti-retour (DD) correspondante, permettant au courant de circuler. Tous les étages composants (52a, 52b, ..52n) finissent par être actionnés, ce qui crée un chemin conducteur à travers les étages (52a, 52b, ...52n) et permet d'assurer la commutation dans le circuit de commutation.
PCT/US1996/015214 1996-09-23 1996-09-23 Commutateur haute tension a semi-conducteur et alimentation a decoupage WO1998012817A1 (fr)

Priority Applications (3)

Application Number Priority Date Filing Date Title
PCT/US1996/015214 WO1998012817A1 (fr) 1996-09-23 1996-09-23 Commutateur haute tension a semi-conducteur et alimentation a decoupage
AU71646/96A AU7164696A (en) 1996-09-23 1996-09-23 Solid-state high voltage switch and switching power supply
EP96933091A EP0927463A4 (fr) 1996-09-23 1996-09-23 Commutateur haute tension a semi-conducteur et alimentation a decoupage

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
PCT/US1996/015214 WO1998012817A1 (fr) 1996-09-23 1996-09-23 Commutateur haute tension a semi-conducteur et alimentation a decoupage

Related Child Applications (1)

Application Number Title Priority Date Filing Date
US09/272,721 Continuation US6008549A (en) 1999-03-19 1999-03-19 Solid-state high voltage switch and switching power supply

Publications (1)

Publication Number Publication Date
WO1998012817A1 true WO1998012817A1 (fr) 1998-03-26

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Application Number Title Priority Date Filing Date
PCT/US1996/015214 WO1998012817A1 (fr) 1996-09-23 1996-09-23 Commutateur haute tension a semi-conducteur et alimentation a decoupage

Country Status (3)

Country Link
EP (1) EP0927463A4 (fr)
AU (1) AU7164696A (fr)
WO (1) WO1998012817A1 (fr)

Cited By (4)

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Publication number Priority date Publication date Assignee Title
GB2378065A (en) * 2001-06-15 2003-01-29 Marconi Applied Technologies High voltage switching apparatus
WO2006137822A2 (fr) 2004-05-04 2006-12-28 Stangenes Industries, Inc Alimentation pulsee haute tension utilisant des commutateurs a l'etat solide
EP3471269A1 (fr) * 2017-10-13 2019-04-17 Samsung SDI Co., Ltd. Circuit de commutation pour connecter et déconnecter un composant électrique entre un potentiel haute tension et un potentiel basse tension
EP4372992A1 (fr) * 2022-11-15 2024-05-22 NXP USA, Inc. Commutateur à transistor cascode

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US5027018A (en) * 1988-09-14 1991-06-25 Eastman Kodak Company High voltage electrophoresis apparatus
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US4370607A (en) * 1980-04-29 1983-01-25 Thomson-Csf Static high voltage switch usable in a switchable high voltage D.C. generator
US5027018A (en) * 1988-09-14 1991-06-25 Eastman Kodak Company High voltage electrophoresis apparatus
US5347166A (en) * 1993-02-01 1994-09-13 Electric Power Research Institute Apparatus and method for switching high power levels

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Cited By (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2378065A (en) * 2001-06-15 2003-01-29 Marconi Applied Technologies High voltage switching apparatus
GB2378065B (en) * 2001-06-15 2004-09-15 Marconi Applied Technologies High voltage switching apparatus
US7256637B2 (en) 2001-06-15 2007-08-14 E2V Technologies (Uk) Limited High voltage switching apparatus
WO2006137822A2 (fr) 2004-05-04 2006-12-28 Stangenes Industries, Inc Alimentation pulsee haute tension utilisant des commutateurs a l'etat solide
EP1766762A2 (fr) * 2004-05-04 2007-03-28 Stangenes Industries, Inc. Alimentation pulsee haute tension utilisant des commutateurs a l'etat solide
EP1766762A4 (fr) * 2004-05-04 2011-01-05 Stangenes Ind Inc Alimentation pulsee haute tension utilisant des commutateurs a l'etat solide
EP3471269A1 (fr) * 2017-10-13 2019-04-17 Samsung SDI Co., Ltd. Circuit de commutation pour connecter et déconnecter un composant électrique entre un potentiel haute tension et un potentiel basse tension
EP4372992A1 (fr) * 2022-11-15 2024-05-22 NXP USA, Inc. Commutateur à transistor cascode

Also Published As

Publication number Publication date
EP0927463A1 (fr) 1999-07-07
EP0927463A4 (fr) 2005-10-19
AU7164696A (en) 1998-04-14

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