US9362995B2 - Transmitter apparatus, receiver apparatus, communication system, communication method, and integrated circuit - Google Patents

Transmitter apparatus, receiver apparatus, communication system, communication method, and integrated circuit Download PDF

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US9362995B2
US9362995B2 US14/002,142 US201214002142A US9362995B2 US 9362995 B2 US9362995 B2 US 9362995B2 US 201214002142 A US201214002142 A US 201214002142A US 9362995 B2 US9362995 B2 US 9362995B2
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signal
dmrs
channel
mobile
perturbation vector
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US20130336282A1 (en
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Hiroshi Nakano
Hiromichi Tomeba
Takashi Onodera
Alvaro Ruiz Delgado
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Sharp Corp
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Sharp Corp
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H04B7/0452—Multi-user MIMO systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H04B7/0456—Selection of precoding matrices or codebooks, e.g. using matrices antenna weighting
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0204—Channel estimation of multiple channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0224—Channel estimation using sounding signals
    • H04L25/0226—Channel estimation using sounding signals sounding signals per se
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/024—Channel estimation channel estimation algorithms
    • H04L25/0242—Channel estimation channel estimation algorithms using matrix methods
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/06—Dc level restoring means; Bias distortion correction ; Decision circuits providing symbol by symbol detection
    • H04L25/067—Dc level restoring means; Bias distortion correction ; Decision circuits providing symbol by symbol detection providing soft decisions, i.e. decisions together with an estimate of reliability
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/003—Arrangements for allocating sub-channels of the transmission path
    • H04L5/0048—Allocation of pilot signals, i.e. of signals known to the receiver
    • H04L5/005—Allocation of pilot signals, i.e. of signals known to the receiver of common pilots, i.e. pilots destined for multiple users or terminals
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H04B7/0456—Selection of precoding matrices or codebooks, e.g. using matrices antenna weighting
    • H04B7/046—Selection of precoding matrices or codebooks, e.g. using matrices antenna weighting taking physical layer constraints into account
    • H04B7/0465—Selection of precoding matrices or codebooks, e.g. using matrices antenna weighting taking physical layer constraints into account taking power constraints at power amplifier or emission constraints, e.g. constant modulus, into account
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J11/00—Orthogonal multiplex systems, e.g. using WALSH codes
    • H04J11/0023—Interference mitigation or co-ordination
    • H04J11/0026—Interference mitigation or co-ordination of multi-user interference
    • H04J11/003—Interference mitigation or co-ordination of multi-user interference at the transmitter

Definitions

  • the present invention relates to a transmitter apparatus, a receiver apparatus, a communication system, a communication method, and an integrated circuit.
  • MIMO Multi-Input Multi-Output
  • MIMO includes Single-User MIMO (SU-MIMO) in which a base-station (BS) apparatus transmits a plurality of signals to a single mobile-station (MS) device at the same timing and on the same frequency, and Multi-User MIMO (MU-MIMO) in which a base station transmits a signal to different mobile-station devices at the same timing and on the same frequency.
  • SU-MIMO Single-User MIMO
  • MS mobile-station
  • MU-MIMO Multi-User MIMO
  • SU-MIMO Since SU-MIMO is unable to multiplex streams more than the number of antennas of a mobile-station device, the maximum number of streams is limited by the number of physical antennas of the mobile-station device. On the other hand, since a base-station apparatus is able to have antennas more than the number of antennas of the mobile-station device, MU-MIMO becomes necessary in order to make the most use of idling antennas of the base-station apparatus.
  • DL down-link
  • LP linear precoding
  • a base-station apparatus orthogonalizes transmit signals by performing a multiplexing operation on a linear filter, and thus removes Multi-User Interference (MUI) between mobile-station devices. This reduces a flexibility of combinations of mobile-station devices that can be spatial multiplexed.
  • MUI Multi-User Interference
  • Nonlinear Precoding (NLP) MU-MIMO is disclosed as another method to implement spatial multiplexing.
  • NLP MU-MIMO a mobile-station device performs a modulo operation to treat, as the same point points, points to which a received signal is shifted in parallel by an integer multiple of a constant width (modulo width) in directions of an in-phase channel (I-ch) and a quadrature channel (Q-ch).
  • the base-station apparatus can add to a modulation signal a signal of any integer multiple of the modulo width (perturbation vector), and reduces transmission power by appropriately selecting the perturbation vector and adding the selected perturbation vector to a signal addressed to each mobile-station device (see Non Patent Literature 2 below).
  • the mobile-station device performs the modulo operation on a received signal, and the base-station apparatus will have a freedom of adding to each modulation signal a signal of any integer multiple of the modulo width.
  • the signal that can be added is referred to as a perturbation vector.
  • VP (Vector Perturbation) MU-MIMO is a method of searching for a perturbation vector that increases power efficiency most, in view of channel states of all mobile-station devices that are spatial multiplexed.
  • the base-station apparatus has a large amount of calculation, but VP MU-MIMO is NLP MU-MIMO scheme that provides excellent characteristics with a full transmit diversity gain (see Non Patent Literature 2 below).
  • THP (Tomlinson-Harashima precoding) MU-MIMO is available and is a method different from VP MU-MIMO.
  • THP MU-MIMO calculates a perturbation vector that is to be successively added to the signal addressed to each mobile-station device, in view of user interference each mobile-station device has suffered.
  • the complexity of a transmission process of the base-station apparatus is low but not all mobile-station devices may obtain full transmit diversity (see Non Patent Literature 3 below).
  • LR-THP is a method of THP MU-MIMO with a process called lattice reduction (LR) added thereto.
  • LR-THP is a method that provides the full transmit diversity gain with an amount of calculation lower than the amount of calculation of VP MU-MIMO (see Non Patent Literature 3 below).
  • a base-station apparatus needs to transmit DMRS (DeModulation Reference Signal) to each mobile-station device.
  • DMRS Demodulation Reference Signal
  • the base-station apparatus performs the same non-linear precoding operation as the data signal on DMRS and then transmits the DMRS, the mobile-station device is unable to estimate channels.
  • the DMRS is a signal that the base-station apparatus uses to notify each mobile-station device in advance of an amplitude and phase of the data signal on which the base-station apparatus has performed the precoding operation through NLP MU-MIMO. If a non-linear precoding operation is performed on the DMRS, the base-station apparatus adds a perturbation vector on the DMRS (or performs the modulo operation on the DMRS), and then transmits the DMRS. The mobile-station device needs to perform the modulo operation on the DMRS as well. For this reason, the mobile-station device needs to know a modulo width needed in the modulo operation in advance.
  • the mobile-station device Since the modulo width is proportional to the amplitude of the modulation signal in a received signal, the mobile-station device needs to know a reception gain of the data signal (a complex gain of a channel) subsequent to the non-linear precoding operation. However, the mobile-station device is unable to know the reception gain (the complex gain of the channel) without estimating channels using the DMRS. More specifically, the mobile-station device is in the situation that “the mobile-station device is unable to acquire the reception gain without estimating the channels using the DMRS, while being unable to estimate the channels using DMRS without knowing that reception gain”. The above-described problem thus arises.
  • the DMRS is transmitted to each mobile-station device using orthogonal radio resources (regions divided in a time direction and in a frequency direction, and if different data signals and different reference signals are assigned to the regions, the regions do not mutually interfere with).
  • the mobile-station device is free from performing the modulo operation on the DMRS, and obtains the reception gain.
  • the radio resources dedicated to the DMRS are needed for the number of mobile-station devices.
  • the overhead involved in the insertion of the DMRS increases accordingly.
  • the present invention has been developed in view of the above problem, and it is an object of the present invention to provide a technique of estimating an amplitude and phase (a complex gain and a reception gain) of a data signal normally in the mobile-station device and minimizing an increase in the overhead involved in the insertion of the DMRS by spatial multiplexing DMRS in the NLP MU-MIMO system.
  • a transmitter apparatus to transmit to a plurality of receiver apparatuses a data signal at the same timing and on the same frequency.
  • the transmitter apparatus includes a transmitter unit that transmits a demodulation reference signal addressed to a first receiver apparatus and a demodulation reference signal addressed to a second receiver apparatus different from the first receiver apparatus at the same timing and on the same frequency.
  • the demodulation reference signals (DMRS) spatial multiplexed are transmitted to all mobile-station apparatuses using a single radio resource.
  • a plurality of demodulation reference signals are transmitted to all the mobile-station apparatuses while enough radio resources are reserved to arrange the data signal.
  • Each mobile-station apparatus (hereinafter referred to as a receiver apparatus) may perform a channel estimation operation using a plurality of demodulation reference signals.
  • the present invention helps reduce the insertion loss of the demodulation reference signal.
  • the transmitter apparatus preferably includes a non-linear precoder that adds to the demodulation reference signal a signal that is an integer multiple of a predetermined signal. Also, the transmitter apparatus may include a non-linear precoder that performs a non-linear precoding operation on the demodulation reference signal.
  • the transmitter unit transmits a plurality of data signals at the same timing and on the same frequency
  • the non-linear precoder performs the same non-linear precoding operation, as the non-linear precoding operation performed on the data signal, the demodulation reference signal.
  • the non-linear precoding operation is performed using the same filter as the filter used on the data signal on the same principle as the principle applied to the data signal.
  • the transmitter apparatus may include a DMRS corrector to correct the demodulation reference signal.
  • the DMRS corrector preferably includes a two-dimensional Euclidean algorithm unit that performs a two-dimensional Euclidean algorithm operation on the demodulation reference signal.
  • the two-dimensional Euclidean algorithm unit may include a difference vector calculator that subtracts from the first demodulation reference signal the second demodulation reference signal, a signal that results from rotating the second demodulation reference signal in phase by 90 degrees, a signal that results from rotating the second demodulation reference signal in phase by 180 degrees, and a signal that results from rotating the second demodulation reference signal in phase by 270 degrees.
  • the present invention relates to a receiver apparatus including a perturbation vector adder that adds to a demodulation reference signal a signal that is an integer multiple of a predetermined width.
  • the receiver apparatus preferably includes an interim channel estimator that estimates a channel in accordance with the demodulation reference signal with the signal added thereto.
  • the receiver may include a perturbation vector candidate selector that selects a plurality of different pieces of the signal and a perturbation vector estimator that selects one of the signals in accordance with a channel estimation result estimated by the interim channel estimator that estimates the channel using each of the signals.
  • the receiver apparatus preferably includes a demodulator that calculates a logarithmic likelihood ratio by soft-estimating a data signal of each of the channel estimate results corresponding to the plurality of different pieces of the signal, and a perturbation vector evaluation value calculator that calculates a variance of each logarithmic likelihood ratio.
  • the perturbation vector estimator selects the signal corresponding to the largest one of the variances.
  • the receiver apparatus preferably includes a two-dimensional Euclidean algorithm unit that applies the two-dimensional Euclidean algorithm to a plurality of demodulation reference signals to calculate an irreducible vector.
  • the receiver apparatus preferably includes a complex gain calculator that calculates a complex gain of a channel using the irreducible vector.
  • the present invention also relates to a communication system.
  • the communication system includes a transmitter apparatus that includes a non-linear precoder that adds to a demodulation reference signal a signal that is an integer multiple of a predetermined signal, and a transmitter unit that transmits the demodulation reference signal and another signal at the same timing and on the same frequency, and a receiver apparatus that includes a perturbation vector adder that adds to the demodulation reference signal a signal that is an integer multiple of a predetermined width.
  • the present invention relates to a communication method.
  • the communication method includes a transmission method that includes a step of adding to a demodulation reference signal a signal that is an integer multiple of a predetermined signal, and a step of transmitting the demodulation reference signal and another signal at the same timing and on the same frequency, and a reception method that includes a step of adding to the demodulation reference signal a signal that is an integer multiple of a predetermined width.
  • the present invention relates to an integrated circuit.
  • the integrated circuit includes a non-linear precoder that adds to a demodulation reference signal a signal that is an integer multiple of a predetermined signal, and a transmitter unit that transmits the demodulation reference signal and another signal at the same timing and on the same frequency.
  • the present invention relates to an integrated circuit.
  • the integrated circuit includes a perturbation vector adder that adds to a demodulation reference signal a signal that is an integer multiple of a predetermined width.
  • the present invention may include a program that causes elements to perform the communication method, and a computer-readable recording medium that has stored the program.
  • FIG. 1 illustrates a concept of a communication system of a first embodiment of the present invention.
  • FIG. 2 illustrates a sequence chart illustrating an example of an operation of the communication system of the present embodiment.
  • FIG. 3 is a function block diagram illustrating a general structure of a base-station apparatus of the present embodiment.
  • FIG. 4 is a schematic diagram illustrating a structure example of a specific signal of the present embodiment.
  • FIG. 5A is a schematic diagram illustrating a structure example of frames of the present embodiment.
  • FIG. 5B illustrates a first structure example of frames transmitted via first through N-th antennas a 101 - n.
  • FIG. 5C illustrates a second structure example of frames transmitted via the first through N-th antennas a 101 - n.
  • FIG. 6 is a function block diagram illustrating a structure example of a mobile-station device of the present embodiment.
  • FIG. 7 is a function block diagram illustrating a DMRS channel estimator in detail.
  • FIG. 8 illustrates examples of positions of two DMRSs (DMRS 1 and DMRS 2) indicated on a signal point plane.
  • FIG. 9 illustrates examples of positions on the signal point plane where an irreducible vector fails to match a signal point of QPSK.
  • FIG. 10 illustrates a configuration example of a two-dimensional Euclidean algorithm unit.
  • FIG. 11 illustrates a configuration example of a DMRS corrector.
  • FIG. 12 illustrates a DMRS channel estimator (a) and a complex gain calculator (b).
  • FIG. 13 illustrates a configuration example of a non-linear precoder.
  • FIG. 14 is a flowchart illustrating an operation of the non-linear precoder.
  • FIG. 15 is a function block diagram illustrating a configuration example of a base-station apparatus of a second embodiment of the present invention.
  • FIG. 1 illustrates a concept of a communication system 1 of a first embodiment of the present invention.
  • the base-station apparatus A 1 transmits a common reference signal (CRS).
  • CRS common reference signal
  • a reference signal as the CRS is stored by the base-station apparatus A 1 and the first through fourth mobile-station devices B 11 through B 14 .
  • Each of the first through fourth mobile-station devices B 11 through B 14 estimates a channel state based on the CRS transmitted by the base-station apparatus A 1 , and then notifies the base-station apparatus A 1 of channel state information based on the estimated channel state.
  • the base-station apparatus A 1 transmits DMRS and a data signal to the first through fourth mobile-station devices B 11 through B 14 .
  • the base-station apparatus A 1 performs a precoding operation on the DMRS and the data signal, and then transmits the DMRS and data signal subsequent to multiplication.
  • the first through fourth mobile-station devices B 11 through B 14 multiplexed estimate the channel state of an equivalent channel having a precoding operation treated as part thereof (hereinafter referred to as an equivalent channel) and acquires a data signal in accordance with equivalent channel state information indicating a channel state of the estimated equivalent channel.
  • FIG. 2 illustrates a sequence chart illustrating an example of an operation of the communication system of the present embodiment.
  • FIG. 2 illustrates the example of the operation of the communication system 1 of FIG. 1 .
  • Step S 101 The base-station apparatus A 1 transmits the CRS to the first through fourth mobile-station devices B 11 through B 14 .
  • the base-station apparatus A 1 then proceeds to step S 102 .
  • Step S 102 The first through fourth mobile-station devices B 11 through B 14 estimate the channel state in accordance with the CRS transmitted in step S 101 . Processing proceeds to step S 103 .
  • Step S 103 The first through fourth mobile-station devices B 11 through B 14 calculate the channel state information in accordance with the channel state estimated in step S 102 . Processing proceeds to step S 104 .
  • Step S 104 The first through fourth mobile-station devices B 11 through B 14 notify the base-station apparatus A 1 of the channel state information calculated in step S 103 . Processing proceeds to step S 105 .
  • Step S 105 The base-station apparatus A 1 calculates a filter to be used in the non-linear precoding operation in accordance with the channel state information transmitted in step S 104 .
  • the base-station apparatus A 1 performs the non-linear precoding operation on the generated DMRS and data signal using the filter, thereby generating DMRS and data signal. Processing proceeds to step S 106 .
  • Step S 106 The base-station apparatus A 1 transmits the signal of DMRS generated in step S 105 to the first through fourth mobile-station devices B 11 , B 12 , B 13 , and B 14 . Processing proceeds to step S 107 .
  • Step S 107 The first through fourth mobile-station devices B 11 through B 14 estimate the equivalent channel in accordance with the signal of DMRS transmitted in step S 106 . Processing proceeds to step S 108 .
  • Step S 108 The base-station apparatus A 1 transmits the data signal generated in step S 105 to each of the first through fourth mobile-station devices B 11 through B 14 . Processing proceeds to step S 109 .
  • Step S 109 The first through fourth mobile-station devices B 11 through B 14 detect and acquire the data signal in accordance with the equivalent channel state information indicating the channel state of the equivalent channel estimated in step S 108 .
  • the first through fourth mobile-station devices B 11 through B 14 are four-multiplexed.
  • N mobile-station devices i.e., a first mobile-station device to an N-th mobile-station device B 11 through B 1 N are multiplexed.
  • the mobile-station devices B 11 through BIN are collectively referred to as a mobile-station device B 1 n.
  • FIG. 3 is a function block diagram illustrating a general structure of the base-station apparatus A 1 of the present embodiment.
  • the base-station apparatus A 1 includes a first antenna a 101 - 1 through an N-th antenna a 101 -N, a first receiver a 102 - 1 through an N-th receiver a 102 -N, a first GI (Guard Interval) remover a 103 - 1 through an N-th GI remover a 103 -N, a first FFT (Fast Fourier Transform) unit a 104 - 1 through an N-th FFT unit a 104 -N, a channel state information acquisition unit a 105 , a filter calculator a 11 , a first encoder a 121 - 1 through an N-th encoder a 121 -N, a first modulator a 122 - 1 through an N-th modulator a 122 -N, a DMRS generator a 124
  • the base-station apparatus A 1 of FIG. 3 uses orthogonal frequency division multiplexing (OFDM) scheme for uplink and downlink, for example.
  • OFDM orthogonal frequency division multiplexing
  • the base-station apparatus A 1 may use time division multiplexing (TDM) scheme or frequency division multiplexing (FDM) on one of the uplink and the down link or on both the uplink and the down link.
  • the first through N-th receivers a 102 - n receive signals transmitted by each mobile-station device B 1 n (a signal of a carrier frequency) via the first through N-th antennas a 101 - n .
  • the signal includes the channel state information.
  • the first through N-th receivers a 102 - n down-convert the received signals, and A/D (analog/digital) convert the down-converted signals, thereby generating baseband digital signals.
  • the first through N-th receivers a 102 - n output the generated digital signals to the first through N-th GI removers a 103 - n.
  • the first through N-th GI removers a 103 - n remove GI from the digital signals input from the first through N-th receivers a 102 - n , and then output the signals with GI removed therefrom to the first through N-th FFT units a 104 - n.
  • the first through N-th FFT units a 104 - n perform FFT transform operations on the signals input from the first through N-th GI removers a 103 - n , thereby generating signals in the frequency domain.
  • the first through N-th FFT units a 104 - n output the generated signals in the frequency domain to the channel state information acquisition unit a 105 .
  • the channel state information acquisition unit a 105 demodulates the signals input from the first through N-th FFT units a 104 - n , and extracts the channel state information from the demodulated information.
  • the channel state information acquisition unit a 105 outputs the extracted channel state information to the filter calculator a 11 .
  • a signal, other than the signal of the channel state information, out of the signals output from the first through N-th FFT units a 104 - n is demodulated by a controller (not illustrated).
  • Control information of the demodulated information is used to control the base-station apparatus A 1 .
  • Data other than the control information of demodulated signal is transmitted to another base-station apparatus, a server apparatus, and the like.
  • the filter calculator a 11 calculates a filter for use in a non-linear precoding operation in accordance with the channel state information input from the channel state information acquisition unit a 105 . Details of a filter calculation process performed by the filter calculator a 11 are described below.
  • the filter calculator a 11 inputs the calculated filter to the non-linear precoder a 13 .
  • the first through N-th encoders a 121 - n error-correction encode the input information bit, and then output the encoded bits to the first through N-th modulators a 122 - n.
  • the first through N-th modulators a 122 - n modulate the encoded bits input from the first through N-th encoders a 121 - n , thereby generating the data signal addressed to the mobile-station devices B 1 n .
  • the first through N-th modulators a 122 - n output the generated data signals to the specific signal constructor a 125 .
  • the base-station apparatus A 1 determines a modulation scheme for use in each of the first through N-th modulators a 122 - n in accordance with the channel state information, and outputs modulation information indicating the determined modulation scheme to the first through N-th modulators a 122 - n .
  • the base-station apparatus A 1 notifies the first through N-th mobile-station devices B 1 n of the modulation information.
  • the DMRS generator a 124 generates DMRSs to be addressed to the first through N-th mobile-station devices B 1 n .
  • the DMRS generator a 124 outputs the generated DMRS to the specific signal constructor a 125 .
  • the specific signal constructor a 125 associates the data signals addressed to the first through N-th mobile-station devices B 1 n input from the first through N-th modulators a 122 - n with the DMRSs addressed to the first through N-th mobile-station devices B 1 n input from the DMRS generator a 124 .
  • Pieces of information associated by the specific signal constructor a 125 and addressed to the first through N-th mobile-station devices B 1 n are referred to as specific signals of the first through N-th mobile-station devices B 1 n .
  • the specific signal constructor a 125 outputs to the non-linear precoder a 13 the specific signals of the first through N-th mobile-station devices B 1 n which are generated through the association operation thereof.
  • the non-linear precoder a 13 performs a non-linear precoding operation on the specific signal (the data signal and the DMRS) of each of the first through N-th mobile-station devices Bin input from the specific signal constructor a 125 .
  • the non-linear precoding operation to be performed by the non-linear precoder a 13 is described in detail below.
  • the non-linear precoder a 13 outputs to the frame constructor a 142 the specific signals that have been non-linearly precoded.
  • the CRS generator a 141 generates the CRS including a known reference signal of the base-station apparatus A 1 and the first through N-th mobile-station devices Bin, and then outputs the generated CRS to the frame constructor a 142 .
  • the frame constructor a 142 maps the specific signal input from the non-linear precoder a 13 and the CRS input from the CRS generator a 141 .
  • the frame constructor a 142 may map the specific signal and the CRS to the same frame or to different frames.
  • the frame constructor a 142 may map only the CRS to a given frame, or may map the CRS and the specific signal to another frame.
  • the base-station apparatus A 1 maps the CRS and the specific signal to a frame in accordance with a predetermined mapping and that the first through N-th mobile-station devices Bin have the knowledge of the mapping in advance.
  • the frame constructor a 142 outputs, to the first through N-th IFFT units a 143 - n , a signal, out of the mapped signals, to be transmitted via the antenna a 101 -non a per frame unit basis.
  • the first through N-th IFFT units a 143 - n perform an IFFT transform operation on the signal input from the frame constructor a 142 , thereby generating a signal in the time domain.
  • the first through N-th IFFT units a 143 - n output the generated signals in the time domain to the first through N-th GI inserters a 144 - n.
  • the first through N-th GI inserters a 144 - n attach guard intervals to the signals input from the first through N-th IFFT units a 143 - n , and then output the signals with the guard intervals attached thereto to the first through N-th transmitters a 145 - n.
  • the first through N-th transmitters a 145 - n D/A convert the signals (baseband digital signals) input from the first through N-th GI inserters a 144 - n .
  • the first through N-th transmitters a 145 - n generate signals on a carrier frequency by up-converting the digital-to-analog converted signal.
  • the first through N-th transmitters a 145 - n transmit the generated signals via the first through N-th antennas a 101 - n.
  • FIG. 4 is a schematic diagram illustrating a structure example of the specific signal of the present embodiment.
  • the abscissa represents time.
  • FIG. 4 illustrates the specific signal output by the specific signal constructor a 125 .
  • FIG. 4 also illustrates the specific signals of the first through N-th mobile-station devices Bin to be transmitted on the same frequency aligned on the time axis.
  • the DMRSs addressed to the first through N-th mobile-station devices Bin are referred to as “DMRS-MSn”.
  • the specific signal is for each of the first through N-th mobile-station devices Bin and includes the DMRS and the data signal.
  • a signal labeled symbol S 11 indicates the DMRS addressed to the mobile-station device B 11 (DMRS-MS1).
  • FIG. 4 indicates that the DMRSs addressed to the first through N-th mobile-station devices Bin are output at the same time from the specific signal constructor a 125 .
  • FIG. 4 also indicates that the data signals addressed to the first through N-th mobile-station devices Bin are also output from the specific signal constructor a 125 at the same time.
  • the data signals assigned to the same time and the DMRSs assigned to the same time are spatial multiplexed by the non-linear precoder a 13 and then transmitted from the base-station apparatus A 1 at the same timing and on the same frequency (i.e., by the same subcarrier of the same OFDM symbol).
  • FIG. 4 also indicates that the DMRS and the data signal are output from the specific signal constructor a 125 at different timings.
  • the DMRSs addressed to the first through N-th mobile-station devices B 1 n are output at time t 1 and time t 2
  • the data signals addressed to the first through N-th mobile-station devices B 1 n are output at time t 3 through time t L .
  • the structure of the specific signal of FIG. 4 is an example only, and the present invention is not limited to this structure.
  • the data signal is output (at t 3 thereafter) after the DMRS addressed to the mobile-station device B 1 n is output.
  • the specific signal constructor a 125 may output the DMRS after outputting the data signal.
  • the specific signal constructor a 125 may output the data signal and the DMRS alternately on the time axis, or in another sequential order.
  • FIG. 5A is a schematic diagram illustrating a structure example of frames of the present embodiment.
  • FIG. 5A illustrates the structure of frames to which the frame constructor a 142 maps the signal.
  • FIG. 5A illustrates the structures of the frames aligned along the time axis and transmitted via the first through N-th antennas a 101 - n.
  • FIG. 5A the CRSs transmitted via the first through N-th antennas a 101 - n are labeled “CRS-Txn”.
  • FIG. 5A also illustrates that each of the frames of the first through N-th antennas a 101 - n includes a CRS and a specific signal (the non-linearly precoded specific signal including the data signal and the DMRS).
  • a signal labeled symbol S 21 indicates a CRS (CRS-Tx1) to be transmitted via the antenna a 101 - 1 .
  • a signal labeled symbol S 22 is a non-linearly precoded specific signal to be transmitted via the antenna a 101 - 1 .
  • FIG. 5A illustrates the CRSs that arranged at different time bands by the frame constructor a 142 and are transmitted via the antenna a 101 - n from the base-station apparatus A 1 .
  • the CRS to be transmitted via the antenna a 101 - 1 is transmitted at time t1
  • the CRS to be transmitted via the antenna a 101 - 2 is transmitted at time t 2 .
  • FIG. 5A also illustrates the CRS and the specific signal that are arranged at different time bands by the frame constructor a 142 .
  • the CRSs to be transmitted via the antenna a 101 - n are arranged in a time band for transmission of time t 1 through time t N
  • the specific signals are arranged in a time band at time t N+1 thereafter.
  • the specific signals to be transmitted via the first through N-th antennas a 101 - n are arranged in the same time band by the frame constructor a 142 .
  • FIG. 5A illustrates the frame structure for exemplary purposes only, and the present invention is not limited to this frame structure.
  • all CRSs to be transmitted via the antennas a 101 - n are arranged in a preceding time band (earlier than t N+1 ), and the specific signals are arranged in a subsequent time band (at t N+1 thereafter).
  • the frame constructor a 142 may arrange the specific signals in the preceding time band, and the CRSs in the subsequent time band.
  • the frame constructor a 142 may arrange the specific signal and the CRS alternately in time bands, or in another sequential order.
  • the frame constructor a 142 may arrange the CRS to be transmitted via the antenna a 101 - 1 in the time band subsequent to time t N+1 and the specific signal in the time band prior to time t N+1 .
  • the base-station apparatus A 1 transmits a frame having only the CRS arranged therewithin before starting the transmission of the specific signal.
  • FIG. 5B and FIG. 5C illustrate a structure example of the frames to be transmitted via the first through N-th antennas a 101 - n .
  • FIG. 4 and FIG. 5A illustrate the DMRS and data signal, or the CRS and specific signal (including the DMRS and data signal) arranged in the time direction.
  • the CRS, DMRS, and data signal may be arranged in a two-dimensional matrix along the time direction (t) and frequency direction (f).
  • FIG. 6 is a function block diagram illustrating a structure example of the mobile-station device B 1 n of the present embodiment.
  • the mobile-station device B 1 n includes an antenna b 101 , a receiver b 102 , a GI remover b 103 , an FFT unit b 104 , a signal separator b 105 , a CRS channel estimator b 107 , a DMRS channel estimator b 12 , a channel compensator b 106 , a modulo calculator b 109 , a demodulator b 110 , a decoder bill, a channel state information generator b 108 , an IFFT unit b 131 , a GI inserter b 132 , and a transmitter b 133 .
  • the receiver b 102 receives a signal transmitted by each mobile-station device B 1 n (a signal on a carrier frequency) via the antenna b 101 .
  • the receiver b 102 down-converts and A/D (analog/digital) converts the received signal, thereby generating a baseband digital signal.
  • the receiver b 102 outputs the generated digital signal to the GI remover b 103 .
  • the GI remover b 103 removes the GI from the digital signal input from the receiver b 102 , and outputs the signal without the GI to the FFT unit b 104 .
  • the FFT unit b 104 performs a fast Fourier transform operation on the signal input from the GI remover b 103 , thereby generating a signal in the frequency domain.
  • the FFT unit b 104 outputs the generated signal in the frequency domain to the signal separator b 105 .
  • the signal separator b 105 demaps the signal input from the FFT unit b 104 in accordance with mapping information from the base-station apparatus A 1 .
  • the signal separator b 105 outputs out of the demapped signal, a CRS to the CRS channel estimator b 107 and a DMRS to the DMRS channel estimator b 12 .
  • the signal separator b 105 outputs a data signal to the channel compensator b 106 .
  • the signal separator b 105 inputs the data signal to the DMRS channel estimator b 12 .
  • the CRS channel estimator b 107 estimates a channel state in accordance with the CRS input from the signal separator b 105 , and then outputs information indicating the estimated channel state to the channel state information generator b 108 .
  • the DMRS channel estimator b 12 estimates a channel state of an equivalent channel that includes as part thereof a filter to be used in the non-linear precoding operation.
  • the DMRS channel estimator b 12 is described in detail below.
  • the DMRS channel estimator b 12 outputs equivalent channel state information indicating the channel state of the estimated equivalent channel to the channel compensator b 106 .
  • the channel compensator b 106 performs channel compensation on the signal input from the signal separator b 105 using the equivalent channel state information input from the DMRS channel estimator b 12 .
  • the channel compensator b 106 outputs the channel compensated signal to the modulo calculator b 109 .
  • the modulo calculator b 109 performs a modulo operation on the signal input from the channel compensator b 106 in accordance with modulation information from the base-station apparatus A 1 .
  • Expression (1-1) indicates the modulo operation that is performed on a signal ⁇ with a modulo width ⁇ .
  • floor(x) represents a maximum integer that does not exceed x
  • Re[ ⁇ ] and Im[ ⁇ ] respectively represent a real part and an imaginary part of a complex number ⁇ .
  • i represents a unit of complex number.
  • ⁇ represents the modulo width.
  • ⁇ is preferably 2 ⁇ 2 1/2 times the mean amplitude of QPSK signal
  • ⁇ is preferably 8 ⁇ 10 1/2 times the 16QAM signal
  • ⁇ is preferably 16 ⁇ 42 1/2 times the 64QAM signal.
  • the modulo width may be different if the base-station apparatus and the mobile-station device use a common value.
  • the modulo calculator b 109 outputs a value as a modulo operation result to the demodulator b 110 .
  • the demodulator b 110 demodulates the signal input from the modulo calculator b 109 in accordance with a modulation scheme indicated by the modulation information from the base-station apparatus A 1 .
  • the demodulator b 110 outputs to the decoder bill the demodulated information (hard-determined encoded bit or soft-estimated value of the encoded bit).
  • the decoder bill acquires an information bit by decoding the information input from the demodulator b 110 , and then outputs the acquired information bit.
  • the channel state information generator b 108 generates the channel state information from the channel state input from the CRS channel estimator b 107 (this operation is referred to as a channel state information generation process).
  • Let h n1 , h n2 , . . . , h nN represent a channel state between the antenna a 101 - 1 and the mobile-station device B 1 n , a channel state between the antenna a 101 - 2 and the mobile-station device B 1 n , . . .
  • the channel state information generator b 108 does not necessarily have to notify the base-station apparatus A 1 of the row vector
  • the channel state information generator b 108 may normalize the row vector by norm
  • the channel state information generator b 108 may notify the base-station apparatus A 1 of a value approximated to be a predetermined value as the channel state information.
  • the channel state information generator b 108 modulates the generated channel state information, and outputs the modulated signal of the channel state information to the IFFT unit b 131 .
  • the IFFT unit b 131 performs an inverse Fourier transform operation on the signal input from the channel state information generator b 108 , thereby generating a signal in the time domain.
  • the IFFT unit b 131 outputs the generated signal in the time domain to the GI inserter b 132 .
  • the GI inserter b 132 attaches the guard interval to the signal input from the IFFT unit b 131 , and outputs the signal with the guard interval attached thereto to the transmitter b 133 .
  • the transmitter b 133 D/A converts the signal (a baseband digital signal) input from the GI inserter b 132 .
  • the transmitter b 133 up-converts the converted signal, thereby generating a signal on a carrier frequency.
  • the transmitter b 133 transmits the generated signal via the antenna b 101 .
  • the filter calculator a 11 of FIG. 3 constructs a channel matrix H from the channel state information input from the channel state information acquisition unit a 105 .
  • H represents is a matrix of N rows and N columns, and a component at a p-th row and a q-th column represents a complex gain of a channel between a p-th mobile-station device B 1 p and a q-th antenna a 101 - q of the base-station apparatus (each of p and q is an integer from 1 to N). More specifically, the channel state acquired from the channel state information from each mobile-station device B 1 p becomes a row vector, and the channel matrix H is generated by arranging a matrix having row vectors corresponding to all the mobile-station devices.
  • the base-station apparatus A 1 If the norm of the channel state information is normalized by the mobile-station device B 1 n , the base-station apparatus A 1 generates the channel matrix H using the channel state information from the mobile-station device B 1 n , that is, using a normalized channel state at each row.
  • sn represent each specific signal input from the specific signal constructor a 125
  • s represent a column vector having each components of all s1 through Sn.
  • the specific signal herein refers to the data signal and the DMRS generated by the DMRS generator a 124 .
  • the value of ⁇ is determined independently of the data signal, but it is necessary to use a DMRS signal having a value common to the base-station apparatus and the mobile-station device.
  • the non-linear precoder a 13 searches for a combination of N-dimensional integer column vectors Z1 and Z2 that minimizes the norm of a transmit signal multiplied by filter W. This operation may be expressed by the following expression.
  • Z 1 ,Z 2) argmin (z1,z2)
  • on the right side is represented by (Z1, Z2). Since each component of z1 and z2 can take any integer, it is difficult to search for all the combinations.
  • a search range of each component of z1 and z2 is thus limited to a predetermined range (for example, to an integer having the absolute value L BS or less: [ ⁇ L BS , ⁇ L BS +1, . . . , ⁇ 2, ⁇ 1, 0, 1, 2, . . . , L BS ⁇ 1, L BS ]).
  • a search operation is preferably performed on points as candidates centered on a signal point Ws, but a method other than this method may be used to search for a point that minimizes the norm.
  • z1 ⁇ +z2 ⁇ is referred to as a perturbation vector.
  • a signal x W(s+Z1 ⁇ +iZ2 ⁇ ) based on the calculated (Z1, Z2) is power-normalized.
  • the base-station apparatus A 1 needs to normalize total transmit power of data signals within a constant number of subcarriers and a constant number of OFDM symbols (referred to as a “power normalization unit”) in order to keep the transmit power constant.
  • the power normalization unit represents the entire frame unit of FIG. 5B and FIG. 5C .
  • Total power P x across the power normalization unit of the power of the data signals x calculated in the non-linear precoding operation is calculated.
  • the non-linear precoder a 13 multiplies the specific signal x (the data signal and DMRS) by the reciprocal of the power normalization coefficient g, and inputs the data signal subsequent to the multiplication to the frame constructor a 142 .
  • Each component of the signal x indicates a transmit signal to be transmitted by each of the first antenna a 101 - 1 through N-th antenna a 101 - 1 .
  • the DMRS channel estimator b 12 may estimate the channel, and the channel compensator b 106 may perform the compensation operation on the amplitude of the data signal by multiplying the amplitude by g. More specifically, a signal can correctly be detected by multiplying the received data signal by g.
  • the non-linear precoder a 13 of the present invention performs on the DMRS the same non-linear precoding operation as the non-linear precoding operation performed on the data signal, including the power normalization.
  • the base-station apparatus A 1 has performed the power normalization, and (2) the channel state changes from when the mobile-station device B 1 n feeds back the channel state information to when the base-station apparatus A 1 transmits the data signal, a reception gain of the data signal (a complex gain of an equivalent channel including the precoding) needs to be freshly estimated during a data signal reception using the DMRS that has undergone the same precoding operation as the precoding operation performed on the data signal.
  • FIG. 7 illustrates the DMRS channel estimator b 12 in detail.
  • the DMRS channel estimator b 12 includes a perturbation vector candidate selector b 121 , a perturbation vector adder b 122 , an interim channel estimating unit b 123 , a channel compensating unit b 124 , a modulo calculator b 125 , a demodulator b 126 , a perturbation vector evaluation value calculator b 127 , and a perturbation vector estimating unit b 128 .
  • Step S 21 The perturbation vector candidate selector b 121 selects a perturbation vector candidate, and input the selected perturbation vector to the perturbation vector adder b 122 .
  • the number of candidates of the perturbation vectors present and expressed in Expression (1-3) is (2L MS +1) 2N .
  • the base-station apparatus A 1 and the mobile-station device B 1 n use a common DMRS modulo width.
  • condition L BS L MS holds preferably, condition L BS >L MS may be acceptable to reduce an amount of calculation in the mobile-station device B 1 n.
  • L BS In THP or LR-THP to be described with reference to a third embodiment, it is difficult to determine L BS , and L MS may be determined depending on the amount of calculation of the mobile-station device B 1 n.
  • the perturbation vector adder b 122 inputs the signal q+Za to the interim channel estimating unit b 123 .
  • Step S 23 the interim channel estimating unit b 123 divides the received DMRSp by the signal q+Za.
  • the interim channel estimating unit b 123 inputs an amplitude
  • Step S 24 The channel compensating unit b 124 performs the channel compensation on the data signal in the frame having the DMRS therewithin using the amplitude and phase input from the interim channel estimating unit b 123 . More specifically, the channel compensating unit b 124 divides the received signal by the complex gain h. The channel compensating unit b 124 inputs the channel-compensated data signal to the modulo calculator b 125 .
  • Step S 25 The modulo calculator b 125 performs a modulo operation on the data signal channel-compensated by the channel compensating unit b 124 .
  • the modulo calculator b 125 uses the data signal modulo width common to the base-station apparatus A 1 and the mobile-station device B 1 n .
  • the modulo calculator b 125 inputs the modulo calculated data signal to the demodulator b 126 .
  • Step S 26 The demodulator b 126 soft-estimates the input modulo operated data signal, and then inputs the soft-estimated value to the perturbation vector evaluation value calculator b 127 .
  • LLR log likelihood ratio
  • y represents the modulo-operated data signal input from the modulo calculator b 125
  • s k represents each of the signal points of each modulation scheme.
  • s m + represents a signal point with a m-th bit thereof modulated through each modulation scheme and being +1
  • s m ⁇ represents a signal point with a m-th bit thereof modulated through each modulation scheme and being 0.
  • ⁇ n 2 represents a sum of variances of noises in I-ch and Q-ch of each modulation signal (i.e., complex Gaussian noise power).
  • the demodulator b 126 inputs, to the perturbation vector evaluation value calculator b 127 , LLR corresponding to each bit assigned to each data signal. More specifically, the demodulator b 126 calculates LLRs of the number equal to the number of data signals ⁇ 2 in the case of QPSK, and calculates LLRs of the number equal to the number of data signals ⁇ 4 in the case of 16QAM, and then inputs the resulting LLRs to the perturbation vector evaluation value calculator b 127 .
  • the perturbation vector evaluation value calculator b 127 calculates a variance of the input LLRs, and input the calculated variance of the LLRs to the perturbation vector estimating unit b 128 .
  • the variance of LLRs is calculated in accordance with the following Expression.
  • V represents the number of data signals soft estimated on each candidate of the perturbation vectors.
  • M represents a bit count assigned to each modulation scheme, and is 2 in QPSK, and 4 in 16 QAM.
  • L v m represents LLR at the m-th bit assigned to a v-th data signal.
  • Step S 28 The perturbation vector estimating unit b 128 selects the largest variance of the variances of LLRs corresponding to perturbation vectors input from the perturbation vector evaluation value calculator b 127 , outputs the amplitude and phase of the data signal (referred to as “equivalent channel state information”) corresponding to the perturbation vector having the largest variance, and thus inputs the equivalent channel state information to the channel compensator b 106 in the DMRS channel estimator b 12 ( FIG. 6 ).
  • a large variance of LLRs indicates that a mutual amount of information transmitted from the base-station apparatus A 1 is the largest when each perturbation vector is assumed.
  • the absolute value of LLR becomes larger as the “probability” that each bit is 1 or 0 is higher.
  • Each perturbation vector is assumed, and the variance of LLRs is calculated.
  • the perturbation vector having the highest “probability” is estimated by selecting a perturbation vector having the largest variance of the LLRs.
  • the channel compensating unit b 124 , the modulo calculator b 125 , and the demodulator b 126 in the DMRS channel estimator b 12 perform the same operations as those of the channel compensator b 106 , the modulo calculator b 109 , and the demodulator b 110 respectively external to the DMRS channel estimator b 12 , the same circuit design may be shared to reduce the circuit scale.
  • a soft-estimated value corresponding to the perturbation vector estimated by the perturbation vector estimating unit b 128 , out of the soft-estimated values calculated by the demodulator b 126 may be used by the decoder bill. This arrangement avoids calculating the soft-estimated value twice, leading to a reduction in the amount of calculation.
  • the base-station apparatus spatial multiplexes the DMRSs through NLP MU-MIMO and transmits the multiplexed DMRSs. Power efficiency is thus increased, and the overhead involved in the insertion of the DMRS is reduced.
  • the data signal that is soft-estimated to estimate the perturbation vector may be a data signal as part of a frame.
  • the perturbation vector may be estimated using an index other than the variance of the LLRs.
  • the mobile-station device B 1 n estimates the likeliest perturbation vector from the candidates of the perturbation vectors.
  • the DMRSs can be spatial multiplexed using NLP MU-MIMO.
  • two apparatuses including a base-station apparatus A 2 and a mobile-station device B 2 n perform a process referred to as “two-dimensional Euclidean algorithm”, and the mobile-station device B 2 n operates with an amount of calculation lower than the amount of calculation of the mobile-station device B 2 n in a first embodiment.
  • the base-station apparatus A 2 of the present embodiment is identical to the base-station apparatus A 1 of the first embodiment except that the base-station apparatus A 2 includes a DMRS corrector a 226 .
  • the mobile-station device B 2 n of the present embodiment is identical to the mobile-station device B 1 n of the first embodiment except that a DMRS channel estimator b 22 in the mobile-station device B 2 n is different in operation from the DMRS channel estimator b 12 .
  • FIG. 15 illustrates a configuration of the base-station apparatus A 2 . As described above, the DMRS corrector a 226 is newly added to the configuration of FIG. 3 .
  • the DMRS corrector a 226 in the base-station apparatus A 2 and the DMRS channel estimator b 22 in the mobile-station device B 2 n are described in detail.
  • Each of the base-station apparatus A 2 and mobile-station device B 2 n of the present embodiment includes a two-dimensional Euclidean algorithm unit that performs a “two-dimensional Euclidean algorithm process”.
  • the two-dimensional Euclidean algorithm unit constitutes one of the features of the present embodiment, and the principle thereof is described first.
  • the two-dimensional Euclidean algorithm is an algorithm to which a standard Euclidean algorithm is extended.
  • the standard Euclidean algorithm is herein referred to as a one-dimensional Euclidean algorithm.
  • the one-dimensional Euclidean algorithm is extended to a complex number.
  • the algorithm to which the one-dimensional Euclidean algorithm is extended is the two-dimensional Euclidean algorithm.
  • a method described herein is to calculate a “base vector (or an irreducible vector)” corresponding to the greatest common divisor of Gaussian integers if two complex numbers, each having an integer real part and an integer imaginary part (Gaussian integers), are present.
  • the two-dimensional Euclidean algorithm is applied to two Gaussian integers of (3+i) and ( ⁇ 1+i).
  • a Gaussian integer of the pair of (3+i) and ( ⁇ 1+i) having a larger norm (absolute value) is added to a Gaussian integer that results from multiplying a Gaussian integer having a smaller norm of the pair by +1, ⁇ 1, +i, and ⁇ i (the multiplication of the Gaussian integer by ⁇ 1, +i, and ⁇ i is interpreted to mean that the Gaussian integer is rotated by +180 degrees, +90 degrees, and +270 degrees, respectively).
  • the Gaussian integers having the smallest norm are 1+i and 1 ⁇ i. If two or more Gaussian integers have the same norm, whichever may be used. For example, 1+i is selected here, and the same operation is performed on ⁇ 1+i and 1+i.
  • the present embodiment features the DMRS channel estimation through the two-dimensional Euclidean algorithm.
  • the principle of the channel estimation is described below.
  • the DMRS serving as a reference in the present embodiment has to be one of the points of the QPSK signal.
  • FIG. 8 illustrates an example of the positions of two DMRSs (referred to as DMRS 1 and DMRS 2) in a signal point plane.
  • the base-station apparatus may now transmit the DMRS 1 at position 3+i, and the DMRS 2 at position ⁇ 1+i, for example.
  • the unit of the signal point plane is set so that the amplitude of the QPSK signal is 2 1/2 . More specifically, the signal points of the QPSK signal are ( ⁇ 1 ⁇ i), and the modulo width is 4 (for convenience of explanation, the unit of amplitude has changed from that of the first embodiment).
  • All of solid circles denoting the DMRS 1 and the DMRS 2 and blank circles other than the solid circles are QPSK signal points and points resulting from adding any perturbation vector to the QPSK signal points.
  • the DMRS 1 is a signal that results from adding a perturbation vector +4 to ⁇ 1+i of the QPSK signal.
  • a complex number 1+i as the “greatest common divisor” is obtained. This means that a minimum lattice vector (irreducible vector) of a lattice point group (solid circles and blank circles) of FIG. 8 is calculated.
  • the irreducible vector corresponds to one of the four QPSK points. If a common factor “a” to complex numbers is multiplied such as a(3+i) and a( ⁇ 1+i) in original signals, the irreducible vector is also multiplied by a, resulting in an irreducible vector a(1+i).
  • the DMRS 1 and the DMRS 2 are multiplied by a complex gain of the channel, namely, h(3+i) and h( ⁇ 1+i). Noise is disregarded herein.
  • the irreducible vector h(1+i) results. This signal is obtained by multiplying the complex gain of the channel by one of the four QPSK points to which no perturbation vector is added. Since it is known that each of the QPSK points has a norm of 2 1/2 , the absolute value
  • the mobile-station device has difficulty in determining which of the four QPSK points ( ⁇ 1 ⁇ i) multiplied by h results in the signal h(1+i) obtained through the two-dimensional Euclidean algorithm unit during the DMRS channel estimation.
  • the phase arg(h) is thus determined on a tentatively selected point (1+i). If the point multiplied by h is one of the other three points ( ⁇ 1+i), (1 ⁇ i), and ( ⁇ 1 ⁇ i), the actual phase difference of the complex gain is definitely one of +90 degrees, +180 degrees, and +270 degrees.
  • the modulo operation is then performed on the DMRS 1 and the DMRS 2.
  • the same modulo operations are performed on I-ch and Q-ch, and are symmetrical with respect to the corresponding axis, and are rotational symmetrical computation by 90 degrees. For this reason, if arg (h) is rotated by an integer multiple of 90 degrees, the modulo operation is performed on each of the DMRS 1 and the DMRS 2 without any problems.
  • the DMRS 1 subsequent to the modulo operation is h(1+i), and the DMRS 2 subsequent to the modulo operation is h( ⁇ 1+i). If noises ⁇ 1 and ⁇ 2 are considered, the DMRS 1 becomes h(1+i)+ ⁇ 1, and the DMRS 2 becomes h( ⁇ 1+i)+ ⁇ 2. Since the mobile-station device has known that the DMRS 1 is (1+i) and the DMRS 2 is ( ⁇ 1+i) prior to the non-precoding operation, the mobile-station device performs the channel estimation from each DMRS as below.
  • ⁇ 1/(1+i) and ⁇ 2/( ⁇ 1+i) are channel estimation errors, and can be reduced by maximum ratio combining the channel estimation results of the DMRS 1 and the DMRS 2.
  • h( ⁇ 1 ⁇ i) corresponding to the irreducible vector cannot be calculated through the two-dimensional Euclidean algorithm in a combination of some DMRSs. If the DMRSs are arranged as illustrated in FIG. 9 , the irreducible vector is 3+3i, and fails to match the signal point of QPSK. To avoid such a case, the base-station apparatus predicts the case of FIG. 9 through the two-dimensional Euclidean algorithm, and then changes the signal points and transmits points free from such a problem.
  • FIG. 10 illustrates the two-dimensional Euclidean algorithm.
  • the two-dimensional Euclidean algorithm unit 300 is present both in the base-station apparatus and the mobile-station device.
  • the two-dimensional Euclidean algorithm unit of the mobile-station device is described first.
  • the two-dimensional Euclidean algorithm unit 300 of FIG. 10 includes a vector storage unit 301 , a difference vector calculator 303 , a difference vector norm calculator 305 , a convergence determining unit 307 , and a norm calculator 309 .
  • Step S 31 The vector storage unit 301 inputs the two input DMRSs to the norm calculator 309 .
  • the two DMRSs are obtained by multiplying the DMRS transmitted from the base-station apparatus by the complex gain h of the channel, and adding noise to the DMRS subsequent to the multiplication.
  • the norm calculator 309 calculates the norms of the two input DMRSs, and input each norm to the vector storage unit 301 .
  • Step S 33 The vector storage unit 301 inputs the norms corresponding to the two input DMRSs to the difference vector calculator 303 .
  • Step S 34 Let “a” represent a vector having a larger norm out of the two input vectors and “b” represent a vector having a smaller norm (the signal obtained when the difference vector calculator 303 recursively performs a calculation on the two input DMRSs is hereinafter referred to as a “vector”).
  • the difference vector calculator 303 also inputs the norms of the vector b and the norm of the vector b to the vector storage unit 301 .
  • Step S 35 The difference vector norm calculator 305 calculates the norms of the four input vectors c1 through c4, inputs the vector having the smallest norm and the norm of that vector to the vector storage unit 301 , and inputs the norm of the vector having the smallest norm to the convergence determining unit 307 .
  • Step S 36 The convergence determining unit 307 compares in magnitude the input norm with a predetermined positive value T. If the norm is larger than T, the convergence determining unit 307 inputs to the vector storage unit 301 information that the two-dimensional Euclidean algorithm has been completed.
  • the norm of 0 is not set as a convergence condition as opposed in the principle of the two-dimensional Euclidean algorithm because the DMRS contains noise in the mobile-station device and the norm of the convergence determining unit 307 does not end up with 0.
  • the noise is typically smaller than the DMRS, and it is thus determined that convergence has reached when the norm becomes smaller than the predetermined constant T.
  • T is a tradeoff of the following two factors, and is preset through computer simulation.
  • the convergence determining unit 307 determines that the norm has not converged because of an error caused by noise, although the convergence determining unit 307 should determine that the norm has converged.
  • T is thus preset in view of the tradeoff of the two factors.
  • Step S 37 The vector storage unit 301 inputs to the difference vector calculator 303 two norms of
  • step S 34 2) norm corresponding to a vector input from the difference vector norm calculator 305 , and repeats the process beginning with step S 34 .
  • Step S 38 Upon receiving from the convergence determining unit 307 information indicating that the two-dimensional Euclidean algorithm has been completed, the vector storage unit 301 outputs the vector b (irreducible vector).
  • FIG. 11 illustrates a configuration example of the DMRS corrector a 226 .
  • the DMRS corrector a 226 includes a two-dimensional Euclidean algorithm unit 300 , an irreducible vector verification unit 320 , and a perturbation vector adder 340 .
  • the DMRS corrector a 226 causes the base-station apparatus A 2 to solve the problem that the irreducible vector fails to match the lattice vector, thereby precluding the problem.
  • the DMRS corrector a 226 acquires the DMRS with the perturbation vector added thereto by the non-linear precoder a 13 .
  • the DMRS acquired by the DMRS corrector a 226 has the perturbation vector added thereto.
  • a signal prior to adding a filter W is (q+z1 ⁇ +iz2 ⁇ ).
  • the DMRSs acquired at a time is two, and the two are paired in advance. Which is paired with which is known by the base-station apparatus A 2 and the mobile-station device B 2 n in advance.
  • the two-dimensional Euclidean algorithm unit 300 inputs to the irreducible vector verification unit 320 an irreducible vector resulting from applying the two-dimensional Euclidean algorithm to the two DMRSs (the irreducible vector is referred to “d”) and the two DMRSs acquired from the non-linear precoder a 13 .
  • the irreducible vector verification unit 320 again inputs the two DMRSs acquired from the non-linear precoder a 13 to the non-linear precoder a 13 .
  • the irreducible vector verification unit 320 inputs the two DMRSs acquired from the non-linear precoder a 13 to the perturbation vector adder 340 .
  • the perturbation vector adder 340 adds the perturbation vector ⁇ ( ⁇ , i ⁇ , or ⁇ i ⁇ ) to one of the two DMRSs acquired from the non-linear precoder a 13 .
  • the perturbation vector adder 340 determines which of the DMRSs the perturbation vector ⁇ is to be added to, in a random fashion using a random number.
  • the perturbation vector adder 340 again inputs to the two-dimensional Euclidean algorithm unit 300 the two DMRSs including the DMRS with the perturbation vector added thereto.
  • the two-dimensional Euclidean algorithm unit 300 performs the two-dimensional Euclidean algorithm using new DMRSs. From then on, the two-dimensional Euclidean algorithm unit 300 , the irreducible vector verification unit 320 , and the perturbation vector adder 340 continuously perform the operations thereof until the irreducible vector verification unit 320 determines that the irreducible vector d is the DMRS prior to the perturbation vector addition, i.e., one of the QPSK signals ( ⁇ 1 ⁇ i).
  • the irreducible vector verification unit 320 outputs and then inputs the two corrected DMRSs to the non-linear precoder a 13 .
  • the non-linear precoder a 13 multiplies the DMRS by a filter W without searching for a new perturbation vector, and multiplies the resulting value by the reciprocal g of the power normalization coefficient g.
  • FIG. 12 ( ⁇ ) illustrates the DMRS channel estimator b 12 in detail.
  • the DMRS channel estimator b 12 includes a two-dimensional Euclidean algorithm unit 300 and a complex gain calculator 350 .
  • the complex gain calculator 350 includes an interim complex gain calculator 351 , a DMRS channel compensator 352 , a DMRS-modulo unit 353 , and a vector divider 354 .
  • the DMRS channel estimator b 12 first acquires the two DMRSs received by the mobile-station device B 2 n .
  • the two DMRSs are a pair of DMRSs (referred to as p 1 and p 2 ) discussed with reference to the DMRS corrector a 226 of the base-station apparatus.
  • the two-dimensional Euclidean algorithm unit 300 performs the two-dimensional Euclidean algorithm on the two DMRSs, thereby calculating an irreducible vector p red .
  • the irreducible vector p red is an irreducible vector already multiplied by the complex gain h, and is thus different from the irreducible vector d calculated by the DMRS corrector a 226 .
  • the irreducible vector p red is a signal that has been obtained by multiplying one of the four QPSK signals with no perturbation vector added thereto by the complex gain h of the channel and by adding an error caused by noise to the multiplied signal.
  • ( pred/2 1/2 ) of the complex gain of the channel can be determined.
  • the interim complex gain calculator 351 inputs the absolute value
  • the DMRS channel compensator 352 performs a channel compensation on the DMRSs p 1 and p 2 using the input absolute value
  • q 1 p 1 /
  • q 2 p 2 /
  • the DMRS channel compensator 352 inputs to the DMRS-modulo unit 353 the channel-compensated DMRSs q 1 and q 2 , the absolute value
  • ⁇ e 1 ⁇ exp (2 ⁇ i ⁇ )) and a signal p mod2 (
  • the DMRS-modulo unit 353 inputs the signals p mod1 and p mod2 to the vector divider 354 .
  • the vector divider 354 inputs the complex gain h of the estimated channel to the channel compensator b 106 .
  • each of the base-station apparatus A 2 and the mobile-station device B 2 n performs the process of the two-dimensional Euclidean algorithm, and the DMRSs can be spatial multiplexed through NLP MU-MIMO using a relatively small amount of calculation on the mobile-station device B 2 n .
  • the overhead involved in the insertion of the DMRS is reduced.
  • Each of the mobile-station device B 1 n of the first embodiment and the mobile-station device B 2 n of the second embodiment performs VP with the non-linear precoder a 13 and the filter calculator a 11 .
  • a third embodiment uses THP that involves a smaller amount of calculation than VP.
  • a base-station apparatus is referred to as a base-station A 3
  • mobile-station devices are referred to as mobile-station devices B 31 through B 3 N
  • any of the mobile-station devices is referred to as a mobile-station device B 3 n.
  • a non-linear precoder is referred to as a non-linear precoder a 33
  • a filter calculator is referred to as a filter calculator a 31 .
  • the filter calculator a 31 first QR decomposes H after calculating the channel matrix H in the same manner as in the first embodiment.
  • H H QR (3-1)
  • R is an upper triangular matrix
  • Q is a unitary matrix
  • H means a complex conjugate transpose of a matrix.
  • A represent a diagonal matrix including only diagonal components of R H .
  • an interference coefficient filter F is calculated using A and R in accordance with the following Expression.
  • F R H A ⁇ 1 ⁇ I (3-3)
  • the filter calculator a 31 inputs the linear filter P 0 and the interference coefficient filter F to the non-linear precoder a 33 .
  • P s represent the mean power of the modulation signal
  • P v represent the mean power of the data signal subsequent to the modulo operation
  • C v is a diagonal matrix having the diagonal components [ P s ,P v ,P v , . . . ,P v ] (3-5) in the order from the top-left corner of the matrix.
  • the filter calculator a 31 inputs to the non-linear precoder a 33 as a new linear filter P a matrix that results from multiplying the linear filter P 0 calculated in Expression (3-2) by g ⁇ 1 . The power normalization described with reference to the first embodiment is not performed then.
  • FIG. 13 illustrates the configuration of the non-linear precoder a 33
  • FIG. 14 is a flowchart illustrating the operation of the non-linear precoder a 33 .
  • the operation of the non-linear precoder a 33 is sequentially described.
  • Step S 1 The interference calculator a 131 and the linear filter multiplier a 134 respectively acquire the interference coefficient filter F and the linear filter P from the filter calculator a 31 .
  • Step S 2 The non-linear precoder a 33 substitutes 1 for n representing the number of a mobile-station device that is calculating the transmit signal.
  • Step S 3 The non-linear precoder a 33 substitutes v1 for the specific signal s1 addressed to a mobile-station device B 31 .
  • Step S 5 The interference calculator a 131 calculates an interference signal f2 received by a mobile-station device B 32 , using v1, in accordance with the following Expression.
  • f 2 F (2,1)* v 1 (3-6) where F(p, q) represents a component at a second row and a first column of the matrix F.
  • Step S 6 An interference subtracter a 133 - 2 subtracts f2 from the specific signal s2 addressed to the mobile-station device B 32 , thereby calculating signal s2 ⁇ f2.
  • Step S 7 A first modulo calculator a 132 - 2 performs the modulo operation on s2 ⁇ f2, thereby calculating a signal v2.
  • Step S 4 The non-linear precoder a 33 increments the value of n by 1.
  • the interference calculator a 131 calculates an interference signal fn received by the mobile-station device n using v1 through v(n ⁇ 1) in accordance with the following Expression.
  • fn F ( n, 1 :n ⁇ 1)*[ v 1 ,v 2 , . . . ,v ( n ⁇ 1)] t (3-7)
  • F(n, 1:n ⁇ 1) represents a row vector indicating components at first column through (n ⁇ 1)-th column at an n-th row of the matrix F.
  • Step S 6 An n-th interference subtracter a 133 - n subtracts fn from a specific signal sn addressed to the n-th mobile-station device B 3 n , thereby calculating sn ⁇ fn.
  • Step S 7 A modulo calculator a 132 - n performs the modulo operation on sn ⁇ fn, thereby calculating a signal vn.
  • the modulo operation reduces transmit signal power addressed to each mobile-station device.
  • the components of the signal x are transmit signals sequentially transmitted via the antennas a 101 - 1 through a 101 -N.
  • the signal x is input to the frame constructor a 142 .
  • THP can also be considered to be a perturbation vector search algorithm as described below.
  • THP is not an algorithm to search for a truly optimal perturbation vector but an algorithm to search for a quasi-optimal perturbation vector using an amount of calculation lower than that for VP.
  • the modulo calculators a 132 - 1 through a 132 -N add, to I-ch and Q-ch of the specific signal s, a signal of an integer multiple of the modulo width.
  • the signal x calculated by the non-linear precoder a 33 is not necessarily a signal, expressed by Expression (1-2), with an optimal perturbation vector added thereto, but is a quasi-optimal signal x because power control through the modulo operation is performed thereon. If the modulo operation is not performed, the norm becomes definitely small in comparison with Ws.
  • the signal x is calculated in accordance with the above-described algorithm. Unlike VP in the first embodiment, searching for all the candidates of perturbation vectors becomes unnecessary. The amount of calculation is thus reduced. In this way, this arrangement minimizes an increase in the overhead involved in the DMRS insertion by spatial multiplexing DMRS in the NLP MU-MIMO system while the mobile-station device can normally estimate the complex gain of the data signal.
  • the present invention is not limited to the configurations and the like illustrated in the accompanied drawings.
  • the configurations and the like may be modified appropriately within the scope where the effects of the present invention are provided.
  • the configurations and the like may be appropriately modified without departing from the scope of the present invention.
  • a program to perform the functions described with reference to the embodiments may be recorded on a computer-readable recording medium.
  • the program recorded on the recording medium may be installed onto a computer system, and the computer system may execute the program. The process of each element is thus performed.
  • the term “computer system” includes OS, and hardware such as a peripheral device.
  • the “computer system” includes a homepage providing environment (or display environment) if a WWW system is used.
  • the term “computer readable recording medium” refers to a portable medium, such as a flexible disk, a magneto-optical disk, ROM, or CD-ROM, or a recording device, such as a hard disk, built into the computer system.
  • the “computer readable recording medium” may include a communication line that holds dynamically the program for a short period of time.
  • the communication line transmits the program via a communication channel such as a network like the Internet or a telephone line.
  • the “computer readable recording medium” may also include a volatile memory in the computer system that may be a server or a client and stores the program for a predetermined period of time.
  • the program may implement part of the above-described function.
  • the part of the above-described function may be used in combination with a program previously recorded on the computer system.
  • the present invention finds applications in communication apparatuses.
  • DRMS generator a 125 . . . specific signal constructor, a 133 . . . interference subtracter, a 132 . . . modulo calculator, a 134 . . . linear filter multiplier, a 141 . . . CRS generator, a 142 . . . frame constructor, a 143 . . . IFFT unit, a 144 . . . GI inserter, a 145 . . . transmitter, a 226 . . . CRS corrector, b 12 . . . DMRS channel estimator, b 101 . . . antenna, b 102 . . .
  • b 103 . . . GI remover b 104 . . . FFT unit, b 105 . . . signal separator, b 106 . . . channel compensator, b 107 . . . CRS channel estimator, b 108 . . . channel state information generator, b 109 . . . modulo calculator, b 110 . . . demodulator, b 111 . . . decoder, b 121 . . . perturbation vector candidate selector, b 122 . . . perturbation vector adder, b 123 . . . interim channel estimating unit, b 124 . . .
  • channel compensating unit b 125 . . . modulo calculator, b 126 . . . demodulator, b 127 . . . perturbation vector evaluation value calculator, b 128 . . . perturbation vector estimating unit, b 131 . . . IFFT unit, b 132 . . . GI inserter, b 133 . . . transmitter, 300 . . . two-dimensional Euclidean algorithm unit, 301 . . . vector storage unit, 303 . . . difference vector calculator, 305 . . . difference vector norm calculator, 307 . . . convergence determining unit, 309 . . . norm calculator, 320 .

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