US9218513B2 - Circuit and method with improved mixer linearity - Google Patents
Circuit and method with improved mixer linearity Download PDFInfo
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- US9218513B2 US9218513B2 US13/724,808 US201213724808A US9218513B2 US 9218513 B2 US9218513 B2 US 9218513B2 US 201213724808 A US201213724808 A US 201213724808A US 9218513 B2 US9218513 B2 US 9218513B2
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- G—PHYSICS
- G06—COMPUTING; CALCULATING OR COUNTING
- G06G—ANALOGUE COMPUTERS
- G06G7/00—Devices in which the computing operation is performed by varying electric or magnetic quantities
- G06G7/12—Arrangements for performing computing operations, e.g. operational amplifiers
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/14—Balanced arrangements
- H03D7/1425—Balanced arrangements with transistors
- H03D7/1441—Balanced arrangements with transistors using field-effect transistors
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/14—Balanced arrangements
- H03D7/1425—Balanced arrangements with transistors
- H03D7/1458—Double balanced arrangements, i.e. where both input signals are differential
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/14—Balanced arrangements
- H03D7/1425—Balanced arrangements with transistors
- H03D7/1466—Passive mixer arrangements
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/16—Multiple-frequency-changing
- H03D7/165—Multiple-frequency-changing at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
- H03D2200/0041—Functional aspects of demodulators
- H03D2200/0066—Mixing
- H03D2200/0074—Mixing using a resistive mixer or a passive mixer
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
- H03D2200/0041—Functional aspects of demodulators
- H03D2200/0088—Reduction of intermodulation, nonlinearities, adjacent channel interference; intercept points of harmonics or intermodulation products
Definitions
- the present invention is in the general field of electronics and relates to improvements in switching, and in some embodiments to improved mixer devices using switches.
- the transmit signal typically occupies a well-defined part of the frequency spectrum and power emitted outside this frequency range is subject to maximum emission limits imposed by regulatory or other requirements. These requirements ensure communication equipment using different parts of the frequency spectrum do not excessively interfere with one another.
- E-UTRA evolved-UMTS Terrestrial Radio Access
- LTE Long Term Evolution
- LTE band 13 transmit band at 777 MHz to 787 MHz
- US public safety band between 769 MHz and 775
- LTE band 1 transmit band at 1920 MHz to 1980 MHz
- PHS Personal Handy-phone System
- FIG. 1 shows a simplified block diagram of a representative architecture of a state-of-the-art LTE transmitter.
- the user data is interleaved with control data (omitted in the diagram) and modulated using a technique called SC-FDMA (Single-Carrier FDMA) which yields a stream of time-domain data symbols. Between these symbols a cyclic prefix is inserted to effectively create a guard time between the data symbols.
- SC-FDMA Single-Carrier FDMA
- the data stream is then up-sampled to a rate multiple times the native LTE symbol rate which by the process known as aliasing again produces unwanted out-of-band emissions. These can be removed using the anti-aliasing filter shown in FIG. 1 .
- the signal can then be converted from the digital into the analog domain using a DAC.
- the radio topology shown is known as a zero-IF architecture where the complex baseband signal is represented by two real-valued signal paths (I and Q) in the analog domain. This type of architecture is very common in low-cost transceiver designs based on CMOS technology.
- the signal is filtered again, mainly to remove DAC quantization noise at the duplex offset for FDD radio bands. Then, the I and Q signal paths are jointly up-converted onto an RF carrier in the IQ modulator block. The resulting RF signal is then amplified and filtered again before being transmitted from the antenna.
- the previously-discussed emission limits set design constraints on a number of blocks shown in the architecture in FIG. 1 .
- the combination of pulse shaping, digital anti-aliasing filter and analog reconstructing filter must suppress out-of-band power at the critical frequency offsets adequately so they make negligible contribution to out-of-band noise after up-conversion to RF and RF amplification.
- all blocks in the analog and RF signal path must be linear enough to avoid spectral re-growth products violating the out-of-band emission limits. This is of particular importance to wide transmit signal allocations where the distance between allocated transmit channel and protected frequency band is less that the bandwidth of the signal and third-order intermodulation products are critical.
- V out c 1 ⁇ V in +c 3 ⁇ V in 3 +c 5 ⁇ V in 5 + . . . .
- V out, I A ⁇ ( c 1 +3 ⁇ 4 c 3 +5 ⁇ 8 c 5 + . . . ) ⁇ cos( ⁇ IF ⁇ t )+ A 3 ⁇ (+1 ⁇ 4 c 3 + 5/16 c 5 + . . . ) ⁇ cos(3 ⁇ IF ⁇ t )+ A 5 ⁇ (+ 1/16 c 5 + . . . .
- FIG. 2 shows the spectrum at RF output. Odd-order non-linearity in the IQ modulator creates unwanted out-of-band products at frequency offsets corresponding to odd multiples of the input signal frequency offset from carrier.
- the rejection of the third-order product at ⁇ LO ⁇ 3 ⁇ IF is given by the ratio 16 c 1 2 /c 3 2 .
- Third-order nonlinearity is often determined and chararacterized by passing two tones of equal amplitude through a nonlinear system and measuring the rejection between unwanted intermodulation product and wanted tones.
- that rejection normally referred to as IM 3
- IM 3 16/9 ⁇ c 1 2 /c 3 2 .
- the 3GPP specification for the critical LTE bands allows reduction of transmitter output power for the most challenging out-of-band emission scenarios. This automatically reduces the effect of non-linearity but also reduces the transmit range of the device. It is therefore highly desirable to meet the out-of-band emission specifications even at maximum allowed output power by dedicated design.
- the most straight-forward way of reducing out-of-band emissions is to employ filtering of the signal in the RF front-end after the power amplifier just before the signal is coupled into the antenna.
- Recent advancements of SAW and FBAR technology have made it possible to achieve significant rejection of unwanted out-of-band power just a few megahertz away from the band edge.
- even the most sophisticated filters are not able to fully reject the discussed narrow-band signals in case of LTE bands 13 and band 1.
- a second RF filter is added before the signal is amplified in the power amplifier to clear up unwanted frequency products in the transmit chain which increases the cost and footprint of the overall solution.
- a well-known technique is to use active current-driven mixer topologies (known as Gilbert mixers) where the input current is linearized through a feedback-loop in the transconductor stage.
- active current-driven mixer topologies known as Gilbert mixers
- negative feedback is created using an operational amplifier, the penalty being that the modulator is now noisy and will not meet the Rx-band noise requirements for LTE FDD bands.
- FIG. 3 shows a simplified block diagram of a passive Transmit (Tx) IQ Modulator, essentially a voltage-mode passive mixer used with 25% duty-cycle local oscillator (LO) I and Q waveforms used, to realize the I+jQ summation in voltage mode at RF.
- Tx passive Transmit
- LO local oscillator
- a passive mixer as this general type of mixer exhibits extremely low noise.
- the dominant noise source is thermal noise of switch on-resistance, which can be as low as 0.575 nV/ ⁇ Hz for 20 ⁇ switch on-resistance.
- Such mixers are also as linear as the switches used within them, with nMOS and CMOS (transmission gate) switches commonly used. It should be noted that in the context of a switch, non-linearity is expressed as the deviation from constant resistance with the magnitude of signal passing through the switch in its on state. Thus a truly linear switch has constant on-resistance regardless of the magnitude of a signal carried.
- the time-varying I and Q input signals V in, IP , V in, IN , V in, QP and V in, QN are periodically sampled by the clock signals ⁇ 1 , ⁇ Q , ⁇ I ′ and ⁇ Q ′.
- the clock signals are periodic with an angular frequency ⁇ LO .
- the clock signals are arranged such that they have roughly 25% duty cycle which means only one pair of switches is on at any moment in time and the other three pairs of switches are off.
- the use of 25% duty cycle has a number of advantages such as improved conversion gain and reduced mixer noise figure.
- the clock signals are illustrated in FIG. 4 .
- any switch non-linearity will get modulated onto the sampled input waveform thereby creating undesirable IF harmonics which are then up-converted to RF.
- the most problematic harmonic is the third harmonic, which appears at RF with an angular frequency ⁇ LO ⁇ 3 ⁇ IF .
- Higher order products also exist but they are generally less problematic than the aforementioned third-order product.
- Equation 3 gives the CMOS switch on-resistance R switch to the first-order.
- K n and K p are process-dependent constants relating to the transconductances of the transistors
- W and L are the width and length of the transistors, respectively
- V gsn is the gate-source voltage of the nMOS
- V sgp is the source-gate voltage of the pMOS
- V Tn and V Tp are the threshold voltages of the two devices, respectively.
- the values of V gsn and V sgp both depend on the input voltage level, V in .
- the on-resistance of the transmission gate is a function of the input voltage when in an on-state; this is the root-cause for mixer nonlinearity.
- the gate-to-channel capacitances are also non-linear and input-dependent, contributing even further to the sampling distortion but can be minimized or even eliminated by connecting the devices' bulk terminal to the source terminal (possible for the nMOS device if a twin-tub process technology is available) and by using non-overlapping clocks which also minimizes the effect of the gate-to-source/drain parasitic capacitances.
- FIG. 5 depicts the non-linear sampling switch in sampling mode. It is shown to have a main current path with a variable series resistance (R S1 ) variable with gate source voltage (V gs ), a gate-source capacitance (C gs ) a gate-drain capacitance (C gd ) and a capacitor (C 1 ) in series with the main current path representing RF amplifier gate capacitance.
- R S1 variable series resistance
- V gs gate source voltage
- C gs gate-source capacitance
- C gd gate-drain capacitance
- C 1 capacitor
- V C ⁇ ⁇ 1 V in ⁇ ( s ) K n ⁇ W L ⁇ ( V gn - V Tn - V in ) + K P ⁇ W L ⁇ ( V in - V gp - ⁇ V Tp ⁇ ) K n ⁇ W L ⁇ ( V gn - V Tn - V in ) + K p ⁇ W L ⁇ ( V in - V gp - ⁇ V Tp ⁇ ) + sC 1 ( 5 )
- Rearranging (5) gives a non-linear expression of V in as equation 6:
- V C ⁇ ⁇ 1 ⁇ ( s ) K n ⁇ W L ⁇ ( V gn - V Tn - V in ) + K p ⁇ W L ⁇ ( V in - V gp - ⁇ V Tp ⁇ ) K n ⁇ W L ⁇ ( V gn - V Tn - V in ) + K p ⁇ W L ⁇ ( V in - V gp - ⁇ V Tp ⁇ ) + sC 1 ⁇ V in . ( 6 )
- a mixer having input nodes for receiving signals to be mixed, and comprising at least one gate arranged to selectively couple an input node to an output node, the gate being arranged to be driven by a sampling clock signal, the gate comprising at least one FET having a respective control node, the mixer having circuitry for applying a voltage between the gate and source nodes of the at least one FET in the on-state, which voltage is independent of input signals at the input nodes.
- the mixer may be a quadrature mixer.
- the gate may comprise a pair of complementary FETS connected to form a transmission gate.
- the circuitry may comprise a capacitor connected in a bootstrap configuration.
- the circuitry may be configured to apply a voltage substantially equal to the sum of the voltage at the input node and the mean of the supply rail voltages to the control node of the FET.
- a switching circuit comprising a transmission gate, the transmission gate comprising an nMOS transistor and a pMOS transistor, each of the transistors having a respective control node, the switching circuit further comprising respective capacitors connected in bootstrap configuration to each control node whereby the on-resistance of the transmission gate is substantially constant and independent of signals gated by the transmission gate.
- a sample-and-hold circuit comprising a switching circuit of the second aspect.
- an analog-to-digital converter comprising a sample-and-hold circuit of the third aspect.
- a method of operating a gating circuit using an input signal comprising adding, to the input signal voltage, a voltage substantially equal to the mean of the supply rail voltages.
- a mixer having input nodes for receiving signals to be mixed, and comprising at least two gates, each arranged to selectively couple a respective input node to an output node, the gates being arranged to be driven by respective sampling clock signals, each gate comprising at least one FET having a respective control node, the mixer having circuitry for applying a voltage between the gate and source nodes of the at least one FET in the on state, which is independent of voltages at the input nodes.
- a mixer having an input node for receiving a voltage, and plural output nodes, the mixer comprising at least two gates, each arranged to selectively couple a respective input node to a respective output node, the gates being arranged to be driven by respective sampling clock signals, each gate comprising at least one FET having a respective control node, the mixer having circuitry for applying a voltage between the gate and source nodes of the at least one FET in the on state, which is independent of voltages at the input node.
- sampling clock signals may be non-overlapping.
- the mixer may in some embodiments be a quadrature mixer.
- the gate comprises a pair of complementary FETS connected to form a transmission gate.
- the circuitry comprises a capacitor connected in a bootstrap configuration.
- FIG. 1 is a simplified block diagram of one example of an LTE transmitter.
- FIG. 2 shows a spectrum at RF output of the transmitter of FIG. 1 when transmitting a narrow band signal and in the presence of non-linearity.
- FIG. 3 shows a simplified block diagram of a passive Transmit (Tx) IQ Modulator.
- FIG. 4 shows clock signals used in the modulator of FIG. 3 .
- FIG. 5 shows an equivalent diagram of a non-linear sampling switch.
- FIG. 6 shows a bootstrap circuit using an nMOS sampling switch.
- FIG. 7 shows a simplified CMOS sampling switch lineariser circuit.
- FIG. 8 is a plot of conventional nMOS, conventional CMOS and linearised CMOS switch resistance against input voltage.
- FIG. 9 shows comparative output spectrum for conventional CMOS switch vs. linearised CMOS switch Tx IQ Modulator.
- FIG. 10 shows linearised Tx IQ Modulator LOIM3-to-Wanted performance with Monte-Carlo process variation at 1.2V and 60° C. and 1.14V and 105° C.
- V drive a virtual drive voltage
- V g V drive +V in (7)
- V C ⁇ ⁇ 1 ⁇ ( s ) K n ⁇ W L ⁇ ( V drive - V Tn ) + K p ⁇ W L ⁇ ( - V drive - ⁇ V Tp ⁇ ) K n ⁇ W L ⁇ ( V drive - V Tn ) + K p ⁇ W L ⁇ ( - V drive - ⁇ V Tp ⁇ ) + sC 1 ⁇ V in ( 8 )
- One method of creating such a virtual drive voltage is to use a capacitor which is pre-charged while the switch is off and then provides a raised gate voltage in the on-state by a bootstrapping scheme.
- the circuit has an nMOS switch ( 61 ) with its current path between drain ( 62 ) and source ( 63 ) controlled by the voltage at its gate ( 64 ).
- the gate is connectable to a reference voltage (V SS ) via a first switch ( 66 ) and to a circuit node ( 67 ) via a second switch ( 68 ).
- the circuit node ( 67 ) is connected to a positive supply rail V DD via a third switch ( 69 ), and to a first, upper as shown, plate ( 70 ) of a capacitor ( 71 ) having a second plate ( 72 ).
- the second, lower as shown, plate ( 72 ) of the capacitor ( 71 ) is connected to an input control node ( 73 ) via a fourth switch ( 74 ) and to a common mode node ( 75 ) via a fifth switch ( 76 ).
- the first, third and fifth switches ( 66 , 69 , 76 ) are controlled by a first clock signal ⁇ ′ and the second and fourth switches ( 68 , 74 ) by a second non-overlapping clock signal ⁇ .
- the common mode node ( 75 ) V cm is set, in this embodiment, to a constant reference approximately half way between the reference voltage V SS and the positive supply V DD and corresponds approximately to the average level of V in . In other embodiments it is derived directly from the supply rails—for example via a divider circuit. In both cases it will be set to be a value between 40% and 60% of the difference between the reference voltage V WW and the positive supply V DD .
- V gn V DD ⁇ V cm +V in (9)
- V gn V DD ⁇ V cm +V in (9)
- the gate-source voltage V gsn of the nMOS switch ( 61 ) is V gn ⁇ V in .
- first, fourth and fifth switches ( 66 , 74 , 76 ) are nMOS devices and second and third switches ( 68 , 69 ) pMOS devices. All of the first-fifth switches are sized an order of magnitude smaller than the dimensions of the nMOS switch to minimize the depletion of the charge of capacitor ( 71 ) via parasitics of the first-fifth switches. The dominant parasitic is then that of the nMOS switch ( 61 ), and the dimensions of the capacitor ( 71 ) are at least an order of magnitude higher than its parasitic capacitance.
- the nMOS device is replaced by a transmission gate as shown in FIG. 7 , and then used in the linearization of a Tx IQ Modulator.
- non-overlapping clocks are used and the pMOS and nMOS devices are sized identically within the CMOS switch to address the mixer issue of signal-dependent charge injection (to a first order) from switch channel capacitance, gate-to-drain overlap capacitance and junction capacitance point of view. Note that these features are specific to mixers and will not all be essential in all switch applications.
- CMOS switch linearization in addition alleviates the effect of signal-dependent switch on-resistance on IF sampling non-linearity and yields better distortion performance over a bootstrapped nMOS switch for IF-to-RF up-conversion applications.
- circuit shown in FIG. 7 represents one of the eight switches shown in the diagram in FIG. 3 . Although the circuit is similar to that of FIG. 6 , it will be described in full for clarity.
- the circuit has CMOS transmission gate ( 101 ) comprising an nMOS switch ( 61 ) and a pMOS switch ( 102 ) having their current paths connected in full parallel.
- the nMOS switch ( 61 ) has its current path between drain ( 62 ) and source ( 63 ) controlled by the voltage at its gate ( 64 ).
- the gate is connectable to a reference voltage (V SS ) via a first switch ( 66 ) and to a circuit node ( 67 ) via a second switch ( 68 ).
- the circuit node ( 67 ) is connected to a positive supply rail V DD via a third switch ( 69 ), and to a first, upper as shown, plate ( 70 ) of a first capacitor ( 71 ) having a second plate ( 72 ).
- the second, lower as shown, plate ( 72 ) of the first capacitor ( 71 ) is connected to an input control node ( 73 ) via a fourth switch ( 74 ) and to a common mode node ( 75 ) via a fifth switch ( 76 ).
- the pMOS switch ( 102 ) likewise has its current path between its drain ( 103 ) and source ( 104 ) controlled by the voltage at its gate ( 105 ).
- the gate ( 105 ) is connectable to the positive supply rail V DD via a sixth switch ( 106 ), and to a first lower, as shown, plate ( 107 ) of a second capacitor ( 108 ) having a second plate ( 109 ) via a seventh switch ( 110 ).
- the second, upper as shown, plate ( 109 ) of the second capacitor is connected to the second plate ( 72 ) of the first capacitor ( 71 ).
- the first plate ( 107 ) of the second capacitor ( 108 ) is connectable to the reference voltage (V SS ) via an eighth switch ( 111 )
- the first, third, fifth, sixth and eighth switches ( 66 , 69 , 76 , 106 , 111 ) are controlled by a first clock signal ⁇ ′ and the second, fourth and seventh switches ( 68 , 74 , 110 ) by a second clock signal ⁇ where ⁇ ′ and ⁇ are non-overlapping.
- the common mode node ( 75 ) is at V cm , defined as the time-average of the input voltage Vin and fixed at roughly the mean voltage of the positive supply rail V DD and the reference voltage V SS .
- the common mode voltage V cm is applied via fifth switch ( 76 ) to the upper plate ( 109 ) of the second capacitor ( 108 ).
- Eighth switch ( 111 ) applies the reference voltage V SS to the lower plate ( 107 ) of the second capacitor ( 108 ).
- the second capacitor ( 108 ) is charged so that its upper plate is at V cm ⁇ V SS above the lower plate ( 107 ).
- clock ⁇ goes high, opening sixth and eight switches, and closing fourth switch ( 74 ) thereby applying V in to the top plate ( 109 ).
- the gate-source voltage of the pMOS switch ( 102 ) is V cm ⁇ V SS .
- FIG. 8 is a plot of conventional nMOS, conventional CMOS and linearised CMOS switch resistance against input voltage. The plots below were created whilst the CMOS switches were in their on-state.
- the pMOS device bulks were connected to the source to eliminate bulk modulation dependence.
- the nMOS device bulks were connected to V SS to emulate a single-tub n-well CMOS process technology and make the circuit more generic.
- FIG. 9 shows comparative output spectrum for conventional CMOS switch vs. linearised CMOS switch Tx IQ Modulator.
- the spectrum shows a wanted tone at 786 MHz, 4 MHz above the carrier at 782 MHz.
- the third-order LOIM3 product is located at 770 MHz. As can be seen the linearization reduces the power in this unwanted tone by more than 30 dB.
- FIG. 10 gives linearised Tx IQ Modulator LOIM3-to-Wanted performance with Monte-Carlo process variation at 1.2V and 60 deg C. and 1.14V and 105 deg C. showing that target ⁇ 66 dBc specification is achieved, enjoyingwhile impossible with conventional CMOS-based Tx IQ Modulator.
- the switch Used in a mixer, the switch allows embodiments to achieve extremely high linearity at the same time as high conversion gain and low noise figure.
- the power consumption is comparable to a standard implementation.
- the design guarantees performance over process corners, temperature and supply voltage which means no calibration is necessary at production.
- out-of-band emissions can be kept low using off-the-shelf components for external filtering and no back-off of transmit power is needed to meet the emission criteria.
- Embodiments of a mixer as described can meet LTE linearity requirements and at the same time adds little noise to the system. It may be designed to have low power consumption and to not require calibration.
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Abstract
Description
V I =A·cos(ΩIF ·t) and
V Q A·sin(ΩIF ·t).
The baseband signal path is typically realized using differential circuitry which has almost negligible even-order non-linearity. However, assume a degree of odd-order non-linearity which can be expressed as a polynomial of the form Vout=c1·Vin+c3·Vin 3+c5·Vin 5+ . . . .
Using the complex tone as an input signal Vin yields the following output signals:
V out, I =A·(c 1+¾c 3+⅝c 5+ . . . )·cos(ΩIF ·t)+A 3·(+¼c 3+ 5/16c 5+ . . . )·cos(3·ΩIF ·t)+A 5·(+ 1/16c 5+ . . . )·cos(5·ΩIF ·t)+ (1)
V out, Q =A·(c 1+¾c 3+⅝c 5+ . . . )·sin(ΩIF ·t)+A 3·(−¼c 3− 5/16c 5+ . . . )·sin(3·ΩIF ·t)+A 5·(+ 1/16c 5+ . . . )·sin(5·ΩIF ·t)+ (2)
When (1) and (2) are up-converted to RF in a quadrature mixer and summed, the resulting spectrum contains frequency products at odd harmonics of the input tone. The tones appear only on one side of the carrier due to the image cancellation occurring in the summation stage of the IQ modulator.
Here, Kn and Kp are process-dependent constants relating to the transconductances of the transistors, W and L are the width and length of the transistors, respectively, Vgsn is the gate-source voltage of the nMOS, Vsgp is the source-gate voltage of the pMOS and VTn and VTp are the threshold voltages of the two devices, respectively. In this case the values of Vgsn and Vsgp both depend on the input voltage level, Vin.
Substituting (3) in (4) and recognizing that Vgs for switch S1 is given by Vgs=Vg−Vin, equation (5) results in the following expression:
Rearranging (5) gives a non-linear expression of Vin as equation 6:
V g =V drive +V in (7)
Then equation (6) becomes
V gn =V DD −V cm +V in (9)
The gate-source voltage Vgsn of the nMOS switch (61) is Vgn−Vin.
Simplifying: V gsn=(V DD −V cm +V in)−V in =V DD −V cm (10)
for most input voltages, within the limits of reasonable and finite on-resistance of the second and fourth switches (68,74), and the switch is guaranteed linear, in other words has a constant on-resistance, since its gate-source voltage is independent of Vin. Typically, first, fourth and fifth switches (66,74,76) are nMOS devices and second and third switches (68,69) pMOS devices. All of the first-fifth switches are sized an order of magnitude smaller than the dimensions of the nMOS switch to minimize the depletion of the charge of capacitor (71) via parasitics of the first-fifth switches. The dominant parasitic is then that of the nMOS switch (61), and the dimensions of the capacitor (71) are at least an order of magnitude higher than its parasitic capacitance.
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US20180219476A1 (en) * | 2017-01-31 | 2018-08-02 | Infineon Technologies Ag | Switched-capacitor circuit and method of operating a switched-capacitor circuit |
US10044321B2 (en) | 2016-08-02 | 2018-08-07 | Samsung Electronics Co., Ltd | System and method for linearizing a transmitter by rejecting harmonics at mixer output |
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US8525574B1 (en) * | 2012-05-15 | 2013-09-03 | Lsi Corporation | Bootstrap switch circuit with over-voltage prevention |
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Cited By (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US9407478B1 (en) * | 2015-08-27 | 2016-08-02 | Telefonaktiebolaget Lm Ericsson (Publ) | Low power and area bootstrapped passive mixer with shared capacitances |
US10044321B2 (en) | 2016-08-02 | 2018-08-07 | Samsung Electronics Co., Ltd | System and method for linearizing a transmitter by rejecting harmonics at mixer output |
US20180219476A1 (en) * | 2017-01-31 | 2018-08-02 | Infineon Technologies Ag | Switched-capacitor circuit and method of operating a switched-capacitor circuit |
US10236765B2 (en) * | 2017-01-31 | 2019-03-19 | Infineon Technologies Ag | Switched-capacitor circuit and method of operating a switched-capacitor circuit |
Also Published As
Publication number | Publication date |
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US20140176224A1 (en) | 2014-06-26 |
EP2747278A1 (en) | 2014-06-25 |
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