US7791909B2 - Quasi-resonant converter and controlling method thereof - Google Patents
Quasi-resonant converter and controlling method thereof Download PDFInfo
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- US7791909B2 US7791909B2 US11/818,421 US81842107A US7791909B2 US 7791909 B2 US7791909 B2 US 7791909B2 US 81842107 A US81842107 A US 81842107A US 7791909 B2 US7791909 B2 US 7791909B2
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
- H02M3/33523—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Definitions
- the present invention relates to a converter. More particularly, the present invention relates to a quasi-resonant converter.
- a converter transforms one DC voltage to at least one other DC voltage.
- the DC output voltage outputted from the converter can be greater or smaller than an input voltage.
- Such a converter is usually used in power electronic devices, particularly, battery power supplies such as a mobile phone or a laptop computer.
- Quasi-resonant converters are widely used at present because a quasi-resonant converter increases power conversion efficiency and reduces electromagnetic interference (EMI).
- EMI electromagnetic interference
- a quasi-resonant converter turns on a switching transistor when the lowest voltage is applied to both ends of the switching transistor due to a resonance. By such a scheme, a switching noise and a switching loss can be reduced in the quasi-resonant converter.
- embodiments of the present invention include a quasi-resonant converter and a controlling method thereof, where a switching frequency of a switching transistor is restricted to a predetermined range regardless of load.
- a quasi-resonant converter can include a primary coil of a transformer, a switch, a switch voltage detector, and a switching controller.
- the primary coil can include a first end electrically coupled to a rectified DC voltage signal.
- the switch is electrically coupled to a second end of the primary coil of the transformer.
- the switch voltage detector senses a first signal denoting voltages at both ends of the switch.
- the switching controller sets a predetermined first period and a second period following the first period. Then, the switching controller turns on the switch at time point where the first signal falls to a minimum voltage if the second period includes the time point, and turns on the switch at a time point of ending of the second period if the second period does not include the time point.
- the minimum voltage may be a voltage at a time when the first signal falls to a minimum after turning off the switch.
- the first period may start at a time when the switch is turned on.
- the switching controller may decide a time of turning off the switch by comparing a signal corresponding to a current flowing through the switch and a signal corresponding to an output voltage of the quasi-resonant converter.
- the delay circuit may include a first resistor and a second resistor electrically coupled in series between the secondary coil and a ground, and a capacitor electrically coupled between a contact node of the first resistor and second resistor and the ground.
- a method for controlling a quasi-resonant converter, which can include a primary coil of a transformer having a first end electrically coupled to a rectified DC voltage signal and a switch electrically coupled to a second end of the primary coil of the transformer.
- a first signal denoting voltages at both ends of the switch can be sensed.
- a predetermined first period and a second period following the first period can be set. Then, the switch can be turned on at time point where the first signal falls to a minimum voltage if the second period includes the time point, and the switch is turned on at a time point of ending of the second period if the second period does not include the time point.
- the method may further include deciding a time of turning off the switch by comparing a signal corresponding to a current flowing through the switch and a signal corresponding to an output voltage of the quasi-resonant converter.
- the minimum voltage may be a voltage of a time when the first signal falls to a minimum after the switch is turned off.
- the first period may be a period starting at a time of turning on the switch.
- FIG. 1 is a diagram illustrating a quasi-resonant converter.
- FIG. 2 is a diagram illustrating signals outputted from each element in FIG. 1 .
- FIG. 3 is a schematic diagram illustrating a drain-source voltage Vds, a signal V 4 , a signal V 5 , and a signal V 6 of FIG. 1 .
- FIG. 4 is a diagram illustrating signals when an output load is higher than the output load of FIG. 2 .
- FIG. 5 is a diagram illustrating signals when an output load is higher than the output load of FIG. 4 .
- FIG. 6 is a graph showing relation between an output load Po and a switching frequency f in a quasi-resonant converter according to an exemplary embodiment of the present invention, and relation between an output load Po and a switching frequency f in a typical quasi-resonant converter.
- an element when it is described that an element is “coupled” to another element, the element may be “directly coupled” to the other element or “electrically coupled” to the other element through a third element.
- FIG. 1 is a diagram illustrating a quasi-resonant converter.
- the quasi-resonant converter can include a power supply 100 , an output unit 200 , a bias voltage supply 300 , a switching controller 400 , and a switching voltage detector 500 .
- the power supply 100 can include a bridge diode BD for rectifying an input AC voltage, a capacitor Cin for smoothing the rectified voltage, and a primary coil L 1 of a transformer connected to one end of the capacitor Cin.
- the power supply 100 transforms an AC voltage to a DC voltage Vin using the bridge diode BD and the capacitor Cin, and supplies power to the secondary side of the transformer, referred to as the output unit 200 , according to a duty of the switching transistor Qsw.
- the output unit 200 can include a secondary coil L 2 of the transformer, a diode D 1 having an anode connected to one end of the secondary coil L 2 of the transformer, and a capacitor C 1 connected between a cathode of the diode D 1 and the ground. A voltage between ends of the capacitor C 1 is referred to as an output voltage Vo.
- the bias voltage supply 300 can include a secondary coil L 3 of a transformer, a diode D 2 having an anode connected to the secondary coil L 3 of the transformer, and a capacitor C 2 , connected between a cathode of the diode D 2 and the ground.
- the switching controller 400 can be embodied as a general IC.
- the bias voltage supply 300 can supply a bias voltage to drive the IC of the switching controller 400 .
- a switching transistor Qsw starts switching, the secondary coil L 3 of the transformer and the diode D 2 become driven, thereby generating a bias voltage Vcc between ends of the capacitor C 2 .
- the switching controller 400 can include a pulse with modulator (PWM) signal generator 410 , a signal generator 420 , a first vibrator 430 , a second vibrator 440 , and a comparator 450 .
- the switching controller 400 can receive a feedback signal Vfb, a sensing signal Vsense that senses a current Ids flowing through the switching transistor Qsw, and an output signal V 5 of the switching voltage detector 500 .
- the switching controller 400 can output a signal VGS for controlling a turn-off/turn-on operation of the switching transistor Qsw.
- the feedback signal Vfb can be a signal having information corresponding to the output voltage Vo, and can be used to decide a time of turning off the switching transistor Qsw. Since a method of generating the feedback signal Vfb is not directly related to the present invention and is well known to a person of ordinary skill in the art, the detailed description thereof will be omitted.
- the PWM signal generator 410 can receive a signal V 3 transmitted from the signal generator 420 , a sensing signal Vsense, and a feedback signal Vfb, and can output a signal VGS for controlling the turn-on/turn-off operation of the switching transistor Qsw.
- the first vibrator 430 can generate a signal V 1 using a signal VGS outputted from the PWM signal generator 410 and can transmit the generated signal V 1 to the signal generator 420 and the second vibrator 440 .
- the second vibrator 440 can generate a signal V 2 using the signal V 1 transmitted from the first vibrator 430 , and can transmit the signal V 2 to the signal generator 420 .
- the signal generator 420 can generate a signal V 3 using the V 1 signal, the V 2 signal, and an output signal V 6 of the comparator 450 , and can transmit the V 3 signal to the PWM signal generator 410 in order to turn on the switching transistor Qsw. If the signal V 2 is in a high state, for example, during a Tw period in FIG. 2 , and if the output signal V 6 of the comparator 450 changes from high to low state, the signal generator 420 can output a short pulse. Also, if the signal V 2 is in high state, and if the output signal V 6 of the comparator 450 does not change from high to low state, the signal generator 420 can output a short pulse at a time when the V 2 signal changes from high to low state.
- the PWM signal generator 410 can output a VGS signal changing from low to high state in order to turn on the switching transistor Qsw when receiving a short pulse from the signal generator 420 .
- the comparator 450 can receive the output signal V 5 of the switching voltage detector 500 through an inverting terminal ⁇ and a reference voltage Vref 1 /Vref 2 through a non-inverting terminal +, and can output a signal V 6 by comparing the output signal V 5 and the reference voltage Vref 1 /Vref 2 .
- the reference voltage Vref 1 can be a voltage higher than the reference voltage Vref 2 .
- the comparator 450 can output a high signal if the signal V 5 , inputted to the inverting terminal + is higher than the reference voltage Vref 1 , and can output a low signal if the signal V 5 is lower than the reference voltage Vref 2 .
- the comparator 450 can sustain the previous signal state if the signal V 5 is in between the reference voltages Vref 1 and Vref 2 .
- Such a comparator 450 can be embodied using a Schmitt Trigger.
- the switching voltage detector 500 can generate a signal V 5 corresponding to a drain-source voltage of the switching transistor Qsw using the voltage V 4 of the secondary coil L 3 of the transformer, and can transmit the generated signal V 5 to the switching controller 400 .
- the switching voltage detector 500 can include resistors R 1 and R 2 , a capacitor C 3 , and a diode D 3 .
- the resistors R 1 and R 2 can be connected between the secondary coil L 3 of the transformer and the ground in series, and the capacitor C 3 and the diode D 3 can be connected between a node, connecting the resistors R 1 and R 2 , and the ground in parallel.
- the secondary coil L 3 of the transformer can reflect the voltage across the primary coil L 1 .
- the voltage across the primary coil L 1 can be a voltage obtained by subtracting the Vin voltage from a drain-source voltage Vds of the switching transistor Qsw. Accordingly, the voltage V 4 across the secondary coil L 3 of the transformer can reflect the drain-source voltage Vds of the switching transistor Qsw.
- the resistors R 1 and R 2 and the capacitor C 3 can function as an RC filter, for example, a delay circuit.
- the voltage V 5 can correspond to a drain source voltage Vds of the switching transistor Qsw as shown in FIG. 3 .
- the diode D 3 can clamp the voltage V 5 not to fall below a predetermined voltage.
- the drain of the switching transistor Qsw can be connected to an end of the primary coil L 1 of the transformer, and a sensing resistor Rsense can be connected between the source of the switching transistor Qsw and the ground.
- a resonance capacitor CR can be additionally connected between the drain and the source of the switching transistor Qsw.
- a parasitic capacitance between the drain and the source of the switching transistor Qsw can be used to induce resonance.
- the quasi-resonant converter will be described under the assumption of using a resonance capacitor CR for convenience.
- the switching transistor Qsw is shown as a MOSFET in FIG. 1 , it can be substituted with other switching transistors that can switch.
- the switching transistor Qsw can become turned on or turned off by being controlled by the output signal VGS of the PWM signal generator 410 .
- FIG. 2 is a diagram illustrating signals outputted by circuit elements in FIG. 1 .
- the switching transistor Qsw is turned on if the output signal V GS of the PWM signal generator 410 changes to a high state. A method of changing the signal V GS to high state will be described in later.
- the current Ids flowing through the switching transistor Qsw increases with a predetermined slope Vin/L 1 .
- the sensing signal Vsense changes in accordance with the current Ids, sensed by the sensing resistor Rsense.
- the sensing signal Vsense is transmitted to the PWM signal generator 410 .
- the PWM signal generator 410 changes the signal V GS from high to low state at a time t 2 by comparing the feedback signal Vfb and the sensing signal Vsense. Accordingly, the switching transistor Qsw is turned off at the time t 2 .
- the switching transistor Qsw When the switching transistor Qsw is turned off at the time t 2 , as shown in (b) of FIG. 2 , the current Is flowing through the diode D 1 is reduced to substantially zero with a slope of ⁇ Vo/L 2 . Also, the drain-source voltage Vds of the switching transistor Qsw increases up to Vin+Vo*Np/Ns, where Np/Ns denotes a turn ratio of the primary side and the secondary side of the transformer.
- the diode D 1 becomes turned off and the secondary coil L 2 changes to high impedance state.
- resonance is induced between the primary coil L 1 of the transformer and the resonance capacitor C R .
- a resonance period is determined by the inductance of the primary coil L 1 and the value of the capacitance of the resonance capacitor C R .
- the first vibrator 430 outputs a signal V 1 according to the signal V GS .
- the first vibrator 430 changes the signal V 1 to high state when the signal V GS changes from low to high state, for example, at the time t 1 in FIG. 2 , and changes the signal V 1 to low state after sustaining the signal V 1 at high state for a predetermined blocking period T B .
- the T B period can be set in a wide range of values.
- the signal generator 420 does not generate a short pulse during the blocking period T B in order to prevent the next turn-on of the switching transistor Qsw.
- the second vibrator 440 outputs a signal V 2 according to the state of the signal V 1 .
- the second vibrator 440 changes the signal V 2 to low state after sustaining the signal V 2 at high state for a sensing period Tw.
- the sensing period Tw starts at a time t 4 when the signal V 1 changes from high to low state.
- the minimum value of a drain-source voltage Vds of the switching transistor Qsw is sensed only during the sensing period Tw.
- the signal generator 420 generates a short pulse when the minimum of the drain-source voltage Vds of the switching transistor Qsw is sensed during the sensing period Tw. Referring to (c) and (e) of FIG.
- the drain-source voltage Vds assumes its minimum within the sensing period Tw at a time t 5 .
- the signal generator 420 generates a short pulse at the time t 5 . If the drain-source voltage Vds does not assume a minimum, during the sensing period Tw as e.g. shown in FIG. 5 , the signal generator 420 generates the short pulse after the sensing period Tw.
- a method of detecting a time when the drain-source voltage Vds assumes a minimum value after resonance starts will be later described in detail with reference to FIG. 3 .
- the signal generator 410 generates a short pulse at the time t 5 , and the PWM signal generator 410 changes the signal V GS to high state at the short pulse. Accordingly, the switching transistor Qsw is turned on at the time t 5 where the drain-source voltage Vds of the switching transistor Qsw is low, thereby reducing the switching loss.
- FIG. 3 is a schematic diagram illustrating the drain-source voltage Vds, a signal V 4 , a signal V 5 , and a signal V 6 of FIG. 1 .
- FIG. 3( a ) illustrates a particular time dependence of the drain-source voltage Vds.
- Such a drain-source voltage Vds makes the primary coil L 1 of the transformer generate a signal falling by as much as the voltage Vin from the drain-source voltage Vds.
- FIG. 3( b ) illustrates that the second coil L 3 of the transformer generates a signal V 4 according to a turn ratio of the transformer.
- the minimum voltage of the drain-source voltage Vds can be sensed using the signal V 4 because the signal V 4 directly reflects the drain-source voltage Vds.
- FIG. 3( c ) illustrates the signal V 5 , generated by the switching voltage detector 500 .
- the signal V 5 is generated by the resistors R 1 and R 2 of the switching voltage detector 500 and a capacitor C 3 .
- the signal V 5 rises and falls, following the signal V 4 by a RC time constant.
- the resistors R 1 and R 2 of the switching voltage detector 500 and the capacitor C 3 function as a delay circuit that delays the signal V 4 to generate the signal V 5 .
- the comparator 450 has two reference voltages Vref 1 and Vref 2 as described above.
- the reference voltage Vref 1 is set as a voltage lower than the maximum voltage of the signal V 5
- the reference voltage Vref 2 is set as the voltage V 5 at the time when the drain-source voltage Vds assumes its minimum after resonance.
- the comparator 450 outputs a high signal if the signal V 5 is higher than the reference voltage Vref 1 , and outputs a low signal if the signal V 5 is lower than the reference voltage Vref 2 .
- the comparator 450 sustains a previous state if the signal V 5 is in between the reference voltages Vref 1 and Vref 2 .
- the output signal V 6 of the comparator 450 changes to high state at a time t 2 ′ and changes from high to low state at a time t 4 ′. Visibly, the signal V 6 changes from high to low state when the signal Vds assumes its minimum.
- the signal generator 420 generates a short pulse at time t 5 of FIG. 2 , when the signal V 2 is in high state and when the signal V 6 changes from high to low state.
- the signal generator 420 does not generate a short pulse in the blocking period T B .
- the signal generator 420 generates a short pulse when the minimum voltage is sensed within the sensing period Tw.
- FIG. 4 is a diagram illustrating signals when an output load is higher than the output load of FIG. 2 .
- FIG. 5 is a diagram illustrating signals when an output load is even higher than the output load of FIG. 4 . Since FIG. 4 and FIG. 5 are analogous to FIG. 2 except the value of the output load, detailed description of duplicated elements will be omitted.
- the switching transistor Qsw becomes turned on when the drain-source voltage Vds assumes its minimum at the first time by inducing the resonance between the primary coil L 1 of the transformer and the resonance capacitor CR.
- the current Ids rises to an even higher peak level compared to FIG. 4 . Therefore, the resonance between the primary coil L 1 of the transformer and the resonance capacitor CR starts even later. Since the blocking period TB and the sensing period Tw are a predetermined value, the minimum of the drain-source voltage Vds is not sensed during the sensing period Tw. In this case, the PWM signal generator 410 forcedly generates a short pulse although the minimum voltage of the signal Vds is not sensed. Accordingly, Ts_max, the maximum of a switching period Ts, becomes T B +Tw.
- the switching period Ts of the switching transistor Qsw will not exceed T B +Tw, although the output load is extremely high. Accordingly, the switching frequency f of the switching transistor Qsw is restricted the following range: 1/( T B +T W ) ⁇ f ⁇ 1 /T B (1)
- FIG. 6 is a graph showing a relation between an output load Po and the switching frequency f in one of the above embodiments of the quasi-resonant converter. The same relation is also shown for a traditional quasi-resonant converter.
- a curve S 100 denotes the switching frequency f varying according to the output load in one of the above embodiments of the quasi-resonant converter.
- a curve S 10 denotes the switching frequency f varying according to the output load in a traditional quasi-resonant converter.
- the switching frequency f is restricted within the range represented by Eq. (1), when the output load Po changes.
- the switching frequency does not exceed the maximum 1/T B
- the switching frequency does not fall below the minimum 1/(T B +T W ). Constraining the switching frequency to the above predetermined range reduces the switching loss as well.
- the switching frequency gradually rises when the output load Po is reduced in traditional quasi-resonant converters, thereby generating a much higher switching loss.
- the switching loss can be reduced by restricting the switching frequency within a predetermined range through the blocking period and the sensing period regardless of load.
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Abstract
Description
1/(T B +T W)<f<1/T B (1)
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KR1020060052860A KR101165386B1 (en) | 2006-06-13 | 2006-06-13 | Qusi-resonat converter and controlling method thereof |
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Also Published As
Publication number | Publication date |
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KR101165386B1 (en) | 2012-07-12 |
KR20070118751A (en) | 2007-12-18 |
US20070285953A1 (en) | 2007-12-13 |
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