US20200091960A1 - Communication device, spread-spectrum receiver, and related method using normalized matched filter for improving signal-to-noise ratio in harsh environments - Google Patents
Communication device, spread-spectrum receiver, and related method using normalized matched filter for improving signal-to-noise ratio in harsh environments Download PDFInfo
- Publication number
- US20200091960A1 US20200091960A1 US16/134,784 US201816134784A US2020091960A1 US 20200091960 A1 US20200091960 A1 US 20200091960A1 US 201816134784 A US201816134784 A US 201816134784A US 2020091960 A1 US2020091960 A1 US 2020091960A1
- Authority
- US
- United States
- Prior art keywords
- signal
- filter
- normalized
- spread
- receiver
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/709—Correlator structure
- H04B1/7093—Matched filter type
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/7097—Interference-related aspects
- H04B1/71—Interference-related aspects the interference being narrowband interference
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/7097—Interference-related aspects
- H04B1/71—Interference-related aspects the interference being narrowband interference
- H04B1/7102—Interference-related aspects the interference being narrowband interference with transform to frequency domain
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04J—MULTIPLEX COMMUNICATION
- H04J11/00—Orthogonal multiplex systems, e.g. using WALSH codes
- H04J11/0023—Interference mitigation or co-ordination
- H04J11/0066—Interference mitigation or co-ordination of narrowband interference
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03828—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties
- H04L25/03834—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties using pulse shaping
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2626—Arrangements specific to the transmitter only
- H04L27/2627—Modulators
- H04L27/264—Pulse-shaped multi-carrier, i.e. not using rectangular window
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2649—Demodulators
- H04L27/26534—Pulse-shaped multi-carrier, i.e. not using rectangular window
- H04L27/2654—Filtering per subcarrier, e.g. filterbank multicarrier [FBMC]
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/709—Correlator structure
- H04B1/7093—Matched filter type
- H04B2001/70935—Matched filter type using a bank of matched fileters, e.g. Fast Hadamard Transform
Definitions
- Embodiments of the present disclosure relate generally to systems, devices, and methods for wireless communication, in particular, for a spread-spectrum receiver using a normalized matched filter for improving a signal-to-noise ratio in harsh environments.
- FB-MC-SS Filter bank multicarrier spread spectrum
- a method of operating a communication device comprises receiving, at a receiver, an incoming RF signal; demodulating, with a demodulator of the receiver, the incoming RF signal to generate a baseband signal; filtering the baseband signal with a normalized matched filter having filter characteristics matched to a pulse-shaping filter of the transmitter that generated the incoming RF signal; and extracting a received signal from a normalized output generated by the normalized matched filter.
- a spread-spectrum receiver for detecting and decoding a signal comprises a demodulator configured to generate a baseband signal responsive to demodulating an incoming RF signal, a normalized matched filter configured to filter the baseband signal with a normalized matched filter having filter characteristics matched to a pulse-shaping filter of the transmitter that generated the incoming RF signal, and a signal extractor configured to extract a received signal from a normalized output generated by the normalized matched filter.
- a communication device comprises a spread-spectrum receiver configured to suppress interference in individual subcarrier bands of a demodulated Filter-Bank Multi-Carrier Spread-Spectrum (FB-MC-SS) signal responsive to normalizing samples of the demodulated FB-MC-SS signal in the frequency domain to a power of unity.
- FB-MC-SS Filter-Bank Multi-Carrier Spread-Spectrum
- FIG. 1 is a schematic diagram of a wireless communication system according to an embodiment of the disclosure.
- FIG. 2 is a block diagram of a communication device according to an embodiment of the disclosure.
- FIG. 3 is a block diagram of an FB-MC-SS transmitter according to an embodiment of the disclosure.
- FIG. 6 is a plot of the magnitude of an output signal from a standard matched filter at the receiver for a single-path channel.
- FIG. 7 is a plot of the magnitude of a standard matched filter output for a two-path channel.
- FIGS. 8A, 8B, and 8C are plots showing the magnitude of a standard matched filter output when the channel has corrupted the received signal for additive white Gaussian noise (AWGN) at various signal-to-noise (SNR) levels.
- AWGN additive white Gaussian noise
- SNR signal-to-noise
- FIG. 9 shows a plot of the magnitude of the standard matched filter output in a case where the received signal is corrupted in a harsh environment.
- FIGS. 10A, 10B, 10C are plots showing the magnitude of the normalized matched filter output for various harsh environments.
- FIG. 11 is a plot of normalized periodic function ⁇ (t) according to an embodiment of the disclosure.
- FIG. 12 is a plot showing a rectangular pulse characterizing a finite impulse response (FIR) filter with coefficients of 1 and its corresponding frequency response.
- FIR finite impulse response
- FIG. 13 is a plot showing a rectangular pulse with added roll-offs at its two sides characterizing an FIR filter and its corresponding frequency response.
- FIG. 14 is a visual representation of forming a window for a prototype filter according to an embodiment of the disclosure.
- FIGS. 15A, 15B, and 15C present a set of sample plots the magnitude of a prototype filter in the frequency domain.
- FIG. 16 shows a set of rectangular windows that may be applied to an incoming signal to extract partitions.
- FIG. 17 shows a set of rectangular windows with raised-cosine roll-offs at the beginning and the end of them for extracting successive partitions of an incoming signal x[n].
- FIG. 18 is a flowchart illustrating a method of operating the normalized matched filter according to an embodiment of the disclosure.
- FIGS. 19A and 19B are plots demonstrating the interference rejection capability of the normalized matched filter.
- FIG. 20 is a simplified block diagram of a receiver according to an embodiment of the disclosure.
- FIG. 21 is a simplified block diagram of a RAKE receiver according to an embodiment of the disclosure.
- a general-purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine.
- a general-purpose processor may be considered a special-purpose processor while the general-purpose processor executes instructions (e.g., software code) stored on a computer-readable medium.
- a processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.
- embodiments may be described in terms of a process that may be depicted as a flowchart, a flow diagram, a structure diagram, or a block diagram. Although a flowchart may describe operational acts as a sequential process, many of these acts can be performed in another sequence, in parallel, or substantially concurrently. In addition, the order of the acts may be re-arranged.
- a process may correspond to a method, a function, a procedure, a subroutine, a subprogram, etc.
- the methods disclosed herein may be implemented in hardware, software, or both. If implemented in software, the functions may be stored or transmitted as one or more instructions or code on computer-readable media.
- Computer-readable media include both computer storage media and communication media, including any medium that facilitates transfer of a computer program from one place to another.
- any reference to an element herein using a designation such as “first,” “second,” and so forth, does not limit the quantity or order of those elements, unless such limitation is explicitly stated. Rather, these designations may be used herein as a convenient method of distinguishing between two or more elements or instances of an element. Thus, a reference to first and second elements does not mean that only two elements may be employed there or that the first element must precede the second element in some manner.
- a set of elements may comprise one or more elements.
- Embodiments of the disclosure include systems, devices, and methods in which interferences and noise from harsh environments may be suppressed using a normalized matched filter that is matched to a transmitter pulse-shaping filter.
- the matched filter of a conventional receiver may be referred to as a “standard matched filter” to be distinguished from what will be referred to as a “normalized matched filter” according to embodiments of the disclosure.
- FIG. 1 is a schematic diagram of a wireless communication system 100 according to an embodiment of the disclosure.
- the wireless communication system 100 includes a first base station 110 A having a first coverage area 112 A that provides wireless communication services for UEs 120 A 1 , 120 A 2 .
- the wireless communication system 100 also includes a second base station 110 B having a second coverage area 112 B that provides wireless communication services for UEs 120 B 1 , 120 B 2 .
- UEs 120 A 1 , 120 A 2 , UEs 120 B 1 , 120 B 2 may operate in half-duplex mode, and the base stations 110 A, 110 B may operate in the full-duplex (FD) mode with self-interference cancellation capabilities.
- FD full-duplex
- Each of the base stations 110 A, 120 B and/or the UEs 120 A 1 , 120 A 2 , 120 B 1 , 120 B 2 may include components such as one or more processors, transmitters, receivers, memory, etc. configured to enable each device to perform the functions described herein in more detail below.
- FIG. 2 is a block diagram of a communication device 200 according to an embodiment of the disclosure.
- the communication device 200 may be any device suitable for communication on one or more spread-spectrum channels, such as, for example, radios, base stations, wireless routers, user equipment such as computers, tablet devices, smart phones, smart watches, automobile communication devices, internet of things (IoT) devices, and other communication devices.
- the communication device 200 may be configured as both a transmitter and a receiver.
- receiving indicates that a device is configured to receive wireless signals as described herein, but should not be interpreted to mean that the communication device 200 only operates as a receiver.
- transmitter indicates that the device is configured to transmit wireless signals as described herein, but should not be interpreted to mean that the communication device 200 only operates as a transmitter.
- the communication device 200 may be configured to execute software programs containing computing instructions and may include one or more processors 212 , memory 214 , and other special purpose modules 216 .
- the special purpose modules 216 may include circuitry, computing instructions, or a combination thereof. Other functional modules are also depicted. In general, all or a portion of the modules may be configured as circuitry, computing instructions, or a combination thereof.
- Some functions to be performed in connection with embodiments of the present disclosure may need to be performed in real-time or near real-time. As a result, specific circuitry may be utilized for some functions such that the functions are performed fast enough. However, some of these functions may be performed by software executed by the processor 212 .
- the processor 212 may be configured for executing a wide variety of operating systems and applications including the computing instructions for carrying out embodiments of the present disclosure.
- the memory 214 may be used to hold computing instructions, data, and other information for performing a wide variety of tasks including performing embodiments of the present disclosure.
- the communication device 200 may also include a group 250 of key processing modules may include a signal generator 252 , a pulse shaping filter 254 , a matched filter 256 , and a max ratio combiner 258 . Functions of these elements are discussed more fully below.
- the communication device 200 may include a spreader 220 configured to spread information 215 such as a data symbol to be transmitted to a remote device. The spreader 220 may feed signals to a modulator 230 to modulate the signals to one or more carrier frequencies and then sent through the circulator 296 for transmission on the antenna 298 .
- the communication device 200 may include a de-spreader 260 may be used to receive demodulated signals from a demodulator 240 to extract data symbols 219 sent by a remote device.
- FIG. 3 is a block diagram of an FB-MC-SS transmitter 300 according to an embodiment of the disclosure.
- the transmitter 300 may be configured to generate an FB-MC-SS signal having a power level at or below a noise level.
- the noise level is a power level below which other communication signals being transmitted in the frequency spectrum would consider noise that would not have a significant impact on their communication signal.
- the FB-MC-SS signal may be considered an underlay control channel (i.e., “underlay,” meaning it is below the noise level) with a relatively large number of subcarriers over the frequency spectrum.
- the number of subcarriers is generally selected to be a large number and may also be selected to be a power of two (i.e., 2 n ), such as, for example, 128, 2048, and 4096 subcarriers.
- the FB-MC-SS signal includes a data symbol modulated onto each subcarrier. Moreover, the same data symbol is redundantly modulated onto each subcarrier. In other words, each subcarrier of the FB-MC-SS signal includes the same data.
- the data symbol modulation may be as simple as a single bit modulated with binary phase shift keying (BPSK).
- BPSK binary phase shift keying
- more complex data symbol modulation techniques may be used for data symbols containing more data bits.
- modulation techniques such as quadrature phase shift keying (QPSK) and quadrature amplitude modulations (QAM), such as QAM 16, QAM 32, and QAM 64, may be used.
- QPSK quadrature phase shift keying
- QAM quadrature amplitude modulations
- these examples are intended to be non-limiting, and other data symbol modulation techniques are contemplated.
- the more simple modulation schemes may allow for easier data symbol recovery and may be more robust in embodiments transmitting at very low power levels.
- the transmitter pulse-shaping filter g(t) may have an impulse response expressed as:
- ⁇ k is the spreading gain at the kth subcarrier band
- h k (t) is a passband filter over the kth subcarrier band.
- the latter is related to a prototype filter h(t) having an impulse response:
- the prototype filter h(t) may be a low pass filter, such as a square-root Nyquist filter (e.g., square-root raised-cosine).
- the square-root Nyquist filter h(t) may be configured such that the zero-crossings of the Nyquist filter h(t)*h(t) are at the interval T.
- the spreading gain factors ⁇ k are a set of complex numbers that may have a unit amplitude (i.e.,
- 2 1) and phase angles that may be uniformly distributed over the interval 0 to 2 ⁇ .
- the set of passband filters h k (t) may have mutually exclusive (i.e., non-overlapping) bands.
- the input signal s(t) includes a train of data symbols s[n], spaced at interval T, that are passed through the pulse-shaping filter g(t) to generate an output signal x(t) that is passed to the modulator.
- the pulse-shaping filter g(t) comprises a bank of N passband filters h 0 (t), h 1 (t), . . . , h N ⁇ 1 (t), with center frequencies f 0 , f 1 , . . . , f N ⁇ 1 .
- the impulse response of the standard matched filter is given by:
- the combined response of the transmitter pulse-shaping filter and the standard matched filter at the receiver may be:
- equation (5) above may be reduced to:
- the subcarrier frequencies f k are given the values:
- ⁇ (t) is a periodic function with a period of T.
- ⁇ (t) is a periodic function with a period of T.
- the exponential terms under the summation in equation (10) have the phase of zero (or some multiple of 2 ⁇ ).
- the exponential terms constructively add to a peak of N.
- these phases may have a phase of ⁇ (or some multiple of 2 ⁇ plus 7 ⁇ ).
- the summation on the right-hand side of equation (10) will be equal to ⁇ N.
- ⁇ (t) is reduced to a small value.
- FIG. 6 is a plot 600 of the magnitude of an output signal from the standard matched filter (SMF) at the receiver for a single-path channel (e.g., under ideal conditions) when a random sequence of binary symbols is passed through the system and there is an ideal channel between the transmitter and receiver.
- SMF standard matched filter
- FIG. 7 is a plot 700 of the magnitude of the standard matched filter (SMF) output for a two-path channel (e.g., under ideal conditions).
- SMF standard matched filter
- the presence of a multipath channel leads to a cluster of pulses instead of each of the pulses in FIG. 6 .
- Any algorithm that can identify the presence of these regular/periodic clusters of pulses, such as the pulses shown in FIG. 7 may be used to identify the presence of an incoming FB-MC-SS data packet.
- the position of these clusters provides information about a timing phase for a fair recovery of the data symbols.
- FIGS. 8A, 8B, and 8C are plots 800 A, 800 B, 800 C that present three cases where the received signal is corrupted by additive white Gaussian noise (AWGN) at signal-to-noise ratio (SNR) values of 0 dB, ⁇ 10 dB, and ⁇ 20 dB, respectively.
- AWGN additive white Gaussian noise
- SNR signal-to-noise ratio
- the regular pulses at the standard matched filter output become less observable (e.g., will be masked out) as the SNR decreases, and naturally the pulses become essentially undetectable as the SNR drops below a certain value.
- the SNR value at which the pulses become undetectable may be lowered by increasing the processing gain of FB-MC-SS (e.g., equivalently, by increasing the number of active subcarriers N).
- plot 800 A shows the magnitude of the standard matched filter output when the channel has corrupted the received signal by AWGN at SNR level of 0 dB.
- FIG. 8B is a plot 800 B showing the magnitude of the standard matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of ⁇ 10 dB.
- FIG. 8C is a plot 800 C showing the magnitude of the standard matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of ⁇ 20 dB.
- the AWGN at the channel output may be at a level that does not mask out the regular pulses at the standard matched filter output, but there is a strong partial-band interference at a considerably higher level that may completely mask out the regular pulses at the standard matched filter output.
- This scenario may be more likely to happen in a harsh environment such as where a jammer may introduce some narrow/partial band interference, or in a case where the FB-MC-SS is used as an underlay control channel in a cognitive radio system.
- overlay communications may introduce some high level partial band interference.
- the communication signals may be hidden within the noise such that the receiver may not be able to distinguish them.
- FIG. 9 shows a plot 900 of the magnitude of the standard matched filter output in a case where the received signal is corrupted in a harsh environment, such as by AWGN at 0 dB SNR and a partial-band interference over 50% of the transmission band at 20 dB above the signal level.
- a harsh environment such as by AWGN at 0 dB SNR and a partial-band interference over 50% of the transmission band at 20 dB above the signal level.
- the regular pulses at the standard matched filter output are masked out by the partial band interference.
- partial band interference i.e., at the presence of AWGN with 0 dB SNR
- the standard matched filter may not provide a discernable output for such harsh environments.
- Embodiments of the disclosure may provide an improvement over the standard matched filter in this scenario, which may be referred to as the normalized matched filter.
- the normalized matched filter may be configured to suppress the partial band interference and cause the regular pulses at the output of the normalized matched filter more clearly observable.
- the normalized matched filter is described more fully below, and is based on the following understanding of the standard matched filter and the mechanism that leads to the regular pulses at its output.
- the mathematical equations (5) through (10) provide some details that explain how these regular pulses are generated.
- each subcarrier has a contribution to one of the complex exponentials that together produce the periodic signal ⁇ (t) that is presented by equation (10).
- the sample plot of ⁇ (t) presented in FIG. 4 and that the positive pulses in FIG. 4 correspond to the time points where all the exponential terms in equation (10) have the phase of zero (or some multiples of 2 ⁇ ).
- the interference suppression of the normalized matched filter is implemented in the frequency domain.
- any frequency domain signal processing may be adopted, the description focuses on the method of overlap-add signal processing.
- the normalized matched filter will be described below, after a brief review of the overlap-add method.
- x[n] is partitioned into adjacent segments of length P.
- the segments are called x 1 [n], x 2 [n], . . . .
- g[n] may be zero-padded to the length L ⁇ P+M ⁇ 1.
- the fast Fourier transform (FFT) of this zero-padded version of g[n] is a sequence of length L and will be named G[k] herein.
- G[k] the fast Fourier transform
- the output y[n] may be formed by time-aligning the segments y 1 [n], y 2 [n], . . . to the positions that match those of the input segments x 1 [n], x 2 [n], . . . and adding the results.
- This overlap-add procedure is a linear filtering operation.
- Embodiments of the disclosure include a normalization process added to the matched filtering operation to form a normalized matched filter configured to reject partial band interference from the input signal.
- the generated frequency domain samples X i [k] provide the information about the spectral content of the time-domain signal segment x i [n].
- the FFT result X i [k] may also be divided into a number of partitions, with each partition covering one of the subcarrier bands in the FB-MC-SS signal.
- the subcarrier bands that are interfered with a strong interference may be characterized by a subset of samples of X i [k] with large amplitude.
- the corresponding samples of X i [k] may be normalized to a power of unity. Assuming that a given subcarrier band is characterized by the frequency index set k 1 ⁇ k ⁇ k 2 , the set of normalized samples are obtained as:
- the portions of the signal spectral may be attenuated that belong to the subcarrier bands that are interfered with a stronger interference. As a result, the interference may also be attenuated proportionally.
- a normalized matched filter may result in an improvement in comparison with a standard matched filter.
- This modified overlap-add method is a non-limiting example of a particular implementation of the normalized matched filter.
- Alternative methods are contemplated and would be apparent to those skilled in the art of signal processing.
- an alternative normalized matched filter may be utilized based on the overlap-save method.
- FIGS. 10A, 10B, 10C are plots 1000 A, 1000 B, 1000 C showing the magnitude of the normalized matched filter output for various harsh environments.
- the received signal is an FB-MC-SS signal that is corrupted by an AWGN at SNR of 0 dB, plus an interference signal that occupies 50% of the transmission band at levels 20 dB, 30 dB, and 50 dB above the signal level, for FIGS. 10A, 10B, and 10C , respectively.
- the prototype filter for each of FIGS. 10A, 10B, and 10C is a rectangular window.
- the normalized matched filter may remove most of the interference from its output. As a result, pulses may be observed and recoverable in most of the scenarios.
- the exception is the case where interference has a very high power in which the interference may not be sufficiently removed (e.g., the case presented in FIG. 10C ).
- Equation (5) describes the combined impulse response between the transmitter input and the standard matched filter output, assuming an ideal channel.
- equation (12) can also be simplified to the same form as equation (7), with:
- FIG. 11 is a plot 1100 of normalized periodic function ⁇ (t) according to an embodiment of the disclosure.
- the normalization coefficients A k are a set of random numbers uniformly distributed over the interval 0 to 1, and the parameter N is equal to 50.
- prototype filter p[n] is a rectangular window.
- This rectangular prototype filter p[n] may have a frequency response that includes a sinc pulse. Such a sinc pulse whose magnitude response is presented in FIG. 12 may have a poor stopband attenuation.
- the sinc pulse may pick up a significant amount of signal power from all parts of the spectrum. Accordingly, when a portion of the spectrum is corrupted by a high-power interference, this interference can impact (e.g., leak to) all samples of X i [k] and as a result may mask out the regular pulses at the output of the normalized matched filter.
- FIGS. 10A, 10B, and 10C give some idea of the impact of the rectangular prototype filter as interference power level increases.
- FIG. 12 is a plot 1200 showing a rectangular pulse 1202 characterizing an FIR filter with coefficients of 1 and its corresponding frequency response 1204 .
- the frequency response 1204 of a rectangular window in the frequency domain may be undesirable as a direct consequence of the sharp edges at the beginning and the end of the rectangular pulse 1202 for the prototype filter p[n].
- FIG. 13 is a plot 1300 showing a rectangular pulse 1302 with added roll-offs 1306 , 1308 at its two sides characterizing an FIR filter and its corresponding frequency response 1304 .
- the frequency response 1304 may be improved relative to FIG. 12 , and as a result, the leakage of a high-power interference to other parts of the transmission band may be reduced.
- the addition of smooth roll-offs 1306 , 1308 to the beginning and the end of a rectangular pulse 1302 that may improve on its frequency domain properties have been adopted in different applications of signal processing.
- orthogonal frequency division multiplexing OFDM
- roll-offs are added to the beginning and the end of each OFDM symbol to improve on the subcarriers out-of-band emissions, so that the signals transmitted from different user equipment nodes in the network minimally interfere with one another.
- smooth roll-offs 1306 , 1308 are added to the prototype filter p[n] to improve on the interference rejection of the normalized matched filter.
- FIG. 12 presented a rectangular pulse 1202 as an FIR filter and its frequency response 1204 .
- FIG. 13 presents a similar rectangular pulse 1302 as an FIR filter after adding raised-cosine roll-offs 1306 , 1308 to the beginning and the end of it and its frequency response, respectively.
- the width of roll-offs in FIG. 13 may be approximately 5% of the width of the rectangular window. Even this small amount of roll-off may contribute to improving the stop-band attenuation of the respective frequency response (compare FIGS. 12 and 13 ) and, as a result, as will be shown through an example later, in reducing the interference leakage in the context of this invention (see description of FIGS. 19 a and 19 b , below).
- FIG. 14 is a visual representation of forming a window 1406 for a prototype filter p(t) with raise-cosine roll-offs at the beginning and the end of the window 1406 through convolution of a rectangular pulse 1402 and a half sine-wave 1404 .
- the visual representation of FIG. 14 may be expressed mathematically as:
- V ⁇ ( f ) T w ⁇ sin ⁇ ⁇ c ⁇ ( fT w ) ⁇ V ⁇ ( f ) ⁇ ⁇
- V ⁇ ( f ) cos ⁇ ( ⁇ ⁇ ⁇ fT v ) 1 - 4 ⁇ f 2 ⁇ T v 2 ( 18 )
- FIGS. 15A, 15B, and 15C present a set of sample plots 1500 A, 1500 B, 1500 C the magnitude of a prototype filter in the frequency domain.
- plot 1500 A shows
- plot 1500 B shows
- FIG. 16 shows
- the prototype filter P(f) is a low-pass filter characterized by a narrow-band main lobe, a number of significant side lobes that expand over a bandwidth of
- transition band is ⁇ 50 dB or less, its transition band may exhibit a width of about
- the windows may be non-overlapping, but, in order not to miss any samples of x[n], the windows may be adjacent to each other.
- FIG. 17 is a set 1700 of sampled rectangular windows w 1 [n], w 2 [n], w 3 [n], that include the added roll-offs at the beginning and the end of windows.
- the rectangular windows w 1 [n], w 2 [n], w 3 [n], . . . must be chosen such that:
- w 0 [n] i.e., the window function of the 0 th partition
- w 0 [n] is effectively the prototype filter p[n].
- Q denotes the number of samples within each of the roll-off intervals. Therefore, recalling the window arrangements in FIG. 17 , one finds that:
- FIG. 18 is a flowchart 1800 illustrating a method of operating the normalized matched filter according to an embodiment of the disclosure.
- the partition index i may first be initialized.
- the portion of the incoming signal that corresponds to the present partition index is taken and the relevant window is applied. The result is the length P sequence x i [n].
- x i [n] is extended to the length L>P by appending zeros to its end and an FFT is applied to the result to convert it to the frequency domain.
- This frequency domain signal is called X i [k].
- the FFT operation may be synthesis filter bank with a rectangular prototype filter.
- the rectangular prototype filter may be replaced with a filter with lower out-of-band emissions.
- segments of X i [k] that correspond to different subcarrier bands of the FB-MC-SS waveform are normalized according to the equation (11). The result is called ⁇ tilde over (X) ⁇ i [k].
- the frequency domain matched filtered sequence ⁇ tilde over (Y) ⁇ i [k] is converted to the time domain through an IFFT operation and the result is overlapped and added to the previous partition of the output (in the time domain).
- the partition index i is incremented by one and operations 1804 through 1812 may be repeated. These operations may be repeated as long as necessary to obtain the output samples of the normalized matched filter that may be used in any packet detection and timing acquisition algorithm for finding the beginning of an FB-MC-SS data packet and obtaining the correct timing phase.
- FIGS. 19A and 19B are plots 1900 A, 1900 B demonstrating the interference rejection capability of the normalized matched filter.
- the results in each plot 1900 A, 1900 B are presented for the scenario a raised-cosine window with 5% roll-offs at both sides is applied to the partitions of the incoming signal.
- the added raised-cosine roll-offs improves the interference suppression capability of the normalized matched filter, as apparent by the comparison below.
- FIG. 19A shows the magnitude of the normalized matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of 0 dB and partial band interference at 50 dB above the signal level.
- FIG. 19A is a comparable result to that of FIG. 10C , but with the roll-offs applied to the partitions of the incoming signal in FIG. 19A but not in FIG. 10C .
- the normalized matched filter is not as effective in rejecting an interference that is 50 dB above the signal level in comparison to that of FIG. 19A .
- the results in FIG. 19A show that with the added roll-offs to the window, the normalized matched filter is more effective in rejecting the interference.
- the synchronization pulses are more apparent at the output shown in FIG. 19A compared with FIG. 10C .
- FIG. 19B shows the magnitude of the normalized matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of 0 dB and partial band interference at 80 dB above the signal level.
- Analog amplifiers with linear performance over a dynamic range of 80 dB are not easy to implement according to conventional methods. Also, analog to digital convertor circuitries with typical 11 or 12 effective bits can cover a linear dynamic range of 60 to 70 dB, only. The normalized matched filter, is therefore, an improvement over conventional systems.
- FIG. 20 is a simplified block diagram of a receiver 2000 according to an embodiment of the disclosure.
- the receiver 2000 is configured to detect and decode in incoming RF signal 2005 (e.g., an FB-MC-SS signal).
- the receiver 2000 includes a demodulator 2010 , normalized matched filter 2020 , carrier recovery unit 2030 , timing recovery unit 2040 , analysis filter bank 2050 , channel estimator 2060 , and maximum-ratio combining unit 2070 .
- the demodulator 2010 receives the RF signal 2005 , and converts the RF signal 2005 to a baseband input 2015 .
- the baseband input 2015 is passed to the analysis filter bank 2050 and the normalized matched filter 2020 .
- the normalized matched filter 2020 is configured to have filter characteristics matched to those of the synthesis filter bank in the transmitter and generates a normalized output as discussed above.
- the output from the normalized matched filter 2020 may be the sequence of high amplitude (i.e., narrow) pulses that can be clearly identified even when the FB-MC-SS signal is at a level below the noise level.
- the combination of the transmit and receive processes may result in a significant and unexpected improvement (both computationally faster and more precise) in timing recovery and frequency lock-in.
- a sequence of narrow pulses 2025 from the normalized matched filter 2020 may occur at regular intervals and carry the necessary information for timing and carrier acquisition as well as for their tracking.
- any carrier and timing recovery method takes advantage of the properties of the modulation method that has been used to construct the transmitted signal. Due to the implementation inaccuracies and channel conditions (e.g., accuracy of components and Doppler shift in the channel), the carrier frequency used at the transmitter and its counterpart frequency at the receiver 2000 are bound to suffer mismatch.
- the carrier recovery unit 2030 finds such mismatches and compensates for the mismatches at the receiver 2000 .
- the normalized matched filter 2020 may be used to develop relatively efficient carrier and timing recovery algorithms.
- the output of the normalized matched filter 2020 includes the sequence of narrow pulses 2025 at the 1 ⁇ 2 spacing of the symbol timing (T/2). These narrow pulses 2025 are present at the time instants that are even multiples of T/2 and correspond to about the center of the data symbols.
- the presence of such pulses depends on the information bits transmitted before and after the time instant.
- the pulse at the odd multiples of T/2 will appear when two consecutive symbols have the same value, otherwise the pulse would be a null.
- This property of the normalized matched filter 2020 enables development of a timing recovery algorithm with a relatively low complexity/processing power.
- the carrier recovery unit 2030 may receive the sequence of narrow pulses 2025 and generate control signals 2035 to develop a phase-locked loop (PLL) between the demodulator 2010 , the normalized matched filter 2020 , and the carrier recovery unit 2030 as is known by those skilled in the art.
- PLL phase-locked loop
- the phase of the baseband input 2015 is known relative to the sequence of narrow pulses 2025 from the normalized matched filter 2020 .
- the timing recovery unit 2040 may use the sequence of narrow pulses 2025 as a synchronization signal to ensure that the demodulated and combined signals from all subcarriers (i.e., the baseband input 2015 ) can be sampled at correct times by the analysis filter bank 2050 .
- the sequence of narrow pulses 2025 containing the timing information and the baseband input 2015 from the demodulator 2010 are passed to the analysis filter bank 2050 .
- the analysis filter bank 2050 extracts the received signal of each of the various subcarriers in parallel and samples them at a proper timing phase based on the timing information.
- the analysis filter bank 2050 performs the inverse process of the synthesis filter bank of the transmitter.
- the output of the analysis filter bank 2050 is a set of signals 2055 corresponding to each of the subcarrier frequencies in the FB-MC-SS signal. Due to the very low power level of the received signal, some of these signals in the set of signals 2055 may not carry accurate information. However, many of the signals in the set of signals 2055 will carry the same information that was transmitted.
- the channel estimator 2060 examines the set of signals 2055 corresponding to each of the subcarrier frequencies to estimate a channel gain and a power of noise plus interference at each of the subcarrier bands by computing the impulse response of each channel in both time and frequency domain.
- Digital Signal Processing (DSP) and Field-Programmable Gate Arrays (FPGAs) may allow for one example of an inexpensive implementation for many of the functions in the receiver 2000 and the channel estimator 2060 in particular.
- the maximum-ratio combining unit 2070 receives the set of signals 2055 from the analysis filter bank 2050 and information 2065 at the output of the channel estimator 2060 to obtain an estimate of the transmitted data symbols.
- the maximum-ratio combining unit 2070 calculates a weighted average of the detected data symbols at the output of the analysis filter bank 2050 . The less noisy outputs are given higher weights and the more noisy outputs are given lower weights.
- the maximum-ratio combining unit 2070 may divide a power estimate of each channel by the mean square of the estimated noise/interference, which may result in either a diminished noise where the signal is stronger or elimination of the certain subcarriers where the noise/interference is higher than the mean noise.
- FIG. 21 is a simplified block diagram of a RAKE receiver 2100 according to an embodiment of the disclosure.
- RAKE receivers are commonly used in direct sequence spread spectrum (DS-SS) systems to reduce the effects of the signal fading due to multipath.
- DS-SS direct sequence spread spectrum
- the common implementation makes use of an analysis filter bank (e.g., analysis filter bank 2010 in FIG. 20 ) to separate portions of the received signal that belong to different subcarrier frequency bands. The maximum ratio combining is then applied to the signal samples at the outputs of the analysis filter bank. By doing so, the subcarrier bands that are corrupted by interference are given lower gains and in this way a receiver with a near optimum performance is constructed.
- Such FB-MC-SS systems are known to have advantage over DS-SS in rejecting partial band interference.
- the RAKE receiver 2100 includes a normalized matched filter 2120 configured as described above with respect to FIG. 18 .
- the normalized matched filter 2120 may also be configured to remove partial band interference without resorting to the use of an analysis filter bank. As a result, the RAKE receiver 2100 may exhibit some performance loss when compared to the receiver 2000 of FIG. 20 , but may have the benefit of having lower complexity.
- the RAKE receiver 2100 makes use of the samples of the output of the normalized matched filter 2120 that correspond to different multi-paths of the channel. These multi-paths are combined together using a maximum ratio combiner.
- the maximum ratio combiner 2170 gives higher gains to the stronger paths and lower gains to weaker paths. In this way, an optimum receiver that maximizes the SNR at its output (detected symbols) is constructed.
- embodiments of the disclosure improve over conventional partial band interference suppression techniques known to the inventors that rely on the use of notch/bandstop filters which require the knowledge or estimation of the position of interference within the transmission band of the spread spectrum signal.
- Embodiments of the disclosure may be configured to suppress the interference without knowing the position of interference by applying a non-linear process to the analyzed signal in the frequency domain.
Landscapes
- Engineering & Computer Science (AREA)
- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Physics & Mathematics (AREA)
- Spectroscopy & Molecular Physics (AREA)
- Power Engineering (AREA)
- Noise Elimination (AREA)
Abstract
Description
- This application is related to U.S. Pat. No. 8,731,027, filed Dec. 5, 2011, issued May 20, 2014; U.S. Pat. No. 8,861,571, filed Dec. 5, 2011, issued Oct. 14, 2014; U.S. Pat. No. 9,215,587, filed Jan. 24, 2014, issued Dec. 15, 2015; U.S. Pat. No. 9,369,866, filed Sep. 26, 2014, issued Jun. 14, 2016; and U.S. Pat. No. 9,559,748, filed Mar. 4, 2016, and issued Jan. 31, 2017, the disclosure of each of which is hereby incorporated herein in its entirety by this reference.
- This invention was made with government support under Contract No. DE-AC07-05-ID14517 awarded by the United States Department of Energy. The government has certain rights in the invention.
- Embodiments of the present disclosure relate generally to systems, devices, and methods for wireless communication, in particular, for a spread-spectrum receiver using a normalized matched filter for improving a signal-to-noise ratio in harsh environments.
- Filter bank multicarrier spread spectrum (FB-MC-SS) methods have previously been used for communication in harsh environments. In FB-MC-SS methods, each data symbol may be spread over N spectrally disjoint subcarrier frequency bands before transmission. At the receiver, a maximum ratio combiner (MRC) may combine the demodulated signals from different subcarrier bands to increase the signal-to-interference-plus-noise ratio (SINR) at the combiner output.
- Assuming the receiver has already identified the presence of a data packet and has obtained the timing phase of the middle point of the received data symbols, methods have been adopted to measure the SINR at each subcarrier band and accordingly calculate the MRC coefficients. For example, method of performing these tasks has been described Haddidin et al., An underlay communication channel for 5G cognitive mesh networks: Packet design, implementation, analysis, and experimental results, IEEE International Conference on Communications (ICC) (2016). The receiver structure presented by Haddidin et al. can operate at some negative SINR, but may fail if part of the frequency band is corrupted by a high level of interference. Under this condition, the receiver may fail to detect the packet and/or identify the symbol timing phase, hence cannot operate successfully. In other words, in harsh environments, packet detection and symbol timing phase estimation are receiver bottle-necks that have not been addressed in prior solutions known to the inventors. Thus, the inventors have identified a need to address this limitation of FB-MC-SS communication methods.
- In some embodiments, a method of operating a communication device is disclosed. The method comprises receiving, at a receiver, an incoming RF signal; demodulating, with a demodulator of the receiver, the incoming RF signal to generate a baseband signal; filtering the baseband signal with a normalized matched filter having filter characteristics matched to a pulse-shaping filter of the transmitter that generated the incoming RF signal; and extracting a received signal from a normalized output generated by the normalized matched filter.
- In some embodiments, a spread-spectrum receiver for detecting and decoding a signal comprises a demodulator configured to generate a baseband signal responsive to demodulating an incoming RF signal, a normalized matched filter configured to filter the baseband signal with a normalized matched filter having filter characteristics matched to a pulse-shaping filter of the transmitter that generated the incoming RF signal, and a signal extractor configured to extract a received signal from a normalized output generated by the normalized matched filter.
- In some embodiments, a communication device comprises a spread-spectrum receiver configured to suppress interference in individual subcarrier bands of a demodulated Filter-Bank Multi-Carrier Spread-Spectrum (FB-MC-SS) signal responsive to normalizing samples of the demodulated FB-MC-SS signal in the frequency domain to a power of unity.
-
FIG. 1 is a schematic diagram of a wireless communication system according to an embodiment of the disclosure. -
FIG. 2 is a block diagram of a communication device according to an embodiment of the disclosure. -
FIG. 3 is a block diagram of an FB-MC-SS transmitter according to an embodiment of the disclosure. -
FIG. 4 is a plot of β(t) when N=50 that is plotted over t/T. -
FIG. 5 is a plot of η(t) when N=50 that is plotted over t/T. -
FIG. 6 is a plot of the magnitude of an output signal from a standard matched filter at the receiver for a single-path channel. -
FIG. 7 is a plot of the magnitude of a standard matched filter output for a two-path channel. -
FIGS. 8A, 8B, and 8C are plots showing the magnitude of a standard matched filter output when the channel has corrupted the received signal for additive white Gaussian noise (AWGN) at various signal-to-noise (SNR) levels. -
FIG. 9 shows a plot of the magnitude of the standard matched filter output in a case where the received signal is corrupted in a harsh environment. -
FIGS. 10A, 10B, 10C are plots showing the magnitude of the normalized matched filter output for various harsh environments. -
FIG. 11 is a plot of normalized periodic function β(t) according to an embodiment of the disclosure. -
FIG. 12 is a plot showing a rectangular pulse characterizing a finite impulse response (FIR) filter with coefficients of 1 and its corresponding frequency response. -
FIG. 13 is a plot showing a rectangular pulse with added roll-offs at its two sides characterizing an FIR filter and its corresponding frequency response. -
FIG. 14 is a visual representation of forming a window for a prototype filter according to an embodiment of the disclosure. -
FIGS. 15A, 15B, and 15C present a set of sample plots the magnitude of a prototype filter in the frequency domain. -
FIG. 16 shows a set of rectangular windows that may be applied to an incoming signal to extract partitions. -
FIG. 17 shows a set of rectangular windows with raised-cosine roll-offs at the beginning and the end of them for extracting successive partitions of an incoming signal x[n]. -
FIG. 18 is a flowchart illustrating a method of operating the normalized matched filter according to an embodiment of the disclosure. -
FIGS. 19A and 19B are plots demonstrating the interference rejection capability of the normalized matched filter. -
FIG. 20 is a simplified block diagram of a receiver according to an embodiment of the disclosure. -
FIG. 21 is a simplified block diagram of a RAKE receiver according to an embodiment of the disclosure. - In the following description, reference is made to the accompanying drawings in which are shown, by way of illustration, specific embodiments in which the disclosure may be practiced. The embodiments are intended to describe aspects of the disclosure in sufficient detail to enable those skilled in the art to make, use, and otherwise practice the disclosure. Furthermore, specific implementations shown and described are only examples and should not be construed as the only way to implement the present disclosure unless specified otherwise herein. It will be readily apparent to one of ordinary skill in the art that the various embodiments of the present disclosure may be practiced by numerous other partitioning solutions. Other embodiments may be utilized and changes may be made to the disclosed embodiments without departing from the scope of the disclosure. The following detailed description is not to be taken in a limiting sense, and the scope of the present invention is defined only by the appended claims.
- In the following description, elements, circuits, and functions may be shown in block diagram form in order not to obscure the present disclosure in unnecessary detail. Conversely, specific implementations shown and described are exemplary only and should not be construed as the only way to implement the present disclosure unless specified otherwise herein. Additionally, block definitions and partitioning of logic between various blocks is exemplary of a specific implementation. It will be readily apparent to one of ordinary skill in the art that the present disclosure may be practiced by numerous other partitioning solutions. For the most part, details concerning timing considerations and the like have been omitted where such details are not necessary to obtain a complete understanding of the present disclosure and are within the abilities of persons of ordinary skill in the relevant art.
- Those of ordinary skill in the art would understand that information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof. Some drawings may illustrate signals as a single signal for clarity of presentation and description. It will be understood by a person of ordinary skill in the art that the signal may represent a bus of signals, wherein the bus may have a variety of bit widths, and the present disclosure may be implemented on any number of data signals including a single data signal.
- The various illustrative logical blocks, modules, and circuits described in connection with the embodiments disclosed herein may be implemented or performed with a general purpose processor, a special purpose processor, a Digital Signal Processor (DSP), an Application Specific Integrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general-purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A general-purpose processor may be considered a special-purpose processor while the general-purpose processor executes instructions (e.g., software code) stored on a computer-readable medium. A processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.
- Also, it is noted that embodiments may be described in terms of a process that may be depicted as a flowchart, a flow diagram, a structure diagram, or a block diagram. Although a flowchart may describe operational acts as a sequential process, many of these acts can be performed in another sequence, in parallel, or substantially concurrently. In addition, the order of the acts may be re-arranged. A process may correspond to a method, a function, a procedure, a subroutine, a subprogram, etc. Furthermore, the methods disclosed herein may be implemented in hardware, software, or both. If implemented in software, the functions may be stored or transmitted as one or more instructions or code on computer-readable media. Computer-readable media include both computer storage media and communication media, including any medium that facilitates transfer of a computer program from one place to another.
- It should be understood that any reference to an element herein using a designation such as “first,” “second,” and so forth, does not limit the quantity or order of those elements, unless such limitation is explicitly stated. Rather, these designations may be used herein as a convenient method of distinguishing between two or more elements or instances of an element. Thus, a reference to first and second elements does not mean that only two elements may be employed there or that the first element must precede the second element in some manner. In addition, unless stated otherwise, a set of elements may comprise one or more elements.
- Embodiments of the disclosure include systems, devices, and methods in which interferences and noise from harsh environments may be suppressed using a normalized matched filter that is matched to a transmitter pulse-shaping filter. Throughout the disclosure, the matched filter of a conventional receiver may be referred to as a “standard matched filter” to be distinguished from what will be referred to as a “normalized matched filter” according to embodiments of the disclosure.
-
FIG. 1 is a schematic diagram of awireless communication system 100 according to an embodiment of the disclosure. Thewireless communication system 100 includes afirst base station 110A having afirst coverage area 112A that provides wireless communication services forUEs wireless communication system 100 also includes asecond base station 110B having asecond coverage area 112B that provides wireless communication services forUEs UEs UEs base stations base stations UEs -
FIG. 2 is a block diagram of acommunication device 200 according to an embodiment of the disclosure. Thecommunication device 200 may be any device suitable for communication on one or more spread-spectrum channels, such as, for example, radios, base stations, wireless routers, user equipment such as computers, tablet devices, smart phones, smart watches, automobile communication devices, internet of things (IoT) devices, and other communication devices. Thecommunication device 200 may be configured as both a transmitter and a receiver. Thus, using the term “receiver” indicates that a device is configured to receive wireless signals as described herein, but should not be interpreted to mean that thecommunication device 200 only operates as a receiver. Similarly, using the term “transmitter” indicates that the device is configured to transmit wireless signals as described herein, but should not be interpreted to mean that thecommunication device 200 only operates as a transmitter. - The
communication device 200 may be configured to execute software programs containing computing instructions and may include one ormore processors 212,memory 214, and otherspecial purpose modules 216. Thespecial purpose modules 216 may include circuitry, computing instructions, or a combination thereof. Other functional modules are also depicted. In general, all or a portion of the modules may be configured as circuitry, computing instructions, or a combination thereof. Some functions to be performed in connection with embodiments of the present disclosure may need to be performed in real-time or near real-time. As a result, specific circuitry may be utilized for some functions such that the functions are performed fast enough. However, some of these functions may be performed by software executed by theprocessor 212. Thus, theprocessor 212 may be configured for executing a wide variety of operating systems and applications including the computing instructions for carrying out embodiments of the present disclosure. Thememory 214 may be used to hold computing instructions, data, and other information for performing a wide variety of tasks including performing embodiments of the present disclosure. - The
communication device 200 may also include agroup 250 of key processing modules may include asignal generator 252, apulse shaping filter 254, a matchedfilter 256, and amax ratio combiner 258. Functions of these elements are discussed more fully below. On the transmitter side, thecommunication device 200 may include aspreader 220 configured to spreadinformation 215 such as a data symbol to be transmitted to a remote device. Thespreader 220 may feed signals to amodulator 230 to modulate the signals to one or more carrier frequencies and then sent through thecirculator 296 for transmission on theantenna 298. On the receiver side, thecommunication device 200 may include a de-spreader 260 may be used to receive demodulated signals from ademodulator 240 to extractdata symbols 219 sent by a remote device. -
FIG. 3 is a block diagram of an FB-MC-SS transmitter 300 according to an embodiment of the disclosure. Thetransmitter 300 may be configured to generate an FB-MC-SS signal having a power level at or below a noise level. The noise level is a power level below which other communication signals being transmitted in the frequency spectrum would consider noise that would not have a significant impact on their communication signal. The FB-MC-SS signal may be considered an underlay control channel (i.e., “underlay,” meaning it is below the noise level) with a relatively large number of subcarriers over the frequency spectrum. The number of subcarriers is generally selected to be a large number and may also be selected to be a power of two (i.e., 2n), such as, for example, 128, 2048, and 4096 subcarriers. - The FB-MC-SS signal includes a data symbol modulated onto each subcarrier. Moreover, the same data symbol is redundantly modulated onto each subcarrier. In other words, each subcarrier of the FB-MC-SS signal includes the same data. As a non-limiting example, the data symbol modulation may be as simple as a single bit modulated with binary phase shift keying (BPSK). In addition, more complex data symbol modulation techniques may be used for data symbols containing more data bits. For example, and without limitation, modulation techniques, such as quadrature phase shift keying (QPSK) and quadrature amplitude modulations (QAM), such as QAM 16, QAM 32, and QAM 64, may be used. Of course, these examples are intended to be non-limiting, and other data symbol modulation techniques are contemplated. Generally, the more simple modulation schemes may allow for easier data symbol recovery and may be more robust in embodiments transmitting at very low power levels.
- The transmitter pulse-shaping filter g(t) may have an impulse response expressed as:
-
- where γk is the spreading gain at the kth subcarrier band, and hk(t) is a passband filter over the kth subcarrier band. The latter is related to a prototype filter h(t) having an impulse response:
-
h k(t)=h(t)e j2πfk t (2) - where fk is the center frequency of the kth subcarrier band. In some embodiments, the prototype filter h(t) may be a low pass filter, such as a square-root Nyquist filter (e.g., square-root raised-cosine). The square-root Nyquist filter h(t) may be configured such that the zero-crossings of the Nyquist filter h(t)*h(t) are at the interval T. The square-root Nyquist filter may include a roll-off factor α=1 for a receiver with a desired suppression of out-of-band spectrum. Other choices of the roll-off factor (α) may be used, which may result in some modification to the receiver design. The spreading gain factors γk are a set of complex numbers that may have a unit amplitude (i.e., |γk|2=1) and phase angles that may be uniformly distributed over the
interval 0 to 2π. In some embodiments, the set of passband filters hk(t) may have mutually exclusive (i.e., non-overlapping) bands. - Referring specifically to
FIG. 3 , the input signal s(t) includes a train of data symbols s[n], spaced at interval T, that are passed through the pulse-shaping filter g(t) to generate an output signal x(t) that is passed to the modulator. According to equation (1) the pulse-shaping filter g(t) comprises a bank of N passband filters h0(t), h1(t), . . . , hN−1(t), with center frequencies f0, f1, . . . , fN−1. Considering equation (1), the impulse response of the standard matched filter is given by: -
- Moreover, considering equation (2) and mat in practical designs h(−t)=h(t) (e.g., ignoring the causality of the impulse response) and hk*(−t)=hk(t). Hence:
-
- With equations (1), (2), and (4), the combined response of the transmitter pulse-shaping filter and the standard matched filter at the receiver may be:
-
- where * denotes linear convolution.
- By design the passband filters hk(t)=h(t)ej2πf
k t and hl(t)=h(t)ej2πfl t may be considered as a pair of bandpass filters. Such a pair of passband filters may either have a common band (e.g., when fk=fl) or cover non-overlapping bands. In the latter case hk(t)=h(t)ej2πfk t*hk(t)=h(t)ej2πfl t=0. Hence, equation (5) above may be reduced to: -
- Manipulating equation (6), and recalling that |γk|2=1, yields:
-
η(t)=β(t)ρ(t) (7) - where ρ(t)=h(t)*h(t), by design, is a Nyquist filter, and
-
- In FB-MC-SS, the subcarrier frequencies fk are given the values:
-
- where T is the time interval between the data symbols s[n]. Substituting these frequency values in equation (8) yields:
-
- Thus, β(t) is a periodic function with a period of T.
-
FIG. 4 is aplot 400 of β(t) when N=50 that is plotted over t/T. As mentioned above, β(t) is a periodic function with a period of T. For t=nT, the exponential terms under the summation in equation (10) have the phase of zero (or some multiple of 2π). Thus, the exponential terms constructively add to a peak of N. For -
- on the other hand, these phases may have a phase of π(or some multiple of 2π plus 7π). Hence, the summation on the right-hand side of equation (10) will be equal to −N. In between these positive and negative peaks, β(t) is reduced to a small value.
-
FIG. 5 is aplot 500 of the combined response η(t) of the transmitter pulse-shaping filter and the standard matched filter when N=50 that is plotted over t/T. Noting that ρ(t) has a main lobe that spans over the range (−T, T) and some small negligible side lobes, the multiplication of β(t) with ρ(t) reduces to effectively three pulses as shown inFIG. 5 . As a result, when the sequence of data symbols s[n] passes through the transmitter pulse-shaping filter g(t) and then through the matched filter at the receiver front-end, the result may be a sequence of persistent positive pulses or negative pulses depending on the sign of the transmitted data symbols s[n]. At the mid-points between these pulses, additional pulses may or may not appear depending of the similarity or dissimilarity of the pair of data symbols at the two sides of these mid-points. -
FIG. 6 is aplot 600 of the magnitude of an output signal from the standard matched filter (SMF) at the receiver for a single-path channel (e.g., under ideal conditions) when a random sequence of binary symbols is passed through the system and there is an ideal channel between the transmitter and receiver. -
FIG. 7 is aplot 700 of the magnitude of the standard matched filter (SMF) output for a two-path channel (e.g., under ideal conditions). The presence of a multipath channel leads to a cluster of pulses instead of each of the pulses inFIG. 6 . Any algorithm that can identify the presence of these regular/periodic clusters of pulses, such as the pulses shown inFIG. 7 , may be used to identify the presence of an incoming FB-MC-SS data packet. Moreover, the position of these clusters provides information about a timing phase for a fair recovery of the data symbols. The article by Sibbett et al., Novel maximum-based timing acquisition for spread-spectrum communications, presented at IEEE Globecom 2016 conference, proposes an algorithm for these tasks; namely, packet detection and timing acquisition. This article by Sibbett et al. is hereby incorporated herein in its entirety by this reference. - In a practical channel, the environment (e.g., noise and interference) adds to the received signal and if noise and/or interference passes a certain level, it can seriously corrupt the received signal and as a result packet detection and timing acquisition may become a difficult task to perform. For example,
FIGS. 8A, 8B, and 8C areplots - Referring specifically to
FIG. 8A ,plot 800A shows the magnitude of the standard matched filter output when the channel has corrupted the received signal by AWGN at SNR level of 0 dB.FIG. 8B is aplot 800B showing the magnitude of the standard matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of −10 dB.FIG. 8C is aplot 800C showing the magnitude of the standard matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of −20 dB. - In another scenario, the AWGN at the channel output may be at a level that does not mask out the regular pulses at the standard matched filter output, but there is a strong partial-band interference at a considerably higher level that may completely mask out the regular pulses at the standard matched filter output. This scenario may be more likely to happen in a harsh environment such as where a jammer may introduce some narrow/partial band interference, or in a case where the FB-MC-SS is used as an underlay control channel in a cognitive radio system. In the latter scenario, overlay communications may introduce some high level partial band interference. Thus, depending on the environment the communication signals may be hidden within the noise such that the receiver may not be able to distinguish them.
-
FIG. 9 shows aplot 900 of the magnitude of the standard matched filter output in a case where the received signal is corrupted in a harsh environment, such as by AWGN at 0 dB SNR and a partial-band interference over 50% of the transmission band at 20 dB above the signal level. In this case, the regular pulses at the standard matched filter output are masked out by the partial band interference. Without partial band interference (i.e., at the presence of AWGN with 0 dB SNR), the result would be the one presented inFIG. 8A . However, as shown inFIG. 9 , the standard matched filter may not provide a discernable output for such harsh environments. - Embodiments of the disclosure may provide an improvement over the standard matched filter in this scenario, which may be referred to as the normalized matched filter. When the incoming signal is corrupted by a partial band interference at a level that otherwise masks out the regular pulses at the standard matched filter output, the normalized matched filter may be configured to suppress the partial band interference and cause the regular pulses at the output of the normalized matched filter more clearly observable.
- The normalized matched filter is described more fully below, and is based on the following understanding of the standard matched filter and the mechanism that leads to the regular pulses at its output. The mathematical equations (5) through (10) provide some details that explain how these regular pulses are generated. In particular, each subcarrier has a contribution to one of the complex exponentials that together produce the periodic signal β(t) that is presented by equation (10). It is also recalled that the sample plot of β(t) presented in
FIG. 4 , and that the positive pulses inFIG. 4 correspond to the time points where all the exponential terms in equation (10) have the phase of zero (or some multiples of 2π). This is a situation where all the terms under summation on the right-hand side of equation (10) are equal to +1 and, thus, add up to a value of N. On the other hand, the negative pulses inFIG. 4 correspond to the time points where all the exponential terms in equation (10) have the phase of π (or some multiples of 2π plus 7π). This is a situation where all the terms under summation on the right-hand side of equation (10) are equal to −1 and, thus, add up to a value of −N. - The interference suppression of the normalized matched filter is implemented in the frequency domain. Although any frequency domain signal processing may be adopted, the description focuses on the method of overlap-add signal processing. The normalized matched filter will be described below, after a brief review of the overlap-add method. The overlap-add method implements the time domain filtering operation y[n]=x[n]*g[n], where x[n] is the input sequence and g[n] is a finite impulse response filter, in the frequency domain, by taking the following operations:
- In a first operation, x[n] is partitioned into adjacent segments of length P. The segments are called x1[n], x2[n], . . . . In a second operation, assuming g[n] has a length of M, g[n] may be zero-padded to the length L≥P+M−1. The fast Fourier transform (FFT) of this zero-padded version of g[n] is a sequence of length L and will be named G[k] herein. In a third operation, for i=1, 2, . . . , repeat the following steps: (a) zero-pad xi[n] to the length of L, and take its FFT with the FFT result being Xi[k], (b) form the frequency domain sequence Yi[k]=Xi[k]G*[k]. This is the matched filtering operation, and (c) convert Yi[k] to the time domain by applying an inverse FFT (IFFT) to it. Call the result yi[n]. In a fourth operation, the output y[n] may be formed by time-aligning the segments y1[n], y2[n], . . . to the positions that match those of the input segments x1[n], x2[n], . . . and adding the results. This overlap-add procedure is a linear filtering operation.
- Embodiments of the disclosure include a normalization process added to the matched filtering operation to form a normalized matched filter configured to reject partial band interference from the input signal. For example, using the example above, in operation (a) of the preceding paragraph, the generated frequency domain samples Xi[k] provide the information about the spectral content of the time-domain signal segment xi[n]. The FFT result Xi[k] may also be divided into a number of partitions, with each partition covering one of the subcarrier bands in the FB-MC-SS signal. The subcarrier bands that are interfered with a strong interference may be characterized by a subset of samples of Xi[k] with large amplitude. To suppress interference in individual subcarrier bands, for each subcarrier band, the corresponding samples of Xi[k] may be normalized to a power of unity. Assuming that a given subcarrier band is characterized by the frequency index set k1≤k≤k2, the set of normalized samples are obtained as:
-
- Repeating this normalization process for all the subcarrier bands, the sample set Xi[k], for k=0, 1, 2, . . . , L−1, is replaced by the corresponding normalized sample set {tilde over (X)}i[k]. Accordingly, after operation (a), the rest of the overlap-add method may continue with the normalized sample set {tilde over (X)}i[k] in place of the original sample set Xi[k]. By normalizing the samples of the overlap-add method, the portions of the signal spectral may be attenuated that belong to the subcarrier bands that are interfered with a stronger interference. As a result, the interference may also be attenuated proportionally. Thus, employing a normalized matched filter may result in an improvement in comparison with a standard matched filter. This modified overlap-add method is a non-limiting example of a particular implementation of the normalized matched filter. Alternative methods are contemplated and would be apparent to those skilled in the art of signal processing. For example, it is contemplated that an alternative normalized matched filter may be utilized based on the overlap-save method.
-
FIGS. 10A, 10B, 10C areplots levels 20 dB, 30 dB, and 50 dB above the signal level, forFIGS. 10A, 10B, and 10C , respectively. The prototype filter for each ofFIGS. 10A, 10B, and 10C is a rectangular window. - Whereas the standard matched filter may fail to reveal the regular pulses at its output, the normalized matched filter may remove most of the interference from its output. As a result, pulses may be observed and recoverable in most of the scenarios. The exception is the case where interference has a very high power in which the interference may not be sufficiently removed (e.g., the case presented in
FIG. 10C ). - Another embodiment described more fully below may further improve the ability to reduce the interference in harsh environments. To understand this improvement, a brief explanation is described herein as to the impact of the subcarrier power normalization in introducing any distortion to the regular pulses at the matched filter output. Equation (5) above describes the combined impulse response between the transmitter input and the standard matched filter output, assuming an ideal channel. By replacing the standard matched filter with the normalized matched filter, the corresponding combined impulse response will be:
-
- where Al is the real and positive normalization factor applied to the frequency domain samples in the lth subcarrier band. Following the same line of derivations that led to equations (7) and (10) above, equation (12) can also be simplified to the same form as equation (7), with:
-
- Similar to the standard β(t) in equation (10), the normalized β(t) in equation (13) is also a periodic function of time. Taking note that the coefficients Ak are real and positive, the periodic function of normalized β(t) in equation (13) has almost the same form (i.e., a sequence of positive and negative impulses) as the standard β(t) in equation (10). For t=nT, the exponential terms under the summation in equation (13) may have the phase of zero (or some multiples of 2π), and thus constructively add to a peak of Σk Ak. For
-
- on the other hand, these phases have the value of π (or some multiples of 2π plus π), hence, the summation on the right-hand side of equation (13) will be equal to −ΣkAk. In between these positive and negative peaks, normalized β(t) vanishes to a small value.
-
FIG. 11 is aplot 1100 of normalized periodic function β(t) according to an embodiment of the disclosure. For this example, the normalization coefficients Ak are a set of random numbers uniformly distributed over theinterval 0 to 1, and the parameter N is equal to 50. - The sample set Xi[k], for k=0, 1, . . . , L−1, are effectively a snap-shot of the outputs of filter bank with a prototype filter p[n]. For the normalized matched filter that has been described so far, prototype filter p[n] is a rectangular window. For example, the prototype filter p[n] may be a finite impulse response filter of length P with prototype filter p[n]=1, for n=0, 1, . . . , P−1. This rectangular prototype filter p[n] may have a frequency response that includes a sinc pulse. Such a sinc pulse whose magnitude response is presented in
FIG. 12 may have a poor stopband attenuation. Therefore, the sinc pulse may pick up a significant amount of signal power from all parts of the spectrum. Accordingly, when a portion of the spectrum is corrupted by a high-power interference, this interference can impact (e.g., leak to) all samples of Xi[k] and as a result may mask out the regular pulses at the output of the normalized matched filter. The results that were presented above inFIGS. 10A, 10B, and 10C give some idea of the impact of the rectangular prototype filter as interference power level increases. -
FIG. 12 is aplot 1200 showing arectangular pulse 1202 characterizing an FIR filter with coefficients of 1 and itscorresponding frequency response 1204. In the process of selecting a prototype filter p[n], thefrequency response 1204 of a rectangular window in the frequency domain may be undesirable as a direct consequence of the sharp edges at the beginning and the end of therectangular pulse 1202 for the prototype filter p[n]. -
FIG. 13 is aplot 1300 showing arectangular pulse 1302 with added roll-offs corresponding frequency response 1304. By adding smooth roll-offs to the beginning and the end of therectangular pulse 1302 of the prototype filter p[n], thefrequency response 1304 may be improved relative toFIG. 12 , and as a result, the leakage of a high-power interference to other parts of the transmission band may be reduced. The addition of smooth roll-offs rectangular pulse 1302 that may improve on its frequency domain properties have been adopted in different applications of signal processing. For example, in orthogonal frequency division multiplexing (OFDM), roll-offs are added to the beginning and the end of each OFDM symbol to improve on the subcarriers out-of-band emissions, so that the signals transmitted from different user equipment nodes in the network minimally interfere with one another. Here, smooth roll-offs - The addition of smooth roll-offs may also contribute to reducing the interference leakage.
FIG. 12 presented arectangular pulse 1202 as an FIR filter and itsfrequency response 1204.FIG. 13 , on the other hand, presents a similarrectangular pulse 1302 as an FIR filter after adding raised-cosine roll-offs FIG. 13 may be approximately 5% of the width of the rectangular window. Even this small amount of roll-off may contribute to improving the stop-band attenuation of the respective frequency response (compareFIGS. 12 and 13 ) and, as a result, as will be shown through an example later, in reducing the interference leakage in the context of this invention (see description ofFIGS. 19a and 19b , below). -
FIG. 14 is a visual representation of forming awindow 1406 for a prototype filter p(t) with raise-cosine roll-offs at the beginning and the end of thewindow 1406 through convolution of arectangular pulse 1402 and a half sine-wave 1404. The visual representation ofFIG. 14 may be expressed mathematically as: -
- where * denotes linear convolution,
-
- The factor
-
- on the right hand side of (16) may be selected such that the identity ∫−∞ ∞v(t)dt=1 holds. Applying a Fourier transform to both sides of (14):
-
-
FIGS. 15A, 15B, and 15C present a set ofsample plots plot 1500A shows |sinc(fTw)|,plot 1500B shows |V(f)|, andFIG. 16 shows -
- The prototype filter P(f) is a low-pass filter characterized by a narrow-band main lobe, a number of significant side lobes that expand over a bandwidth of
-
- and a number of well-attenuated side lobes at higher frequencies. If the stopband of the prototype filter is over the frequencies in which
-
- is −50 dB or less, its transition band may exhibit a width of about
-
- as shown in
FIG. 15B . This may give an idea of how far (in frequency) a high-power interferer may have noticeable impact on other parts of the spectrum. -
FIG. 16 shows aset 1600 of sampled rectangular windows w1[n], w2[n], w3[n], . . . that may be applied to an incoming signal x[n] to extract the partitions x1[n]=w1[n]x[n], x2[n]=w2[n]x[n], x3[n]=w3[n]x[n], . . . . The windows may be non-overlapping, but, in order not to miss any samples of x[n], the windows may be adjacent to each other. -
FIG. 17 is aset 1700 of sampled rectangular windows w1[n], w2[n], w3[n], that include the added roll-offs at the beginning and the end of windows. The rectangular windows w1[n], w2[n], w3[n], . . . must be chosen such that: -
- factoring x[n] out of the summation on the left-hand side of equation (19), the result simplifies to:
-
- For reference below, it is assumed that w0[n] (i.e., the window function of the 0th partition) is non-zero over the
interval 0≤n≤P−1. It is noted that that w0[n] is effectively the prototype filter p[n]. In addition, Q denotes the number of samples within each of the roll-off intervals. Therefore, recalling the window arrangements inFIG. 17 , one finds that: -
w i[n]=p[n−i(P−Q)]. (21) -
FIG. 18 is aflowchart 1800 illustrating a method of operating the normalized matched filter according to an embodiment of the disclosure. Atoperation 1802, the partition index i may first be initialized. Atoperation 1804, the portion of the incoming signal that corresponds to the present partition index is taken and the relevant window is applied. The result is the length P sequence xi[n]. - At
operation 1806, xi[n] is extended to the length L>P by appending zeros to its end and an FFT is applied to the result to convert it to the frequency domain. This frequency domain signal is called Xi[k]. The FFT operation may be synthesis filter bank with a rectangular prototype filter. In another embodiment, the rectangular prototype filter may be replaced with a filter with lower out-of-band emissions. - At
operation 1808, segments of Xi[k] that correspond to different subcarrier bands of the FB-MC-SS waveform are normalized according to the equation (11). The result is called {tilde over (X)}i[k]. - At
operation 1810, matched filtering is applied to the normalized frequency domain samples {tilde over (X)}i[k] to obtain {tilde over (Y)}i[k]={tilde over (X)}i[k]G*[k]. - At
operation 1812, the frequency domain matched filtered sequence {tilde over (Y)}i[k] is converted to the time domain through an IFFT operation and the result is overlapped and added to the previous partition of the output (in the time domain). - At
operation 1814, the partition index i is incremented by one andoperations 1804 through 1812 may be repeated. These operations may be repeated as long as necessary to obtain the output samples of the normalized matched filter that may be used in any packet detection and timing acquisition algorithm for finding the beginning of an FB-MC-SS data packet and obtaining the correct timing phase. -
FIGS. 19A and 19B areplots plot -
FIG. 19A shows the magnitude of the normalized matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of 0 dB and partial band interference at 50 dB above the signal level.FIG. 19A is a comparable result to that ofFIG. 10C , but with the roll-offs applied to the partitions of the incoming signal inFIG. 19A but not inFIG. 10C . As observed inFIG. 10C (without the addition of the roll-offs to the window), the normalized matched filter is not as effective in rejecting an interference that is 50 dB above the signal level in comparison to that ofFIG. 19A . In contrast, the results inFIG. 19A show that with the added roll-offs to the window, the normalized matched filter is more effective in rejecting the interference. As a result, the synchronization pulses are more apparent at the output shown inFIG. 19A compared withFIG. 10C . -
FIG. 19B shows the magnitude of the normalized matched filter output when the channel has corrupted the received signal with an AWGN at SNR level of 0 dB and partial band interference at 80 dB above the signal level. Analog amplifiers with linear performance over a dynamic range of 80 dB are not easy to implement according to conventional methods. Also, analog to digital convertor circuitries with typical 11 or 12 effective bits can cover a linear dynamic range of 60 to 70 dB, only. The normalized matched filter, is therefore, an improvement over conventional systems. -
FIG. 20 is a simplified block diagram of areceiver 2000 according to an embodiment of the disclosure. Thereceiver 2000 is configured to detect and decode in incoming RF signal 2005 (e.g., an FB-MC-SS signal). Thereceiver 2000 includes ademodulator 2010, normalized matchedfilter 2020,carrier recovery unit 2030, timingrecovery unit 2040,analysis filter bank 2050,channel estimator 2060, and maximum-ratio combining unit 2070. Thedemodulator 2010 receives theRF signal 2005, and converts theRF signal 2005 to abaseband input 2015. Thebaseband input 2015 is passed to theanalysis filter bank 2050 and the normalized matchedfilter 2020. - The normalized matched
filter 2020 is configured to have filter characteristics matched to those of the synthesis filter bank in the transmitter and generates a normalized output as discussed above. With the special selection of the parameters discussed above relative to the transmitter, the output from the normalized matchedfilter 2020 may be the sequence of high amplitude (i.e., narrow) pulses that can be clearly identified even when the FB-MC-SS signal is at a level below the noise level. The combination of the transmit and receive processes may result in a significant and unexpected improvement (both computationally faster and more precise) in timing recovery and frequency lock-in. - A sequence of
narrow pulses 2025 from the normalized matchedfilter 2020 may occur at regular intervals and carry the necessary information for timing and carrier acquisition as well as for their tracking. In general, any carrier and timing recovery method takes advantage of the properties of the modulation method that has been used to construct the transmitted signal. Due to the implementation inaccuracies and channel conditions (e.g., accuracy of components and Doppler shift in the channel), the carrier frequency used at the transmitter and its counterpart frequency at thereceiver 2000 are bound to suffer mismatch. - The
carrier recovery unit 2030 finds such mismatches and compensates for the mismatches at thereceiver 2000. To accomplish this carrier recovery, the normalized matchedfilter 2020 may be used to develop relatively efficient carrier and timing recovery algorithms. The output of the normalized matchedfilter 2020 includes the sequence ofnarrow pulses 2025 at the ½ spacing of the symbol timing (T/2). Thesenarrow pulses 2025 are present at the time instants that are even multiples of T/2 and correspond to about the center of the data symbols. - On the other hand, at the time instants that are odd multiples of T/2, the presence of such pulses depends on the information bits transmitted before and after the time instant. In other words, the pulse at the odd multiples of T/2 will appear when two consecutive symbols have the same value, otherwise the pulse would be a null. This property of the normalized matched
filter 2020 enables development of a timing recovery algorithm with a relatively low complexity/processing power. Once the pulses at the even multiples of T/2 are recognized, any carrier offset in the demodulated signal can be identified by comparing the relative phases of the pulses at the successive time instants, leading to carrier frequency recovery. - The
carrier recovery unit 2030 may receive the sequence ofnarrow pulses 2025 and generatecontrol signals 2035 to develop a phase-locked loop (PLL) between the demodulator 2010, the normalized matchedfilter 2020, and thecarrier recovery unit 2030 as is known by those skilled in the art. As the PLL locks on, the phase of thebaseband input 2015 is known relative to the sequence ofnarrow pulses 2025 from the normalized matchedfilter 2020. As a result, thetiming recovery unit 2040 may use the sequence ofnarrow pulses 2025 as a synchronization signal to ensure that the demodulated and combined signals from all subcarriers (i.e., the baseband input 2015) can be sampled at correct times by theanalysis filter bank 2050. - The sequence of
narrow pulses 2025 containing the timing information and thebaseband input 2015 from thedemodulator 2010 are passed to theanalysis filter bank 2050. Theanalysis filter bank 2050 extracts the received signal of each of the various subcarriers in parallel and samples them at a proper timing phase based on the timing information. Theanalysis filter bank 2050 performs the inverse process of the synthesis filter bank of the transmitter. Thus, the output of theanalysis filter bank 2050 is a set ofsignals 2055 corresponding to each of the subcarrier frequencies in the FB-MC-SS signal. Due to the very low power level of the received signal, some of these signals in the set ofsignals 2055 may not carry accurate information. However, many of the signals in the set ofsignals 2055 will carry the same information that was transmitted. - The
channel estimator 2060 examines the set ofsignals 2055 corresponding to each of the subcarrier frequencies to estimate a channel gain and a power of noise plus interference at each of the subcarrier bands by computing the impulse response of each channel in both time and frequency domain. Digital Signal Processing (DSP) and Field-Programmable Gate Arrays (FPGAs) may allow for one example of an inexpensive implementation for many of the functions in thereceiver 2000 and thechannel estimator 2060 in particular. - The maximum-
ratio combining unit 2070 receives the set ofsignals 2055 from theanalysis filter bank 2050 andinformation 2065 at the output of thechannel estimator 2060 to obtain an estimate of the transmitted data symbols. The maximum-ratio combining unit 2070 calculates a weighted average of the detected data symbols at the output of theanalysis filter bank 2050. The less noisy outputs are given higher weights and the more noisy outputs are given lower weights. To perform the combining, the maximum-ratio combining unit 2070 may divide a power estimate of each channel by the mean square of the estimated noise/interference, which may result in either a diminished noise where the signal is stronger or elimination of the certain subcarriers where the noise/interference is higher than the mean noise. -
FIG. 21 is a simplified block diagram of aRAKE receiver 2100 according to an embodiment of the disclosure. RAKE receivers are commonly used in direct sequence spread spectrum (DS-SS) systems to reduce the effects of the signal fading due to multipath. In the case of FB-MC-SS systems, the common implementation makes use of an analysis filter bank (e.g.,analysis filter bank 2010 inFIG. 20 ) to separate portions of the received signal that belong to different subcarrier frequency bands. The maximum ratio combining is then applied to the signal samples at the outputs of the analysis filter bank. By doing so, the subcarrier bands that are corrupted by interference are given lower gains and in this way a receiver with a near optimum performance is constructed. Such FB-MC-SS systems are known to have advantage over DS-SS in rejecting partial band interference. - The
RAKE receiver 2100 includes a normalized matchedfilter 2120 configured as described above with respect toFIG. 18 . The normalized matchedfilter 2120 may also be configured to remove partial band interference without resorting to the use of an analysis filter bank. As a result, theRAKE receiver 2100 may exhibit some performance loss when compared to thereceiver 2000 ofFIG. 20 , but may have the benefit of having lower complexity. - The
RAKE receiver 2100 makes use of the samples of the output of the normalized matchedfilter 2120 that correspond to different multi-paths of the channel. These multi-paths are combined together using a maximum ratio combiner. Themaximum ratio combiner 2170 gives higher gains to the stronger paths and lower gains to weaker paths. In this way, an optimum receiver that maximizes the SNR at its output (detected symbols) is constructed. - Thus, embodiments of the disclosure improve over conventional partial band interference suppression techniques known to the inventors that rely on the use of notch/bandstop filters which require the knowledge or estimation of the position of interference within the transmission band of the spread spectrum signal. Embodiments of the disclosure may be configured to suppress the interference without knowing the position of interference by applying a non-linear process to the analyzed signal in the frequency domain.
- While the present disclosure has been described herein with respect to certain illustrated embodiments, those of ordinary skill in the art will recognize and appreciate that it is not so limited. Rather, many additions, deletions, and modifications to the illustrated embodiments may be made without departing from the scope of the disclosure as hereinafter claimed, including legal equivalents thereof. In addition, features from one embodiment may be combined with features of another embodiment while still being encompassed within the scope of the disclosure. Further, embodiments of the disclosure have utility with different and various detector types and configurations.
Claims (22)
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US16/134,784 US10581481B1 (en) | 2018-09-18 | 2018-09-18 | Communication device, spread-spectrum receiver, and related method using normalized matched filter for improving signal-to-noise ratio in harsh environments |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US16/134,784 US10581481B1 (en) | 2018-09-18 | 2018-09-18 | Communication device, spread-spectrum receiver, and related method using normalized matched filter for improving signal-to-noise ratio in harsh environments |
Publications (2)
Publication Number | Publication Date |
---|---|
US10581481B1 US10581481B1 (en) | 2020-03-03 |
US20200091960A1 true US20200091960A1 (en) | 2020-03-19 |
Family
ID=69645696
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US16/134,784 Active US10581481B1 (en) | 2018-09-18 | 2018-09-18 | Communication device, spread-spectrum receiver, and related method using normalized matched filter for improving signal-to-noise ratio in harsh environments |
Country Status (1)
Country | Link |
---|---|
US (1) | US10581481B1 (en) |
Families Citing this family (2)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US10833900B2 (en) * | 2018-12-17 | 2020-11-10 | Lockheed Martin Corporation | Joint estimation of communication channel effects in communication receivers |
US20230028791A1 (en) * | 2021-07-22 | 2023-01-26 | Qualcomm Incorporated | Dynamic shaping filter indications |
Family Cites Families (43)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3849730A (en) | 1973-06-20 | 1974-11-19 | Bell Telephone Labor Inc | Carrier recovery in vestigial sideband data receivers |
US5204901A (en) | 1991-08-01 | 1993-04-20 | General Electric Company | Public key cryptographic mechanism |
US6154484A (en) | 1995-09-06 | 2000-11-28 | Solana Technology Development Corporation | Method and apparatus for embedding auxiliary data in a primary data signal using frequency and time domain processing |
US6950388B2 (en) | 1996-08-22 | 2005-09-27 | Tellabs Operations, Inc. | Apparatus and method for symbol alignment in a multi-point OFDM/DMT digital communications system |
DE69725995T2 (en) | 1996-08-29 | 2004-11-11 | Cisco Technology, Inc., San Jose | SPATIAL SIGNAL PROCESSING FOR TRANSMISSION SYSTEMS |
US6272121B1 (en) | 1997-03-03 | 2001-08-07 | Omnipoint Corporation | Spread spectrum communication system using DECT protocol |
US7430257B1 (en) | 1998-02-12 | 2008-09-30 | Lot 41 Acquisition Foundation, Llc | Multicarrier sub-layer for direct sequence channel and multiple-access coding |
US6741551B1 (en) | 1999-01-11 | 2004-05-25 | International Business Machines Corporation | Hybrid TDMA/CDMA system based on filtered multitone modulation |
JP2000244743A (en) | 1999-02-23 | 2000-09-08 | Sony Corp | Filter bank constitution method and filter bank device |
US6628728B1 (en) | 1999-04-28 | 2003-09-30 | Cyntrust Communications, Inc. | Nyquist filter and method |
FR2799073B1 (en) | 1999-09-29 | 2002-01-18 | France Telecom | METHOD FOR TRANSMITTING A BFDM / OQAM SIGNAL, CORRESPONDING MODULATION AND DEMODULATION METHODS AND DEVICE |
US7639597B2 (en) | 2000-07-19 | 2009-12-29 | Steve J Shattil | Method and apparatus for transmitting signals having a carrier-interferometry architecture |
US7558310B1 (en) | 2001-01-09 | 2009-07-07 | Urbain Alfred von der Embse | Multi-scale code division frequency/wavelet multiple access |
US7336693B2 (en) | 2001-05-08 | 2008-02-26 | Sony Corporation | Communication system using ultra wideband signals |
KR100547843B1 (en) | 2001-07-13 | 2006-02-01 | 삼성전자주식회사 | Apparatus and method for controling transmission power in mobile telecommunications system |
US7161987B2 (en) | 2001-09-26 | 2007-01-09 | Conexant, Inc. | Single-carrier to multi-carrier wireless architecture |
US7123934B1 (en) | 2002-02-15 | 2006-10-17 | Sprint Communications Company L.P. | Telecommunications overlay system |
US7099299B2 (en) | 2002-03-04 | 2006-08-29 | Agency For Science, Technology And Research | CDMA system with frequency domain equalization |
US7627056B1 (en) | 2002-03-29 | 2009-12-01 | Scientific Research Corporation | System and method for orthogonally multiplexed signal transmission and reception on a non-contiguous spectral basis |
US6651408B1 (en) | 2002-07-19 | 2003-11-25 | New Holland North America, Inc. | Adjustable force net brake mechanism and wrapping material dispensing assembly |
KR100461547B1 (en) | 2002-10-22 | 2004-12-16 | 한국전자통신연구원 | Transceiver for ds/cdma mimo antenna systems utilizing full receiver diversity |
US7180963B2 (en) | 2002-11-25 | 2007-02-20 | Ali Corporation | Digital receiver capable of processing modulated signals at various data rates |
JP4472692B2 (en) | 2003-02-19 | 2010-06-02 | パナソニック株式会社 | Receiver and method for digital multicarrier transmission |
CN1739274A (en) | 2003-02-19 | 2006-02-22 | 松下电器产业株式会社 | Receiving apparatus and method for digital multi-carrier transmission |
US20040196780A1 (en) | 2003-03-20 | 2004-10-07 | Chin Francois Po Shin | Multi-carrier code division multiple access communication system |
US7394844B1 (en) | 2003-05-19 | 2008-07-01 | Xtendwave, Inc. | Code division multiple access employing orthogonal filter banks |
US7443917B2 (en) | 2003-09-02 | 2008-10-28 | Data Jce Ltd | Method and system for transmission of information data over a communication line |
US7532723B2 (en) | 2003-11-24 | 2009-05-12 | Interdigital Technology Corporation | Tokens/keys for wireless communications |
US7787545B2 (en) | 2004-01-12 | 2010-08-31 | Xtendwave, Inc. | Multi-channel wavelet codec |
ES2885101T3 (en) | 2004-01-29 | 2021-12-13 | Neo Wireless Llc | Procedures and apparatus for superimposing direct sequence and multi-carrier spread spectrum signals in a broadband wireless communication system |
US7174141B2 (en) | 2004-04-30 | 2007-02-06 | Nokia Corporation | Apparatus, and associated method, for facilitating communications in a radio communication system through use of ultrawide band signals |
FR2882479B1 (en) | 2005-02-22 | 2007-04-20 | Thales Sa | METHOD AND DEVICE FOR SYNCHRONIZING RECTILIGNED OR QUASI-RECTILINE LINKS IN THE PRESENCE OF INTERFERENCE |
DE602005012504D1 (en) | 2005-08-22 | 2009-03-12 | Panasonic Corp | Combined OFDM and wavelet multicarrier transceiver |
US7873125B2 (en) | 2006-11-30 | 2011-01-18 | Broadcom Corporation | Method and system for sliding window phase estimator for WCDMA automatic frequency correction |
US8472304B2 (en) | 2009-03-31 | 2013-06-25 | Mitsubishi Electric Research Laboratories, Inc. | Carrier allocation and time sharing for OFDMA/TDMA networks |
CN102362453B (en) | 2009-04-01 | 2014-07-16 | 日本电信电话株式会社 | Wireless transmission method, wireless transmission system, and transmission apparatus and reception apparatus of wireless transmission system |
EP2273687B1 (en) | 2009-07-01 | 2016-12-07 | Telefonaktiebolaget LM Ericsson (publ) | Multi-path timing tracking and impairment modeling for improved grake receiver performance in mobility scenarios |
FR2951046B1 (en) * | 2009-10-02 | 2011-10-14 | Conservatoire Nat Des Arts Et Metiers Cnam | MULTI-DATA DIGITAL DATA TRANSMISSION SYSTEMS AND TRANSMISSION METHODS USING SUCH SYSTEMS |
US20110188671A1 (en) | 2009-10-15 | 2011-08-04 | Georgia Tech Research Corporation | Adaptive gain control based on signal-to-noise ratio for noise suppression |
CN101867548B (en) | 2010-05-24 | 2014-01-15 | 北京科技大学 | Blind frequency tracking algorithm based on multi-carrier of filter bank |
US8731027B2 (en) * | 2011-12-05 | 2014-05-20 | Battelle Energy Alliance, Llc | Methods and apparatuses using filter banks for multi-carrier spread-spectrum signals |
US8938041B2 (en) | 2012-12-18 | 2015-01-20 | Intel Corporation | Techniques for managing interference in multiple channel communications system |
US9838782B2 (en) * | 2015-03-30 | 2017-12-05 | Bose Corporation | Adaptive mixing of sub-band signals |
-
2018
- 2018-09-18 US US16/134,784 patent/US10581481B1/en active Active
Also Published As
Publication number | Publication date |
---|---|
US10581481B1 (en) | 2020-03-03 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US9559748B2 (en) | Methods and apparatuses using filter banks for multi-carrier spread spectrum signals | |
US11394594B2 (en) | Method and apparatus for high data rate long range acoustic and RF communication using chirp waveform modulation | |
US20110103430A1 (en) | Autocorrelation-Based Multi-Band Signal Detection | |
US10581481B1 (en) | Communication device, spread-spectrum receiver, and related method using normalized matched filter for improving signal-to-noise ratio in harsh environments | |
CN110933007B (en) | Device and method for eliminating interference aiming at OFDM broadband signal | |
Laraway et al. | HF band filter bank multi-carrier spread spectrum | |
Laraway et al. | Experimental Comparison of FB-MC-SS and DS-SS in HF Channels | |
Liu et al. | Iterative blind OFDM parameter estimation and synchronization for cognitive radio systems | |
Carron et al. | Comparison of two modulation techniques using frequency domain processing for in-house networks | |
Kanterakis | A novel technique for narrowband/broadband interference excision in ds-ss communications | |
Carrick et al. | An optimal filter for signals with time-varying cyclostationary statistics | |
Islam et al. | Preamble-based improved channel estimation for multiband UWB system in presence of interferences | |
Kumar et al. | Blind symbol rate estimation by exploiting cyclostationary features in wavelet domain | |
An et al. | Classification of single-and multi-carrier signals using CNN based deep learning | |
Ijiga | Channel estimation techniques for filter bank multicarrier based transceivers for next generation of wireless networks | |
US20230261922A1 (en) | Systems, devices, and methods for multi-band spread spectrum communication | |
Khan et al. | Performance comparison of wavelet packet modulation and ofdm over multipath wireless channel with narrowband interference | |
Anttonen et al. | Low complexity phase-unaware detectors based on estimator-correlator concept | |
Xu et al. | Broadband cognitive radio transmission based on sub-channel sensing and NC-OFDM | |
Arslan | Multidimensional signal analysis | |
de Lucena et al. | Optimum detection of non-orthogonal QAM signals with spectral overlapping | |
Robertson et al. | A Constant Envelope OFDM Implementation on GNU Radio | |
Jazdzewski et al. | Detection of direct sequence spread spectrum signals in the presence of harmonic and narrowband interferences | |
KOLLURU et al. | CFO & PAPR Estimation for OFDM/OQAM Systems Based on Sub Channel Signals | |
Carron et al. | COMPARISON OF TWO MODULATION TECHNIQUES USING FREQUENCY DOMAIN zyxwvutsrqpon |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
FEPP | Fee payment procedure |
Free format text: ENTITY STATUS SET TO UNDISCOUNTED (ORIGINAL EVENT CODE: BIG.); ENTITY STATUS OF PATENT OWNER: SMALL ENTITY |
|
FEPP | Fee payment procedure |
Free format text: ENTITY STATUS SET TO SMALL (ORIGINAL EVENT CODE: SMAL); ENTITY STATUS OF PATENT OWNER: SMALL ENTITY |
|
AS | Assignment |
Owner name: UNITED STATES DEPARTMENT OF ENERGY, DISTRICT OF COLUMBIA Free format text: CONFIRMATORY LICENSE;ASSIGNOR:BATTELLE ENERGY ALLIANCE;REEL/FRAME:047419/0717 Effective date: 20181023 |
|
AS | Assignment |
Owner name: BATTELLE ENERGY ALLIANCE, LLC, IDAHO Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:MORADI, HUSSEIN;DRIGGS, JONATHAN D.;MAJID, ARSLAN J.;AND OTHERS;SIGNING DATES FROM 20180906 TO 20180918;REEL/FRAME:050853/0429 |
|
AS | Assignment |
Owner name: BATTELLE ENERGY ALLIANCE, LLC, IDAHO Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:SIBBETT, TAYLOR;REEL/FRAME:051165/0802 Effective date: 20191203 |
|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |
|
MAFP | Maintenance fee payment |
Free format text: PAYMENT OF MAINTENANCE FEE, 4TH YR, SMALL ENTITY (ORIGINAL EVENT CODE: M2551); ENTITY STATUS OF PATENT OWNER: SMALL ENTITY Year of fee payment: 4 |