TWI648960B - New architecture design of millimeter-band wireless communication base station antenna - Google Patents
New architecture design of millimeter-band wireless communication base station antenna Download PDFInfo
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Abstract
為應付未來無線通訊數據傳輸的龐大需求，頻率波段必須往上提升到毫米波段，例如：28GHz、38GHz、60GHz、73GHz等等。本發明基於對天線技術、通道特性、通訊系統及陣列訊號處理的綜合理解，提出一種新的天線架構及簡易可行的訊號處理方法，可以大大地降低天線製作成本及訊號處理的運算量，並可獲得優異的傳輸性能。天線架構分成兩部分：高頻(RF)端之天線陣列部分及基帶端的陣列訊號處理部分。在高頻(RF)端之天線陣列部分，將全數的天線元素(element)分成幾組固定的次陣列(sub-array)，次陣列的參數包括天線元素的數目、瞄準的方位、以及相鄰次陣列之傳輸線相角的關係，這些參數的選擇則是基於對通道特性的瞭解，由基地台所處的環境及應用情境來設計。有別於傳統的相位陣列天線(phase array)需使用大量的數位相移器(digital phase shifter)，本架構在RF天線端完全不使用主動元件，亦沒有控制電路，因此全然沒有***損失(inserting loss)及耗電散熱的問題，與傳統的全數位多輸入輸出(multiple input multiple output)相較，本 發明採用次陣列的方式可大大地降低端點(port)的數目，減少送收機(transceiver)、類比轉數位(A/D)、數位轉類比(D/A)所需的數量，降低系統成本。 In order to meet the huge demand for future wireless communication data transmission, the frequency band must be increased to the millimeter wave band, for example: 28GHz, 38GHz, 60GHz, 73GHz, and so on. Based on a comprehensive understanding of antenna technology, channel characteristics, communication system, and array signal processing, the present invention proposes a new antenna architecture and a simple and feasible signal processing method, which can greatly reduce the antenna manufacturing cost and the amount of signal processing operations. Get excellent transmission performance. The antenna architecture is divided into two parts: the antenna array part at the high frequency (RF) end and the array signal processing part at the baseband end. At the high-frequency (RF) antenna array, the entire antenna element is divided into several fixed sub-arrays. The parameters of the sub-array include the number of antenna elements, the orientation of the target, and the neighbors. The relationship between the phase angles of the transmission lines of the sub-array. The selection of these parameters is based on the understanding of the channel characteristics, and is designed by the environment and application scenario of the base station. Unlike traditional phase array antennas, which require a large number of digital phase shifters, this architecture does not use active components at the RF antenna end and has no control circuit, so there is no insertion loss at all. Compared with the traditional multiple input multiple output, this problem The invention of the sub-array method can greatly reduce the number of endpoints, reduce the number of transmitters, analog to digital (A / D), and digital to analog (D / A), and reduce the system. cost.
在基帶端的陣列訊號處理部分，本設計利用無線通訊系統特有的指標訊號(pilot signal)，這種資源使得通道狀態響應(channel state information)可以獲得，本發明依據次陣列的特性及通道狀態響應，提出非常簡易的陣列訊號處理方式，可大量降低手機端及基地台端的數值運算量，但仍可獲得優異的性能。 In the array signal processing part of the baseband end, the original design uses a pilot signal unique to the wireless communication system. This resource makes channel state information available. The present invention is based on the characteristics of the sub-array and the channel state response. A very simple array signal processing method is proposed, which can greatly reduce the number of numerical calculations on the mobile phone side and the base station side, but still obtain excellent performance.
Description
本發明提出一種使用固定式次陣列(sub-array)方式的全新基地台天線架構及非常簡易的陣列訊號處理方法，可以應用在未來毫米波段無線寬頻通訊系統。 The invention proposes a new base station antenna architecture using a fixed sub-array method and a very simple array signal processing method, which can be applied to future millimeter wave wireless broadband communication systems.
傳統的時域多重接續(time domain multiple access,TDMA)系統，它在RF端使用相位陣列天線(phased array antenna)並使用大量的數位相移器(digital phase shifter)來調整波束的方向，相移器含有主動元件，串接多級的二極體開關來達到相位的改變，並經多級的功率結合器/分割器(power combiner/divider)後輸出/輸入，這些相移器及結合器將造成額外的***損失(insertion loss)，而且每一元素需要接一個相移器，當天線元素增加，將造成製作成本增加、控制電路複雜、體積增加，電力消耗及散熱等諸多的問題。 A traditional time domain multiple access (TDMA) system, which uses a phased array antenna at the RF end and uses a large number of digital phase shifters to adjust the beam direction and phase shift. The converter contains active components, which are connected in series with multi-stage diode switches to achieve phase change, and output / input after multi-stage power combiner / divider. These phase shifters and combiners will It causes additional insertion loss, and each element needs to be connected to a phase shifter. When the antenna elements increase, it will cause many problems such as increased production costs, complicated control circuits, increased volume, power consumption and heat dissipation.
傳統全數位化處理的多輸入輸出(multiple input multiple output,MIMO)技術各個端點(port)多使用相同特性之單元，當天線元素增加，端點的送收單元(transceiver unit)、類比轉數位(A/D)、數位轉類比(D/A)的數量也成比例的增加，這將大幅增加硬體的製作成本。在軟體部分，端 點數的大量增加，將造成基帶端訊號處理運算量的沉重負擔。 Traditional multiple input multiple output (MIMO) technology of traditional full digital processing uses multiple units with the same characteristics at each endpoint. When the number of antenna elements increases, the transceiver unit and analog to digital conversion of the endpoint The number of (A / D) and digital-to-analog (D / A) has also increased proportionally, which will greatly increase the production cost of the hardware. In the software section, the end The large increase in the number of points will cause a heavy burden on the baseband signal processing operations.
本發明基於對天線技術、通道特性、通訊系統及陣列訊號處理的綜合理解，提出一種新的天線架構及簡易可行的訊號處理方法，可以大大地降低天線製作成本及訊號處理的運算量，並可獲得優異的傳輸性能。天線架構分成兩部分：高頻(RF)端之天線陣列部分及基帶端的陣列訊號處理部分。在天線高頻(RF)端全不使用主動元件，沒有控制電路，因此全然沒有***損失的問題，在天線端將全數的元素分成幾組固定的次陣列(sub-array)，每一次陣列由數個天線元素所組成，次陣列的設計可依應用環境的情況來設計，但一經設計製作後，在該環境下就固定不變，亦即對於不同來向的用戶，天線端就固定不變，波束的調整，則在基頻端來處理。每一個次陣列為一個端點(port)，每一個端點連接一個送收機單元(transceiver unit)，將高頻信號經混波器降至基頻並經類比轉數位轉換成基帶的數位信號。各端點的基帶信號再做陣列信號處理而完成。與全基帶處理的多輸入輸出系統相比較，多輸入輸出系統的每個端點多是相同的單元，有相同的輻射特性，每個天線元素為一端點，因此端點的數目為天線元素的數目。本發明所提的新架構，端點數為次陣列的個數，因此可以大大降低送收機及類比轉數位(A/D)、數位轉類比(D/A)的數目，降低系統的成本。在基帶端的陣列訊號處理部分，本設計利用無線通訊系統特有的指標訊號(pilot signal)，這種資源使得通道狀態響應(channel state information)可以獲得，本發明依據次陣列的特性及通道狀態響應，提出非常簡易的陣列訊號處理方式，可大量降低手機端及基地台端的數值運算量，但仍可獲得優異的性能。 Based on a comprehensive understanding of antenna technology, channel characteristics, communication system, and array signal processing, the present invention proposes a new antenna architecture and a simple and feasible signal processing method, which can greatly reduce the antenna manufacturing cost and the amount of signal processing operations. Get excellent transmission performance. The antenna architecture is divided into two parts: the antenna array part at the high frequency (RF) end and the array signal processing part at the baseband end. No active components are used at the high-frequency (RF) end of the antenna, and there is no control circuit, so there is no problem of insertion loss. At the antenna end, all the elements are divided into fixed sub-arrays. Each array consists of Composed of several antenna elements, the design of the sub-array can be designed according to the application environment, but once designed and manufactured, it will be fixed in that environment, that is, for users from different directions, the antenna end will be fixed. Beam adjustment is handled at the fundamental frequency. Each sub-array is an endpoint, and each endpoint is connected to a transceiver unit. The high-frequency signal is reduced to the fundamental frequency by the mixer and converted to digital signals of baseband by analog to digital conversion. . The baseband signals at each endpoint are processed by array signal processing. Compared with the full-baseband multi-input-output system, each endpoint of the multiple-input-output system is mostly the same unit and has the same radiation characteristics. Each antenna element is an endpoint, so the number of endpoints is that of the antenna element. number. The new architecture proposed by the present invention has the number of endpoints as the number of sub-arrays, so the number of senders and analog to digital (A / D) and digital to analog (D / A) can be greatly reduced, and the cost of the system is reduced. . In the array signal processing part of the baseband end, the original design uses a pilot signal unique to the wireless communication system. This resource makes channel state information available. The present invention is based on the characteristics of the sub-array and the channel state response. A very simple array signal processing method is proposed, which can greatly reduce the number of numerical calculations on the mobile phone side and the base station side, but still obtain excellent performance.
如圖1所示，本發明的毫米波段無線通訊基地台天線架構具有多個次陣列100、多個相位差傳輸線單元140、多個升/降頻單元150、多個權重乘法器單元160及一第一結合器170，各次陣列100均具有多個天線110、多個相位差傳輸線單元120、一第二結合器130及一次陣列連接埠，其中，各天線110均經由一相位差傳輸線單元120與第二結合器130的一分支端耦接，且第二結合器130具有一總和端以與該次陣列連接埠耦接；各相位差傳輸線單元140的一端係與一所述次陣列連接埠耦接，而另一端則與一升/降頻單元150耦接；各升/降頻單元150的一端係與一相位差傳輸線單元140耦接，而另一端則與一權重乘法器單元160耦接，且各升/降頻單元150均具有一降頻轉換器151、一類比數位轉換器152、一數位類比轉換器153及一升頻轉換器154，其中，升/降頻單元150係藉由降頻轉換器151及類比數位轉換器152執行一下行操作，及藉由數位類比轉換器153及升頻轉 換器154執行一上傳操作；各權重乘法器單元160的一端係與一升/降頻單元150耦接，而另一端則與第一結合器170耦接；以及第一結合器170具有多個分支端以分別與各權重乘法器單元160耦接，以及一總和端以提供一整體天線連接埠；所述多個次陣列110在一預定的角度範圍內有多個不同的對準方向，且每兩個相鄰次陣列110的增益場型均有重疊。 As shown in FIG. 1, the antenna architecture of the millimeter-wave wireless communication base station of the present invention has multiple sub-arrays 100, multiple phase-difference transmission line units 140, multiple up / down frequency units 150, multiple weight multiplier units 160, and First coupler 170, each secondary array 100 has multiple antennas 110, multiple phase-difference transmission line units 120, a second combiner 130, and a primary array port, wherein each antenna 110 passes through a phase-difference transmission line unit 120 Is coupled to a branch end of the second coupler 130, and the second coupler 130 has a summing end to be coupled to the secondary array port; one end of each phase difference transmission line unit 140 is connected to a secondary array port And the other end is coupled to an up / down unit 150; one end of each up / down unit 150 is coupled to a phase difference transmission line unit 140, and the other end is coupled to a weight multiplier unit 160 And each up / down unit 150 has a down converter 151, an analog digital converter 152, a digital analog converter 153 and an up converter 154, wherein the up / down unit 150 is borrowed Down-converter 151 and analog digital Converter 152 performs a downlink operation, and by digital to analog converter 153, and frequency transfer The converter 154 performs an upload operation; one end of each weight multiplier unit 160 is coupled to an up / down frequency unit 150, and the other end is coupled to the first combiner 170; and the first combiner 170 has multiple The branch ends are respectively coupled to the weight multiplier units 160, and a summing end is provided to provide an overall antenna port; the plurality of sub-arrays 110 have a plurality of different alignment directions within a predetermined angle range, and The gain field patterns of every two adjacent sub-arrays 110 overlap.
另外，第n+1個第一相位差傳輸線單元140和第n個第一相位差傳輸線單元140的相位差為：βn+1-βn=(γn-γn+1)-kdtn cos((αn+αn+1)/2)，其中，n為正整數，βn+1為第n+1個第一相位差傳輸線單元140在第n+1個次陣列100的指向角度為αn+1時之相位，βn為第n個第一相位差傳輸線單元140在第n個次陣列100的指向角度為αn時之相位，γ n+1為第n+1個次陣列100在指向角度為(αn+αn+1)/2時所產生之一複數場相位，γ n為第n個次陣列100在指向角度為(αn+αn+1)/2時所產生之一複數場相位，k=2π/λ，λ為波長，d為二相鄰天線110之間隔，tn為第n個次陣列100的天線110的數目。 In addition, the phase difference between the n + 1th first phase difference transmission line unit 140 and the nth first phase difference transmission line unit 140 is: β n + 1 -β n = (γ n -γ n + 1 ) -kdt n cos ((α n + α n + 1 ) / 2), where n is a positive integer and β n + 1 is the orientation of the n + 1th first phase difference transmission line unit 140 at the n + 1th sub-array 100 Phase when the angle is α n + 1 , β n is the phase when the pointing angle of the n-th sub-array 100 in the n-th sub-array 100 is α n , and γ n + 1 is the n + 1 A complex field phase generated when the sub-array 100 has a pointing angle of (α n + α n + 1 ) / 2, γ n is the n-th sub-array 100 at a pointing angle of (α n + α n + 1 ) / A complex field phase generated at 2 o'clock, k = 2 π / λ , λ is the wavelength, d is the interval between two adjacent antennas 110, and t n is the number of antennas 110 of the n-th sub-array 100.
另外，在頻域雙工(FDD)系統下，一手機端可利用指標訊號得到各個次陣列100的通道響應值，俾以由N個次陣列100所產生的N個通道響應值及N-1組二相鄰次陣列100所產生的N-1個合成通道響應值進行一增益比較運算以決定由那一或那些次陣列100傳送訊號。 In addition, under the frequency domain duplex (FDD) system, a mobile phone can use the index signal to obtain the channel response values of each sub-array 100, using the N channel response values and N-1 generated by the N sub-arrays 100. The N-1 synthetic channel response values generated by the two adjacent sub-arrays 100 are subjected to a gain comparison operation to determine which signal or signals are transmitted by the sub-arrays 100.
如圖1，只考慮一維的天線陣列，假設由M個元素組成的均勻天線陣列，將之分成N個次陣列，第n個次陣列有t n 個元素(element)，，其 指向角度為α n ，這t n 個元素的權重(weighting)為 ，，k=2π/λ，λ為波長，d為 元素之間的間隔。次陣列的各個元素乘上權重後，經功率結合器相加，之後再經一段傳輸線到達次陣列的端點，該段傳輸所產生的相位為β n ，其示意圖如圖1所示。本架構的參數有M、N、t n 、α n 或、以及β n 。這些參數的選擇與無線通信系統的應用情境有關，例如室內環境如房間尺寸、形狀、長寬比值、使用者的分布情況；室外環境如街道的長寬比、群眾聚集的周遭環境等等。 As shown in Fig. 1, considering only a one-dimensional antenna array, suppose a uniform antenna array composed of M elements is divided into N sub-arrays, and the n-th sub-array has t n elements. , Its pointing angle is α n , and the weighting of these t n elements is , , K = 2 π / λ , λ is the wavelength, and d is the interval between the elements. Each element of the secondary array is multiplied by weights, added by a power combiner, and then reached the endpoint of the secondary array via a section of transmission line. The phase generated by this section of transmission is β n , and its schematic diagram is shown in FIG. 1. The parameters of this architecture are M, N, t n , α n or , And β n . The selection of these parameters is related to the application scenario of the wireless communication system, such as indoor environment such as room size, shape, aspect ratio, and user distribution; outdoor environment such as street aspect ratio, surrounding environment where people gather, and so on.
假設僅有一個路徑波，其來向為θ，第n個次陣列的功率增益場型(power gain pattern)可表示成。 值得注意的是，該架構完全沒有主動元件，次陣列之相鄰元素間的相位角差，以及傳輸線的相位角β n ，皆可經由調整傳輸線的長度來獲得。不管使用者的位置在哪裡，高頻天線的參數都固定不變，天線波束的調整及控制由基帶端來做數位信號處理。因此高頻端的天線結構不會增加任何的***損失。 Assuming that there is only one path wave, and its direction is θ, the power gain pattern of the nth sub-array can be expressed as . It is worth noting that this architecture has no active components at all, and the phase angle difference between adjacent elements of the sub-array , And the phase angle β n of the transmission line can be obtained by adjusting the length of the transmission line. Regardless of the user's location, the parameters of the high-frequency antenna are fixed. The adjustment and control of the antenna beam is performed by the baseband end for digital signal processing. Therefore, the antenna structure at the high frequency end does not increase any insertion loss.
每個次陣列端點輸出的高頻信號經過送收機降到基頻，並經類比轉數位轉成數值之後需做數位信號處理。在無線通信系統中，手機端會間隔子載波、間隔時間，或全部子載波間隔時間的傳送指標信號(pilot signal)，由已知的指標信號及測量到的量測值，可以求出或估測出那個時刻每一個子載波的頻率響應(sub-carrier frequency)或通道響應(channel response)，亦即通道頻率響應是可以得知的。令第n個次陣列的通道頻率響應表示為H n (ω)， 在做陣列信號處理時，我們就將第n個次陣列的輸出信號乘上一個權重為其自己的共軛複數，然後將N個值相加，並做正規化後即得到總和信號H T (ω)。 The high-frequency signal output from the end of each secondary array is reduced to the fundamental frequency by the sender and receiver, and then converted to a digital value by analog rotation to digital signal processing. In the wireless communication system, the mobile phone will transmit the pilot signal at intervals of the subcarriers, the interval time, or the interval of all subcarriers. The known indicator signals and measured values can be calculated or estimated. The frequency response (sub-carrier frequency) or channel response (channel response) of each sub-carrier at that moment is measured, that is, the channel frequency response can be known. Let the channel frequency response of the nth sub-array be expressed as H n ( ω ). When doing array signal processing, we multiply the output signal of the n-th sub-array by a weight to its own conjugate complex number, and then After the N values are added and normalized, the sum signal H T ( ω ) is obtained.
若天線陣列的來向波僅有一個從θ方向進來的入射波源，則
|H n (ω,θ)|2對θ作圖，即為第n個次陣列的功率增益場型，|H T (ω,θ)|2則為這N個次陣列之功率增益場型的總和。 | H n (ω, θ) | 2 is plotted against θ, which is the power gain field pattern of the nth sub-array, | H T (ω, θ) | 2 is the power gain field pattern of the N sub-arrays Sum.
上傳與下傳的載波頻率可依使用分時雙工(time division duplex,TDD)或分頻雙工(frequency division duplex,FDD)而有所不同。若使用分時雙工，則上下傳的載波頻率相同，因此上下傳有相同的通道響應。若使用分頻雙工，則上下傳的載波頻率不同，其通道響應亦不相同，因此兩種系統的信號處理方式亦不相同。如果是分時雙工系統，因為H n (ω)是可以得到的，因此我們可以將要傳送的數位信號X(ω)，在第n個次陣列的通道，乘上
的數值信號，經數位轉類比轉換成類比信號，再經送收機升頻
至高頻頻率後到達次陣列的輸入端，此時第n個次陣列輸入端的信號為
，再反向傳輸到次陣列的tn個元素，再輻射出去到達使用
者端，因為從次陣列端點到使用者端的通道響應為Hn(ω)，因此N個次陣列，亦即所有天線元素輻射到使用者端的總和為
因為為純量，使用者端可以獲得X(ω)的資訊，而整個天線 系統的通道增益為，由上面的分析，亦證明了分時雙工系 統，滿足了電波傳播的互易性(reciprocity)。 because Being scalar, the user can obtain X ( ω ) information, and the channel gain of the entire antenna system is From the above analysis, it is also proved that the time division duplex system satisfies the reciprocity of radio wave propagation.
如果使用分頻雙工系統，上傳下傳使用不同的載波頻率，載波頻率不同，通道響應會有極大的不同，因此不能使用上傳通道響應的共軛複數來做權重。 If a frequency-division duplex system is used, different carrier frequencies are used for upload and download, and the channel response will be greatly different for different carrier frequencies. Therefore, the conjugate complex number of the upload channel response cannot be used for weighting.
在LTE系統的標準，定義了數種標準類型的預編碼矩陣(precoding matrix)，在用戶端量測了各元素端的通道響應，分別乘上所有類型的預編碼矩陣，選擇一組有最佳性能的預編碼矩陣回報給基地台，基地台就使用那組預編碼矩陣來做權重，LTE-Advance已經定義了元素個數為8的預編碼矩陣，其款型高達16*16個，在手機端要執行選擇預編碼矩陣的運算負擔非常繁重。 In the standard of the LTE system, several standard types of precoding matrices are defined. The channel response of each element end is measured at the user end, and all types of precoding matrices are multiplied to select a group with the best performance. The precoding matrix is reported to the base station, and the base station uses that set of precoding matrices for weighting. LTE-Advance has defined a precoding matrix with 8 elements, and its models are up to 16 * 16. The computational burden of performing the selection of the precoding matrix is very heavy.
本發明提出下鏈基帶端的信號處理法則，以及在高頻端天線傳輸線相位的設計，讓手機端的運算負擔變得非常簡易而又有可以接受的傳輸性能。 The present invention proposes a signal processing rule at the baseband end of the downlink, and the design of the phase of the antenna transmission line at the high-frequency end, which makes the computational load on the mobile phone end very simple and has acceptable transmission performance.
100‧‧‧次陣列 100‧‧‧times array
110‧‧‧天線 110‧‧‧antenna
120‧‧‧相位差傳輸線單元 120‧‧‧phase difference transmission line unit
130‧‧‧第二結合器 130‧‧‧Second Coupler
140‧‧‧相位差傳輸線單元 140‧‧‧phase difference transmission line unit
150‧‧‧升/降頻單元 150‧‧‧ Up / Down Frequency Unit
151‧‧‧降頻轉換器 151‧‧‧ Downconverter
152‧‧‧類比數位轉換器 152‧‧‧ Analog Digital Converter
153‧‧‧數位類比轉換器 153‧‧‧ Digital Analog Converter
154‧‧‧升頻轉換器 154‧‧‧ Upconverter
160‧‧‧權重乘法器單元 160‧‧‧ weight multiplier unit
170‧‧‧第一結合器 170‧‧‧first coupler
M‧‧‧天線陣列元素總數 M‧‧‧ Total number of antenna array elements
N‧‧‧次陣列總數 Total number of N‧‧‧ arrays
t n ‧‧‧第n個次陣列元素個數 t n ‧‧‧nth array element number
α n ‧‧‧第n個次陣列指向角度 α n ‧‧‧nth array pointing angle
k=2π/λ‧‧‧為波數 k = 2 π / λ is the wave number
d‧‧‧為元素之間的間隔 d‧‧‧ is the interval between elements
λ‧‧‧為波長 λ‧‧‧ is the wavelength
=kdcosαn = kdcosα n
βn‧‧‧第n個次陣列傳輸線的相角 β n ‧‧‧ phase angle of the nth array transmission line
ωn‧‧‧第n個次陣列的權重 ω n ‧‧‧ weight of the nth sub-array
圖1是本發明提出使用次陣列方式的基地台天線架構。 FIG. 1 is a base station antenna architecture using a sub-array method proposed by the present invention.
圖2是安裝環境(1)長形馬路，基地台置於馬路的一端示意圖。 Fig. 2 is a schematic diagram of the installation environment (1) a long road with the base station placed at one end of the road.
圖3~圖4是本發明於安裝環境(1)與文獻方法的功率增益場型比較圖。 FIG. 3 to FIG. 4 are comparison diagrams of power gain field types in the installation environment (1) and the literature method of the present invention.
圖5是安裝環境(2)長形馬路，基地台置於馬路之一側示意圖。 Fig. 5 is a schematic diagram of the installation environment (2) a long road with the base station placed on one side of the road.
圖6~圖7是本發明於安裝環境(2)與文獻方法的功率增益場型比較圖。 FIG. 6 to FIG. 7 are comparison diagrams of power gain field types in the installation environment (2) and the literature method of the present invention.
圖8是本發明將次陣列個數減少為四個，安裝環境(1)及安裝環境(2)示意圖。 FIG. 8 is a schematic diagram of the installation environment (1) and the installation environment (2) by reducing the number of secondary arrays to four according to the present invention.
圖9~圖12是本發明於圖8與文獻方法的功率增益場型比較圖。 FIG. 9 to FIG. 12 are comparison diagrams of power gain field patterns of the present invention in FIG. 8 and the literature method.
方式1： Way 1:
N組次陣列每次只使用一組次陣列，每一組的次陣列有其功率增益場型(gain pattern)，依序輪流使用不同的次陣列傳送，使用者端將收到N個接收值，從N個接收值中取有最大值的那一個次陣列來負責發射，其它組的次陣列就不使用，這時所對應的權重向量為[0..010..0]，如果基地台到使用者端只有一條直接波，則基地台天線在各個方向的功率增益就是這N條次陣列的功率增益(gain pattern)曲線取最大值。 N sets of sub-arrays use only one set of sub-arrays at a time. Each set of sub-arrays has its power gain pattern. It uses different sub-arrays to transmit in turn. The user terminal will receive N received values. The sub-array with the maximum value from the N received values is responsible for transmitting, and the sub-arrays of other groups are not used. At this time, the corresponding weight vector is [0..010..0]. There is only one direct wave at the user end, so the power gain of the base station antenna in all directions is the maximum value of the power pattern of the N sub-arrays.
方式2： Way 2:
N組次陣列每次取相鄰的兩個次陣列同時發射，其餘的次陣列則不發射，這時基帶數位信號處理所對應的權重向量為[0..0110..0]/；假設基定台天線和使用者僅有一條直接波，使用者所在的方向為θ，則第n個次陣列發射的波在θ方向的複數場值為
第(n+1)個次陣列發射的波在θ的複數場值為
Hn(θ)和Hn+1(θ)的總和為
我們要調整βn和βn+1，使得|HTn(θ)|在有興趣的θ範圍內有較高的場值，我們選擇當θ=(αn+αn+1)/2時，兩項的相位相等，此時，HTn(θ=(αn+αn+1)/2)的功率增益將是兩個次陣列功率增益的總和，因此將有最大的增益。 We want to adjust β n and β n + 1 so that | H Tn (θ) | has a higher field value in the range of θ of interest. We choose when θ = (α n + α n + 1 ) / 2 , The phases of the two terms are equal. At this time, the power gain of H Tn (θ = (α n + α n + 1 ) / 2) will be the sum of the power gains of the two sub-arrays, so there will be a maximum gain.
設
γ n 、γ n+1都可以求得，則βn、βn+1的選擇為γ n +βn=γ n+1+kdtn cos((αn+αn+1)/2)+βn+1或者βn+1-βn=(γ n -γ n+1)-kdtn cos((αn+αn+1)/2)。 Both γ n and γ n +1 can be obtained, and the choice of β n and β n + 1 is γ n + β n = γ n +1 + kdt n cos ((α n + α n + 1 ) / 2) + β n + 1 or β n + 1 -β n = ( γ n -γ n +1 ) -kdt n cos ((α n + α n + 1 ) / 2).
上式說明了兩相鄰次陣列所應補償的量，主要的補償量是補償兩相鄰次陣列在空間傳播時路徑差所引起的相位差。 The above formula illustrates the amount that two adjacent sub-arrays should compensate. The main compensation amount is to compensate the phase difference caused by the path difference between two adjacent sub-arrays in space propagation.
假設β1的值為任意的定值，我們可以由上式逐步的決定β2,β3,...,βN，這些相位角度決定之後，我們可以畫出兩相鄰次 陣列同時發射時的功率增益場型曲線。依序輪流發射，使用端將獲得(N-1)個接收值，選取最大值所對應的(n,n+1)組合，並回報給基地台端，之後在傳送數值信號時，就使用那兩個次陣列來發射，每種組合有一條增益場型的曲線，總共可以獲得(N-1)條曲線，將(N-1)條曲線都取最大值，即為這種架構方式的增益場型。 Assuming that the value of β 1 is arbitrary, we can gradually determine β 2 , β 3 , ..., β N by the above formula. After these phase angles are determined, we can draw two adjacent sub-arrays transmitting at the same time. Curve of the power gain field. The transmission is performed in turn. The user will obtain (N-1) received values, select the (n, n + 1) combination corresponding to the maximum value, and report it to the base station. When transmitting a numerical signal, use the two Each sub-array is used to transmit, and each combination has a curve of gain field type. A total of (N-1) curves can be obtained. The maximum value of (N-1) curves is the gain field of this architecture. type.
以上的分析是決定基地台端高頻天線的設計過程。在實際的通信系統，基地台端點發送指標(pilot)信號，手機端會隨時收到各個次陣列的通道響應(channel response)，上述所謂的輪流發射或兩組同時發射及求最大值的動作，其實是在手機端，依隨時量到的通道響應，以軟體的方式進行運算及選取，並將選取的組合回報基地台，讓基地台在傳送數值信號時，決定使用那一組合，因此亦可就方式1及方式2共(2N-1)個值，選取最大值後將對應的次陣列組合回報給基地台。 The above analysis determines the design process of the high-frequency antenna at the base station. In an actual communication system, the base station endpoint sends a pilot signal, and the mobile phone terminal will receive the channel response of each sub-array at any time. In fact, on the mobile phone side, according to the channel response that can be measured at any time, the calculation and selection are performed in software, and the selected combination is reported back to the base station, so that the base station decides which combination to use when transmitting the numerical signal, so it can also Regarding the mode 1 and mode 2 (2N-1) values in total, the corresponding sub-array combination is returned to the base station after selecting the maximum value.
方式3： Way 3:
基地台端各次陣列埠端單獨傳送指標信號時，用戶端可以量到第m個port到用戶端的通道響應Hm(ω)，我們將Hm(ω)乘上一個
權重值。然後將這些值相加得到
上式的意義代表在基地台端的權重向量(weighting vector)為(ω1 ,…,ωm)，從用戶端天線的接收值，我們要找一組的{ωm}，
使得|yT(ω)|有最大值。很顯然地，當
會有最大值，這樣的組合方式稱為相同
增益組合(equal gain combining)，但要將該ωm傳回基地台，要花用相當多的資源。如果我們對ωm,Pm以及做一些限制，就可以簡化問題。首先，我們限制ωm ，其中，
Pm {0,1,…,P-1}，例如P=4，
8，則
，我們限定ω1=1=ej0°，且令，因為
Hm(ω)為手機端量到的通道響應，因此|Hm(ω)|及αm皆為已知。我們選擇ωm，使得|yT(ω)|有最大值，因此
欲使|y T (ω)|有最大值，則上式各項的相位角應盡量的同相，因此P m 的選擇就是使
由已知的α m ，以及上面的關係式，就可以很容易的求得對應的P m 。將所解出的P m ，回報給基地台做為基地台端的Precoding vector，在傳送訊號時，基地台使用對應之Precoding vector{ω 1,...ω m }，用戶端收到的訊號為y T (ω)。由於{ω m }及{H m (ω)}皆已獲得，欲解出所傳送的信號，可將y T (ω)X乘上y T *(ω)，亦 即y T (ω)X．(Σω m *．H m *(ω))=(Σω m H m (ω)X)．(Σω m *．H m *(ω))=|Σω m H m (ω)|2．X From the known α m and the above relation, the corresponding P m can be easily obtained. Report the resolved P m to the base station as the Precoding vector at the base station. When transmitting the signal, the base station uses the corresponding Precoding vector { ω 1 , ... ω m }. The signal received by the client is y T ( ω ). Since { ω m } and { H m ( ω )} have been obtained, to solve the transmitted signal, y T ( ω ) X can be multiplied by y T * ( ω ), that is, y T ( ω ) X. (Σ ω m * . H m * ( ω )) = (Σ ω m H m ( ω ) X ). (Σ ω m * . H m * ( ω )) = | Σ ω m H m ( ω ) | 2 . X
因為|Σω m H m (ω)|2為一純量，因此所傳送的信號X即可解出來。本方法求解{ω m }的方法非常簡單，所需的運算量亦很輕微，不似LTE選擇Precoding matrix的方法，要從眾多的候選名單中，經過繁複的計算，才能選出最佳的結果。而且本方法要將求得的{ω m }傳回基地台，所需的資訊量亦非常之少，所需的位元數僅為M log2 P，因此本方法在實際應用上，是簡易可行的。 Since | Σ ω m H m ( ω ) | 2 is a scalar quantity, the transmitted signal X can be solved. The method of solving { ω m } by this method is very simple, and the amount of calculation required is also very small. Unlike the method of LTE selecting the Precoding matrix, it is necessary to select the best result after arduous calculation from a large number of candidate lists. Moreover, this method needs to return the { ω m } obtained to the base station, the amount of information required is very small, and the number of bits required is only M log 2 P. Therefore, this method is simple in practical application feasible.
值得一提的是，如果P值越大，本方式所選擇的ω m ，就愈接近equal-gain combining的共軛相位，而上行信號所使用的結合方式為maximal ratio combining，因此當P愈大，上下行的增益場型就可以越接近。 It is worth mentioning that if the value of P is larger, the ω m selected by this method is closer to the conjugate phase of equal-gain combining, and the combination method used for uplink signals is maximum ratio combining, so when P is larger The closer the up and down gain field patterns can be.
考慮一長型街道，基地台置於馬路的一端，如圖2所示，擬通信的涵蓋範圍為整條街道，馬路遠端與天線中線的夾角較小，越靠近基地台的馬路兩側夾角越大，但距離基地台較近，為了涵蓋較遠的範圍及維持可接受的品質，陣列增益的需求應是距離遠夾角小的部分要有較高的增益，而距離近夾角大的區域容許有較小的增益。我們先只考慮水平方向的陣列增益，仰角方向暫不考慮，我們先設計的陣列天線含有八個次陣列，各次陣列 的元素數目及所對準的方向亦列於表一及圖2，各次陣列在各方向的功率增益場型亦於圖3(a)，上行信號經過最大比值結合(maximal ratio combining)，之後的陣列增益場型示於圖3(b)，如果使用相位陣列，並調整相移器，使對準用戶方向，再扣除5dB的相移器***損失後，其增益場型亦示於圖中。 Consider a long street with the base station at one end of the road. As shown in Figure 2, the coverage area to be communicated is the entire street. The angle between the far end of the road and the center line of the antenna is smaller. The larger the angle, but the closer the base station, in order to cover the longer range and maintain acceptable quality, the array gain needs to have a higher gain from the part with the smaller included angle and the region with the larger included angle. Smaller gains are allowed. We first consider the array gain in the horizontal direction, but not in the elevation direction. The array antenna we designed first contains eight sub-arrays. The number of elements and the aligned directions are also listed in Table 1 and Figure 2. The power gain field pattern of each array in each direction is also shown in Figure 3 (a). The uplink signals are subjected to maximum ratio combining, and then The array gain field pattern is shown in Figure 3 (b). If a phase array is used and the phase shifter is adjusted so that it is aligned with the user's direction, the 5dB phase shifter insertion loss is subtracted. The gain field pattern is also shown in the figure. .
圖中顯示，夾角越小，或距離越遠處，本方法可以有較佳的增益，而夾角較大，距離較近的兩側，本方法的增益較小，但因距離近，增益小並不會影響系統的傳輸性能。 The figure shows that the smaller the angle, or the farther away, the better the method can be, and the larger the angle, the closer the two sides, the smaller the gain of this method. Does not affect the transmission performance of the system.
在下行部分，結合相鄰兩個次陣列同時發送時，其增益場型如圖4(a)所示，採方式一、二並採2N-1=15條曲線取最大值後的結果示於圖4(b)，採方式三(並令P=8)、使用相位陣列並對準用戶端、以及使用傳統八個元素的MIMO，使用LTE-advance規格所訂的預編碼矩陣，由(rank=1)16x16=256種可能中選出最好的結果，亦顯示於圖中。需注意的是手機端選擇預編碼矩陣所需的運算量負擔極大。 In the downstream part, when combining two adjacent sub-arrays to send at the same time, the gain field type is shown in Figure 4 (a). The first and second methods are used together and 2N-1 = 15 curves are taken to obtain the maximum value. The results are shown in Figure 4 (a). Figure 4 (b), using the third method (and let P = 8), using a phase array and aligning the user side, and using the traditional eight-element MIMO, using the precoding matrix defined by the LTE-advance specification, = 1) 16x16 = 256 out of 256 possibilities. The best result is also shown in the figure. It should be noted that the calculation load required to select a precoding matrix on the mobile phone end is extremely heavy.
圖5是將基地台置於馬路的一側，並向馬路遠處的兩端發射，此時陣列天線場型的設計要點是夾角接近90°的兩端距離較 遠，需有較大的增益，而夾角較小的範圍距離較近，可以使用較小的增益，我們亦設計了八個次陣列的基地台天線，各次陣列天線的元素個數，瞄準方向示於表二及圖6，每個次陣列天線的增益場型亦示於圖6(a)，八個次陣列天線的信號經過最大比例結合後的結果示於圖6(b)，使用相位陣列，並瞄準用戶的方向，以及信號的八個元素的MIMO，經最大比例結合後的結果亦顯示於圖中。值得注意的是本例左右兩端次陣列天線瞄準的方向是兩端，而這兩次陣列有幾乎相同的場型，因此總增益在兩端增強很多。 Figure 5 shows the base station placed on one side of the road and transmitting to both ends of the road. At this time, the design point of the array antenna field is that the distance between the two ends is close to 90 °. It needs a large gain, and the range with a small included angle is relatively close. A small gain can be used. We also designed eight base station antennas, the number of elements in each array antenna, and the aiming direction. It is shown in Table 2 and Figure 6. The gain field type of each sub-array antenna is also shown in Figure 6 (a). The results of the signals of the eight sub-array antennas after the maximum proportion are shown in Figure 6 (b). Array, and aiming at the direction of the user, as well as the eight elements of the signal's MIMO, the results of the combined maximum ratio are also shown in the figure. It is worth noting that in this example, the aiming directions of the secondary array antennas at the left and right ends are both ends, and the two arrays have almost the same field pattern, so the total gain is greatly enhanced at both ends.
在下行部分，採方式二，結合相鄰兩個次陣列同時發送，其增益場型如圖7(a)所示，擇方式一、二並採2N-1=15條曲線取最大值後的結果示於圖7(b)，值得注意的是，在左右兩端方向，下行的增益遠不及上行的增益，因為我們只取相鄰兩個次陣列同時發射，雖然第一次陣列與第八次陣列有接近的場型，但方式二並沒有將第一次陣列和第八次陣列同時發送的情況考慮進去。使用方式三，並令P=8、使用相位陣列並對準用戶端、以及使用八個元素的MIIMO，並從16*16個Precoding matrix找出最佳 組合後的結果均標示於圖中，圖中顯示方式三的效果優於方式一、二，因為它利用了全部次陣列相結合的結果。 In the downstream part, the second method is adopted, which combines two adjacent sub-arrays to transmit at the same time. The gain field type is shown in Figure 7 (a). The second and second methods are used to select 2N-1 = 15 curves and take the maximum value. The results are shown in Figure 7 (b). It is worth noting that in the left and right directions, the gain of the downlink is far less than the gain of the uplink, because we only take two adjacent sub-arrays to transmit at the same time. The secondary array has a close field type, but the second array does not take into account the case where the first array and the eighth array are sent at the same time. Use method three, and let P = 8, use phase array and align the user side, and use eight elements of MIIMO, and find the best from 16 * 16 Precoding matrix The combined results are shown in the figure. The effect of mode 3 shown in the figure is better than modes 1 and 2, because it uses the result of combining all sub-arrays.
不管是上行還是下行，本發明所使用的方法，在左右兩端方向的增益都比相位陣列和8個端點的MIMO高不少，但在夾角較小、距離較近的範圍，增益則較小。但因距離短，雖然增益較小，也不太影響傳輸的效能。 Regardless of uplink or downlink, the method used in the present invention has a much higher gain in the left and right ends than the phase array and 8-terminal MIMO, but in the range with a small included angle and a short distance, the gain is relatively high. small. However, due to the short distance, although the gain is small, it does not affect the transmission performance.
或許有人會質疑，使用八個端點，需八組的收發單元，A/D，D/A，其成本太高，我們若將陣列天線的總元素減少，並將次陣列個數減少為四個，如圖8所示，重複前面的方法，次陣列的參數，瞄準方向示於表三、表四，所得的結果示於圖9至圖12，用另二種傳統方式所得的結果亦顯示於圖中，以茲比較。 Some people may question that using eight endpoints requires eight sets of transceiver units, A / D, D / A. The cost is too high. If we reduce the total elements of the array antenna and reduce the number of sub-arrays to four As shown in Figure 8, repeat the previous method. The parameters of the secondary array and the aiming directions are shown in Tables 3 and 4. The results obtained are shown in Figures 9 to 12, and the results obtained by the other two traditional methods are also shown. In the figure, comparison is made here.
上面的例子，只考慮水平方向的結果，然而次陣列天線的特 性亦可顯現在仰角的場型上，例如同樣的樓層高度，距離近，仰角就很大，距離遠，仰角就很小，我們亦可根據基地台所處的周圍環境，來選擇次陣列所對準的仰角及使用的元素個數，對於近距離有高樓時，對準的仰角稍大，且元素個數較少，可獲得較寬的波束，涵蓋近處的高樓。距離遠的高樓，仰角變小許多，可以增加垂直方向元素的個數，以增加遠距離的增益，所瞄準的仰角角度亦可以接近水平，以涵蓋遠處的高樓。表五亦列了各個次陣列的元素個數(包括水平方向及垂直方向)及所瞄準的仰角角度及水平角度。 The above example only considers the results in the horizontal direction. The characteristics can also be displayed on the field of the elevation angle. For example, the same floor height, the distance is short, the elevation angle is large, and the distance is long, the elevation angle is small. We can also choose the position of the secondary array according to the surrounding environment of the base station. The quasi-elevation angle and the number of elements are used. For high-rise buildings at a close distance, the aligned elevation angle is slightly larger and the number of elements is smaller. A wider beam can be obtained to cover nearby high-rise buildings. High-rise buildings at long distances have much smaller elevation angles. You can increase the number of elements in the vertical direction to increase the gain at long distances. The target elevation angle can also be close to horizontal to cover distant high-rise buildings. Table 5 also lists the number of elements in each sub-array (including horizontal and vertical directions) and the elevation and horizontal angles aimed at.
上述例子顯示固定式次陣列的概念具有相當大的彈性，可以根據基地台的周遭環境及應用情境，適當地設計次陣列的參數。天線公司亦可根據數種典型的情境，預先設計數款次陣列的類型以供客戶選擇。 The above example shows that the concept of a fixed sub-array has considerable flexibility. The parameters of the sub-array can be appropriately designed according to the surrounding environment of the base station and the application situation. Antenna companies can also pre-design several types of sub-arrays for customers to choose according to several typical scenarios.
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Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102110883A (en) * | 2010-12-01 | 2011-06-29 | 西安空间无线电技术研究所 | Beam-forming method for forming array antenna of variable beam |
| TW201433005A (en) * | 2013-01-25 | 2014-08-16 | Intel Corp | Apparatus, system and method of wireless communication via an antenna array |
| TW201534063A (en) * | 2013-12-18 | 2015-09-01 | Alcatel Lucent | Beamforming apparatuses, methods and computer programs for a base station transceiver and a mobile transceiver |
| CN105009473A (en) * | 2013-03-05 | 2015-10-28 | Lg电子株式会社 | Method of reporting channel state information for vertical beamforming in a multicell based wireless communication system and apparatus therefor |
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| TW201433005A (en) * | 2013-01-25 | 2014-08-16 | Intel Corp | Apparatus, system and method of wireless communication via an antenna array |
| CN105009473A (en) * | 2013-03-05 | 2015-10-28 | Lg电子株式会社 | Method of reporting channel state information for vertical beamforming in a multicell based wireless communication system and apparatus therefor |
| TW201534063A (en) * | 2013-12-18 | 2015-09-01 | Alcatel Lucent | Beamforming apparatuses, methods and computer programs for a base station transceiver and a mobile transceiver |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| TWI710785B (en) * | 2019-05-17 | 2020-11-21 | 何忠誠 | High resolution spatial angle scanning radar system and its design method |
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