TWI556559B - A Bidirectional DC - DC Converter with Adaptive Phase Shift Angle Control Mechanism - Google Patents
A Bidirectional DC - DC Converter with Adaptive Phase Shift Angle Control Mechanism Download PDFInfo
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本發明係有關於雙向直流-直流轉換器，特別是關於一種具自適應相移角控制機制之雙向直流-直流轉換器。The present invention relates to a bidirectional DC-DC converter, and more particularly to a bidirectional DC-DC converter having an adaptive phase shift angle control mechanism.
能源短缺是全球面臨的重要問題，為解決這個問題，再生能源的開發是不可或缺的。但再生能源容易受天候因素影響，使得供電不穩定且發電量難以預估，為解決這個問題，微電網系統(Micro-Grid System)乃被提出。圖1為微電網系統架構圖，微電網系統能在負載需求低於再生能源所提供的能量時，將多餘的能量透過雙向轉換器儲存在電池中，或是將能量饋回市電，而在負載需求高於再生能源所提供的能量時，也能將電池的能量透過雙向轉換器提供給負載使用，若是電池儲能不足，也可以從市電饋入能量以滿足負載所需。如此一來，不但增加了能量的可用性，同時也解決了能源的供需問題，故雙向轉換器在微電網系統內為重要元件。Energy shortage is an important issue facing the world. To solve this problem, the development of renewable energy is indispensable. However, the regenerative energy is easily affected by weather factors, making the power supply unstable and the power generation amount difficult to predict. To solve this problem, the Micro-Grid System has been proposed. Figure 1 shows the microgrid system architecture. The microgrid system can store excess energy in the battery through the bidirectional converter when the load demand is lower than the energy provided by the regenerative energy, or feed the energy back to the mains. When the demand is higher than the energy provided by the renewable energy source, the energy of the battery can also be supplied to the load through the bidirectional converter. If the battery is insufficiently stored, the energy can be fed from the commercial power to meet the load. In this way, not only the energy availability is increased, but also the supply and demand of energy is solved, so the bidirectional converter is an important component in the microgrid system.
另外，自1973年後世界經歷了三次的石油危機，國際原油的價格常居高不下。有鑑於此，各車廠開始研發各式油電混合車(Hybrid electric vehicle, HEV)、插電式油電混合車(Plug-in Hybrid electric vehicle, PHEV)以及純電動車(Battery electric vehicle, BEV)，以達到環保及降低燃料成本之目的。圖2為電動車的電力介面，油電混合車在剛起步或是低速行駛時，車輛動力會由電池供給以減少燃油的使用，在一般行駛時，車輛動力則會由燃油給予，而在上坡或是高速行駛時，車輛動力會由燃油和電池合併給予，使車輛有更大的動力，在車輛煞車時，則會回收剩餘動能為電池充電。若由電動車內部電力介面來看，電池係透過雙向升降壓直流-直流轉換器與直流匯流排連接。車輛加速時，如果電池端電壓高於直流匯流排端電壓，電池為降壓放電；反之，電池為升壓放電。而車輛煞車時，如果電池端電壓高於直流匯流排端電壓，電池為升壓充電；反之，電池為降壓充電。另外，車輛到電網(Vehicle to Grid, V2G)模式是指當電動汽車長時間靜置，車載電池所儲存電能可銷售給電網，而當車載電池需要充電，充電所需電能則需由電網提供。在V2G充電模式，如果整流匯流排端電壓高於電池端電壓，電池為降壓充電；反之，電池為升壓充電。在V2G放電模式，如果整流匯流排端電壓高於電池端電壓，電池為升壓放電；反之，電池為降壓放電。由此可知雙向升降壓直流-直流轉換器亦為電動車重要裝置之一。In addition, since 1973, the world has experienced three oil crises, and the price of international crude oil is often high. In view of this, various car manufacturers began to develop various hybrid electric vehicle (HEV), Plug-in Hybrid electric vehicle (PHEV) and Battery electric vehicle (BEV). To achieve environmental protection and reduce fuel costs. Figure 2 shows the power interface of an electric vehicle. When the hybrid vehicle is just starting or running at a low speed, the vehicle power is supplied by the battery to reduce the use of fuel. In normal driving, the vehicle power is given by the fuel. When driving on a slope or at a high speed, the vehicle's power will be combined by the fuel and the battery, so that the vehicle has more power. When the vehicle is braking, the remaining kinetic energy is recovered to charge the battery. According to the internal power interface of the electric vehicle, the battery is connected to the DC bus through a bidirectional buck-boost DC-DC converter. When the vehicle is accelerating, if the battery terminal voltage is higher than the DC bus terminal voltage, the battery is step-down discharge; otherwise, the battery is boosted discharge. When the vehicle is braking, if the battery terminal voltage is higher than the DC bus terminal voltage, the battery is boosted and charged; otherwise, the battery is stepped and charged. In addition, the vehicle to Grid (V2G) mode means that when the electric vehicle is left standing for a long time, the stored energy of the vehicle battery can be sold to the power grid, and when the vehicle battery needs to be charged, the power required for charging needs to be provided by the power grid. In the V2G charging mode, if the rectifier bus terminal voltage is higher than the battery terminal voltage, the battery is step-down charging; otherwise, the battery is boost charging. In the V2G discharge mode, if the rectifier bus terminal voltage is higher than the battery terminal voltage, the battery is boosted discharge; otherwise, the battery is step-down discharge. It can be seen that the bidirectional buck-boost DC-DC converter is also one of the important devices of electric vehicles.
一般的雙向直流-直流轉換器係以功率開關取代二極體，如圖3所示。由於傳統的雙向直流-直流轉換器只能在一方向升壓而在另一方向降壓，其使用的範圍會受到限制。有些文獻分析比較了傳統雙向轉換器、C’uk轉換器與SEPIC轉換器，其架構如圖4(a)-4(c)所示。傳統雙向轉換器的電路架構簡單且電路被動元件少，而效率也優於C’uk轉換器與SEPIC轉換器。若要將雙向直流-直流轉換器運用在電動車上，除了要克服體積與重量的限制，也要有在雙向靈活傳遞能量的能力。有些文獻提出非隔離型雙向直流-直流轉換器，如圖5所示，非隔離型雙向直流-直流轉換器改善了傳統型雙向直流-直流轉換器只能單向升降壓的缺點，另外有文獻提出的雙向直流-直流轉換器的電壓增益是傳統直流-直流轉換器的兩倍，而且開關上的電壓應力只有輸入電壓的一半，因而有更寬的電壓轉換範圍。有的文獻除了提出具零電壓切換能力之雙向直流-直流轉換器電路架構，也推導出功率與電路規格的關係式，更探討了在不同負載的情況下，相移控制機制對整體電路效率的影響。而且非隔離型雙向直流-直流轉換器相較於如圖6所示之隔離型雙向直流-直流轉換器，不僅電路架構簡單而且體積較小，更加適合應用在電動車上。A typical bidirectional DC-DC converter replaces the diode with a power switch, as shown in Figure 3. Since the conventional bidirectional DC-DC converter can only be boosted in one direction and stepped down in the other direction, its range of use is limited. Some literature analyses compare traditional bidirectional converters, C’uk converters and SEPIC converters, the architecture of which is shown in Figures 4(a)-4(c). Traditional bidirectional converters have a simpler circuit structure and fewer passive components, and are more efficient than C’uk converters and SEPIC converters. To apply a bidirectional DC-DC converter to an electric vehicle, in addition to overcoming the limitations of volume and weight, there is also the ability to flexibly transfer energy in both directions. Some literatures propose non-isolated bidirectional DC-DC converters. As shown in Figure 5, the non-isolated bidirectional DC-DC converters have improved the shortcomings of traditional bidirectional DC-DC converters that can only be unidirectionally buck-boost. The proposed bidirectional DC-DC converter has twice the voltage gain of a conventional DC-DC converter, and the voltage stress on the switch is only half of the input voltage, thus having a wider voltage conversion range. Some literatures not only propose a bidirectional DC-DC converter circuit architecture with zero voltage switching capability, but also derive the relationship between power and circuit specifications, and discuss the efficiency of the phase shift control mechanism for the overall circuit under different load conditions. influences. Moreover, the non-isolated bidirectional DC-DC converter has better circuit structure and smaller volume than the isolated bidirectional DC-DC converter shown in FIG. 6, and is more suitable for application in an electric vehicle.
本發明之主要目的在於提供一雙向升降壓直流-直流轉換器以應用在一微電網系統或一電動車中。SUMMARY OF THE INVENTION A primary object of the present invention is to provide a bidirectional buck-boost DC-DC converter for use in a microgrid system or an electric vehicle.
為達到上述目的，一種具自適應相移角控制機制之雙向直流-直流轉換器乃被提出，其具有：To achieve the above object, a bidirectional DC-DC converter with an adaptive phase shift angle control mechanism is proposed, which has:
一全橋式轉換電路，具有一第一半橋開關電路、一第二半橋開關電路、及耦接於該第一半橋開關電路和該第二半橋開關電路之間之一電感器，其中，該第一半橋開關電路具有二通道端以與一直流電壓源耦接、一第一控制端以與一第一控制信號耦接、以及一第二控制端以與一第二控制信號耦接，該第二半橋開關電路具有二通道端以與一電池單元耦接、一第三控制端以與一第三控制信號耦接、以及一第四控制端以與一第四控制信號耦接，以及該第一控制信號係與該第二控制信號電位互補且該第三控制信號係與該第四控制信號電位互補；a full bridge conversion circuit having a first half bridge switching circuit, a second half bridge switching circuit, and an inductor coupled between the first half bridge switching circuit and the second half bridge switching circuit, The first half bridge switch circuit has a two-channel end coupled to the DC voltage source, a first control terminal coupled to a first control signal, and a second control terminal coupled to a second control signal. The second half-bridge switch circuit has a two-channel terminal coupled to a battery unit, a third control terminal coupled to a third control signal, and a fourth control terminal coupled to a fourth control signal. Coupling, and the first control signal is complementary to the second control signal potential and the third control signal is complementary to the fourth control signal potential;
一第一信號處理電路，依該直流電壓源之一電壓產生一第一電壓信號；a first signal processing circuit for generating a first voltage signal according to a voltage of the DC voltage source;
一第二信號處理電路，依該電池單元之一電壓產生一第二電壓信號以及依該電池單元之一電流產生一電流信號；以及a second signal processing circuit for generating a second voltage signal according to a voltage of the battery unit and generating a current signal according to a current of the battery unit;
一控制單元，儲存有一韌體程式，用以執行一充電程序或一放電程序，該充電程序和該放電程序均包含藉由一比例-積分-微分運算調整該第一控制信號和該第三控制信號間之一相移角以對該全橋式轉換電路進行一電壓轉換操作，其中，該充電程序係依該第二電壓信號及該電流信號調整該相移角，且該放電程序係依該第一電壓信號調整該相移角。a control unit storing a firmware program for executing a charging program or a discharging program, the charging program and the discharging program each comprising adjusting the first control signal and the third control by a proportional-integral-derivative operation a phase shift angle between the signals to perform a voltage conversion operation on the full bridge conversion circuit, wherein the charging program adjusts the phase shift angle according to the second voltage signal and the current signal, and the discharging program is The first voltage signal adjusts the phase shift angle.
在一實施例中，該充電程序係一降壓程序或一升壓程序。In one embodiment, the charging program is a buck program or a boost program.
在一實施例中，該放電程序係一降壓程序或一升壓程序。In one embodiment, the discharge program is a buck program or a boost program.
在一實施例中，該充電程序和該放電程序均包含一類比至數位轉換運算。In one embodiment, both the charging program and the discharging program include an analog to digital conversion operation.
在一實施例中，該充電程序和該放電程序均進一步包含一濾波運算。In an embodiment, the charging program and the discharging program each further comprise a filtering operation.
為使 貴審查委員能進一步瞭解本發明之結構、特徵及其目的，茲附以圖式及較佳具體實施例之詳細說明如后。The detailed description of the drawings and the preferred embodiments are set forth in the accompanying drawings.
在雙向直流-直流轉換的應用中，能量的傳遞都需要透過一雙向升/降壓直流-直流轉換器，而電池端電壓的高低將會決定該雙向升/降壓直流-直流轉換器的操作模式。在電池充電時，假若電池端電壓低於匯流排端電壓，則該雙向升/降壓直流-直流轉換器會操作在降壓型充電模式；反之，則操作在升壓型充電模式。在電池放電時，假若電池端電壓低於匯流排端電壓，則該雙向升/降壓直流-直流轉換器會操作在升壓型放電模式；反之，則操作在降壓型放電模式。In bidirectional DC-DC conversion applications, energy transfer is required through a bidirectional step-up/step-down DC-DC converter, and the voltage at the battery terminal determines the operation of the bidirectional step-up/step-down DC-DC converter. mode. When the battery is charged, if the battery terminal voltage is lower than the bus terminal voltage, the bidirectional step-up/step-down DC-DC converter operates in the step-down charging mode; otherwise, it operates in the step-up charging mode. When the battery is discharged, if the battery terminal voltage is lower than the bus terminal voltage, the bidirectional step-up/step-down DC-DC converter operates in the boost mode discharge mode; otherwise, it operates in the buck mode discharge mode.
本發明之雙向升/降壓直流-直流轉換器主要分為兩部分，第一部分為硬體架構，其具有架構簡單、被動元件數少且可藉由一橋式功率開關組的切換改變能量傳輸方向等優點，使轉換器可操作於升/降壓型充/放電模式。第二部分為控制架構，因傳統類比轉換器控制元件容易受到工作環境溫度及濕度影響，造成元件特性老化，因此本發明使用一微處理器(例如Microchip公司針對切換式電源應用場合所開發之dsPIC33FJ16GS502微處理器)作為控制核心，將輸出電壓及電流經由信號處理電路回授至該微處理器，再經由該微處理器內部運算產生控制訊號來驅動該橋式功率開關組，達到雙向直流-直流轉換器數位化控制。本發明之轉換器可以適應性地調整該橋式功率開關組之對應驅動信號間的相移角以達到柔切(soft switching)驅動的效果，從而減少切換損耗、提高系統效率。The bidirectional step-up/step-down DC-DC converter of the invention is mainly divided into two parts. The first part is a hardware structure, which has a simple structure, a small number of passive components and can change the energy transmission direction by switching of a bridge type power switch group. The advantages are that the converter can operate in the up/down type charge/discharge mode. The second part is the control architecture. Because the traditional analog converter control components are susceptible to the temperature and humidity of the working environment, causing the components to deteriorate, the present invention uses a microprocessor (such as the dsPIC33FJ16GS502 developed by Microchip for switching power applications). As a control core, the microprocessor outputs the output voltage and current to the microprocessor via the signal processing circuit, and then generates a control signal through the internal operation of the microprocessor to drive the bridge power switch group to achieve bidirectional DC-DC. Converter digital control. The converter of the present invention can adaptively adjust the phase shift angle between the corresponding driving signals of the bridge power switch group to achieve the effect of soft switching driving, thereby reducing switching loss and improving system efficiency.
請參照圖7，其繪示本發明之具自適應相移角控制機制之雙向直流-直流轉換器之一實施例。如圖7所示，該雙向直流-直流轉換器具有一全橋式轉換電路100、一第一信號處理電路110、一第二信號處理電路120、以及一控制單元130。Referring to FIG. 7, an embodiment of a bidirectional DC-DC converter with adaptive phase shift angle control mechanism of the present invention is illustrated. As shown in FIG. 7, the bidirectional DC-DC converter has a full bridge conversion circuit 100, a first signal processing circuit 110, a second signal processing circuit 120, and a control unit 130.
全橋式轉換電路100具有一第一半橋開關電路101、一第二半橋開關電路102、及耦接於該第一半橋開關電路101和該第二半橋開關電路102之間之一電感器103，其中，該第一半橋開關電路101具有二通道端A、B以與一直流電壓源V DC耦接、一第一控制端以與一第一控制信號S 1耦接、以及一第二控制端以與一第二控制信號S 2耦接，該第二半橋開關電路102具有二通道端C、D以與一電池單元200耦接、一第三控制端以與一第三控制信號S 3耦接、以及一第四控制端以與一第四控制信號S 4耦接，以及該第一控制信號S 1係與該第二控制信號S 2電位互補且該第三控制信號S 3係與該第四控制信號S 4電位互補，亦即，一控制信號為高電位/低電位時，另一控制信號為低電位/高電位。 The full bridge conversion circuit 100 has a first half bridge switching circuit 101 , a second half bridge switching circuit 102 , and one of the first half bridge switching circuit 101 and the second half bridge switching circuit 102 . Inductor 103, wherein the first half bridge switch circuit 101 has two channel ends A, B coupled to the DC voltage source V DC , a first control terminal coupled to a first control signal S 1 , and A second control terminal is coupled to a second control signal S 2 , the second half bridge switch circuit 102 has two channel ends C, D coupled to a battery unit 200, and a third control terminal to The third control signal S 3 is coupled, and a fourth control terminal is coupled to a fourth control signal S 4 , and the first control signal S 1 is complementary to the second control signal S 2 and the third control The signal S 3 is complementary to the potential of the fourth control signal S 4 , that is, when one control signal is high/low, the other control signal is low/high.
第一信號處理電路110係依該直流電壓源V DC之一電壓V 1產生一第一電壓信號S V1。 The first signal processing circuit 110 generates a first voltage signal S V1 according to the voltage V 1 of the DC voltage source V DC .
第二信號處理電路120係依該電池單元200之一電壓V 2產生一第二電壓信號S V2以及依該電池單元200之一電流I產生一電流信號S I。 The second signal processing circuit 120 generates a second voltage signal S V2 according to a voltage V 2 of the battery unit 200 and generates a current signal S I according to a current I of the battery unit 200.
控制單元130，儲存有一韌體程式，用以執行一充電程序或一放電程序。該充電程序和該放電程序均包含一類比至數位轉換單元131、一濾波運算單元132、一比例-積分-微分運算單元133、以及一脈衝寬度調變運算單元134、以及一驅動單元135。另外，該充電程序可為一降壓程序或一升壓程序，且該放電程序可為一降壓程序或一升壓程序。The control unit 130 stores a firmware program for executing a charging program or a discharging program. The charging program and the discharging program each include an analog-to-digital conversion unit 131, a filtering operation unit 132, a proportional-integral-derivative operation unit 133, a pulse width modulation operation unit 134, and a drive unit 135. In addition, the charging program can be a buck program or a boost program, and the discharging program can be a buck program or a boost program.
類比至數位轉換單元131係用以對第一電壓信號S V1或第二電壓信號S V2、電流信號S I執行一類比-數位轉換運算；濾波運算單元132係用以對類比至數位轉換單元131之輸出執行一濾波運算；比例-積分-微分運算單元133係用以調整該第一控制信號S 1和該第三控制信號S 3間之一相移角，以經由脈衝寬度調變運算單元134及驅動單元135輸出第一控制信號S 1、第二控制信號S 2、第三控制信號S 3、以及第四控制信號S 4，從而對該全橋式轉換電路100進行一電壓轉換操作，其中，該充電程序係依該第二電壓信號S V2及該電流信號S I調整該相移角，且該放電程序係依該第一電壓信號S V1調整該相移角。 The analog-to-digital conversion unit 131 is configured to perform an analog-to-digital conversion operation on the first voltage signal S V1 or the second voltage signal S V2 and the current signal S I ; the filtering operation unit 132 is configured to analog to the digital conversion unit 131 The output performs a filtering operation; the proportional-integral-differential operation unit 133 is configured to adjust a phase shift angle between the first control signal S 1 and the third control signal S 3 to pass the pulse width modulation operation unit 134 And the driving unit 135 outputs the first control signal S 1 , the second control signal S 2 , the third control signal S 3 , and the fourth control signal S 4 to perform a voltage conversion operation on the full bridge conversion circuit 100, wherein The charging program adjusts the phase shift angle according to the second voltage signal S V2 and the current signal S I , and the discharging program adjusts the phase shift angle according to the first voltage signal S V1 .
以下將對本發明雙向升/降壓直流-直流轉換器的原理做詳細說明：The principle of the bidirectional step-up/step-down DC-DC converter of the present invention will be described in detail below:
I.硬體架構與電路操作分析I. Hardware Architecture and Circuit Operation Analysis
1.1硬體架構1.1 hardware architecture
本發明之全橋式轉換電路100，如圖8之示意圖所示，是由一顆電感、兩顆電容和四顆功率開關所組成。本發明係經由比較判斷電池端電壓與匯流排端電壓後，控制四顆功率開關切換，改變能量傳遞方向，讓所有功率開關均可操作在零電壓切換，以提升電路整體效率。The full bridge conversion circuit 100 of the present invention, as shown in the schematic diagram of FIG. 8, is composed of an inductor, two capacitors, and four power switches. The invention controls the four power switches by comparing and determining the battery terminal voltage and the bus terminal voltage, and changes the energy transmission direction, so that all the power switches can be operated at zero voltage switching to improve the overall efficiency of the circuit.
1.2充放電等效電路分析電路操作分析1.2 Charge and discharge equivalent circuit analysis circuit operation analysis
圖9為全橋式轉換電路100之等效模型，以下將對其充電模式與放電模式進行分析。FIG. 9 is an equivalent model of the full bridge conversion circuit 100. The charging mode and the discharging mode will be analyzed below.
1.2.1充電模式等效模型1.2.1 Charging mode equivalent model
充電模式有四個主要操作狀態：The charging mode has four main operating states:
充電模式狀態(一)：S 1、S 4導通，S 2、S 3截止； Charging mode state (1): S 1 , S 4 are turned on, and S 2 and S 3 are turned off;
充電模式狀態(二)：S 1、S 3導通，S 2、S 4截止； Charging mode state (2): S 1 , S 3 are turned on, and S 2 and S 4 are turned off;
充電模式狀態(三)：S 2、S 3導通，S 1、S 4截止； Charging mode state (3): S 2 and S 3 are turned on, and S 1 and S 4 are turned off;
充電模式狀態(四)：S 2、S 4導通，S 1、S 3截止。 Charging mode state (4): S 2 and S 4 are turned on, and S 1 and S 3 are turned off.
接下來將針對充電模式四個狀態進行分析。Next, four states of the charging mode will be analyzed.
充電模式狀態(一)：Charging mode status (1):
如圖10(a)所示，在充電模式狀態(一)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 10(a), in the state of charge mode (1), the following equation of state can be obtained via the Khöhoff voltage and current law:
(1) (1)
(2) (2)
(3) (3)
充電模式狀態(二)：Charging mode status (2):
如圖10(b)所示，在充電模式狀態(二)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 10(b), in the charging mode state (2), the following equation of state can be obtained via the Khöhehoff voltage and current law:
(4) (4)
(5) (5)
(6) (6)
充電模式狀態(三)：Charging mode status (3):
如圖10(c)所示，在充電模式狀態(三)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 10(c), in the state of charge mode (3), the following equation of state can be obtained via Kelschoff's voltage and current law:
(7) (7)
(8) (8)
(9) (9)
充電模式狀態(四)：Charging mode status (4):
如圖10(d)所示，在充電模式狀態(四)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 10(d), in the state of charge mode (4), the following equation of state can be obtained via the Khöhehoff voltage and current law:
(10) (10)
(11) (11)
(12) (12)
充電模式狀態由(1)至(12)式，可推導出 、 、 在一個週期內之平均值分別如(13)至(15)式所示，其中 為 、 為 、 為 、 為 、 、 為開關S 1與S 3的責任週期。 The charging mode status is from (1) to (12) and can be derived. , , The average value over a period is shown in (13) to (15), respectively. for , for , for , for , , The duty cycle for switches S 1 and S 3 .
(13) (13)
(14) (14)
(15) (15)
根據電感的伏特-秒平衡定律，可推導出(16)式。According to the volt-second equilibrium law of the inductance, equation (16) can be derived.
(16) (16)
整理(16)式後，可得電池端電壓與匯流排端電壓之關係式為After finishing (16), the relationship between the battery terminal voltage and the bus terminal voltage is
(17) (17)
在充電模式中，開關S 1、S 2及S 3、S 4均為互補訊號，開關S 3與S 4責任週期為固定50 %，由(17)式可知，若要達成降壓之功能，開關S 1責任週期必須小於50 %，亦即S 3和S 1之間須有一相移角f CD使開關S 1的責任週期小於50 %；若要達成升壓之功能，開關S 1責任週期必須大於50 %，亦即S 3和S 1之間須有一相移角f CU使開關S 1的責任週期大於50 %。 In the charging mode, the switches S 1 , S 2 and S 3 , S 4 are all complementary signals, and the duty cycle of the switches S 3 and S 4 is fixed at 50%. From (17), it can be known that if the function of step-down is to be achieved, The duty cycle of switch S 1 must be less than 50%, that is, there must be a phase shift angle f CD between S 3 and S 1 to make the duty cycle of switch S 1 less than 50%; to achieve the boost function, switch S 1 duty cycle Must be greater than 50%, that is, there must be a phase shift angle f CU between S 3 and S 1 to make the duty cycle of switch S 1 greater than 50%.
1.2.2放電模式等效模型1.2.2 discharge mode equivalent model
放電模式有四個主要操作狀態：The discharge mode has four main operating states:
放電模式狀態(一)：S 2、S 3導通，S 1、S 4截止； Discharge mode state (1): S 2 and S 3 are turned on, and S 1 and S 4 are turned off;
放電模式狀態(二)：S 1、S 3導通，S 2、S 4截止； Discharge mode state (2): S 1 and S 3 are turned on, and S 2 and S 4 are turned off;
放電模式狀態(三)：S 1、S 4導通，S 2、S 3截止； Discharge mode state (3): S 1 , S 4 are turned on, and S 2 and S 3 are turned off;
放電模式狀態(四)：S 2、S 4導通，S 1、S 3截止。 Discharge mode state (4): S 2 and S 4 are turned on, and S 1 and S 3 are turned off.
接下來將針對放電模式四個狀態進行分析。Next, four states of the discharge mode will be analyzed.
放電模式狀態(一)：Discharge mode state (1):
如圖11(a)所示，在放電模式狀態(一)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 11(a), in the discharge mode state (1), the following equation of state can be obtained via the Khöhoff voltage and current law:
(18) (18)
(19) (19)
(20) (20)
放電模式狀態(二)Discharge mode state (2)
如圖11(b)所示，在放電模式狀態(二)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 11(b), in the discharge mode state (2), the following equation of state can be obtained via the Khöhehoff voltage and current law:
(21) (twenty one)
(22) (twenty two)
(23) (twenty three)
放電模式狀態(三)Discharge mode state (3)
如圖11(c)所示，在放電模式狀態(三)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 11(c), in the discharge mode state (3), the following equation of state can be obtained via the Khöhoff voltage and current law:
(24) (twenty four)
(25) (25)
(26) (26)
放電模式狀態(四)Discharge mode state (4)
如圖11(d)所示，在放電模式狀態(四)中，經由克希荷夫電壓及電流定律可得到以下之狀態方程式：As shown in Fig. 11(d), in the discharge mode state (4), the following equation of state can be obtained via the Khöhehoff voltage and current law:
(27) (27)
(28) (28)
(29) (29)
放電模式由(18)至(29)式，可推導出 、 、 在一週期內之平均值為(30)至(32)式，其中 為 、 為 、 為 、 為 、 、 為開關S 1與S 3的責任週期。 The discharge mode is from (18) to (29) and can be derived. , , The average value over the one-week period is (30) to (32), where for , for , for , for , , The duty cycle for switches S 1 and S 3 .
(30) (30)
(31) (31)
(32) (32)
根據電感的伏特-秒平衡定律，可推導出(33)式。According to the volt-second equilibrium law of the inductance, equation (33) can be derived.
(33) (33)
整理(33)式後，可得匯流排端電壓與電池端電壓之關係式。After finishing the (33) type, the relationship between the bus terminal voltage and the battery terminal voltage can be obtained.
(34) (34)
在放電模式中，開關S 1、S 2及S 3、S 4均為互補訊號，開關S 1與S 2的責任週期為固定50 %，由(34)式可得知，若要達成降壓之功能，開關S 3的責任週期必須小於50 %，亦即S 1和S 3之間須有一相移角f DD使開關S 3的責任週期小於50 %；若要達成升壓之功能，開關S 3責任週期必須大於50 %，亦即S 1和S 3之間須有一相移角f DU使開關S 3的責任週期大於50 %。 In the discharge mode, the switches S 1 , S 2 and S 3 , S 4 are complementary signals, and the duty cycle of the switches S 1 and S 2 is fixed at 50%, which can be known from the equation (34). The function, the duty cycle of switch S 3 must be less than 50%, that is, there must be a phase shift angle f DD between S 1 and S 3 to make the duty cycle of switch S 3 less than 50%; to achieve the boost function, the switch The S 3 duty cycle must be greater than 50%, that is, there must be a phase shift angle f DU between S 1 and S 3 such that the duty cycle of switch S 3 is greater than 50%.
1.3雙向升降壓直流-直流轉換器動作分析1.3 Bidirectional buck-boost DC-DC converter action analysis
1.3.1充電模式動作分析1.3.1 Charging mode action analysis
為防止開關同時導通造成電路誤動作，因此在開關切換時加入延遲時間(Dead Time)，而藉由S 1～S ４導通切換以達成充電動作。充電模式可細分成十二個時序狀態。 In order to prevent the circuit from malfunctioning due to the simultaneous conduction of the switch, a delay time (Dead Time) is added during the switching of the switch, and the switching is performed by S 1 - S 4 to achieve the charging operation. The charge mode can be subdivided into twelve timing states.
狀態一 ( t 0≤ t ＜ t 1 )： State one ( t 0 ≤ t < t 1 ):
在 t=t 0 時，等效電路如圖12(a)所示，由於上一狀態D b-s1導通，開關S 1零電壓導通，S 4則是延續上一狀態持續導通，S 2、S 3截止，電感上的跨壓為V bus，匯流排端提供能量對電感儲能，電感電流呈線性上升，負載端能量由C bat提供。當S 4截止則進入下一狀態。 When t=t 0 , the equivalent circuit is as shown in Fig. 12(a). Since the previous state D b-s1 is turned on, the zero-voltage of the switch S 1 is turned on, and the S 4 is continuously turned on in the previous state, S 2 , S 3 cutoff, the voltage across the inductor is V bus , the energy at the bus bar provides energy to the inductor, the inductor current rises linearly, and the energy at the load end is provided by C bat . When S 4 is turned off, it enters the next state.
狀態二 ( t 1≤ t ＜ t 2 )： State two ( t 1 ≤ t < t 2 ):
在 t=t 1 時，等效電路如圖12(b)所示，開關S 1導通，S 2、S 3、S 4截止。依據楞次定律，電感電流必須保持連續性，故電感電流延續上一狀態之電流方向持續流動，因此電感電流對C oss3放電、對C oss4充電，直到C oss3兩端電壓放至零、C oss4兩端電壓充至V bat，開關S 3的本體二極體D b-s3才會導通。當D b-s3導通則進入下一個狀態。 When t = t 1 , the equivalent circuit is as shown in Fig. 12 (b), the switch S 1 is turned on, and S 2 , S 3 , and S 4 are turned off. According to Lenz's law, the inductor current continuity must be maintained, so that the inductor current continues the current state of a continuously flowing direction, thus discharging the inductor current C oss3 for C oss4 charged until the voltage across C oss3 put to zero, C oss4 When the voltage at both ends is charged to V bat , the body diode D b-s3 of the switch S 3 is turned on. When D b-s3 is turned on, it enters the next state.
狀態三 ( t 2≤ t ＜ t 3 )： State three ( t 2 ≤ t < t 3 ):
在 t=t 2 時，等效電路如圖12(c)所示，開關S 1導通，S 2、S 3、S 4截止。由於開關S 3的本體二極體D b-s3導通，S 3上的跨壓箝位在接近零電位，若能立即導通開關S 3，S 3會操作在零電壓切換，若開關S 3未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當S 3導通則進入下一狀態。 When t = t 2 , the equivalent circuit is as shown in Fig. 12 (c), the switch S 1 is turned on, and S 2 , S 3 , and S 4 are turned off. Since the body diode D b-s3 of the switch S 3 is turned on, the voltage across the clamp on S 3 is close to zero potential. If the switch S 3 can be turned on immediately, S 3 will operate at zero voltage switching if the switch S 3 is not Immediate turn-on will cause additional switching losses, reducing overall circuit efficiency. When S 3 is turned on, it enters the next state.
狀態四 ( t 3≤ t ＜ t 4 )： State four ( t 3 ≤ t < t 4 ):
在 t=t 3 時，等效電路如圖12(d)所示，開關S 1延續上一狀態持續導通、S 3零電壓導通，S 2、S 4截止，電感上的跨壓為V bus-V bat，若是降壓型，V bus電壓大於V bat，因此電感繼續儲能，電感電流呈線性上升。若是升壓型，V bus電壓小於V bat，因此電感開始釋能，電感電流呈線性下降。而匯流排端能量經由S 1及S 3對C bat充電，並提供能量給負載。當S 1截止則進入下一狀態。 When t=t 3 , the equivalent circuit is as shown in Fig. 12(d), the switch S 1 continues to be in the previous state, the S 3 zero voltage is turned on, the S 2 and S 4 are turned off, and the voltage across the inductor is V bus . -V bat , if it is step-down, the V bus voltage is greater than V bat , so the inductor continues to store energy and the inductor current rises linearly. If it is a boost type, the V bus voltage is less than V bat , so the inductor begins to release energy and the inductor current decreases linearly. The bus end energy charges C bat via S 1 and S 3 and provides energy to the load. When S 1 is turned off, it goes to the next state.
狀態五 ( t 4≤ t ＜ t 5 )： State five ( t 4 ≤ t < t 5 ):
在 t=t 4 時，等效電路如圖12(e)所示，開關S 3導通，S 1、S 2、S 4截止，電感電流需保持連續性，故電感電流延續上一狀態之電流方向持續流動，因此電感電流對C oss1充電、C oss2放電，直到C oss1兩端電壓充至V bus、C oss2兩端電壓放至零，開關S 2的本體二極體D b-s2才會導通。當D b-s2導通則進入下一狀態。 When t=t 4 , the equivalent circuit is as shown in Fig. 12(e), the switch S 3 is turned on, S 1 , S 2 , and S 4 are turned off, and the inductor current needs to be continuous, so the inductor current continues the current in the previous state. The direction continues to flow, so the inductor current charges C oss1 and C oss2 , until the voltage across C oss1 is charged to V bus , the voltage across C oss2 is zero, and the body diode D b-s2 of switch S 2 will Turn on. When D b-s2 is turned on, it enters the next state.
狀態六 ( t 5≤ t ＜ t 6 )： State six ( t 5 ≤ t < t 6 ):
在 t=t 5 時，等效電路如圖12(f)所示，開關S 3導通，S 1、S 2、S 4截止，由於開關S 2的本體二極體D b-s2導通，S 2上的跨壓箝位在接近零電位，若能立即導通開關S 2，S 2會操作在零電壓切換，若開關S 2未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當S 2導通則進入下一狀態。 When t=t 5 , the equivalent circuit is as shown in Fig. 12(f), the switch S 3 is turned on, and S 1 , S 2 , and S 4 are turned off, since the body diode D b-s2 of the switch S 2 is turned on, S 2 clamp the voltage across the potential close to zero, if the switch is turned on immediately S 2, S 2 will operate in zero voltage switching, when the switch S 2 is not turned on immediately, will result in additional switching losses, reducing the overall efficiency of the circuit . When S 2 is turned on, it enters the next state.
狀態七 ( t 6≤ t ＜ t 7 )： State seven ( t 6 ≤ t < t 7 ):
在 t=t 6 時，等效電路如圖12(g)所示，開關S 2零電壓導通、S 3則是延續上一個狀態導通，S 1、S 4截止，電感上的跨壓為-V bat，因此電感開始釋能，電感電流呈線性下降，並提供能量給負載端使用。在此狀態必須有足夠的時間讓電感電流持續下降到負，讓電感電流改變方向，否則無法讓下一狀態開關達到零電壓切換。當S 3截止則進入下一狀態。 At t = t 6, the equivalent circuit of FIG. 12 (g), the zero voltage switch S 2 is turned on, S 3 is a continuation of a turn-on state, S 1, S 4 is turned off, the voltage across the inductor is - V bat , so the inductor begins to release energy, the inductor current drops linearly, and provides energy to the load end. In this state, there must be enough time for the inductor current to continue to drop to negative, so that the inductor current changes direction, otherwise the next state switch cannot be switched to zero voltage. When S 3 is turned off, the next state is entered.
狀態八 ( t 7≤ t ＜ t 8 )： State eight ( t 7 ≤ t < t 8 ):
在 t=t 7 時，等效電路如圖12(h)所示，開關S 2導通，S 1、S 3、S 4截止。電感電流在上一狀態改變電流方向，為保持連續性，因此電感電流對C oss3充電、C oss4放電，直到C oss3兩端電壓充至V bat、C oss4兩端電壓放至零，開關S 4的本體二極體D b-s4才會導通。當D b-s4導通則進入下一狀態。 At t = t 7, the equivalent circuit in FIG. 12 (h), the switch S 2 is turned on, S 1, S 3, S 4 are turned off. The inductor current changes the current direction in the previous state. To maintain continuity, the inductor current charges C oss3 and C oss4 discharges until the voltage across C oss3 is charged to V bat and the voltage across C oss4 is zero. Switch S 4 The body diode D b-s4 will be turned on. When D b-s4 is turned on, it enters the next state.
狀態九 ( t 8≤ t ＜ t 9 )： State nine ( t 8 ≤ t < t 9 ):
在 t=t 8 時，等效電路如圖12(i)所示，開關S 2導通，S 1、S 3、S 4截止。由於開關S 4的本體二極體D b-s4導通，S 4上的跨壓箝位在接近零電位，若能立即導通開關S 4，S 4會操作在零電壓切換，若開關S 4未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當S 4導通則進入下一狀態。 When t = t 8 , the equivalent circuit is as shown in Fig. 12 (i), the switch S 2 is turned on, and S 1 , S 3 , and S 4 are turned off. Since the body diode D b-s4 of the switch S 4 is turned on, the voltage across the S 4 clamp is close to zero potential. If the switch S 4 can be turned on immediately, the S 4 will operate at zero voltage switching if the switch S 4 is not Immediate turn-on will cause additional switching losses, reducing overall circuit efficiency. When S 4 is turned on, it enters the next state.
狀態十 ( t 9≤ t ＜ t 10 )： State ten ( t 9 ≤ t < t 10 ):
在 t=t 9 時，等效電路如圖12(j)所示，開關S 2延續上一狀態持續導通、S 4零電壓導通，S 1、S 3截止。此時電感上的跨壓為零，電感電流方向保持不變，負載端能量由C bat提供。當S 2截止則進入下一狀態。 At t = t 9, the equivalent circuit of FIG. 12 (j), the switch S 2 continued on a continuous conduction state, S 4 zero voltage, S 1, S 3 is turned off. At this time, the voltage across the inductor is zero, the direction of the inductor current remains unchanged, and the energy at the load end is provided by C bat . When S 2 is turned off, it goes to the next state.
狀態十一 ( t 10≤ t ＜ t 11 )： State eleven ( t 10 ≤ t < t 11 ):
在 t=t 10 時，等效電路如圖12(k)所示，開關S 4導通，S 1、S 2、S 3截止。電感電流需保持連續性，故電感電流延續上一狀態之電流方向持續流動，因此電感電流對C oss1放電、C oss2充電，直到C oss1兩端電壓放至零、C oss2兩端電壓充至V bus，開關S 1的本體二極體D b-s1才會導通。當D b-s1導通則進入下一狀態。 When t=t 10 , the equivalent circuit is as shown in Fig. 12(k), the switch S 4 is turned on, and S 1 , S 2 , and S 3 are turned off. Inductor current required to maintain the continuity of the continuation of the inductor current so that the current direction of a state of continuous flow, and therefore the inductor current to discharge C oss1, C oss2 charge until the voltage across C oss1 put to zero, the voltage across C oss2 charged to V Bus , the body diode D b-s1 of the switch S 1 will be turned on. When D b-s1 is turned on, it enters the next state.
狀態十二 ( t 11≤ t ＜ t 12 )： State twelve ( t 11 ≤ t < t 12 ):
在 t=t 11 時，等效電路如12(l)所示，開關S 4導通，S 1、S 2、S 3截止。由於開關S 1的本體二極體D b-s1導通，S 1上的跨壓箝位在接近零電位，若能立即導通開關S 1，S 1會操作在零電壓切換，若開關S 1未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當t=t 12時，完成充電模式一個切換週期，並回到t=t 1，持續循環相同電路動作。在充電模式一個切換週期內，四個功率開關均可達到零電壓切換。 At t = t 11, the equivalent circuit 12 (l), the switch is turned on S 4, S 1, S 2, S 3 is turned off. Since the body diode D b-s1 of the switch S 1 is turned on, the voltage across the clamp on S 1 is close to zero potential. If the switch S 1 can be turned on immediately, S 1 will operate at zero voltage switching if the switch S 1 is not Immediate turn-on will cause additional switching losses, reducing overall circuit efficiency. When t=t 12 , the charging mode is completed for one switching cycle, and returns to t=t 1 to continuously cycle the same circuit action. Four power switches can achieve zero voltage switching during one switching cycle of the charging mode.
1.3.2放電模式動作分析1.3.2 Discharge mode action analysis
為防止開關同時導通造成電路誤動作，因此在開關切換時加入延遲時間，而藉由S 1～S ４導通切換以達成放電動作。放電模式可細分成十二個時序狀態。 In order to prevent the switch from being turned on at the same time, the circuit is malfunctioned. Therefore, the delay time is added when the switch is switched, and the switching is performed by S 1 to S 4 to achieve the discharge operation. The discharge mode can be subdivided into twelve timing states.
狀態一 ( t 0≤ t ＜ t 1 )： State one ( t 0 ≤ t < t 1 ):
在 t=t 0 時，等效電路如圖13(a)所示，開關S 2延續上一狀態持續導通，由於上一狀態D b-s3導通，S 3零電壓導通，S 1、S 4截止，電感上的跨壓為V bat，電池端提供能量對電感儲能，電感電流呈線性上升，負載端能量由C bus提供。當S 2截止則進入下一狀態。 When t=t 0 , the equivalent circuit is as shown in Fig. 13(a), the switch S 2 continues to be turned on in the previous state, and since the previous state D b-s3 is turned on, the S 3 zero voltage is turned on, S 1 , S 4 At the cutoff, the voltage across the inductor is V bat , the battery terminal provides energy to the inductor to store energy, the inductor current rises linearly, and the load end energy is provided by C bus . When S 2 is turned off, it goes to the next state.
狀態二 ( t 1≤ t ＜ t 2 )： State two ( t 1 ≤ t < t 2 ):
在 t=t 1 時，等效電路如圖13(b)所示，開關S 3導通，S 1、S 2、S 4截止。依據楞次定律，電感電流必須保持連續性，故電感電流延續上一狀態之電流方向持續流動，因此電感電流對C oss1放電、對C oss2充電，直到C oss1兩端電壓放至零、C oss2兩端電壓充至V bus，開關S 1的本體二極體D b-s1才會導通。當D b-s1導通則進入下一狀態。 When t = t 1 , the equivalent circuit is as shown in Fig. 13 (b), the switch S 3 is turned on, and S 1 , S 2 , and S 4 are turned off. According to Lenz's law, the inductor current continuity must be maintained, so that the inductor current is a continuation of the direction of the current state of continuous flow, and therefore the inductor current to discharge C oss1 for C oss2 charged until the voltage across C oss1 put to zero, C oss2 The voltage at both ends is charged to V bus , and the body diode D b-s1 of the switch S 1 is turned on. When D b-s1 is turned on, it enters the next state.
狀態三 ( t 2≤ t ＜ t 3 )： State three ( t 2 ≤ t < t 3 ):
在 t=t 2 時，等效電路如圖13(c)所示，開關S 3導通，S 1、S 2、S 4截止。由於開關S 1的本體二極體D b-s1導通，S 1上的跨壓箝位在接近零電位，若能立即導通開關S 1，S 1會操作在零電壓切換，若開關S 1未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當S 1導通則進入下一狀態。 When t = t 2 , the equivalent circuit is as shown in Fig. 13 (c), the switch S 3 is turned on, and S 1 , S 2 , and S 4 are turned off. Since the body diode D b-s1 of the switch S 1 is turned on, the voltage across the clamp on S 1 is close to zero potential. If the switch S 1 can be turned on immediately, S 1 will operate at zero voltage switching if the switch S 1 is not Immediate turn-on will cause additional switching losses, reducing overall circuit efficiency. When S 1 is turned on, it enters the next state.
狀態四 ( t 3≤ t ＜ t 4 )： State four ( t 3 ≤ t < t 4 ):
在 t=t 3 時，等效電路如圖13(d)所示，開關S 3延續上一狀態持續導通、S 1零電壓導通，S 2、S 4截止，電感上的跨壓為V bat-V bus，若是降壓型，V bat電壓大於V bus，因此電感繼續儲能，電感電流呈線性上升。若是升壓型，V bat電壓小於V bus，因此電感開始釋能，電感電流呈線性下降。而電池端能量經由S 3及S 1對C bus充電，並提供能量給負載。當S 3截止則進入下一狀態。 When t=t 3 , the equivalent circuit is as shown in Fig. 13(d), the switch S 3 continues to be on in the previous state, the S 1 zero voltage is turned on, the S 2 and S 4 are turned off, and the voltage across the inductor is V bat -V bus , if it is step-down, the V bat voltage is greater than V bus , so the inductor continues to store energy and the inductor current rises linearly. In the case of boost mode, the V bat voltage is less than V bus , so the inductor begins to release energy and the inductor current decreases linearly. The battery end energy charges the C bus via S 3 and S 1 and provides energy to the load. When S 3 is turned off, the next state is entered.
狀態五 ( t 4≤ t ＜ t 5 )： State five ( t 4 ≤ t < t 5 ):
在 t=t 4 時，等效電路如圖13(e)所示，開關S 1導通，S 2、S 3、S 4截止，電感電流需保持連續性，故電感電流延續上一狀態之電流方向持續流動，因此電感電流對C oss3充電、C oss4放電，直到C oss3兩端電壓充至V bat、C oss4兩端電壓放至零，開關S 4的本體二極體D b-s4才會導通。當D b-s4導通則進入下一狀態。 When t=t 4 , the equivalent circuit is as shown in Fig. 13(e), the switch S 1 is turned on, S 2 , S 3 , and S 4 are turned off, and the inductor current needs to be continuous, so the inductor current continues the current in the previous state. The direction continues to flow, so the inductor current charges C oss3 and C oss4 discharges until the voltage across C oss3 is charged to V bat and the voltage across C oss4 is zero, and the body diode D b-s4 of switch S 4 will Turn on. When D b-s4 is turned on, it enters the next state.
狀態六 ( t 5≤ t ＜ t 6 )： State six ( t 5 ≤ t < t 6 ):
在 t=t 5 時，等效電路如圖13(f)所示，開關S 1導通，S 2、S 3、S 4截止，由於開關S 4的本體二極體D b-s4導通，S 4上的跨壓箝位在接近零電位，若能立即導通開關S 4，S 4會操作在零電壓切換，若開關S 4未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當S 4導通則進入下一狀態。 When t=t 5 , the equivalent circuit is as shown in Fig. 13(f), the switch S 1 is turned on, and S 2 , S 3 , and S 4 are turned off, since the body diode D b-s4 of the switch S 4 is turned on, S the voltage across the clamp 4 close to zero potential, if immediately turn on the switch S 4, S 4 will be operating at the zero-voltage switching, when the switch S 4 is not turned on immediately, will result in additional switching losses, so reducing the overall efficiency of the circuit . When S 4 is turned on, it enters the next state.
狀態七 ( t 6≤ t ＜ t 7 )： State seven ( t 6 ≤ t < t 7 ):
在 t=t 6 時，等效電路如圖13(g)所示，S 1是延續上一個狀態導通、開關S 4零電壓導通，S 2、S 3截止，電感上的跨壓為-V bus，因此電感開始釋能，電感電流呈線性下降，並提供能量給負載端使用。在此狀態必須有足夠的時間讓電感電流持續下降到負，讓電感電流改變方向，否則無法讓下一狀態開關達到零電壓切換。當S 1截止則進入下一狀態。 When t=t 6 , the equivalent circuit is as shown in Fig. 13(g), S 1 is continuous in the previous state, zero voltage is turned on in switch S 4 , S 2 and S 3 are off, and the voltage across the inductor is -V Bus , so the inductor begins to release energy, the inductor current drops linearly, and provides energy for the load end. In this state, there must be enough time for the inductor current to continue to drop to negative, so that the inductor current changes direction, otherwise the next state switch cannot be switched to zero voltage. When S 1 is turned off, it goes to the next state.
狀態八 ( t 7≤ t ＜ t 8 )： State eight ( t 7 ≤ t < t 8 ):
在 t=t 7 時，等效電路如圖13(h)所示，開關S 4導通，S 1、S 2、S 3截止。電感電流在上一狀態改變電流方向，為保持連續性，因此電感電流對C oss1充電、C oss2放電，直到C oss1兩端電壓充至V bus、C oss2兩端電壓放至零，開關S 2的本體二極體D b-s2才會導通。當D b-s2導通則進入下一狀態。 At t = t 7, the equivalent circuit of FIG. 13 (h), the switch is turned on S 4, S 1, S 2, S 3 is turned off. The inductor current changes the current direction in the previous state. In order to maintain continuity, the inductor current charges C oss1 and C oss2 discharges until the voltage across C oss1 is charged to V bus , and the voltage across C oss2 is set to zero. Switch S 2 The body diode D b-s2 will be turned on. When D b-s2 is turned on, it enters the next state.
狀態九 ( t 8≤ t ＜ t 9 )： State nine ( t 8 ≤ t < t 9 ):
在 t=t 8 時，等效電路如圖13(i)所示，開關S 4導通，S 1、S 2、S 3截止。由於開關S 2的本體二極體D b-s2導通，S 2上的跨壓箝位在接近零電位，若能立即導通開關S 2，S 2會操作在零電壓切換，若開關S 2未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當S 2導通則進入下一狀態。 At t = t 8, the equivalent circuit of FIG. 13 (i), the switch is turned on S 4, S 1, S 2, S 3 is turned off. Since the body diode D b-s2 of the switch S 2 is turned on, the voltage across the S 2 clamp is close to zero potential. If the switch S 2 can be turned on immediately, the S 2 will operate at zero voltage switching if the switch S 2 is not Immediate turn-on will cause additional switching losses, reducing overall circuit efficiency. When S 2 is turned on, it enters the next state.
狀態十 ( t 9≤ t ＜ t 10 )： State ten ( t 9 ≤ t < t 10 ):
在 t=t 9 時，等效電路如圖13(j)所示，開關S 2零電壓導通、S 4則是延續上一狀態持續導通，S 1、S 3截止。此時電感上的跨壓為零，電感電流方向保持不變，負載端能量由C bus提供。當S 4截止則進入下一狀態。 At t = t 9, the equivalent circuit of FIG. 13 (j), the zero voltage switch S 2 is turned on, S 4 is a continuation of the continuous conduction state, S 1, S 3 is turned off. At this time, the voltage across the inductor is zero, the direction of the inductor current remains unchanged, and the energy at the load end is provided by C bus . When S 4 is turned off, it enters the next state.
狀態十一 ( t 10≤ t ＜ t 11 )： State eleven ( t 10 ≤ t < t 11 ):
在 t=t 10 時，等效電路如圖13(k)所示，開關S 2導通，S 1、S 3、S 4截止。電感電流需保持連續性，故電感電流延續上一狀態之電流方向持續流動，因此電感電流對C oss3放電、C oss4充電，直到C oss3兩端電壓放至零、C oss4兩端電壓充至V bat，開關S 3的本體二極體D b-s3才會導通。當D b-s3導通則進入下一狀態。 When t = t 10 , the equivalent circuit is as shown in Fig. 13 (k), the switch S 2 is turned on, and S 1 , S 3 , and S 4 are turned off. The inductor current needs to be continuous, so the inductor current continues to flow in the current direction of the previous state. Therefore, the inductor current charges C oss3 and C oss4 until the voltage across C oss3 is zero, and the voltage across C oss4 is charged to V. Bat , the body diode D b-s3 of the switch S 3 will be turned on. When D b-s3 is turned on, it enters the next state.
狀態十二 ( t 11≤ t ＜ t 12 )： State twelve ( t 11 ≤ t < t 12 ):
在 t=t 11 時，等效電路如圖13(l)所示，開關S 2導通，S 1、S 3、S 4截止。由於開關S 3的本體二極體D b-s3導通，S 3上的跨壓箝位在接近零電位，若能立即導通開關S 3，S 3會操作在零電壓切換，若開關S 3未立即導通，將會造成額外的開關損耗，使電路整體效率降低。當t=t 12時，完成放電模式一個切換週期，並回到t=t 1，持續循環相同電路動作，在放電模式一個切換週期內，四個功率開關均可達到零電壓切換。 In time t = t 11, the equivalent circuit of FIG. 13 (l), the switch S 2 is turned on, S 1, S 3, S 4 are turned off. Since the body diode D b-s3 of the switch S 3 is turned on, the voltage across the clamp on S 3 is close to zero potential. If the switch S 3 can be turned on immediately, S 3 will operate at zero voltage switching if the switch S 3 is not Immediate turn-on will cause additional switching losses, reducing overall circuit efficiency. When t=t 12 , the discharge mode is completed for one switching cycle, and returns to t=t 1 to continuously cycle the same circuit action. In one switching cycle of the discharge mode, the four power switches can reach zero voltage switching.
II相移機制II phase shift mechanism
根據前面所描述之模式分析可得知，在降壓型充電模式，功率開關信號S 3會對S 1相移固定角度f CD；在升壓型充電模式，功率開關信號S 3則是對S 1相移固定角度f CU。在降壓型放電模式，功率開關信號S 1會對S 3相移固定角度f DD；在升壓型放電模式，功率開關信號S 1則是對S 3相移固定角度f DU。而最佳相移角度f的大小則可由雙向升降壓直流-直流轉換器輸出功率計算得到。 According to the mode analysis described above, in the step-down charging mode, the power switch signal S 3 will be phase shifted by a fixed angle f CD for S 1 ; in the boost type charging mode, the power switch signal S 3 is for S 1 phase shift fixed angle f CU . In the buck mode discharge mode, the power switch signal S 1 is phase shifted by a fixed angle f DD for S 3 ; in the boost mode discharge mode, the power switch signal S 1 is phase shifted by a fixed angle f DU for S 3 . The optimum phase shift angle f can be calculated from the output power of the bidirectional buck-boost DC-DC converter.
2.1相移原理與分析2.1 Phase shift principle and analysis
在相同輸入電壓、輸出電壓但不同負載情況下，令i LL(t)為輕載電感電流、i LH(t)為滿載電感電流。由於相移角度f固定，因此輕載電感電流與滿載電感電流的峰對峰值及有效值相同。然而固定相移角度會造成轉換器在輕載效率不佳，其主因為輕載時，過大的電感電流有效值會使得電感損耗和功率開關導通損變大，使得電路在輕載狀況下效率不彰。 For the same input voltage, output voltage but different load, let i LL (t) be the light load inductor current and i LH (t) be the full load inductor current. Since the phase shift angle f is fixed, the peak-to-peak value and the effective value of the light-load inductor current and the full-load inductor current are the same. However, the fixed phase shift angle will cause the converter to have poor light load efficiency. Because of the light load, the excessive IGBT effective value will make the inductor loss and power switch conduction loss become larger, making the circuit less efficient under light load conditions. Zhang.
在相同輸入電壓、輸出電壓與負載狀況下，相移角度越大，可傳輸功率、電感電流的峰對峰值及有效值越大。然而過大的相移角度會使得電感損耗和功率開關導通損變大，改善此損耗的方法有兩種，其一為增加電感量，使電感的峰對峰值及有效值降低，但電感值增加，電感銅損及成本也隨之提升；另一種方法為隨著輸出負載改變而調變相移角度，使功率開關流過較低的電感電流有效值，降低電感損耗和功率開關導通損。當輸出負載增加，電感電流直流準位向上提升，使得電感電流無法下降到負值，讓電感電流改變方向，導致開關無法達到零電壓切換，因此可採取隨著輸出負載改變而調變相移角度，除了可以解決無法零電壓切換之問題，也可以提升轉換器在各種負載情況的整體效率。Under the same input voltage, output voltage and load condition, the larger the phase shift angle, the larger the peak-to-peak value and the effective value of the transmittable power and the inductor current. However, an excessive phase shift angle will cause the inductor loss and the power switch conduction loss to be large. There are two ways to improve the loss. One is to increase the inductance, so that the peak-to-peak value and the effective value of the inductor are reduced, but the inductance value is increased. Inductor copper loss and cost are also increased; another method is to adjust the phase shift angle as the output load changes, so that the power switch flows through the lower inductor current RMS, reducing the inductor loss and power switch conduction loss. When the output load increases, the DC current of the inductor current rises upwards, so that the inductor current cannot fall to a negative value, causing the inductor current to change direction, causing the switch to fail to achieve zero voltage switching. Therefore, the phase shift angle can be adjusted as the output load changes. In addition to solving the problem of zero voltage switching, the overall efficiency of the converter under various load conditions can also be improved.
2.2相移角度與電感值設計2.2 Phase shift angle and inductance value design
從2.1節可得知相移角度和電感值會影響轉換器的整體效率，因此需要推導相移角度與電感關係式，並進行電路分析及控制最佳化設計。根據前面之模式分析，在一個週期 T s 的每個區間的電感電流變化量如下所示。 It can be seen from Section 2.1 that the phase shift angle and the inductance value affect the overall efficiency of the converter. Therefore, it is necessary to derive the relationship between the phase shift angle and the inductance, and perform circuit analysis and control optimization design. According to the previous mode analysis, the amount of change in the inductor current in each interval of one cycle T s is as follows.
(35) (35)
平均功率P average之定義如(36)式所示，再將(35)式代入(36)式，經由電感電流積分後，可計算出輸入平均功率P in以及輸出平均功率P out，如(37)式所示。 The average power P average is defined as shown in equation (36), and the equation (35) is substituted into equation (36). After integrating the inductor current, the input average power P in and the output average power P out can be calculated, such as (37). ) as shown.
(36) (36)
(37) (37)
從(37)式可發現，輸入功率、輸出功率與開關時間t 1、t 2、t 3有關，故求出開關時間t 1、t 2、t 3的最大值，即可求出最大平均功率。假設在理想狀態下，轉換器無任何損耗，因此輸入、輸出功率會等於平均功率，如(38)式所示。 It can be found from equation (37) that the input power and output power are related to the switching times t 1 , t 2 , and t 3 , so the maximum values of the switching times t 1 , t 2 , and t 3 are obtained, and the maximum average power can be obtained. . Assume that under ideal conditions, the converter has no loss, so the input and output power will be equal to the average power, as shown in equation (38).
(38) (38)
由(35)式可得知在t=t 3時，電感電流為-I o，如(39)式所示。整理(39)式後，可得到t 1、t 2之關係式，如(40)式所示。 It can be seen from equation (35) that the inductor current is -I o at t = t 3 , as shown in equation (39). After finishing the formula (39), the relationship of t 1 and t 2 can be obtained, as shown by the formula (40).
(39) (39)
(40) (40)
將(40)式代入(37)式後，可得到P average(t 1)和P average(t 2)，令 、 ，即可得到t 1,max與t 2,max，如(41)式所示。 Substituting (40) into (37) gives P average (t 1 ) and P average (t 2 ), , , you can get t 1,max and t 2,max as shown in (41).
(41) (41)
將(41)式代入(37)式，即可得最大平均功率P average,max(42)式。 By substituting the formula (41) into the equation (37), the maximum average power P average,max (42) can be obtained.
(42) (42)
根據(42)式，可藉由所定義之最大平均功率反推導出電感的感值。另外，t 3跟f的關係式，如(43)式所示。 According to the formula (42), the inductance of the inductance can be derived by inversely deriving the maximum average power defined. In addition, the relational expression of t 3 and f is as shown in the formula (43).
(43) (43)
根據(42)式與(43)式，可得知隨著需求負載功率的改變，調整t 3的大小便可達到升降壓充電模式與升降壓放電模式之最佳化相移角度控制。 According to (42) and formula (43) formula, that may change as the load power demand, adjust the size of t 3 can achieve step-down phase of the charging mode and the best mode of elevating the discharge pressure control shift angle.
2.3零電壓切換條件2.3 zero voltage switching conditions
電感儲能必需大於兩個MOSFET的寄生電容儲能，才能在開關截止時達到開關零電壓切換，如(44)式所示。The energy storage of the inductor must be greater than the parasitic capacitance of the two MOSFETs to achieve switching zero voltage switching when the switch is turned off, as shown in equation (44).
(44) (44)
將(44)式整理後，可得零電壓切換條件式。若MOSFET的寄生電容為100 pF，則將V bu s 為380 V、電感為1.5 mH、MOSFET的寄生電容為100 pF代入(45)式，可計算出I o只要大於108 mA，在每個操作點都可以達到零電壓切換。 After sorting (44), a zero voltage switching condition can be obtained. If the parasitic capacitance of the MOSFET is 100 pF, then V bu s is 380 V, the inductance is 1.5 mH, and the parasitic capacitance of the MOSFET is 100 pF (45), and I o can be calculated as long as it is greater than 108 mA in each operation. Zero voltage switching can be achieved at all points.
(45) (45)
2.4相移角度設計2.4 phase shift angle design
雙向升降壓轉換器會操作於降壓型充電模式、升壓型充電模式、降壓型放電模式與升壓型放電模式共四種操作模式。根據1.2節可得知，當電路啟動時需給予一個初始相移角度f，電路才可以正常動作。The bidirectional buck-boost converter operates in four operating modes: buck-type charge mode, boost-type charge mode, buck-type discharge mode, and step-up discharge mode. According to Section 1.2, it is known that when the circuit is started, an initial phase shift angle f is required to allow the circuit to operate normally.
在降壓型充電模式部分，由(17)式電池端電壓與匯流排端電壓的關係式可得到(46)式。若在一例子中，將一電池端電壓與匯流排端電壓規格代入(46)式，得知功率開關S 1的責任週期最少要42 %才能達到本案所需降壓比，則將(46)式代入(47)式即可得初始相移角度需大於28.8度。 In the step-down charging mode section, the equation (46) can be obtained from the relationship between the (17) type battery terminal voltage and the bus terminal voltage. If, in an example, a battery terminal voltage and a bus terminal voltage specification are substituted into equation (46), it is known that the duty cycle of the power switch S 1 is at least 42% to achieve the required step-down ratio in this case, then (46) Substituting the formula (47) can achieve an initial phase shift angle of more than 28.8 degrees.
(46) (46)
(47) (47)
根據2.2節所推導之(42)式與(43)式，將所需之雙向升降壓直流-直流轉換器之設計規格代入，可得降壓型充電之最大平均功率與相移角度之關係。本發明之相移調整機制會隨著不同的負載功率需求調變相移角度，直到最大額定功率為止。According to the formulas (42) and (43) deduced in Section 2.2, the design specifications of the required bidirectional buck-boost DC-DC converter are substituted, and the relationship between the maximum average power and the phase shift angle of the step-down charging can be obtained. The phase shift adjustment mechanism of the present invention modulates the phase shift angle with different load power requirements up to the maximum rated power.
在升壓型充電模式部分，由(17)式電池端電壓與匯流排端電壓的關係式可得到(48)式。若在一例子中，將一電池端電壓與匯流排端電壓規格代入(48)式，可得知功率開關S 1的責任週期最少要55 %才能達到本發明所需升壓比，則將(48)式代入(49)式即可得初始相移角度需大於18度。 In the boost type charging mode portion, the equation (48) can be obtained from the relationship between the battery terminal voltage of the (17) type and the bus terminal voltage. If, in an example, a battery terminal voltage and a bus terminal voltage specification are substituted into equation (48), it can be known that the duty cycle of the power switch S 1 is at least 55 % to achieve the required boost ratio of the present invention. 48) Substituting into equation (49), the initial phase shift angle needs to be greater than 18 degrees.
(48) (48)
(49) (49)
根據2.2節所推導之(42)式與(43)式，將一雙向升降壓直流-直流轉換器之設計規格代入，可得升壓型充電之最大平均功率與相移角度之關係。本發明之相移調整機制會隨著不同的負載功率需求調變相移角度，直到最大額定功率為止。According to the formulas (42) and (43) deduced in Section 2.2, the design specifications of a bidirectional buck-boost DC-DC converter are substituted, and the relationship between the maximum average power of the boost type charging and the phase shift angle can be obtained. The phase shift adjustment mechanism of the present invention modulates the phase shift angle with different load power requirements up to the maximum rated power.
在降壓型放電模式部分，由(34)式匯流排端電壓與電池端電壓的關係式可得到(50)式。在一例子中，將一匯流排端電壓與電池端電壓規格代入(50)式，得知功率開關S 3的責任週期最少要45 %才能達到本案所需降壓比，則將(50)式代入(51)式即可得初始相移角度需大於18度。 In the step-down type of discharge mode, the equation (50) can be obtained from the relationship between the voltage of the bus terminal of the type (34) and the voltage of the battery terminal. In an example, a bus terminal voltage and a battery terminal voltage specification are substituted into the equation (50), and it is known that the duty cycle of the power switch S 3 is at least 45% to achieve the required step-down ratio in the present case, and then the equation (50) Substituting (51) can achieve an initial phase shift angle greater than 18 degrees.
(50) (50)
(51) (51)
根據2.2節所推導之(42)式與(43)式，將一雙向升降壓直流-直流轉換器之設計規格代入，可得降壓型放電之最大平均功率與相移角度之關係。本發明之相移調整法會隨著不同的負載功率需求調變相移角度，直到最大額定功率為止。According to the formulas (42) and (43) deduced in Section 2.2, the design specifications of a bidirectional buck-boost DC-DC converter are substituted, and the relationship between the maximum average power of the step-down discharge and the phase shift angle can be obtained. The phase shift adjustment method of the present invention modulates the phase shift angle with different load power requirements up to the maximum rated power.
在升壓型放電模式部分，由(34)式匯流排端電壓與電池端電壓的關係式可得到(52)式。在一例子中，將一匯流排端電壓與電池端電壓規格代入(52)式，得知功率開關S 3的責任週期最少要59 %才能達到本案所需升壓比，則將(52)式代入(53)式即可得初始相移角度需大於32.4度。 In the step-up type of discharge mode, the equation (52) can be obtained from the relationship between the voltage of the bus terminal (34) and the voltage of the battery terminal. In an example, a bus terminal voltage and a battery terminal voltage specification are substituted into the equation (52), and it is known that the duty cycle of the power switch S 3 is at least 59% to achieve the required boost ratio in the present case, and then (52) Substituting (53) can achieve an initial phase shift angle of more than 32.4 degrees.
(52) (52)
(53) (53)
根據2.2節所推導之(42)式與(43)式，將一雙向升降壓直流-直流轉換器之設計規格代入，可得降壓型放電之最大平均功率與相移角度之關係。本發明之相移調整法會隨著不同的負載功率需求調變相移角度，直到最大額定功率為止。According to the formulas (42) and (43) deduced in Section 2.2, the design specifications of a bidirectional buck-boost DC-DC converter are substituted, and the relationship between the maximum average power of the step-down discharge and the phase shift angle can be obtained. The phase shift adjustment method of the present invention modulates the phase shift angle with different load power requirements up to the maximum rated power.
III.韌體設計III. Firmware design
韌體程式設計流程Firmware programming process
圖14-18為本發明所採用控制程式其一實施例之相關流程圖。14-18 are related flowcharts of an embodiment of a control program employed in the present invention.
如圖14所示，主程式部分一開始會先設定資料記憶體初始值、堆疊指標設定、初始工作暫存器、暫存器初始值設定、內部振盪器初始值設定、輸出輸入埠設定、模組致能(PWM、ADC等)及中斷向量設定，接著進入無窮迴圈等待中斷向量旗標發生。當中斷向量旗標發生，會先讀取輸入電壓V bus、輸出電壓V bat與輸出電流值I bat，分別送入FIR濾波器進行濾波，接著判斷是否進入充電模式，為避免過於頻繁切換模式，造成電路損壞，因此加入了遲滯區間(hystersis)進行判斷，其中心點為380V、正膝點(positive trip point)為382V、負膝點(negative trip point)為378V，當匯流排電壓低於378V，代表負載需求大於能量供給，因此電池進行放電模式以提供能量給負載使用，接下來判斷電池端電壓大小，若大於380V則進入降壓型放電模式，反之則進入升壓型放電模式。若匯流排電壓高於382V，代表再生能源過剩，因此必須將多餘能量存入電池，進行充電模式，接下來判斷電池端電壓大小，若大於380V則進入升壓型充電模式，反之則進入降壓型充電模式。 As shown in Figure 14, the main program part first sets the data memory initial value, stacking index setting, initial working register, scratchpad initial value setting, internal oscillator initial value setting, output input setting, and mode. The group enables (PWM, ADC, etc.) and interrupt vector settings, and then enters the infinite loop to wait for the interrupt vector flag to occur. When the interrupt vector flag occurs, the input voltage V bus , the output voltage V bat and the output current value I bat are first read and sent to the FIR filter for filtering, and then it is determined whether to enter the charging mode, in order to avoid switching the mode too frequently, The circuit is damaged, so the hystersis is added for judgment. The center point is 380V, the positive trip point is 382V, and the negative trip point is 378V. When the bus voltage is lower than 378V. , represents that the load demand is greater than the energy supply, so the battery performs the discharge mode to provide energy for the load to use, and then determines the battery terminal voltage level. If it is greater than 380V, it enters the buck mode discharge mode, and vice versa, enters the boost mode discharge mode. If the busbar voltage is higher than 382V, it means that there is excess regenerative energy. Therefore, it is necessary to store excess energy in the battery for charging mode. Next, determine the voltage of the battery terminal. If it is greater than 380V, it will enter the boost charging mode. Otherwise, it will enter the buck. Type charging mode.
如圖15-18所示，進入升降壓充放電模式後，會先設定四個開關的責任週期，接著判斷輸出電流I o位於哪一設定範圍，給予相對應的相移角度，接著進入增量型PID運算，算出新的開關週期。 As shown in Figure 15-18, after entering the buck-boost charging and discharging mode, the duty cycle of the four switches is set first, then it is determined which setting range the output current I o is located, the corresponding phase shift angle is given, and then the increment is entered. Type PID calculation to calculate a new switching cycle.
IV. 實驗結果IV. Experimental results
圖19為本發明之自適應相移調整機制與固定相移100度之作法之一效率曲線比較圖。從圖19中可觀察出，本發明在任何負載情況下的效率皆大於等於固定相移100度之作法，因此可驗證本發明之技術可提升轉換電路之效率。FIG. 19 is a comparison diagram of an efficiency curve of an adaptive phase shift adjustment mechanism and a fixed phase shift of 100 degrees according to the present invention. It can be observed from Fig. 19 that the efficiency of the present invention under any load condition is greater than or equal to a fixed phase shift of 100 degrees, so that the technique of the present invention can be verified to improve the efficiency of the conversion circuit.
亦即，本發明之雙向升降壓直流-直流轉換器具有電路硬體架構簡單、被動元件數少、藉由功率開關的切換可改變能量傳輸方向、使轉換器均可操作於升/降壓型充放電模式、且其控制方式係數位控制方式等優點。此外，本發明亦提出一相移調整機制，可隨著不同負載功率需求調變相移角度，除有效降低導通損耗，亦可大幅提升輕載時的轉換效率，並採取開關柔切方式，減少切換損耗以提高系統效率。That is, the bidirectional buck-boost DC-DC converter of the present invention has a simple circuit hardware structure, a small number of passive components, and can change the energy transmission direction by switching the power switch, so that the converter can be operated on the step-up/step-down type. Charge and discharge mode, and its control mode coefficient bit control mode. In addition, the present invention also proposes a phase shift adjustment mechanism, which can adjust the phase shift angle with different load power requirements, in addition to effectively reducing the conduction loss, can also greatly improve the conversion efficiency at light load, and adopt a switch soft cut mode to reduce switching. Loss to increase system efficiency.
本案所揭示者，乃較佳實施例，舉凡局部之變更或修飾而源於本案之技術思想而為熟習該項技藝之人所易於推知者，俱不脫本案之專利權範疇。The disclosure of the present invention is a preferred embodiment. Any change or modification of the present invention originating from the technical idea of the present invention and being easily inferred by those skilled in the art will not deviate from the scope of patent rights of the present invention.
綜上所陳，本案無論就目的、手段與功效，在在顯示其迥異於習知之技術特徵，且其首先發明合於實用，亦在在符合發明之專利要件，懇請 貴審查委員明察，並祈早日賜予專利，俾嘉惠社會，實感德便。In summary, this case, regardless of its purpose, means and efficacy, is showing its technical characteristics that are different from the conventional ones, and its first invention is practical and practical, and it is also in compliance with the patent requirements of the invention. I will be granted a patent at an early date.
100‧‧‧全橋式轉換電路
110‧‧‧第一信號處理電路
120‧‧‧第二信號處理電路
200‧‧‧電池單元
130‧‧‧控制單元
101‧‧‧第一半橋開關電路
102‧‧‧第二半橋開關電路
103‧‧‧電感器
131‧‧‧類比至數位轉換單元
132‧‧‧濾波運算單元
133‧‧‧比例-積分-微分運算單元
134‧‧‧脈衝寬度調變運算單元
135‧‧‧驅動單元100‧‧‧Full bridge conversion circuit
110‧‧‧First signal processing circuit
120‧‧‧second signal processing circuit
200‧‧‧ battery unit
130‧‧‧Control unit
101‧‧‧First half bridge switching circuit
102‧‧‧Second half bridge switching circuit
103‧‧‧Inductors
131‧‧‧ analog to digital conversion unit
132‧‧‧Filtering unit
133‧‧‧Proportional-Integral-Derivative Unit
134‧‧‧ pulse width modulation unit
135‧‧‧ drive unit
圖1為一微電網系統架構圖。 圖2繪示一電動車的電力介面。 圖3繪示以功率開關取代二極體的一般雙向直流-直流轉換器。 圖4(a)-4(c)分別繪示一傳統雙向轉換器、一C’uk轉換器與一SEPIC轉換器。 圖5繪示一非隔離型雙向直流-直流轉換器。 圖6繪示一隔離型雙向直流-直流轉換器。 圖7繪示本發明之具自適應相移角控制機制之雙向直流-直流轉換器之一實施例。 圖8繪示圖7所示之全橋式轉換電路之一示意圖。 圖9繪示圖7所示之全橋式轉換電路之一等效模型。 圖10(a)-10(d)繪示本發明之充電模式狀態(一)至充電模式狀態(四)。 圖11(a)-11(d)繪示本發明之放電模式狀態(一)至放電模式狀態(四)。 圖12(a)-12(l)繪示本發明在充電模式之十二個時序狀態下的等效電路。 圖13(a)-13(l)繪示本發明在放電模式之十二個時序狀態下的等效電路。 圖14-18為本發明所採控制程式其一實施例之相關流程圖。 圖19為本發明之自適應相移調整機制與固定相移100度之作法之一效率曲線比較圖。Figure 1 is a diagram of a microgrid system architecture. 2 illustrates a power interface of an electric vehicle. FIG. 3 illustrates a general bidirectional DC-DC converter in which a diode is replaced by a power switch. 4(a)-4(c) respectively show a conventional bidirectional converter, a C'uk converter and a SEPIC converter. FIG. 5 illustrates a non-isolated bidirectional DC-DC converter. FIG. 6 illustrates an isolated bidirectional DC-DC converter. FIG. 7 illustrates an embodiment of a bidirectional DC-DC converter with adaptive phase shift angle control mechanism of the present invention. FIG. 8 is a schematic diagram of a full bridge conversion circuit shown in FIG. 7. FIG. 9 is a diagram showing an equivalent model of the full bridge conversion circuit shown in FIG. 7. 10(a)-10(d) illustrate the state of charge mode (1) to the state of charge mode (4) of the present invention. 11(a)-11(d) illustrate the discharge mode state (1) to the discharge mode state (4) of the present invention. 12(a)-12(l) illustrate equivalent circuits of the present invention in twelve timing states of the charging mode. Figures 13(a)-13(l) illustrate equivalent circuits of the present invention in twelve timing states of the discharge mode. 14-18 are flowcharts related to an embodiment of the control program of the present invention. FIG. 19 is a comparison diagram of an efficiency curve of an adaptive phase shift adjustment mechanism and a fixed phase shift of 100 degrees according to the present invention.
100‧‧‧全橋式轉換電路 100‧‧‧Full bridge conversion circuit
110‧‧‧第一信號處理電路 110‧‧‧First signal processing circuit
120‧‧‧第二信號處理電路 120‧‧‧second signal processing circuit
200‧‧‧電池單元 200‧‧‧ battery unit
130‧‧‧控制單元 130‧‧‧Control unit
101‧‧‧第一半橋開關電路 101‧‧‧First half bridge switching circuit
102‧‧‧第二半橋開關電路 102‧‧‧Second half bridge switching circuit
103‧‧‧電感器 103‧‧‧Inductors
131‧‧‧類比至數位轉換單元 131‧‧‧ analog to digital conversion unit
132‧‧‧濾波運算單元 132‧‧‧Filtering unit
133‧‧‧比例-積分-微分運算單元 133‧‧‧Proportional-Integral-Derivative Unit
134‧‧‧脈衝寬度調變運算單元 134‧‧‧ pulse width modulation unit
135‧‧‧驅動單元 135‧‧‧ drive unit
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| Publication number | Priority date | Publication date | Assignee | Title |
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| CN112825455A (en) * | 2019-11-01 | 2021-05-21 | 上海瞻芯电子科技有限公司 | Control method for power cycle circuit |
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| TW201720031A (en) | 2017-06-01 |
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