TWI388138B - Joint circular array antenna and mfs-mc-dsss technologies to estimate the doa of a very low signal-to-noise ratio air target for bistatic radar - Google Patents
Joint circular array antenna and mfs-mc-dsss technologies to estimate the doa of a very low signal-to-noise ratio air target for bistatic radar Download PDFInfo
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本發明是一種關於雙態相列雷達之目標來向估測接收機架構與裝置，且特別是一種結合環形陣列天線與多載波直接展頻序列倍頻率展頻技術之低訊雜比目標來向估測接收機架構方法與裝置。The invention relates to a target-oriented estimation receiver architecture and device for a two-state phased radar, and in particular to a low-to-noise ratio target combined with a circular array antenna and a multi-carrier direct spread spectrum sequence frequency spreading technique. Receiver architecture method and apparatus.
雙態雷達系統是指雷達發射端與雷達接收端不在同一地理位置，所以雷達接收端無法直接得知發射端傳送的波形。在雷達接收端會同時接收直接來自發射端的直接路徑(line-of-sight,LOS)參考信號與飛行目標之反射回波信號，其中直接路徑參考信號的功率會遠大於目標回波信號的功率，傳統上會使用兩組高方向性(directivity)天線分別接收不同到達方向(direction of arrival，DOA)的直接路徑參考信號與目標回波信號，將目標回波信號與參考信號分開。而本發明則是採用一組環形陣列天線，可以16組等寬波束以涵蓋360°之水平域，藉由多模式數位波束成型處理將目標回波信號與參考信號分開，並消除特定方向的干擾信號。因此相較於傳統的兩組高方向性天線做法，環形陣列天線的多模式數位波束成型處理可以同時接收15個方向的反射回波信號並提高雙態雷達系統的抗干擾性能。The two-state radar system means that the radar transmitting end is not in the same geographical position as the radar receiving end, so the radar receiving end cannot directly know the waveform transmitted by the transmitting end. At the receiving end of the radar, the line-of-sight (LOS) reference signal directly from the transmitting end and the reflected echo signal of the flying target are simultaneously received, wherein the power of the direct path reference signal is much larger than the power of the target echo signal. Traditionally, two sets of high directivity antennas are used to receive direct path reference signals and target echo signals of different directions of arrival (DOA), respectively, to separate the target echo signals from the reference signals. The present invention uses a set of circular array antennas, which can cover 16 sets of equal-width beams to cover the horizontal range of 360°, separate the target echo signals from the reference signals by multi-mode digital beamforming, and eliminate interference in specific directions. signal. Therefore, compared with the traditional two sets of high directional antennas, the multi-mode digital beamforming process of the ring array antenna can simultaneously receive the reflected echo signals in 15 directions and improve the anti-interference performance of the two-state radar system.
傳統雙態雷達系統的同調定位(coherent location)做法是利用一組目標回波信號與參考信號所到達時間差(time difference of arrival,TDOA)估測目標位置。但是低訊雜比的目標回波信號TDOA的誤差量會遠大於高訊雜比直接路徑參考信號TDOA的誤差量，因此在實施上大多是利用多個不同地理位置所接收的目標回波信號TDOA，或是利用接收來自多個不同地理位置發射端直接路徑參考信號TDOA減少估測目標位置的誤差量。傳統的TDOA做法並沒有利用高訊雜比直接路徑參考信號來改善低訊雜比的目標回波信號TDOA的誤差量，因此本發明揭示藉由高訊雜比的直接路徑參考信號與低訊雜比的目標回波信號的交相關處理估測目標來向。本發明並揭示應用倍頻率展頻(multiple frequency spreader,MFS)於多載波直接展頻序列(MC-DSSS)調變波形，進一步提高交相關處理的脈波壓縮增益，降低目標偵測誤判率。The coherent location of the traditional two-state radar system is to estimate the target position by using a set of target echo signals and a reference signal time difference of arrival (TDOA). However, the error amount of the target echo signal TDOA of the low signal-to-noise ratio is much larger than the error amount of the high-signal-to-noise direct path reference signal TDOA. Therefore, in practice, the target echo signal TDOA received by a plurality of different geographical locations is mostly used. Alternatively, the amount of error of the estimated target position is reduced by receiving the direct path reference signal TDOA from a plurality of different geographical locations. The traditional TDOA method does not utilize the high signal ratio direct path reference signal to improve the error amount of the low echo ratio target echo signal TDOA. Therefore, the present invention discloses a direct path reference signal and a low signal miscellaneous ratio by high signal ratio. The intersection of the target echo signals is compared to the estimated target arrival. The invention also discloses applying a multiple frequency spreader (MFS) to a multi-carrier direct spread spectrum sequence (MC-DSSS) modulation waveform, further improving the pulse compression gain of the cross correlation processing, and reducing the false detection rate of the target detection.
鑒於以上所述先前技術之缺點，本發明之主要目的便是在於提供一種雙態相列雷達低訊雜比目標來向估測架構方法及裝置，其利用高訊雜比的直接路徑參考信號與低訊雜比的目標回波信號的交相關處理估測目標反射回波信號來向。In view of the above-mentioned shortcomings of the prior art, the main object of the present invention is to provide a two-state phased radar low signal-to-noise ratio target-oriented estimation architecture method and apparatus, which utilizes a high signal-to-noise ratio direct path reference signal and low The inter-correlation processing of the target echo signal of the signal-to-noise ratio estimates the direction of the target reflected echo signal.
本發明之另一目的在於提供一種雙態相列雷達低訊雜比目標來向估測接收機之架構方法及裝置，其採用一組環形陣列天線，利用等寬波束以涵蓋360°之水平域，藉由多模式數位波束成型處理將目標回波信號與參考信號分開，並消除特定方向的干擾信號，提高雙態雷達系統的抗干擾性能與訊雜比。Another object of the present invention is to provide a two-state phased radar low signal-to-noise ratio target-oriented estimation receiver architecture method and apparatus, which uses a set of ring array antennas, and uses a uniform beam to cover a horizontal range of 360°. The multi-mode digital beamforming process separates the target echo signal from the reference signal and eliminates the interference signal in a specific direction, thereby improving the anti-interference performance and the signal-to-noise ratio of the two-state radar system.
本發明之另一目的在於提供一種雙態相列雷達低訊雜比目標來向估測架構方法及裝置，應用倍頻率展頻於多載波直接展頻序列(MFS-MC-DSSS)調變波形，進一步提高交相關處理的脈波壓縮增益，降低目標偵測誤判率。Another object of the present invention is to provide a method and apparatus for estimating a low-to-noise ratio target of a two-state phased radar, and applying a frequency multiplication to a multi-carrier direct spread spectrum sequence (MFS-MC-DSSS) modulation waveform. Further improve the pulse compression gain of the correlation processing and reduce the false detection rate of the target detection.
本發明之另一目的在於提供一種雙態相列雷達低訊雜比目標來向估測架構方法及裝置，至少包含：(1).發射出一多載波直接展頻序列倍頻率展頻調變波形，提高交相關處理的脈波壓縮增益，降低目標偵測誤判率；(2).一組環形陣列天線；(3).利用環形陣列天線的多模式數位波束成型處理將目標回波信號與參考信號分開，並消除特定方向的干擾信號，提高雙態雷達系統的抗干擾性能；(4).利用直接路徑參考信號與目標回波信號的交相關處理估測反射回波信號目標來向。Another object of the present invention is to provide a method and apparatus for estimating a low-to-noise ratio target of a two-state phased radar, comprising at least: (1) transmitting a multi-carrier direct spread spectrum sequence frequency spread spectrum modulation waveform , improve the pulse compression gain of the cross-correlation processing, reduce the false detection rate of the target detection; (2) a set of ring array antennas; (3) multi-mode digital beamforming processing using the circular array antenna to the target echo signals and reference The signals are separated, and the interference signals in a specific direction are eliminated, and the anti-interference performance of the two-state radar system is improved; (4) The target direction of the reflected echo signals is estimated by using the correlation processing between the direct path reference signal and the target echo signal.
為進一步對本發明有更深入的說明，乃藉由以下圖示、圖號說明及發明詳細說明，冀能對 貴審查委員於審查工作有所助益。In order to further explain the present invention, it will be helpful to review the review by the following illustrations, illustrations, and detailed descriptions of the invention.
茲配合下列之圖式說明本發明之詳細結構，及其連結關係，以利於 貴審委做一瞭解。The detailed structure of the present invention and its connection relationship will be described in conjunction with the following drawings to facilitate an understanding of the audit committee.
以下即配合所附之圖式，詳細揭露說明本發明之結合環形陣列天線與多載波直接展頻序列倍頻率展頻技術之低訊雜比目標來向估測接收機架構方法與裝置之實施例。本發明並推導雙態相列雷達低訊雜比目標來向估測接收機直接路徑接收信號與反射回波信號之交相關輸出公式，以模擬方式驗證本發明之估測架構與裝置性能。實施方式分為五大項分述之：In the following, with reference to the accompanying drawings, an embodiment of the method and apparatus for estimating the receiver architecture of the low-to-noise ratio target combining the circular array antenna and the multi-carrier direct spread spectrum multiple frequency spread spectrum technique of the present invention will be described in detail. The invention also deduces the low-signal-to-noise ratio target of the two-state phased radar to estimate the correlation output expression of the direct path received signal and the reflected echo signal of the receiver, and simulates the estimated architecture and device performance of the present invention in an analog manner. The implementation method is divided into five major items:
環形陣列天線低訊雜比目標來向估測接收機架構如圖1、10及圖14所示，本發明之結合環形陣列天線與多載波直接展頻序列倍頻率展頻技術之低訊雜比目標來向估測接收裝置，其係包括有：一多載波直接展頻序列發射機1，該多載波直接展頻序列發射機更係包括有：一多載波直接展頻序列倍頻率展頻波形產生器11，脈波產生器輸出第m 脈波經過直接序列展頻器輸出直接序列展頻信號，該多載波直接展頻序列倍頻率展頻波形產生器11更係包括有一脈波產生器111、一直接序列展頻器112、一四相相移鍵控信號調變器113、一倍頻率展頻器114、一64-IFFT(快速傅立葉轉換)115及一數位/類比轉換器116；以及一升頻器12，用以放大該多載波直接展頻序列倍頻率展頻波形產生器的輸出基頻信號並將基頻信號升頻至載波頻率，轉變為射頻信號；一環形天線接收機2，該環形天線接收機更係包括有：一環形陣列天線21，具有複數個天線元，該複數個天線元可以至少二個以上為一組子陣列天線，若干個子陣列天線可用以接收360度方向的信號；一線性補償前置處理器22，接受環形陣列天線21信號，並使天線元接收信號經過延遲線進行線性補償前置處理，使其等效為一非等間距之線性陣列天線；一降頻轉換器23，接收該線性補償前置處理器22信號轉換為一基頻信號，該降頻轉換器23更係包括有一柴比雪夫窗戶處理器231、一降頻器232及類比/數位轉換器233；一零化處理器24，去除特定方向的干擾信號，並產生基頻反射回波信號與基頻參考信號，該零化處理器24係由一多模式數位波束成型器241所構成，該多模式數位波束成型器更係包括：一多波束成型模組2411、一振幅比較目標來向估測模組2412及一零化指向波束成型模組2413；一移動目標指示處理器25，用以去除零化處理器24所產生之低訊雜比的基頻反射回波信號，該基頻反射回波信號係指散射體所產生的反射雜波；一倍頻率解展頻電路26，用以獲利若干倍數之頻率展頻增益，該倍頻率解展頻電路更係包括有一倍頻率解展頻器262、一64-FFT 261及一四相相移鍵控信號調解器263；一交相關處理器27，估測低訊雜比飛行目標反射回波信號之目標來向，該交相關處理器27更係包括有一64-FFT271、一振幅相關處理器272、一64點循環位移器273、一倍頻率解展器274、一四相相移鍵控信號調解器275、一直接序列解展頻器276及一數位頻率合成器277；一目標偵測器28，用以判斷該環形天線接收機2若干個子陣列天線何者為輸出最大功率，以初步判定飛行目標的方向，該目標偵測器28係由一目標偵測之最大值判斷器281所構成；以及一目標來向估測器29，根據一多模式數位波束成型處理裝置之目標來向估測模式細部判定飛行目標的方向。Circular array antenna low signal-to-noise ratio target-to-estimation receiver architecture, as shown in Figures 1, 10 and 14, the low-to-noise ratio target of the combined ring array antenna and multi-carrier direct spread spectrum multiple frequency spread spectrum technique of the present invention The estimated receiving device comprises: a multi-carrier direct spread spectrum sequence transmitter 1 , the multi-carrier direct spread spectrum sequence transmitter further comprises: a multi-carrier direct spread spectrum sequence frequency spread spectrum waveform generator 11. The pulse wave generator outputs the mth pulse wave to output a direct sequence spread spectrum signal through the direct sequence spreader, and the multicarrier direct spread frequency sequence frequency spread spectrum waveform generator 11 further includes a pulse wave generator 111 and a a direct sequence spreader 112, a quadrature phase shift keying signal modulator 113, a double frequency spreader 114, a 64-IFFT (fast Fourier transform) 115, and a digital/analog converter 116; and one liter The frequency converter 12 is configured to amplify an output baseband signal of the multi-carrier direct spread spectrum multiple frequency spread spectrum waveform generator and up-convert the base frequency signal to a carrier frequency to be converted into a radio frequency signal; a loop antenna receiver 2, the Loop antenna connection The machine further includes: a ring array antenna 21 having a plurality of antenna elements, wherein the plurality of antenna elements can be at least two or more of a set of sub-array antennas, and the plurality of sub-array antennas can be used to receive signals in a 360-degree direction; The compensation pre-processor 22 receives the signal of the ring array antenna 21, and causes the antenna element receiving signal to undergo linear compensation pre-processing through the delay line to make it equivalent to a non-equal-spaced linear array antenna; a down converter 23 The signal received by the linear compensation pre-processor 22 is converted into a baseband signal, and the down converter 23 further includes a Chebyshev window processor 231, a downconverter 232, and an analog/digital converter 233; The annihilation processor 24 removes the interference signal in a specific direction and generates a fundamental frequency reflected echo signal and a fundamental frequency reference signal. The annihilation processor 24 is formed by a multi-mode digital beamformer 241, the multi-mode digital position. The beamformer further includes: a multi-beam forming module 2411, an amplitude comparison target-to-estimation module 2412, and a zero-pointing beamforming module 2413; a moving target indication The processor 25 is configured to remove the fundamental frequency reflected echo signal generated by the zeroing processor 24, and the fundamental frequency reflected echo signal refers to the reflected clutter generated by the scatterer; The frequency circuit 26 is configured to obtain a plurality of frequency spread gains, and the double frequency despreading circuit further comprises a frequency sweeper 262, a 64-FFT 261 and a four-phase phase shift keying signal to mediate The intersection processor 263 is configured to estimate the target direction of the low-to-noise ratio target echo echo signal. The cross-correlation processor 27 further includes a 64-FFT 271, an amplitude correlation processor 272, and a 64 point. a cyclic shifter 273, a frequency despreader 274, a four-phase phase shift keying signal conditioner 275, a direct sequence despreading 276 and a digital frequency synthesizer 277; a target detector 28 for Determining which of the plurality of sub-array antennas of the loop antenna receiver 2 is the output maximum power to initially determine the direction of the flight target, the target detector 28 is composed of a target detection maximum value determiner 281; and a target direction Estimator 29, based on a multi-mode digital Beam forming process of the target device to determine the direction of flight details to estimate the target pattern.
藉由上述結構，環形陣列天線多模式數位波束成型處理，產生第j 波束(Beamj )接收直接路徑(line-of-sight,LOS)的接收信號，做為雙態雷達的參考信號s t (t )，產生第k 波束(Beamk )接收飛行目標之反射回波信號s r (t )與參考信號s t (t )進行交相關(cross correlation)處理，估測低訊雜比飛行載具反射回波信號之目標來向。環形陣列天線的實際天線元排列並不在同一空間的直線上，天線元接收信號必須先經過延遲線進行線性補償前置處理，使其等效為一非等間距之線性陣列天線。Beamk 與Beamj 輸出的反射目標回波信號s r (t )與直接路徑參考信號s t (t )經過降頻轉換為基頻信號分別進行零化處理，去除特定方向的干擾信號，產生基頻目標反射回波信號u r (t )與基頻直接路徑參考信號u t (t )。其中低訊雜比的u r (t )可以藉由移動目標指示(moving target indication,MTI)處理去除靜止散射體(scatter)所產生的反射雜波(clutter)，輸出u' r (t )。最後u' r (t )與u t (t )進行交相關處理與目標來向估測。With the above structure, the circular array antenna multi-mode digital beamforming process generates a j- beam (Beam j ) receiving line-of-sight (LOS) received signal as a reference signal of the two-state radar s t ( t ), generating a kth beam (Beam k ) receiving a flying target reflected echo signal s r ( t ) and a reference signal s t ( t ) for cross correlation processing, estimating a low signal ratio flying carrier The target of the reflected echo signal is directed. The actual antenna element arrangement of the ring array antenna is not in a straight line in the same space, and the antenna element receiving signal must first undergo linear compensation pre-processing through the delay line to make it equivalent to a non-equidistant linear array antenna. The reflected target echo signal s r ( t ) output by Beam k and Beam j and the direct path reference signal s t ( t ) are subjected to down-conversion to the fundamental frequency signal to be respectively zeroed, and the interference signal in a specific direction is removed to generate a basis. The frequency target reflects the echo signal u r ( t ) and the fundamental frequency direct path reference signal u t ( t ). The u r ( t ) of the low signal-to-noise ratio can remove the reflected clutter generated by the stationary scatter by the moving target indication (MTI) process, and output u' r ( t ). Finally, u' r ( t ) and u t ( t ) are cross-correlated and target-oriented.
低訊雜比目標來向估測接收機採用環形等波束寬陣列天線的架構如圖2所示，以16個天線元涵蓋360°水平方位。為了滿足等波束寬的需求，選定4個天線元的陣列為一子陣列天線，如圖3所示。實際天線元排列並不在同一空間的直線上，中間的兩天線元間距為d ，則兩旁的天線元間距縮短為d cos(π/8)，而天線元與天線陣列軸線的距離為d sin(π/8)。為了天線場型分析上的需求，進行子陣列天線之相位補償The architecture of the low-to-noise ratio target to the estimated receiver using a circular beam-width array antenna is shown in Figure 2, with 16 antenna elements covering the 360° horizontal azimuth. In order to meet the requirements of equal beamwidth, the array of four antenna elements is selected as a sub-array antenna, as shown in FIG. The actual antenna element arrangement is not on the straight line of the same space. The distance between the two antenna elements in the middle is d , and the distance between the antenna elements on both sides is shortened to d cos(π/8), and the distance between the antenna element and the axis of the antenna array is d sin ( π/8). Phase compensation for subarray antennas for antenna field analysis requirements
使其等效為一非等間距之線性陣列天線，如圖4所示。補償後的等效子天線陣列之天線元間距將變為Make it equivalent to a non-equal spacing linear array antenna, as shown in Figure 4. The antenna element spacing of the compensated equivalent sub-antenna array will become
天線元權值I m 分別為[0.48,1,1,0.48]。16個天線元共產生16個波束寬23°之等寬波束以涵蓋360°之水平域，並且再假設每個天線元之天線增益等於1，則相對應的每一等寬波束之場型如圖5所示，本文將以此線性陣列模型近似原非線性陣列，以進行後續之訊號處理。實現架構如圖6所示，x m ~x m +3 為等寬多波束陣列天線中相鄰的四天線元所接收訊號。The antenna element weights I m are [0.48, 1, 1, 0.48], respectively. 16 antenna elements collectively generate 16 equal beam beams with a beam width of 23° to cover the horizontal domain of 360°, and further assume that the antenna gain of each antenna element is equal to 1, then the corresponding field type of each equal-width beam is As shown in Figure 5, this paper will approximate the original nonlinear array with this linear array model for subsequent signal processing. The implementation architecture is shown in Figure 6. x m ~ x m +3 is the received signal of the adjacent four antenna elements in the equal-width multi-beam array antenna.
本發明利用16個波束加上振幅比較法尋向法則，進行目標方向估測，先以兩波束的狀況來說明振幅比較的尋向原理，因為不同的兩相鄰波束，只是負責不同的方位範圍估測，都是採用一樣的尋向原理，因此以兩波束來說明已經可以代表其一般性。等寬多波束陣列天線中的相鄰兩波束之天線場型可以分別表示為
差場型循向原理是選擇兩個不同指向的波束成型對接收的目標來向信號輸出功率之比值，因為在dB軸上為兩個波束成型的場型之差值，因此稱之為差場型。因為差場型的輸出功率與接收信號功率的大小無關，所以通道衰減造成接收信號功率的變化不會影響對差場型循向。而對應兩波束峰值之間的方位範圍，即可視為負責的尋向範圍，也就是所謂的瞬時視野(field of view)，在波束較寬時，瞬 時視野較廣。瞬時視野對尋向準確度的設計有相當的影響，一般而言，瞬時視野愈寬，尋向的準確度愈差。兩波束振幅比較的尋向系統方塊如圖9所示。The difference field type tracking principle is to select the ratio of the beam-forming of two different directions to the output power of the received target to the signal, because the difference between the two beamformed field types on the dB axis is called the difference field type. . Since the output power of the differential field type is independent of the magnitude of the received signal power, the channel attenuation causes a change in the received signal power without affecting the differential field type. Corresponding to the range of azimuth between the peaks of the two beams, it can be regarded as the responsible homing range, which is called the field of view. When the beam is wide, the instantaneous The time horizon is wider. Instantaneous field of view has a considerable impact on the design of homing accuracy. In general, the wider the instantaneous field of view, the worse the accuracy of homing. The homing system block of the two beam amplitude comparison is shown in FIG.
多模式數位波束成型處理器硬體架構如圖10所示，依功能區分為三個模組，多波束成型模組2411、振幅比較目標來向估測模組2412與零化指向波束成型模組2413。多波束天線之零化指向波束成型之目的在於要消除特定方向之高能量干擾訊號，其作法是利用多波束天線之其他天線組作為輔助天線組，並將輔助天線組的主波束指向干擾訊號源方向，再將輔助天線組所收到之訊號乘上一權值和主要天線組所收到的訊號相加，如圖9所示。為了共用相同的電路模組，多模式數位波束成型器的特定方向零化指向功能，直接使用多波束成型產生的16個波束之中的兩個相鄰指向波束(分別以BF M
與BF S
表示)的線性組合產生特定方向零化之指向波束的線性組合BF null
=BF M
+w
(θ n
)BF S
(5)產生特定零化方向θ n
的指向波束場型，其中零化權值
環形陣列天線多載波直接序列倍頻率展頻(Multiple frequency speading-multi carrier direct sequence spread spectrum,MFS-MC-DSSS)雙態相列雷達系統方塊圖，如圖14所示。多載波直接展頻序列倍頻率展頻雷達發射機的脈波產生器輸出第m 脈波經過直接序列展頻器輸出直接序列展頻信號C m ={c 0 c 1 Kc 127 }。直接序列展頻信號C m 經過四相相移鍵控(QPSK)調變產生多載波直接展頻序列倍頻率展頻波形的64個子載波複數信號B m ={b 0 b 1 Kb 63 }，輸入64點反快速傅立葉轉換(Inverse Fast Fourier Transform,IFFT)，64-IFFT產生MFS-MC-DSSS信號u mc (n )。u mc (n )經過數位轉類比器(D/A)輸出多載波直接展頻序列倍頻率展頻基頻調變信號u mc (t )再經過混波器升頻f c ，則多載波直接展頻序列倍頻率展頻雷達發射微波信號可表示為A block diagram of a dual-frequency speading-multi-carrier direct sequence spread spectrum (MFS-MC-DSSS) two-state phase-series radar system, as shown in FIG. The multi-carrier direct spread spectrum multiple frequency spread spectrum radar transmitter pulse generator output m- th pulse through the direct sequence spreader output direct sequence spread spectrum signal C m = { c 0 c 1 K c 127 }. The direct sequence spread spectrum signal C m is subjected to quadrature phase shift keying (QPSK) modulation to generate 64 subcarrier complex signals B m ={ b 0 b 1 K b 63 } of the multicarrier direct spread sequence multiple frequency spread spectrum waveform, The 64-point Inverse Fast Fourier Transform (IFFT) is input, and the 64-IFFT generates the MFS-MC-DSSS signal u mc ( n ). u mc ( n ) is outputted by a digital transcoder (D/A) multi-carrier direct spread spectrum sequence frequency spread spectrum fundamental frequency modulation signal u mc ( t ) and then up-converted by the mixer f c , then multi-carrier direct Spread spectrum sequence frequency spread spectrum radar transmitting microwave signal can be expressed as
其中f c 為載波頻率，T sym 為基頻信號的時間寬度。Where f c is the carrier frequency and T sym is the time width of the baseband signal.
雙態相列雷達系統之多路徑環境如圖1所示，若直接路徑入射Beamj 的角度為θ t ，Beamj 的指向角度為θ j ，則接收的參考信號可以表示為The multipath environment of the two-state phased radar system is shown in Figure 1. If the angle of the direct path incident Beam j is θ t and the pointing angle of Beam j is θ j , the received reference signal can be expressed as
其中P t 為參考信號功率，R t 為直接路徑距離，c 為光速，I (t )為干擾信號，n t (t )為接收機雜訊。假設發射信號s (t )為全向(isotropic)且非分散(non-dispersive)之窄頻(narrow band)信號。在遠場(far field)條件下入射陣列元(array element)，此時信號波前為平面波。環狀天線之子陣列天線陣列是由四個天線元組成，其陣列manifold向量a (θ)可以表示為Where P t is the reference signal power, R t is the direct path distance, c is the speed of light, I ( t ) is the interference signal, and n t ( t ) is the receiver noise. It is assumed that the transmitted signal s ( t ) is an isotropic and non-dispersive narrow band signal. An array element is incident under far field conditions, in which case the signal wavefront is a plane wave. The sub-array antenna array of the loop antenna is composed of four antenna elements, and the array manifold vector a (θ) thereof can be expressed as
其中波長λ =c/f c 。而靜止散射體產生的非直接路徑(Non-line-of-sight,NLOS)接收回波信號可以表示為Wherein the wavelength λ = c/f c . The non-direct-of-sight (NLOS) received echo signal generated by the stationary scatterer can be expressed as
其中g i 為路徑衰減。假設只有單一目標，若以發射信號時間為參考點(i.e.t =0)，目標與發射端之距離為R 0 ，目標與接收端的環狀陣列天線之距離為R r 、相對速度為v 。若目標反射信號入射波束(Beam)k 的角度為θ r ，波束k 的指向角度為θ k ，則目標反射回波信號Where g i is the path attenuation. Assume that there is only a single target. If the time of the transmitted signal is used as the reference point (ie t =0), the distance between the target and the transmitting end is R 0 , and the distance between the target and the ring-shaped array antenna at the receiving end is R r and the relative speed is v . If the angle of the target reflected signal incident beam (Beam) k is θ r and the pointing angle of the beam k is θ k , the target reflected echo signal
其中為直接路徑同調(coherent)干擾，其中D (t )為一時變的時間延遲。令接收端相對目標距離R r ，相對目標速度v ，當目標相對於發射端的距離R 0 遠大於vT prt ，T prt 為雷達脈波重覆週期，則among them For direct path coherent interference, where D ( t ) is a time-varying time delay. Let the receiving end relative to the target distance R r , relative to the target speed v , when the distance R 0 of the target relative to the transmitting end is much larger than vT prt , and T prt is the radar pulse repetition period, then
而I DP (t )為從傳送端以θ n 入射波束k 的直接路徑同調(coherent)干擾，P r 為接收回波信號功率。因為直接路徑同調(coherent)干擾功率會遠大於非直接路徑接收回波信號(P r <<P t )，即使同調干擾的入射角度θ n 落在a H (θ k )產生的指向波束k 的-10dB旁波瓣，如圖5所示，仍不足以消除同調干擾。w (θ n )為多模式數位波束成型的零化指向波束權值，可以產生超過-80dB的零點壓抑，零化θ n 入射的強同調干擾信號並維持指向θ k 。MTI濾波器則可以消除靜止散射體產生的干擾信號I (t )。當θ t =θ j ,θ r =θ k ，則 I DP ( t ) is the direct path coherent interference from the transmitting end with the θ n incident beam k , and P r is the received echo signal power. Since the direct path coherent interference power is much larger than the indirect path reception echo signal ( P r << P t ), even if the coherent interference incidence angle θ n falls on the pointing beam k generated by a H (θ k ) The -10dB sidelobe, as shown in Figure 5, is still not sufficient to eliminate coherent interference. w (θ n ) is the zero-directed beam weight of multi-mode digital beamforming, which can generate zero-point suppression of more than -80dB, zeroing the strong coherent interference signal of θ n incident and maintaining the pointing θ k . The MTI filter eliminates the interference signal I ( t ) generated by the stationary scatterer. When θ t = θ j , θ r = θ k , then
其中a r 為天線增益，若以接收的直接路徑S參考信號s t (t )的接收時間為參考點，並忽略雜訊，則目標反射的回波信號s r (t )可以表示為Where a r is the antenna gain. If the reception time of the received direct path S reference signal s t ( t ) is taken as the reference point and the noise is ignored, the target reflected echo signal s r ( t ) can be expressed as
其中s r (t )相對於s t (t )的相位延遲為Wherein the phase delay of s r ( t ) relative to s t ( t ) is
角頻率函數為The angular frequency function is
由(14)可知雙態雷達參考信號s t (t )與回波信號s r (t )經過混波器降到基頻，再經過類比/數位轉換器輸出基頻回波信號u t (n )與u r (n )。都卜勒頻率估測器是根據MFS-MC-DSSS基頻回波信號u r (n )的64個子載波頻率相對於MFS-MC-DSSS基頻參考信號u t (n )的位移量計算出都卜勒頻率估測值f' D 。以都卜勒頻率f' D 補償回波信號u r (n )獲得都卜勒補償信號u d (n )。u d (n )經過64-FFT(快速傅立葉轉換)解調為64個子載波複數信號。使用最大似然法則(Maximum Likelihood,ML)解碼獲得展頻信號。將解回的展頻信號與參考的展頻信號進行解展頻。特別要注意的是參考信號與回波信號的雜訊n t (n )與n r (n )為統計獨立，所以雜訊n t (n )與n r (n )導致的ML解碼錯誤為不相關，解展頻的自相關處理輸出可以消除雜訊E t 與E r 。換言之，即使在低訊雜比時，當完成正確都卜勒頻率估測與補償，解展頻信號經由脈波壓縮處理仍會產生最大輸出功率，並經由多模式波束成型處理估測目標方向。It can be seen from (14) that the two-state radar reference signal s t ( t ) and the echo signal s r ( t ) are down-converted to the fundamental frequency by the mixer, and then output the fundamental frequency echo signal u t ( n ) through the analog/digital converter. ) with u r ( n ). The Doppler frequency estimator is calculated based on the displacement of the 64 subcarrier frequencies of the MFS-MC-DSSS fundamental frequency echo signal u r ( n ) relative to the MFS-MC-DSSS fundamental reference signal u t ( n ). The Doppler frequency estimate f' D . The Doppler compensation signal u d ( n ) is obtained by compensating the echo signal u r ( n ) at the Doppler frequency f' D . u d ( n ) is demodulated into 64 subcarrier complex signals by 64-FFT (fast Fourier transform) . Use the Maximum Likelihood (ML) decoding to obtain the spread spectrum signal . The spread spectrum signal that will be solved Spread spectrum signal with reference Perform de-spreading. It is important to note that the noises n t ( n ) and n r ( n ) of the reference and echo signals are statistically independent, so the ML decoding errors caused by the noises n t ( n ) and n r ( n ) For irrelevance, the de-spread autocorrelation processing output can eliminate the noise E t and E r . In other words, even at the low signal-to-noise ratio, when the correct Doppler frequency estimation and compensation is completed, the despread signal will still generate the maximum output power via the pulse compression process, and the target direction is estimated via the multi-mode beamforming process.
如圖14所示，，將展頻序列C m 進行四相相移鍵控信號(Quadrature Phase Shift Keying，QPSK)調變(星雲編碼)，令星雲編碼器的輸出為As shown in FIG. 14, the spread spectrum sequence C m is subjected to Quadrature Phase Shift Keying (QPSK) modulation (nebula coding), so that the output of the nebula encoder is
式中函數Mapping {}代表星雲編碼器的映射函數，當函數為QPSK調變時，是將長度L =log2 4=2的二位元序列映射到一個複數，而直接序列信號之碼長度P =64L 。MFS-MC-DSSS離散基頻訊號為The function Mapping {} represents the mapping function of the nebula encoder. When the function is QPSK modulation, the two-bit sequence of length L = log 2 4=2 is mapped to a complex number, and the code length of the direct sequence signal is P. =64 L . The MFS-MC-DSSS discrete fundamental frequency signal is
由(7)與(13)可知參考信號與回波信號分別經過混波器降頻到基頻可以表示為It can be seen from (7) and (13) that the reference signal and the echo signal are respectively down-converted to the fundamental frequency by the mixer, which can be expressed as
其中目標徑向速度v 遠小於光速c ，所以且為都卜勒頻率。若類比轉數位器以取樣率f s 對參考信號u t (t )與回波信號u r (t )進行取樣，則Where the target radial velocity v is much smaller than the speed of light c , so And For the Doppler frequency. If the analog to digital counter samples the reference signal u t ( t ) and the echo signal u r ( t ) at a sampling rate f s , then
其中表示取最接近x的整數。若以接收的直接路徑參考基頻信號u t (t )的接收時間為參考點，則(20)可以表示為among them Represents the integer closest to x. If the received time of the received direct path reference baseband signal u t ( t ) is taken as a reference point, then (20) can be expressed as
利用離散傅立葉轉換(discrete-time Fourier transform,DFT)之變數擴張與時間延遲性質，由(18)可得The variable expansion and time delay properties of discrete-time Fourier transform (DFT) are obtained by (18)
將(22)代入(21)，可得Substituting (22) into (21), you can get
其中，Δf =f s /64為相鄰子載波之間隔頻率。among them , Δ f = f s /64 is the interval frequency of adjacent subcarriers.
由(23)可知子載波b k 在接收端會具有n d 點的相位差與k D 點的頻率位移。64-FFT輸出的第k' 個子載波函數值可表示為It can be seen from (23) that the subcarrier b k has a phase difference of n d point and a frequency shift of the k D point at the receiving end. The value of the k'th subcarrier function of the 64-FFT output can be expressed as
因為分別為τ與v 的函數代入(24)可得because Substituting functions for τ and v respectively (24)
則第k' 個子載波函數值之QPSK解調器輸出序列為Then the k'th subcarrier function value The QPSK demodulator output sequence is
式中函數deMapping { }代表QPSK解調器的映射函數，此函數將一個複數映射到長度L 的序列，則64個子載波函數值之星雲解碼輸出序列重新組成回波信號的展頻碼序列與訓練信號的序列做解展頻，產生脈波壓縮輸出功率值。The function deMapping { } represents the mapping function of the QPSK demodulator, which maps a complex number to a sequence of length L , and the nebula decoded output sequence of 64 subcarrier function values reconstitutes the spread spectrum code sequence of the echo signal. Sequence with training signals The solution spread spectrum is generated to generate a pulse compression output power value.
MFS-MC-DSSS雷達波形之混淆函數可以用(25)、(26)與(27)式產生。因為用來估測距離的反射回波信號為低訊雜比，恢復的直接序列碼具有很高的位元錯誤率(以QPSK調變為例，在-10dB訊雜比的位元錯誤率為0.2738)，而無法產生足夠的脈波壓縮功率。因此本發明提出以倍頻率展頻方式降低恢復的直接序列碼的位元錯誤率。SF 倍頻率展頻處理是將相鄰的SF 個子載波傳送相同QPSK信號：The aliasing function of the MFS-MC-DSSS radar waveform can be generated using equations (25), (26), and (27). Because the reflected echo signal used to estimate the distance is a low signal-to-noise ratio, the recovered direct sequence code It has a very high bit error rate (in the case of QPSK modulation, the bit error rate of the signal-to-noise ratio is -0.2738), and it cannot generate enough pulse compression power. Therefore, the present invention proposes to reduce the recovered direct sequence code by frequency doubling. Bit error rate. The SF frequency spread spectrum processing is to transmit the same QPSK signal to adjacent SF subcarriers:
在接收端做倍頻率解展頻Doing frequency multiplication spread spectrum at the receiving end
所以可以獲得SF 倍頻率展頻增益。以-16dB訊雜比的QPSK調變為例，經過八倍頻率展頻可以提高9dB的訊雜比，位元錯誤率可以降低為0.229。因為使用四個天線元，多模式波束成型產生的最大天線增益為6dB，位元錯誤率可以進一步降低為0.09838。因此目標偵測機率P D 會大於1-0.09838=0.9016。Therefore, the SF frequency spread gain can be obtained. In the example of QPSK modulation with a signal-to-noise ratio of -16dB, the signal-to-noise ratio of 9dB can be improved by eight times frequency spreading, and the bit error rate can be reduced to 0.229. Since four antenna elements are used, the maximum antenna gain produced by multimode beamforming is 6 dB, and the bit error rate can be further reduced to 0.09838. Therefore, the target detection probability P D will be greater than 1-0.09838=0.9016.
多載波直接序列倍頻率展頻(MFS-MC-DSSS)雙態雷達系統模擬參數表，如表1所示。圖15~18為在SNRr =-16dB,SNRt =4dB訊雜比條件下，不同頻率展頻倍率(SF =1,2,4,8)的MFS-MC-DSSS交相關函數。由圖15~18可知頻率展頻八倍抗雜訊之性能最好，-16dB的低雜訊比時仍然可以達到1.6dB的脈波壓縮增益，展頻四倍可以達到1dB的脈波壓縮增益，而當低於兩倍的展頻倍率則無法產生足夠的脈波壓縮增益。由環形陣列天線之多波束數位波束成型處理規格確定其解析度為波束寬度(22.5°)，再由多模式數位波束成型處理之目標來向估測模式進行目標來向細估。振幅比較輸出的差場型有相同的角斜率為1.3dB/度。Multi-carrier direct sequence frequency spread spectrum (MFS-MC-DSSS) two-state radar system simulation parameter table, as shown in Table 1. Figure 15~18 shows the MFS-MC-DSSS cross-correlation function for different frequency spread ratios ( SF = 1, 2, 4, 8) with SNR r = -16dB and SNR t = 4dB. It can be seen from Fig. 15~18 that the performance of the frequency spread spectrum is eight times better than that of the noise. The low noise ratio of -16dB can still achieve the pulse compression gain of 1.6dB, and the spread spectrum can achieve the pulse compression gain of 1dB by four times. However, when the frequency is less than twice the spread frequency, sufficient pulse compression gain cannot be generated. The multi-beam digital beamforming processing specification of the circular array antenna determines its resolution to be the beam width (22.5°), and then the target of the multi-mode digital beamforming process is used to make a target estimation to the estimation mode. The difference field of the amplitude comparison output has the same angular slope of 1.3 dB/degree.
綜上所述，本發明之結構特徵及各實施例皆已詳細揭示，而可充分顯示出本發明案在目的及功效上均深富實施之進步性，極具產業之利用價值，且為目前市面上前所未見之運用，依專利法之精神所述，本發明案完全符合發明專利之要件。In summary, the structural features and embodiments of the present invention have been disclosed in detail, and can fully demonstrate that the present invention has deep progress in the purpose and efficacy of the present invention, and has great industrial value, and is currently The unprecedented use in the market, according to the spirit of the patent law, the invention is fully in line with the requirements of the invention patent.
唯以上所述者，僅為本發明之較佳實施例而已，當不能以之限定本發明所實施之範圍，即大凡依本發明申請專利範圍所作之均等變化與修飾，皆應仍屬於本發明專利涵蓋之範圍內，謹請 貴審查委員明鑑，並祈惠准，是所至禱。The above is only the preferred embodiment of the present invention, and the scope of the present invention is not limited thereto, that is, the equivalent variations and modifications made by the scope of the present invention should still belong to the present invention. Within the scope of the patent, I would like to ask your review committee to give a clear understanding and pray for it. It is the prayer.
1‧‧‧多載波直接展頻序列發射機1‧‧‧Multi-carrier direct spread spectrum sequence transmitter
11‧‧‧多載波直接展頻序列倍頻率展頻波形產生器11‧‧‧Multi-carrier direct spread spectrum sequence frequency spread spectrum generator
111‧‧‧脈波產生器111‧‧‧ Pulse generator
112‧‧‧直接序列展頻器112‧‧‧Direct Sequence Spreader
113‧‧‧四相相移鍵控信號調變器113‧‧‧Quad Phase Shift Keying Signal Transducer
114‧‧‧倍頻率展頻器114‧‧‧ times frequency spreader
115‧‧‧64-IFFT115‧‧‧64-IFFT
116‧‧‧數位/類比轉換器116‧‧‧Digital/Analog Converter
12‧‧‧升頻器12‧‧‧Upconverter
2‧‧‧環形天線接收機2‧‧‧loop antenna receiver
21‧‧‧環形陣列天線21‧‧‧Circular Array Antenna
22‧‧‧線性補償前置處理器22‧‧‧Linear compensation preprocessor
23‧‧‧降頻轉換器23‧‧‧down converter
231‧‧‧柴比雪夫窗戶處理器231‧‧‧Chebyshev window processor
232‧‧‧降頻器232‧‧‧Downer
233‧‧‧類比/數位轉換器233‧‧‧ Analog/Digital Converter
24‧‧‧零化處理器24‧‧‧Zero processor
241‧‧‧多模式數位波束成型器241‧‧‧Multi-mode digital beamformer
2411‧‧‧多波束成型模組2411‧‧‧Multibeam Forming Module
2412‧‧‧振幅比較目標來向估測模組2412‧‧‧Amplitude comparison target to estimate module
2413‧‧‧零化指向波束成型模組2413‧‧‧Zeroized beamforming module
25‧‧‧移動目標指示處理器25‧‧‧Mobile Target Indication Processor
26‧‧‧倍頻率解展頻電路26‧‧‧ times frequency despreading frequency circuit
261‧‧‧64-FFT261‧‧‧64-FFT
262‧‧‧倍頻率解展頻器262‧‧‧ times frequency despreader
263‧‧‧四相相移鍵控信號調解器263‧‧‧Quad Phase Shift Keying Signal Mediator
27‧‧‧交相關處理器27‧‧‧Related processor
271‧‧‧64-FFT271‧‧‧64-FFT
272‧‧‧振幅相關處理器272‧‧‧Amplitude related processor
273‧‧‧64點循環位移器273‧‧64-bit cycle shifter
274‧‧‧倍頻率解展器274‧‧‧ times frequency despreader
275‧‧‧四相相移鍵控信號調解器275‧‧‧Quad Phase Shift Keying Signal Mediator
276‧‧‧直接序列解展頻器276‧‧‧Direct Sequence Despreader
277‧‧‧數位頻率合成器277‧‧‧Digital Frequency Synthesizer
28‧‧‧目標偵測器28‧‧‧Target detector
281‧‧‧目標偵測之最大值判斷器281‧‧‧Maximum value detector for target detection
29‧‧‧目標來向估測器29‧‧‧Target to the estimator
第1圖為低訊雜比目標來向估測接收機架構示意圖Figure 1 is a schematic diagram of the low-to-noise ratio target to estimate the receiver architecture.
第2圖為環形16個天線元等波束寬陣列天線架構Figure 2 is a beamwidth array antenna structure with 16 antenna elements
第3圖為天線元分組示意圖Figure 3 is a schematic diagram of antenna element grouping
第4圖為陣列天線經路徑補償示意圖Figure 4 is a schematic diagram of the path compensation of the array antenna.
第5圖為等寬波束之場型圖Figure 5 is a field diagram of a beam of equal width
第6圖為數位波束成型實現架構圖Figure 6 is a digital beamforming implementation architecture diagram
第7圖為多波束成型之相鄰兩波束場型圖Figure 7 is the adjacent two beam pattern of multi-beamforming
第8圖為相鄰兩波束之差場型圖Figure 8 is the difference field diagram of two adjacent beams.
第9圖為多模式數位波束成型器架構圖Figure 9 is a multi-mode digital beamformer architecture diagram
第10圖為多模式數位波束成型器硬體架構圖Figure 10 is a multi-mode digital beamformer hardware architecture diagram
第11圖為主要陣列天線場型圖Figure 11 shows the main array antenna pattern
第12圖為輔助陣列天線場型圖Figure 12 is the auxiliary array antenna pattern
第13圖為合成陣列天線場型圖Figure 13 is a synthetic array antenna pattern
第14圖為多載波直接展頻序列倍頻率展頻雙態相列雷達傳接機裝置方塊圖Figure 14 is a block diagram of a multi-carrier direct spread spectrum sequence double frequency spread spectrum binary state phased radar relay device
第15圖為多載波直接展頻序列倍頻率展頻交相關函數圖(SF =8)Figure 15 is a multi-carrier direct spread spectrum sequence frequency spread spectrum cross correlation function graph ( SF = 8)
第16圖為多載波直接展頻序列倍頻率展頻交相關函數圖(SF =4)Figure 16 is a multi-carrier direct spread spectrum sequence frequency spread spectrum cross correlation function graph ( SF = 4)
第17圖為多載波直接展頻序列倍頻率展頻交相關函數圖(SF =2)Figure 17 is a multi-carrier direct spread spectrum sequence frequency spread spectrum correlation function graph ( SF = 2)
第18圖為多載波直接展頻序列倍頻率展頻交相關函數圖(SF =1)Figure 18 is a multi-carrier direct spread spectrum sequence frequency spread spectrum correlation function graph ( SF =1)
1．．．多載波直接展頻序列發射機1. . . Multi-carrier direct spread spectrum sequence transmitter
11．．．多載波直接展頻序列倍頻率展頻波形產生器11. . . Multi-carrier direct spread spectrum sequence frequency spread spectrum waveform generator
12．．．升頻器12. . . Upconverter
2．．．環形天線接收機2. . . Loop antenna receiver
21．．．環形陣列天線twenty one. . . Circular array antenna
22．．．線性補償前置處理器twenty two. . . Linear compensation preprocessor
23．．．降頻轉換器twenty three. . . Down converter
24．．．零化處理器twenty four. . . Zero processor
25．．．移動目標指示處理器25. . . Moving target indication processor
26．．．倍頻率解展頻電路26. . . Double frequency despreading frequency circuit
27．．．交相關處理器27. . . Interrelated processor
28．．．目標偵測器28. . . Target detector
29．．．目標來向估測器29. . . Target to estimator
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| CN111521968B (en) * | 2020-05-22 | 2022-05-20 | 南京理工大学 | Underdetermined DOA estimation method based on target space diversity |
| TWI806456B (en) * | 2022-03-01 | 2023-06-21 | 瑞昱半導體股份有限公司 | Signal receiving apparatus and operation method thereof having phase compensation mechanism |
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| CN104158777B (en) * | 2014-08-28 | 2017-09-08 | 中国民航大学 | United orthogonal projects the rangefinder pulse interference suppression method with circulation beam forming |
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