TWI321392B - A high-efficiency current-source inverter using resonant technique - Google Patents

Description

1321392 九、發明說明： 【發明所屬之技術領域】 本發明所涉及之技術領域包含電力電子、直流/交流轉 換技術及綠色能源科技之範疇，但其主要在於以直流電源 (如燃料電池、風力、太陽能等）作為輸入電源，將其轉換 為交流電源，以提供緊急電源或一般交流電器負載使用； 本裝置由交流正弦電壓命令與回授電壓之誤差，於每一週 期中控制箝制開關導通時間，利用耦合電感產生電流源 (Current Source)，經過全橋開關正、負週期導引對輸出電 容充電，調整電壓上升或下降之幅度以累積交流弦波輸出 電壓。 【先前技術】 目前市面上將直流電轉成60Hz交流電壓之產品大致 分成兩類；第一類是應用於交流馬達之變頻器，利用馬達 之線圈電感特性，將正弦脈波寬度調變電壓波形產生近似 正弦電流，這種架構不能適用於電阻性或電容性負載，嚴 格來說變頻器不能供應一般家電及電腦產品。第二類即針 對前者缺點所推出產品，典型的商品為不斷電設備（UPS) 為代表其架構圖如圖1 (a)所示。與前類比較，輸出端增加 電感串聯電容之LC濾波電路，並增加回授控制，使輸出 電壓固定以克服負載及輸入電壓變動，這類型產品最大優 點為架構簡單而且成本低廉，另外增加電池及充、放電電 路，以提供市電以外之緊急電源。以目前台灣製造UPS之 1321392 - 技術及產品佔有率，如台達電子與飛瑞，已名列世界前茅。 . 即使如此，仍有若干特性仍須繼續改善。首先LC濾波電 路有諸多負載限制：第一點濾波電感貫穿整個輸出電流， 且考慮偏移二階諧振電路中半功率頻率（_3dB)響應，電感 值及容量較大，一般市售UPS所使用之電感值約在w/f範 圍，濾波電感提高會增加產品重量與能量轉換損失。第二 ' 點，加諸於電感兩端電壓I·沿/汾為直流電壓與輸出電容之 差，其值於正弦峰值附近最小，受限滤波電感值，導致正 Φ 弦電壓峰值轉折點失真，產生高諧波成分，即使提高濾波 電壓仍無法避免。此電感提供濾波功能，但同時限制非線 性負載，瞬間負載變化之調節能力。第三點，若干負載將 . 危及驅動電路，如半波整流性負載或高電感性負載，主要 因LC濾波電路之正負半周波形對稱性，以及高電感性負 載改變二階濾波之頻率響應，輸出正弦電壓過低，進而必 須調高直流電壓準位，系統可能過壓而燒毁。第四點，非 電阻性負載之電壓波形失真率，一般又稱為整體諧波失真 φ 率（Total Harmonic Distortion，THD)，遠高於電阻性負載。 這歸咎於原先設計之二階濾波電路特性不能滿足非電阻性 負載如電容性、電感性及非線性負載。總而言之，目前反 流器的產品，不能全面提供各類型的負載，對於消費者不 但不便，而且顯有使用者能夠分辨負載類型，只能各憑運 氣買不斷電設備，也許購買加大一級容量，是一個可行之 _ 道。 . 除此之外，開關切換損失隨著切換頻率提高而增加， 1321392 - 系統效率因而降低。許多廠商已經開始引用各種柔性切換 . 技術應用於大功率IGBT開關元件，若干文獻中已證明可 降低PWM切換損失進而提高切換頻率，改善輸出電壓波 形。相對於傳統正弦脈波寬度調變電壓波形，電流源反流 器之正弦電壓，大部分控制電流源對電容充電以累積正弦 ' 電壓，可以承受各種不同負載與頻率變化，但為何市面上 ' 少有此種產品，分析原因乃電流源之電感太大，控制電感 迴路與柔性切換技術不易實現，故而效率不高。 | 一般具有柔性切換的諧振電路是經常引用之技術，電 壓諧振型之電流值與電流諧振之電壓值，在諧振之過程中 易發生異常情形，一旦電壓超過額定，即立刻損壞半導體 開關，因此發展電壓增益不受負載影響之諧振電路[1]，[2]。 但由於譜振電路之能量利用率太低，將近一半能量會流回 電源測，於是後續專家學者提出具有電壓箝制電路的半諧 振電壓源反流器，以減少開關切換應力並使用脈波寬度調 變技術來控制反流器輸出波形P]-[5]。受限於開關導通時 • 間需搭配半諧振週期，於是發展部份諧振電路[6],[7]，將諧 振波形侷限於開關導通或截止交越區間，因此切換頻率可 以為定值，責任週期不必配合諧振時間。新近有學者研究 將濾波電感置放於直流電源側與全橋反流器之間，如圖1(b) 所示，降低電感值並提高動態響應能力，形成電流源反流 器架構，用以改善傳統電壓源架構之缺點。但電流截止所 產生切換電壓值常數倍於系統電壓，易危及開關元件，是 •故，開關元件耐壓值必須相對提升，成本大幅增加，不利 於此架構之_ 技術之瓶<7_應用。最近提出電流源柔性切換架構[8]解決該 ^ 1 項’此乃應用於太陽能發電系統饋入市電之反流 具有零電壓切換(Zer。v。1邮 梦入：兒 w切換（Zero Current Switching，ZCS)之效果， 減少市文功率因數接近一，但電感之體積及數量仍有待 裝置設^刀換頻率低（5kHz)，造成高漣波輸出電壓。然因 二該f置I的為直接饋入市電，並不直接供應給負載，所 哭=構^供此系統之電源品質已經優於目前電壓源反流 二，2續提出電壓箝制與柔性切換雙效果之反流器機 雷壓箱J:於三相交流馬達驅動器’利用變壓器二次側做 路電流，整體電感值降低且轉換效率高，基 大，現電流源架構之特點，惟電流源之電感環流太 獻中：C:、不易’理論上是變壓器銅損會很高，但文 q |變壓器損失。除電壓波形漣波高外，此架構 :::點是開關需承受四倍直流電源電壓，開關成本過 冋驅董子象為感應馬達，不利於之消費性產品應用。 卜新開發低頻高階諧振之正弦電壓反流器[10]， 此種木構非丰簡單，利用幾何平均頻率（。⑶则此河⑽ Frequency)之特性並於開迴路狀態下，電壓調節率有不錯 表現，然㈣振電感容量太大以及三切波成分太高是其 缺』電$箝制與柔性切換效能之電流源正弦電壓反流器 [11]，Π2]彻低激磁電感變壓器—次側作為電流源電感， 將開關而承又四倍直流電源電壓降為兩倍，其全橋四個開 關同寺”有令電壓與零電流切換。然而，該四個開關必須 1321392 造成輪出電;之短路=是週期反相切換時’ _冓中全橋開關所串聯二極體阻斷 =同:::行:_車效能，此為大部分== 肌盗/、同之缺點。其次，該電流源電 需谷量降低，但為確保箝制開關導通時電」九; :::換1，漣波電流高，且因切換頻4 =加;: ;’令。取後’為大幅提高麵合係數壓 、於 =困難之三明治繞法，而且二次側繞組電;:= 二達輸出電磨三分之一，這種高環流未導 入輸出端，導致輕載轉換效率特別低。 =:=r論來限制系_，並：= 負載直接向頻切換充電控制，累積正弦波電㈣ 之 tut全部功率半導體_元件具零_或零電流 柔性切換特性，進而提高反流器之輸出效率。 本發明改善先前技術之原理及對照功效如下·· 電愿箝制技術、搞合電感電流能量互遞以及於言皆振 反〜器令並聯諧振技術，使得全部功率半導體開關及二 極體均有柔性切換特性，最高轉換效率大於咖； 2.運用本發明電覆箝制技術，可降低所有功率半導體開關 兀件之耐壓規格，其咖約等於或小於輸入之直流電源 1321392 電壓，其值遠低於上述文獻記載。 3. 具再生式剎車功能：由於本發明之全橋開關不需串聯二 極體，並提供一不需控制之剎車二極體迴路，使得交流 能量得以逆流至直流電源端，更進一步箝制所有系統元 件之電壓。 4. 本發明所需之耦合電感容量與體積均小於一般電流源架 構，且耦合電感可操作於連續電流模式並能快速調整電 流以供應負載所需，可提高切換頻率有助降低輸出電容 之容量。當功率半導體開關觸發信號導通時，若二次側 繞組電流在觸發信號導通前已降到零，則為電流不連續 模式（Discontinue Current Mode, DCM)，因此自然形成 導通具有ZCS現象；同理，若為電流連續模式（Continue Current Mode, CCM)，受限耦合電感能量傳遞影響，仍 然形成導通具有ZCS特性。 5. 全橋反流器之功率半導體開關可省略習用架構之四個串 接二‘體，且利用並聯諧振特性，來處理輸出電壓極性 交越換向，不需透過任何開關元件，因此可以省略習用 電流源反流器常用之串聯二極體； 6. 輸出端可省略習用高容量之串聯濾波電感，電流源直接 對輸出負載及濾波電容充電，因此可接受各種電感性、 電容性、非線性及瞬間變化之負載，且輸出電壓波形失 真率及傅立葉頻譜分析均優於傳統脈波寬度調變架構。 備註：參考文獻 [1] F. J. Lin, R. Y. Duan, and J. C. Yu, aAn ultrasonic motor drive 12 1321392 using a current-source parallel-resonant inverter with energy feedback,Trans. Power Electron., vol. 14, no. 1, pp. 31-42, 1999.1321392 IX. INSTRUCTIONS: [Technical Fields of the Invention] The technical field to which the present invention relates includes power electronics, DC/AC conversion technology, and green energy technology, but mainly uses DC power sources (such as fuel cells, wind power, Solar energy, etc.) is used as an input power source to convert it to an AC power source to provide emergency power or general AC electrical load. The device controls the on-time of the clamp switch in each cycle by the error of the AC sinusoidal voltage command and the feedback voltage. The current source is generated by the coupled inductor, and the output capacitor is charged by the full-bridge switching positive and negative cycle guidance, and the voltage rise or fall is adjusted to accumulate the AC sine wave output voltage. [Prior Art] At present, the products that convert DC power into 60Hz AC voltage on the market are roughly divided into two categories; the first type is an inverter applied to an AC motor, which uses a coil inductance characteristic of the motor to generate a sinusoidal pulse width modulation voltage waveform. Approximate sinusoidal current, this architecture cannot be applied to resistive or capacitive loads. Strictly speaking, inverters cannot supply general household appliances and computer products. The second category is the product launched for the shortcomings of the former. The typical product is the uninterruptible power equipment (UPS) as its structural diagram is shown in Figure 1 (a). Compared with the former class, the output increases the LC filter circuit of the inductor series capacitor, and increases the feedback control to make the output voltage fixed to overcome the load and input voltage variation. The biggest advantage of this type of product is that the architecture is simple and the cost is low, and the battery and Charge and discharge circuits to provide emergency power supplies other than utility power. At present, Taiwan's UPS 1321392 - technology and product share, such as Delta Electronics and Feirui, has been ranked among the best in the world. Even so, there are still a number of features that still need to continue to improve. First, the LC filter circuit has many load limitations: the first point filter inductor runs through the entire output current, and considers the half-power frequency (_3dB) response in the offset second-order resonant circuit, the inductance value and the capacity are large, and the inductor used in the commercially available UPS is generally used. The value is approximately in the w/f range, and increased filter inductance increases product weight and energy conversion losses. The second 'point, applied to the voltage across the inductor I · along / 汾 is the difference between the DC voltage and the output capacitance, the value is the smallest near the sinusoidal peak, the limited filter inductance value, resulting in the positive Φ string voltage peak turning point distortion, resulting High harmonic components, even if the filter voltage is increased, cannot be avoided. This inductor provides filtering, but at the same time limits the ability to adjust for non-linear loads and transient load changes. Third, some loads will endanger the drive circuit, such as half-wave rectifying load or high-inductive load, mainly due to the positive and negative half-cycle waveform symmetry of the LC filter circuit, and the high-inductive load changing the frequency response of the second-order filter, output sine If the voltage is too low, the DC voltage level must be increased and the system may be over-pressed and burned. Fourth, the voltage waveform distortion rate of non-resistive loads is also commonly referred to as Total Harmonic Distortion (THD), which is much higher than resistive load. This is due to the fact that the previously designed second-order filter circuit characteristics cannot satisfy non-resistive loads such as capacitive, inductive and non-linear loads. All in all, the current products of the inverter cannot fully provide various types of loads. It is not only inconvenient for the consumers, but also the users can distinguish the types of loads. Only the luck can be used to buy the uninterrupted equipment. Is a feasible way. In addition, the switching loss increases as the switching frequency increases, 1321392 - system efficiency is thus reduced. Many manufacturers have begun to cite various flexible switching techniques. The technology is applied to high-power IGBT switching components, and several documents have been shown to reduce PWM switching losses, increase switching frequency, and improve output voltage waveform. Compared to the traditional sinusoidal pulse width modulation voltage waveform, the sinusoidal voltage of the current source inverter, most of the control current source charges the capacitor to accumulate the sinusoidal voltage, can withstand various load and frequency changes, but why the market is less There is such a product, the reason for the analysis is that the inductance of the current source is too large, and the control inductance loop and the flexible switching technology are not easy to realize, so the efficiency is not high. The resonant circuit with flexible switching is a frequently cited technology. The voltage value of the voltage resonance type and the voltage value of the current resonance are prone to abnormal conditions during the resonance process. Once the voltage exceeds the rated value, the semiconductor switch is damaged immediately, so the development The voltage gain is not affected by the load of the resonant circuit [1], [2]. However, since the energy utilization rate of the spectral vibration circuit is too low, nearly half of the energy will flow back to the power supply measurement, so follow-up experts have proposed a semi-resonant voltage source inverter with a voltage clamping circuit to reduce the switching stress and use the pulse width modulation. Variable technology to control the inverter output waveform P]-[5]. When the switch is turned on, it needs to be matched with the semi-resonant period. Therefore, some resonant circuits [6], [7] are developed, and the resonant waveform is limited to the switch on or off crossover interval. Therefore, the switching frequency can be fixed. The period does not have to match the resonance time. Recently, some scholars have studied to place the filter inductor between the DC power supply side and the full-bridge inverter. As shown in Figure 1(b), the inductor value is reduced and the dynamic response capability is improved to form a current source inverter structure. Improve the shortcomings of traditional voltage source architectures. However, the switching voltage value generated by the current cutoff is twice as large as the system voltage, which jeopardizes the switching components. Therefore, the withstand voltage value of the switching component must be relatively increased, and the cost is greatly increased, which is not conducive to the architecture of the technology_7_Application . Recently, a current source flexible switching architecture [8] has been proposed to solve this problem. This is applied to the solar power system to feed the commercial power. The reverse current has zero voltage switching (Zer.v. 1 mail dream: Zero Current Switching) , ZCS) effect, reducing the power factor of the city is close to one, but the volume and quantity of the inductor still have to be set to a low frequency (5 kHz), resulting in a high chopping output voltage. However, the f is set to direct feed. The mains supply is not directly supplied to the load. The power supply for this system is better than the current voltage source reverse flow. 2 Continued to propose a voltage clamp and flexible switching dual-effect inverter machine. The three-phase AC motor driver uses the secondary side of the transformer to make the current, the overall inductance value is reduced and the conversion efficiency is high, the base is large, and the current source architecture is characterized. However, the current source inductance loop is too concentrated: C:, not easy to 'theory On the transformer copper loss will be very high, but the loss of the transformer q. In addition to the voltage waveform ripple high, this architecture::: point is the switch has to withstand four times the DC power supply voltage, the switch cost over the drive Dongzi Xiang for the induction horse It is not conducive to the application of consumer products. Bu Xin developed a sinusoidal voltage inverter with low-frequency high-order resonance [10]. This kind of wood structure is not simple, and it uses the characteristic of geometric mean frequency ((3) then this river (10) Frequency) In the open loop state, the voltage regulation rate has a good performance, but (4) the vibration inductance capacity is too large and the three-cut wave component is too high is the lack of electric current clamped and flexible switching efficiency current source sinusoidal voltage inverter [11], Π 2 The low-inductive inductive transformer—the secondary side acts as the current source inductor, which reduces the voltage of the switch and the quadruple DC power supply by a factor of two, and the four bridges of the full bridge are switched to the voltage and zero current. However, the four The switch must be 1321392 to cause the wheel to be powered out; the short circuit = is the period of the reverse phase switching ' _ 冓 全 full bridge switch series diode blocking = same ::: line: _ car performance, this is the majority == muscle Secondly, the current source needs to reduce the amount of electricity, but to ensure that the clamp switch is turned on when electricity is turned on; ::: for 1, the chopping current is high, and because of the switching frequency 4 = plus; 'Order. After taking' is a sandwich that greatly increases the face factor pressure, and = difficult Winding method, and the secondary side winding electricity;:= Two-point output electric grinding one-third, this high circulating current is not introduced into the output end, resulting in light load conversion efficiency is particularly low. =:=r on the limit system _, and := Load direct-to-frequency switching charge control, cumulative sinusoidal power (4) tut all power semiconductors_components with zero- or zero-current flexible switching characteristics, thereby improving the output efficiency of the inverter. The present invention improves the principle and comparison of the prior art The functions are as follows: · The electric clamp technology, the inductive current energy transfer and the parallel resonance technology make the power switching switch and the diode have flexible switching characteristics, and the highest conversion efficiency is greater than the coffee; 2. Using the electric capping technology of the present invention, the withstand voltage specification of all power semiconductor switch components can be reduced, and the coffee is approximately equal to or less than the input DC power supply 1321392 voltage, which is much lower than the above document. 3. Regenerative brake function: Since the full bridge switch of the present invention does not require a series diode and provides an uncontrolled brake diode circuit, the AC energy can be reversed to the DC power supply terminal, further clamping all systems. The voltage of the component. 4. The coupling inductor capacity and volume required by the present invention are smaller than the general current source architecture, and the coupled inductor can operate in a continuous current mode and can quickly adjust the current to supply the load, and the switching frequency can help reduce the capacity of the output capacitor. . When the power semiconductor switch trigger signal is turned on, if the secondary side winding current has dropped to zero before the trigger signal is turned on, it is a discontinuous current mode (DCM), so naturally forming a conduction has a ZCS phenomenon; similarly, In the case of Continuous Current Mode (CCM), the limited coupling inductance energy transfer effect still forms conduction with ZCS characteristics. 5. The power semiconductor switch of the full-bridge inverter can omit the four serial two-body of the conventional architecture, and use the parallel resonance characteristic to handle the polarity crossover of the output voltage. It does not need to pass any switching elements, so it can be omitted. Conventional current source inverter commonly used series diode; 6. The output terminal can omit the conventional high-capacity series filter inductor, the current source directly charges the output load and the filter capacitor, so it can accept various inductive, capacitive, nonlinear And the transient load, and the output voltage waveform distortion rate and Fourier spectrum analysis are better than the traditional pulse width modulation architecture. Remarks: References [1] FJ Lin, RY Duan, and JC Yu, aAn ultrasonic motor drive 12 1321392 using a current-source parallel-resonant inverter with energy feedback, Trans. Power Electron., vol. 14, no. Pp. 31-42, 1999.

[2] F. J. Lin, R. Y. Duan, R. J. Wai and C. M. Hong, 4iLLCC resonant inverter for piezo-electric ultrasonic motor drive,5, IEE Proc. Electric Power Appl, vol. 146, no. 5, pp. 479-487, 1999.[2] FJ Lin, RY Duan, RJ Wai and CM Hong, 4iLLCC resonant inverter for piezo-electric ultrasonic motor drive, 5, IEE Proc. Electric Power Appl, vol. 146, no. 5, pp. 479-487, 1999 .

[3] L. Malesani, P. Tenti, P. Tomasin，and V· Toigo, “High efficiency quasi-resonant DC link three-phase power inverter for full-range FWM,^ IEEE Trans. Ind. Αρρί, vol. 31, pp. 141-147, 1995.[3] L. Malesani, P. Tenti, P. Tomasin, and V· Toigo, “High efficiency quasi-resonant DC link three-phase power inverter for full-range FWM, ^ IEEE Trans. Ind. Αρρί, vol. 31 , pp. 141-147, 1995.

[4] V. V. Deshpande, and S. R. Doradla, “A new topology for parallel resonant DC link with reduced peak voltage,IEEE Trans. Ind.[4] V. V. Deshpande, and S. R. Doradla, “A new topology for parallel resonant DC link with reduced peak voltage, IEEE Trans. Ind.

Appl., vol. 32, pp. 310-307, 1996.Appl., vol. 32, pp. 310-307, 1996.

[5] S. Chen, and T. A. Lipo,44A novel soft-switched PWM inverter for AC motor drivers,IEEE Trans. Power Electron.^ vol. 11, pp. 653-659, 1996.[5] S. Chen, and T. A. Lipo, 44A novel soft-switched PWM inverter for AC motor drivers, IEEE Trans. Power Electron. ^ vol. 11, pp. 653-659, 1996.

[6] P· C. Theron, and J. A. Ferreira, “The zero voltage switching partial series resonant converter,^ IEEE Trans. Ind. Αρρί, vol 31, pp. 879-886, 1995.[6] P. C. Theron, and J. A. Ferreira, “The zero voltage switching partial series resonant converter, ^ IEEE Trans. Ind. Αρρί, vol 31, pp. 879-886, 1995.

[7] C. S· Moo, Y. C. Chuang, and C. R. Lee, “A new power - factor -correction circuit for electronic ballasts with series-load resonant inverter,5, IEEE Trans. Power Electron., vol. 13, pp. 273-278, 1998.[7] C. S. Moo, YC Chuang, and CR Lee, “A new power - factor -correction circuit for electronic ballasts with series-load resonant inverter, 5, IEEE Trans. Power Electron., vol. 13, pp. 273-278, 1998.

[8] R. Itoh, K. Ishizaka, H. Oishi and H. Okada, “Soft-switched current-source inverter for single-phase utility interfaces,” 13 1321392[8] R. Itoh, K. Ishizaka, H. Oishi and H. Okada, “Soft-switched current-source inverter for single-phase utility interfaces,” 13 1321392

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[9] H· Ishikawa, and Y· Murai，“A novel soft-switched PWM current source inverter with voltage clamped circuit/5 IEEE Trans. Power Electron., vol. 15, no. 6, pp. 1081-1087, 2000.[9] H. Ishikawa, and Y· Murai, “A novel soft-switched PWM current source inverter with voltage clamped circuit/5 IEEE Trans. Power Electron., vol. 15, no. 6, pp. 1081-1087, 2000 .

[10] R. J. Wai，R. Y. Duan, J. D. Lee，and L. W. Liu，“High-efficiency fuel cell power inverter with soft-switching resonant technique,5, IEEE Transactions on Energy Conversion, vol. 20, no. 2, pp. 485-492, 2005.[10] RJ Wai, RY Duan, JD Lee, and LW Liu, "High-efficiency fuel cell power inverter with soft-switching resonant technique, 5, IEEE Transactions on Energy Conversion, vol. 20, no. 2, pp. 485 -492, 2005.

[11] R. J. Wai and R. Y· Duan，“High-efficiency power conversion for low power fuel cell generation system,IEEE Trans. Power Electronics, vol. 20, no. 4, pp. 847-856, 2005.[11] R. J. Wai and R. Y. Duan, "High-efficiency power conversion for low power fuel cell generation system, IEEE Trans. Power Electronics, vol. 20, no. 4, pp. 847-856, 2005.

[12] R. J. Wai, R. Y. Duan, and L. W. Liu, UA current-source sine wave voltage driving circuit via voltage-clamping and soft-switching techniques,R. O. C. Conference on Electrical Power Engineering, PartC-3,pp. 749-753,2003.[12] RJ Wai, RY Duan, and LW Liu, UA current-source sine wave voltage driving circuit via voltage-clamping and soft-switching techniques, ROC Conference on Electrical Power Engineering, PartC-3, pp. 749-753, 2003 .

[13] R. J. Wai and R. Y. Duan, “High-efficiency DC/DC converter with high voltage gain，” IEE Proc. Electric Power Applications, vol. 152, no. 4, pp. 793-802, July 2005. 【發明内容】 如圖2所示為本發明所揭示之高效率諧振式電流源反 流器’其中包含一略高於交流峰值之直流電源1〇1 ; —箝 制電路102 :由兩個箝制開關η、巧、兩個箝制二極體Di、 A ’ 一剎車二極體/¾及一箝制電容&所組成；一耦合電路 i〇3，由由_合雷咸 -整流二極體sc與:次側繞〜以及 全桥門關+ 咕振反流器電路104 :由四個 組-驅動電路';Γ;ί出⑽^ 驅動訊號。 負貝回授控制與輸出六個開關之 經箝半週時，電流由直流電源⑻ 合電感Z；之K /」、Γ2,及輕合電路103之轉 之全橋開關輸由譜振反流器電路104 為正弦波負半週輸出===電。同理，欲產生電壓 譜振反流器電路104:入推^電路102之箱制開關^與 輸出電容Q反向充電同時觸發導通，對 高頻切換電流源電路路 古泣帝.κ, 耦D電路103之一次側繞組々介於 01之電壓心、箝制電路102之箝制電容c之電 =以及譜振反流器電路1〇4之輸出電容三。 之Γ須以電流源作媒介，以“輸出 -=繞叫之漏感特性，達成全部功率半導財性 =與電㈣制之效果；譜振反流器電路1(^四個= 開關η負責導引正半週期之電流源電流路1導 之王橋關7Γ及7Γ則為負半週之路徑；魅反流器電路 W振電感4與輸出電容Q形成—組二階並聯 用電感電流落後電叙特性，由魏電感U輸出電容^ 1321392 電壓t極性換向。 ，發明之「高效⑽振式電絲反流器」之各點波形 時序。電路工作模式，分別如圖3與圖4所示，以下將以 ^兩^容逐段說日紅作原理，為簡化電路分析，所有 開關几件及二極體導通壓降忽略不記。另外 :=)解:略專有名詞不至於冗長，電路歸屬圖號“口… =路101),略之，直接對照說明所屬圖式即可明瞭： 模式一：時間（ί〇1)箝制開關7]、r2導通一段時間’、 式開關7導通後—段時⑽㈣電 耦合電感卜次側繞組4電知，從直 -電U ’貫穿箝制開關认巧、符制電容c。以及搞 電感7之一次側繞組〜’流入譜振反流器電路之 截止狀態，令極::二=二 可表示為 ^為正時，則一次側繞組心之跨壓νω[13] RJ Wai and RY Duan, “High-efficiency DC/DC converter with high voltage gain,” IEE Proc. Electric Power Applications, vol. 152, no. 4, pp. 793-802, July 2005. As shown in FIG. 2, the high-efficiency resonant current source inverter disclosed in the present invention includes a DC power supply 1〇1 which is slightly higher than the AC peak; the clamp circuit 102: is composed of two clamp switches η, Qiao , two clamping diodes Di, A 'one brake diode / 3⁄4 and a clamp capacitor & a coupling circuit i〇3, consisting of _He Lei Xian-rectifier diode sc and: secondary side Winding ~ and full bridge door closing + 反 inverting circuit 104: by four groups - drive circuit '; Γ; ί (10) ^ drive signal. When the negative feedback control and output of the six switches are half-cycled, the current is supplied by the DC power supply (8) and the inductance Z; the K / ", Γ 2, and the full-bridge switching of the light-synchronous circuit 103 are transmitted by the spectrum. The circuit 104 is sinusoidal negative half cycle output === electricity. Similarly, the voltage spectrum inverting circuit 104 is to be generated: the box switch of the input circuit 102 and the output capacitor Q are reversely charged and simultaneously triggered to conduct, and the high frequency switching current source circuit is the ancient crying. κ, coupling The primary winding 々 of the D circuit 103 is between the voltage core of 01, the voltage of the clamp capacitor c of the clamp circuit 102, and the output capacitance of the spectral inverter circuit 1〇4. It is necessary to use the current source as the medium to achieve the effect of the output-= around the leakage inductance, to achieve the full power semi-conductivity = electricity (four) system; spectrum oscillator circuit 1 (^ four = switch η Responsible for guiding the positive half cycle of the current source current path 1 guide Wangqiaoguan 7Γ and 7Γ is the negative half cycle path; the charm inverter circuit W vibration inductance 4 and the output capacitor Q form - group second-order parallel inductor current behind Electrical characteristics, by Wei inductor U output capacitor ^ 1321392 voltage t polarity commutation. Invented "high efficiency (10) vibrating wire reactor" waveform timing. Circuit operation mode, as shown in Figure 3 and Figure 4, respectively In the following, the following two principles will be used to describe the process of red, and in order to simplify the circuit analysis, the voltage drop of all the switches and diodes is ignored. In addition: =) Solution: The slightly proper noun is not too long. The circuit belongs to the figure number “口...=路101). If it is omitted, it can be clearly defined by the following description: Mode 1: Time (〇1) Clamp switch 7], r2 is turned on for a while, and switch 7 is turned on. After the period - (10) (four) electrically coupled inductors, the secondary winding 4 is known, from the straight-electric U's through the clamp Close the coincidence, the capacitor c. And the inductive state of the primary side of the inductor 7 ~ 'into the spectral inverting circuit of the off state, so that the pole:: two = two can be expressed as ^ is the timing, then the primary winding core Trans-voltage νω

Ld · did !dt = vLd = vm + vco - v . ° (1) it為輸入ΐ直流電源電壓、為輸出交流電壓，等號 電流谷又:，-次側繞組、 因此電流姻分不至於太高，可以 :側，並有效降低籍制開關之導通責任週期= 失。另外’箝制開關叫之跨壓分別為導叫 VTl - Vco - VD2 (2.a) 16 ；T2 = Vc〇 ~ VD, ；T2 = Vc〇 ~ VD, (2.b) 示據上式可传到推制二極體A及&之跨壓〜與％如下所 (3.a) (3.b) (4) VD\ = ν〇〇 - VT2 VD2 = vc〇 - Vr, 輸出交流電壓V。可計算為 V〇 - it^d ~ Ll ~ i。) · dt 之為交流負載電流，此模式止 中，為譜振電感々電流 於箝制電容C。電壓'。放電至零：特 模式^ :時間“〜〜)箝制二極體ΑA導㉟ ’瞀容C°放電至接近零伏特時，依據式(3.a)及式(3.b、 =二極體端跨壓由逆偏降至零伏特，再 關順形成兩箝制二極體零電壓切換導通。此時籍制 開關7]及Γ2觸發信號仍j道_ 體後再相互並聯狀態，平均八二：二:J f關串聯二極 节. 十巧刀擔流向輸出電容Q之充電電 核式二.日夺間〇2〜’3)箝制開關；及^觸發信號截止 =箝制開，及明發信號截止後，全橋開關（及&—仍 寺、通，耦合電感一次側繞組、受限漏感續流因素，其 徑轉流料、C。及…對箝制電容&與輸出電容 充電。由於’因此當箝制電容c。電壓傷上升時： 只有微幅上升。依據式(2.a)及式(2 b)所示，箝制開 電壓分別等於箝制二極體Α導通電壓加上電容 〜當電壓一漸上昇，表示籍制開關截止 17 備零電壓切換特性，同時箝制電心充 感的能量’並於下-週期之模式 fM. Ll 傳运、.、。輸出電容Q，箝制電容c。電^可表式為 (5)Ld · did !dt = vLd = vm + vco - v . ° (1) it is the input ΐ DC power supply voltage, the output AC voltage, the equal-number current valley again:, - the secondary winding, so the current marriage is not too High, can: side, and effectively reduce the conduction cycle of the system switch = loss. In addition, the 'clamp switch' is called VTl - Vco - VD2 (2.a) 16; T2 = Vc〇~ VD, ; T2 = Vc〇~ VD, (2.b) To push the diode A and & the cross-pressure ~ and % as follows (3.a) (3.b) (4) VD\ = ν〇〇- VT2 VD2 = vc〇- Vr, output AC voltage V . It can be calculated as V〇 - it^d ~ Ll ~ i. ● dt is the AC load current. In this mode, it is the spectral inductance 々 current and clamp capacitor C. Voltage'. Discharge to zero: special mode ^: time "~~" clamp diode ΑA lead 35 '瞀 C ° discharge to near zero volts, according to equation (3.a) and formula (3.b, = diode The end-span voltage is reduced from zero to volts by reverse bias, and then the two voltages of the two clamps are switched to form a turn-on voltage. At this time, the trigger signals of the switch 7] and Γ2 are still j-channel and then parallel with each other, the average is 82 : 2: J f off series diodes. 10 Qiao knife to the output capacitor Q charging electric core type 2. Day 〇 〇 2 ~ '3) clamp switch; and ^ trigger signal cutoff = clamp open, and Mingfa After the signal is cut off, the full bridge switch (and & still temple, pass, coupled inductor primary winding, limited leakage inductance freewheeling factor, its diameter converter material, C. and ... charge the clamp capacitor & output capacitor Because 'so when clamping capacitor c. When the voltage injury rises: only a slight rise. According to equations (2.a) and (2 b), the clamp-on voltage is equal to the clamp diode turn-on voltage plus capacitor ~ When the voltage rises gradually, it means that the switch is turned off and the voltage is switched, and the energy of the charge is clamped. Cycle mode fM. Ll transport, .,. Output capacitor Q, clamp capacitor c. Electric ^ can be expressed as (5)

Jd-dt t2<t<t4 此時一次側繞組心之跨壓νω可表示為 vLd=Vm-vco-v〇 _ (6) 依據上式所示，輕合電感—次側繞組&在極性點處為正· 壓’感應至二次側繞組電以亦相同情形，因輪合電ς ，整流二極體巧仍為逆偏，二次側繞組~無電流路徑。 模式四：時間（?3〜幻耦合電感二次侧電流開始導通 本〃模式始於箝制電容Ce電壓％。時，依據方程式 (6)計算，耦合電感一次側繞組心之電壓極性開始反相，在 非極性點處為正電壓，感應至二次側繞組電感~亦相同情 形，因此耦合電路之整流二極體巧為順偏狀態。依據磁通 不滅定律，二次側繞組L/開始產生電流，在此期間，二次 側繞組^電流，跟隨一次側繞組心一同流入諧振反流器 電路，其關係式可表示為 L(^d(max) - Ldid + / Π\ 其中’ ~(胃> 電流G之最高值’兩者電流^及匕一長一消，本 模式止於電流&降為零，而電流。達到最高點。 由於箝制電容C。可以充分吸收漏感能量，以及磁通得由二 次側繞組•釋放’漏感對糸統影響不高。為簡化分析，暫 不考慮漏感能量[13]。令輕合電感一、二次繞組匝數分別 1321392 為與，則匝數比π可表示為 %仏 ⑻ 在此模式期間，二次側繞組電壓-V£/等於V。， 繞組Ld電壓vu為 可反推一次側 v (9) 依據克希荷夫電壓定律，本模式之方程式可以表示如下 VIN ~Vco+ — ~Vo=() η (10) 整理上式可得 Vco =1^ΪΝ +( l)Vo η (11) 由於V。為交流電壓，加絕對值以簡化分析 ^co=VIN+{Xln-l)\Vo\ (12) 上式等號之第二項為負數，由於箝制電容電壓Ve£)跨在箝制 開關兩端，其電壓最高值發生在輸出交流電壓\為零時， • 可以得到開關耐壓規格為Jd-dt t2<t<t4 At this time, the cross-voltage νω of the primary winding core can be expressed as vLd=Vm-vco-v〇_ (6) According to the above formula, the light-inductive inductance-secondary winding & At the point, the positive and negative voltages are induced to the secondary winding. In the same situation, the rectifying diode is still reversed due to the rotation of the electric motor, and the secondary winding has no current path. Mode 4: Time (?3~ phantom-coupled inductor secondary current begins to conduct. The local mode starts from the clamp capacitor Ce voltage %. When calculating according to equation (6), the voltage polarity of the primary winding core of the coupled inductor begins to reverse. The positive voltage at the non-polar point and the inductance to the secondary winding are also the same, so the rectifying diode of the coupling circuit is in a forward-biased state. According to the law of flux inactivation, the secondary winding L/ begins to generate current. During this period, the secondary side winding ^ current flows along with the primary side winding core into the resonant inverter circuit, and its relationship can be expressed as L(^d(max) - Ldid + / Π\ where '~(stomach &gt The highest value of the current G 'both currents ^ and 匕 one long, the current mode stops at the current & fall to zero, and the current reaches the highest point. Due to the clamping capacitor C, the leakage energy can be fully absorbed, and the magnetic Passing the secondary winding and releasing 'leakage inductance' has little effect on the system. To simplify the analysis, the leakage inductance energy is not considered for the time being. [13] Let the light-combined inductance first and second winding turns are respectively 1321392 and then The turns ratio π can be expressed as %仏(8) During the mode, the secondary winding voltage -V£/ is equal to V. The winding Ld voltage vu is reversible on the primary side v (9). According to the Khrhew voltage law, the equation of this mode can be expressed as follows VIN ~Vco+ — ~ Vo=() η (10) Finishing the above formula gives Vco = 1^ΪΝ +( l)V η (11) Since V is the AC voltage, add the absolute value to simplify the analysis ^co=VIN+{Xln-l)\ Vo\ (12) The second term of the upper equal sign is a negative number. Since the clamp capacitor voltage Ve() is across the clamp switch, the highest voltage value occurs when the output AC voltage is zero. • The switch withstand voltage can be obtained. Specifications are

Vn(max) = V7'2(max) = ^1N (13) 因此本架構之箝制開關耐壓與輸入電壓相同，同理全橋四 個開關7；+、77、7；+及7；-，因輸出端為電容器且有飛輪二極 體之電壓箝制效能，開關耐壓與輸出交流電壓V。相同。 模式五：時間〇4〜〇電流9由最高點開始下降 當一次側繞組k電流G降為零時，耦合電感能量全部透 過二次側繞組電流&傳送至輸出電容仏，因此該電流由最 19 1321392 - 高點開始下降。本模式止於箝制開關7]、Γ2觸發信號導通時。 . 模式六：時間〇5〜?。）箝制開關7；、Γ2觸發信號導通 此時箝制開關7；、Γ2觸發信號導通，若二次側繞組電流 b在本模式開始前已降到零，則為電流不連續模式 (Discontinue Current Mode, DCM )，因此自然形成導通具 有ZCS現象。同理，若為電流連續模式（Continue Current Mode，CCM )，受限一次側繞組&漏感及兩側繞組能量傳遞 影響，如同模式四中相同的電氣特性，因此仍然形成導通 φ 具有ZCS特性。 此外，在諧振反流器部分，諧振電感A的電流Q落後 輸出交流電壓約90度電氣角，因此當輸出交流電壓V。在 零伏特附近時，電流L為最高值，可以抽出輸出電容q的 能量以完成零交越換向目的，此過程不需經過開關與控 制，因此可以省略一般電流源常用之串聯二極體，同時為 避免空載時所發生二階諧振之高增益電壓，諧振頻率應避 開60Hz。為確保換向成功，以避免省略串聯二極體所造成 • 飛輪路徑之短路電流，設計時諧振電感電流能量必須高於 輸出電容位能，可表示為 ^l}LL(mdix.) ^ ^o(max) ( 1 4) 一般電流源架構之全橋開關所串聯二極體阻斷負載反 饋至直流電源之路徑，必須單獨作一交流電阻式剎車迴 路。當輸出電壓高於直流電源電壓時，本發明之剎車二極 _ 體1)3自然導通並提供之能量反饋路徑，此情形發生於負載 * 為交流電動機之飛輪慣量，以及電感性負載改變諧振頻率 20 1321392 ‘所邊成巧電壓；㈣車二極體將纟能量牽引至直流電源電壓 • 吸收’除可箱制所有開關電壓，再生式剎車功能有效利用 源裘負栽之能量，提高能源利用率。 由上迷說明可知，多數開關二極體及開關導通與截止 時’㈣具有ZCs與zvs特性’因此在理論分析上，本發 明所述=路可以獲得高轉換效率。 雖、、丨本發明已前述較佳實施例揭示，然其並非用以限 定本發明’任何熟習此技藝者，再不脫離本發明之精神和 參範園内，當可作各種之變動與修改，因此本發明之保護範 圍當視後附之申請專利範圍所界定者為準。 【實施方式】 本發明主要元件之所有功率半導體開關選用 MOSFET，編號為讯^^心導通電阻/^㈣^^^’耐壓 250V以及額定電流44A，包裝形式為TO-247。依據方程 式（13)可以得到開關最高对壓規格，本發明實施例目的在 鲁於控制輸出電壓之峰值為156V，換算成有效值為ll〇v， 設定額定輸出規格為AC 110V60Hz 500W之電源規格。而 本發明其它參數設計及元件選用提供如下： 直流輸入電壓：170V 交流輸出電壓：AC ll〇V 60Hz 切換頻率：50kHz 耦合電感7；之一次侧繞組心及二次側繞組& : 100 W及 200你，一、二次繞組匝數％與％ : 8.5及12匝，core-EE55， 21 1321392 - 柄合係數k = 0.98 . 箝制開關 7]、：Γ2及全橋開關 7；+、Γ;、7；+、77 : POWER MOSFET IRFP264，F^=250F，/?頌撕）=60mQ，/D=44d，TO-247 輸出電容Q及諧振電感4 : 6.8W及1.034//Vn(max) = V7'2(max) = ^1N (13) Therefore, the clamp switch of this architecture has the same withstand voltage as the input voltage, and the same bridge has four switches 7; +, 77, 7; + and 7; Because the output is a capacitor and has the voltage clamping performance of the flywheel diode, the switch withstand voltage and the output AC voltage V. the same. Mode 5: Time 〇4~〇current 9 starts to fall from the highest point. When the primary side winding k current G drops to zero, the coupled inductor energy is transmitted through the secondary side winding current & to the output capacitor 仏, so the current is the most 19 1321392 - The high point begins to fall. This mode ends when the trigger switch 7] and Γ2 trigger signal are turned on. Mode 6: Time 〇 5~?. The clamp switch 7;, Γ 2 trigger signal is turned on at this time, the clamp switch 7; Γ 2 trigger signal is turned on, if the secondary side winding current b has dropped to zero before the start of the mode, then the current discontinuous mode (Discontinue Current Mode, DCM), so naturally forming conduction has a ZCS phenomenon. Similarly, if the current continuous mode (CCM), the limited primary winding & leakage inductance and the energy transfer of the two sides of the winding, like the same electrical characteristics in mode four, so still form conduction φ with ZCS characteristics . Further, in the resonant inverter portion, the current Q of the resonant inductor A is delayed by an electrical angle of about 90 degrees from the output AC voltage, so when the AC voltage V is output. When the voltage L is near the highest value, the current L is the highest value, and the energy of the output capacitor q can be extracted to complete the zero-crossing commutation. This process does not need to be switched and controlled, so the series diode commonly used in general current sources can be omitted. At the same time, in order to avoid the high gain voltage of the second-order resonance occurring at no load, the resonant frequency should avoid 60Hz. In order to ensure the success of the commutation, to avoid omitting the short circuit current of the flywheel path caused by the series diode, the resonant inductor current energy must be higher than the output capacitor potential energy, which can be expressed as ^l}LL(mdix.) ^ ^o (max) (1 4) The path of the series diode of the full-bridge switch of the general current source architecture to block the load feedback to the DC power supply must be used as an AC resistance brake circuit. When the output voltage is higher than the DC power supply voltage, the brake diode _ body 1) 3 of the present invention naturally turns on and provides an energy feedback path, which occurs when the load * is the flywheel inertia of the AC motor, and the inductive load changes the resonant frequency. 20 1321392 'The smart voltage is on the side; (4) The car diode draws the energy of the 至 to the DC power supply voltage. • Absorbs all the switching voltages except the box. The regenerative brake function effectively utilizes the energy of the source and energy to improve energy efficiency. . As can be seen from the above description, most of the switching diodes and switches have the ZCs and zvs characteristics when turned on and off (4). Therefore, in theoretical analysis, the present invention can achieve high conversion efficiency. Although the present invention has been disclosed in the foregoing preferred embodiments, it is not intended to limit the invention to those skilled in the art, and various changes and modifications may be made without departing from the spirit and scope of the invention. The scope of the invention is defined by the scope of the appended claims. [Embodiment] All the power semiconductor switches of the main components of the present invention are selected from MOSFETs, and the number is the signal conduction resistance / ^ (four) ^ ^ ^ ' withstand voltage 250V and rated current 44A, and the package form is TO-247. According to the equation (13), the highest switching voltage specification can be obtained. The purpose of the embodiment of the present invention is to control the output voltage to a peak value of 156V, convert it into an effective value of ll〇v, and set a rated output specification of AC 110V60Hz 500W. The other parameter design and component selection of the present invention are as follows: DC input voltage: 170V AC output voltage: AC ll〇V 60Hz Switching frequency: 50kHz Coupling inductor 7; primary winding core and secondary winding & : 100 W and 200 you, primary and secondary winding turns % and %: 8.5 and 12 匝, core-EE55, 21 1321392 - shank combination coefficient k = 0.98. Clamp switch 7],: Γ 2 and full bridge switch 7; +, Γ; , 7; +, 77 : POWER MOSFET IRFP264, F ^ = 250F, /? 颂 tear) = 60mQ, / D = 44d, TO-247 output capacitor Q and resonant inductor 4: 6.8W and 1.034 / /

箝制電容Q : 0.04wF 箝制二極體 A、/)2及巧：SFA1604G 及 SFA1608G 反流器諧振頻率：60Hz 圖5表示本發明所揭示之高效率諧振式電流源反流器 φ 各元件之60Hz實測波形圖，圖5(a)中為無載時輸出電壓 及耦合電感7；—次側繞組心電流&，其中輸出電壓係由頻 率50kHz高頻切換依PI比例積分方式累積而成，其微量高 頻成分之漣波電壓已改善，且電流G很小。輸出電壓V。為零 值附近之區域；因換向能量完全由諧振電感A提供，電流& 幾乎為零，可證明在無載時本發明之高效率諧振式電流源 反流器其能量損失極低。圖5(b)為輸出電壓及諧振電感 電流L之波形，諧振電感4電感值為1.034//，電流Q峰值 • 為0.39A，得知其容量為30.5VAR，雖然電感值很高，但實 際的體積相當小。圖5(c)中為輸出功率400W之輸出電容C, 端電壓V。及一次側繞組Zy電流。為簡化分析一次側繞組電 流Q於穩態時之平均值，箝制電容電壓％暫不考慮，依據 克希荷夫電壓及電流定律Clamping Capacitor Q: 0.04wF Clamping Diode A, /) 2 and QC: SFA1604G and SFA1608G Inverter Resonant Frequency: 60Hz Figure 5 shows the high-efficiency resonant current source inverter φ disclosed by the present invention. The measured waveform diagram, Figure 5 (a) is the output voltage and coupled inductor 7 when no load; the secondary side winding current &, the output voltage is accumulated by the frequency 50kHz high frequency switching according to the PI proportional integral method, The chopping voltage of the trace high frequency component has been improved, and the current G is small. Output voltage V. The area near the zero value; since the commutation energy is completely provided by the resonant inductor A, the current & is almost zero, which proves that the high efficiency resonant current source of the present invention has an extremely low energy loss when no load is present. Figure 5(b) shows the waveform of the output voltage and the resonant inductor current L. The resonant inductor 4 has an inductance of 1.034//, and the current Q peak value is 0.39A. The capacitance is 30.5VAR. Although the inductance is high, the actual value is actually The volume is quite small. In Figure 5(c), the output capacitor C with an output power of 400W and the terminal voltage V. And the primary side winding Zy current. In order to simplify the analysis of the average value of the primary winding current Q at steady state, the clamp capacitor voltage % is not considered at this time, according to the Khreshofer voltage and current law.

Vo=VLL=LL-diLL/dt = VCL (15) " 以及 U ~ ^CL + hL + Zo (16) 22 1321392 輸出電壓νβ為AC 110V 60Hz，因此v0可表示為156sin377i’ 流經負載/?(假設為3〇Ω)、輸出電容Q及諧振電感4之電流 tb及L可分別表示為；Vo=VLL=LL-diLL/dt = VCL (15) " and U ~ ^CL + hL + Zo (16) 22 1321392 The output voltage νβ is AC 110V 60Hz, so v0 can be expressed as 156sin377i' flowing through the load /? (assumed to be 3 Ω), the output capacitor Q and the resonant inductor 4 currents tb and L can be expressed as;

156sin377r R156sin377r R

5.2sin377/A (17)5.2sin377/A (17)

156sin377i XCL156sin377i XCL

=Q · 156 cos377i = 0.39 sin (377/ + 90°) A z.cx ==Q · 156 cos377i = 0.39 sin (377/ + 90°) A z.cx =

156sin377i156sin377i

XX

= 0.39 sin (377卜 90。）A (18) (19)= 0.39 sin (377 卜 90.) A (18) (19)

LL φ 故輸出電壓為零值附近之區域，流經負載之電流為零， 電流G及Q大小相同但相位差180度，因此輸出電容(^之 電壓換向能量可完全由諧振電感A提供，所以不需由輸入 . 電源提供換向所需電流路徑，耦合電感一次側繞組A/之 電流G幾乎為零，僅需供應諧振過程的能量損失。換向完 成後之，箝制開關7；、Γ2觸發信號導通期間，依據方程式（18) 所示，輸出電容Q之充電電流正比cos函數，是故電流心最 高；此外，輸出電壓v。峰值則僅需供應負載Λ之電流，此區 • 間之電流G與負載輕重有關。圖5(d)中為全橋開關7；+之電壓 να+及電流ί，由於全橋開關採60Hz低頻切換，全橋開關之 電流G導通與截止時，該開關兩端呈現零電壓狀態，完全 無切換損失，且全橋開關輸出端為電容器，加上飛輪二極 體所提供之迴路，使得開關耐壓箝制與輸出交流電壓V。相 同。圖5(e)為箝制開關7；之電壓νη與電流ζη，圖中顯示箝制 開關電壓％在截止時具有電壓箝制特性，由於箝制電容電 ' 壓、跨接於箝制開關兩端，其電壓最高值發生在輸出交流 23 1321392 - 電壓〜為零時，由波形顯示開關所需承受之電壓與推導方 ， 程式（13)相符。綜合圖5電路實作波形所示，諧振反流器 電路已有效控制輸出電壓V。之零交越波形，且在該交越區 域附近，所有開關幾乎沒有電流通過。本發明之高效率諧 振式電流源反流器所有功率半導體開關耐壓皆箝制與輸入 電壓相同，所承受電壓約等於或小於電源電壓，電壓箝制 效能優於習用反流器[11]。 圖6表示本發明所揭示之高效率諧振式電流源反流器 φ 各元件高頻操作之實測波形圖，其中圖6(a)所示為操作於 連續模式之耦合電感7； —、二次側繞組電流6及9交越波 形，受限兩側繞組漏感影響，當箝制電容電壓 時，依據方程式（6)計算，耦合電感7；—次側繞組~之電壓 極性開始反相，在非極性點處為正電壓，感應至二次側繞 組&亦相同情形，因此耦合電路之整流二極體D/為順偏狀 態。依據磁通不滅定律，二次側繞組&開始產生電流，在 此期間，二次侧繞組&電流&，跟隨一次側繞組4電流心一 • 同流入諧振反流器電路，兩繞組電流關係式係依據方程式 (7)產生變化。其交越變化情形，如圖6(b)為一次側繞組心電 流上升與二次側繞組&電流Z/下降，以及圖6(c)為一次側 繞組心電流^下降與二次側繞組&電流^上升。圖6(d)為整 流二極體巧之電壓vD/、電流&波形，由實驗波形顯示，整 流二極體义電壓V#波形於導通及截止時皆有柔性切換效 '果，且同時存在零電壓及零電流切換；此外整流二極體電 • 壓、因高壓震盪所產生之突波電壓經加入緩震電路 24 1321392 (snubber)與數值模擬~660伏特相比已降為560伏特 波電壓經緩振電路改善後可降低二極體耐壓規格。圖6犬 為箝制開關5之電壓vn及電流Z n波形，箝制開關5於^通(e > 截止時皆有柔性切換效果’加上可使用耐壓規格較低 率半導體開關’可以提高本發明之轉換效率。 _ 圖7表示本發明所揭示之高效率譜振式電流源反法器 於無载及各種負載之電壓％、電流^實測波形圖’並分二^ 傅立葉頻率分析以計算總諧波失真率（THD)。圖7(a)為無載 時’其中基頻（6〇Hz)之值為40.6dB㉟乎與交流電壓有钱 相同，總言皆波失真率(而)為L〇%。圖7(b)負載為非線性 整流性負載（h_，C = 47〇MF) ’其中基頻（6〇叫之值為 40.3犯’總譜波失真率（丁_為L8%。圖7(c)負 咸 性負載（^50ω，ζ = 5〇μ〇，實測總諧波失真率（THD)古^ 13.92%，高於國際標準值5%。圖7(d)則為圖％ = 載波形改善後之輸出電壓圖形。若將輪出電容&兩端並聯 ⑽電：：以充分供應電感性負載所 總譜波失降為議，解決供應電感性負：： 生波形失真問=® 7(e)為反流器瞬間加載3此電版性負 載之電壓ν’Κ波形圖，依據實驗所示於瞬 ： 輸出電壓〜峰值部份僅些許失真。综合圖7之各波肝出 本發明所揭不之錢率難式電流源反流器，可接^ 電感性、電容性、非線性及瞬間變化之負* ,= 波形_波失真率及傅立葉頻譜分析均優於上述之參^ 獻0 25 1321392 - 圖8表示本發明所揭示之高效率諧振式電流源反流器 . 與參考文獻[11]之轉換效率圖，該圖顯示本發明之高效率諧 振式電流源反流器轉換效率高於97%，在輕載及重載時均 優於參考文獻[11 ]，尤其在輕載時，利用I皆振換向技術以及 二次側繞組丄/電流全部導入輸出端，大幅減少開關電流之 環流成分，因此最高可提昇轉換效率約達8%。 本發明經電路實作，實現高效能之電流正弦電壓轉換 電路，綜合特點如下： φ 1.利用耦合電感一、二次側能量傳遞方式控制電流源，使 得全部半導體開關及二極體均有柔性切換特性，最高轉 換效率大於97%，且經各種負載測試後，本架構之反流 器總諧波失真率（THD)皆低於2%以下； 2. 運用電壓箝制技術，可降低功率半導體開關元件之耐壓 規格，所有開關所承受電壓約等於或小於電源電壓； 3. 耦合電感容量與體積小於一般電流源架構，其能量全部 傳送至輸出端，無環流問題； • 4.耦合電感操作在連續電流模式下，可以提高切換頻率以 降低感值，同時開關仍具有柔性切換效果； 茲將電路實作元件柔性切換特性彙整表1所示。 26 1321392LL φ, so the output voltage is near the zero value, the current flowing through the load is zero, the current G and Q are the same but the phase difference is 180 degrees, so the voltage commutation energy of the output capacitor (^ can be completely provided by the resonant inductor A, Therefore, it is not necessary to provide the current path required for commutation by the input. The current G of the primary winding A/ of the coupled inductor is almost zero, and only the energy loss of the resonance process needs to be supplied. After the commutation is completed, the switch 7 is clamped; During the turn-on of the trigger signal, according to equation (18), the charging current of the output capacitor Q is proportional to the cos function, so the current core is the highest; in addition, the output voltage v. The peak value only needs to supply the current of the load ,, The current G is related to the light weight of the load. In Figure 5(d), the full bridge switch 7; the voltage να+ and the current ί of the +, due to the 60Hz low frequency switching of the full bridge switch, the current G of the full bridge switch is turned on and off, the switch The two terminals exhibit zero voltage state, no switching loss at all, and the full bridge switch output is a capacitor, and the circuit provided by the flywheel diode makes the switch withstand voltage clamped the same as the output AC voltage V. Figure 5 (e ) is the voltage of the clamp switch 7; the voltage νη and the current ζη, the figure shows that the clamp switch voltage % has a voltage clamping characteristic when it is cut off. Since the clamped capacitor is electrically connected and connected across the clamp switch, the highest voltage value occurs at the output. AC 23 1321392 - When the voltage is zero, the voltage required by the waveform display switch is consistent with the derivation, program (13). As shown in the actual waveform of Figure 5, the resonant inverter circuit has effectively controlled the output voltage V. The zero-crossing waveform, and almost no current flows through all the switches in the vicinity of the crossover region. The high-efficiency resonant current source inverter of the present invention is clamped to the same voltage as the input voltage of all power semiconductor switches. The voltage clamping performance is equal to or lower than the power supply voltage, and the voltage clamping performance is better than that of the conventional inverter [11]. Fig. 6 is a diagram showing the measured waveform of the high-frequency operation of each component of the high-efficiency resonant current source inverter φ disclosed by the present invention, wherein 6(a) shows the coupled inductor 7 operating in continuous mode; —, the secondary side winding currents 6 and 9 crossover waveforms, limited leakage inductance on both sides of the winding, when clamped When the voltage is calculated, according to equation (6), the coupled inductor 7; the voltage polarity of the secondary winding ~ begins to be inverted, and the positive voltage is at the non-polar point, and the secondary winding & is also the same, so the coupling circuit The rectifying diode D/ is in a forward bias state. According to the law of magnetic flux immortality, the secondary winding & begins to generate current, during which the secondary winding & current & follows the primary winding 4 current core • With the inflowing resonant inverter circuit, the relationship between the two winding currents changes according to equation (7). The crossover changes, as shown in Figure 6(b), the primary winding current rises and the secondary winding & current Z/descent, and Fig. 6(c) shows the primary side winding current drop and the secondary side winding & current ^ rise. Figure 6(d) shows the voltage vD/, current & waveform of the rectified diode. The waveform of the rectifier diode V# waveform has a flexible switching effect when it is turned on and off. There is zero voltage and zero current switching; in addition, the voltage of the rectified diode and the surge voltage generated by the high voltage oscillation are reduced to 560 volts by adding the cushioning circuit 24 1321392 (snubber) compared with the numerical simulation of ~660 volts. The voltage can be reduced by the vibration damping circuit to reduce the diode withstand voltage specifications. Figure 6 shows the voltage vn and current Z n waveform of the clamp switch 5, and the clamp switch 5 is connected to the ^ (E > flexible switching effect at the end of the cut-off plus a lower-rate semiconductor switch with a withstand voltage specification can improve the present Conversion efficiency of the invention. _ Figure 7 shows the high-efficiency spectral-type current source counter-cursor disclosed in the present invention. The voltage and current waveforms of the unloaded and various loads are measured and divided into two ^ Fourier frequency analysis to calculate the total Harmonic distortion rate (THD). Figure 7 (a) is no load when 'the fundamental frequency (6 〇 Hz) value is 40.6dB35 is the same as the AC voltage, the total wave distortion rate (and) is L 〇%. Figure 7(b) The load is a nonlinear rectifying load (h_, C = 47〇MF) 'where the fundamental frequency (6 〇 is 40.3 guilty 'total spectral distortion rate (D = 8%). Figure 7 (c) negative salt load (^50ω, ζ = 5〇μ〇, measured total harmonic distortion rate (THD) is ancient ^ 13.92%, higher than the international standard value of 5%. Figure 7 (d) is the figure % = The output voltage pattern after the carrier shape is improved. If the round-out capacitor & both ends are connected in parallel (10), the total spectral wave loss is fully supplied to the inductive load. Inductive Negative:: Raw Waveform Distortion == 7(e) is the inverter's instantaneous load 3 The voltage of this electro-plate load ν'Κ waveform diagram, according to the experiment shown in the instant: Output voltage ~ peak part only a little distortion According to the various wave of the liver of Fig. 7, the difficult current source current reflux device disclosed by the invention can be connected with negative inductance of inductive, capacitive, nonlinear and transient changes, = waveform_wave distortion rate and Fourier spectrum analysis is superior to the above-mentioned reference 0 25 1321392 - Figure 8 shows the high efficiency resonant current source inverter disclosed in the present invention. and the conversion efficiency diagram of reference [11], which shows the present invention. The high-efficiency resonant current source inverter has a conversion efficiency higher than 97%, which is superior to the reference [11] at light load and heavy load, especially at light load, using I-commutation commutation technology and secondary winding The 丄/current is all introduced into the output terminal, which greatly reduces the circulating current component of the switching current, so the conversion efficiency can be increased up to about 8%. The present invention implements a high-performance current sinusoidal voltage conversion circuit through a circuit, and the comprehensive features are as follows: φ 1 Using coupled inductors, secondary side energy The transfer mode controls the current source, so that all semiconductor switches and diodes have flexible switching characteristics, the maximum conversion efficiency is greater than 97%, and the total harmonic distortion (THD) of the inverter of the architecture is low after various load tests. 2% or less; 2. Using voltage clamping technology, the voltage specification of power semiconductor switching components can be reduced, and all switches are subjected to voltages equal to or less than the power supply voltage; 3. The coupled inductor capacity and volume are smaller than the general current source architecture, and its energy All transmitted to the output, no circulation problem; • 4. Coupled inductor operation in continuous current mode, can increase the switching frequency to reduce the sense of value, while the switch still has a flexible switching effect; the circuit is implemented as a component flexible switching characteristics summary table 1 is shown. 26 1321392

表1主要元件柔性切換特性 零電壓切換(zvs) 零電流切換(zcs) 元件符號 導通 截止 導通 截止 Ά、τ2 〇 〇 〇 Τ:、Ta-、Tb+、Tb- 〇 〇 〇 〇 D'、D2、D3 〇 〇 〇 〇 Df 〇 〇 〇 〇 27Table 1 Flexible switching characteristics of main components Zero voltage switching (zvs) Zero current switching (zcs) Component symbol turn-on turn-off turn-off Ά, τ2 〇〇〇Τ:, Ta-, Tb+, Tb- 〇〇〇〇D', D2 D3 〇〇〇〇Df 〇〇〇〇27

【圖式簡單說明】 表示習用正弦電壓反流器架構圖。 表不本發明所揭示之高效率譜振式電流源反流器路 方塊圖。 表不本發明所揭示之高效率諧振式電流源反流器各 點波形時序圖。 表不本發明所揭示之高效率譜振式電流源反流器工 作模式圖。 表不本發明所揭示之高效率諧振式電流源反流器各 元件之60Hz實測波形圖。 表不本發明所揭示之高效率諧振式電流源反流器各 元件高頻操作之實測波形圖。 表示本發明所揭示之尚效率譜振式電流源反流器於 無載及各種負載之實測波形圖。 表不本發明所揭示之高效率諧振式電流源反流器與 參考文獻[11]之轉換效率圖。 【主要元件符號說明】 1〇ι :直流電源 1〇2 :箝制電路 103 :耦合電路 104 :譜振反流器電路 105 :控制及驅動電路 7]:箝制電路之功率半導體開關（簡稱箝制開關) 28 1321392 、 r2 :箝制電路之功率半導體開關（簡稱箝制開關） . 7；+、77、7；+及7；-:諧振反流器之功率半導體開關（簡稱全橋 開關） 7；:具高激磁電流之變壓器（簡稱耦合電感） A：:耦合電感7；之耦合係數（簡稱耦合係數） b :耦合電感C之一次側繞組（簡稱一次側繞組）[Simple diagram of the diagram] shows the conventional sinusoidal voltage inverter architecture diagram. The block diagram of the high efficiency spectral oscillator current source inverter circuit disclosed in the present invention is shown. The waveform timing diagram of each point of the high efficiency resonant current source inverter disclosed in the present invention is shown. The working mode diagram of the high efficiency spectral oscillator current source inverter disclosed in the present invention is shown. The 60 Hz measured waveform of each component of the high efficiency resonant current source inverter disclosed in the present invention is shown. The measured waveforms of the high frequency operation of each component of the high efficiency resonant current source inverter disclosed in the present invention are shown. The measured waveform diagram of the efficiency spectrum oscillator current source inverter disclosed in the present invention is unloaded and various loads. The conversion efficiency diagram of the high efficiency resonant current source inverter disclosed in the present invention and reference [11] is shown. [Main component symbol description] 1〇ι: DC power supply 1〇2: Clamp circuit 103: Coupling circuit 104: Spectral inverter circuit 105: Control and drive circuit 7]: Power semiconductor switch for clamp circuit (referred to as clamp switch) 28 1321392 , r2 : Power semiconductor switch for clamp circuit (referred to as clamp switch) . 7; +, 77, 7; + and 7; -: power semiconductor switch of resonant inverter (referred to as full bridge switch) 7; Transformer for magnetizing current (referred to as coupled inductor) A:: Coupling inductor 7; Coupling coefficient (referred to as coupling coefficient) b: Primary winding of coupled inductor C (referred to as primary winding)

Zy :耦合電感7；之二次側繞組（簡稱二次側繞組） k:諧振反流器電路之諧振電感 φ Θ:箝制電路之第一箝制二極體 A:箝制電路之第二箝制二極體 A:箝制電路之剎車二極體 A:耦合電路之整流二極體 Ci :諧振反流器電路之輸出電容 G:箝制電路之箝制電容Zy: coupled inductor 7; secondary winding (abbreviated as secondary winding) k: resonant inductor φ of resonant inverter circuit Θ: first clamped diode of clamp circuit A: second clamped diode of clamp circuit Body A: Brake diode of clamp circuit A: Rectifier diode of coupling circuit Ci: Output capacitor of resonant inverter circuit G: Capacitor of clamp circuit

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Claims (1)

1321392 十、申請範圍： 1 一種高效率譜振式電流源反流器，其中包含 一箝制電路：由兩個箝制開關、兩個箝制二極體、一剎 車二極體及一箝制電容所組成，主要是控制耦合電感之 電流與回昇能量； 一耦合電路：由耦合電感之一、二次側繞組及一整流二 極體所組成，一次側繞組限制直流電壓源之電流，並將 其導通期間所儲存能量藉由耦合電感傳遞至二次側繞 組，繼續釋放至諧振.反流器電路； 一諧振反流器電路：由四個全橋開關、一輸出電容及一 諧振電感所組成，主要導引一次側繞組電流至輸出電 容，以累積交流輸出電壓； 一控制驅動電路：係將弦波命令電壓與輸出電壓作閉迴 路控制，最後輸出至兩箝制開關及四個全橋開關所需之 驅動訊號； 耦合電感電路為輸入直流電源與諧振反流器電路之間 之緩衝電路，將兩者電壓源之壓差跨於耦合電感，並以 電流源方式呈現，該電流源電流經粉制電路之兩對稱紐 制開關及箝制二極體組合路徑，流入諧振反流器電路； 譜振反流為電路將來自直流電源端之電流導引至輸出 電容，並控制輸出電容之電壓極性，以產生交流電壓； 箝制電路除起斷反流器電路之電流之外，並以電壓箝制 搭配耦合電感兩繞組特性，使得全部功率半導體開關及 二極體均有柔性切換特性；箝制電路可降低功率半導體 30 1321392 器，其中電壓箝制電路中，剎車二極體主要提供輸出電 壓高於直流電源電壓之能量反饋路徑，此情形發生於負 載為交流電動機之飛輪慣量，以及電感性負載改變諧振 頻率所造成高電壓；剎車二極體將其能量牽引至直流電 源電壓端吸收，除可箝制所有開關電壓，並有效利用源 至負載之回昇式能量。 321321392 X. Scope of application: 1 A high-efficiency spectral oscillator current source inverter consisting of a clamp circuit consisting of two clamp switches, two clamp diodes, a brake diode and a clamp capacitor. Mainly to control the current and rebound energy of the coupled inductor; a coupling circuit consisting of one of the coupled inductor, the secondary winding and a rectifying diode, the primary winding limits the current of the DC voltage source, and turns it on during the conduction period. The stored energy is transferred to the secondary winding by the coupled inductor and continues to be discharged to the resonant. Inverter circuit; a resonant inverter circuit: consisting of four full bridge switches, an output capacitor and a resonant inductor, mainly guiding The primary winding current is output to the output capacitor to accumulate the AC output voltage; a control drive circuit: the closed-loop control of the sine wave command voltage and the output voltage, and finally the driving signals required for the two clamp switches and the four full-bridge switches The coupled inductor circuit is a buffer circuit between the input DC power supply and the resonant inverter circuit, and the voltage difference between the two voltage sources is coupled across the coupling The inductor is represented by a current source, and the current source current flows into the resonant inverter circuit through the two symmetric switch and the clamped diode combination path of the powder circuit; the spectral backflow is that the circuit will come from the DC power supply terminal. The current is directed to the output capacitor and controls the voltage polarity of the output capacitor to generate an AC voltage. The clamp circuit not only interrupts the current of the inverter circuit, but also clamps the two winding characteristics of the coupled inductor with voltage clamping, so that all power semiconductor switches Both the diode and the diode have flexible switching characteristics; the clamping circuit can reduce the power semiconductor 30 1321392, wherein in the voltage clamping circuit, the brake diode mainly provides an energy feedback path with an output voltage higher than the DC power supply voltage, which occurs when the load is The flywheel inertia of the AC motor and the high voltage caused by the inductive load changing the resonant frequency; the brake diode draws its energy to the DC power supply voltage terminal for absorption, in addition to clamping all switching voltages, and effectively utilizing the source-to-load return energy . 32