201246768 六、發明說明： 【發明所屬之技術領域】 〇〇本發明係有關於一種單開關電流源混合共振式轉換 器，特別係設有輸入電源連接儲能電感串聯功率開關，再 於功率開關上並聯分流電容及一組共振槽，共振槽係由共 振電感串聯共振電容後，再並聯另一共振電容所組成，該 共振槽連接橋式整流器，最後並聯低通濾波器及負載；如 此，利用單一個功率開關來做高頻的切換，可達到減少功 率開關的切換損失，並具有柔性切換的特性，同時提高轉 換器的操作效率，另共振槽特性阻抗及切換頻率可調的特 性，可增加彈性調整空間，而調整切換頻率的大小，可控 制輸出的電壓與電流，且藉由共振槽之二共振電容比值^ 調整’則可達到不同性質的轉換器。 【先前技術】 ^按，目前為改善傳統PWM轉換器的半導體元件在高頻 環境時，會產生切換損失的不良影響，以及能得到更高的 效率，近些年來，有許多關於柔切轉換器技術，即零電壓 切換(Zero Voltage Switching ; ZVS)與零電流切換(Zero201246768 VI. Description of the invention: [Technical field to which the invention pertains] The present invention relates to a single-switch current source hybrid resonant converter, in particular, an input power supply connected to an energy storage inductor series power switch, and then a power switch Parallel shunt capacitor and a set of resonant tanks, the resonant tank is composed of a resonant inductor series resonant capacitor, and then connected with another resonant capacitor, the resonant tank is connected to the bridge rectifier, and finally the parallel low-pass filter and the load; A power switch for high-frequency switching can reduce the switching loss of the power switch, and has the characteristics of flexible switching, while improving the operating efficiency of the converter, and the characteristic impedance of the resonant tank and the adjustable switching frequency can increase the elasticity. Adjusting the space and adjusting the switching frequency can control the voltage and current of the output, and the converter of different characteristics can be achieved by adjusting the ratio of the resonance capacitance of the resonant tank. [Prior Art] ^In order to improve the switching loss of the semiconductor components of the conventional PWM converter in the high-frequency environment, and to obtain higher efficiency, in recent years, there have been many about the flexible converter. Technology, Zero Voltage Switching (ZVS) and Zero Current Switching (Zero
Current Switching ; ZCS)的報告相繼的發表出來；例如， 串聯共振式轉換器(Series-Resonant Converter)、並聯共 振式轉換ll(Parallel-Resonant Converter)、串並聯共振 式轉換器（Series-Parallel Resonant Converter)，Current Switching; ZCS) reports have been published; for example, Series-Resonant Converter, Parallel-Resonant Converter, Series-Parallel Resonant Converter ),
Class-E轉換器，半共振式轉換器（Quasi_ Res〇nantClass-E converter, semi-resonant converter (Quasi_ Res〇nant
Converter) ’ 多共振式轉換器（Multi- Resonant Converter)，以及柔式切換（Soft—Switching)轉換器等， 都是利用開關元件在切換時，以正弦波或半正弦波形式來 產生共振，使得開關再換相時，處於零電壓或是零電流的 狀態，使得共振式轉換器在開關上的切換損失，或是開關 201246768 較於傳統簡的硬式轉換器而言，都可以得到 •生員的改.，緣此’本發明人有鑑於f知存在有如上 明：究、改良，遂得以首先發明本發明。 本發明之主要目的’係在提供一種利用單一個功率開 關來做高頻的切換，可達到減少功率開_切換損失，並 具有柔性娜的特性’啊提高機$賴作效率，另共 振槽特性阻抗及切換鮮可調的特性，可增加彈性調整空 間’而碰祕頻率的大小，可控制輸出的電壓與電流， 且藉由共振狀二共振電容雜_整，射達到不同性 質的轉換器之單開關電流源混合共振式轉換器。 本發明之特徵係在：輸入電源連接儲能電感串聯功率 開關’再於神_上並聯分流電容及—組共振槽，共振 槽係由共振電感㈣共振電容後，再並聯另—共振電容所 組成，該共振槽連接橋式整流H，最後並聯低财波器及 負載。 4 ° 【實施方式】 有關本發明為達上述之使用目的與功效，所採用之技 術手段，茲舉出較佳可行之實施例，並配合圖式所示， 述如下： Μ 本發明之實施例，請參閱第一、二圖所示，主要係設 有輸入電齡Α連接儲能械U聯功翔.再於功率2 心上並聯分流電容ς及一組共振槽！，共振槽J係由共振 電感ζ串聯共振電容Ci後’再並聯另一共振電容^所組成， 201246768 該共振槽1連接橋式整流器2，橋式整流器2係設有數二極 體（D1〜D4)所連接組成，最後並聯低通渡波器3及負載r， 低通濾、波II 3係設魏波電m线紅所連接組成。 使用日τ ，月參閱第一、二圖所示，首先在輸入電源^ (電 源側）輸人—直流電壓，經過儲能電感遂將直流電壓轉換 成電流源’再驅動功率開關蝴換導通，功率開關罐選擇 M0SFET電晶體開關，其内寄生之反向二極體可配合 作時流經功相，之逆向紐，共_ 1係由共振電感 尤、串聯共振電容c:與並聯共振電容[所組成，其輸入端由 分流電容ς供給能量，而輸出端並聯於橋式整流器2，以將 兩頻交流賴轉為直流龍，另_端係域波電感^與 慮波電奋⑽組成的低賴波器3,經由低賴波器3將高 頻雜訊齡後，可得到—穩定的直流電壓提供給負載^因 電路是操作在高頻的環境’所哺出端橋式整流H 2之整 ⑽一極體（D1〜D4)所需的逆向恢復時間需很快的恢復，才 能配合高頻的操作模式’故採用快速恢復二極體(f晰 Recovery)或是蕭特基二極體(Sch〇ttky)二極體。 本發明之工作模式分別為： 一、工作模式一(叫^<<yi/)，如第三圖所示： 驅動電⑸由低電位轉為高電位時，功率開關乂導 通’功率開關^上的電流〜由負值開始上升，並聯於功率開 關^的分流電容ς上之電^為零，故分流電私上並無; 201246768 _L過’由於電流K之電流值從負值轉變為零，所以有 一反向電流流經功率開關wb時功率開·上的恤，也 由負值開始上升’因共振賴〜跨壓在橋式整流器2之二 極體Ah上’咖二極動2私呈截止㈣，另一组二 極私與A並無跨I所以餘。流經二極私^，使得 二極體A與形成順向偏壓而導通，當電流^之電流值 上升至零時，進入工作模式二。 一、工作模式二（％ — <%)，如第四圖所示： 驅動電壓L為高電位，此時功率開關&導通，電流w 之電"!!_值大於零’故流經功率開關&，並聯於功率開關^上 的刀机電谷ς之電‘_為零’故分流電容上並無電流流 過’因共振電壓L跨壓在橋式整流n 2之二極體^和^ 上’所以二極體A與A呈截止狀態，另-組二極體/),與〇3並 無跨壓’所以電流^流經二極體A與A，使得二極體 形成順向偏壓鱗通，當共振槽丨之共振诚£1流々共振 至零時，進入工作模式三。 二、工作模式三（奴^奴<咚），如第五圖所示： 驅動電壓L維持在高電位，此時功率開關&導通，電 流ϋ之電流值大於零，故流經功率開關&，並聯於功率 開關S,上的分流電容C|之電流/c，為零，故分流電容ς上並無 電流流過，因共振電壓^跨壓在橋式整流器2之二極體A 和仏上，所以二極體^與仏呈截止狀態，另一組二極體^與 201246768 A並無跨壓’所以電流，·。流經二極體A與&，使得二極私 與a形成川員向偏壓而導通，共振電流開始逆向流經功率開 關*w回共振電容Cp，當共振槽i之共振電容電壓v由正 cp 值下降至零時，進入工作模式四。 四、工作模式四（岣S似<%)，如第六圖所示： 驅動電壓L維持在高電位，此時功率開關^導通，電 流b之電流值大於零，故流經功率開關5，並聯於功率開 關s上的刀"宁L電谷ς之電流^為零，故分流電容Cl上並無電 /❹α過’ ϋ共振槽i並聯之共振電容賴、為貞值，故橋 式整流器2之二極私與D3_止雜，共振電流為電軚 減掉電流，等於電流’而流經二極私動4，使得二極體 A與A形成順向偏壓而導通，共振電.逆向流經功率開關 •V再流回於共振電容q，t並聯於功糊歡上之分流電 容c,電流&由零開始上升時，進入工作模式五。 五、工作模式五Uv⑽，如第七圖所示： 驅動電壓fcs由局電位轉為低電位，此時功率開關乂截 止’功率開關^上之電流〜為零，因功率開關續止時，使 電流流經功率開_路徑形成斷路，電流ϋ之電流值大 於零’ 谷電壓％由零開始上升，電流流經分流電容^ 開始對分流電私充電，由於共振電壓〜小於零，故橋式 整流器2之二極體⑽仏呈截止狀態，共振電流為電軚減 掉電流纟等於電‘，喊經二極私私，使得二極私 201246768 與A形成順向偏璧而導通，當共振電容電^由負值上升 至零時，進入工作模式六。 工作模式六（⑽奴j，如第八圖所示·· .¾動電·ω維持在低電位，此時功率開關續止，功 。開關4上之電^為零’電流Κ之電流值大於零，分流 電容電壓々持續提升，電流流經分流電容ς繼續對分流電容 ς充電’由於共振電壓以、於零，故橋式整流器2之二極 體Α與Ζ)3呈截止狀態，共振電流為電流^減掉電流,。等於電 流：，而流經二極私料，使得二極體Α與⑽成順向偏 壓而導通’當共魏4雜至科，進U作模式七。 七、工作模式七（W%)，如第九圖所示： 驅動電氣維持在低電位，此時功率_截止，功 率_上之電^為零，電流^之電流值大於零，電流 流經分流電容ς繼續對分流電紅充電，由於共振電狄小 於零’故橋式整流器2之二極體D與&呈戴止狀態，共振 電流為電流、減掉電^等於電‘，而流經二極體· 仏’使得二極體仏與仏形成順向偏壓而導通，當並聯於開 關上之分流電容電U降轉時，進人工作模式八。 八、工作模式八U以，如 驅動電⑸維持在低電位，此時功率開_截止’功 率開_|上之她，為零’電流ϋ之電流值由零開始下降 為負值，分流電容電壓，始由峰值下降，由於共Converter) 'Multi-Resonant Converters, as well as Soft-Switching converters, use switching elements to generate resonances in the form of sine or half sine waves when switching. When the switch is re-synchronized, it is in a state of zero voltage or zero current, which makes the switching loss of the resonant converter on the switch, or the switch 201246768 can be obtained by the student than the conventional simple hard converter. Therefore, the inventors of the present invention have invented the present invention first in view of the above-mentioned findings and improvements. The main object of the present invention is to provide a high-frequency switching by using a single power switch, which can reduce the power on-switching loss, and has the characteristics of flexible ' 'ah, improve the efficiency of the machine, and the characteristics of the resonant tank. Impedance and switching of the freshly adjustable characteristics can increase the elastic adjustment space' and the frequency of the collision frequency, can control the output voltage and current, and achieve the different properties of the converter by the resonance of the two resonance capacitors. Single-switch current source hybrid resonant converter. The invention is characterized in that: the input power is connected to the energy storage inductor series power switch 'again the god _ upper parallel shunt capacitor and the group resonance tank, the resonance tank is composed of the resonance inductor (four) resonance capacitor, and then parallel and another resonance capacitor The resonant tank is connected to the bridge rectifier H, and finally the low-frequency inverter and the load are connected in parallel. 4 ° [Embodiment] The present invention is directed to the above-described objects of use and efficacy, and the preferred embodiments are described, and are as follows: Μ Embodiments of the present invention Please refer to the first and second figures. The main system is equipped with input electric age Α connection energy storage device U 联功翔. Then parallel the shunt capacitor 功率 and a set of resonant slots on the power 2 core! The resonant tank J is composed of a resonant inductor ζ series resonant capacitor Ci and then 'parallel to another resonant capacitor ^, 201246768. The resonant tank 1 is connected to the bridge rectifier 2, and the bridge rectifier 2 is provided with a plurality of diodes (D1~ D4) The connected components, the final parallel low-pass waver 3 and the load r, the low-pass filter, and the wave II 3 system are connected by Weibo electric m-line red. Use the day τ, month, as shown in the first and second figures, first input the power-DC voltage on the input power supply ^ (power supply side), convert the DC voltage into a current source through the energy storage inductor 再, and then drive the power switch to switch on and off. The power switch can selects the M0SFET transistor switch, and the anti-parasitic reverse diode can cooperate with the working phase, and the reverse phase, the total _ 1 is composed of the resonant inductor, the series resonant capacitor c: and the parallel resonant capacitor. The input terminal is supplied with energy by a shunt capacitor ,, and the output terminal is connected in parallel to the bridge rectifier 2 to convert the two-frequency alternating current into a DC dragon, and the other _ end system domain wave inductor ^ and the wave wave electric power (10) are low. After the high frequency noise is passed through the low frequency oscillating device 3, the sifter 3 can obtain a stable DC voltage to be supplied to the load, and the circuit is operated in a high frequency environment to feed the bridge rectifier R 2 The reverse recovery time required for the entire (10) one-pole (D1 to D4) needs to be quickly restored to match the high-frequency operation mode. Therefore, a fast recovery diode (F-Recovery) or a Schottky diode is used. (Sch〇ttky) diode. The working modes of the present invention are as follows: 1. Working mode 1 (called ^<<yi/), as shown in the third figure: When the driving power (5) changes from low potential to high potential, the power switch 乂 conducts 'power switch The current on ^ is increased by a negative value, and the power connected to the shunt capacitor 功率 of the power switch ^ is zero, so the shunt power is not private; 201246768 _L over 'because the current value of the current K changes from a negative value to a negative value Zero, so there is a reverse current flowing through the power switch wb when the power is turned on and the shirt is also started to rise by a negative value 'because of the resonance 〜~cross pressure on the diode rectifier 2 of the bridge rectifier 2 'Cai 2 pole 2 The private deadline is (4), and the other group is not cross-I. Passing through the two poles, the diode A is turned on by forming a forward bias, and enters the operating mode 2 when the current value of the current rises to zero. First, the working mode 2 (% - <%), as shown in the fourth figure: The driving voltage L is high, at this time the power switch & conduction, the current of the electric current "!!_ value is greater than zero' After the power switch &, parallel to the power switch ^, the electromechanical valley is electrically '_zero', so there is no current flowing through the shunt capacitor' due to the resonance voltage L across the voltage in the bridge rectifier n 2 The body ^ and ^ on 'so the diodes A and A are in the off state, the other - group diode /), and the 〇 3 does not have a crossover 'so the current ^ flows through the diodes A and A, making the diode A forward biased scale is formed, and when the resonance of the resonant tank is resonating to zero, it enters the working mode three. Second, the working mode three (slave slaves < 咚), as shown in the fifth figure: The driving voltage L is maintained at a high potential, at this time the power switch & conduction, the current value of the current 大于 is greater than zero, so flow through the power switch &, the current /c of the shunt capacitor C| in parallel with the power switch S, is zero, so no current flows through the shunt capacitor ,, because the resonant voltage is across the diode A of the bridge rectifier 2 And 仏, so the diodes and 仏 are in the off state, the other group of diodes ^ and 201246768 A have no cross-over 'so current, ·. Flowing through the diodes A and &, so that the two poles and the a formation of the Sichuanese are biased and turned on, the resonant current begins to flow backward through the power switch *w back to the resonant capacitor Cp, when the resonant capacitor voltage v of the resonant tank i is When the positive cp value drops to zero, it enters working mode four. Fourth, the working mode four (岣S like <%), as shown in the sixth figure: The driving voltage L is maintained at a high potential, at this time the power switch ^ is turned on, the current value of the current b is greater than zero, so it flows through the power switch 5 The current connected to the power switch s on the power switch s is zero, so there is no electric / ❹α over the shunt capacitor C. The resonant capacitor of the resonant tank i is parallel, which is a 贞 value, so the bridge type The two poles of the rectifier 2 are separated from the D3_, and the resonant current is the electric current minus the current, which is equal to the current' and flows through the two-pole teleport 4, so that the diodes A and A form a forward bias and conduct, and the resonant power Reverse flow through the power switch • V and then flow back to the resonant capacitor q, t parallel to the shunt capacitor c on the power paste, the current & when starting from zero, enter the working mode five. 5. Working mode Five Uv(10), as shown in the seventh figure: The driving voltage fcs changes from the local potential to the low potential. At this time, the power switch 乂 turns off the current on the power switch ^ is zero, because the power switch continues, The current flows through the power on_path to form an open circuit, and the current value of the current 大于 is greater than zero'. The valley voltage % starts from zero, and the current flows through the shunt capacitor ^ to start charging the shunt power. Since the resonance voltage is less than zero, the bridge rectifier The 2nd polar body (10) is in an off state, the resonant current is the electric power, the current is reduced, the electric power is equal to the electric power, and the second pole private 201266768 and A form a forward bias and turn on, when the resonant capacitor is electrically ^ When the negative value rises to zero, it enters the working mode six. Working mode six ((10) slave j, as shown in the eighth figure... 3⁄4 moving power · ω is maintained at a low potential, at this time the power switch continues, work. The electric current on switch 4 is zero 'current 电流 current value If it is greater than zero, the shunt capacitor voltage 々 continues to increase, and the current flows through the shunt capacitor ς to continue charging the shunt capacitor ' 'Because the resonance voltage is at zero, the diode rectifier 之 and Ζ of the bridge rectifier 2 are turned off, resonance The current is current ^ minus current. It is equal to the current:, and flows through the dipole private material, so that the diode Α and (10) become forward biased and turn on ‘When the total Wei 4 is to the branch, enter U for mode VII. 7. Working mode seven (W%), as shown in the ninth figure: The driving electrical is maintained at a low potential. At this time, the power_off, the power_on is zero, the current value of the current is greater than zero, and the current flows. The shunt capacitor ς continues to charge the shunting red, because the resonant dipole is less than zero', so the diode D of the bridge rectifier 2 and the & is in a wear state, the resonant current is the current, the subtraction is equal to the electric ', and the flow The diode ·' makes the diode 仏 and 仏 form a forward bias and conducts. When the shunt capacitor U connected in parallel with the switch is turned down, it enters the working mode eight. Eight, the working mode eight U to, such as the drive power (5) to maintain a low potential, at this time power on_off' power on _| on her, zero 'current ϋ current value from zero to a negative value, shunt capacitance The voltage begins to fall from the peak due to the total
CP 201246768 小於零，橋式整流器2之- 極體A與a呈截止狀態，共振 電流為電軚減掉電^等於·，電流流經二極•與 4使付-極體歧⑽成順向偏壓而導通，當共振槽丄 亚聯電容電壓、上升至零時，進入工作模式九。 九、工作模式九U心^)，如第十一圖所示： 驅動電壓p·維持在低雷 寸仕低電位’此時功率開關s,截止，功 率開歐上之電^為零，電流之電流值為負值，分流 電容電I維持下降狀態，共振電壓、由零開始上升，因丘 振電壓、跨壓在橋式整流器2之二極動加j，所以二 極體A細4呈截止雜4—組二極動，與A並無跨壓， 戶⑽電流,。流經二極私和，使得二極體A私形成順向 偏壓而導通，當分法雷交蕾^腐、 電電二'下降至零時，驅動電壓匕由 赠位轉為高電位’此時功率開關補導通後回到工作模 式一，完成-週工作週期的循環；此工作模式中只需要一 個雙向的導通開關，在此電路中是使用_et的電晶體開 關作為切換開關，由於咖了内部就有-個寄生的二極 體’所以獨要再另外並聯—個二極體，可以減少電路使 用的7L件’因為當開關在切換導通與截止時，分流電容電 h都是由零開始上升或是下降至零，都是在零的時候才 、刀換的鱗，所以糊在纽與導通時並無跨有電壓， 達到了零電壓切換，降低了開關在切換上的切換損失，且 有柔性切換的特性。 /、 201246768 而驅動電壓L與分流電容電壓^^，實測波形圖，如第十 二圖所示，其 CH1:X 軸：5//s/div、Y軸：10V/div; CH2:X軸：5/zs/div、Y軸：100V/div。 而分流電容電壓!^,與電流&實測波形圖，如第十三圖所 示，其CH1:X轴：5#s/div、Y軸：100V/div; CH2:X軸：5//s/div、Y軸：2A/div。 而分流電容電壓vc，與分流電容電流iCi實測波形圖，如第 十四圖所示，其 CH1:X軸：5/zs/div、Y軸：100V/div; CH2 : X軸：5#s/div、Y軸：2A/div。 而分流電容電壓與共振電流^實測波形圖，如第十五 圖所示，其 CH1:X軸：5/zs/div、Y軸：100V/div; CH2 : X 軸：5#s/div、Y 軸：2A/div。 而共振電壓vc„與共振電流^實測波形圖，如第十六圖所 示，其CH1:X軸：5/zs/div、Y軸：50V/div; CH2 : X軸：5//s/div、 Y軸：2A/div。 而共振並聯電容電壓vc,，與電流、實測波形圖，如第十 七圖所示，其 CHI : X 軸：5//s/div、Y 軸：50V/div ; CH2 : X軸：5/zs/div、Y軸：2A /div。 而共振串聯電容電壓vCv與電流/Cs實測波形圖，如第十 八圖所示，其 CHI : X 軸：5/zs/div、Y 軸：50V/div ; CH2:X軸：5#s/div、Y軸：2A/div。 而輸入共振槽1前端電壓^與輸出電壓^實測波形 201246768 圖，如第十九圖所示，其CHI : X軸：5/zs/div、Y軸： 100V/div ; CH2 : X 軸：5#s/div、Υ軸：50V /div。 而共振槽輸出電壓^與輸出電流實測波形圖，如第 二十圖所示，其 CHI : X 軸：5#s/div、Υ 軸：50V/div ; CH2 : X 轴·· 5/z s/div、Y 軸·· 2A/div。 而二極體A、A上電壓、,％與電流“實測波形圖， 如第二十一圖所示，其 CHI :X 軸：5/zs/div、Y 軸：50V/div; CH2 : X 軸：5 // s/div、Y 軸：2A/div。 而二極體A、A上電壓VD2,VW與電流‘k實測波形圖， 如第二十二圖所示，其 CHI: X 軸：5//s/div、Y 轴：50V/div; CH2 : X 軸：5 // s/div、Y 軸：2A/div。 而儲能電感電壓~與電流實測波形圖，如第二十三 圖所示，其 CH1 ·· X 軸：5//s/div、Y 軸：100V/div ; CH2 : X 軸：5/zs/div、Y軸：2A/div。 而共振電感電壓與電流/1實測波形圖，如第二十四圖 所示，其 CHI : X 軸：5//s/div、Y 軸：200V/div ; CH2: X 軸：5 // s/div、Y 軸：2A/div。 而濾波電感電壓\與電流、實測波形圖，如第二十五 圖所示，其 CHI : X 轴：5/zs/div、Y 軸：50V/div ; CH2 : X 轴：5//s/div、Y 軸：2A/div。 而負載並聯電容電壓vc<)與電流實測波形圖，如第二 十六圖所示，其 CHI : X 轴：5/zs/div、Y 軸：10V/div ; CH2 : X 軸：5 // s/div、Y 軸：2A/div。 11 201246768 示，其cm 電，實測波形圖，*第二十七圖所 CH2 J:5=1V、Y軸:50職V; 本發明經由選擇二,、”由：2纏V。 頻率，使功率開關以及共振 降低功率關d 轉於零電縣零電流的狀態，以 關㈣散熱問題===功率損失，亦可改善功率開 本發明升直流轉換直流的效率。 式轉換器工作於高:摔：f式器，係由於共振 流器中，因為Ε類4用在直流對交流的換 換損失減少，還有單一個開關，所以切 路可以比一般的共振電路以I類共振電 ’曰§竭關在導通之刖開關上的跨壓就降為跫，# pq _ ;導通的時候不會與開關上的電二產 i持===關上的導通電流必須在開 =為零&樣才不會和開關上的跨麗產生重疊導致截止 本，明所係由E類換流器電路的負載尺端增加一 式整流器2,當共振槽1輸出端的經過橋式整流器2將會被 整流成直流電’再經過-組域波電心和舰電容組 成的低通濾、波器3，過濾、成-個穩定的直流電給負載只，並 且可以藉由調整切換頻率來控制輸出電流與輸出電壓，此 電，不僅電路構造鮮且控制電路設計容易，由於電路僅 須單一個功率開關s ’有別於一般傳統的D類共振式轉換器 的雙開關，所以可以降低開關的切換損失，並且有柔性;刀 12 201246768 $的特性’因此可財效崎低切換損失以及提高轉換器 操作效率’另共振;ft 1特性阻抗及切換頻率可調的特性， 可增加彈性調整空間，而調整切換頻率的大小，可控制輸 出的電壓與電流，且藉由共振槽丨之二共 值的調整，則可達到不同性質的轉換器。电合i P比 ’ ί㈣實施例確實已能達到所預期之目的 及使用功效，且未見有_結構魏公知、公 ϊ本合τ專利之申請要件，爰依法提出申 心明早曰審、，，口，並核賜專利，實深任成荷。 【圖式簡單說明】 第一圖所示係為本發明實施例之電路圖。 第二圖所示係為本發明實施例之方塊圖。 第三圖所示係為本發明實施例卫作模式—之等效電路圖。 ^四圖所示係為本發明實施例1作模式二之等效電路圖。 =五圖所不係為本發明實施例卫作模式三之等效電路圖。 所示係為本發明實施例工作模式四之等效電路圖。 第七圖所示係為本發明實施例I作模式五之等 第八圖所示係為本發明實施例工作 ° 第九圖所示係為本發明實df電路圖。 第十圈所示係為本發明實施例工之^效電路围。 :卜_蝴她物 k與分流電容 第十二圖所示係為本發明實施例驅動 電壓'實測波形圖。 電容電壓VC,與電流 第十三圖所示係為本發明實施例分流 匕·實測波形圖。 13 201246768 第十四圖所示係為本發明實施例分流電容電壓vC|與分流 電容電流/c,實測波形圖。 第十五圖所示係為本發明實施例分流電容電壓VC|與共振 電流^實測波形圖。 第十六圖所示係為本發明實施例共振電壓VC„與共振電流 ^實測波形圖。 第十七圖所示係為本發明實施例共振並聯電容電壓vc„與 電流k實測波形圖。 第十八圖所示係為本發明實施例共振串聯電容電壓vCs與 電流，c,實測波形圖。 第十九圖所示係為本發明實施例輸入共振槽前端電壓V„ 與輸出電壓V6實測波形圖。 第二十圖所示係為本發明實施例共振槽輸出電壓與輸 出電流實測波形圖。 第二十一圖所示係為本發明實施例二極體A、Z)3上電壓 〜&與電流^ω實測波形圖。 第二十二圖所示係為本發明實施例二極體A、上電壓 vβ2,vw與電流‘“實測波形圖。 第二十三圖所示係為本發明實施例儲能電感電壓與電 流k實測波形圖。 第二十四圖所示係為本發明實施例共振電感電壓Vi與電. 流G實測波形圖。 第二十五圖所示係為本發明實施例濾波電感電壓與電 流~實測波形圖。 14 201246768 第二十六圖所示係為本發明實施例負載並聯電容電壓％ 與電流k實測波形圖。 第二十七圖所示係為本發明實施例輸出電壓與電流實 測波形圖。 【主要元件符號說明】 1 共振槽 2 橋式整流器 3 低通濾、波器 4輸入電源 s 功率開關 L 共振電感 D1〜D4二極體 Κ濾波電感 k、’C,、、’(：/,、，、乙 h儲能電感 ς分流電容 共振電容 R負載 c〇 濾波電容 lDlflD3、lD2，lD4、hf、lL〇、lC〇 VGS ' VC, ' VC;, ' VQ ' ' V6 ' VDpVD3 ' VD2,VM ' VLf ' VA ' V,〇 ' Vc〇 v0電壓 15CP 201246768 is less than zero, the bridge rectifier 2 - the polar body A and a are in the off state, the resonant current is the electric power minus the electric ^ equals ·, the current flows through the dipole • and the 4 makes the pay-polar body (10) in the forward direction It is biased and turned on. When the resonant tank 丄 sub-capacitor voltage rises to zero, it enters the operating mode nine. Nine, working mode nine U heart ^), as shown in the eleventh figure: The driving voltage p · is maintained at low low level low potential ' at this time power switch s, cut off, power on the open ^ ^ ^, current The current value is a negative value, the shunt capacitor I maintains a falling state, and the resonance voltage starts to rise from zero. Since the voltage of the hill and the voltage across the bridge rectifier 2 are increased by j, the diode A is thin. The cut-off 4-group is two-pole, and there is no cross-over with A, and the household (10) current. Passing through the poles of the two poles, the diode A is privately formed into a forward bias and turned on. When the method is divided into lei and er, the electric power is lowered to zero, and the driving voltage is changed from the gift to the high potential. When the power switch is turned on, it returns to the working mode one, and completes the cycle of the weekly working cycle; in this working mode, only one bidirectional conduction switch is needed. In this circuit, the transistor switch of _et is used as the switching switch, because the coffee There is a parasitic diode inside, so it is necessary to connect another diode in parallel, which can reduce the 7L parts used in the circuit. Because when the switch is switched on and off, the shunt capacitor h is zero. Starting to rise or fall to zero, the scales are changed at zero time, so there is no voltage across the paste and conduction, reaching zero voltage switching, reducing the switching loss of the switch in switching. And has the characteristics of flexible switching. /, 201246768 and the driving voltage L and the shunt capacitor voltage ^^, the measured waveform diagram, as shown in the twelfth figure, its CH1: X axis: 5 / / s / div, Y axis: 10V / div; CH2: X axis : 5 / zs / div, Y axis: 100V / div. The shunt capacitor voltage!^, and the current & measured waveform diagram, as shown in the thirteenth figure, its CH1: X axis: 5 # s / div, Y axis: 100V / div; CH2: X axis: 5 / / s/div, Y axis: 2A/div. The shunt capacitor voltage vc, and the shunt capacitor current iCi measured waveform diagram, as shown in the fourteenth figure, its CH1: X axis: 5 / zs / div, Y axis: 100V / div; CH2: X axis: 5 # s /div, Y axis: 2A/div. The shunt capacitor voltage and the resonant current ^ measured waveform diagram, as shown in the fifteenth figure, its CH1: X axis: 5 / zs / div, Y axis: 100V / div; CH2: X axis: 5 # s / div, Y axis: 2A/div. The resonance voltage vc „ and the resonance current ^ measured waveform diagram, as shown in the sixteenth figure, its CH1: X axis: 5 / zs / div, Y axis: 50V / div; CH2: X axis: 5 / / s / Div, Y axis: 2A/div. Resonant shunt capacitor voltage vc, and current, measured waveform diagram, as shown in Figure 17, CHI: X axis: 5 / / s / div, Y axis: 50V / Div ; CH2 : X axis: 5/zs/div, Y axis: 2A / div. And resonant series capacitor voltage vCv and current / Cs measured waveform, as shown in Figure 18, its CHI: X axis: 5 / Zs/div, Y axis: 50V/div; CH2: X axis: 5#s/div, Y axis: 2A/div. Input voltage of front end of resonant tank 1 and output voltage ^ measured waveform 201246768, as shown in the 19th As shown in the figure, its CHI: X axis: 5/zs/div, Y axis: 100V/div; CH2: X axis: 5#s/div, Υ axis: 50V / div. Resonant slot output voltage ^ and output current The measured waveform diagram, as shown in the twentieth diagram, has its CHI: X axis: 5#s/div, Υ axis: 50V/div; CH2: X axis·· 5/zs/div, Y axis·· 2A/div And the voltage, % and current on the diodes A and A, "measured waveform diagram, as shown in the twenty-first figure, its CHI: X axis: 5 / zs / div, Y : 50V / div; CH2: X axis: 5 // s / div, Y-axis: 2A / div. The voltage waveforms of VD2, VW and current 'k on diodes A and A, as shown in Fig. 22, are CHI: X axis: 5//s/div, Y axis: 50V/div; CH2 : X axis: 5 // s/div, Y axis: 2A/div. The stored energy inductor voltage ~ and current measured waveform diagram, as shown in the twenty-third figure, its CH1 · · X axis: 5 / / s / div, Y axis: 100V / div; CH2: X axis: 5 / zs /div, Y axis: 2A/div. The resonant inductor voltage and current / 1 measured waveform diagram, as shown in Figure 24, its CHI: X axis: 5 / / s / div, Y axis: 200V / div; CH2: X axis: 5 / s /div, Y axis: 2A/div. The filter inductor voltage \ and current, the measured waveform diagram, as shown in the twenty-fifth figure, its CHI: X axis: 5 / zs / div, Y axis: 50V / div; CH2: X axis: 5 / / s / Div, Y axis: 2A/div. The load shunt capacitor voltage vc<) and the current measured waveform diagram, as shown in the twenty-sixth figure, its CHI: X axis: 5/zs/div, Y axis: 10V/div; CH2: X axis: 5 // s/div, Y axis: 2A/div. 11 201246768 shows, its cm electricity, measured waveform diagram, * twenty-seventh figure CH2 J: 5 = 1V, Y axis: 50 job V; the present invention by selecting two, "by: 2 wrapped V. frequency, so The power switch and the resonance reduce the power off d to the zero current state of the zero electricity county, to turn off (4) heat dissipation problem === power loss, and also improve the power of the invention to increase the efficiency of the direct current conversion DC. The converter works at high: F: The f-type device is due to the fact that in the resonator, because the Ε class 4 is used for the DC-to-AC exchange loss reduction, there is also a single switch, so the circuit can be a type I resonant power than the general resonant circuit. § The cross-pressure on the switch after the turn-on is reduced to 跫, # pq _ ; when it is turned on, it will not be held with the electric switch on the switch. === The turn-on current must be on = zero & The sample will not overlap with the cross on the switch, resulting in the cutoff of the load. The rectifier is added to the load scale end of the E-type converter circuit. When the output of the resonant tank 1 passes through the bridge rectifier 2 will be rectified. It is a low-pass filter and waver 3 composed of a DC-sub-group wave core and a ship capacitor. Filtering, forming a stable DC power to the load only, and adjusting the switching frequency to control the output current and the output voltage. This electric circuit not only has a simple circuit structure and the control circuit design is easy, since the circuit only needs a single power switch s ' Different from the traditional dual switch of the D-type resonant converter, it can reduce the switching loss of the switch and has flexibility; the characteristic of the knife 12 201246768 $ can therefore reduce the switching loss and improve the operating efficiency of the converter. Another resonance; ft 1 characteristic impedance and adjustable switching frequency characteristics, can increase the elastic adjustment space, and adjust the switching frequency to control the output voltage and current, and by the adjustment of the two values of the resonance slot Converters of different natures can be achieved. The embodiment of the electric coupling i P is better than the ' ί 四 四 四 四 四 四 四 四 四 四 四 四 四 实施 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏 魏Put forward Shen Xinming’s early review, and mouth, and grant a patent, and Shen Shen Ren Chenghe. [Simplified illustration] The first figure shows the embodiment of the present invention. The second figure is a block diagram of an embodiment of the present invention. The third figure is an equivalent circuit diagram of the security mode of the embodiment of the present invention. The four figures are shown in the first embodiment of the present invention. The equivalent circuit diagram of the mode 2 is not equivalent to the equivalent circuit diagram of the mode 3 of the embodiment of the present invention. The figure is an equivalent circuit diagram of the working mode 4 of the embodiment of the present invention. The eighth embodiment of the present invention is the fifth embodiment of the present invention, which is shown in the figure of the present invention. The ninth figure is a circuit diagram of the actual df of the present invention. The tenth circle is shown in the embodiment of the present invention.效果 _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ Capacitor Voltage VC, and Current FIG. 13 is a shunt and measured waveform diagram of the embodiment of the present invention. 13 201246768 The fourteenth figure shows the shunt capacitor voltage vC| and the shunt capacitor current /c, which are measured waveforms. The fifteenth figure is a waveform diagram of the shunt capacitor voltage VC| and the resonance current ^ according to the embodiment of the present invention. Figure 16 is a waveform diagram of the resonance voltage VC „ and the resonance current ^ in the embodiment of the present invention. Figure 17 is a waveform diagram of the resonant parallel capacitor voltage vc „ and the current k in the embodiment of the present invention. Figure 18 is a diagram showing the resonant series capacitor voltage vCs and current, c, measured waveforms of the embodiment of the present invention. FIG. 19 is a waveform diagram of the input front end voltage V„ and the output voltage V6 of the embodiment of the present invention. FIG. 20 is a waveform diagram of the output voltage and output current of the resonant tank according to the embodiment of the present invention. The twenty-first figure shows the measured waveforms of the voltages ~& and the current ^ω on the diodes A, Z) 3 of the embodiment of the present invention. The twenty-second figure shows the diode of the embodiment of the present invention. A, the upper voltage vβ2, vw and the current '" measured waveform diagram. The twenty-third figure is a measured waveform of the energy storage inductor voltage and the current k according to the embodiment of the present invention. The twenty-fourth embodiment is a waveform diagram of the resonant inductor voltage Vi and the electric current G of the embodiment of the present invention. Figure 25 is a diagram showing the filtered inductor voltage and current ~ measured waveforms according to an embodiment of the present invention. 14 201246768 The twenty-fifth figure shows the measured waveform of the load parallel capacitor voltage % and current k according to the embodiment of the present invention. The twenty-seventh figure is a waveform diagram of the measured voltage and current of the embodiment of the present invention. [Main component symbol description] 1 Resonant slot 2 Bridge rectifier 3 Low-pass filter, wave 4 input power s Power switch L Resonant inductor D1 ~ D4 diode Κ filter inductor k, 'C,,, '(:/, , , , b, energy storage inductor, shunt capacitor, resonant capacitor, R load, c〇 filter capacitor, lDlflD3, lD2, lD4, hf, lL〇, lC〇VGS 'VC, 'VC;, 'VQ ' 'V6 'VDpVD3 ' VD2, VM ' VLf ' VA ' V,〇' Vc〇v0 voltage 15