201218603 六、發明說明： 【發明所屬之技術領域】 本發明是有關於一種升壓轉換器’特別是指一種節省元 件成本的高升壓比轉換器。 【先前技術】 升壓轉換器（Boost Converter)廣泛應用於各種電器，舉 凡電池、不斷電系統（UPS)、光伏系統或是太陽能發電設備， 皆需使用直流升壓轉換器將低壓直流轉為高壓直流輸出。其 中，不斷電系統與光伏系統用較高電壓的轉換電路架構，主 要是將低壓轉為高壓直流後再轉為交流輸出。 已知高升壓比電路有利用磁性元件耦合繞組，或是利用 電何幫浦（Charge Pump)與開g式電容（Swhched ca_㈣ 來進饤電壓疊加’甚至是以上兩種之混合；也有其他相當複 雜的架構，甚至使用二個以上的主動開關和大量被動元件， 但轉換效率*佳，歧^能使帛於低功率應用。201218603 VI. Description of the Invention: [Technical Field] The present invention relates to a boost converter', particularly to a high boost ratio converter which saves component cost. [Prior Art] Boost Converter is widely used in various electrical appliances. For batteries, uninterruptible power systems (UPS), photovoltaic systems, or solar power generation equipment, DC boost converters are required to convert low-voltage DC to High voltage DC output. Among them, the higher voltage conversion circuit architecture of the uninterruptible power system and the photovoltaic system mainly converts the low voltage into high voltage direct current and then converts it into an alternating current output. High-boost ratio circuits are known to use magnetic components to couple windings, or to use a charge pump and an open-g capacitor (Swhched ca_(4) to add voltage to the voltage combination' or even a mixture of the above two; there are other rather complicated The architecture even uses more than two active switches and a large number of passive components, but the conversion efficiency is good, which can be used in low power applications.
前述架構有其缺點，亦有其他形式之電路架構，但是需 要額外的隔離驅動電路，如此將徒增系統複雜度，即便有高 升壓比，但卻無法精簡設計。 【發明内容】 LJ此，本發明之目的，即在提供 種免除複雜設計 一-… 卞Μ 節省元件成本的高升壓比轉換器 於是’本發明高升壓 ^ , 发匕锝換器電連接於一電源及一負載 之間’該尚升壓比轉換薄白 第，… 轉換15包含-輸入電容、-第-電感、- 第一順向導通元件、一坌 第-项向導通元件、—第二電感、一 201218603 跨接電容、一第一開關元件、一第二開關元件及一輸出電容。 該輸入電容具有一電連接於該電源的第一端及一接地 的第二端；該第一電感具有一與該電源電連接的第一端及一 第二端；該第一順向導通元件具有一與該第一電感的第二端 電連接的第一端及一第二端。 該第二順向導通元件具有一與該電源電連接的第一端 及一第二端；該第二電感具有一與該第二順向導通元件的第 二端電連接的第一端，及一與該第一順向導通元件的第二端 電連接的第二端；該跨接電容電性連接在該第一電感的第二 端與該第二順向導通元件的第二端之間。 該第一開關元件具有一接地的第一端及一與該第一順 向導通元件的第二端電連接的第二端；該第二開關元件具有 一與該第二電感的第二端電連接的第一端及一與該負載電 連接的第一端，該輸出電容具有一電連接於該負載與該第二 開關元件的第二端之間的第一端及一接地的第二端。 當該第一開關元件導通及該第二開關元件不導通，此時 形成兩個電流迴路，其中之一電流迴路的電流由該輸入電容 流經該第一順向導通元件及該第二順向導通元件，且該第一 順向導通元件及該第二順向導通元件被順偏導通，令該跨接 電容為充電狀態且其充電電壓為電源之電壓，該第一電感、 該第二電感同時跨電源之電壓而激磁，另一電流迴路的電流 由該輸出電容流經該負載；當該第一開關元件不導通及該第 二開關7G件導通，電流由該輸入電容流經該第一電感、該跨 接電容及該第二電感’且該第一電感及該第二電感為去磁， 201218603 該跨接電容為放電。 本發明高升壓比轉換器藉由前述元件，利賴似現有升 壓轉換器的控制模式及簡易的設計，可達到節省元件成本的 功效》 【實施方式】 有關本發明之前述及其他技術内容、特點與功效，在以 下配合參考圖式之較佳實施例的詳細說明中將可清楚的呈 現。The aforementioned architecture has its shortcomings, and there are other forms of circuit architecture, but additional isolation drive circuitry is required, which increases the complexity of the system, even with high boost ratios, but does not simplify the design. SUMMARY OF THE INVENTION LJ, the object of the present invention is to provide a high-boost ratio converter that saves component cost by providing a complex design-... ' high voltage booster of the present invention, and electrical connection of the converter Between a power supply and a load 'the boost ratio is thinner, the conversion 15 includes - input capacitance, - first inductance, - first forward conduction component, one first - guide conduction component, The second inductor, a 201218603 jumper capacitor, a first switching component, a second switching component, and an output capacitor. The input capacitor has a first end electrically connected to the power source and a second end connected to the ground; the first inductor has a first end and a second end electrically connected to the power source; the first forward conducting component There is a first end and a second end electrically connected to the second end of the first inductor. The second forward conducting component has a first end and a second end electrically connected to the power source; the second inductor has a first end electrically connected to the second end of the second forward conducting component, and a second end electrically connected to the second end of the first forward conducting component; the jumper capacitor is electrically connected between the second end of the first inductor and the second end of the second forward conducting component . The first switching element has a first end connected to the ground and a second end electrically connected to the second end of the first forward conducting element; the second switching element has a second end electrically connected to the second inductor a first end of the connection and a first end electrically connected to the load, the output capacitor having a first end electrically connected between the load and the second end of the second switching element and a grounded second end . When the first switching element is turned on and the second switching element is not turned on, two current loops are formed, wherein a current of one of the current loops flows from the input capacitor through the first forward conducting component and the second forward direction a conducting component, wherein the first forward conducting component and the second forward conducting component are turned on, the charging capacitor is in a charging state, and the charging voltage is a voltage of the power source, the first inductor and the second inductor Simultaneously, the voltage is excited across the voltage of the power source, and the current of the other current loop flows through the load; when the first switching element is not turned on and the second switch 7G is turned on, the current flows through the first input capacitor. The inductor, the jumper capacitor and the second inductor 'and the first inductor and the second inductor are demagnetized, and the jumper capacitor is discharged in 201218603. The high boost ratio converter of the present invention can achieve the effect of saving component cost by utilizing the aforementioned components, and the control mode and simple design of the existing boost converter can be achieved. [Embodiment] The foregoing and other technical contents related to the present invention The features and advantages of the invention will be apparent from the following detailed description of the preferred embodiments.
參閲圖i，本發明之較佳實施例中，高升壓比轉換器_ 電連接於-電源及-負載a之間，且高升壓比轉換器⑽包 含一輸入電容一第一電感A、一第—順向導通元❹1、 -第二順向導通元件A、一第二電感々、_跨接電容。、一 第-開關元❹,、-第二開關元件狀—輸出電容〇 輸入電容C；具有_電連接於電源的第—端U及一接地 的第二端12;第一電感^具有一與電源電連接的第—端Η 二：端22;第一順向導通元件Α具有—與第 第二端22電連接的第一端31及一第二端U。 第二順向導通元件Α具有一與電源電連接的第一端Μ 及一第一端42;第二電咸右—你哲 具有肖第二順向導通元 第二端42電連接的第-端5卜及-與第-順向導通元“ 的第二端32電連接的第二端52;跨接電μ電性連接在第 電感糊二端22與第二順向導通元件Α的第二 間0 第一端61及—與第一順 第一開關元件2,具有一接地的 201218603 二導::件明第二端32電連接的第二端62;第二開關元 U一與第二電感A的第二端52電連接的第一端71及 一與負載㈣連接的第二端72;輪出電^具有—電連接於 負載績第二開關元件込的第二端72之間的第一端以及― 接地的第二端82。 本較佳實施例中，第—順向導通元件A及第二順向導通 元件A均為二極體，各第一端31、41皆為？極，各第二端 32、42皆為n極；第—開關元件⑽第二開關元件㈣各 第一端6卜71與各第二端62、72之間各反向連接一二極體 A,、A2 ;此外’第一開關元件β及第二開關元件仏皆為n 型金氧半場效電晶體’其閘極則受控制以決定第一開關元件 β及第二開關元件a導通與否’各第一端6卜71皆為源極， 各第二端62、72皆為汲極。 參閱圖2，本發明高升壓比轉換器1〇〇之控制架構包括 一比較器ιοί、一場可程式閘陣列（FPGA)1〇2 一半橋式閘驅 動器（Half-bridge gate driver) 103 及一電壓驅動器（v〇itage driver)104,其中，前述的控制架構可參考2〇〇6年申請人於 IEEE APEC’06會議發表的論文“以場可程式閘陣列的前饋 轉換器應用於記數為基礎的脈波寬度調變的控制系統 (Applying a counter-based PWM control scheme to an FPGA-based SR forward converter)，’。比較器 101 取得電壓 驅動器104之輸出訊號與輸入電壓比較後產生回饋控制訊 號VFB ’場可程式閘陣列102接收比較器101之回饋控制訊 號VFB並產生脈波控制訊號]^1及M2,藉此對應驅動第一開 201218603 關：件⑽第二開關元件a ’此外，附加半橋式閉驅動器 亦可驅動第一開關元件β及第二開關元件込。 參閲圖3至圓6皆工作於連續導通模式，且根據第一電 感A與第二電感£2電感量大小區分各種狀態，且各種狀_且 有的操作模式分析如下。 心八 第—狀態··假設電感A之電感值等於電感&之電感值， 且操作於滿載（1GG%)，具有二種操作模式介紹如下。 參閱圖3A，第-模式中，第一開關元件g導通及第二 開關元件込不導通，此時形成兩個電流迴路，其中之一電流 迴路的電流由輸人電容Q流經第-順向導通元件A及第二 順向導通7C件&，且第—順向導通元件A及第二順向導通元 件A被順偏導通，令跨接電容q為充電狀態且其充電電壓為 電源㈣壓’第一電感Ζι、第二電感&同時跨電源⑽ 壓而激磁，另一電流迴路的電流由該輸出電容〇。流經負載 及m順向導通元件n順向導通元件域順 ，導通（forward-based)’跨接電容〇;為充電狀態，且跨接電 谷ς的充電電壓為輸入電壓匕’第一電感A、第二電感々同 時跨輸入電壓4而激磁（magnetized)，此時輸出能量由輸 出電容C。提供。 參閱圖3B，第二模式中，第一開關元件β不導通及第 二開關元件&導通，電流由該輸人電容&流經該第一電感 A、該跨接電容Ce及該第二電“，且第—電感^及第二電 感々為去磁，該跨接電容&為放電。 假設第一模A中的第一電感4與第二電感&的跨壓分 201218603 別為Zil-0W及’第二模式在第一電感/1及第二電感4的跨 壓分別為及K2_0fF ;既然跨接電容Ce的電壓L等於輸入 電壓匕，因此第一模式中的第一電感^與第二電感4的跨壓 么別為及L-CW等於輸入電壓匕，根據伏秒平衡原理 (v〇ltage-second balance) ’ Ζ)χΚ=(ΐ-£〇>^_，第二模式在第 一電感Α及第二電感Α的跨壓4._0汗及K2_0JV可表示成：Referring to FIG. 1, in a preferred embodiment of the present invention, a high boost ratio converter _ is electrically connected between a power supply and a load a, and a high boost ratio converter (10) includes an input capacitor and a first inductor A. , a first - forward guide pass ❹ 1, - a second forward conduction component A, a second inductance 々, _ jumper capacitor. a first-switch element ❹, a second switch element--output capacitor 〇 input capacitor C; having a first end U electrically connected to the power source and a grounded second end 12; the first inductor ^ has a The first end of the power supply is connected to the second end 22; the first forward conducting element has a first end 31 and a second end U electrically connected to the second end 22. The second forward conducting component Α has a first end 电 and a first end 42 electrically connected to the power source; the second electric ampere right-you have the second electrical connection of the second cistern passing end 42 a terminal end 52 and a second end 52 electrically connected to the second end 32 of the first-directional conductive element; the first electrical connection between the second inductor 22 and the second forward conducting component The two first ends 61 and the first first switching element 2 have a grounded 201218603 two-conductor: a second end 62 electrically connected to the second end 32; the second switching element U and The second end 52 of the second inductor A is electrically connected to the first end 71 and the second end 72 connected to the load (4); the turn-off power is electrically connected between the second end 72 of the second switching element The first end and the grounded second end 82. In the preferred embodiment, the first forward conducting component A and the second forward conducting component A are diodes, and each of the first ends 31, 41 is The second end 32, 42 are all n poles; the first switching element (10), the second switching element (4), the first end 6b 71 and each of the second ends 62, 72 are respectively connected to a diode A, A2; The first switching element β and the second switching element 仏 are both n-type MOS field-effect transistors, the gates of which are controlled to determine whether the first switching element β and the second switching element a are turned on or not. Each of the second ends 62, 72 is a drain. Referring to Figure 2, the control architecture of the high boost ratio converter of the present invention includes a comparator ιοί, a programmable gate array (FPGA). 1〇2 Half-bridge gate driver 103 and a voltage driver (v〇itage driver) 104, wherein the aforementioned control architecture can refer to the IEEE APEC '06 meeting for 2-6 years. Published paper "Applying a counter-based PWM control scheme to an FPGA-based SR forward converter", 'The application of a counter-based PWM control scheme to an FPGA-based SR forward converter' . Comparing the output signal of the voltage driver 104 with the input voltage, the comparator 101 generates a feedback control signal VFB. The field programmable gate array 102 receives the feedback control signal VFB of the comparator 101 and generates pulse wave control signals ^1 and M2. Corresponding drive first opening 201218603 OFF: member (10) second switching element a ' In addition, the additional half bridge type closed driver can also drive the first switching element β and the second switching element 込. Referring to Figures 3 through 6, all operate in the continuous conduction mode, and the various states are distinguished according to the magnitude of the first inductance A and the second inductance, and the various modes of operation are analyzed as follows. Heart 8 - State · Assume that the inductance of inductor A is equal to the inductance of inductor & and operates at full load (1GG%). Two modes of operation are described below. Referring to FIG. 3A, in the first mode, the first switching element g is turned on and the second switching element is not turned on. At this time, two current loops are formed, and one of the current loop currents flows through the first-forward direction from the input capacitor Q. The conduction element A and the second forward conduction 7C piece & and the first forward conduction element A and the second forward conduction element A are turned on, so that the jump capacitor q is in a charging state and the charging voltage is a power source (4) The voltage of the first inductor 第二ι, the second inductor & is simultaneously excited across the power source (10), and the current of the other current loop is 〇 by the output capacitor. Flow through the load and the m-direction conduction component n is forward-directed to the component, and the forward-based capacitor is connected; the state of charge is charged, and the charging voltage across the electric valley is the input voltage 匕 'first inductance A. The second inductor mag is simultaneously magnetized across the input voltage 4, and the output energy is output capacitor C at this time. provide. Referring to FIG. 3B, in the second mode, the first switching element β is non-conducting and the second switching element is turned on, and current flows from the input capacitor & through the first inductor A, the jumper capacitor Ce, and the second Electrically, and the first inductor and the second inductor 去 are demagnetized, and the jumper capacitor & is discharged. It is assumed that the first inductor 4 and the second inductor & The voltage across the Zil-0W and the 'second mode at the first inductor /1 and the second inductor 4 is respectively K2_0fF; since the voltage L across the capacitor Ce is equal to the input voltage 匕, the first inductor in the first mode^ The cross-voltage with the second inductor 4 is equal to and the L-CW is equal to the input voltage 匕, according to the volt-second balance principle (v〇ltage-second balance) 'Ζ) χΚ=(ΐ-£〇>^_, second The mode voltage between the first inductance 第二 and the second inductance 4 4._0 汗 and K2_0JV can be expressed as:
ViA-〇FF = 反 VL2~〇FF =公式 1 在第二模式，輸出電壓K可表示成： — ^L\-0FF + ^Ll-OFF + VCe+Vin 公式 2 將公式1代入公式2產生以電壓轉換比率表示式為： K _ 2 vin Ι-D 公式 3 第二狀態：假設電感A之電感值等於電感々之電感值， 且操作於輕載（10%) ’具有五種操作模式介紹如下。 參閱圖4A，第一模式中，第一開關元件Q導通及第二 開關元件込不導通，此時，第一順向導通元件A、第二順向 導通元件A被順偏導通，跨接電容q的充電電壓為輸入電壓 匕，第一電感A、第二電感々同時跨輸入電壓匕而激磁，輸 出能量由輸出電容Q提供’且跨接電容q為充電狀態。 參閱圖4B，第二模式中，第一開關元件^不導通及第 二開關元件込導通’電流由該輸入電容Q流經該第一電感 '、該跨接電容(^及該第二電感Z2，且第一電感七及第二電 感々為去磁，該跨接電容（；為放電。 參閱圖4C，第三模式中，第一開關元件^仍不導通及 第一開關元件β2仍導通，第一順向導通元件孕、第二順向導 201218603 通元件z>2仍被順偏導通，此時，輸出電壓匕釋放至輸出端， 且第電感A及第一電感A為激磁（magnetized)至反方向，該 跨接電容q為充電。 參閱圖4D，第四模式中，第一開關元件Q導通及第二 開關元件込不導通，第一順向導通元件A、第二順向導通元 件A被反向偏壓，此時，第一電感A、第二電感心在反方向 為去磁，且跨接電容Ce為充電狀態。 參閱圖4E，第五模式中，第一開關元件&仍導通及第 二開關元件込仍不導通，第一順向導通元件A、第二順向導 通元件A被反向偏壓，此時，跨接電容&為放電狀態，造成 第一電感A、第二電感矣為激磁。 第二狀態·假设電感&之電感值大於電感尽之電感值’ 且操作於滿載（100%)，具有三種操作模式介紹如下。 參閱圖5A，第一模式中，第一開關元件&導通及第二 開關元件込不導通，此時，第一順向導通元件A、第二順向 導通元件A被順偏導通，跨接電容&的充電電壓為輸入電壓 匕，第一電感々、第二電感乓同時跨輸入電壓&而激磁，因 第一電感心之電感值大於第二電感&之電感值，電流小 於 IL2 〇 參閱圖5B，第二模式中，第一開關元件^不導通及第 一開關元件込導通，由於第一電感马之電感值大於第二電感 A之電感值，使得第二模式初始的電流Iu小於電流，依 照電流定律，電流Ια之一部分迫使第二順向導通元件A順 向偏壓導通，因此第一電感马被跨接電容^的跨壓繼續正向 201218603 激磁’電流iL丨繼續增加，第-雷戍r达上 $ —電感12為去磁，電流Il2下降 直到電流IL丨等於電流IL2時進入第三模式。 參閱圖5C，第三模式中，第一開關元件⑽導通及第 二開關元件&導通，此時輪出端能量由輸入端加上第一電感 4、第二電感與跨接電容ς的能量所提供，在此模式下， 第-電感Α、第-電感12為去磁，跨接電容q為放電。 第四狀態：假設電感4之電感值小於電感A之電感值， 且操作於滿載⑽％)，具有三種操作模式介紹如下。ViA-〇FF = inverse VL2~〇FF = formula 1 In the second mode, the output voltage K can be expressed as: — ^L\-0FF + ^Ll-OFF + VCe+Vin Equation 2 Substituting Equation 1 into Equation 2 The voltage conversion ratio is expressed as: K _ 2 vin Ι-D Equation 3 Second state: Assume that the inductance of inductor A is equal to the inductance of inductor ,, and operate at light load (10%) 'There are five modes of operation as follows . Referring to FIG. 4A, in the first mode, the first switching element Q is turned on and the second switching element is not turned on. At this time, the first forward conducting component A and the second forward conducting component A are turned on, and the capacitor is connected across the capacitor. The charging voltage of q is the input voltage 匕, the first inductor A and the second inductor 激 are simultaneously excited across the input voltage ,, the output energy is provided by the output capacitor Q and the jumper capacitor q is in a charged state. Referring to FIG. 4B, in the second mode, the first switching element is not turned on and the second switching element is turned on. The current flows from the input capacitor Q through the first inductor, and the jumper capacitor (^ and the second inductor Z2) And the first inductor 7 and the second inductor 々 are demagnetized, and the jumper capacitor is a discharge. Referring to FIG. 4C, in the third mode, the first switching component is still not turned on and the first switching component β2 is still turned on. The first forward conduction component is pregnant, and the second forward guide 201218603 through component z>2 is still turned on. At this time, the output voltage 匕 is released to the output terminal, and the first inductor A and the first inductor A are magnetized to In the reverse direction, the jumper capacitor q is charged. Referring to FIG. 4D, in the fourth mode, the first switching element Q is turned on and the second switching element is not turned on, and the first forward conducting component A and the second forward conducting component A are When the voltage is reverse biased, the first inductor A and the second inductor are demagnetized in the opposite direction, and the jump capacitor Ce is in a charged state. Referring to FIG. 4E, in the fifth mode, the first switching element & The conduction and the second switching element 込 are still not turned on, and the first forward conduction element A The second forward conducting component A is reverse biased. At this time, the jump capacitor & is in a discharged state, causing the first inductor A and the second inductor 矣 to be excited. The second state assumes that the inductance & inductance value is greater than The inductor has the inductance value ' and operates at full load (100%). There are three modes of operation as follows. Referring to FIG. 5A, in the first mode, the first switching element & and the second switching element are not turned on. The first forward conducting component A and the second forward conducting component A are turned on, and the charging voltage across the capacitor & is the input voltage, and the first inductor and the second inductor are simultaneously excited across the input voltage & Because the inductance value of the first inductor core is greater than the inductor value of the second inductor &amplifier, the current is less than IL2. Referring to FIG. 5B, in the second mode, the first switching component is not turned on and the first switching component is turned on, due to the first The inductance value of the inductance horse is greater than the inductance value of the second inductance A, so that the initial current Iu of the second mode is smaller than the current. According to the current law, one part of the current Ια forces the second forward conduction element A to be forward biased, so the first The inductor horse is connected to the capacitor ^ across the voltage and continues to forward 201218603. The excitation 'current iL丨 continues to increase, the first - thunder r reaches $ - the inductor 12 is demagnetized, and the current Il2 drops until the current IL 丨 equals the current IL2. Referring to FIG. 5C, in the third mode, the first switching element (10) is turned on and the second switching element is turned on, and the wheel end energy is added to the input terminal by the first inductor 4, the second inductor and the jumper capacitor. The energy of ς is provided. In this mode, the first-inductance Α and the first-inductor 12 are demagnetized, and the jumper capacitor q is a discharge. The fourth state: assume that the inductance value of the inductor 4 is smaller than the inductance value of the inductor A, and the operation At full load (10)%), there are three modes of operation as described below.
參閱圖6A，第一模式中，第-開關元件e,導通及第二 開關元件02不導通’此時，第—順向導通元件ZV第二順向 導通元件A被順偏導通，跨接電容Q的充電電壓為輸入電壓 ίη第€感A第—電感同時跨輸入電壓匕而激磁，因 第一電感A之電感值小於第二電感^電感值，電流^大 於 IL2。 參閱圖6B，第二模式中，第一開關元件Q不導通及第 二開明元件22導通，由於第-電感A之電感值小於第二電感 々之電感I，使得第二模式初始的電流^大於電流匕，依 照電流定律，電流I, , > _ 、， L1之。卩分迫使第一順向導通元件旱順 向偏壓導通’因此第H2被跨接電容ce的跨壓繼續正向 激磁’電流U續增加，第—電感&為去磁，電流^下降， 直到電流1L1等於電流IL2時進人第三模式。 參閱圖6C，第三模式中，第-開關元件β不導通及第 開關兀件導通，此時輸出端能量由輸入端加上第一電感 A、第-電感12與跨接電容q的能量所提供，在此模式下， 10 201218603 第電感A、第二電感12為去磁，跨接電容為放電。 參閱圖7至圖9，為本較佳實施例之實驗結果，實驗的 没&條件為10伏至16伏，輸出電壓為60伏，直流電流為 1安培’開關頻率為195kHz，配合圖1，第一順向導通元件 A、第二順向導通元件A的型號為STPS15H100CB，第一開 關兀*件3、第二開關元件込的型號為IRF3710ZS ;半橋式閘 驅動器103的型號為HIP2101 ;跨接電容(^的容值為270// F，輸出電容C；的容值為330 // F ;輸入電容的容值為1800 β F ’場可程式閘陣列102的型號為EP1C3T100。 參閱圖7A至圖7C，本較佳實施例中，第一電感&之電 感值等於第二電感乓之電感值’圖7A為無負載（no load)、 圖7B為半載（half load)，及圖7C為滿載（rated load)，且圖 7A至圖7C各圖中具有第一電感電流iu、第二電感電流 iu、第一開關電壓Vgs]及跨接電容電壓Vc。的量測結果， 可觀察到第一電感電流IL1幾乎等於第二電感電流Iu。 參閱圖8A至圖8C ’為本較佳實施例中，第一電感a之 電感值大於第一電感A之電感值，圖8A為無負載、圖8B 為半載’及圖8C為滿載’且圖8A至圖8C各圖中具有第一 電感電流iLi、第二電感電流、第一開關電壓Vgsi及跨接 電容電壓Vc。的量測結果’可觀察到第一電感電流Iu大於 第二電感電流IL2。 參閱圖9A至圖9C，為本較佳實施例中，第一電感及之 電感值小於第二電感A之電感值，圖9A為無負載、圖9B 為半載，及圖9C為滿載’且圖9A至圖9c各圖中具有第一 201218603 電感電流1L丨、第二電感電流iLz、第一開關電壓丨及跨接 電谷電壓VC。的量測結果，可觀察到第一電感電流小於 第二電感電流Iu。 綜上所述’本發明高升壓比轉換器100藉由前述元件組 成，利用類似現有升壓轉換器的控制模式及簡易的設計，可 節省元件成本，故確實能達成本發明之目的。 惟以上所述者，僅為本發明之較佳實施例而已，當不能 以此限定本發明實施之範圍，即大凡依本發明中請專利範圍 及發明說明内容所作之簡單的等效變化與修飾，皆仍屬本發 明專利涵蓋之範圍内。 【圖式簡單說明】 圖1是一電路圖，說明本發明高升壓比轉換器之較佳實 施例； 圖2是一電路圖，說明本發明高升壓比轉換器之較佳實 施例之控制架構； 圖3A至圖3B是電路圖，說明本較佳實施例的第一狀 態的兩種操作模式的電流流向； 圖4A至圖4E是電路圖，說明本較佳實施例的第二狀 態的五種操作模式的電流流向； 圖5A至圖5C是電路圖，說明本較佳實施例的第三狀 態的三操作模式的電流流向； 圖6A至圖6C是電路圖，說明本較佳實施例的第四狀 態的各操作模式的電流流向； 圖7A至圖7C是波形圖，說明本較佳實施例的第一及 12 201218603Referring to FIG. 6A, in the first mode, the first switching element e, the conducting and the second switching element 02 are not conducting. At this time, the second forward conducting component ZV is forwardly turned on, and the capacitor is connected across the capacitor. The charging voltage of Q is the input voltage ίη. The first inductance - the inductor is excited simultaneously across the input voltage, because the inductance of the first inductor A is smaller than the inductance of the second inductor, and the current ^ is greater than IL2. Referring to FIG. 6B, in the second mode, the first switching element Q is non-conducting and the second enlightening element 22 is turned on. Since the inductance of the first inductor A is smaller than the inductance I of the second inductor, the initial current of the second mode is greater than Current 匕, according to the current law, current I, , > _,, L1. The enthalpy forces the first forward conduction component to be forward biased. Therefore, the second H2 is continuously excited by the voltage across the capacitor ce. The current U continues to increase. The first inductance & is demagnetization, and the current ^ decreases. The third mode is entered until the current 1L1 is equal to the current IL2. Referring to FIG. 6C, in the third mode, the first switching element β is not turned on and the first switching element is turned on. At this time, the energy of the output end is added by the input end to the energy of the first inductor A, the first inductor 12 and the jump capacitor q. Provided, in this mode, 10 201218603 The first inductor A and the second inductor 12 are demagnetized, and the jumper capacitor is discharged. Referring to FIG. 7 to FIG. 9 , the experimental results of the preferred embodiment are as follows: the experimental conditions are 10 volts to 16 volts, the output voltage is 60 volts, the direct current is 1 ampere, and the switching frequency is 195 kHz. The model of the first forward conduction element A and the second forward conduction element A is STPS15H100CB, the model of the first switch 兀*3, the second switching element 为 is IRF3710ZS, and the model of the half bridge brake actuator 103 is HIP2101; The capacitance of the jumper capacitor (^ is 270//F, the output capacitor C; the capacitance is 330 // F; the capacitance of the input capacitor is 1800 β F 'The field programmable gate array 102 is model EP1C3T100. 7A to 7C, in the preferred embodiment, the inductance value of the first inductor & is equal to the inductance value of the second inductor pa. FIG. 7A is a no load, FIG. 7B is a half load, and FIG. 7B is a half load. 7C is a rated load, and the measurement results of the first inductor current iu, the second inductor current iu, the first switching voltage Vgs, and the crossover capacitor voltage Vc in each of FIGS. 7A to 7C can be measured. It is observed that the first inductor current IL1 is almost equal to the second inductor current Iu. Referring to FIG. 8A to FIG. In the embodiment, the inductance value of the first inductor a is greater than the inductance value of the first inductor A, FIG. 8A is no load, FIG. 8B is half load ' and FIG. 8C is full load', and FIG. 8A to FIG. The measurement result of the inductor current iLi, the second inductor current, the first switching voltage Vgsi, and the jump capacitor voltage Vc can be observed that the first inductor current Iu is greater than the second inductor current IL2. Referring to FIG. 9A to FIG. 9C, In the preferred embodiment, the inductance of the first inductor and the inductance are smaller than the inductance of the second inductor A, FIG. 9A is no load, FIG. 9B is half load, and FIG. 9C is full load' and each of FIGS. 9A to 9c has The first 201218603 inductor current 1L 丨, the second inductor current iLz, the first switching voltage 丨 and the crossover valley voltage VC. The first inductor current can be observed to be smaller than the second inductor current Iu. The high boost ratio converter 100 of the present invention is composed of the aforementioned components, and utilizes a control mode similar to the conventional boost converter and a simple design, thereby saving component costs, and thus the object of the present invention can be achieved. , which is only a preferred embodiment of the present invention The scope of the invention is not limited thereto, that is, the simple equivalent changes and modifications made in the scope of the invention and the description of the invention are still within the scope of the invention. 1 is a circuit diagram illustrating a preferred embodiment of a high step-up ratio converter of the present invention; FIG. 2 is a circuit diagram illustrating a control architecture of a preferred embodiment of the high step-up ratio converter of the present invention; FIG. Figure 3B is a circuit diagram illustrating current flow in two modes of operation of the first state of the preferred embodiment; Figures 4A through 4E are circuit diagrams illustrating current flow in five modes of operation of the second state of the preferred embodiment 5A to 5C are circuit diagrams illustrating current flow in a three-operation mode of a third state of the preferred embodiment; and FIGS. 6A to 6C are circuit diagrams illustrating respective operation modes of the fourth state of the preferred embodiment; Current flow direction; FIG. 7A to FIG. 7C are waveform diagrams illustrating the first and 12 201218603 of the preferred embodiment
第二電感值相等之無負載、半載及滿載的量測結果； 圖8A至圖8C是波形圖，說明本較佳實施例的第一電 感值大於第二電感值之無負載、半載及滿載的量測結果；及 圖9A至圖9C是波形圖，說明本較佳實施例的第一電 感值小於第二電感值之無負載、半載及滿載的量測結果。 13 201218603 【主要元件符號說明】 100 •高升壓 比轉換器 A ... …第一 順 向 導 通元件 101 •比較器 D2". …第二 順 向 導 通元件 102 •場可程式閘1 举列 久、 Dbi ·-— 極 體 103 •半橋式 閘驅動器 V··· ……第 一 電 感 104 •電壓驅 動器 L2 ……第 二 電 感 11 ' 21 ' 31 ' 41 ' 51 ' 61、 Μ, ' m2 ·脈波控制 訊號 71、 81 · •第一端 … ……第 一 開 關 元件 12、 22、 32、42 、52、 62 ' 込… ……第 二 開 關 元件 72 ' 82 . .第二端 R…. ……胃 載 Q.· •輸入電 容 κ ··· ...... ^'J 入 電 壓 ce.· •跨接電 容 Vfb · ......Θ 饋 控 制 訊號 C0 ·· •輸出電 容 v0 ··· ......iA. ¥m 出 電 壓 14The second inductance value is equal to the unloaded, half load and full load measurement results; FIG. 8A to FIG. 8C are waveform diagrams illustrating the no-load, half load and the first inductance value of the preferred embodiment being greater than the second inductance value. The measurement result of the full load; and FIG. 9A to FIG. 9C are waveform diagrams illustrating the measurement results of no load, half load and full load of the first inductance value of the preferred embodiment being smaller than the second inductance value. 13 201218603 [Description of main component symbols] 100 • High boost ratio converter A... First forward conduction element 101 • Comparator D2"....Second forward conduction element 102 • Field programmable gate 1 Long, Dbi ·-- pole body 103 • Half bridge gate driver V···...first inductor 104 •voltage driver L2 ...second inductor 11 ' 21 ' 31 ' 41 ' 51 ' 61, Μ, ' m2 Pulse wave control signals 71, 81 · • First end ... ... first switching element 12, 22, 32, 42 , 52, 62 ' 込 ... ... second switching element 72 ' 82 . . second end R... ...... stomach load Q.· • input capacitance κ ··· ...... ^'J input voltage ce.· • jumper capacitor Vfb · ...... 馈 feed control signal C0 ·· • output Capacitor v0 ··· ......iA. ¥m output voltage 14