200847133 九、發明說明： 【發明所屬之技術領域】 本發明之具體貫施例係關於音頻處理，更特定言之係關 於音頻訊號之分析。 【先前技術】 存在許多用於將一音頻訊號分為若干次頻帶並導出隨時 間變化之頻率相依振幅與相位特徵之解決方案。範例包括 視窗型快速傅立葉變換/反快速傅立葉變換（fft/ifft)系統 以及並聯若干組有限脈衝回應（F J R)濾波器組與無限脈衝 回應⑽)濾波器組。不㉟，此等習知解決方案全部遭受缺 陷0 不利地，視窗型FFT系統僅針對各頻帶提供一單一固定 '、寬…通$在底部處採用一高解析度選擇一從低頻率至 項率所知加之頻寬。例如，在工⑽以處，需要一具有 頻見之濾波器（組）。不過，此意味著在8kHz處使用 5〇HZ頻見，而在8kHz處一更寬頻寬（例如400Hz)可能更 恰當。因此，土卜莖多 等系、、先無法提供與人感覺相匹配之靈活 性0 視窗型FFT系访+ 在高頻率處W另一缺點係稀疏取樣視窗型F F T系統 、 不充足精細頻率解析度在應用修改（例如， 樂I雜Γ抑Γ)條件下可導致不可採用人工產物（例如，”音 躍大I ”）。、精由明顯減少視窗型框架間之重疊大小"FFT跳 工產物數（即’增加超取樣）可在某—程度上減少人 。遺憾的係，FFT系統之計算成本隨超取樣增加 127033.doc 200847133 而增加。同樣地，遽'波器組之FIR子類亦係計算昂貴，其 係由於各次頻帶中取樣脈衝回應之卷積（其可導致高潛 時)。例如’ -具有256個樣本之視窗之系統將需要脚 乘法及一 128個樣本之潛時，若該視窗為對稱的話。 IIR子類由於其遞迴性質而計算較不昂貴，不過僅採用 實值濾波H係數之實施方案在實現近完美重建方面存在困 難’尤其係修改次頻帶訊號的話。此外，需要用於各次頻 帶之相位與振幅補償以及時間對齊以便在輸出處產生一平 坦頻率回應。難以結合實值訊號執行相位補償，因為其丟 失用於以精細時間解析度直接計算振幅與相位之正交分 量。用以決定振幅與頻率之最普通方法係在各級輸出上應 用一 Hilbert變換。不過需要一額外用於在實值濾波器組中 計算Hilbert變換之計算步驟，且其係計算昂貴的。 因此，需要與現有系統相比計算較不昂貴的用於分析與 重建一音頻訊號同時提供低端對端潛時，以及用於時間頻 率解析度之必需自由度的系統及方法。 【發明内容】 本發明之具體實施例提供用於音頻訊號處理的系統及方 法。在範例性具體實施例中，使用一複值濾波器之渡波器 級聯來將一輸入音頻訊號分解為複數個次頻帶訊號。在一 具體實施例中，採用該濾波器級聯之一複值濾波器對一輸 入訊號進行濾、波以產生一第一已濾波訊號。從該輸入訊號 中減去該弟一已濾'波Λ號以導出一第一次頻帶訊號。接 著，藉由該濾波器級聯之下一複值濾波器處理該第一已渡 127033.doc 200847133 波訊號以產生下—已濾波訊號。重複程序直到利用該級聯 中之最後複值濾波器。在某些具體實施例中，㈣複值渡 波器係單極複值渡波器。200847133 IX. DESCRIPTION OF THE INVENTION: TECHNICAL FIELD OF THE INVENTION The specific embodiments of the present invention relate to audio processing, and more particularly to the analysis of audio signals. [Prior Art] There are a number of solutions for dividing an audio signal into sub-bands and deriving frequency-dependent amplitude and phase characteristics that change over time. Examples include a window-type fast Fourier transform/inverse fast Fourier transform (fft/ifft) system and parallel sets of finite impulse response (F J R) filter banks and infinite impulse response (10) filter banks. No. 35, these conventional solutions all suffer from defect 0. The windowed FFT system only provides a single fixed 'width for each frequency band', wide ... pass at the bottom with a high resolution to select a low frequency to the term rate Know the bandwidth. For example, in the workplace (10), a filter (group) with a frequency is required. However, this means that a 5 Hz HZ frequency is used at 8 kHz, and a wider bandwidth (e.g., 400 Hz) at 8 kHz may be more appropriate. Therefore, the soil stems are more than the first line, and can not provide the flexibility to match the human perception. 0 Window type FFT system access + At high frequencies, another disadvantage is the sparse sampling window type FFT system, insufficient fine frequency resolution. Under the condition of applying modification (for example, music I), artificial products (for example, "sound hop I") may not be used. It is possible to reduce the number of overlaps between window frames by significantly reducing the number of FFT products (ie, increasing the number of oversamplings) to some extent. Unfortunately, the computational cost of the FFT system increases with the increase in oversampling by 127033.doc 200847133. Similarly, the FIR subclass of the 遽's wave group is also computationally expensive due to the convolution of the sampled impulse responses in each sub-band (which can result in high latency). For example, a system with a window of 256 samples would require foot multiplication and a potential of 128 samples if the window is symmetrical. The IIR subclass is less expensive to calculate due to its recursive nature, but implementations that only use real-valued filtered H-factors have difficulties in achieving near-perfect reconstructions, especially when modifying sub-band signals. In addition, phase and amplitude compensation and time alignment for each band are required to produce a flat frequency response at the output. It is difficult to perform phase compensation in conjunction with real-valued signals because their loss is used to directly calculate the amplitude and phase orthogonal components with fine time resolution. The most common method for determining amplitude and frequency is to apply a Hilbert transform at each stage of the output. However, an additional computational step for computing the Hilbert transform in a real-valued filter bank is required and is computationally expensive. Therefore, there is a need for a less expensive system and method for analyzing and reconstructing an audio signal while providing low end-to-end latency, as well as the necessary degrees of freedom for time-frequency resolution, as compared to existing systems. SUMMARY OF THE INVENTION Embodiments of the present invention provide systems and methods for audio signal processing. In an exemplary embodiment, a wave cascade of complex value filters is used to decompose an input audio signal into a plurality of sub-band signals. In one embodiment, an input signal is filtered using a filter cascading complex value filter to generate a first filtered signal. The younger filtered 'wave number' is subtracted from the input signal to derive a first frequency band signal. Then, the first 127033.doc 200847133 wave signal is processed by the filter cascade to generate a down-filtered signal. Repeat the procedure until the last complex value filter in the cascade is utilized. In some embodiments, the (IV) complex-valued waver is a single-pole complex-valued waver.
一旦分解該輸入訊號，便可藉由一重建模組來處理該等 次頻帶訊號。該重建模組係經組態用以在該等次頻帶訊號 之或夕個上執行一相位對齊。該重建模組亦可經組態用 以在該等次頻帶訊號之—或多個上執行振幅補償。此外， 可藉由該重建餘在料次頻帶訊號之—《多個上執行一 時間延遲。對該等已補償及/或時間已延遲次頻帶訊號之 實數部分求和以產生一重建音頻訊號。 【貫施方式】 本《月之具體貝鈿例提供用於一音頻訊號之近完美重建 的系統及方法。該範例性I統利用—遞迴遽波器組來產生 正交輸出。在範例性具體實施例中，該濾波器組包含複數 個複值濾'波器。在其他具體實施例中，該濾、波器組包含複 數個單極複值濾波器。 —參考圖1 ’顯示一範例性系統100,在該系統1〇〇中可實 打本發明之具體實施例。㈣統丨⑽可為能夠處理音頻訊 號之任何裝置’例如但不受限於，蜂巢式電話、助聽器、 揚聲器電話、電腦或任何其他裝置。㈣統⑽亦可表示 此等裝置之任何裝置之一音頻路徑。 系統100包含一音頻處理引擎1()2…音頻來源⑽、— 調節模組1〇6及一音頻槽108。其他與音頻訊號之重 之組件可提供於系統1〇〇中。此外，雖然系統剛兒明—從 127033.doc 200847133 圖1之各組件至下一者之資料邏輯行進，但替代具體實施 例可包含系統100之各種經由一或多個匯流排或其他元件 而耦合之組件。 範例性音頻處理引擎102處理經由音頻來源104而輸入之 輸入（音頻）5孔號。在一具體實施例中，音頻處理引擎102包 含儲存於一裝置上之軟體，該裝置係藉由一通用處理器來 操作。在各種具體實施例中，音頻處理引擎1〇2包含一分 析濾波器組模組110、一修改模組丨丨2及一重建模組丨丨4。 應注思’在音頻處理引擎1 〇2中可提供更多、更少或功能 等效模組。例如，可將模組110至114之一或多個組合為很 少模組且仍提供相同功能性。 音頻來源104包含接收輸入（音頻）訊號之任何裝置。在 某些具體實施例中，音頻來源1〇4係經組態用以接收類比 音頻訊號。在一範例中，音頻來源1〇4係一耦合至類比至 數位（A/D)轉換器之麥克風。該麥克風係經組態用以接收 類比音頻訊號，同時該A/D轉換器取樣該等類比音頻訊號 以將該等類比音頻訊號轉換為適於進一步處理之數位音頻 訊號。在其他範例中，該音頻來源1〇4係經組態用以接收 類比音頻訊號，同時該調節模組1〇6包含該A/D轉換器。在 替代具體實施例中，音頻來源1〇4係經組態用以接收數位 音頻訊號。例如，音頻來源1〇4係一能夠讀取儲存於硬碟 或其他媒體形式上之音頻訊號資料之磁碟裝置。其他具體 實施例可利用其他形式之音頻訊號感測/捕獲裝置。 調節模組106預處理輸入訊號（即，不需要輪入訊號之分 127033.doc 200847133 解之任何處理）。在一具體實施例中，調節模組丨〇6包含一 自動增盈控制。調節模組106亦可執行誤差校正與雜訊濾 波。調節模組106可包含用於預處理音頻訊號之其他組件 與功能。 分析濾波器組模組11 〇將接收到之輸入訊號分解為複數 個次頻帶訊號。在某些具體實施例中，可直接使用來自分 析濾波為組模組11 〇之輸出（例如，用於視覺顯示）。將結合 圖2更詳細地論述分析濾波器組模組110。在範例性具體實 施例中，各次頻帶訊號表示一頻率分量。 I巳例性修改模組112透過個別分析路徑從分析濾波器組 杈組11 0接收各次頻帶訊號。修改模組i丨2可基於個別分析 路徑修改/調整該等次頻帶訊號。在一範例中，修改模組 112對透過特定分析路徑接收到之次頻帶訊號中之雜訊進Once the input signal is resolved, the sub-band signals can be processed by a reconstruction module. The reconstruction module is configured to perform a phase alignment on or after the sub-band signals. The reconstruction module can also be configured to perform amplitude compensation on one or more of the sub-band signals. In addition, a time delay can be performed on the plurality of sub-band signals by the reconstruction. The real part of the compensated and/or time delayed subband signals is summed to produce a reconstructed audio signal. [Complex method] This month's specific shell example provides a system and method for near-perfect reconstruction of an audio signal. The exemplary I system utilizes a recursive chopper group to generate quadrature outputs. In an exemplary embodiment, the filter bank includes a plurality of complex value filters. In other embodiments, the filter bank includes a plurality of unipolar complex filters. - An exemplary system 100 is shown with reference to Figure 1 in which a specific embodiment of the present invention can be implemented. (d) Reconciliation (10) may be any device capable of processing audio signals' such as, but not limited to, a cellular telephone, a hearing aid, a speakerphone, a computer, or any other device. (d) System (10) may also indicate an audio path to any of these devices. The system 100 includes an audio processing engine 1() 2...audio source (10), an adjustment module 1〇6, and an audio slot 108. Other components that are important to the audio signal can be provided in the system. In addition, although the system is just described as being logically advanced from the various components of FIG. 1 to the next, alternative embodiments may include various couplings of system 100 via one or more busbars or other components. The components. The example audio processing engine 102 processes the input (audio) 5 hole number entered via the audio source 104. In one embodiment, audio processing engine 102 includes software stored on a device that is operated by a general purpose processor. In various embodiments, the audio processing engine 102 includes an analysis filter bank module 110, a modification module 丨丨2, and a reconstruction module 丨丨4. It should be noted that more, fewer or functional equivalent modules are available in the Audio Processing Engine 1 〇2. For example, one or more of modules 110-114 can be combined into very few modules and still provide the same functionality. Audio source 104 contains any device that receives input (audio) signals. In some embodiments, the audio source 1〇4 is configured to receive analog audio signals. In one example, the audio source 1〇4 is coupled to a microphone of an analog to digital (A/D) converter. The microphone is configured to receive an analog audio signal, and the A/D converter samples the analog audio signals to convert the analog audio signals into digital audio signals suitable for further processing. In other examples, the audio source 1〇4 is configured to receive an analog audio signal, and the adjustment module 1〇6 includes the A/D converter. In an alternate embodiment, the audio source 1〇4 is configured to receive digital audio signals. For example, the audio source 1 is a disk device capable of reading audio signal data stored on a hard disk or other media form. Other embodiments may utilize other forms of audio signal sensing/capturing devices. The adjustment module 106 preprocesses the input signal (i.e., does not require any rounding of the signal 127033.doc 200847133 for any processing). In one embodiment, the adjustment module 丨〇6 includes an automatic gain control. The adjustment module 106 can also perform error correction and noise filtering. Adjustment module 106 can include other components and functions for pre-processing audio signals. The analysis filter bank module 11 decomposes the received input signal into a plurality of sub-band signals. In some embodiments, the output from the analysis filter to the group module 11 can be used directly (e.g., for visual display). The analysis filter bank module 110 will be discussed in greater detail in conjunction with FIG. In an exemplary embodiment, each sub-band signal represents a frequency component. The exemplary modification module 112 receives the sub-band signals from the analysis filter bank group 110 through individual analysis paths. The modification module i丨2 can modify/adjust the sub-band signals based on the individual analysis paths. In one example, the modification module 112 enters the noise in the sub-band signal received through the specific analysis path.
細地論述重建模組丨丨4。 行濾波。在另一範例中， ，一從特定分析路徑接收到之次頻Discuss the reconstruction module 丨丨4 in detail. Line filtering. In another example, a secondary frequency received from a particular analysis path
訊號之不可採用部分。The unusable part of the signal.
某些具體實施例中， 出董建音頻訊號之任何裝置。在 頻槽108輸出一類比重建音頻訊 127033.doc 200847133 號。例如，音頻槽108可包含一數 一揚殊"數位至類比(D/A)轉換器與 :㈣此範例中，該D / A轉換器係經組態用以接收 ::處理引擎102之重建音頻訊號並將其轉換為類比 重建。曰頻訊號。該揚聲器可接著接收並輸出該類比重建音 頻訊號。音頻槽108可包含任何類比輸出裝置，其包括： 不受限於頭戴式耳機、耳機或助聽器。或者，音頻槽⑽ 包含該D/A轉換器及—經組態用以耗合至外部音頻裝即列In some embodiments, any device of the Dong Jian audio signal is output. An analogy of the reconstructed audio signal 127033.doc 200847133 is output in the frequency bin 108. For example, the audio slot 108 can include a number of "digital" analog-to-digital (D/A) converters with: (d) In this example, the D/A converter is configured to receive:: processing engine 102 Rebuild the audio signal and convert it to an analog reconstruction.曰 Frequency signal. The speaker can then receive and output the analog to reconstruct the audio signal. The audio slot 108 can include any analog output device including: not limited to a headset, earphone, or hearing aid. Alternatively, the audio slot (10) contains the D/A converter and is configured to be used to external audio packs
如，揚聲器、頭戴式耳機、耳機、助聽器)之音頻輸出 璋。 在替代具體實施例中，音頻槽108輸出—數位重建音頻 訊號。在另一範例中，音頻槽1〇8係—磁碟裝置，其中可 將該重建音頻訊號儲存於一硬碟或其他媒體上。在替代具 體實施例中，音頻槽108係可選的且音頻處理引擎1〇2產2 該重建音頻訊號用於進一步處理（圖1未顯示）。 現在參考圖2，更詳細顯示範例性分析濾波器組模組 110。在範例性具體實施例中，分析據波器組模組11〇接收 一輸入訊號202，且透過一系列濾波器2〇4處理該輸入訊號 202以產生複數個次頻帶訊號或分量（例如1>1至？6)。許多 濾波器204可包含該分析濾波器組模組11()。在範例性具體 實施例中’濾波器204係複值濾波器。在其他具體實施例 中，濾波器204係一階濾波器（例如單極複值）。圖3進一步 論述濾波器204。 在範例性具體實施例中，將濾波器204組織成一渡波器 級聯，藉此一濾波器204之一輸出變為該級聯中下一渡波 127033.doc -11 - 200847133 器204中的一輸入。因此，輸入訊號2〇2係饋送至一第一濾 波器2(Ma。藉由一第一計算節點2〇6a從輸入訊號2〇2中減 去第一濾波器204a之一輸出訊號ρι以產生一輸出D1。輸出 D1表不進入第一濾波器2〇4a中之訊號與第一濾波器2〇乜後 之訊號間之差訊號。 在替代具體實施例中，不使用計算節點2〇6來決定次頻 帶訊號就可實現濾波器級聯之益處。即，舉例而言，可直For example, audio output from speakers, headphones, headphones, hearing aids. In an alternate embodiment, the audio slot 108 outputs a digitally reconstructed audio signal. In another example, the audio slot is a disk device in which the reconstructed audio signal can be stored on a hard disk or other medium. In an alternate embodiment, the audio slot 108 is optional and the audio processing engine 1 2 produces the reconstructed audio signal for further processing (not shown in Figure 1). Referring now to Figure 2, an exemplary analysis filter bank module 110 is shown in greater detail. In an exemplary embodiment, the analysis packet module 11 receives an input signal 202 and processes the input signal 202 through a series of filters 2〇4 to generate a plurality of sub-band signals or components (eg, 1> 1 to 6). A number of filters 204 can include the analysis filter bank module 11(). In an exemplary embodiment, filter 204 is a complex valued filter. In other embodiments, filter 204 is a first order filter (e.g., unipolar complex). The filter 204 is further discussed in FIG. In an exemplary embodiment, filter 204 is organized into a waver cascade, whereby one of the outputs of one filter 204 becomes an input in the next wave 127033.doc -11 - 200847 133 of the cascade. . Therefore, the input signal 2〇2 is fed to a first filter 2 (Ma. The first computing node 2〇6a subtracts one of the output signals ρι of the first filter 204a from the input signal 2〇2 to generate An output D1. The output D1 represents a difference signal between the signal in the first filter 2〇4a and the signal after the first filter 2. In an alternative embodiment, the computing node 2〇6 is not used. The benefits of filter cascading can be achieved by determining the sub-band signal. That is, for example, straight
接使用各濾波為204之輸出來表示輸出處或欲顯示之訊號 之能量。 由於分析濾波器組模組110之級聯結構，輸出訊號卩丨現 在係一進入級聯中下一濾波器20仆中之輸入訊號。和與第 一濾波器204a相關聯之程序類似，藉由下一計算節點2〇計 從輸入訊號pi中減去下-遽波器2_之—輸出（即p2)以獲 得下-頻帶或頻道（即輸出的）。此下_頻道強調本渡波^ 204b與先刖濾波器204a之截止頻率間之頻率。此程序透過 該級聯之濾波器204之其餘者繼續。 ^ 在一具體實施例中，將該級聯中之滹、、古 τ <應渡杰集分離為八個 -組。因此可在不同八個一組中之對應據波器(位於一類 似位置）間共用濾波器參數與係數。在 仕果國專利申請案序 號09/534,682中詳細說明此程序。 、 係單極複值據波琴。 例如’濾波器2G4可包含採用複值運作之—階數位或類比 濾波器。全體地，濾波器2〇4之輸出表 ^ v 码衣不音頻訊號之次頻 V分量。由於計算節點200，各輸出# ^ 、 囬表不一次頻帶，且所 127033.doc -12- 200847133 有屮 扣之和表示整個輸入訊號202。由於級聯濾波器2〇4係 一階，痛 λ工 々々 以什算費用可比級聯濾波器204為二階或二階以 /之。十异費用少很多。此外，藉由改變一階濾波器可 、易仏改彳文音頻訊號所擷取之各次頻帶。在其他具體實 施例中，濾波器204係複值濾波器且不必為單極。 在其他具體實施例中，修改模組112(圖1}可在必要時處 理汁异節點206之輸出。例如，修改模組112可半波整流已 ，波次頻帶。此外，可調整輸出之增益以壓縮或擴展一動 “範圍。在某些具體實施例中，藉由另一濾波器2〇4鏈/級 聯加以處理之前，可降低取樣任何濾波器2〇4之輸出。 在範例性具體實施例中，濾波器2〇4係具有經設計用以 產生所需頻道解析度之截止頻率之無限脈衝回應（IIR)濾波 :。濾波器204可在複數音頻訊號上採用各種係數執行連 續Hilbert變換以便在特定次頻帶内抑制或輸出訊號。 圖3係一方塊圖，其解說本發明之一範例性具體實施例 中之此訊號流。傳遞濾遣器204之輸出^ι[η]^_[η]分 別用作級聯中下一濾波器2〇4之輸入與 X1 mag[η+1 ]。項η識別欲從音頻訊號擷取之次頻帶，其中 "η”係假定為一整數。由於„R濾波器2〇4係遞迴式，所以濾 波器之輸出可基於先前輸出而改變。可在訊號之實分量之 求和之後、之前或期間對輸入訊號之虛分量（例如Ximag[n]) 求和。在一具體實施例中，可藉由複數一階差分方程式 來說明濾波器 2〇4，其中 b =r_Z*exp(i*theta一P)且 a=_r_p*exp(i*theta_p)*、"係一樣本 127033.doc -13- 200847133 索引。 在本具體實施例中’ MgM係一增益因數。應注意，可在 不影響極與零位置之任何地方應用該增益因數。在替代具 體實施例中，已將音頻訊號分解為次頻帶訊號之後，可藉 由修改模組112(圖1)來應用該增益。 現在參考圖4，針對一音頻訊號之每六個次頻帶顯示 一範例性量值與相位之對數顯示。量值與相位資訊係基於 來自分析濾波器組模組^(^圖丨）之輸出。即，圖4所示振幅 係來自計算節點206(圖2)之輸出（即輸出〇1至〇6)。在本範 例中’为析濾波器組模組1 i 〇正在結合一從8 至此沿之 頻率範圍之23 5個次頻帶以一 16kHz取樣速率運作。此分析 濾、波器組模組110之端對端潛時係17 3 ms。 在某些具體實施例中，需要在高頻率處具有一寬頻率回 應且在低頻率處具有一窄頻率回應。因為本發明之具體實 2係可適於許多音頻來源1G4(圖υ，所以可使用不同頻 ^之不同頻寬。因此，可獲得高頻率處具有寬頻寬之快 古回應及低頻率處具有窄、短頻寬之緩慢回應 有相對較低潛時（例如12ms)更適於人耳之回應。 - 之考圖5，顯示—分析耳蝎設計之每級量值與相位 至P6)&/ °圖5所示振幅係圖2之渡波器204之輸出（例如P1 〇鮮說依據本發明 作。為〜 一〜丹腹員她例之重建模組114之 靶例性具體實施例中，對齊各次 執行振幅補待 貝f衹唬之相位 貝、移除各次頻帶訊號之複數部分、然後在 127033.doc -14- 200847133 要時藉由使各次頻帶訊號延遲來對齊時間以獲得一平坦重 建頻譜並減少脈衝回應分散。 口為濾波恭使用複合訊號（例如實數與虛數部分），所以 可針對任何樣本導出相位。此外，亦可藉由A== 來計算振幅。因此，數學上使得音頻訊號 重建更容易。作為此方法之結果，任何樣本之振幅與相 位可很容易用於進一步處理（即至修改模組112(圖丨））。 ( 由於次頻帶訊號之脈衝回應可具有不同群組延遲，所以 、 僅僅對分析濾波器組模組110(圖1)之輸出求和可能不會提 供音頻訊號之準確重建。因此，可使一次頻帶之輸出延遲 該次頻帶之脈衝回應峰值時間以便所有次頻帶濾波器在同 一時間瞬時具有其脈衝回應包絡最大值。 在脈衝回應波形最大值之時間比所需群組延遲更遲的一 具體實施例中，將濾波器輸出與一複數常數相乘以便脈衝 回應之實數部分在所需群組延遲處具有一局部最大值。 如圖所示，重建模組114從修改模組112(圖1)接收次頻帶 訊號602(例如SG、Sj Sm)。接著將係數604(例如知、^及 am)應用於次頻帶訊號。係數包含一固定複數因數（即包含 一實數與虛數部分）。或者，可在分析濾波器組模組丨丨〇内 將係數604應用於次頻帶訊號。將係數應用於各次頻帶訊 號會對齊次頻帶訊號之相位並補償各振幅。在範例性具體 實施例中，該等係數係預定的。係數之應用之後，藉由一 實值模組606(即Re{ })丟棄虛數部分。 接著藉由一延遲Z·1 608使次頻帶訊號之各實數部分延 127033.doc -15- 200847133 遲。此延遲提供交叉次頻帶對齊。在一具體實施例中，延 遲Ζ·ι 608提供一分接頭延遲。該延遲之後，在一求和節點 610中對個別次頻帶訊號求和，導致一值。接著將部分重 建訊號載送至下一求和節點61〇中並應用於下一已延遲次 頻帶訊號。該程序繼續直到對所有次頻帶訊號求和，導致 重建音頻訊號。因此重建音頻訊號係適於音頻槽1 圖 1)。儘管顯示延遲Ζ·ι 608係在對次頻帶訊號求和之後描 述，不過重建模組114之運作順序可互換。 圖7解說一基於圖4與圖5之範例之重建曲線圖。藉由重 建模組114(圖1)所執行之相位對齊、振幅補償及針對交又 次頻帶對齊之延遲之後，藉由組合各濾波器2〇4(圖2)之輪 出獲得重建（即重建音頻訊號）。因此，該重建曲線圖相= 較平坦。 現在參考圖8，提供一範例性用於音頻訊號處理之方法 之流程圖800。在步驟802中，將一音頻訊號分解為若干次 頻帶訊號。在範例性具體實施例中，#由分析濾波器組模 組11〇(圖1)處理該音頻訊號。該處理包含透過一濾波器 2〇4(圖2)級聯對該音頻訊號進行濾波，各濾波器2〇4之輸出 導致個別輸出206處之一次頻帶訊號。在一具體實施例 中，濾波器204係複值濾波器。在另一具體實施例中，濾 波益2 0 4係單極複值濾波器。 次頻帶分解之後，在步驟804中透過修改模組112(圖丨）處 理次頻帶訊號。在範例性具體實施例中，修改模組112(圖 υ調整輸出之增益以壓縮或擴展一動態範圍。在某些具體 127033.doc -16- 200847133 實施例中，修改模組m可抑制不可採用次頻帶訊號。 :步驟_中，一重建模組114(圖υ接著在各次頻帶訊號 執仃相位與振幅補償。在一具體實施例中，藉由將一 數係數應用於該次頻帶訊號執行相位與振幅補償。之後在 ν驟808中將已補償次頻帶訊號之虛數部分丢棄。在其他 具體實施例中，保留已補償次頻帶訊號之虛數部分。 使用已補償次頻帶訊號之實數部分，在步驟81〇中針對 :叉次頻帶對齊使次頻帶訊號延遲。在一具體實施例中， 藉由利用重建模組114中之-延遲線獲得該延遲。 在步驟812中，對已延遲次頻帶訊號求和以獲得一重建 汛號。在範例性具體實施例中，各次頻帶訊號/片段表示 一頻率。 以上已參考範例性具體實施例說明本發明之具體實施 例。熟知此項技術者應明白，在不背離本發明之更廣泛範 嚀下，可進行各種修改且可使用其他具體實施例。因此， 期望依據該等範例性具體實施例之此等及其他變化為本發 明所涵蓋。 Χ 【圖式簡單說明】 圖1係一採用本發明之具體實施例之系統的範例性方塊 圖； 圖2係本發明之一範例性具體實施例中之分析濾波器組 模組之範例性方塊圖； 圖3解說依據一具體實施例的該分析濾波器組模組之一 濾波器； 127033.doc •17- 200847133 圖4針對每六個（6)次頻帶解說次頻帶轉移函數之量值與 相位之對數顯示； 圖5針對每六（6)級解說累積濾波器轉移函數之量值與相 位之對數顯示； 圖6解說該範例性重建模組之運作； 及The output of each filter is used to represent the energy at the output or the signal to be displayed. Due to the cascading structure of the analysis filter bank module 110, the output signal is now an input signal entering the servant of the next filter 20 in the cascade. Similar to the procedure associated with the first filter 204a, the next compute node 2 subtracts the output of the down-chopper 2_ from the input signal pi (ie, p2) to obtain the down-band or channel. (ie output). This lower channel emphasizes the frequency between the local wave 204b and the cutoff frequency of the pre-filter 204a. This program continues through the rest of the cascaded filter 204. ^ In a specific embodiment, the 级, 古 τ < 宜渡杰集 in the cascading is separated into eight groups. Therefore, filter parameters and coefficients can be shared between corresponding data filters (located in a similar position) in different groups of eight. This procedure is described in detail in the official patent application Serial No. 09/534,682. , is a unipolar complex value of the Boqin. For example, 'filter 2G4 can include an order-order or analog filter that operates with complex values. Overall, the output table of the filter 2〇4 is not the sub-frequency V component of the audio signal. Due to the calculation node 200, each output #^, the return table does not have a frequency band, and the sum of 127033.doc -12-200847133 has the entire input signal 202. Since the cascading filter 2 〇 4 is a first order, the pain λ 々々 is comparable to the cascading filter 204 as a second or second order. Ten different costs are much less. In addition, each frequency band captured by the audio signal can be easily changed by changing the first-order filter. In other embodiments, filter 204 is a complex valued filter and does not have to be a single pole. In other embodiments, the modification module 112 (FIG. 1) can process the output of the juice node 206 as necessary. For example, the modification module 112 can half-wave rectify the wave band. In addition, the gain of the output can be adjusted. To compress or extend the motion "range. In some embodiments, the output of any filter 2〇4 can be reduced before being processed by another filter 2〇4 chain/cascade. In an exemplary implementation In the example, filter 2〇4 has infinite impulse response (IIR) filtering designed to produce a cutoff frequency of the desired channel resolution: Filter 204 can perform continuous Hilbert transform using various coefficients on the complex audio signal so that The signal is suppressed or outputted in a particular sub-band. Figure 3 is a block diagram illustrating the signal stream in an exemplary embodiment of the present invention. The output of the pass filter 204 is ^ι[η]^_[η Used as the input of the next filter 2〇4 in the cascade and X1 mag[η+1 ]. The item η identifies the sub-band to be extracted from the audio signal, where "n" is assumed to be an integer. „R filter 2〇4 is recursive, so filter The output of the device can be changed based on the previous output. The imaginary component of the input signal (eg, Ximag[[]] can be summed after, before, or during the summation of the real components of the signal. In a particular embodiment, The complex first-order difference equation is used to describe the filter 2〇4, where b = r_Z*exp(i*theta_P) and a=_r_p*exp(i*theta_p)*, " is the same as this 127033.doc -13 - 200847133 Index. In the present embodiment 'MgM is a gain factor. It should be noted that this gain factor can be applied anywhere without affecting the pole and zero positions. In an alternative embodiment, the audio signal has been decomposed into times. After the band signal, the gain can be applied by modifying module 112 (FIG. 1). Referring now to Figure 4, an exemplary magnitude and phase logarithmic display is displayed for every six sub-bands of an audio signal. The phase information is based on the output from the analysis filter bank module ^ (^ 图). That is, the amplitude shown in Figure 4 is from the output of the compute node 206 (Figure 2) (ie, outputs 〇1 to 〇6). In the example, 'for the filter bank module 1 i 〇 is combining one from 8 to this edge The frequency range of 23 sub-bands operates at a sampling rate of 16 kHz. The end-to-end latency of the analysis filter and wave bank module 110 is 17 3 ms. In some embodiments, at high frequencies It has a wide frequency response and a narrow frequency response at low frequencies. Since the specific embodiment of the present invention can be adapted to many audio sources 1G4 (Fig. 所以, different bandwidths of different frequencies can be used. The fast response with wide bandwidth at high frequencies and the slow response with narrow and short bandwidth at low frequencies have relatively low latency (eg 12ms) and are more suitable for human ear response. - Figure 5, showing - analyzing the magnitude and phase of each level of the deaf design to P6) & / ° The amplitude shown in Figure 5 is the output of the waver 204 of Figure 2 (eg, P1 is said to be in accordance with the present invention) In the specific example of the reconstruction module 114 of the example of the ～1 丹 腹 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她 她At 127033.doc -14- 200847133, the time is adjusted by delaying the signal of each sub-band to obtain a flat reconstructed spectrum and reduce the impulse response dispersion. The port uses a composite signal (such as real and imaginary parts) for filtering. The phase is derived for any sample. In addition, the amplitude can also be calculated by A ==. Therefore, the audio signal reconstruction is mathematically made easier. As a result of this method, the amplitude and phase of any sample can be easily used for further processing ( That is, to modify the module 112 (Fig. )). (Because the impulse response of the sub-band signal can have different group delays, only the summation of the output of the analysis filter bank module 110 (Fig. 1) may not be mentioned. Accurate reconstruction of the audio signal. Therefore, the output of the primary band can be delayed by the pulse response peak time of the sub-band so that all sub-band filters have their impulse response envelope maximum at the same time instant. In a specific embodiment where the desired group delay is later, the filter output is multiplied by a complex constant such that the real portion of the impulse response has a local maximum at the desired group delay. As shown, re-modeling Group 114 receives subband signal 602 (e.g., SG, Sj Sm) from modification module 112 (Fig. 1). Coefficients 604 (e.g., known, ^, and am) are applied to the subband signal. The coefficients include a fixed complex factor (i.e., Include a real and imaginary part.) Alternatively, the coefficient 604 can be applied to the sub-band signal in the analysis filter bank module. The application of the coefficient to each sub-band signal will align the phase of the sub-band signal and compensate for the amplitude. In an exemplary embodiment, the coefficients are predetermined. After the application of the coefficients, the imaginary part is discarded by a real-valued module 606 (ie, Re{ }). The real portion of the sub-band signal is delayed by 127033.doc -15-200847133 by a delay Z·1 608. This delay provides cross-subband alignment. In one embodiment, the delay 提供·ι 608 provides a tap delay. After the delay, the individual sub-band signals are summed in a summing node 610, resulting in a value. The partial reconstruction signal is then carried to the next summing node 61 and applied to the next delayed sub-band signal. The program continues until all sub-band signals are summed, resulting in the reconstruction of the audio signal. Therefore, the reconstructed audio signal is adapted to the audio slot 1 Figure 1). Although the display delay Ζ·ι 608 is described after summing the sub-band signals, However, the order of operation of the reconstruction module 114 is interchangeable. FIG. 7 illustrates a reconstruction graph based on the examples of FIGS. 4 and 5. After the phase alignment, amplitude compensation, and delay for the sub-band alignment performed by the reconstruction module 114 (FIG. 1), reconstruction is achieved by combining the rotations of the filters 2〇4 (FIG. 2) (ie, reconstruction). Audio signal). Therefore, the reconstructed graph phase = flatter. Referring now to Figure 8, a flow chart 800 of an exemplary method for audio signal processing is provided. In step 802, an audio signal is decomposed into a number of sub-band signals. In an exemplary embodiment, # is processed by the analysis filter bank module 11 (Fig. 1). The process includes filtering the audio signal through a cascade of filters 2〇4 (Fig. 2), and the output of each filter 2〇4 results in a primary band signal at the individual output 206. In a specific embodiment, filter 204 is a complex valued filter. In another embodiment, the filter benefit is a unipolar complex value filter. After the sub-band decomposition, the sub-band signal is processed by the modification module 112 (Fig. 804) in step 804. In an exemplary embodiment, the module 112 is modified (the gain of the output is adjusted to compress or expand a dynamic range. In some embodiments 127033.doc -16-200847133, the modified module m can be suppressed from being adopted. Sub-band signal. In step _, a reconstruction module 114 (Fig. 仃 then performs phase and amplitude compensation in each sub-band signal. In a specific embodiment, by applying a coefficient to the sub-band signal Phase and amplitude compensation. The imaginary part of the compensated sub-band signal is then discarded in step 808. In other embodiments, the imaginary part of the compensated sub-band signal is retained. Using the real part of the compensated sub-band signal, In step 81A, the sub-band signal is delayed for the sub-band alignment. In a specific embodiment, the delay is obtained by using a delay line in the reconstruction module 114. In step 812, the delayed sub-band is used. Signal summation to obtain a reconstruction nickname. In an exemplary embodiment, each sub-band signal/segment represents a frequency. The invention has been described above with reference to exemplary embodiments. The embodiments are well known to those skilled in the art, and various modifications may be made and other specific embodiments may be employed without departing from the scope of the invention. And other variations are encompassed by the present invention. Χ [Simplified illustration of the drawings] FIG. 1 is an exemplary block diagram of a system embodying a specific embodiment of the present invention; FIG. 2 is an analysis of an exemplary embodiment of the present invention. An exemplary block diagram of a filter bank module; FIG. 3 illustrates a filter of the analysis filter bank module in accordance with an embodiment; 127033.doc • 17- 200847133 FIG. 4 for every six (6) sub-bands Explain the logarithm of the magnitude and phase of the subband transfer function; Figure 5 illustrates the logarithm of the magnitude and phase of the cumulative filter transfer function for every six (6) stages; Figure 6 illustrates the operation of the exemplary reconstruction module; and
圖7解說該音頻訊號之一範例性重建之曲線圖表示 圖8係一範例性用於重建音頻訊號之方法之流程圖 【主要元件符號說明】 100 系統 102 104 106 108 110 112 114 204 204a 204b 206 206a 206b 606 610 音頻處理引擎 音頻來源 調節模組 音頻槽 分析濾波器組模組 修改模組 重建模組 濾波器 第一濾波器 下一濾波器 計算節點/輪出 第一計算節點 丁一計算節點 實值模組 求和節點 127033.doc -18-7 is a flow chart showing an exemplary reconstruction of the audio signal. FIG. 8 is a flow chart of an exemplary method for reconstructing an audio signal. [Main element symbol description] 100 System 102 104 106 108 110 112 114 204 204a 204b 206 206a 206b 606 610 Audio Processing Engine Audio Source Adjustment Module Audio Slot Analysis Filter Bank Module Modification Module Reconstruction Module Filter First Filter Next Filter Computation Node/Turn Out First Computation Node Ding One Computation Node Value module summation node 127033.doc -18-