JPS6336474Y2 - - Google Patents
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- Publication number
- JPS6336474Y2 JPS6336474Y2 JP1980048353U JP4835380U JPS6336474Y2 JP S6336474 Y2 JPS6336474 Y2 JP S6336474Y2 JP 1980048353 U JP1980048353 U JP 1980048353U JP 4835380 U JP4835380 U JP 4835380U JP S6336474 Y2 JPS6336474 Y2 JP S6336474Y2
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- level
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- input terminal
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- Prior art date
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- 238000001514 detection method Methods 0.000 claims description 31
- 239000012530 fluid Substances 0.000 description 7
- 238000010586 diagram Methods 0.000 description 6
- 239000003990 capacitor Substances 0.000 description 5
- 230000000630 rising effect Effects 0.000 description 5
- 238000005259 measurement Methods 0.000 description 4
- 230000003111 delayed effect Effects 0.000 description 3
- 230000007423 decrease Effects 0.000 description 2
- 230000000694 effects Effects 0.000 description 2
- 241000251468 Actinopterygii Species 0.000 description 1
- 230000005669 field effect Effects 0.000 description 1
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- Measuring Volume Flow (AREA)
- Arrangements For Transmission Of Measured Signals (AREA)
Description
【考案の詳細な説明】
この考案は例えば超音波流量計における受信パ
ルスのように時間と共に振幅が変化する交流受信
波を検出する信号検出回路に関する。[Detailed Description of the Invention] This invention relates to a signal detection circuit that detects an alternating current received wave whose amplitude changes with time, such as a received pulse in an ultrasonic flowmeter.
超音波流量計においては第１図に示すようにパ
イプ１１内に流体が満管状態で流れ、そのパイプ
１１内の流体の流れ方向にずらされ、かつパイプ
１１の互に反対側に送受波器１２及び１３が設け
られ、それ等送受波器間をパイプ１１内の流体を
通つて超音波パルスが伝播する時間よりその流体
の流量を測定する。即ち今パイプ１１の内径を
Ｄ、超音波送受波器１２及び１３を結ぶ直線とパ
イプ１１の直径とのなす角度をθ、パイプ１１内
を流れる流体の流速をｖ、超音波パルスの音速を
ｃとすると、送受波器１２により送受波器１３へ
の流体の流れに逆らう方向における超音波パルス
の伝播時間t1は
t1＝Ｄ／cosθ／ｃ−ｖ sinθ
となる。逆に超音波パルスが送受波器１３から送
受波器１２に伝播する伝播時間、即ち流れの順方
向に伝播する伝播時間t2は
t2＝Ｄ／cosθ／ｃ＋ｖ sinθ
となる。この送受波器１２，１３間の超音波パル
スの伝播時間の流れに逆らう方向と順方向とのそ
れぞれの逆数の差は
△ｆ＝１／t2−１／t1＝sin2θ／Ｄｖ
となり、流速ｖに比例した出力を得ることができ
る。従つて一方の超音波送受波器より送波したパ
ルスを他方の超音波送受波器で受信する迄の時間
t1及びt2を測定し、先の△ｆを演算すれば流量が
測定できる。この場合の超音波パルスの受信波形
は例えば第２図に示すように時間と共にだんだん
振幅が大きくなつた後に再び小さくなるような波
形であり、超音波パルスの到達時点を正しく検出
できるかどうかによつて測定精度が大きく左右さ
れる。 In an ultrasonic flowmeter, as shown in FIG. 1, fluid flows in a pipe 11 in a full state, and the transducer/receiver is shifted in the flow direction of the fluid in the pipe 11, and transducers are installed on opposite sides of the pipe 11. 12 and 13 are provided, and the flow rate of the fluid is measured from the time it takes for an ultrasonic pulse to propagate through the fluid in the pipe 11 between the transducers. That is, now the inner diameter of the pipe 11 is D, the angle between the straight line connecting the ultrasonic transducers 12 and 13 and the diameter of the pipe 11 is θ, the flow velocity of the fluid flowing inside the pipe 11 is v, and the sound speed of the ultrasonic pulse is c. Then, the propagation time t 1 of the ultrasonic pulse in the direction opposite to the flow of fluid from the transducer 12 to the transducer 13 is t 1 =D/cos θ/c−v sin θ. Conversely, the propagation time t 2 for the ultrasonic pulse to propagate from the transducer 13 to the transducer 12, ie, the propagation time t 2 for the ultrasonic pulse to propagate in the forward direction of the flow, is t 2 =D/cos θ/c+v sin θ. The difference in the reciprocals of the propagation time of the ultrasonic pulse between the transducers 12 and 13 in the direction opposite to the flow and in the forward direction is △f=1/t 2 -1/t 1 = sin2θ/Dv, and the flow velocity An output proportional to v can be obtained. Therefore, the time it takes for a pulse transmitted from one ultrasonic transducer to be received by the other ultrasonic transducer
The flow rate can be measured by measuring t 1 and t 2 and calculating the aforementioned Δf. In this case, the received waveform of the ultrasonic pulse is a waveform in which the amplitude gradually increases over time and then decreases again as shown in Figure 2, and it depends on whether the arrival point of the ultrasonic pulse can be detected correctly. This greatly affects measurement accuracy.
従来においてはその受信波のレベルが一定の検
出レベルl1を越えた時にその受信波の検出時点と
していた。しかしこれは受信パルスの振幅によつ
て検出時点が可成り変動する。即ち第２図の波形
の一部を拡大して第３図に示すように曲線１４の
場合はパルスの立上り点t0から検出レベルl1を交
叉する迄の期間φ1であるが、これより振幅が小
さい曲線１５のような受信パルスの場合はこの検
出レベルl1迄の期間はφ2となり、更にレベルが低
い曲線１６のようになると曲線１４，１５と同一
位相のサイクルの部分では検出レベルl1と交叉せ
ず、次のサイクルにおいて検出レベルl1を通過す
るため、時点t0から検出レベルl1を越えるまでの
期間はt0より360゜以上遅れたφ3と著しく大きくな
る。したがつて、従来においてはAGC回路によ
つて受信信号の振幅が一定になるように制御し、
一定の検出レベルl1に対して、受信パルスの立上
りt0から一定の位相で検出できるようにしてい
た。しかし受信パルスの包絡線が変化すると個々
の波の波高値が変化してしまう。即ち例えば第４
図に示すように一定振幅にしても受信パルスの包
絡線が曲線１７，１８，１９のようにその立上り
が急なものと、なだらかなものとによつては検出
レベルl1と交叉する点がそれぞれ異なる。 Conventionally, the detection point of the received wave was determined when the level of the received wave exceeded a certain detection level l1 . However, the detection time varies considerably depending on the amplitude of the received pulse. That is, in the case of curve 14, as shown in FIG . 3 by enlarging a part of the waveform in FIG . In the case of a received pulse with a small amplitude such as curve 15, the period up to this detection level l 1 is φ 2 , and when the level becomes even lower as shown in curve 16, the detection level decreases in the part of the cycle that has the same phase as curves 14 and 15. Since it does not intersect with l 1 and passes the detection level l 1 in the next cycle, the period from time t 0 until it exceeds the detection level l 1 becomes extremely long, with φ 3 delayed by more than 360° from t 0 . Therefore, in the past, an AGC circuit was used to control the amplitude of the received signal to be constant.
For a fixed detection level l1 , detection was made possible at a fixed phase from the rising edge t0 of the received pulse. However, when the envelope of the received pulse changes, the peak value of each wave changes. That is, for example, the fourth
As shown in the figure, even if the amplitude is constant, the envelope of the received pulse may cross the detection level l 1 depending on whether the rise is steep as shown in curves 17, 18, and 19, or when it is gentle. Each one is different.
第１図に示した流量計のような場合においては
その流体の流れに逆らう場合とこれに沿う場合と
によつてその受信超音波パルスの波形が変化し、
即ち包絡線が変化し、また流速によつてもその包
絡線が変化する。このような包絡線の変化は先に
述べた流量測定の誤差、この場合はスパン誤差と
なる。 In a case like the flowmeter shown in Fig. 1, the waveform of the received ultrasonic pulse changes depending on whether it goes against the flow of the fluid or follows it.
That is, the envelope changes, and the envelope also changes depending on the flow velocity. Such a change in the envelope results in the above-mentioned flow rate measurement error, in this case a span error.
例えば超音波パルスの搬送波の周波数を0.5M
Hz、θを25゜、管内径Ｄを1000mm、音速ｃを1450
ｍ／秒とすると、超音波の検出位相1゜について
5.6×10-9秒となる。検出位相が変化するために
起る誤差は波高値に対する検出レベルによつて変
化し、45゜の位相位置では波高値１％の変化に対
して0.58゜変化し約15mm／秒の流速測定誤差とな
る。スパンが1000mm／秒の場合は1.5％の測定誤
差となり、AGCにより検出受信波の波高値を±
１％以内に保つ事は難かしく、受信波の包絡線が
変化すれば誤差は一層大きくなる。従つて従来に
おいてはスパンが1000mm／秒の場合は数％の誤差
が生じていると云える。 For example, the frequency of the ultrasonic pulse carrier wave is 0.5M
Hz, θ is 25°, tube inner diameter D is 1000mm, sound speed c is 1450
If it is m/sec, then the detection phase of ultrasonic wave is 1°.
It becomes 5.6×10 -9 seconds. The error caused by the change in the detection phase changes depending on the detection level relative to the peak value. At a phase position of 45°, a 1% change in the peak value changes by 0.58°, resulting in a flow velocity measurement error of approximately 15 mm/sec. Become. If the span is 1000 mm/sec, there will be a measurement error of 1.5%, and AGC will adjust the peak value of the detected received wave to ±
It is difficult to maintain it within 1%, and if the envelope of the received wave changes, the error will become even larger. Therefore, it can be said that conventionally, when the span is 1000 mm/sec, an error of several percent occurs.
ところで例えば第５図に示すように振幅の大き
さに拘らず常にゼロレベルのところ、つまりゼロ
交叉点は一致したものとなる。よつてゼロを横切
るところを検出すれば良いが、受信パルス波の最
初のゼロ交叉点を検出しようとしても、その前に
雑音成分が存在しているため、ゼロ交叉点が多数
あつてどれが受信パルスの最初のゼロ交叉か検出
することはできない。受信回路中にヒステリシス
を持たせたり、波器によつて雑音を除去するこ
とも考えられるが、そのような手段の挿入によつ
てもとの受信波形に対し位相差を与えることにな
り、正しい立上り点の検出はできない。 By the way, as shown in FIG. 5, for example, the zero level, that is, the zero crossing point always coincides regardless of the magnitude of the amplitude. Therefore, it is enough to detect the point where the received pulse wave crosses zero, but even if you try to detect the first zero crossing point of the received pulse wave, there are noise components before that, so there are many zero crossing points and it is difficult to detect which one is the received pulse wave. It is not possible to detect the first zero crossing of the pulse. It may be possible to add hysteresis to the receiving circuit or remove noise using a waveform, but inserting such means will give a phase difference to the original received waveform, making it difficult to correct the noise. The rising point cannot be detected.
このような点から、第６図に示す信号検出回路
が考案された（実開昭54−127555号公報）。以下、
この回路について説明すると、図において入力端
子２１は被検出信号が印加される端子であり、被
検出信号は同入力端子２１を介して比較回路２２
の反転入力端子へ供給される。一方、比較回路２
２の非反転入力端子には基準電源（出力レベル
l1）２３の出力が抵抗２４を介して供給され、あ
るいは接地電位がスイツチ２５を介して供給され
る。この場合、スイツチ２５は比較回路２２の出
力によつてそのオン／オフが制御されるもので、
比較回路２２の出力が“ロー”レベルの時は“オ
ン”状態となり、“ハイ”レベルの時は“オフ”
状態となる。また、比較回路２２の出力は出力端
子２６から検出信号として出力される。 From this point of view, the signal detection circuit shown in FIG. 6 was devised (Japanese Utility Model Application Publication No. 127555/1983). below,
To explain this circuit, the input terminal 21 in the figure is a terminal to which a detected signal is applied, and the detected signal is passed through the input terminal 21 to the comparator circuit 22.
is supplied to the inverting input terminal of On the other hand, comparison circuit 2
The reference power supply (output level
l 1 ) 23 is supplied via a resistor 24, or the ground potential is supplied via a switch 25. In this case, the switch 25 is turned on/off by the output of the comparator circuit 22.
When the output of the comparator circuit 22 is at a “low” level, it is in an “on” state, and when it is at a “high” level, it is in an “off” state.
state. Further, the output of the comparison circuit 22 is outputted from the output terminal 26 as a detection signal.
このように構成された回路において、入力端子
２１に被検出信号が印加される以前においては、
比較回路２２の出力は“ハイ”レベルの状態にあ
り、したがつてスイツチ２５は開状態にあり、比
較回路２２の非反転入力端子に基準電源２３の出
力電圧（レベルl1）が供給されている。なお、こ
れは次の理由による。すなわち、もし比較回路２
２の出力が“ロー”レベルにありスイツチ２５が
“オン”状態にあつたとしても、入力端子２１の
雑音信号が接地レベルより小となつた時点で比較
回路２２が反転し、その出力が“ハイ”レベルと
なり、スイツチ２５が“オフ”状態となる。この
結果、比較回路２２の非反転入力端子に基準電源
２３の出力電圧が供給され、またこの出力電圧は
入力端子２１の雑音信号レベルよりはるかに大で
あり、したがつて、以後比較回路２２の出力は
“ハイ”レベルを続ける。このような状態におい
て、入力端子２１に被検出信号、例えば超音波振
動子によつて受波した超音波パルスを増幅器によ
つて増幅し、更にAGC回路によつつて一定レベ
ルとした信号（第７図イ参照）が印加されると、
同被検出信号のレベルが前記レベルl1より大とな
つた時点t1において、比較回路２２の出力が“ロ
ー”レベルに反転し（第７図ロ参照）、スイツチ
２５が“オン”となり、比較回路２２の非反転入
力端子に接地電位が供給される。そして、時点t2
において被検出信号のレベルが接地電位を横切る
と、比較回路２２の出力が再び“ハイ”レベルに
反転し、したがつて基準電源２３の出力が再び比
較回路２２の非反転入力端へ供給される。以下、
同様な動作が繰返えされ、これにより比較回路２
２の出力として第７図ロに示すパルス信号が得ら
れる。 In the circuit configured in this way, before the detected signal is applied to the input terminal 21,
The output of the comparator circuit 22 is at a "high" level, so the switch 25 is in an open state, and the output voltage (level l 1 ) of the reference power supply 23 is supplied to the non-inverting input terminal of the comparator circuit 22. There is. This is due to the following reason. That is, if comparison circuit 2
Even if the output of the switch 25 is at the "low" level and the switch 25 is in the "on" state, the comparator circuit 22 is inverted as soon as the noise signal at the input terminal 21 becomes lower than the ground level, and its output becomes " The signal becomes "high" level, and the switch 25 becomes "off". As a result, the output voltage of the reference power supply 23 is supplied to the non-inverting input terminal of the comparator circuit 22, and this output voltage is much higher than the noise signal level of the input terminal 21. The output continues at “high” level. In such a state, a signal to be detected at the input terminal 21, for example, an ultrasonic pulse received by an ultrasonic transducer, is amplified by an amplifier, and then a signal (the first signal) is made to a constant level by an AGC circuit. (see Figure 7 A) is applied,
At time t1 when the level of the detected signal becomes higher than the level l1 , the output of the comparator circuit 22 is inverted to a "low" level (see FIG. 7B), and the switch 25 is turned "ON". A ground potential is supplied to the non-inverting input terminal of the comparison circuit 22. And at time t 2
When the level of the detected signal crosses the ground potential at , the output of the comparator circuit 22 is again inverted to the "high" level, and therefore the output of the reference power supply 23 is again supplied to the non-inverting input terminal of the comparator circuit 22. . below,
Similar operations are repeated, and as a result, the comparator circuit 2
The pulse signal shown in FIG. 7B is obtained as the output of step 2.
そして、以上の動作から明らかなように、この
信号検出回路によれば、被検出信号が始めてレベ
ルl1を超えた後最初に接地レベルを横切る点（時
点t2）を、比較回路２２の出力パルスの最初の立
上り時点として検出することができる。 As is clear from the above operation, according to this signal detection circuit, the point at which the detected signal crosses the ground level for the first time after exceeding the level l1 (time t2 ) is determined by the output of the comparator circuit 22. It can be detected as the first rising edge of the pulse.
しかしながら、第６図に示す信号検出回路には
未だ次の様な欠点がある。すなわち、スイツチ２
５としては通常FET（電界効果トランジスタ）等
が用いられるが、この場合、FETのゲート容量
等により同FETのオン／オフ動作が遅れ、この
ため比較電圧Vr（第６図参照）に対する正帰還効
果が失なわれる期間が生じ、トリガミスの原因と
なる。また、比較回路２２の非反転入力端子から
みたインピーダンス（比較電圧Vrを加える回路
のインピーダンス）はスイツチ（FET）２５が
“オフ”の場合に高い値となる。このため、抵抗
２４、比較回路２２の入力容量、FETのドレイ
ンソース間容量等により比較電圧Vrの立上りが
遅れ、この結果比較回路２２の出力が実際には第
７図ロに示すようなパルス波形とならず第７図ハ
に示すように立上りのなまつた波形となつてしま
う。 However, the signal detection circuit shown in FIG. 6 still has the following drawbacks. In other words, switch 2
5 is usually a FET (field effect transistor), etc., but in this case, the on/off operation of the FET is delayed due to the gate capacitance of the FET, and this causes a positive feedback effect on the comparison voltage Vr (see Figure 6). There is a period during which the signal is lost, causing a trigger error. Further, the impedance seen from the non-inverting input terminal of the comparator circuit 22 (the impedance of the circuit that applies the comparison voltage Vr) has a high value when the switch (FET) 25 is "off". Therefore, the rise of the comparison voltage Vr is delayed by the resistor 24, the input capacitance of the comparison circuit 22, the drain-source capacitance of the FET, etc., and as a result, the output of the comparison circuit 22 actually has a pulse waveform as shown in FIG. Instead, the waveform has a slow rise as shown in FIG. 7C.
そして、これにより比較回路２２の立上り時点
を正確に検出し得なくなる。 This makes it impossible to accurately detect the rising edge of the comparison circuit 22.
この考案は上述した鑑点から、第６図に示す信
号検出回路を更に改良し、被検出信号のゼロクロ
ス点（第７図イにおける時点t2）を正確に検出し
得るようにした信号検出回路を提供するものであ
る。 Based on the above considerations, this invention is a signal detection circuit that further improves the signal detection circuit shown in FIG. 6 and can accurately detect the zero-crossing point (time t 2 in FIG. 7A) of the detected signal. It provides:
以下、図面を参照しこの考案の実施例について
説明する。 Hereinafter, embodiments of this invention will be described with reference to the drawings.
第８図はこの考案の第１の実施例を示すもの
で、この図において第６図に示す回路各部と対応
する部分には同一の符号が付してある。この図に
示す信号検出回路が第６図に示す信号検出回路と
異なる点は比較回路２２の出力端子と非反転入力
端子との間にコンデンサ３０（帰還回路）が介挿
されていることである。そして、このコンデンサ
３０によつて比較回路２２の出力の立上り、立下
りが比較回路２２の非反転入力端子へ正帰還さ
れ、これにより、比較回路２２の出力が第７図ニ
に示すように立上り、立下りの共に急峻な波形と
なる。すなわち、コンデンサ３０を介挿すること
により、スイツチ２５の遅れあるいは比較電圧
Vrを加える回路のインピーダンス（抵抗４）に
よる遅れを補償することができ、これにより、第
７図イに示す時点t2を正確に検出することが可能
となる。 FIG. 8 shows a first embodiment of this invention, and in this figure, parts corresponding to those of the circuit shown in FIG. 6 are given the same reference numerals. The signal detection circuit shown in this figure differs from the signal detection circuit shown in FIG. 6 in that a capacitor 30 (feedback circuit) is inserted between the output terminal and the non-inverting input terminal of the comparison circuit 22. . The rising and falling edges of the output of the comparator circuit 22 are fed back positively to the non-inverting input terminal of the comparator circuit 22 by this capacitor 30, so that the output of the comparator circuit 22 rises as shown in FIG. , the waveform becomes steep both at the falling edge. That is, by inserting the capacitor 30, the delay of the switch 25 or the comparison voltage can be reduced.
It is possible to compensate for the delay due to the impedance (resistance 4) of the circuit that applies Vr, thereby making it possible to accurately detect time t 2 shown in FIG. 7A.
第９図はこの考案の第２の実施例を示すもの
で、この図に示す実施例は比較回路２２の入力端
子の極性が第８図に示す実施例と逆の場合であ
る。すなわち、第９図に示す信号検出回路におい
ては、入力端子２１に得られる被検出信号が比較
回路２２の非反転入力端子に供給され、比較電圧
Vrが比較回路２２の反転入力端子に供給される。
また、スイツチ２５は比較回路２２の出力が“ロ
ー”レベルの時“オフ”状態となり、“ハイ”レ
ベルの時“オン”状態となる。さらに、比較回路
２２の出力がインバータ３１およびコンデンサ３
０を介して比較回路２２の反転入力端子へ正帰還
される。そして、この信号検出回路においても第
８図に示す信号検出回路と全く同様の作用効果を
得ることができる。 FIG. 9 shows a second embodiment of this invention, in which the polarity of the input terminal of the comparator circuit 22 is opposite to that of the embodiment shown in FIG. That is, in the signal detection circuit shown in FIG. 9, the detected signal obtained at the input terminal 21 is supplied to the non-inverting input terminal of the comparison circuit 22, and the comparison voltage is
Vr is supplied to the inverting input terminal of the comparison circuit 22.
Further, the switch 25 is in the "off" state when the output of the comparison circuit 22 is at the "low" level, and is in the "on" state when the output is at the "high" level. Furthermore, the output of the comparison circuit 22 is connected to the inverter 31 and the capacitor 3.
0 to the inverting input terminal of the comparator circuit 22. In this signal detection circuit as well, it is possible to obtain exactly the same effects as the signal detection circuit shown in FIG.
以上説明したように、この考案によれば比較回
路の出力を同比較回路の入力端子へ正帰還したの
で、被検出信号のゼロクロス点を正確に検出する
ことができ、これにより被検出信号の到達時点を
正確に検知することができる。したがつて、超音
波流量計、超音波レベル計、魚群探知器、その他
の超音波パルス機器、あるいはレーダ等の高周波
パルス機器等における信号検出に用いて特に好適
である。 As explained above, according to this invention, since the output of the comparator circuit is positively fed back to the input terminal of the comparator circuit, it is possible to accurately detect the zero-crossing point of the signal to be detected. It is possible to accurately detect the point in time. Therefore, it is particularly suitable for use in signal detection in ultrasonic flow meters, ultrasonic level meters, fish finders, other ultrasonic pulse devices, or high frequency pulse devices such as radar.
第１図は超音波流量計のパイプと送受波器との
関係を示す図、第２図は超音波流量計の受信パル
スの包絡線を示す図、第３図は受信波の特定レベ
ルの検出の説明に供する波形図、第４図は受信パ
ルスの包絡線の変化の検出点への影響を示す図、
第５図は超音波流量計の受信パルスのゼロクロス
点が振幅の大小にかかわらず一定となることを示
す図、第６図は従来の信号検出回路の一例を示す
回路図、第７図イ〜ハは共に第６図に示す回路の
動作を説明するための波形図、第７図ニはこの考
案による信号検出回路の動作を説明するための波
形図、第８図、第９図は各々この考案の第１、第
２の実施例を示す回路図である。
２２……比較回路、２５……スイツチ、３０…
…コンデンサ、３１……インバータ。
Figure 1 shows the relationship between the pipe and transducer of the ultrasonic flowmeter, Figure 2 shows the envelope of the received pulse of the ultrasonic flowmeter, and Figure 3 shows the detection of a specific level of the received wave. FIG. 4 is a diagram showing the influence of changes in the envelope of the received pulse on the detection point.
Fig. 5 is a diagram showing that the zero-crossing point of the received pulse of an ultrasonic flowmeter remains constant regardless of the magnitude of the amplitude, Fig. 6 is a circuit diagram showing an example of a conventional signal detection circuit, and Fig. 7 C is a waveform diagram for explaining the operation of the circuit shown in FIG. 6, FIG. 7 D is a waveform diagram for explaining the operation of the signal detection circuit according to this invention, and FIGS. FIG. 3 is a circuit diagram showing first and second embodiments of the invention. 22...Comparison circuit, 25...Switch, 30...
...Capacitor, 31...Inverter.
Claims (1)
回路と、基準電圧あるいは接地電位を前記比較回
路の出力に基づいて選択的に前記比較回路の他方
の入力端子に供給する切換手段と、前記比較回路
の出力を前記比較回路の他方の入力端子へ正帰還
する帰還回路とを具備してなる信号検出回路。 a comparison circuit to which a detected signal is supplied to one input terminal; a switching means to selectively supply a reference voltage or a ground potential to the other input terminal of the comparison circuit based on the output of the comparison circuit; and the comparison circuit. A signal detection circuit comprising: a feedback circuit that positively feeds back an output of the circuit to the other input terminal of the comparison circuit.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP1980048353U JPS6336474Y2 (en) | 1980-04-10 | 1980-04-10 |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP1980048353U JPS6336474Y2 (en) | 1980-04-10 | 1980-04-10 |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS56152398U JPS56152398U (en) | 1981-11-14 |
| JPS6336474Y2 true JPS6336474Y2 (en) | 1988-09-27 |
Family
ID=29643346
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP1980048353U Expired JPS6336474Y2 (en) | 1980-04-10 | 1980-04-10 |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPS6336474Y2 (en) |
Family Cites Families (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS56221Y2 (en) * | 1976-01-07 | 1981-01-07 | ||
| JPS537715U (en) * | 1976-07-06 | 1978-01-23 |
-
1980
- 1980-04-10 JP JP1980048353U patent/JPS6336474Y2/ja not_active Expired
Also Published As
| Publication number | Publication date |
|---|---|
| JPS56152398U (en) | 1981-11-14 |
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