JPS6252823B2 - - Google Patents
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- Publication number
- JPS6252823B2 JPS6252823B2 JP4504580A JP4504580A JPS6252823B2 JP S6252823 B2 JPS6252823 B2 JP S6252823B2 JP 4504580 A JP4504580 A JP 4504580A JP 4504580 A JP4504580 A JP 4504580A JP S6252823 B2 JPS6252823 B2 JP S6252823B2
- Authority
- JP
- Japan
- Prior art keywords
- phase
- output
- transformer
- leakage current
- angular frequency
- Prior art date
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- 238000009413 insulation Methods 0.000 claims description 20
- 238000000034 method Methods 0.000 claims description 12
- 230000010355 oscillation Effects 0.000 claims description 7
- 101710170230 Antimicrobial peptide 1 Proteins 0.000 description 3
- 229910001219 R-phase Inorganic materials 0.000 description 3
- 239000002131 composite material Substances 0.000 description 3
- 238000001514 detection method Methods 0.000 description 3
- 238000010586 diagram Methods 0.000 description 3
- NCGICGYLBXGBGN-UHFFFAOYSA-N 3-morpholin-4-yl-1-oxa-3-azonia-2-azanidacyclopent-3-en-5-imine;hydrochloride Chemical compound Cl.[N-]1OC(=N)C=[N+]1N1CCOCC1 NCGICGYLBXGBGN-UHFFFAOYSA-N 0.000 description 2
- 102100036464 Activated RNA polymerase II transcriptional coactivator p15 Human genes 0.000 description 2
- 102100034033 Alpha-adducin Human genes 0.000 description 2
- 101710170231 Antimicrobial peptide 2 Proteins 0.000 description 2
- 102100024348 Beta-adducin Human genes 0.000 description 2
- 101000713904 Homo sapiens Activated RNA polymerase II transcriptional coactivator p15 Proteins 0.000 description 2
- 101000799076 Homo sapiens Alpha-adducin Proteins 0.000 description 2
- 101000689619 Homo sapiens Beta-adducin Proteins 0.000 description 2
- 101000629598 Rattus norvegicus Sterol regulatory element-binding protein 1 Proteins 0.000 description 2
- 230000018199 S phase Effects 0.000 description 2
- 229910004444 SUB1 Inorganic materials 0.000 description 2
- 101150023658 SUB4 gene Proteins 0.000 description 2
- 101100381996 Saccharomyces cerevisiae (strain ATCC 204508 / S288c) BRO1 gene Proteins 0.000 description 2
- 101100488882 Saccharomyces cerevisiae (strain ATCC 204508 / S288c) YPL080C gene Proteins 0.000 description 2
- 230000008878 coupling Effects 0.000 description 2
- 238000010168 coupling process Methods 0.000 description 2
- 238000005859 coupling reaction Methods 0.000 description 2
- 102100034004 Gamma-adducin Human genes 0.000 description 1
- 101000799011 Homo sapiens Gamma-adducin Proteins 0.000 description 1
- 229910004438 SUB2 Inorganic materials 0.000 description 1
- 101150080287 SUB3 gene Proteins 0.000 description 1
- 101100311330 Schizosaccharomyces pombe (strain 972 / ATCC 24843) uap56 gene Proteins 0.000 description 1
- 230000002950 deficient Effects 0.000 description 1
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- 238000005259 measurement Methods 0.000 description 1
- 101150018444 sub2 gene Proteins 0.000 description 1
Classifications
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R27/00—Arrangements for measuring resistance, reactance, impedance, or electric characteristics derived therefrom
- G01R27/02—Measuring real or complex resistance, reactance, impedance, or other two-pole characteristics derived therefrom, e.g. time constant
- G01R27/16—Measuring impedance of element or network through which a current is passing from another source, e.g. cable, power line
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- Physics & Mathematics (AREA)
- General Physics & Mathematics (AREA)
- Measurement Of Resistance Or Impedance (AREA)
- Emergency Protection Circuit Devices (AREA)
Description
本発明は交流三相3線式電路、特に三相3線変
圧器2次一端接地式電路の活線型絶縁抵抗、対地
浮遊容量の測定方法に関する。
従来三相3線2次一端接地式電路の絶縁抵抗測
定の第1の方法として、例えば雑誌「図説電気」
の1979年第20巻1号24〜28ページに示された方法
があるが、これは接地線を切断してそこに直列に
低周波印加用のトランスと漏洩電流検出用抵抗を
挿入接続する方法である。この方法によれば漏洩
電流中の有効分を検出することにより非接地相電
路の並列合成絶縁抵抗を得ることができるが、接
地線に上述工事を追加する必要にせまられるため
簡単に、測定を実施することができない。また接
地線にトランス、漏洩電流検出用抵抗が挿入され
るため接地線のインピーダンスが高くなる傾向に
あり、一般の接地系の接地抵抗にはしかるべき規
定値が定められているためのぞましいものとは言
えない。
また電路に地絡があつた場合、地絡電流は直接
接地線に流れるための上述のトランス、漏洩電流
検出用抵抗の大容量のものを必要とするため多く
の問題がある。
また従来の三相3線電路で2次一端接地式電路
の絶縁抵抗測定の第2の方法として上述の雑誌
(同年、同巻、同号の18〜23ページ)に示されて
いる方法がある。これは商用周波数による漏洩電
流中の各非接地相毎の有効成分をもとめ、それら
の和によつて非接地相の並列合成絶縁抵抗を求め
る方法であるが、同文中にも示されている通り、
非接地相2線の対地浮遊容量が等しくない場合こ
の不平衡分のみ誤差要因となつて正しい並列合成
絶縁抵抗測定値を得ることができない。
本発明の方法は上記問題をすべて解決するだけ
でなく、各非接地相電路の絶縁抵抗、対地浮遊容
量等をそれぞれ測定できるため、どの非接地相電
路の絶縁が悪いか直接判断できるものである。
以下図面を参照しながら本発明を詳細に説明す
る。
第1図は本発明の原理を示す図であり三相変圧
器2次一端接地の例を示している。
TR1は三相3線式変圧器で2次三角形回路で
ある。S相が接地線ELで接地されている。Z1、
2、3は負荷である。I1,I2,I3はそれぞれ、R相、
S相、T相の負荷電流である。R1,R2ならびに
C1,C2はそれぞれ非接地電路R相、T相の絶縁
抵抗ならびに対地浮遊容量である。ig1,ig2は、
非接地電路のそれぞれから絶縁抵抗、対地浮遊容
量を介して流出する漏洩電流である。したがつ
て、接地線ELが貫通する零相変流器ZCT出力に
は広く知られているように漏洩電流ig1+ig2が得
られる。
ところでR相の電圧をE0sinω0t、T相の電圧
をE0sin(ω0t−)とする。
ここでω0は商用角周波数、=120゜であ
る。したがつて、漏洩電流ig1+ig2の角周波数ω
0の成分をig10+ig20とすれば
ig10+ig20=E0/R1sinω0t+ω0C1E0cosω0t+E0/R2sin(ω0t−)
+ω0C2E0cos(ω0t−) ……
となる。式に相当する電流は零相変流器ZCT
の出力を中心角周波数ω0のバンドパスフイルタ
F1に通すことにより得られる。
漏洩電流ig10+ig20とR相の電圧E0sinω0tとの
積をとると、
(ig10+ig20)E0sinω0t=E0 2/2R1+E0 2/2R2cos+ω0C2E0 2/2sin
−E0 2/2R1cos2ω0t+ω0C1E0 2/2sin2ω0t
−E0 2/2R2cos(2ω0t−)+ω0C2E0 2/2
sin(2ω0t−) ……
となる。式の直流分をAとすれば、式から
A=E0 2/2R1+E0 2/2R2cos+ω0C2
E0 2/2sin……
となる。これは漏洩電流中の角周波数ω0におけ
るR相の有効分である。
また、漏洩電流ig10+ig20とT相の電圧E0sin
(ω0t−)との積をとると、
(ig10+ig20)E0sin(ω0t−)=E0 2/2R2+E0 2/2R1cos−ω0C1E0 2/2sin
E0 2/2R1cos(2ω0t−)+ω0C1E0 2/2
sin(2ω0t−)−E0 2/2R2cos2(ω0t−) ……
となる。式の直流分をBとすれば、式から
B=E0 2/2R2+E0 2/2R1cos−ω0C1
E0 2/2sin……
となる。これは漏洩電流中の角周波数ω0におけ
るT相の有効分である。ところで直流分Aと直流
分Bとの和をPとすれば
P=A+B
、式から
=E0 2(1+cos)/2・(1/R1+1/R2)−ω0E0 2sin/2・(C1−C2) ……
となる。したがつてPは対地浮遊容量C1,C2が
等しい場合式の第2項は零となるから、このと
きに限りPの値は絶縁抵抗R1,R2の並列合成抵
抗値に逆比例した値となる。
ところでR相の電圧を90゜移相器で位相を90゜
だけシフトすることによりE0cosω0tを得ること
ができるが、これと漏洩電流ig10+ig20との積を
とると
(ig10+ig20)・E0cosω0t=ω0C1E0 2/2−E0 2/2R2sin+ω0C2E0 2/2cos
+E0 2/2R1sin2ω0t+ω0C1E0 2/2cos2ω0t
+E0 2/2R2sin(2ω0t−)+ω0C2E0 2/2cos
(2ω0t−) ……
となる。式の直流分をCとすれば、式から
C=ω0C1E0 2/2−E0 2/2R2sin+ω0
C2E0 2/2cos……
となる。これは漏洩電流中の角周波数ω0におけ
るR相の無効分である。
またT相の電圧を90゜移相器で位相を90゜だけ
シフトすることによりE0cos(ω0t−)を得る
ことができるが、これと漏洩電流ig10+ig20との
積をとると、
(ig10+ig20)・E0cos(ω0t−)=E0 2/2R1sin+ω0C1E0 2/2cos+ω0C2E0 2/2
+E0 2/2R1sin(2ω0t−〓)+ω0C1E0 2/2cos
(2ω0t−)+E0 2/2R2sin2(ω0t−)
+ω0C2E0 2/2cos2(ω0t+) ……
となる。式の直流分をDとすれば式から
D=E0 2/2R1sin+ω0C1E0 2/2cos+
ω0C2E0 2/2……
となる。これは漏洩電流中の角周波数ω0におけ
るT相の無効分である。
ところで直流分Cと直流分Dとの和をQとすれ
ば、
Q=C+D
式、式から
=ω0E0 2(1+cos)/2・(C1+C2)+E0 2sin/2(1/R1−1/R2) ……
となる。したがつてQは絶縁抵抗R1,R2が等し
い場合、式の第2項は零となるから、このとき
限りQの値は対地浮遊容量の並列合成値に比例し
た値となることが分る。
第1図においてトランスTR2は接地線ELが貫
通する低周波発振トランス又は発振器出力をパワ
ーアンプに接続した結合トランスであり、OSCC
は低周波発振回路である。かくすることにより接
地線ELには角周波数ω1の微弱な低周波電圧
e1sinω1tを誘起させることができる。したがつ
て、この電圧によつて流れる漏洩電流をig11+
ig21とすれば、これは零相変流器ZCT出力を中心
角周波数ω1のバンドパスフイルタF2に通すこ
とにより、得ることができる。
すなわち、
ig11+ig21=e1/R1sinω1t+ω1C1e1cosω1t
+e1/R2sinω1t+ω1C2e1cosω1t ……
となる。漏洩電流ig11+ig21と上述の微弱な低周
波電圧e1sinω1tに比例した電圧e0sinω1tとの積を
とると
(ig11+ig21)e0sinω1t=e1e0/2(1/R1+1/R2)−(e0e1/2cos2ω1t
+ω1e0e1(C1+C2)/2・sin2ω0t ……
式の直流分をFとすれば、式から
F=e1e0/2(1/R1+1/R2)……
これは漏洩電流中の角周波数ω1における有効分
である。また上述の電圧e0sinω1tを90゜移相器で
90゜だけシフトすることによりe0cosω1tを得る
ことができるが、これと漏洩電流ig11+ig21との
積をとると
(ig11+ig21)・e0cosω1t=ω1e0e1/2(C1+C2)+e0e1/2(1/R1+1/R2)
sin2ω1t+ω1e0e1/2(C1+C2)・cos2ω1t ……
式の直流分をGとすれば、式から
G=ω1e0e1/2(C1+C2) ……
となる。これは漏洩電流中の角周波数ω1におけ
る無効分である。
式のPに式のFを代入し整理すれば、
ω0E0 2/2sin(C1−C2)=E0 2/e0e1(1−cos)・F−P ……
となる。
、式から
ω0E0 2sin・C1=E0 2/e0e1(1−cos)・F−P+E0 2/e0e1・ω0/ω1・sin・G ……
ω0E0 2・sin・C2=E0 2/e0e1・ω0/ω1・sin・G−〔E0 2/e0e1(1−cos)・F−P〕…
…
すなわち、、の右辺の計算を行なうことによ
り対地浮遊容量C1,C2に比例した値を得ること
ができる。
式のQに式のGを代入して整理すれば
E0 2sin/2(1/R1−1/R2)=Q−ω0/ω1・E0 2/e0e1(1−cos)・G ……
式と式から
E0 2sin1/R1=E0 2/e0e1sin・F+Q=ω0/ω1E0 2/e0e1(1+cos)・G ……
E0 2sin1/R2=Q−ω0/ω1・E0 2/e0e1(1+cos)・G−E0 2/e0e1sin・F ……
すなわち、、の右辺の計算を行なうことによ
り絶縁抵抗R1,R2に逆比例した値を得ることが
できる。
cos120゜=−1/2、
The present invention relates to a method for measuring the live line insulation resistance and stray capacitance to ground of an AC three-phase three-wire electric line, particularly a three-phase three-wire transformer secondary one-end grounded electric line. The first method for measuring the insulation resistance of a conventional three-phase, three-wire, secondary, one-end, grounded circuit is described, for example, in the magazine "Illustrated Denki".
There is a method shown in 1979, Vol. 20, No. 1, pages 24-28, in which the grounding wire is cut and a transformer for low frequency application and a resistor for leakage current detection are inserted and connected in series. It is. According to this method, it is possible to obtain the parallel composite insulation resistance of the non-grounded phase circuit by detecting the effective component in the leakage current, but since it is necessary to add the above-mentioned work to the grounding wire, it is not easy to measure. cannot be implemented. In addition, since a transformer and leakage current detection resistor are inserted into the grounding wire, the impedance of the grounding wire tends to increase. I can not say. Further, when a ground fault occurs in the electrical path, many problems arise because the above-mentioned transformer and leakage current detection resistor of large capacity are required because the ground fault current flows directly to the ground line. A second method for measuring the insulation resistance of a conventional three-phase three-wire circuit with a secondary one-end grounding system is the method described in the above-mentioned magazine (pages 18-23 of the same year, same volume, and same issue). . This is a method of determining the effective component of each non-grounded phase in the leakage current due to the commercial frequency, and then calculating the parallel combined insulation resistance of the non-grounded phases by summing them. ,
If the ground stray capacitances of the two non-grounded phase wires are not equal, this unbalanced amount alone becomes an error factor, making it impossible to obtain a correct parallel composite insulation resistance measurement value. The method of the present invention not only solves all of the above problems, but also measures the insulation resistance, stray capacitance, etc. of each non-grounded phase circuit, so it can directly determine which non-grounded phase circuit has poor insulation. . The present invention will be described in detail below with reference to the drawings. FIG. 1 is a diagram showing the principle of the present invention, and shows an example of grounding one end of the secondary of a three-phase transformer. TR1 is a three-phase three-wire transformer with a second-order triangular circuit. The S phase is grounded with the grounding wire EL. Z 1 ,
2 and 3 are loads. I 1 , I 2 , I 3 are R phase,
These are the S-phase and T-phase load currents. R 1 , R 2 and
C 1 and C 2 are the insulation resistance and ground stray capacitance of the R-phase and T-phase non-grounded circuits, respectively. ig 1 and ig 2 are
This is the leakage current that flows out from each ungrounded circuit via insulation resistance and ground stray capacitance. Therefore, as is widely known, a leakage current ig 1 +ig 2 is obtained at the zero-phase current transformer ZCT output through which the grounding wire EL passes. By the way, the voltage of the R phase is E0sinω0t, and the voltage of the T phase is E0sin(ω0t−). Here, ω0 is the commercial angular frequency, = 120°. Therefore, the angular frequency ω of the leakage current ig 1 + ig 2
If the component of 0 is ig 10 + ig 20 , then ig 10 + ig 20 =E 0 /R 1 sinω 0 t+ω 0 C 1 E 0 cosω 0 t+E 0 /R 2 sin(ω 0 t−) +ω 0 C 2 E 0 cos (ω 0 t−) …… becomes. The current corresponding to the formula is zero phase current transformer ZCT
The output of is passed through a bandpass filter with center angular frequency ω 0 .
Obtained by passing through F 1 . Taking the product of the leakage current ig 10 + ig 20 and the R-phase voltage E 0 sinω 0 t, we get (ig 10 + ig 20 )E 0 sinω 0 t=E 0 2 /2R 1 +E 0 2 /2R 2 cos + ω 0 C 2 E 0 2 /2sin -E 0 2 /2R 1 cos2ω 0 t+ ω 0 C 1 E 0 2 /2sin2ω 0 t -E 0 2 / 2R 2 cos (2ω 0 t-) + ω 0 C 2 E 0 2 /2 sin (2ω 0 t−) ……. If the DC component of the equation is A, then from the equation A=E 0 2 /2R 1 +E 0 2 /2R 2 cos + ω 0 C 2
E 0 2 /2sin... This is the effective component of the R phase at the angular frequency ω 0 in the leakage current. Also, leakage current ig 10 + ig 20 and T phase voltage E 0 sin
(ω 0 t−), (ig 10 + ig 20 )E 0 sin(ω 0 t−)=E 0 2 /2R 2 +E 0 2 /2R 1 cos−ω 0 C 1 E 0 2 / 2sinE02 / 2R1cos ( 2ω0t - ) + ω0C1E02 / 2sin ( 2ω0t -)- E02 / 2R2cos2 ( ω0t - ) ... If the DC component of the equation is B, then from the equation B=E 0 2 /2R 2 +E 0 2 /2R 1 cos−ω 0 C 1
E 0 2 /2sin... This is the effective part of the T phase at angular frequency ω 0 in the leakage current. By the way, if the sum of DC component A and DC component B is P, then P=A+B, and from the formula =E 0 2 (1+cos)/2・(1/R 1 +1/R 2 )−ω 0 E 0 2 sin/ 2・(C 1 − C 2 ) ...... Therefore, P is inversely proportional to the parallel combined resistance value of the insulation resistances R 1 and R 2 because the second term in the equation becomes zero when the ground stray capacitances C 1 and C 2 are equal. will be the value. By the way, by shifting the phase of the R-phase voltage by 90° using a 90° phase shifter, E 0 cosω 0 t can be obtained, but if we take the product of this and the leakage current ig 10 + ig 20 , we get (ig 10 +ig 20 )・E 0 cosω 0 t=ω 0 C 1 E 0 2 /2−E 0 2 /2R 2 sin+ω 0 C 2 E 0 2 /2cos +E 0 2 /2R 1 sin2ω 0 t+ω 0 C 1 E 0 2 / 2cos2ω0t + E02 / 2R2sin ( 2ω0t - ) + ω0C2E02 / 2cos ( 2ω0t - )... If the DC component of the equation is C, then from the equation C=ω 0 C 1 E 0 2 /2−E 0 2 /2R 2 sin+ω 0
C 2 E 0 2 /2cos... This is the reactive component of the R phase at the angular frequency ω 0 in the leakage current. Also, by shifting the phase of the T-phase voltage by 90° using a 90° phase shifter, E 0 cos (ω 0 t−) can be obtained, and the product of this and the leakage current ig 10 + ig 20 is obtained. and (ig 10 + ig 20 )・E 0 cos (ω 0 t−) = E 0 2 /2R 1 sin + ω 0 C 1 E 0 2 /2 cos + ω 0 C 2 E 0 2 /2 + E 0 2 /2R 1 sin ( 2ω 0 t−〓)+ω 0 C 1 E 0 2 /2cos (2ω 0 t−) +E 0 2 /2R 2 sin2 (ω 0 t−) +ω 0 C 2 E 0 2 /2cos2 (ω 0 t+) ... becomes. If the DC component of the equation is D, then from the equation D=E 0 2 /2R 1 sin+ω 0 C 1 E 0 2 /2cos+
ω 0 C 2 E 0 2 /2... This is the reactive component of the T phase at angular frequency ω 0 in the leakage current. By the way, if the sum of DC component C and DC component D is Q, then from the formula Q=C+D, =ω 0 E 0 2 (1+cos)/2・(C 1 +C 2 )+E 0 2 sin/2(1 /R 1 -1/R 2 ) ... It becomes. Therefore, when the insulation resistances R 1 and R 2 are equal, the second term in the equation becomes zero, so it can be seen that only in this case, the value of Q is proportional to the parallel composite value of the stray capacitance to ground. Ru. In Figure 1, the transformer TR2 is a low frequency oscillation transformer through which the grounding wire EL passes, or a coupling transformer in which the oscillator output is connected to the power amplifier, and the OSCC
is a low frequency oscillation circuit. As a result, a weak low frequency voltage with an angular frequency ω 1 is applied to the grounding wire EL.
e 1 sinω 1 t can be induced. Therefore, the leakage current flowing due to this voltage is ig 11 +
ig 21 , this can be obtained by passing the zero-phase current transformer ZCT output through a bandpass filter F 2 with a center angular frequency ω 1 . That is, ig 11 + ig 21 = e 1 /R 1 sin ω 1 t + ω 1 C 1 e 1 cos ω 1 t +e 1 /R 2 sin ω 1 t + ω 1 C 2 e 1 cos ω 1 t .... Taking the product of the leakage current ig 11 + ig 21 and the voltage e 0 sin ω 1 t proportional to the weak low-frequency voltage e 1 sin ω 1 t mentioned above, (ig 11 + ig 21 ) e 0 sin ω 1 t = e 1 e 0 /2(1/R 1 +1/R 2 )-(e 0 e 1 /2cos2ω 1 t +ω 1 e 0 e 1 (C 1 +C 2 )/2・sin2ω 0 t... Let F be the DC component of the equation. For example, from the formula F=e 1 e 0 /2 (1/R 1 +1/R 2 )...
This is the effective component at angular frequency ω 1 in the leakage current. Also, the voltage e 0 sinω 1 t mentioned above is changed using a 90° phase shifter.
By shifting by 90 degrees, e 0 cosω 1 t can be obtained, but when this is multiplied by the leakage current ig 11 + ig 21 , (ig 11 + ig 21 )・e 0 cosω 1 t=ω 1 e 0 e 1 /2 (C 1 + C 2 ) + e 0 e 1 /2 (1/R 1 + 1/R 2 ) sin2ω 1 t + ω 1 e 0 e 1 /2 (C 1 + C 2 )・cos2ω 1 t ... of the formula If the DC component is G, then from the formula, G=ω 1 e 0 e 1 /2 (C 1 +C 2 )... This is the reactive component at angular frequency ω 1 in the leakage current. If we substitute F in the formula for P in the formula and rearrange it, we get ω 0 E 0 2 /2sin(C 1 −C 2 )=E 0 2 /e 0 e 1 (1−cos)・F−P …… . , from the formula, ω 0 E 0 2 sin・C 1 =E 0 2 /e 0 e 1 (1−cos)・FP+E 0 2 /e 0 e 1・ω 0 /ω 1・sin・G ... ω 0 E 0 2・sin・C 2 =E 0 2 /e 0 e 1・ω 0 /ω 1・sin・G− [E 0 2 /e 0 e 1 (1−cos)・FP]…
... That is, by calculating the right-hand side of , it is possible to obtain values proportional to the ground stray capacitances C 1 and C 2 . Substituting G in the equation for Q in the equation and rearranging, E 0 2 sin/2 (1/R 1 -1/R 2 )=Q-ω 0 /ω 1・E 0 2 /e 0 e 1 (1 -cos)・G... From the formula and formula, E 0 2 sin1/R 1 = E 0 2 /e 0 e 1 sin・F+Q=ω 0 /ω 1 E 0 2 /e 0 e 1 (1+cos)・G... …E 0 2 sin1/R 2 =Q−ω 0 /ω 1・E 0 2 /e 0 e 1 (1+cos)・G−E 0 2 /e 0 e 1 sin・F …… In other words, the right side of By performing calculations, values that are inversely proportional to the insulation resistances R 1 and R 2 can be obtained. cos120゜=-1/2,
【式】を、、
、式に代入して整理すると、
となる。
第2図は、第1図のフイルタF1,F2出力なら
びに発振器OSCC出力から′〜′に相当する電
圧を得るための実施例を示している。
R相電圧をトランスTR4を介してかけ算器
MULT1の一方の入力に加え、他の入力端にフ
イルタF1の出力を加えることによりかけ算器
MULT1の出力には式に相当する信号が得ら
れる。かけ算器MULT1の出力をローパスフイ
ルタLPF1に加えることにより、その出力には
式で表わされる直流分Aが得られる。同様にT相
電圧をトランスTR3を介してかけ算器MULT3
の一方の入力端に加え、他の入力端にフイルタ
F1の出力を加えることによりかけ算器MULT3
の出力には式に相当する信号が得られる。かけ
算器MULT3の出力をローパスフイルタLPF3
に加えることにより、その出力には式に相当す
る直流分Bが得られる。ローパスフイルタLPF
1,LPF2の出力を加算器ADD1のそれぞれの
入力に加えることにより加算器ADDの出力には
式で表わされる信号Pが得られる。
トランスTR4の出力を90゜移相器PS1に加え
て90゜位相をシフトさせ、90゜移相器PS1とバ
ンドパスフイルタF1の出力の積をかけ算器
MUML2でとることにより、かけ算器MULT2
の出力には、式に相当する信号が得られる。か
け算器MULT2の出力をローパスフイルタLPF
2に加えることにより、その出力には式のCに
相当する直流分が得られる。またトランス3の出
力を90゜移相器PS2に加えて90゜位相をシフト
させ、90゜位相器P2とバンドパスフイルタF1
の出力の積をかけ算器MULT4でとることによ
り、かけ算器MULT4の出力には式に相当す
る信号が得られる。かけ算器MULT4の出力を
ローパスフイルタLPF4に加えることにより、そ
の出力には式のDに相当する直流分が得られ
る。
ローパスフイルタLPF2とローパスフイルタ
LPF4の出力を加算器ADD2で加算することに
より、その出力には式に相当するQが得られ
る。
一方、バンドパスフイルタF2の出力と発振回
路OSCC出力との積をかけ算器MULT5でとるこ
とによりかけ算器MULT5の出力には式に相
当する信号が得られる。かけ算器MULT5の出
力をローパスフイルタLPF5に加えることによ
り、その出力には式のFに相当する直流分が得
られる。一方、OSCC出力を90゜移相器PS3で
90゜位相をシフトさせその出力とバンドパスフイ
ルタF2の出力との積をかけ算器MULT6でとる
ことによりその出力には式に相当する信号が得
られる。かけ算器MULT6の出力をローパスフ
イルタLPF6に加えることにより、その出力には
式に相当する直流分Gが得られる。
一般にe0e1<E0 2となるため、ローパスフイル
タLPF5の出力は利得E0 2/e0e1のアンプAMP1
にて増幅する。またローパスフイルタLPF6の出
力は利得E0 2/e0e1・ω0/ω1のアンプAM2に
て増幅する。
アンプAMP1の出力を係数器CF3で1/2倍
し、引算器SUB1で加算器ADD1の出力との差
をとる。一方、アンプAMP2の出力を係数器CF
4で2/√3倍し、加算器ADD3にて加算するとそ
の出力OUT1には式′に相当する対地浮遊容量
C1に相当する値を得る。一方引算器SUB1の出
力と係数器CF4の出力との差を引算器SUB2で
とることにより、その出力OUT2には式′に相
当する対地浮遊容量C2に比例した値を得る。
加算器ADD2の出力とアンプAMP2の出力を
係数器CF1で1/2倍した値との差を引算器SUB4
でとる。アンプAMP1の出力を係数器CR2で2/
√3倍し、それとの和を加算器ADD4でとること
によりその出力OUT3には′式に相当する絶縁
抵抗R1に逆比例した値を得る。またひき算器
SUB4と係数器CF2との差をひき算器SUB3で
とることによりその出力OUT4には式′に相当
する、絶縁抵抗R2に逆比例した値を得る。
上述の通り本発明の方法は、零相変流器
ZCT、発振器トランスTR2等を分割型鉄心とす
れば接地線を切断することなく容易に絶縁抵抗、
対地浮遊容量を検出することができる。またトラ
ンスTR2と発振回路OSCCとの接続回路に直列
抵抗を挿入することにより、この直列抵抗の両端
電圧によつて零相変流器ZCT出力を兼用するこ
ともできる。
本発明の方法によればどの非接地電路が不良か
すぐに判断できるため保安業務に対する効果は著
しいものである。Substituting [formula] into the formula and rearranging it, we get becomes. FIG. 2 shows an embodiment for obtaining voltages corresponding to ' to ' from the outputs of the filters F 1 and F 2 and the oscillator OSCC of FIG. 1. Multiplier for R phase voltage via transformer TR4
In addition to one input of MULT1, by adding the output of filter F1 to the other input terminal, a multiplier can be created.
A signal corresponding to the formula is obtained at the output of MULT1. By adding the output of the multiplier MULT1 to the low-pass filter LPF1, the DC component A expressed by the formula is obtained at its output. Similarly, the T-phase voltage is applied to multiplier MULT3 via transformer TR3.
In addition to one input end of the
Multiplier MULT3 by adding the output of F 1
A signal corresponding to Eq. is obtained at the output of . The output of multiplier MULT3 is passed through low pass filter LPF3.
By adding , a DC component B corresponding to the equation is obtained in the output. low pass filter LPF
By adding the outputs of 1 and LPF2 to the respective inputs of adder ADD1, a signal P expressed by the following equation is obtained at the output of adder ADD. The output of transformer TR4 is applied to 90° phase shifter PS1 to shift the phase by 90°, and the product of the output of 90° phase shifter PS1 and bandpass filter F1 is multiplied by a multiplier.
Multiplier MULT2 by taking with MUML2
At the output, a signal corresponding to Eq. The output of multiplier MULT2 is passed through low pass filter LPF.
2, the DC component corresponding to C in the equation is obtained in the output. In addition, the output of transformer 3 is applied to a 90° phase shifter PS2 to shift the phase by 90°, and a 90° phase shifter P2 and a bandpass filter F1
By taking the product of the outputs in multiplier MULT4, a signal corresponding to the expression is obtained at the output of multiplier MULT4. By adding the output of the multiplier MULT4 to the low-pass filter LPF4, a DC component corresponding to D in the equation is obtained in the output. Low pass filter LPF2 and low pass filter
By adding the outputs of the LPF4 with the adder ADD2, the output has a Q corresponding to the equation. On the other hand, by multiplying the output of the bandpass filter F2 and the output of the oscillation circuit OSCC by the multiplier MULT5, a signal corresponding to the formula is obtained at the output of the multiplier MULT5. By adding the output of the multiplier MULT5 to the low-pass filter LPF5, a DC component corresponding to F in the equation is obtained in the output. On the other hand, the OSCC output is connected to the 90° phase shifter PS3.
By shifting the phase by 90 degrees and multiplying the output by the output of the bandpass filter F2 using a multiplier MULT6, a signal corresponding to the equation can be obtained at its output. By adding the output of the multiplier MULT6 to the low-pass filter LPF6, a DC component G corresponding to the equation is obtained at the output. Generally, e 0 e 1 < E 0 2 , so the output of the low-pass filter LPF5 is the amplifier AMP1 with gain E 0 2 /e 0 e 1.
Amplify it. The output of the low-pass filter LPF6 is amplified by an amplifier AM2 with a gain of E 0 2 /e 0 e 1 ·ω 0 /ω 1 . The output of the amplifier AMP1 is multiplied by 1/2 by the coefficient multiplier CF3, and the difference with the output of the adder ADD1 is obtained by the subtractor SUB1. On the other hand, the output of amplifier AMP2 is
4 is multiplied by 2/√3 and added by adder ADD3, and the output OUT1 has a stray capacitance to ground corresponding to the formula '.
Get the value corresponding to C 1 . On the other hand, by taking the difference between the output of the subtracter SUB1 and the output of the coefficient multiplier CF4 in the subtracter SUB2, a value proportional to the ground stray capacitance C 2 corresponding to equation ' is obtained as the output OUT2. Subtractor SUB4 calculates the difference between the output of adder ADD2 and the value obtained by multiplying the output of amplifier AMP2 by 1/2 using coefficient unit CF1.
Take it. The output of amplifier AMP1 is 2/
By multiplying by √3 and calculating the sum with the adder ADD4, the output OUT3 obtains a value inversely proportional to the insulation resistance R1 corresponding to the equation ''. Also a subtracter
By taking the difference between SUB4 and coefficient multiplier CF2 with subtractor SUB3, the output OUT4 obtains a value inversely proportional to insulation resistance R2 , which corresponds to equation '. As mentioned above, the method of the present invention uses a zero-phase current transformer.
If ZCT, oscillator transformer TR2, etc. are made with a split core, insulation resistance can be easily adjusted without cutting the grounding wire.
Can detect ground stray capacitance. Furthermore, by inserting a series resistor in the connection circuit between the transformer TR2 and the oscillation circuit OSCC, the output of the zero-phase current transformer ZCT can also be used by the voltage across the series resistor. According to the method of the present invention, it is possible to immediately determine which non-grounded electrical circuit is defective, so the effect on security operations is significant.
第1図は本発明の原理を示す説明図、第2図は
本発明の原理による実施例を示す図
TR1:三相変圧器(2次三角回路)、Z1〜3:
負荷、R1〜2:絶縁抵抗、C1〜2:対地浮遊容量、
ZCT:零相変流器、TR2:低周波発振トランス
又は結合トランス、OSCC:発振回路、F1,F2:
バンドパスフイルタ、TR3,4:トランス、
MULT1〜6:かけ算器、PS1〜3:移相器、
LPF1〜6:ローパスフイルタ、AMP1〜2:アン
プ、ADD1〜3:加算器、SUB1〜4:引算器、
CF1〜4:係数器。
Fig. 1 is an explanatory diagram showing the principle of the present invention, and Fig. 2 is a diagram showing an embodiment according to the principle of the present invention. TR1: Three-phase transformer (secondary triangular circuit), Z 1 to 3 :
Load, R1 ~2 : Insulation resistance, C1 ~2 : Stray capacitance to ground,
ZCT: Zero-phase current transformer, TR2: Low frequency oscillation transformer or coupling transformer, OSCC: Oscillation circuit, F 1 , F 2 :
Bandpass filter, TR3,4: transformer,
MULT1~6: Multiplier, PS1 ~3 : Phase shifter,
LPF 1~6 : Low pass filter, AMP 1~2 : Amplifier, ADD 1~3 : Adder, SUB 1~4 : Subtracter,
CF 1~4 : Coefficient unit.
Claims (1)
抵抗ならびに対地浮遊容量の測定において接地線
を貫通する低周波電圧の印加されているトランス
または低周波電圧を発振する発振器の発振トラン
スにより該トランスから商用角周波数ω0とはこ
となる角周波数ω1の電圧を印加し、接地線を貫
通する零相変流器出力に含まれる角周波数ω0に
おける漏洩電流と角周波数ω1における漏洩電流
を分離検出し、該角周波数ω0における漏洩電流
のうち、一対の非接地電路の位相における有効分
ならびに無効分をそれぞれの位相について求める
と共に該角周波数ω1における漏洩電流の有効分
ならびに無効分とを求めることにより、各非接地
電路の絶縁抵抗ならびに対地浮遊容量に比例した
量を活線状態で測定することを特徴とする三相3
線式電路の絶縁抵抗ならびに対地浮遊容量測定方
法。1. When measuring the insulation resistance and stray capacitance to ground of the secondary one-end grounded circuit of a three-phase three-wire transformer, use a transformer to which a low-frequency voltage is applied through the grounding wire or an oscillation transformer of an oscillator that oscillates a low-frequency voltage. A voltage with an angular frequency ω 1 different from the commercial angular frequency ω 0 is applied from the transformer, and the leakage current at the angular frequency ω 0 and the leakage current at the angular frequency ω 1 included in the zero-phase current transformer output that passes through the grounding wire are The current is detected separately, and the effective and reactive components of the leakage current at the angular frequency ω 0 are determined for each phase of the pair of ungrounded electrical circuits, and the effective and reactive components of the leakage current at the angular frequency ω 1 are determined for each phase. A three-phase three-phase three-phase three-phase method characterized in that an amount proportional to the insulation resistance and stray capacitance to ground of each ungrounded circuit is measured in a live line state by determining the
Method for measuring insulation resistance and stray capacitance to ground of wire electrical circuits.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP4504580A JPS56141568A (en) | 1980-04-04 | 1980-04-04 | Method for measuring insulation resistance and floating capacity to ground of single-phase 3-wire type electric circuit |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP4504580A JPS56141568A (en) | 1980-04-04 | 1980-04-04 | Method for measuring insulation resistance and floating capacity to ground of single-phase 3-wire type electric circuit |
Publications (2)
Publication Number | Publication Date |
---|---|
JPS56141568A JPS56141568A (en) | 1981-11-05 |
JPS6252823B2 true JPS6252823B2 (en) | 1987-11-06 |
Family
ID=12708389
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP4504580A Granted JPS56141568A (en) | 1980-04-04 | 1980-04-04 | Method for measuring insulation resistance and floating capacity to ground of single-phase 3-wire type electric circuit |
Country Status (1)
Country | Link |
---|---|
JP (1) | JPS56141568A (en) |
Families Citing this family (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS58198765A (en) * | 1981-12-14 | 1983-11-18 | Toyo Commun Equip Co Ltd | Simple measuring method of insulation resistance of live wire circuit |
JPS6154462A (en) * | 1984-08-24 | 1986-03-18 | Midori Anzen Kk | Measuring method of ground insulating resistance of cable way |
KR20180102542A (en) * | 2016-01-08 | 2018-09-17 | 미쓰비시덴키 가부시키가이샤 | Insulation resistance measuring device |
-
1980
- 1980-04-04 JP JP4504580A patent/JPS56141568A/en active Granted
Also Published As
Publication number | Publication date |
---|---|
JPS56141568A (en) | 1981-11-05 |
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