JPS6152176A - Vector controlling method of induction motor - Google Patents
Vector controlling method of induction motorInfo
- Publication number
- JPS6152176A JPS6152176A JP59173713A JP17371384A JPS6152176A JP S6152176 A JPS6152176 A JP S6152176A JP 59173713 A JP59173713 A JP 59173713A JP 17371384 A JP17371384 A JP 17371384A JP S6152176 A JPS6152176 A JP S6152176A
- Authority
- JP
- Japan
- Prior art keywords
- command
- speed
- frequency
- current
- torque
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
Description
【発明の詳細な説明】
〔発明の利用分野〕
本発明は誘導電動機のベクトル制御方法に係９、特に速
度検出器や磁束検出器を用いることなく誘導電動機を高
速応答性を確保して高精度に制御するベクトル制御方法
に関する。[Detailed Description of the Invention] [Field of Application of the Invention] The present invention relates to a vector control method for an induction motor9, and in particular, the present invention relates to a method for vector control of an induction motor, and in particular, to ensure high-speed response and high precision of an induction motor without using a speed detector or a magnetic flux detector. The present invention relates to a vector control method for controlling.
誘導電動機を周波数変換器ちるいはインバータを用いて
速度制御する装置において、高速応τ高精度な制御を可
能にするベクトル制御が知られている。ベクトル制御は
、かご形誘導電動機によシ直流電動機と同等の性能を得
る制御方式として発展してきたものでちゃ、いわば直流
機の整流機能を電気的制御により行い゛、磁束ベクトル
と電流ベクトルとの直交関係を作って回転速度を制御し
ようとするものである。Vector control, which enables high-speed response and highly accurate control, is known in devices that control the speed of an induction motor using a frequency converter or an inverter. Vector control has been developed as a control method that uses squirrel cage induction motors to achieve performance equivalent to that of DC motors.In other words, the rectification function of a DC motor is electrically controlled, and the magnetic flux vector and current vector are This is an attempt to control the rotational speed by creating an orthogonal relationship.
従来のベクトル制御方式は、すべり周波数制御を基本と
し、電動機の回転速度に応じてインバータの出力周波数
を制御する方式であるので、速度検出器（または回転位
置検出器）並びに検出器とインバータ装置間の信号ケー
ブルが必要となり、装置が複雑化したシ、また既設の誘
導機に適用する場合の手間が必要となるなどの欠点があ
った。The conventional vector control method is based on slip frequency control and controls the output frequency of the inverter according to the rotational speed of the motor. The disadvantages of this method are that it requires additional signal cables, which makes the device complicated, and that it takes time and effort to apply it to an existing induction machine.
一方、このような不具合を解決するため、電動機の端子
電圧を検出して電動機磁束を演算し、これに従って電動
機電流（１次電流）と周波数を制御して、速度センナを
使わない、いわゆるセンサレス速度制御方式が発表され
ている（例えば、昭和５９年電気学会全国大会予稿集ｐ
、７３１．ｒすベシ周波数を用いた誘導電動機のＰＧレ
スベクトル制御方式」、昭和５・７年電気学会全国大会
予；ｆａ’６集ｐ、７０９．「鍍導電勲機の二次鎖交磁
束演算による磁界オリエント制御方式（ＴＧを用いない
ベクトル制御））。On the other hand, in order to solve such problems, the motor magnetic flux is calculated by detecting the terminal voltage of the motor, and the motor current (primary current) and frequency are controlled accordingly, so-called sensorless speed control that does not use a speed sensor. Control methods have been announced (for example, Proceedings of the 1981 National Conference of the Institute of Electrical Engineers of Japan, p.
, 731. PG-less vector control method for induction motors using RS frequency, 1930 and 1930 IEEJ National Conference Preliminaries; FA'6, p. 709. "Magnetic field orientation control method using secondary linkage magnetic flux calculation (vector control without using TG) of the Kanden Denshu machine".
しかし、上述のセンサレス速度制御方弐においても、電
動機の電圧を検出するための絶縁変圧器における鉄心飽
和現象並びに磁束演算のだめの積分器におけるトリット
の問題が内在しており、このために、特に低周波運転時
において十分な検出演算Ｎ度が得られず、安定な運転が
行えないという不具合がある。However, even in the above-mentioned sensorless speed control method, there are inherent problems such as iron core saturation phenomenon in the isolation transformer for detecting the voltage of the motor and trit in the integrator for magnetic flux calculation. There is a problem that a sufficient detection calculation N degree cannot be obtained during frequency operation, and stable operation cannot be performed.
本発明は、速度検出器や電圧検出器等の検出器を用いる
ことなく、高精度で誘導電動機の回転速度を制御しうる
ベクトル制御方法を提供することを目的とする。An object of the present invention is to provide a vector control method that can control the rotational speed of an induction motor with high precision without using a detector such as a speed detector or a voltage detector.
上記目的を達成するために、本発明は誘導電動圧の位相
基準に対し同位相の成分に応じて変換器の出力電圧及び
周波数を制御し、かつ上記電流成分に応じて定まる周波
数指令に応じて電動機の回転速度を制御するようにした
点に特徴を有する。In order to achieve the above object, the present invention controls the output voltage and frequency of a converter according to components in the same phase with respect to the phase reference of the induced electric voltage, and controls the output voltage and frequency of a converter according to a frequency command determined according to the above current component. The feature is that the rotational speed of the electric motor is controlled.
次に、本発明による誘導電動機のベクトル制御方法の実
施例を図面に基づいて説明する。Next, an embodiment of a vector control method for an induction motor according to the present invention will be described based on the drawings.
第１図に本発明による誘導電動機のＰＷＭ （パルス幅
変調）方式を用いた制御回路を示す。第１図において、
符号１はインバータを示しており、ＰＷＭ制御回路３に
より制御されて所定パルス幅の出力電圧を誘導電動機２
に供給する。ＰＷＭ制御回路３は電圧指令演算器５から
の電圧指令ｖ１＊（正弦波信号）と搬送波信号（三角波
信号）を比較し、その比較結果に応じてインバータ１の
スイッチング素子をオン、オフ制御し、インバータ１の
出力電圧をパルス幅制御する。発振器４Ｉｒ！、加算点
１３からの周波数指令ω１に比例した周波数で振幅が一
定な正弦波信号（誘導起電力の位相基準）を出力する。FIG. 1 shows a control circuit using a PWM (pulse width modulation) method for an induction motor according to the present invention. In Figure 1,
Reference numeral 1 indicates an inverter, which is controlled by the PWM control circuit 3 and supplies an output voltage of a predetermined pulse width to the induction motor 2.
supply to. The PWM control circuit 3 compares the voltage command v1* (sine wave signal) from the voltage command calculator 5 with the carrier wave signal (triangular wave signal), controls the switching elements of the inverter 1 on and off according to the comparison result, The output voltage of the inverter 1 is pulse width controlled. Oscillator 4Ir! , outputs a sine wave signal (phase reference for induced electromotive force) with a constant amplitude and a frequency proportional to the frequency command ω1 from the addition point 13.
電圧指令演算器５は発振器４の出力信号と前記周波数指
令ω！＊を演算し、振幅が周波数指令ω−に比例して変
化する′ル圧指令ｖ１”を出力する。加算点６は電動機
回転速度の指令値ｎ＊と演算値弁の偏差を出力する。７
は速度偏差増幅器、８は電流検出器である。電流成分検
出器９は醒動機電流の成分であって、インバータ出力電
圧（誘導起電力）の位相基準に対して同位相の成分子、
を検出する。加算点１０は速度偏差増幅器７からのトル
ク指令Ｔ＊と前記同位相成分子、との偏差を取り出す。The voltage command calculator 5 receives the output signal of the oscillator 4 and the frequency command ω! *, and outputs a 'ruple pressure command v1' whose amplitude changes in proportion to the frequency command ω-.Addition point 6 outputs the deviation between the command value n* of the motor rotation speed and the calculated value valve.7
8 is a speed deviation amplifier, and 8 is a current detector. The current component detector 9 detects a component of the wake-up motor current, which is in phase with the phase reference of the inverter output voltage (induced electromotive force);
Detect. An addition point 10 takes out the deviation between the torque command T* from the speed deviation amplifier 7 and the same phase component element.
電流偏差増幅器１１は上記トルク指令Ｔ”と同位相成分
■、との偏差に応じて周波数制御信号Δωを出力する。The current deviation amplifier 11 outputs a frequency control signal Δω according to the deviation between the torque command T'' and the in-phase component (2).
変化率制限器１２は速度指令ｎ の変化率を制御し取り
出す。加算点１３は電流偏差増幅器１１及び変化率制限
器１２がらの信号Δω及びｎ＊＊を加算し、周波数指令
ω−を出力する。関数発生器１４は電流成分検出器、に
基づいてすべり周波数信号ωＢを出力する。加算点１５
は周波数指令信号ω、からすべり周波数信号らを減算し
、回転速度の推定値弁を出力する。The rate of change limiter 12 controls and takes out the rate of change of the speed command n. Addition point 13 adds signals Δω and n** from current deviation amplifier 11 and rate of change limiter 12, and outputs frequency command ω-. The function generator 14 outputs a slip frequency signal ωB based on the current component detector. Additional points 15
subtracts the slip frequency signal etc. from the frequency command signal ω, and outputs an estimated value of the rotational speed.
次に、本発明の基本であるインバータ制御方式の原理に
ついて述べる。誘導電動機の一般等価回路を第３図（ａ
）に示す。αは任意に選べるので、１次の漏れリアクタ
ンスを零にするようにαを定めると、第３図の）の等節
回路が得られる。ここに、α＝ Ｘ ｔ ／ ｘ、
・・・（１）ｘ′Ｉ＋１：Ｘ１＝Ｘ
ｍ＋ＸＩ ・・・（２）ｔ 、
（Ｘａ ＋ｘｌン（Ｘｌ＋Ｘ２ ）Ｘ２千
・・・（３）ｘ、ｌｌ
このとき、励磁電流ｉ≦と１次電流ｉＩ及びトルクＴの
関係は次式にて示される。Next, the principle of the inverter control method that is the basis of the present invention will be described. The general equivalent circuit of an induction motor is shown in Figure 3 (a
). Since α can be arbitrarily selected, if α is determined so as to make the first-order leakage reactance zero, the equinodal circuit shown in FIG. 3 can be obtained. Here, α=Xt/x,
...(1)x'I+1:X1=X
m+XI...(2)t,
(Xa +xln(Xl+X2)X2,000
(3) x, ll At this time, the relationship between the exciting current i≦, the primary current iI, and the torque T is expressed by the following equation.
” ” ”十子卦” ｔ） ：Ｔ。” ” Tenzi Trigrams t) :T.
Ｒ２＋ ｒ Ｓ ｘ ２ °°惰）３ｐ 愕
（Ｓｘ二）２
’ｒ＝ 〜 ・ −・
Ｉｌ’ｏｌ ２ ・・・（５ンω
Ｓ Ｒ１＋（Ｓ Ｘ２’） ”
Ｖ Ｉ ＝ Ｅ ｔ’ ＋ １３−ｔ（１＋ ’、
１“４′−三さヒ； 、 コ１５ ・・・（６ン
Ｒ２＋γ５ｘ２
Ｅ Ｉ’ ＝Ｔ Ｘ ｒａ Ｉ ｏ
・・・（７）ここに、ｐは極対数でちる。励磁電流ｉ
６が−定及び電動機定数が不変であると仮定すると、１
次電流ｉ＋及びトルクＴはすべ９角周波数ωＢのみの関
数である。したがって、トルクＴに応じてすべり角周波
数ωＳ及び１次ｕｉｒ流工」が一義的に定まる。すなわ
ち、すべり角周波数ωＳに応じてトルクＴ及び１次社流
Ｉ＋’を制御ｒることができることを意味する。R2+ r S x 2 °° inertia) 3p shock
(Sx2)2'r= ~ ・ −・
Il'ol 2...(5nω)
S R1+(S
1"4'-three sahi; , Ko15...(6nR2+γ5x2 E I' =T
...(7) Here, p is the number of polar logarithms. Excitation current i
Assuming that 6 is - constant and the motor constant is unchanged, then 1
The secondary current i+ and the torque T are all functions only of the angular frequency ωB. Therefore, the slip angular frequency ωS and the first-order UIR flow rate are uniquely determined according to the torque T. That is, it means that the torque T and the primary flow I+' can be controlled according to the slip angular frequency ωS.
一方、励磁電流Ｉ６が一定であるだめの条件は、電圧ｌ
Ｅ＋’ｌを周波数ω１に比例するように制御することに
より得られ、（６）式に示すように、１次電圧Ｖ＋に１
？：Ｆ、、抵抗降下凡Ｉ１１を加味することにより得ら
れるっ
以上の関係に基づいて、第１図に示すイノバータ制御回
路の判御動作について説明する。On the other hand, the condition for the excitation current I6 to be constant is that the voltage l
It is obtained by controlling E+'l so that it is proportional to the frequency ω1, and as shown in equation (6), 1 is applied to the primary voltage V+.
? The control operation of the inverter control circuit shown in FIG. 1 will be explained based on the above relationship obtained by taking into account the resistance drop I11.
まず、励磁ｉｉ流ＩＩｏ’ｌを一定に保つ動作について
述べる。発（辰器４は、周波数指令ω」”に比例した周
波数の正弦信号（訪導起心力の位相基準）を発生し、社
圧指令発生器５において周波数指令ωｌ＊と前述の正弦
波信号が乗算され、周波数指令の−に比例した大きさと
周波数をもつ起電力指令Ｃ１８が作られる。さらに、第
１図中に破線にて示すように、４流検出信号１１に基つ
いて起電力指令ｅ１“に１次抵抗降下Ｒｔｌ＋を加算し
、１次社圧指令Ｖ＋＊（３相正弦波信号）が取り出され
る。First, the operation of keeping the excitation flow IIo'l constant will be described. (The radial generator 4 generates a sine signal (phase reference of the induced centripetal force) with a frequency proportional to the frequency command ω'', and in the social pressure command generator 5, the frequency command ωl* and the above-mentioned sine wave signal are generated. The electromotive force command C18 is multiplied and has a magnitude and frequency proportional to - of the frequency command.Furthermore, as shown by the broken line in FIG. The primary resistance drop Rtl+ is added to the primary resistance drop Rtl+, and the primary pressure command V+* (three-phase sine wave signal) is extracted.
ＰνＹＭ制御回路３においでｒｌ：１次電圧指令Ｖ−と
搬送波信号・が比較さ１１．周知のパルス幅制御が行わ
れる。これによジインバータ１の出力賦圧はその基本疲
労の瞬時値が１次゛、ｄ圧指令Ｖｌ”に比例するように
制御される。このようにして電動機電圧は（６）式に従
い制御され、励磁電流ｌ Ｉｏ′ｌは一定に保たれる。In the PνYM control circuit 3, rl: primary voltage command V- and carrier wave signal are compared.11. Well-known pulse width control is performed. As a result, the output pressure of the di-inverter 1 is controlled so that the instantaneous value of its basic fatigue is linearly proportional to the d-pressure command Vl. In this way, the motor voltage is controlled according to equation (6). , the excitation current l Io'l is kept constant.
次に、電動機の４流、トルク及び回転速反の制御動作に
ついて述べる。Next, the control operations for the four currents, torque, and rotational speed of the electric motor will be described.
１次電流の大きさ１■１１．１次成流Ｈ、／の励磁電流
１ｏ’に直交な成分（励磁電流ＩＯ’に対して位相差が
９０度すなわち誘導起電力５１′に同位相の成分ＪＩ、
及びトルクＴと励磁電流ｌＩｏ’ｌの関係は次式にて示
される。Magnitude of primary current 1 ■ 11. Primary current H, /component perpendicular to excitation current 1o' (phase difference is 90 degrees with respect to excitation current IO', that is, component in phase with induced electromotive force 51') JI,
The relationship between the torque T and the excitation current lIo'l is expressed by the following equation.
波数ωＳに応じて制御可能である。またトルクＴと電流
成分Ｉｑは比例の関係にちり、１４をトルりとみなすこ
とができる。It can be controlled according to the wave number ωS. Further, the torque T and the current component Iq are in a proportional relationship, and 14 can be regarded as the torque.
Ｉ、は電流成分検出器９において次式に従い検出される
。I is detected by the current component detector 9 according to the following equation.
Ｉ 、 ＝８． ・ｉ、ｙ ＋ｅｐ ・ｉｐ
−（１１）ここに、１１１＝ｉυ
ｉβ＝ −二（ｉｖ ｉｗ）
ｉＵ−ｉｗ：Ｕ相〜Ｗ相電流の瞬時値
ｅ 、Ｉ＝ ｅ ｕ ＝−ｓｉｎω、ｔｅｆｉ ＝ 二
（ｅ ｖ −ｅｙ ） ＝ｃｏｓωＬｔでちり、ｅｕ−
ｅｗはＵ−Ｗ相の誘導起電力の位相基準信号であって、
振幅が一定、位相が起電力と同位相の信号である。なお
、電圧指令演算器５において、これらの信号と周波数指
令ω１＊、！：が乗算され、各相の起電力指令ｅ＊が作
られる。I, =8.・i, y +ep ・ip
-(11) Here, 111=iυ iβ= -2(iv iw) iU-iw: Instantaneous value e of U-phase to W-phase current, I= eu =-sinω, tefi = 2(ev-ey) = cosωLt, dust, eu-
ew is a phase reference signal of induced electromotive force of U-W phase,
This is a signal with a constant amplitude and the same phase as the electromotive force. Note that in the voltage command calculator 5, these signals and the frequency commands ω1*, ! : is multiplied, and the electromotive force command e* of each phase is created.
このようにして検出された・電流成分工、は、加算点１
０においてトルク指令Ｔ＊と突き合わされ、調節され、
これに従いトルクはＴ２に比例して制御される。この際
、■、−ω８系の制御ゲインを十分高く選ぶことにより
、トルりをＴ＊に追従して高速応答に制御することがで
きる。The current component detected in this way is an addition point of 1
0, compared with the torque command T* and adjusted,
Accordingly, the torque is controlled in proportion to T2. At this time, by selecting the control gains of the -ω8 system sufficiently high, the torque can be controlled to follow T* and have a high-speed response.
一方、すべり角周波数ω８は電流成分１．と（９）式の
関係があるため、■、に基づいてω［Ｉを求めることが
できる。これは発生器１４にて行われる。On the other hand, the slip angular frequency ω8 is the current component 1. Since there is a relationship in equation (9), ω[I can be found based on . This is done in generator 14.
なお、ω＋＋ Ｉ。％性の一例を第４図に示す。定格
トルク以内であれば、電流成分Ｉ、はすペシ角周波数ω
８にほぼ比例するため、関数発生器１４を省略すること
もできる。なお、ωＢに対する１次電流ＩＩ、１の特性
を同図に示す。■、と１ＩｔｌはωＳが正の範囲につい
てみれば略一致しており、へ
１工１１を用いてもωａｌｃ求められる。ｌＩｔ ｌは
電流検出信号１１の大きさより検出される。In addition, ω++ I. An example of the percentage is shown in FIG. If the torque is within the rated torque, the current component I, the angular frequency ω
Since the function generator 14 is approximately proportional to 8, the function generator 14 can be omitted. Note that the characteristic of the primary current II,1 with respect to ωB is shown in the same figure. (2) and 1Itl are approximately the same in the range where ωS is positive, and ωalc can be obtained even if 11 is used. lIt l is detected from the magnitude of the current detection signal 11.
△
回転速度の推定値（演算値）ｎは、このすべり角周波数
ωＳを用いて次式より求められる。△ The estimated value (calculated value) n of the rotational speed is obtained from the following equation using this slip angular frequency ωS.
／＼ ＊ ／＼
ｎ＝ｋ（ω！−ω８） ・・・（１２）こ
こ［、に：比例定数
この演算は加算点１５において行われる。/\*/\n=k(ω!-ω8) (12) Here [, to: proportionality constant This operation is performed at addition point 15.
連関偏差増幅器７からは、速度指令ｎ＊と推定値令との
偏差に応じてトルク指令Ｔ・が出力され、さらに前述の
ようにしてトルク指令Ｔ＊に比例ししたように、回転速
度推定値ｎは、１欠周波数ωＬからすべり角周波数ωＩ
ｉを差し引いて演算されるため、誘導電動機に特有なす
ベシによる回転速度の低下が補正される。したがって、
従来のＶ／ｆ制御に比べて高精度な速度制御を行うこと
ができる。The associated deviation amplifier 7 outputs a torque command T according to the deviation between the speed command n* and the estimated value command, and further outputs the rotational speed estimated value in proportion to the torque command T* as described above. n is the slip angular frequency ωI from the missing frequency ωL
Since the calculation is performed by subtracting i, the reduction in rotational speed due to the bias characteristic of induction motors is corrected. therefore,
It is possible to perform speed control with higher accuracy than conventional V/f control.
なお、変化率制御器１２は速度指令ｎ＊の変更に伴い、
周波数指令ωｔを変化させるためのもので、これにより
定常時において前述の周波数制御信号Δωがすベシ角周
波数ωａの指令としての意味をもつようになる。Note that the rate of change controller 12 changes the speed command n* according to the change in speed command n*.
This is for changing the frequency command ωt, so that the above-mentioned frequency control signal Δω has a meaning as a command for the angular frequency ωa during steady state.
以上のように本実施例によれば、電動機取付けの速度検
出器や電動機電圧検出器を用いることなしに、高速応答
高精度な速度制御を行うことができる。As described above, according to this embodiment, speed control with high speed response and high accuracy can be performed without using a speed detector or a motor voltage detector attached to the motor.
次に、第２図に本発明の他の実施例を示す。第１図のも
のと異なるところは、励磁電流Ｉ Ｉｏ’ Ｉを検出し
、その指令値工Ｏ＊ との偏差に応じてインバータ出力
電圧を制御するところにおる。前述のようにｌ Ｅｌ’
ｌ／ω１を一定に制御すれば励磁電流ＩＩｏ１を一定
に保つことができるが、実際にはインバータ１及びＰＷ
Ｍ制御回路３において、１次電圧指令Ｖ−から実際の出
力電圧Ｖｌまでの入出力特性に非線形性がちシ、出力電
圧１／ｌを必ずしも１次電圧指令Ｖ の通りに制御でき
ない。そのため励磁電流ＩＩｏ′１が変動することがあ
る。そこで、本実施例においては、励磁電流ＩＩｏ’ｌ
を検出し、この励磁電流ＩＩｏ’ｌが常に所定の値とな
るように１次電圧指令ｖ１＊の振幅を修正制御する。Next, FIG. 2 shows another embodiment of the present invention. The difference from the one in FIG. 1 is that the excitation current IIo'I is detected and the inverter output voltage is controlled according to the deviation from the command value O*. As mentioned above, l El'
If l/ω1 is controlled constant, the excitation current IIo1 can be kept constant, but in reality, the inverter 1 and PW
In the M control circuit 3, the input/output characteristics from the primary voltage command V- to the actual output voltage Vl tend to be nonlinear, and the output voltage 1/l cannot necessarily be controlled in accordance with the primary voltage command V1. Therefore, the excitation current IIo'1 may fluctuate. Therefore, in this embodiment, the exciting current IIo'l
is detected, and the amplitude of the primary voltage command v1* is corrected and controlled so that this excitation current IIo'l always has a predetermined value.
第２図にないて、１６は電動機電流の成分であってイン
バータ出力電圧（誘導起電力）の位相基準に対して９０
度位相差の励磁電流成分１１ｏ’ｌを検出する励磁電流
成分検出器、１７ｉｉ、励磁電流成分ＩＩｏ’ｌ とそ
の指令値Ｉｏとの偏差を取り出す加算点、１８はその偏
差を増幅する励磁電流偏差増幅器、１９は該増幅器１８
の出力信号と周波数指令ωＬ＊を乗算し、１次電圧指令
ｖ１２の振幅設定信号を出力する乗算器である。他の１
〜１５の部品は第１図のものと同一物であるので説明を
省略する。In Fig. 2, 16 is a component of the motor current, which is 90% relative to the phase reference of the inverter output voltage (induced electromotive force).
An excitation current component detector 17ii detects the excitation current component 11o'l with a degree phase difference, an addition point that extracts the deviation between the excitation current component IIo'l and its command value Io, and 18 an excitation current deviation that amplifies the deviation. amplifier, 19 is the amplifier 18;
This is a multiplier that multiplies the output signal by the frequency command ωL* and outputs the amplitude setting signal of the primary voltage command v12. other 1
Parts 1 to 15 are the same as those shown in FIG. 1, so their explanation will be omitted.
以下、本実施例の動作を述べる。まず、励磁電流検出器
１６は次式に従いｌＩｏ’ｌを検出する。The operation of this embodiment will be described below. First, the excitation current detector 16 detects lIo'l according to the following equation.
ＩＩｏ’１＝＝ｅ、ｙ−ｒａ−ｅａ−ｒｐここで、ｅ、
、ｅβ及びｉｌｌ、ｉβは前述した通υである。IIo'1==e,y-ra-ea-rpwhere, e,
, eβ and ill, iβ are the above-mentioned general υ.
次に、上述した通りに励磁電流指令値Ｉｏ と励磁電流
ＩＩｏ’ｌの偏差を励磁電流偏差増幅器１８において増
幅し、乗算器１９において周波数指令ω１″＆乗算して
１次電圧指令ｙ−の振幅設定信号を作る。電圧指令演算
器５は、この設定信号と発振器４からの誘導起電力の位
相基準信号を乗算し、Ｕ−Ｗ相の電圧指令Ｖｌを出力す
る。その他の動作は第１図において述べた内容と同一で
ちる。このようにして、励磁電流ＩＩｏ’ｌが常に励磁
電流指令値Ｉｏ”に一致するように制御されるため、前
述した励磁電流ＩＩｏ勺の変動を防止できる。Next, as described above, the deviation between the excitation current command value Io and the excitation current IIo'l is amplified in the excitation current deviation amplifier 18, and the multiplier 19 multiplies the frequency command ω1''& to obtain the amplitude of the primary voltage command y-. A setting signal is generated.The voltage command calculator 5 multiplies this setting signal by the phase reference signal of the induced electromotive force from the oscillator 4, and outputs the U-W phase voltage command Vl.Other operations are shown in FIG. In this way, the excitation current IIo'l is controlled so as to always match the excitation current command value Io'', so the above-described fluctuation in the excitation current IIo' can be prevented.
なお、以上に述べた各実施例においては、アナログ制御
回路を用いた例について説明したが、マイクロプロセッ
サによるディジタル制御を用いた装置においても本発明
を適用でき、前述と同様の効果が得られる。また、以上
の各実施例は、ＰＷＭインバータ装置への適用例である
が、１ＦＬＵ形インバータ及びサイクロコンバータなど
の他の変換器を用いた装置にも同様に適用できることは
いうまでもない。In each of the embodiments described above, an example using an analog control circuit has been described, but the present invention can also be applied to a device using digital control by a microprocessor, and the same effects as described above can be obtained. Moreover, although each of the above embodiments is an example of application to a PWM inverter device, it goes without saying that it can be similarly applied to devices using other converters such as a 1FLU type inverter and a cycloconverter.
以上述べた如く、本発明によれば、速度検出器や電圧検
出器等の検出器類を用いることなく、高精度で誘導電動
機の回転速度を制御することができる。As described above, according to the present invention, the rotational speed of an induction motor can be controlled with high precision without using any detectors such as a speed detector or a voltage detector.
第１図は本発明の一実施例を示す制御回路ブロック図、
第２図は本発明の他の実施例を示す１σり御回路ブロッ
ク図、第３図（ａ）は誘導電動機の一般等価回路、（ｂ
）は１次漏れリアクタンスを消去した場合の等価回路、
第４図は制御動作を説明するためのすベシ角周波数と電
流との関係を示す特性図である。
１゛・・・インバータ、２・・・誘導電動機、４・・・
発振器、冠流成分検出器、１１・・・電流偏差増幅器、
Ｉｑ・・・電流成分、ωｌ・・・周波数指令。FIG. 1 is a control circuit block diagram showing one embodiment of the present invention;
FIG. 2 is a block diagram of a 1σ control circuit showing another embodiment of the present invention, FIG. 3(a) is a general equivalent circuit of an induction motor, and FIG.
) is the equivalent circuit when the primary leakage reactance is eliminated,
FIG. 4 is a characteristic diagram showing the relationship between the overall angular frequency and the current for explaining the control operation. 1... Inverter, 2... Induction motor, 4...
Oscillator, coronary flow component detector, 11... current deviation amplifier,
Iq...Current component, ωl...Frequency command.
Claims (1)
出力電圧を電圧指令に応じて制御する誘導電動機のベク
トル制御方法において、前記誘導電動機の１次電流の成
分であつて前記変換器の出力電圧の位相基準に対し同位
相の電流成分に応じて前記変換器の出力電圧及び周波数
の制御し、かつ上記電流成分に応じて定まる周波数指令
に応じて電動機の回転速度を制御することを特徴とする
誘導電動機のベクトル制御方法。1. In a vector control method for an induction motor in which the output voltage of a converter that supplies variable frequency alternating current to the induction motor is controlled according to a voltage command, a component of the primary current of the induction motor that is the output of the converter. The output voltage and frequency of the converter are controlled according to a current component that is in phase with respect to a voltage phase reference, and the rotational speed of the motor is controlled according to a frequency command determined according to the current component. vector control method for induction motors.
Priority Applications (4)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP59173713A JPS6152176A (en) | 1984-08-21 | 1984-08-21 | Vector controlling method of induction motor |
| EP85110379A EP0175154B1 (en) | 1984-08-21 | 1985-08-19 | Method of controlling inverter-driven induction motor |
| DE8585110379T DE3584603D1 (en) | 1984-08-21 | 1985-08-19 | METHOD FOR CONTROLLING AN INDUCTION MOTOR DRIVEN BY A INVERTER. |
| US06/766,945 US4680526A (en) | 1984-08-21 | 1985-08-19 | Method of controlling inverter-driven induction motor |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP59173713A JPS6152176A (en) | 1984-08-21 | 1984-08-21 | Vector controlling method of induction motor |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS6152176A true JPS6152176A (en) | 1986-03-14 |
| JPH0344509B2 JPH0344509B2 (en) | 1991-07-08 |
Family
ID=15965748
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP59173713A Granted JPS6152176A (en) | 1984-08-21 | 1984-08-21 | Vector controlling method of induction motor |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPS6152176A (en) |
Cited By (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH01110092A (en) * | 1987-10-22 | 1989-04-26 | Fuji Electric Co Ltd | Variable speed controller for induction motor |
| JPH0232788A (en) * | 1987-06-12 | 1990-02-02 | Hitachi Ltd | Control of motor and motor control device |
| JPH02184286A (en) * | 1989-01-09 | 1990-07-18 | Toyo Electric Mfg Co Ltd | Speed detection system of induction motor |
| JPH06225574A (en) * | 1994-01-10 | 1994-08-12 | Hitachi Ltd | Method and apparatus for controlling motor |
| EP1075078A3 (en) * | 1999-08-05 | 2003-12-03 | Sew-Eurodrive GmbH & Co. KG | Control method for generation of voltage vector and control system for an inverter |
-
1984
- 1984-08-21 JP JP59173713A patent/JPS6152176A/en active Granted
Cited By (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH0232788A (en) * | 1987-06-12 | 1990-02-02 | Hitachi Ltd | Control of motor and motor control device |
| JPH01110092A (en) * | 1987-10-22 | 1989-04-26 | Fuji Electric Co Ltd | Variable speed controller for induction motor |
| JPH0773440B2 (en) * | 1987-10-22 | 1995-08-02 | 富士電機株式会社 | Variable speed controller for induction motor |
| JPH02184286A (en) * | 1989-01-09 | 1990-07-18 | Toyo Electric Mfg Co Ltd | Speed detection system of induction motor |
| JPH06225574A (en) * | 1994-01-10 | 1994-08-12 | Hitachi Ltd | Method and apparatus for controlling motor |
| EP1075078A3 (en) * | 1999-08-05 | 2003-12-03 | Sew-Eurodrive GmbH & Co. KG | Control method for generation of voltage vector and control system for an inverter |
Also Published As
| Publication number | Publication date |
|---|---|
| JPH0344509B2 (en) | 1991-07-08 |
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