JPS597259B2 - Kouatsuhatsu Seisouchi - Google Patents
Kouatsuhatsu SeisouchiInfo
- Publication number
- JPS597259B2 JPS597259B2 JP14506075A JP14506075A JPS597259B2 JP S597259 B2 JPS597259 B2 JP S597259B2 JP 14506075 A JP14506075 A JP 14506075A JP 14506075 A JP14506075 A JP 14506075A JP S597259 B2 JPS597259 B2 JP S597259B2
- Authority
- JP
- Japan
- Prior art keywords
- voltage
- flyback transformer
- waveform
- order
- high voltage
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
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- Details Of Television Scanning (AREA)
Description
【発明の詳細な説明】
本発明ぱテレビジョン受信機に用いられる高圧レギユレ
ーシヨンを改善した高圧発生装置に関するものである。DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a high voltage generator with improved high voltage regulation used in a television receiver.
従来より高圧レギユレーシヨンを改善する手段として種
々の方法が知られている。Various methods have been known to improve high pressure regulation.
例えば、可飽和リアクターを使用して高圧負荷変動に従
いパルス巾を変化して高圧を安定化する方法、高圧負荷
変動に従い直流電源電圧を変化てせる方法、高圧整流器
の後にブリーダー抵抗を接続する方法等がある。しかし
、これらの方法でぱ無駄な消費電力を必要以上に要した
ヤ、部品点数が多くコスト高になる等の欠点があつた。
以下ブリーダー抵抗を接続する方法を例にとり説明する
。第１図ぱ一般的な水平出力回路を示し、ここで１ぱ水
平出力トランジスター、２ぱダンパーダイオード、３ぱ
共振コンデンサー、４ぱ偏向ヨーク、５ぱＳ字補正及び
直流カットコンデンサー、６ぱフライバックトランス、
□ばフライバックトランプ スの一次巻線、８ぱ二次巻
線、９ぱ高圧整流ダイオードをそれぞれ示す。For example, a method of using a saturable reactor to stabilize high voltage by changing the pulse width according to high voltage load fluctuations, a method of changing the DC power supply voltage according to high voltage load fluctuations, a method of connecting a bleeder resistor after a high voltage rectifier, etc. There is. However, these methods have drawbacks such as unnecessary power consumption and high cost due to the large number of parts.
The method of connecting a bleeder resistor will be explained below as an example. Figure 1 shows a general horizontal output circuit, where 1 is a horizontal output transistor, 2 is a damper diode, 3 is a resonant capacitor, 4 is a deflection yoke, 5 is an S-curve correction and DC cut capacitor, and 6 is a flyback. Trance,
□ indicates the primary winding of the flyback transceiver, 8 indicates the secondary winding, and 9 indicates the high voltage rectifier diode.
これを等価回路で表わすと第２図の如くなる。This can be expressed as an equivalent circuit as shown in FIG.
ここでＳＷぱ水平トランジスター１とダンパーダイオー
ド２に相当し、Ｌ７は偏向ヨーク４とフラ麿 イバツク
トランス６の一次巻線７のインダクタンスを並列接続し
た等価インダクタンス、Ｃｌぱ共振コンデンサー、Ｌ２
ぱ一次巻線と二次巻線との間のリーケージインダクタン
ス、Ｃ２ぱ二次巻線の対地容量である。第２図の等価回
路において、ク ＳＷが開いた時の対地容量Ｃ２の両端
に発生する電圧ＶＣ２ぱ（１式で表わされる。ＥＢＴＳ
Ｉｃｔ２β２
ＶＣ２＝−−〔ｓｉｎ（αを−φ１）２ α（β”−α
’）５ α
一 −ｓｉｎ（βを−φ２）〕 ・・・・・・・・・（
１）βただし
ここで、α，βはこの共振回路の共振角周波数、αぱ基
本角周波数、βは高調波角周波数を表わし、β／αが奇
数のＮにおいて（２）式を満足する値の時走査期間にリ
ンギング電圧が生じないというのが良く知られた高調波
同調理論である。Here, SW corresponds to the horizontal transistor 1 and damper diode 2, L7 is the equivalent inductance obtained by connecting the inductance of the primary winding 7 of the deflection yoke 4 and the horizontal back transformer 6 in parallel, and Cl is the resonant capacitor.
P is the leakage inductance between the primary winding and the secondary winding, and C2 is the ground capacitance of the secondary winding. In the equivalent circuit of Fig. 2, the voltage VC2 generated across the ground capacitance C2 when the switch is open is expressed by the following equation: EBTS
Ict2β2 VC2=--[sin(α-φ1)2 α(β”-α
') 5 α - sin (β - φ2)] ・・・・・・・・・(
1) β Here, α and β are the resonant angular frequencies of this resonant circuit, α is the fundamental angular frequency, β is the harmonic angular frequency, and β/α is a value that satisfies equation (2) for an odd number of N. According to the well-known harmonic tuning theory, no ringing voltage occurs during the scanning period when .
（２）式に訃いて、Ｎ＝３が３次同調、Ｎ＝５が５次同
調であ！），．Ｎによ．！２Ｎ次同調と呼ばれている。Based on equation (2), N = 3 is 3rd order tuning, and N = 5 is 5th order tuning! ),. To N. ! This is called 2N-order tuning.
各次数によりβ／αはＴｒＴｓの比によつて変化する。
例えば、Ｔｒ−１２μＳｌｌ４μｓの場合（水平繰返し
周期は６３．５μｓとする）各次数でのβ／αは表１の
ようになる。上記（１）式かられかるように出力ＶＣ２
の基本波成分と高調波成分の比はα／βとなＶ１表１か
られかるように同調次数により決定されるほぼ一定の値
を持つ。For each order, β/α changes depending on the ratio of TrTs.
For example, in the case of Tr-12μSll4μs (the horizontal repetition period is 63.5μs), β/α at each order is as shown in Table 1. As can be seen from the above equation (1), the output VC2
The ratio of the fundamental wave component to the harmonic component is α/β, which is determined by the tuning order and has a substantially constant value as seen from V1 Table 1.
又、出力波形についてＩｊ（１）式かられかるように３
，７，１１次同調では第３図ａの如くノ（３次同調の場
合を図示）ＶＣ２の波形の中心では凸状波形となＶｌ５
，９，ｌ３次同調では第３図ｂの如く（５次同調の場合
を図示）凹状波形となる。In addition, as for the output waveform, as can be seen from equation Ij (1), 3
, 7th and 11th order tuning, the center of the VC2 waveform is a convex waveform, as shown in Figure 3a (the case of 3rd order tuning is shown).
, 9, l Third-order tuning results in a concave waveform as shown in FIG. 3b (the case of fifth-order tuning is shown).
ここでは３次，５次同調の場合を例にとｖ両者の比較と
レギユレーシヨンの関係を説明する。第３図Ａｉｊ３次
同調、ｂは５次同調の場合の出力波形を示したもので、
波形を見ても明らかなように５次同調波形は３次同調波
形に比べて幅が広い。この為、負荷を取つた時５次同調
の方が３次同調と比較してダイオード流通角が広くな勺
高圧レギユレーシヨンが良い事は明らかである。このよ
うに、レギユレーシヨンを良くする方法として５，９次
同調を用いる事が多くなつて来た。しかし、このままで
は第３図ｂかられかるように波形が双峰となつている為
、小電流領域では双峰部が第４図のようにけずり取られ
るまでの電圧変動率が大きい欠点があつた。Here, a comparison of both v and the relationship between regulation will be explained using the cases of third-order and fifth-order tuning as examples. Figure 3 shows the output waveform for Aij third-order tuning, and b shows the output waveform for fifth-order tuning.
As is clear from the waveform, the fifth-order tuned waveform has a wider width than the third-order tuned waveform. Therefore, when the load is removed, it is clear that 5th-order tuning is better than 3rd-order tuning because it has a wider diode flow angle and higher voltage regulation. In this way, 5th and 9th order tuning has been increasingly used as a method to improve regulation. However, as it is, the waveform becomes double peaks as shown in Figure 3b, so in the small current region there is a drawback that the voltage fluctuation rate is large until the double peaks are cut off as shown in Figure 4. Ta.
この影響を無くす為、第５図に示すように高圧ダイオー
ドの後にブリーダー抵抗１０を挿入して常に一定電流を
流し双峰部を切ｖ取つた状態で用いなければ十分な効果
を得る事は出来ない。この模様を第６図に示す。Ａがブ
リーダー抵抗の無い場合、Ｂはブリーダー抵抗有の場合
のレギユレーシヨン特性である。このように、ブリーダ
ー抵抗を入れればブラウン管電流以外の電流を流す為、
ブリーダー抵抗１０の抵抗値をＲとし高圧をＥＨＴとす
るとＥＨＴ２／Ｒの電力損失を生じる事となる。その上
高圧部よりアースに抵抗を接続する為、十分な耐圧を持
つた高圧抵抗を使用しなければならず、絶縁にも十分な
考慮を払わなければならないから当然コストも上がｖ信
頼性にも欠ける事となる。本発明の目的は上記した従米
技術の欠点を無くし、有効に高圧レギユレーシヨンを改
善することにある。In order to eliminate this effect, a bleeder resistor 10 is inserted after the high-voltage diode as shown in Figure 5, and a constant current is constantly passed and the double peaks are cut off. Otherwise, a sufficient effect cannot be obtained. do not have. This pattern is shown in FIG. A is the regulation characteristic when there is no bleeder resistance, and B is the regulation characteristic when there is bleeder resistance. In this way, if you insert a bleeder resistor, a current other than the cathode ray tube current will flow, so
If the resistance value of the bleeder resistor 10 is R and the high voltage is EHT, a power loss of EHT2/R will occur. Furthermore, since the resistor is connected to the ground from the high voltage section, a high voltage resistor with sufficient withstand voltage must be used, and sufficient consideration must be given to insulation, which naturally increases cost and reduces reliability. will also be missing. The object of the present invention is to eliminate the above-mentioned drawbacks of conventional techniques and to effectively improve high pressure regulation.
５，９，１３次等の（４Ｋ＋１）次同調では出力波形の
中心で基本波と高調波の位相が反対とな９出力波形は双
峰波形とな９中心でへこみを生じるが、このへこみの程
度は高調波成分の大きさにより変化する。In (4K+1) order tuning such as 5th, 9th, and 13th orders, the phase of the fundamental wave and harmonics are opposite at the center of the output waveform, and the 9 output waveform is a bimodal waveform with a dent at the 9 center. The degree changes depending on the magnitude of the harmonic components.
この関係を５次同調の場合を例にとｖ示したのが第７図
である。ここで、Ｐは基本波成分に対する高調波成分の
比である。前述の（１）式と表１のβ／αよりわかるよ
うにＰはほぼ０．２２であＶ１深いへこみを生じている
が高調波成分をなんらかの方法で小Ａ〈寸Ａ車が出乎れ
げへｉム量を減少せしめる事が出米る。又、３，７，１
１次等の（４Ｋ−１）次同調では逆にとがｖ波形となる
が、同様に高調波成分を小さくすればとがりの程度は小
さくなる。本発明の特徴はフライバツクトランスの一次
巻線と直列に周波数特性を持つた減衰回路を挿入し、高
調波成分を効果的に減衰せしめ出力波形のへこみもしく
はとがり特性をやわらげ高圧レギユレーシヨンを改善し
ようとすることにある。FIG. 7 shows this relationship using the case of fifth-order tuning as an example. Here, P is the ratio of harmonic components to fundamental components. As can be seen from the above equation (1) and β/α in Table 1, P is approximately 0.22, causing a deep dent in V1. It is possible to reduce the amount of heat generated. Also, 3, 7, 1
Conversely, in (4K-1) order tuning such as first order, the peak becomes a v waveform, but similarly, if the harmonic components are made smaller, the degree of the peak becomes smaller. The feature of the present invention is to insert an attenuation circuit with frequency characteristics in series with the primary winding of the flyback transformer, to effectively attenuate harmonic components, soften the concave or peaked characteristics of the output waveform, and improve high voltage regulation. It's about doing.
以下図において本発明を説明する。The invention will be explained with reference to the following figures.
第８図は本発明の一実施例の水平偏向高圧発生回路を示
すもので、第１図と同一物は同一番号を付し説明を省略
する。本発明の実施例と第１図に示す従来例との違いは
フライバツクトランス６の一次巻線７の低圧側とＢ電圧
との間にコイル１１、抵抗１２よりなるＬＲ並列回路を
接続した点にある。FIG. 8 shows a horizontal deflection high voltage generating circuit according to an embodiment of the present invention. Components that are the same as those in FIG. The difference between the embodiment of the present invention and the conventional example shown in FIG. 1 is that an LR parallel circuit consisting of a coil 11 and a resistor 12 is connected between the low voltage side of the primary winding 7 of the flyback transformer 6 and the B voltage. It is in.
上記ＬＲのインピーダンス周波数特性を示したのが第９
図でａ曲線はＬ，ｂ曲線ぱＲのインピーダンス曲線を示
しているＬＲのインピーダンス関係よりわかるように、
基本波成分（ｄ成分）の大部分ぱインピーダンスの低い
Ｌを流れＲを流れる割合は少ない為抵抗損失は小さくダ
ンピングはほとんど受けないが、高調波成分（β成分）
ぱ大部分インピーダンスの低いＲを流れる為大きな抵抗
損失を生じる。The 9th one shows the impedance frequency characteristics of the above LR.
In the figure, the a curve shows the impedance curve of L, b curve and R. As can be seen from the LR impedance relationship,
Most of the fundamental wave component (d component) flows through L, which has low impedance, and a small proportion flows through R, so the resistance loss is small and there is almost no damping, but the harmonic component (β component)
Since most of the current flows through R, which has low impedance, a large resistance loss occurs.
この抵抗損失の為、フライバツクトランスの帰線出力波
形の高調波成分はダンピングを受け、基本波成分に比べ
て大きく減衰する。この為、出力波形のへこみ量（（４
Ｋ＋１）次同調の場合）もしくばとがｖ量（（４Ｋ−１
）次同調の場合Ｘは小さくなジ波形は平担に近づき小電
流領域での電圧変動を押さえる事が出米る。な｝、高調
波成分の減衰量はＬＲの各定数を適当に選ぶ事により調
整出来るが、ＬＲダンピング効果のききを良くし抵抗損
失を出米るだけ押さえるには、αＬ＜ＲくβＬの関係に
ＬＲ定数を選ぶ事が望ましい。第８図の実施例ではＬＲ
並列回路をフライバツクトランスの一次巻線の低圧側に
挿入したが、第１３図に示す如く一次巻線の中間に接続
しても同一効果が得られる事ぱいうまでも無い。Due to this resistance loss, the harmonic components of the retrace output waveform of the flyback transformer are damped and are greatly attenuated compared to the fundamental wave components. For this reason, the amount of dent in the output waveform ((4
In the case of K+1) order tuning) or the value of v ((4K-1
) In the case of second-order tuning, the j waveform with a small X becomes almost flat, making it possible to suppress voltage fluctuations in the small current region. The amount of attenuation of harmonic components can be adjusted by appropriately selecting each constant of LR, but in order to improve the LR damping effect and suppress resistance loss to a certain extent, the relationship αL < R × βL is established. It is desirable to select the LR constant. In the embodiment of FIG.
Although the parallel circuit is inserted on the low voltage side of the primary winding of the flyback transformer, it goes without saying that the same effect can be obtained by connecting it to the middle of the primary winding as shown in FIG.
又、本発明では高圧側に接続しても同一効果ぱ得られる
。又、第１０図は他の実施例を示したもので、フライバ
ツクトランスの二次巻線を８１・・・８１−１ノ８ｎと
高圧整流ダイオード９によりｎ分割した場合である．こ
の場合ｎケの二次巻線をそれぞれどういう高調波次数で
同調させるかは任意であＶ１このような複数同調の場合
にもＬＲ定数を適当に選んでやればレギユレーシヨンの
改善が出米る事はいうまでもない。Further, in the present invention, the same effect can be obtained even when connected to the high voltage side. FIG. 10 shows another embodiment in which the secondary winding of the flyback transformer is divided into n parts by 81...81-1 and 8n and a high-voltage rectifier diode 9. In this case, the harmonic order at which each of the n secondary windings is tuned is arbitrary.V1 Even in the case of multiple tuning like this, the regulation can be improved by appropriately selecting the LR constant. Needless to say.
又、必要なら複数個直列に接続しても良い。本発明は以
上のように、従米のように数多くの高価な部品を必要と
せず安価なコイル、抵抗のみですみ大きなコストダウン
が可能となる上信頼性も向上する。Moreover, if necessary, a plurality of them may be connected in series. As described above, the present invention does not require a large number of expensive parts unlike the conventional method, and requires only an inexpensive coil and resistor, making it possible to significantly reduce costs and improve reliability.
又、消費電力についてもＱの高い共振を減衰させる方法
なので少なくてすみ大巾な低減が可能である。更に、走
査期間に発生する有害な振動（リンギング）を抑圧出来
リンギングによる損失の増加、他回路への誘導妨害、水
平水カトランジスノ一のコレクノ一電流の増加等を防止
する事が出来る。この事を第１１図，第１２図について
説明する。第１１図はフライバツクトランス６の二次側
出力電圧波形を水平１周期について示したもので、ａ図
ぱ本発明のＬＲ回路の無い場合、ｂ図は本発明のＬＲ回
路を挿入した場合を表わす。ＬＲ回路の無い場合ぱ走査
期間に発生する有害なリンギング電圧イは走査期間の間
ほとんど減衰なく継続する。このリンギング電圧の角周
波数をｒとすると、ｒ′−βの関係がある為、ＬＲ回路
を挿入すればｂ図に示すように帰線期間の出力波形が平
らとなるだけでなく走査期間のリンギング電圧イも減衰
する。第１２図は走査期間のリンギングの有害性と本発
明の効果を水平出力トランジスターのコレクター電流を
１例に取ｖ図示したもので、ａ図はＬＲ回路の無い場合
、ｂ図はＬＲ回路を挿入した場合のコレクノ一電流波形
を示す。走査期間にリンギング電流が重畳されると位相
によつてぱａ図に示すように最大コレクタ電流ＩｃＰは
ＩＣＰ２のように増加する。Furthermore, since the method is used to attenuate high-Q resonance, power consumption can be reduced significantly. Further, harmful vibrations (ringing) occurring during the scanning period can be suppressed, and it is possible to prevent an increase in loss due to ringing, induction interference to other circuits, and an increase in current in the horizontal water transducer. This will be explained with reference to FIGS. 11 and 12. FIG. 11 shows the secondary output voltage waveform of the flyback transformer 6 for one horizontal period, and FIG. represent. Without the LR circuit, the harmful ringing voltage that would occur during the scanning period would continue with little attenuation during the scanning period. If the angular frequency of this ringing voltage is r, there is a relationship of r'-β, so inserting an LR circuit not only flattens the output waveform during the retrace period as shown in figure b, but also eliminates the ringing during the scanning period. Voltage A also attenuates. Figure 12 illustrates the harmful effects of ringing during the scanning period and the effects of the present invention, taking the collector current of a horizontal output transistor as an example. The correct current waveform is shown in this case. When a ringing current is superimposed on the scanning period, the maximum collector current IcP increases as ICP2 depending on the phase as shown in Fig. a.
リンギング位相は非常に不安定で高圧負荷の変動、回路
定数のバラツキにより変化する為リンギング電流は小さ
くする事が必要とされている。本発明によれば走査期間
の終りではリンギング電流ぱｂ図に示すようにほとんど
雰に減衰させる事が出来１ｅｐの増加を防ぐ事が出米る
。The ringing phase is extremely unstable and changes due to fluctuations in the high voltage load and variations in circuit constants, so it is necessary to keep the ringing current small. According to the present invention, at the end of the scanning period, the ringing current can be attenuated to almost zero as shown in the graph, and an increase in 1ep can be prevented.
第１図は一般的な水平偏向高圧発生回路図、第２図ぱそ
の等価回路、第３図，第４図は同回路説明の為の波形図
、第５図ぱブリーダー抵抗を使用した水平偏向高圧発生
回路図、第６図は同回路説明の為の特性図、第７図は本
発明説明の為の波形図、第８図は本発明の一実施例を示
す水平偏向高圧発生回路図、第９図は本発明の動作原理
を説明する為のインピーダンス特性曲線、第１０図及び
第１３図は本発明の他の実施例を示す水平偏向高圧発生
回路図、第１１図，第１２図は本発面の他の効果を説明
する為の波形図である。
６・・・フライバツクトランス、７・・・一次巻線、８
・・・二次巻線、１１・・・コイル、１２・・・抵抗。Figure 1 is a general horizontal deflection high voltage generation circuit diagram, Figure 2 is the equivalent circuit of PAS, Figures 3 and 4 are waveform diagrams to explain the circuit, and Figure 5 is horizontal deflection using a PA bleeder resistor. A high voltage generation circuit diagram; FIG. 6 is a characteristic diagram for explaining the circuit; FIG. 7 is a waveform diagram for explaining the present invention; FIG. 8 is a horizontal deflection high voltage generation circuit diagram showing an embodiment of the present invention; FIG. 9 is an impedance characteristic curve for explaining the operating principle of the present invention, FIGS. 10 and 13 are horizontal deflection high voltage generation circuit diagrams showing other embodiments of the present invention, and FIGS. 11 and 12 are FIG. 3 is a waveform diagram for explaining another effect of the present invention. 6... Flyback transformer, 7... Primary winding, 8
...Secondary winding, 11...Coil, 12...Resistance.
Claims (1)
ックトランスの１次巻線に接続された水平出力トランジ
スターと、この水平出力トランジスターに並列に接続さ
れた偏向ヨーク及び共振コンデンサーと、フライバック
トランスの２次巻線に接続された高圧整流ダイオードを
具え、フライバックトランスの２次巻線に発生する帰線
出力電圧の基本波成分とその高調波成分を整流ダイオー
ドによつて整流することにより高電圧を発生する高圧発
生装置において、高調波成分に対し高インピーダンスに
なるコイルと、抵抗とからなる並列回路が上記１次巻線
に対して直列に接続され、帰線出力電圧の先端が平坦化
されていることを特徴とする高圧発生装置。1 At least a flyback transformer, a horizontal output transistor connected to the primary winding of this flyback transformer, a deflection yoke and a resonant capacitor connected in parallel to this horizontal output transistor, and a secondary winding of the flyback transformer. A high-voltage transformer is equipped with a high-voltage rectifier diode connected to the flyback transformer, and generates a high voltage by rectifying the fundamental wave component and its harmonic components of the retrace output voltage generated in the secondary winding of the flyback transformer. In the generator, a parallel circuit consisting of a coil that has high impedance against harmonic components and a resistor is connected in series with the primary winding, and the tip of the retrace output voltage is flattened. Characteristic high pressure generator.
Priority Applications (15)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP14506075A JPS597259B2 (en) | 1975-12-08 | 1975-12-08 | Kouatsuhatsu Seisouchi |
| FI763461A FI61377C (en) | 1975-12-08 | 1976-12-01 | HOEGSPAENNINGSGENERATOR |
| US05/747,324 US4112337A (en) | 1975-12-08 | 1976-12-03 | High voltage generator |
| CA267,235A CA1081844A (en) | 1975-12-08 | 1976-12-06 | High voltage generator |
| GB51005/76A GB1573808A (en) | 1975-12-08 | 1976-12-07 | High voltage generator |
| IT69923/76A IT1072145B (en) | 1975-12-08 | 1976-12-07 | HIGH VOLTAGE GENERATOR PARTICULARLY FOR TELEVISION |
| AU20307/76A AU499222B2 (en) | 1975-12-08 | 1976-12-07 | High voltage generator |
| MX767903U MX3504E (en) | 1975-12-08 | 1976-12-07 | HIGH VOLTAGE GENERATOR |
| TR19271A TR19271A (en) | 1975-12-08 | 1976-12-07 | HIGH VOLTAGE GENERATOERUE |
| FR7636843A FR2335112A1 (en) | 1975-12-08 | 1976-12-07 | VERY HIGH VOLTAGE GENERATOR |
| DE2655466A DE2655466B2 (en) | 1975-12-08 | 1976-12-07 | High voltage generator for cathode ray tubes, preferably for television receivers and EDP display devices |
| AR265751A AR216753A1 (en) | 1975-12-08 | 1976-12-07 | HIGH VOLTAGE GENERATOR |
| PH19213A PH12818A (en) | 1975-12-08 | 1976-12-08 | High voltage generator |
| HK591/81A HK59181A (en) | 1975-12-08 | 1981-12-03 | High voltage generator |
| MY184/82A MY8200184A (en) | 1975-12-08 | 1982-12-30 | High voltage generator |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP14506075A JPS597259B2 (en) | 1975-12-08 | 1975-12-08 | Kouatsuhatsu Seisouchi |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS5269523A JPS5269523A (en) | 1977-06-09 |
| JPS597259B2 true JPS597259B2 (en) | 1984-02-17 |
Family
ID=15376438
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP14506075A Expired JPS597259B2 (en) | 1975-12-08 | 1975-12-08 | Kouatsuhatsu Seisouchi |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPS597259B2 (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH0510176Y2 (en) * | 1987-03-30 | 1993-03-12 |
-
1975
- 1975-12-08 JP JP14506075A patent/JPS597259B2/en not_active Expired
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH0510176Y2 (en) * | 1987-03-30 | 1993-03-12 |
Also Published As
| Publication number | Publication date |
|---|---|
| JPS5269523A (en) | 1977-06-09 |
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