JPS59165980A - Vector control system of induction motor - Google Patents

Abstract

PURPOSE:To enable to control to always perpendicularly cross a magnetic flux to the secondary current including the interference variation due to the angular frequency of a power source by correcting to cancel the mutual interference due to the primary current. CONSTITUTION:A correction calculator 3 calculates the correction value for correcting the interference due to alpha-phase primary current i1alpha to the value passed through a coefficient unit 31 of the primary resistor r1 to alpha-phase primary current set value i*alpha to control the secondary magnetic flux lambda2 constantly for setting the beta-phase primary voltage e1beta. Further, it adds a correcting value for correcting the interference due to the beta-phase primary current i*1beta to the value passed through a coefficient unit 35 of the primary resistor r1 to the set value i*1alpha for setting the alpha-phase primary voltage e1. Thus, the motor 1 can be controlled for speed and torque in noninterference between the magnetic flux and the secondary current.

Description

【発明の詳細な説明】 本発明は、誘導直動機のベクトル制御方式に関する。[Detailed description of the invention] The present invention relates to a vector control system for an induction motor.

近年、誘導電動機の連応性ン向上する制御方式として、
電動機の一？：に電流乞励磁電流と二次電流とに５＋け
て制御し、二？Ｘ磁束と二次電流ベクトルを常に直交さ
せろことで直流機と同等の応答性２得ようとするベクト
ル制御方式が提案されている。In recent years, as a control method to improve the coordination of induction motors,
An electric motor? : The current is controlled by 5+ times the excitation current and the secondary current, and 2? A vector control method has been proposed that attempts to obtain the same responsiveness as a DC machine by always making the X magnetic flux and the secondary current vector orthogonal.

しかし、実際ｆ便世する電力変換装置にパルス幅’ＪＧ
［ｌ（ｐｗＭ）方式インバータなどの電圧形インパータ
ン便用−「ろと、−次電流ゲ闇制御すると言っても′電
圧が操作量となるため、周波数ン高くした冨速運転時に
設定通りの一次電流が流ｎなくなって応答性が悪くなり
、積度良い可変速制御が雛しくなる問題があった。However, in practice, the pulse width 'JG' is
[For voltage type impartan convenience such as l(pwM) type inverter - Even though it is said to control the secondary current, the voltage is the manipulated variable, so when operating at high speed with a high frequency, the There was a problem in that the current flow stopped, resulting in poor responsiveness and poor variable speed control with good integration.

本発明は、′電動機の一次藏圧匍１１′ＩｇＩにおいて
、二次磁束外と二次電流分との間に互いの干渉分ンキャ
ンセルで致ろベクトル匍制御とでろことにより、従来の
間趙点乞解消したベクトル制御方式ン提供ｆろことン目
的とする。The present invention has the advantage that, in the primary pressure force 11'IgI of the electric motor, interference vector control is performed between the secondary magnetic flux and the secondary current by canceling each other. The purpose of the present invention is to provide a vector control method that eliminates the need for control.

以下、本発明の詳細な説明に続いて実施ｉ＋１１　’２
詳：洲に説明ｆろ。The detailed description of the present invention is followed by implementation i+11 '2.
Details: Please explain to Su.

まず、読４審＠磯ンー矢電圧に同期して回転才ろα−β
軸で表わした電圧刀根式は以下の第山式になるし、発生
トルクＴは第（２）式になる。First of all, 4th referee @ Iso-n - Rotating in synchronization with the arrow voltage α-β
The voltage root equation expressed by the axis is the following equation, and the generated torque T is the equation (2).

（以下余白） Ｔ−一（λ′２βｌ／　２Ｃｔ−λ′２ｃｔ１′２β）
　　・・・・・・・２）ここで、各記号は以下に示ｆ渚
清である。(Left below) T-1 (λ'2βl/ 2Ct-λ'2ct1'2β)
......2) Here, each symbol is shown below.

ｅ、−一次区圧（α、β或汁）ｅ′２に欠４圧（α、β
或汁）１、電−次′心流（α、β成汁）λ′２；二／′
に出東（α、β成分）ｒ、；−矢抵抗　　　　　ｒ、暮
二矢低抗Ｍ　１１万／ｌ［インダクタンス　　Ｌ２；二
次インダクタンスＬ、Ｌ２−　ｙ２ Ｌσ；専＋１Ｉｉｉ湘れインダクタンス（Ｌσ＝□）２ Ｌｌ；−次インダクタンス　　Ｐｉｅ５＋ｊ己号−４士
５ ω】；電源角叫彼故　　　ｂ、１；ロータ川周波数１′
２−二仄こ流（α、β成分） 上述のｔｌｌ　、　＋２１式はブロック線図で表わすと
第１図に示すようになり、二相鑞圧ｅ　　、ｅ　　に対
し１α　　１β て−／に電Ｚ危と二次磁束のα軸、β軸成分１．ヶ、１
．β。e, - Primary pressure (α, β or juice) e′2 is missing 4 pressure (α, β
2/'
Inductance (α, β components) r, ;-arrow resistance r, Kurejiya low resistance M 110,000/l [Inductance L2; Secondary inductance L, L2- y2 Lσ; Special +1Iii submergence inductance (Lσ = □ )2 Ll; -order inductance Pie5+jself-4shi5 ω]; power supply angle therefore b, 1; rotor river frequency 1'
2-2 current (α, β components) The above equation tll, +21 is expressed in a block diagram as shown in Figure 1, and for two-phase solder pressures e and e, 1α 1β and -/ Z-axis and α-axis and β-axis components of secondary magnetic flux 1. 1, 1
．． β.

λ′２ａ、λ′２β及びトルクＴン発生する誘導直動機
の等価ブロック図になる。This is an equivalent block diagram of an induction direct motor that generates λ'2a, λ'2β and torque T.

ここで、−次道圧に同期して回るα、βｌ！ａはどのよ
５な立相（Ｃ定めても良いが、α軸化二次磁束の方間に
定めろと、二次電流がβ軸に一致する条件、′ｆなわち
、二？′に電流が磁束と直交でる条件はベクトル、１１
Ｉ、埋理論で明らかにされているように、λ′２β−〇 であり、かつ−矢周波叔ω。は である。Here, α, βl, which rotates in synchronization with the -next road pressure! a can be set in any five phases (C can be determined, but it should be set in the direction of the α-axis secondary magnetic flux, and the condition that the secondary current coincides with the β-axis, 'f, 2?') The condition for the current to be perpendicular to the magnetic flux is the vector, 11
I, as revealed by the buried theory, λ′2β−〇, and −arrow frequency wave ω. It is.

このよ５に、α、β＠３を定めると、−次也流１１のα
軸成す１．Ｃｔ（＝一定）は磁束λ′２ＶＣ相当する一
ｒＫ醒流であり、β軸成分１．β（エニ欠イ冗１′２　
に相当ｆる一次醒流となる。If α and β @ 3 are determined in this way 5, then α of -jiya style 11
Shaft 1. Ct (=constant) is a 1rK current corresponding to the magnetic flux λ'2VC, and the β-axis component 1. β (any lack of meaning 1'2
It becomes a primary wake flow corresponding to f.

次に、上述の３１　、１４１式の条件乞第１図のブロッ
ク線図に入れろと第２１ｇ１に示すブロック線図になる
。そなわち、第１図におげろ６点は・４）式の関Ｉ糸か
ら零に皿し卸されろ。０点はλ′２β＝０であるからこ
の点につなかろ者シ工芋て零であり、同様Ｋ　１′２（
ｆ＝０からｄ点につながる罐も零であるしλ′２ｉ−一
定であるからその機外であるｂ点も零である。そして、
第１図の破線ブロックへの部分ぞ同様の条件下で計ｉｆ
ると、Ｌ２／ｒ２−τ２　、　Ｌ（ｒ　＝　Ｌ（Ｌ＋Ｌ
２−Ｍ２）　／　Ｌ２３として、 一τ２Ｐ　＋１ ＝□Ｌ１ω０１１α τ２Ｐ＋１ ここで１１α＝一定であるからｐ　＝　ｎとおいて＝Ｌ
、ωｏ１．α このようにして、第１図のブロック線図は第２１メ１の
ブロック線図になる。第２図から明らかなように、二次
磁束λ’２ａはα相−次電圧ｅ、（１によって一義的に
設定できずにβ相−仄軍流１．βによる＋Ｌ、ω。１．
β分の干渉があるし、二次電流１２βはβ相−次′ボ圧
ｅ、βによって一猛的に設定できずにα相−次ル流１　
による−Ｌ、ω。１．ｃｔ　汁の干渉かある。Next, if the conditions of formulas 31 and 141 mentioned above are included in the block diagram of FIG. 1, the block diagram shown in No. 21g1 will be obtained. In other words, the 6 points in Figure 1 can be translated into zero from the Seki I thread of equation 4). 0 point is λ'2β = 0, so anyone who does not connect to this point is zero, and similarly K 1'2 (
The can connected from f=0 to point d is also zero, and since λ'2i is constant, point b, which is outside the machine, is also zero. and,
If the part to the dashed line block in Figure 1 is calculated under similar conditions,
Then, L2/r2-τ2, L(r = L(L+L
2-M2) / L23, - τ2P +1 = □L1ω011α τ2P+1 Here, since 11α = constant, p = n = L
,ωo1. α In this way, the block diagram of FIG. 1 becomes the block diagram of the 21st block diagram. As is clear from FIG. 2, the secondary magnetic flux λ'2a cannot be uniquely set by the α-phase-order voltage e,
There is interference by β, and the secondary current 12β cannot be set due to the β phase-next pressure e, β, and the
According to −L, ω. 1. ct There may be interference from the juice.

１α そこで、本発明においては、−次′市流１１゜及び１１
βによる干＃分χ予め補償した７Ｉｉ１し１ｇ１着にな
るよう一次電圧”、（ｔ　１　ｅｊβン桶正する。こり
梱正には一次電圧ｅ、ｃｔＶｃ力ロ算されろり。ω。１
．〕乞見込んで該電圧８．（Ｉの設定にり。ω。１．β
ン眞算しておき、−欠電圧ｅ、βに成算されるＬ１ω。1α Therefore, in the present invention, −next′ market flow 11° and 11
Correct the primary voltage so that it becomes 7Ii1 and 1g1 after pre-compensated by β.The primary voltage e and ctVc force are calculated for ω.1
．． ] Please note that the voltage 8. (Depending on the setting of I.ω.1.β
-L1ω is calculated and added to -shortage voltage e and β.

１．ｃｔン見込んで拶市圧ｅ、βの設定１ｃＬ、ω。１
．α？り０彦しておき、さらに電圧ｅ１βに対して電流
１．βが［１／　（Ｌ、Ｐ＋　ｒ、）　〕’ｙｊの一次
遅れ？伴なうことから該電圧ｅ、ｃｔの補正演算に遅れ
分も廿めた補正乞しておく。この帰旧により、二次磁束
λ′２ａと二次′ｆｔ鑞１２βぞ非干渉に訓ｆ卸する一
矢電圧ｅ１α＋ｅ１βを設定ｆることかで羨ろ。1. Set the market pressure e, β to 1 cL, ω, assuming ct. 1
．． α? 0hiko, and furthermore, the current 1. Is β the first-order lag of [1/ (L, P+ r,)]'yj? Therefore, the correction calculations for the voltages e and ct should be corrected to account for the delay. By this return, it is possible to set the voltage e1α+e1β to prevent the secondary magnetic flux λ'2a and the secondary flux 12β from interfering with each other.

第３図は本発明の一実施例乞示すブロック図である。電
動機１にｐＷＭ方式インバータ２から′直圧制御による
一矢覗圧乞供給して該遊動の１に磁束と二仄砿流とが硬
いＶｃｉσダするよう拙御するにおいて、α、β相電圧
ｅ１Ｃｔｌ　ｅ、βの設定に抽正演真回・各３によって
前述の補正ン施す。補正演算回・６３はα相−次電圧ｅ
、ｃｔの設定に、電動機の二仄逼東λ′２ぞ一定に制御
ｆろためのα相−次イ流設定；１σ１□に一次砥抗ｒ１
の係数４３．？通した値・て対してβ相−次市流１．β
による。第２図に示す干渉分ン補正ｆろための鋪正値ン
炊算しておく。この（ｉ＠旧値は、二次電流１′２ン制
御才ろためのβ相−矢′社流設定１１α１でβｖＣ第２
図に示す一次遅れ汁［１／（Ｌ、ｐ＋ｒ、）］　とその
係ｎｒ、Ｙ乗するＭ分器３□によって該１、βに対ｆる
遅れ汁補王１直１３＊＊ ぞ叙出し、この補正ｌｌｌＴ１．βに電源角周波斂ω０
乞乗算器３３で乗算し、この乗算結果に係数として等価
漏れインダクタンスＬσヶ持つ係数器３４乞ｙｆｊｌ　
して得る。FIG. 3 is a block diagram showing one embodiment of the present invention. When the electric motor 1 is supplied with direct voltage control from the pWM inverter 2, and is carefully controlled so that the floating magnetic flux and the two currents become hard Vciσ, the α and β phase voltages e1Ctl e The above-mentioned correction is applied to the settings of , β by 3 each of the abstraction and derivation times. Correction calculation time 63 is α phase-order voltage e
, ct setting, α phase-order current setting for constant control of the electric motor's 2nd axis λ'2; primary grinding force r1 at 1σ1□
The coefficient of 43. ? For the value passed through β phase - next market flow 1. β
by. The positive value for the interference correction shown in FIG. 2 is calculated in advance. This (i @ old value is the β phase for secondary current 1'2 - Ya' company style setting 11α1 and βvC 2nd value)
The first-order delayed juice [1/(L, p+r, )] shown in the figure and its coefficient nr, M divider 3□ raised to the power of Y are used to calculate the delayed juice corresponding to f for the 1 and β. , this correction lllT1. β has a power angular frequency difference ω0
The multiplier 33 multiplies the result, and the multiplier 34 has an equivalent leakage inductance Lσ as a coefficient.
and get it.

また、補正演算回路３は、β相−欠′区圧ｅ１βの設定
に設定１旧１．βに一次砥ｒにｒｌの係数器３，２通し
た値に対して、α相−１矢戚流１１．工による第２図に
示イ干渉鋒乞補正ｆろための＠ＩＥ値ン７ＪＯ算し不 てお（。このｐＨｌ正値は電流設定値１１αに解源用周
波数ω０ン乗算器３゜で乗算し、この栄算桔呆に係数と
して一次インダクタンスＬ１ヶ待つ；系叔６３７乞通し
て得ろ。Further, the correction calculation circuit 3 sets the β phase-missing section pressure e1β to 1 old 1. α phase-1 arrow flow 11. The @IE value for the interference correction shown in Fig. 2 is not calculated (this pHl positive value is multiplied by the current setting value 11α and the solution source frequency ω0 multiplier 3°). However, I will wait for the primary inductance L1 as a coefficient for this Eisan calculation; get it by asking 637 uncles.

β相−次藏流設定値１雪βは速度設定値４と電動機１に
結合ｆろ速度検出器４の検出１１σ（ロータ肉周波数ω
ｒ）との偏差ン比（り１１積汁演算（ｐ’ｒ）ｆろ速度
調節器５の出力として得ろ。也源角周汲叔ωＯは角周彼
数演算回＠６によって得る。この演算回路６は、設定値
１雪βに第２図の遅れ汁を補正した設定１１ｒ１）と設
定１ｉ　ｉ１αの除算ンする割算器６．と、この除′ａ
皓果１１β／１１Ｃｔｖｃ係敬１　／　７２　Ｙ　掛算
ｆ、ｂ係数器６２　と７有してすべり角周波数ω８ヶ算
出し、このすべり角周波数ωＳにロータ角周波数ω、ン
力り算して屯諒角周彼数ω０娑得ろ。この割算器６１　
と係数器６２Ｖｃよるすべり角周波数ω８の算出は、前
述の、４１式右辺第２項中に前述の次の条件及び第２図
からλ’２ｄ　＝　１４ａ・Ｍ　’！ｌ’代入して１．
ｐ／（１，（１’τ２）に置換さ肚ろ。β phase - Next flow set value 1 snow β is coupled to speed set value 4 and motor 1 f Detection 11σ of speed detector 4 (rotor flesh frequency ω
Obtain the deviation ratio (p'r) from the angle r) as the output of the filter speed regulator 5. Obtain the source angle and circumference ωO by the angle and circumference calculation times @6. This calculation The circuit 6 is a divider 6. which divides the set value 1 snow β by the setting 11r1) corrected for the delay in FIG. 2 and the setting 1i i1α. And this removal'a
Result 11β/11Ctvc relation 1/72 Y multiplier f, b coefficient multiplier 62 and 7 to calculate slip angular frequency ω8, rotor angular frequency ω and n force are multiplied by this slip angular frequency ωS to obtain the sum. Get the number of angles around the corner ω0. This divider 61
The slip angular frequency ω8 is calculated by the coefficient unit 62Vc using the following conditions mentioned above in the second term on the right side of Equation 41 and from FIG. 2, λ'2d = 14a·M'! Substitute l' to 1.
Be sure to replace it with p/(1, (1'τ2)).

２ τ２−一 ２ このように干渉分が補正されたα、β相の一次電圧ｅ、
Ｃｔ’＋’Ｂ＋βは相電圧演算回路７において２相−３
相変換がなされ、インバータ２Ｑ）３相電圧役定値、＊
、　、　ｅＺ　、　六　　が取出され、この設定値πよ
るＰＷＭ彼形の一次軍圧市制御によって電動様１には磁
束と二次電流ン非干＃にした速度又Ｖエトルク部］御が
実現されゐ。なに、相ぼ圧屓算回・烙７ｖｃ％ける２相
−３相変喚のためて、市ｗ用周波数（・タ、乞便っだ三
角ｉ列数発生回路８から正弦波ＳＩＮωＱｔ、及び余弦
波ＣＯ８ωｏｔケ得ている。また、インノく一夕２にお
けろｐｗｇ１反形乞得るために、′屯源角周彼叙ω０ン
便つ−た三角波発生回路９から拶ω口に同呵した定数倍
の三角波（！１′取出し、この三角波と設定重圧ｅａｌ
　Ｑｈ！　８．、、　　とのレベル比較によってＰＷＭ
彼形を得ている。2 τ2--2 The primary voltage e of the α and β phases with the interference corrected in this way,
Ct'+'B+β is 2-phase -3 in the phase voltage calculation circuit 7.
Phase conversion is performed, inverter 2Q) 3-phase voltage nominal value, *
. Wow. What, for the 2-phase to 3-phase change that takes 7vc% of the phase pressure calculation cycle, the sine wave SINωQt from the triangular i-series number generation circuit 8, A cosine wave CO8ωot is obtained.Also, in order to obtain the opposite form of pwg1 in Innoku Ichiyo 2, I also received a message from the triangular wave generation circuit 9, which was sent by 'Tunyuan Kakushu's description ω0'. Take out the triangular wave (!1'), which is multiplied by the constant, and combine this triangular wave with the set pressure eal
Qh! 8. , , PWM by comparing the level with
He's getting his shape.

以上のとおり、本発明によれば、電圧形インバータ７１
史って誘導電動４幾ケベクトル市１」御ｆろにおいて、
革ＩＩＪＪ機の励（丑電流設定値１青、と二次電流設定
値ｉテβからインバータの設定重圧”、Ｌｌ、　Ｉ　ｅ
１βン得るのに電動機の一次市流１　、　　によろ相互
１α　１１β 干＃５＋馨キャンセルする補正ケするため、市源角周波
依ω０による干渉汁変動も含めて磁束と二欠電流ベクト
ルヶ常に直交させろ制御がｏＪ能となり、広いＭｌｌ　
（卸範囲に渡って正確なベクトル匍１＠ができる幼果が
ある。しかも、Ｌ、／ｒ、の時定数ン持つ一次遅れ汁も
含めた一次電圧及び角周波数ω０の補正になって高精度
の制御が可能となる。As described above, according to the present invention, the voltage source inverter 71
In the history of induction motors 4 and vector city 1,
Excitation of IIJJ machine (current setting value 1 blue, and secondary current setting value i te β to inverter setting pressure", Ll, I e
To obtain 1β, the motor's primary flow 1 and mutual 1α 11β are canceled.In order to compensate for cancellation, the magnetic flux and the two-way current vector, including the interference fluctuation due to the source angular frequency ω0, should always be orthogonal. Control becomes OJ function, wide Mll
(There is a young fruit that can produce an accurate vector 1@ over the wholesale range. Moreover, it is highly accurate because it corrects the primary voltage and angular frequency ω0 including the first-order lag fluid with a time constant of L, /r. control becomes possible.

【図面の簡単な説明】[Brief explanation of drawings]

第１図シエ２相電圧ｅ　　、ｅ　　に対する誘導電動１
α　　１β 機の等１曲ブロック図、第２図は誘導電動機のベクトル
制御におげろ等価ブロック図、第３図は本発明の一実施
例乞示−ｆ′１ＩｉＩＩ御装置ブロック図である。 １・・・誘導電動機、２・・・電圧形インバータ、３・
・・補正演算回路、４・・・速度検出器、５・・・速度
調節器、６・・・角周波数演算回路、７・・・相電圧演
算回路、８・・・三角関数発生回路、９・・・三角波発
生回路。Figure 1 Induction motor 1 for two-phase voltages e and e
FIG. 2 is an equivalent block diagram for vector control of an induction motor, and FIG. 3 is a block diagram of a control device for an embodiment of the present invention. 1... Induction motor, 2... Voltage type inverter, 3...
... Correction calculation circuit, 4 ... Speed detector, 5 ... Speed regulator, 6 ... Angular frequency calculation circuit, 7 ... Phase voltage calculation circuit, 8 ... Trigonometric function generation circuit, 9 ...Triangular wave generation circuit.

Claims (1)

【特許請求の範囲】 Ｉｌｌ　　誘導電＠機χ亀圧形インバータで駆動し、誘
導電動機の磁束５＋ケ設定するα相電圧θ１αと二次盲
、流汁ン設定才ろβ相゛亀圧ｅ１βから２相３相変換に
よって上記電圧形インバータのａ、ｂ。 Ｃ相３相准圧設定値ｅへ、と、−〇　を得る誘導電動機
のベクトル制御方式において、誘導ｔ　ｒ＋の１′幾の
磁＃、汁？設定するα相−次電流設定１可己αに一次抵
抗ｒ、に設定′ｆろ係ａ器乞通した値から誘尋電＠機の
二次心流汁を設定するβ相−欠電流設定値１１βに等１
曲漏れインダクタンスＬσと一矢砥抗ｒ、の比Ｌσ／ｒ
１の時定ａ？持ちかつｒ。 倍した一次遅れに設定した係数器２通した値六−に電圧
形インバータの角周波数設定器０Ｊｏを乗算しかつ上記
インダクタンスＬσに設定する係数器ケ通した値を畝算
して上記電圧ｅ１αン求め、上記設定値１１β乞−次抵
抗ｒ１に設定する係数＊ 器乞通した値に上記設定値１．ＣｔＫ上記角周波数ｗａ
　Ｙ乗算しかつ一次インダクタンスＬ１に設定する係数
器ン通、した値をカロ算して上記電圧ｅ、βン求めるこ
とン特徴とする誘導電動機のベクトル制御方式。 ＊＊＊ １２）上記設定置１．βから求めた上記値１．βン上記
設定値踏ヶで割算した１直乞＝次インタリタンスＬ２と
二次抵抗ｒ２の比ｒ　２／Ｌ　２に設定する係数器を通
した１１Ｉｉｊに誘導心動機のロータ角周波数検出値ｖ
ｖｒン加算して上記角周波敬設定餉００を求めること乞
特徴とする特許請求の範囲第１項記載の誘導電動機のベ
クトル制御方式。[Claims] Ill The induction motor is driven by a tortoise pressure type inverter, and the magnetic flux 5+ of the induction motor is set from the α phase voltage θ1α and the secondary blind, the flow rate is set from the β phase to the tortoise pressure e1β. a, b of the above voltage source inverter by two-phase three-phase conversion. In the vector control method of an induction motor to obtain the C-phase 3-phase quasi-pressure set value e, and -〇, the induction motor 1' of the induction t r+ has a magnetic #, ? Set the α phase - secondary current setting 1 Set the primary resistance r to the self α, set the filtration coefficient a, and set the secondary heart flow of the dielectric @ machine β phase - missing current setting equal to the value 11β1
Ratio Lσ/r of bending leakage inductance Lσ and single arrow resistance r
1 time constant a? Hold and r. Multiply the angular frequency setter 0Jo of the voltage source inverter by the value 6- passed through the coefficient machine set to the multiplied first-order lag, and calculate the value passed through the coefficient machine set to the above inductance Lσ to calculate the voltage e1α. Calculate the above set value 11β and set the coefficient * to the next resistance r1. Set the above set value 1. CtK above angular frequency wa
A vector control method for an induction motor characterized in that the voltages e and β are obtained by multiplying by Y and passing through a coefficient set to the primary inductance L1, and calculating the resulting values. *** 12) Above settings 1. The above value obtained from β1. The rotor angular frequency detected value v of the induction motor is set to 11Iij through a coefficient unit set to 2/L.
2. The vector control method for an induction motor according to claim 1, wherein the angular frequency setting value 00 is determined by adding the angular frequency.