JPH09223978A - Adaptive interference canceler - Google Patents
Adaptive interference cancelerInfo
- Publication number
- JPH09223978A JPH09223978A JP2700896A JP2700896A JPH09223978A JP H09223978 A JPH09223978 A JP H09223978A JP 2700896 A JP2700896 A JP 2700896A JP 2700896 A JP2700896 A JP 2700896A JP H09223978 A JPH09223978 A JP H09223978A
- Authority
- JP
- Japan
- Prior art keywords
- signal
- interference
- desired signal
- output
- despreading
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
- 230000003044 adaptive effect Effects 0.000 title claims abstract description 20
- 230000002194 synthesizing effect Effects 0.000 claims description 10
- 230000015572 biosynthetic process Effects 0.000 claims description 4
- 238000003786 synthesis reaction Methods 0.000 claims description 4
- 230000005540 biological transmission Effects 0.000 abstract description 10
- 238000001514 detection method Methods 0.000 abstract description 7
- 102100024023 Histone PARylation factor 1 Human genes 0.000 abstract 1
- 101001047783 Homo sapiens Histone PARylation factor 1 Proteins 0.000 abstract 1
- 101000964789 Homo sapiens Zinc finger protein 83 Proteins 0.000 abstract 1
- 230000002411 adverse Effects 0.000 abstract 1
- 238000000034 method Methods 0.000 description 8
- 238000010586 diagram Methods 0.000 description 7
- 230000000875 corresponding effect Effects 0.000 description 5
- 239000011159 matrix material Substances 0.000 description 4
- 239000008186 active pharmaceutical agent Substances 0.000 description 3
- 238000004891 communication Methods 0.000 description 3
- 239000002131 composite material Substances 0.000 description 3
- 238000010295 mobile communication Methods 0.000 description 2
- 238000012986 modification Methods 0.000 description 2
- 230000004048 modification Effects 0.000 description 2
- 230000001276 controlling effect Effects 0.000 description 1
- 230000002596 correlated effect Effects 0.000 description 1
- 230000003111 delayed effect Effects 0.000 description 1
- 238000005562 fading Methods 0.000 description 1
- 230000001360 synchronised effect Effects 0.000 description 1
Abstract
Description
【０００１】[0001]
【発明の属する技術分野】この発明は、通信システムに
用いられる適応干渉キャンセラに関するものであり、特
に、直接拡散符号分割多元接続方式（以下「ＤＳ／ＣＤ
ＭＡ方式」と記す）において、加入者容量の拡大と安定
性の向上に寄与する適応干渉キャンセラに関する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an adaptive interference canceller used in a communication system, and more particularly to a direct sequence code division multiple access system (hereinafter referred to as "DS / CD").
"MA method"), the present invention relates to an adaptive interference canceller that contributes to expansion of subscriber capacity and improvement of stability.
【０００２】[0002]
【従来の技術】移動体通信では、限られた周波数資源で
多数の加入者を収容できるアクセス方式が望まれる。Ｄ
Ｓ／ＣＤＭＡ方式は、広帯域の無線チャンネルを多数の
加入者で共有し、各加入者が互いに直交した拡散符号を
用いて通信を行なう通信方式である。このＤＳ／ＣＤＭ
Ａ方式では、狭帯域の変調波（１次変調）を拡散符号を
用いてさらに変調（２次変調）して出力信号を生成す
る。２次変調波と１次変調波の帯域幅の比は拡散利得と
呼ばれる。変調の際に用いた拡散符号と同一の符号で受
信信号を逆拡散すると、一次変調波が得られるが、他の
加入者からの信号は広帯域のままであるから、１次変調
波の復調の段階では拡散利得分だけ信号対干渉電力比が
向上する。2. Description of the Related Art In mobile communication, an access method capable of accommodating a large number of subscribers with limited frequency resources is desired. D
The S / CDMA system is a communication system in which a wide range of wireless channels are shared by a large number of subscribers and each subscriber communicates using mutually orthogonal spreading codes. This DS / CDM
In method A, a narrow band modulated wave (first-order modulation) is further modulated (secondary modulation) using a spread code to generate an output signal. The bandwidth ratio between the secondary modulated wave and the primary modulated wave is called the spreading gain. When the received signal is despread with the same code as the spreading code used for modulation, a primary modulated wave can be obtained, but the signals from other subscribers remain in a wide band, so demodulation of the primary modulated wave is not possible. At the stage, the signal-to-interference power ratio is improved by the spreading gain.
【０００３】受信信号を逆拡散する過程にあっては、所
望信号の拡散符号と対応したマッチドフィルタによって
受信信号から所望信号成分が抽出される。この場合、理
想的には、所望信号成分に干渉信号成分は含まれない
が、各信号の各拡散符号に相関がある場合や、各拡散符
号の位相が同期していない場合、あるいは多重波伝搬が
生じている場合等では、前記フィルタの他の信号に対す
る直交性が崩れて所望信号成分に干渉信号成分が含ま
れ、伝送特性が劣化してしまう。このため、伝送特性を
改善する技術として、適応干渉キャンセラが知られてい
た。In the process of despreading the received signal, the desired signal component is extracted from the received signal by the matched filter corresponding to the spreading code of the desired signal. In this case, ideally, the desired signal component does not include the interference signal component, but when the spreading codes of the signals are correlated, the phases of the spreading codes are not synchronized, or the multiple wave propagation is performed. In the case where the above occurs, the orthogonality with respect to other signals of the filter is broken, the interference signal component is included in the desired signal component, and the transmission characteristic is deteriorated. Therefore, an adaptive interference canceller has been known as a technique for improving the transmission characteristic.
【０００４】この適応干渉キャンセラの内、干渉に係わ
る他の信号の拡散符号を予め知る必要がないものとし
て、直交化フィルタを用いた適応干渉キャンセラがあ
る。そこでは、直交化フィルタによって受信信号から干
渉信号成分が抽出され、前記所望信号成分から干渉信号
成分を差し引くことにより、干渉信号成分がキャンセル
されていた。Among these adaptive interference cancellers, there is an adaptive interference canceller using an orthogonalization filter that does not need to know the spread code of other signals related to interference in advance. Here, an interference signal component is extracted from the received signal by an orthogonalization filter, and the interference signal component is canceled by subtracting the interference signal component from the desired signal component.
【０００５】[0005]
【発明が解決しようとする課題】ところで、この直交化
フィルタは、予め計算されたタップ係数を持つマッチド
フィルタ群を有し、それらの出力に基づいて干渉信号成
分を生成している。しかし、多重波の存在するフェージ
ング伝搬路において、固定のタップ係数を持つマッチド
フィルタ群は、所望信号との直交性を厳密に維持できな
い。このため、直交化フィルタの出力たる干渉信号成分
には、所望信号成分が内在する。したがって、従来の適
応干渉キャンセラでは、伝送特性の改善が不十分であ
り、また、安定に動作しないといった欠点があった。By the way, this orthogonalization filter has a matched filter group having pre-calculated tap coefficients, and generates an interference signal component based on the outputs thereof. However, in a fading channel where multiple waves exist, a matched filter group having a fixed tap coefficient cannot strictly maintain orthogonality with a desired signal. Therefore, a desired signal component is inherent in the interference signal component that is the output of the orthogonalization filter. Therefore, the conventional adaptive interference canceller has drawbacks such that the transmission characteristics are not sufficiently improved and the adaptive interference canceller does not operate stably.
【０００６】本発明は上述した事情に鑑がみてなされた
ものであり、その目的とするところは、伝送特性を改善
でき、かつ安定性に優れた適応干渉キャンセラを提供す
ることにある。The present invention has been made in view of the above-mentioned circumstances, and an object of the present invention is to provide an adaptive interference canceller capable of improving transmission characteristics and excellent in stability.
【０００７】[0007]
【課題を解決するための手段】上記課題を解決するため
請求項１に記載の発明にあっては、入力信号から干渉信
号成分を除去するキャンセラにおいて、所望信号の拡散
符号に基づいて、入力信号を逆拡散する所望信号逆拡散
手段と、前記所望信号の拡散符号と直交した各拡散符号
に基づいて、入力信号を各々逆拡散する干渉信号逆拡散
手段と、前記干渉信号逆拡散手段の各出力から所望信号
成分を除去する所望信号除去手段と、前記所望信号除去
手段の各出力に基づいて、合成係数を生成する合成係数
生成手段と、前記合成係数に基づいて、前記干渉信号逆
拡散手段の各出力を合成する干渉信号合成手段と、前記
所望信号逆拡散手段の出力と前記干渉信号合成手段の出
力の差分を算出する干渉信号キャンセル手段とを備えた
ことを特徴とする。In order to solve the above problems, according to the invention of claim 1, in a canceller for removing an interference signal component from an input signal, the input signal is based on a spread code of a desired signal. A desired signal despreading means for despreading, and an interference signal despreading means for despreading an input signal based on each spreading code orthogonal to the spreading code of the desired signal, and each output of the interference signal despreading means A desired signal removing means for removing a desired signal component from the output signal, a composite coefficient generating means for generating a composite coefficient based on each output of the desired signal removing means, and an interference signal despreading means based on the composite coefficient. An interference signal combining means for combining the respective outputs, and an interference signal canceling means for calculating a difference between the output of the desired signal despreading means and the output of the interference signal combining means are provided.
【０００８】また、請求項２に記載の発明にあっては、
前記合成係数生成手段は、前記干渉信号キャンセル手段
の出力電力が最小となるように、前記合成係数を適応的
に生成することを特徴とする。Further, in the invention described in claim 2,
The synthesis coefficient generation means adaptively generates the synthesis coefficient so that the output power of the interference signal cancellation means is minimized.
【０００９】また、請求項３に記載の発明にあっては、
前記所望信号除去手段は、前記所望信号逆拡散手段の出
力を検波して所望信号を生成する所望信号検波手段と、
前記所望信号に基づいて、その共役複素所望信号を生成
する共役複素生成手段と、前記干渉信号逆拡散手段の各
出力と共役複素所望信号を各々乗算する第１の乗算手段
と、前記第１の乗算手段の各出力から低域周波数成分を
除去する各高域通過フィルタと、前記各高域通過フィル
タの出力と前記所望信号を乗算する第２の乗算手段とを
備えたことを特徴とする。Further, in the invention according to claim 3,
The desired signal removing means detects the output of the desired signal despreading means to generate a desired signal, and a desired signal detecting means,
Conjugate complex generation means for generating the conjugate complex desired signal based on the desired signal, first multiplication means for multiplying each output of the interference signal despreading means by the conjugate complex desired signal, and the first The present invention is characterized by including high-pass filters for removing low-frequency components from the outputs of the multiplying means, and second multiplying means for multiplying the output of the high-pass filters by the desired signal.
【００１０】[0010]
【発明の実施の形態】１．実施形態の構成 以下、図面を参照してこの発明の実施形態の構成につい
て説明する。図１はこの発明の一実施形態に係わる適応
干渉キャンセラのブロック図である。図１において、Ｉ
Ｎは入力端子であり、ここに受信信号ｒが供給される。
また、ＯＵＴは出力端子であり、そこから出力信号が出
力される。ＭＦ０は所望信号逆拡散手段であり、所望信
号の拡散符号に対応したマッチドフィルタで構成され、
受信信号ｒに逆拡散処理を施して出力信号ｘ0(i)を生成
する。なお、「i」はある時刻を特定するための変数で
ある。所望信号逆拡散手段ＭＦＯは、例えば、図２に示
すトランスバーサルフィルタによって構成される。図２
において、受信信号ｒは、そのチップ周期ＴCと遅延時
間が一致する遅延回路１０1〜１０mによって順次遅延さ
れる。そして、各タップの出力が計数器１１0〜１１mを
介して加算器１２に供給される。この例では、計数器１
１0〜１１mの係数を、所望信号の拡散符号に対応して設
定する。このため、加算器１２の出力は所望信号成分と
なり、これが所望信号逆拡散手段ＭＦ０の出力信号ｘ
0(i)となる。BEST MODE FOR CARRYING OUT THE INVENTION 1. Configuration of Embodiment Hereinafter, a configuration of an embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram of an adaptive interference canceller according to an embodiment of the present invention. In FIG. 1, I
N is an input terminal to which the received signal r is supplied.
OUT is an output terminal from which an output signal is output. MF0 is a desired signal despreading means, and is composed of a matched filter corresponding to the spreading code of the desired signal,
The received signal r is subjected to despreading processing to generate an output signal x 0 (i). In addition, "i" is a variable for specifying a certain time. The desired signal despreading means MFO is composed of, for example, a transversal filter shown in FIG. FIG.
In, the received signal r is sequentially delayed by the delay circuits 10 1 to 10 m whose delay time matches the chip period T C. Then, the output of each tap is supplied to the adder 12 via the counters 11 0 to 11 m . In this example, the counter 1
Coefficients of 10 to 11 m are set corresponding to the spread code of the desired signal. Therefore, the output of the adder 12 becomes a desired signal component, which is the output signal x of the desired signal despreading means MF0.
It becomes 0 (i).
【００１１】また、図１に示すＭＦＧは干渉信号逆拡散
手段であり、マッチドフィルタ群ＭＦ１〜ＭＦｎから構
成される。各マッチドフィルタＭＦ１〜ＭＦｎは、所望
信号逆拡散手段ＭＦ０と同様にトランスバーサルフィル
タによって構成され、その各タップ係数は、所望信号の
拡散符号と直交する拡散符号に対応する。このため、伝
送路において直交性が維持される場合には、マッチドフ
ィルタＭＦ１〜ＭＦｎの出力信号ｘ1(i)〜ｘn(i)は干渉
信号成分に対応した信号となる。The MFG shown in FIG. 1 is an interference signal despreading means, which is composed of matched filter groups MF1 to MFn. Each of the matched filters MF1 to MFn is composed of a transversal filter like the desired signal despreading means MF0, and each tap coefficient thereof corresponds to a spreading code orthogonal to the spreading code of the desired signal. Therefore, when the orthogonality is maintained in the transmission path, the output signals x 1 (i) to x n (i) of the matched filters MF1 to MFn are signals corresponding to the interference signal components.
【００１２】また、ＣＢＮは干渉信号合成手段であり、
各マッチドフィルタＭＦ１〜ＭＦｎの出力信号ｘ1(i)〜
ｘn(i)を合成してキャンセル用干渉信号成分を生成す
る。図３は干渉信号合成手段ＣＢＮのブロック図であ
る。図３において、出力信号ｘ1(i)〜ｘn(i)が係数器２
０1〜２０nを介して加算器２１に供給されると、それら
が加算され、干渉信号成分が生成される。なお、係数器
２０1〜２０nの係数は、後述する係数ベクトルＷ(i)に
よって制御される。このため、係数ベクトルＷ(i)を適
切に設定することによって、干渉信号合成手段ＣＢＮは
キャンセル用干渉信号成分を生成できる。CBN is an interference signal synthesizing means,
Output signal x 1 (i) of each matched filter MF1 to MFn
x n (i) is combined to generate a canceling interference signal component. FIG. 3 is a block diagram of the interference signal combining means CBN. In FIG. 3, the output signals x 1 (i) to x n (i) are the coefficient units 2
When supplied to the adder 21 via 0 1 to 20 n , they are added to generate an interference signal component. The coefficients of the coefficient units 20 1 to 20 n are controlled by a coefficient vector W (i) described later. Therefore, by appropriately setting the coefficient vector W (i), the interference signal combining means CBN can generate the canceling interference signal component.
【００１３】また、ＤＳＣは所望信号除去手段であっ
て、ここで、出力信号ｘ1(i)〜ｘn(i)から所望信号成分
が除去され、出力信号ｘ1'(i)〜ｘn'(i)が生成される。
図４は所望信号除去手段ＤＳＣのブロック図である。同
図に示すように所望信号除去手段ＤＳＣは、所望信号検
波手段ＤＳＣ−ＤＥＴ、複素共役生成手段ＣＣ、乗算器
３０1〜３０n，３１1〜３１n、およびハイパスフィルタ
ＨＰＦ1〜ＨＰＦnから構成される。The DSC is a desired signal removing means, in which the desired signal component is removed from the output signals x 1 (i) to x n (i), and the output signals x 1 ′ (i) to x n. '(i) is generated.
FIG. 4 is a block diagram of the desired signal removing means DSC. As shown in the figure, the desired signal removing means DSC includes the desired signal detecting means DSC-DET, the complex conjugate generating means CC, the multipliers 30 1 to 30 n , 31 1 to 31 n , and the high pass filters HPF 1 to HPF n. Composed.
【００１４】所望信号検波手段ＤＳＣ−ＤＥＴは所望信
号を判定する検波回路で構成される。複素共役生成手段
ＣＣは所望信号に複素共役処理を施し、複素共役所望信
号を生成する。なお、複素共役処理自体は周知であるか
ら、その具体的な構成については説明を省略する。ま
た、乗算器３０1〜３０nは、出力信号ｘ1〜ｘnを逆変調
し、そこに含まれている所望信号成分を低域周波数に変
換する手段として機能する。ハイパスフィルタＨＰＦ1
〜ＨＰＦnは、低域周波数に変換された所望信号成分を
十分除去できるカットオフ周波数を有する。乗算器３１
1〜３１nは、ハイパスフィルタＨＰＦ1〜ＨＰＦnの各出
力信号を所望信号で再度変調するが、ハイパスフィルタ
ＨＰＦ1〜ＨＰＦnによって所望信号成分は除去されるか
ら、それらの出力信号ｘ1'(i)〜ｘn'(i)は干渉信号成分
のみを有する信号となる。The desired signal detection means DSC-DET is composed of a detection circuit for determining a desired signal. The complex conjugate generating means CC performs a complex conjugate process on the desired signal to generate a complex conjugate desired signal. Since the complex conjugate process itself is well known, the description of its specific configuration will be omitted. Further, the multipliers 30 1 to 30 n function as means for inversely modulating the output signals x 1 to x n and converting the desired signal components contained therein into low frequency bands. High pass filter HPF 1
~ HPF n has a cutoff frequency that can sufficiently remove the desired signal component converted to the low frequency range. Multiplier 31
1 to 31 n remodulate the output signals of the high-pass filters HPF 1 to HPF n with the desired signals, but the desired signal components are removed by the high-pass filters HPF 1 to HPF n , so that the output signals x 1 ' (i) to x n '(i) are signals having only interference signal components.
【００１５】次に、ＣＡＮは減算器によって構成される
干渉信号キャンセル手段であり、所望信号逆拡散手段Ｍ
Ｆ０の出力信号から干渉信号合成手段ＣＢＮの出力信号
を減算する。ところで、所望信号成分と干渉信号成分
（雑音）は、互いに独立であるから、所望信号逆拡散手
段ＭＦ０の出力電力は、これらの和になる。したがっ
て、干渉信号合成手段ＣＢＮの出力信号が干渉信号成分
のみであれば、干渉信号キャンセル手段ＣＡＮによっ
て、干渉信号成分が完全にキャンセルされる。Next, CAN is an interference signal canceling means composed of a subtracter, and a desired signal despreading means M.
The output signal of the interference signal synthesizing means CBN is subtracted from the output signal of F0. By the way, since the desired signal component and the interference signal component (noise) are independent of each other, the output power of the desired signal despreading means MF0 is the sum of them. Therefore, if the output signal of the interference signal synthesizing unit CBN is only the interference signal component, the interference signal canceling unit CAN completely cancels the interference signal component.
【００１６】また、ＡＡは係数ベクトル生成手段であっ
て、出力信号ｘ1'(i)〜ｘn'(i)に基づいて干渉信号合成
手段ＣＢＮに用いられる係数ベクトルＷ(i)を生成す
る。ここで、所望信号逆拡散手段ＭＦ０の出力電力は所
望信号成分と干渉信号成分の和となるため、所望信号逆
拡散手段ＭＦ０の出力信号から干渉信号成分がキャンセ
ルされるように係数ベクトルＷを制御すると、干渉信号
キャンセル手段ＣＡＮの出力電力は最小となる。一方、
干渉信号キャンセル手段ＣＡＮの出力電力が最小となる
ように係数ベクトルＷ(i)を制御すれば、干渉信号成分
をキャンセルすることができる。このため、係数ベクト
ル生成手段ＡＡは、適応アルゴリズムを用いて、干渉信
号キャンセル手段ＣＡＮの出力電力が最小となるように
係数ベクトルＷ(i)を生成する。最小化アルゴリズムに
は、例えば、ＬＭＳアルゴリズムやＲＬＳアルゴリズム
を用いる。また、ＤＥＴは検波手段であり、これにより
干渉信号キャンセル手段ＣＡＮの出力信号が検波され所
望信号が生成される。AA is a coefficient vector generating means for generating a coefficient vector W (i) used in the interference signal synthesizing means CBN based on the output signals x 1 '(i) to x n ' (i). . Here, since the output power of the desired signal despreading means MF0 is the sum of the desired signal component and the interference signal component, the coefficient vector W is controlled so that the interference signal component is canceled from the output signal of the desired signal despreading means MF0. Then, the output power of the interference signal canceling means CAN becomes the minimum. on the other hand,
The interference signal component can be canceled by controlling the coefficient vector W (i) so that the output power of the interference signal canceling means CAN is minimized. Therefore, the coefficient vector generation means AA uses the adaptive algorithm to generate the coefficient vector W (i) so that the output power of the interference signal cancellation means CAN becomes the minimum. As the minimization algorithm, for example, the LMS algorithm or the RLS algorithm is used. Further, DET is a detection means, which detects the output signal of the interference signal cancellation means CAN to generate a desired signal.
【００１７】２．実施形態の動作 以下、図面を参照してこの発明の実施形態の全体動作に
ついて説明する。この例では、受信信号ｒにおいて、シ
ンボル周期をＴ，チップ周期をＴCとし、時刻ｔ＝ｉＴ
＋ｊＴCにおける受信信号ｒのサンプル値をｒ(i,j)と表
わすこととする。この場合、受信信号ｒ(i,j)の実数部
分は、受信信号の同相成分の振幅、虚数部分は受信信号
の直交成分の振幅を各々指示する。[0017] 2. Operation of the Embodiment Hereinafter, the overall operation of the embodiment of the present invention will be described with reference to the drawings. In this example, in the received signal r, the symbol period is T, the chip period is T C, and time t = iT
The sample value of the received signal r at + jT C will be represented as r (i, j). In this case, the real part of the received signal r (i, j) indicates the amplitude of the in-phase component of the received signal, and the imaginary part indicates the amplitude of the quadrature component of the received signal.
【００１８】図１において、まず、受信信号ｒ(i,j)は
所望信号逆拡散手段ＭＦ０に供給される。この所望信号
逆拡散手段ＭＦ０を構成する各係数器１１0〜１１m（図
２参照）の各タップ係数をｖ0(j)とおくと、所望信号逆
拡散手段ＭＦ０の出力信号ｘ0(i)は次式で与えられる。 ｘ0(i)＝Σjｒ(i,j)ｖ0(j) また、同様に、干渉信号逆拡散手段ＭＦＧを構成する第
ｋマッチドフィルタＭＦｋにおいて、その各タップ係数
をｖk(j)で表わすものとする。タップ係数ｖk(j)には、
上述したように所望信号の拡散符号と直交した符号が割
り当てられるから、干渉信号逆拡散手段ＭＦＧの出力信
号ｘ1(i)〜ｘn(i)は、干渉信号成分に対応したものとな
る。なお、干渉信号逆拡散手段ＭＦＧの出力信号をベク
トルＸT(i)で表わすと、この出力信号ベクトルＸT(i)は
次式で与えられる。但し、「T」は転置行列を指示する
記号である。 ＸT(i)＝［ｘ1(i)，ｘ2(i)，…，ｘn(i)］In FIG. 1, the received signal r (i, j) is first supplied to the desired signal despreading means MF0. Letting v 0 (j) be each tap coefficient of each coefficient unit 11 0 to 11 m (see FIG. 2) constituting the desired signal despreading means MF0, the output signal x 0 (i of the desired signal despreading means MF0 ) Is given by the following equation. x 0 (i) = Σ j r (i, j) v 0 (j) Similarly, in the k-th matched filter MFk forming the interference signal despreading means MFG, each tap coefficient is v k (j). Shall be represented by. For the tap coefficient v k (j),
Since the code orthogonal to the spreading code of the desired signal is assigned as described above, the output signals x 1 (i) to x n (i) of the interference signal despreading means MFG correspond to the interference signal components. When the output signal of the interference signal despreading means MFG is represented by a vector X T (i), this output signal vector X T (i) is given by the following equation. However, " T " is a symbol indicating a transposed matrix. X T (i) = [x 1 (i), x 2 (i), ..., X n (i)]
【００１９】ここで、干渉信号合成手段ＣＢＮを構成す
る係数器２０1〜２０nの各タップ係数をｗ1(i)〜ｗn(i)
で表わし、これをベクトルＷT(i)で表わすと、係数ベク
トルＷT(i)は次式で与えられる。 ＷT(i)＝［ｗ1(i)，ｗ2(i)，…，ｗn(i)］ この場合、干渉信号合成手段ＣＢＮの出力信号、すなわ
ち、干渉信号成分はＷH(i)ＸT(i)となる。この出力信号
が干渉信号キャンセル手段ＣＡＮに供給されると、干渉
信号合成手段ＣＢＮにおいて、ｘ0(i)とＷH(i)ＸT(i)の
差分が算出され、ｘ0(i)−ＷH(i)ＸT(i)が検波手段ＤＥ
Ｔに入力される。一方、干渉信号除去手段ＤＳＣは、干
渉信号逆拡散手段ＭＦＧの出力信号ベクトルＸT(i)から
所望信号を除去する操作を行なう。Here, the tap coefficients of the coefficient units 20 1 to 20 n forming the interference signal synthesizing means CBN are represented by w 1 (i) to w n (i).
When this is represented by the vector W T (i), the coefficient vector W T (i) is given by the following equation. W T (i) = [w 1 (i), w 2 (i), ..., W n (i)] In this case, the output signal of the interference signal combining means CBN, that is, the interference signal component is WH (i). It becomes X T (i). When this output signal is supplied to the interference signal canceling means CAN, the interference signal synthesizing means CBN calculates the difference between x 0 (i) and WH (i) X T (i), and x 0 (i) − W H (i) X T (i) is the detection means DE
Input to T. On the other hand, the interference signal removing means DSC performs an operation of removing a desired signal from the output signal vector X T (i) of the interference signal despreading means MFG.
【００２０】この動作について図４を参照しつつ説明す
る。いま、加入者０からｎまでに対応した各信号を複素
送信シンボルで表わした場合、それらのベクトルＳTを
次式で与えるものとする。 ＳT＝［ｓ0(i)，ｓ1(i)，…，ｓn(i)］ なお、この例にあっては、ｓ0(i)が所望信号の複素送信
シンボルに対応するものである。このベクトルＳとｎ行
×ｎ＋１列の行列Ｒを用いると、干渉信号逆拡散手段Ｍ
ＦＧの出力信号ベクトルＸ(i)は、Ｘ(i)＝ＲＳと表わす
ことができる。この行列Ｒは干渉信号逆拡散手段ＭＦＧ
のタップ係数と各信号の拡散符号および各拡散符号の位
相差から定まる。特に各拡散符号に位相差がない場合、
行列Ｒの各要素（Ｒ）klは次式で与えられる。但し、ｕ
l(j)は、各信号の拡散符号に対応する。 （Ｒ）kl＝Σjｖk(j)ｕl(j)。 こうして生成された出力信号ベクトルＸ(i)が乗算器３
０1〜３０nに供給される。This operation will be described with reference to FIG. Now, when each signal corresponding to the subscribers 0 to n is represented by a complex transmission symbol, their vector S T is given by the following equation. S T = [s 0 (i), s 1 (i), ..., s n (i)] In this example, s 0 (i) corresponds to the complex transmission symbol of the desired signal. is there. Using this vector S and the matrix R of n rows × n + 1 columns, the interference signal despreading means M
The output signal vector X (i) of the FG can be expressed as X (i) = RS. This matrix R is the interference signal despreading means MFG.
It is determined from the tap coefficient of, the spreading code of each signal, and the phase difference of each spreading code. Especially when there is no phase difference in each spreading code,
Each element (R) kl of the matrix R is given by the following equation. Where u
l (j) corresponds to the spread code of each signal. (R) kl = Σ j v k (j) u l (j). The output signal vector X (i) thus generated is multiplied by the multiplier 3
0 1 to 30 n .
【００２１】一方、干渉信号キャンセル手段ＣＡＮの出
力信号が、所望信号検波手段ＤＳＣ−ＤＥＴに供給され
ると、そこで判定が行なわれ、所望信号が生成される。
さて、この所望信号の複素送信シンボル判定値をｑ0(i)
で表わし、複素共役を記号＊で表わすものとすれば、複
素共役生成手段ＣＣの出力信号はｑ0 *(i)となる。これ
が、各乗算器３０1〜３０nに供給され、出力信号ベクト
ルＸ(i)と乗積される。この乗積結果は次式で与えられ
る。 ｑ0 *(i)Ｘ(i)＝ｑ0 *(i)ＲＳ＝ＲＳ' 但し、Ｓ'T＝［ｓ0(i)ｑ0 *(i)，ｓ1(i)ｑ0 *(i)…，ｓ
n(i)ｑ0 *(i)］である。On the other hand, when the output signal of the interference signal canceling means CAN is supplied to the desired signal detecting means DSC-DET, a judgment is made there and a desired signal is generated.
Now, let the complex transmission symbol decision value of this desired signal be q 0 (i)
And the complex conjugate is represented by the symbol *, the output signal of the complex conjugate generating means CC is q 0 * (i). This is supplied to each of the multipliers 30 1 to 30 n and multiplied with the output signal vector X (i). This product result is given by the following equation. q 0 * (i) X ( i) = q 0 * (i) RS = RS ' However, S' T = [s 0 (i) q 0 * (i), s 1 (i) q 0 * (i ) ..., s
a n (i) q 0 * ( i)].
【００２２】この操作によって、出力信号ベクトルＸ
(i)に逆変調が施されると、 所望信号成分ｓ0(i)ｑ
0 *(i)はシンボルレートに比べて低い周波数に電力が集
中する。ここで、所望信号が正しく判定されており、ま
た、送信信号が単位円上にあるとすれば、 Ｓ'T＝［１，ｓ1(i)ｑ0 *(i)，…，ｓn(i)ｑ0 *(i)］ となる。この場合、ｓ1(i)ｑ0 *(i)，…，ｓn(i)ｑ0 *(i)
の各成分は、送信シンボル系列が異なるのでランダムプ
ロセスとなり、一定値とはならない。こうして生成され
た乗算器３０1〜３０nの各出力信号ＲＳ'が、ハイパス
フィルタＨＦＰ1〜ＨＦＰnに供給されると、そこで低域
周波数に逆変調され一定値となった所望信号成分が除去
される。このため、ハイパスフィルタＨＦＰ1〜ＨＦＰn
の出力信号をＲＳ''で表わすと、Ｓ''T＝［０，ｓ1(i)
ｑ0 *(i)，…，ｓn(i)ｑ0 *(i)］となる。By this operation, the output signal vector X
When inverse modulation is applied to (i), desired signal component s 0 (i) q
In 0 * (i), power is concentrated at a frequency lower than the symbol rate. Here, if the desired signal is correctly determined and the transmitted signal is on the unit circle, then S ′ T = [1, s 1 (i) q 0 * (i), ..., S n ( i) q 0 * (i)]. In this case, s 1 (i) q 0 * (i), ..., s n (i) q 0 * (i)
Since the transmission symbol sequences are different, each component of is a random process and does not have a constant value. When the output signals RS ′ of the multipliers 30 1 to 30 n generated in this way are supplied to the high-pass filters HFP 1 to HFP n , the desired signal components that are inversely modulated to a low frequency and have a constant value are removed there. To be done. Therefore, the high pass filters HFP 1 to HFP n
If the output signal of R is represented by RS '', S '' T = [0, s 1 (i)
q 0 * (i), ..., s n (i) q 0 * (i)].
【００２３】次に、乗算器３１1〜３１nにおいて、出力
信号ＲＳ''と所望信号ｑ0(i)の乗積が行なわれ、これに
より出力信号ＲＳ''に再度変調が施される。乗算器３１
1〜３１nの出力信号ベクトルＸ'(i)をＲＳ'''で表わす
ならば、Ｘ'(i)＝ＲＳ'''＝ｑ0(i)ＲＳ''となる。ここ
で、Ｓ'''Tは次式で与えられる。 Ｓ'''T＝［０，ｓ1(i)ｑ0 *(i)ｑ0(i)，…，ｓn(i)ｑ0 *(i)ｑ0(i)］ ＝［０，ｓ1(i)，…，ｓn(i)］ ところで、上述したように干渉信号逆拡散手段ＭＦＧの
出力信号ベクトルＸ(i)は、Ｘ(i)＝ＲＳであり、また、
ＳT＝［ｓ0(i)，ｓ1(i)，…，ｓn(i)］であった。ここ
で、Ｓ'''TとＳTを比較すると、Ｓ'''Tではｓ0(i)が０
となっていることが判る。このことは、出力信号ベクト
ルＸ'(i)が、出力信号ベクトルＸ(i)から所望信号成分
を除去したものとなっていることを意味する。Next, in the multipliers 31 1 to 31 n , the product of the output signal RS ″ and the desired signal q 0 (i) is performed, whereby the output signal RS ″ is re-modulated. Multiplier 31
If the output signal vector X ′ (i) of 1 to 31 n is represented by RS ″ ′, X ′ (i) = RS ′ ″ = q 0 (i) RS ″. Here, S ″ ′ T is given by the following equation. S ″ ′ T = [0, s 1 (i) q 0 * (i) q 0 (i), ..., s n (i) q 0 * (i) q 0 (i)] = [0, s 1 (i), ..., s n (i)] By the way, as described above, the output signal vector X (i) of the interference signal despreading means MFG is X (i) = RS, and
S T = [s 0 (i), s 1 (i), ..., S n (i)]. Here, 'A comparison of T and S T, S' S '' s the '' T 0 (i) is 0
It turns out that This means that the output signal vector X '(i) is obtained by removing the desired signal component from the output signal vector X (i).
【００２４】この後、図１に示す係数ベクトル生成手段
ＡＡは、出力信号ベクトルＸ'(i)に基づいて、干渉信号
キャンセル手段ＣＡＮの出力電力が最小となるように係
数ベクトルＷ(i)を求める。以下、適応アルゴリズムに
ＬＭＳアルゴリズムを用いた場合を説明する。干渉信号
キャンセル手段ＣＡＮの出力信号は、上述したようにｙ
(i)＝ｘ0(i)−ＷH(i)ＸT(i)となる。ＬＭＳアルゴリズ
ムのステップサイズをμとおくと、次式で与えられる更
新式によって、係数ベクトルＷ(i)が算出される。 Ｗ(i+1)=Ｗ(i)+μＸ'(i)ｙ＊（ｉ）After that, the coefficient vector generating means AA shown in FIG. 1 determines the coefficient vector W (i) based on the output signal vector X '(i) so that the output power of the interference signal canceling means CAN becomes the minimum. Ask. The case where the LMS algorithm is used as the adaptive algorithm will be described below. The output signal of the interference signal canceling means CAN is y as described above.
(i) = x 0 and becomes (i) -W H (i) X T (i). Assuming that the step size of the LMS algorithm is μ, the coefficient vector W (i) is calculated by the update equation given by the following equation. W (i + 1) = W (i) + μX '(i) y * (i)
【００２５】以上説明したように本実施形態にあって
は、所望信号除去手段ＤＳＣによって干渉信号逆拡散手
段ＭＦＧの各出力信号から所望信号成分を除去し、それ
らに基づいて係数ベクトルＷ（ｉ）を生成したので、干
渉信号合成手段ＣＢＮの出力信号には所望信号成分が含
まれない。このため、干渉信号キャンセル手段ＣＡＮに
よって、干渉信号成分を確実にキャンセルすることがで
きる。また、干渉信号キャンセル手段ＣＡＮによって所
望信号成分がキャンセルされることがないので、干渉信
号キャンセル手段ＣＡＮの出力電力を最小化するように
適応制御を行なっても、干渉信号キャンセル手段ＣＡＮ
の出力電力を一定値に維持することができ、制御システ
ムの安定性を向上することができる。As described above, in the present embodiment, the desired signal removing means DSC removes the desired signal component from each output signal of the interference signal despreading means MFG, and based on them, the coefficient vector W (i) is obtained. , The desired signal component is not included in the output signal of the interference signal synthesizing means CBN. Therefore, the interference signal canceling means CAN can reliably cancel the interference signal component. Further, since the desired signal component is not canceled by the interference signal canceling means CAN, even if adaptive control is performed so as to minimize the output power of the interference signal canceling means CAN, the interference signal canceling means CAN.
Output power can be maintained at a constant value, and the stability of the control system can be improved.
【００２６】３．変形例 本発明は上述した実施形態に限定されるものでなく、例
えば以下のように種々の変形が可能である。 上記実施形態において、所望信号検波手段ＤＳＣ−Ｄ
ＥＴ（図４参照）の替わりに検波手段ＤＥＴを用いても
良い。この場合には検波手段ＤＥＴの出力信号を複素共
役生成手段ＣＣと乗算器３１1〜３１nに直接供給すれば
良い。[0026] 3. Modifications The present invention is not limited to the above-described embodiment, and various modifications are possible, for example, as follows. In the above embodiment, the desired signal detection means DSC-D
The detection means DET may be used instead of ET (see FIG. 4). In this case, the output signal of the detecting means DET may be directly supplied to the complex conjugate generating means CC and the multipliers 31 1 to 31 n .
【００２７】上記実施形態は、ＤＳ／ＣＤＭＡ方式に
用いられる適応干渉キャンセラを一例として説明した
が、本発明はこれに限定されるものではなく、拡散符号
を用いた他の通信方式における適応干渉キャンセラに適
用しても良いことは勿論である。In the above embodiment, the adaptive interference canceller used in the DS / CDMA system is described as an example, but the present invention is not limited to this, and the adaptive interference canceller in another communication system using the spread code. Of course, it may be applied to.
【００２８】[0028]
【発明の効果】以上説明したように、本発明に係わる発
明特定事項によれば、干渉信号逆拡散手段の出力信号に
含まれる干渉信号成分を確実に除去できるため、所望信
号成分のＳ／Ｎを改善することができ、また、動作の安
定性を向上することができる。As described above, according to the features of the invention relating to the present invention, the interference signal component contained in the output signal of the interference signal despreading means can be reliably removed, so that the S / N ratio of the desired signal component is reduced. Can be improved, and the stability of operation can be improved.
【図１】 本発明の一実施形態に係わる適応干渉キャン
セラのブロック図である。FIG. 1 is a block diagram of an adaptive interference canceller according to an embodiment of the present invention.
【図２】 同実施形態に用いられる所望信号拡散手段の
ブロック図である。FIG. 2 is a block diagram of a desired signal spreading unit used in the same embodiment.
【図３】 同実施形態に用いられる干渉信号合成手段の
ブロック図である。FIG. 3 is a block diagram of an interference signal synthesizing unit used in the same embodiment.
【図４】 同実施形態に用いられる所望信号除去手段の
ブロック図である。FIG. 4 is a block diagram of a desired signal removing unit used in the same embodiment.
ｒ 入力信号 ＭＦ０ 所望信号逆拡散手段 ＭＦＧ 干渉信号逆拡散手段 ＤＳＣ 所望信号除去手段 ＡＡ 合成係数生成手段 ＣＢＮ 干渉信号合成手段 ＣＡＮ 干渉信号キャンセル手段 ＤＳＣ−ＤＥＴ 所望信号検波手段 ＣＣ 共役複素生成手段 ３０1〜３０n 乗算器（第１の乗算手段） ３１1〜３１n 乗算器（第２の乗算手段） ＨＰＦ1〜ＨＰＦn ハイパスフィルタ（高域通過フィル
タ）r input signal MF0 desired signal despreading means MFG interference signal despreading means DSC desired signal removing means AA combining coefficient generating means CBN interference signal combining means CAN interference signal canceling means DSC-DET desired signal detecting means CC conjugate complex generating means 30 1 to 30 n multiplier (first multiplication means) 31 1 to 31 n multiplier (second multiplication means) HPF 1 to HPF n high-pass filter (high-pass filter)
フロントページの続き (71)出願人 596020233 榎本 和宏 京都府京都市右京区太秦上刑部町16−５ (72)発明者 鈴木 博 東京都港区虎ノ門二丁目10番１号 エヌ・ ティ・ティ移動通信網株式会社内 (72)発明者 吉田 進 京都府宇治市木幡北畠10−７ (72)発明者 村田 英一 大阪府枚方市禁野本町２丁目11 枚方合同 宿舎1022 (72)発明者 榎本 和宏 京都府京都市右京区太秦上刑部町16−５Front Page Continuation (71) Applicant 596020233 Kazuhiro Enomoto 16-5 Uzumasa Shangyobucho, Ukyo-ku, Kyoto City, Kyoto Prefecture (72) Inventor Hiroshi Suzuki 2-10-1 Toranomon, Minato-ku, Tokyo, NTT Mobile Communications Ami Co., Ltd. (72) Inventor Susumu Yoshida 10-7 Kohata Kitahata, Uji City, Kyoto Prefecture (72) Inventor Eiichi Murata 2-chome, 11-chome Hinokatacho, Hirakata City, Osaka Prefecture 1022 (72) Kazuhiro Enomoto Kyoto 16-5 Uzumasa Joubecho, Ukyo-ku, Kyoto Prefecture
Claims (3)
ャンセラにおいて、 所望信号の拡散符号に基づいて、入力信号を逆拡散する
所望信号逆拡散手段と、 前記所望信号の拡散符号と直交した各拡散符号に基づい
て、入力信号を各々逆拡散する干渉信号逆拡散手段と、 前記干渉信号逆拡散手段の各出力から所望信号成分を除
去する所望信号除去手段と、 前記所望信号除去手段の各出力に基づいて、合成係数を
生成する合成係数生成手段と、 前記合成係数に基づいて、前記干渉信号逆拡散手段の各
出力を合成する干渉信号合成手段と、 前記所望信号逆拡散手段の出力と前記干渉信号合成手段
の出力の差分を算出する干渉信号キャンセル手段とを備
えたことを特徴とする適応干渉キャンセラ。1. A canceller for removing an interference signal component from an input signal, a desired signal despreading means for despreading an input signal based on a spreading code of the desired signal, and each spreading orthogonal to the spreading code of the desired signal. Based on the code, the interference signal despreading means for despreading the input signal, a desired signal removing means for removing a desired signal component from each output of the interference signal despreading means, and each output of the desired signal removing means Based on the synthetic coefficient, an interference signal synthesizing means for synthesizing the respective outputs of the interference signal despreading means, and an output of the desired signal despreading means and the interference. An adaptive interference canceller, comprising: interference signal canceling means for calculating a difference between outputs of the signal combining means.
キャンセル手段の出力電力が最小となるように、前記合
成係数を適応的に生成することを特徴とする請求項１に
記載の適応干渉キャンセラ。2. The adaptive interference canceller according to claim 1, wherein the synthesis coefficient generation means adaptively generates the synthesis coefficient so that the output power of the interference signal cancellation means is minimized. .
成する所望信号検波手段と、 前記所望信号に基づいて、その共役複素所望信号を生成
する共役複素生成手段と、 前記干渉信号逆拡散手段の各出力と共役複素所望信号を
各々乗算する第１の乗算手段と、 前記第１の乗算手段の各出力から低域周波数成分を除去
する各高域通過フィルタと、 前記各高域通過フィルタの出力と前記所望信号を乗算す
る第２の乗算手段とを備えたことを特徴とする請求項１
または２に記載の適応干渉キャンセラ。3. The desired signal removing means detects the output of the desired signal despreading means to generate a desired signal, and the conjugate complex desired signal based on the desired signal. Conjugate complex generation means, first multiplication means for multiplying each output of the interference signal despreading means by each conjugate complex desired signal, and each high frequency component for removing low frequency components from each output of the first multiplication means. 2. A high-pass filter, and second multiplication means for multiplying the output of each high-pass filter by the desired signal.
Or the adaptive interference canceller according to 2.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2700896A JP2912866B2 (en) | 1996-02-14 | 1996-02-14 | Adaptive interference canceller |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2700896A JP2912866B2 (en) | 1996-02-14 | 1996-02-14 | Adaptive interference canceller |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPH09223978A true JPH09223978A (en) | 1997-08-26 |
| JP2912866B2 JP2912866B2 (en) | 1999-06-28 |
Family
ID=12209092
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP2700896A Expired - Fee Related JP2912866B2 (en) | 1996-02-14 | 1996-02-14 | Adaptive interference canceller |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JP2912866B2 (en) |
-
1996
- 1996-02-14 JP JP2700896A patent/JP2912866B2/en not_active Expired - Fee Related
Also Published As
| Publication number | Publication date |
|---|---|
| JP2912866B2 (en) | 1999-06-28 |
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