JPH0865083A - Poliphase circuit type phase shift device - Google Patents
Poliphase circuit type phase shift deviceInfo
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- JPH0865083A JPH0865083A JP19525094A JP19525094A JPH0865083A JP H0865083 A JPH0865083 A JP H0865083A JP 19525094 A JP19525094 A JP 19525094A JP 19525094 A JP19525094 A JP 19525094A JP H0865083 A JPH0865083 A JP H0865083A
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- resistor
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Abstract
Description
【０００１】[0001]
【産業上の利用分野】本発明はポリフェーズ回路型位相
推移器に関し、特に互いに１８０度位相の異った２対・
４入力信号から互いに９０度位相の異った４出力信号を
得ることを目的とするポリフェーズ回路型位相推移器に
関する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a polyphase circuit type phase shifter, and more particularly to two pairs of phase shifters each having a phase difference of 180 degrees.
The present invention relates to a polyphase circuit type phase shifter for obtaining four output signals having phases different from each other by 90 degrees from four input signals.
【０００２】[0002]
【従来の技術】従来、音声周波数信号を単側帯波（ＳＳ
Ｂ）変調する方式の一つに、同相および９０度位相が推
移した一対の音声周波数信号によって、同じく同相およ
び９０度位相の推移した一対の搬送波信号をそれぞれ平
衡変調した後、これら平衡変調された二つの信号を合成
する方式がある。このＳＳＢ変調方式においては、音声
周波数信号および搬送波信号ともに、二つの信号間に正
確な９０度位相差を保つ必要がある。しかし、音声周波
数信号は広い帯域を有するので、この帯域全体に亘って
二つの信号間に９０度の位相差を保つのは困難な問題で
あった。この問題を解決するために、抵抗器とコンデン
サとからなる回路網であるポリフェーズ回路型位相推移
器が発表された（例えば、文献１：Ｅｌｅｃｔｒｉｃａ
ｌ Ｃｏｍｍｕｎｉｃａｔｉｏｎ，Ｖｏｌ．４８，Ｎ
ｏ．１＆Ｎｏ．２，ｐｐ２１〜ｐｐ２５，１９７３，Ｅ
ｎｇｌａｎｄ；文献２：ＨＡＭ Ｊｏｕｒｎａｌ，Ｎ
ｏ．４５，ｐｐ１０８〜ｐｐ１１８，１９８６，日
本））。2. Description of the Related Art Conventionally, a voice frequency signal is converted into a single sideband (SS
B) As one of the modulation methods, a pair of voice frequency signals whose in-phase and 90-degree phase are shifted is balanced-modulated on a pair of carrier signals which are also shifted in-phase and 90-degree phase, respectively, and then balanced-modulated. There is a method of combining two signals. In this SSB modulation method, it is necessary to maintain an accurate 90-degree phase difference between the two signals for both the voice frequency signal and the carrier signal. However, since the voice frequency signal has a wide band, it is a difficult problem to maintain the phase difference of 90 degrees between the two signals over the entire band. In order to solve this problem, a polyphase circuit type phase shifter, which is a circuit network including a resistor and a capacitor, has been announced (for example, Reference 1: Electrica).
l Communication, Vol. 48, N
o. 1 & No. 2, pp21 to pp25, 1973, E
ngland; Reference 2: HAM Journal, N
o. 45, pp108 to pp118, 1986, Japan)).
【０００３】以下、従来技術について、図１の本発明に
係わるポリフェーズ回路型位相推移器の回路図を参照し
て説明する。The prior art will be described below with reference to the circuit diagram of the polyphase circuit type phase shifter according to the present invention shown in FIG.
【０００４】本発明に係わるポリフェーズ回路型の位相
推移器１は、４つの入力端子１１，１２，１３および１
４に音声周波数信号ｅ１１，ｅ１２，ｅ１３およびｅ１
４をそれぞれ受け、４つの出力端子ｑ１，ｑ２，ｑ３お
よびｑ４に音声周波数信号ｅｑ１，ｅｑ２，ｅｑ３およ
びｅｑ４をそれぞれ出力する。入力端子１１と出力端子
ｑ１との間にはＰ（Ｐは１以上の整数）個の抵抗器Ｒ１
１，Ｒ２１，Ｒ３１，Ｒ４１，…，Ｒｐ１を縦続に接続
し、入力端子１２と出力端子ｑ２との間にもＰ個の抵抗
器Ｒ１２，Ｒ２２，Ｒ３２，Ｒ４２，…，Ｒｐ２を縦続
に接続し、入力端子１３と出力端子ｑ３との間にもＰ個
の抵抗器Ｒ１３，Ｒ２３，Ｒ３３，Ｒ４３，…，Ｒｐ３
を縦続に接続し、入力端子１４と出力端子ｑ４との間に
もＰ個の抵抗器Ｒ１４，Ｒ２４，Ｒ３４，Ｒ４４，…，
Ｒｐ４を縦続に接続している。なお、符号ｑは（Ｐ＋
１）の数を表すことになる。いま、入力端子１１と出力
端子ｑ１との間を第１行，抵抗器Ｒ１１，Ｒ１２，Ｒ１
３およびＲ１４の並びを第１列というように定義する
と、この位相推移器１は４行・Ｐ列の回路網である。A polyphase circuit type phase shifter 1 according to the present invention has four input terminals 11, 12, 13 and 1.
4 the audio frequency signals e11, e12, e13 and e1
4 and outputs voice frequency signals eq1, eq2, eq3 and eq4 to four output terminals q1, q2, q3 and q4, respectively. Between the input terminal 11 and the output terminal q1, there are P (P is an integer of 1 or more) resistors R1.
1, R21, R31, R41, ..., Rp1 are connected in cascade, and P resistors R12, R22, R32, R42, ..., Rp2 are also connected in cascade between the input terminal 12 and the output terminal q2. , P resistors R13, R23, R33, R43, ..., Rp3 between the input terminal 13 and the output terminal q3.
Are connected in series, and P resistors R14, R24, R34, R44, ... Are also connected between the input terminal 14 and the output terminal q4.
Rp4 is connected in cascade. The symbol q is (P +
It represents the number of 1). Now, between the input terminal 11 and the output terminal q1, the first row, resistors R11, R12, R1
When the arrangement of 3 and R14 is defined as the first column, this phase shifter 1 is a network of 4 rows and P columns.
【０００５】位相推移器１は、上述の抵抗器に加えて、
隣接する行との間に循環的に接続されるコンデンサを各
列に備える。即ち、第１列では、コンデンサＣ１１が第
１行の入力端子１１と第４行の抵抗器Ｒ１４・Ｒ２４接
続点（以下、抵抗器Ｒ１４の出力端というように表現す
る）との間に、コンデンサＣ１２が第２行の入力端子１
２と第１行の抵抗器Ｒ１１の出力端との間に、コンデン
サＣ１３が第３行の入力端子１３と第２行の抵抗器Ｒ１
２の出力端との間に、コンデンサＣ１４が第４行の入力
端子１４と第３行の抵抗器Ｒ１３の出力端との間にそれ
ぞれ接続される。以下、第２列ないし第Ｐ列も同様に、
コンデンサＣ２１ないしＣｐ１は第１行と第４行との間
に、コンデンサＣ２２ないしＣｐ２は第２行と第１行と
の間に、コンデンサＣ２３ないしＣｐ３は第３行と第２
行との間に、コンデンサＣ２４ないしＣｐ４は第４行と
第３行との間にそれぞれ接続される。The phase shifter 1 includes, in addition to the resistors described above,
Each column is provided with a capacitor that is cyclically connected to adjacent rows. That is, in the first column, the capacitor C11 is connected between the input terminal 11 in the first row and the connection point between the resistors R14 and R24 in the fourth row (hereinafter, referred to as the output end of the resistor R14). C12 is the input terminal 1 on the second row
2 and the output terminal of the resistor R11 of the first row, a capacitor C13 is provided between the input terminal 13 of the third row and the resistor R1 of the second row.
The capacitor C14 is connected between the input terminal 14 of the fourth row and the output terminal of the resistor R13 of the third row. Hereinafter, the same applies to the second to Pth columns.
The capacitors C21 to Cp1 are located between the first and fourth rows, the capacitors C22 to Cp2 are located between the second row and the first row, and the capacitors C23 to Cp3 are located between the third row and the second row.
Between the rows, capacitors C24 to Cp4 are connected between the fourth and third rows, respectively.
【０００６】なお、列数Ｐが１の場合は、当然Ｐ＋１＝
２列以降の抵抗器およびコンデンサは接続されず、第１
列の抵抗器およびコンデンサは出力端子ｑ１，ｑ２，ｑ
３およびｑ４に直結される。この場合、出力端子符号の
ｑはＰ＋１＝２と読み替えることになる。以下、Ｐが２
以上の場合も同様である。When the number of columns P is 1, naturally P + 1 =
The resistors and capacitors in the second and subsequent rows are not connected, and the first
The resistors and capacitors in the columns are output terminals q1, q2, q
Directly connected to 3 and q4. In this case, q of the output terminal code is read as P + 1 = 2. Below, P is 2
The same applies to the above cases.
【０００７】ここで、上記抵抗器の抵抗値および上記コ
ンデンサの容量値は、各列ごとに同じ値に設定する。即
ち、抵抗器Ｒ１１，Ｒ１２，Ｒ１３およびＲ１４は互い
に同じ抵抗値Ｒ１を、抵抗器Ｒ２１，Ｒ２２，Ｒ２３お
よびＲ２４は互いに同じ抵抗値Ｒ２を、抵抗器Ｒ３１，
Ｒ３２，Ｒ３３およびＲ３４は互いに同じ抵抗値Ｒ３
を、抵抗器Ｒ４１，Ｒ４２，Ｒ４３およびＲ４４は互い
に同じ抵抗値Ｒ４を、抵抗器Ｒｐ１，Ｒｐ２，Ｒｐ３お
よびＲｐ４は互いに同じ抵抗値Ｒｐを有する。同様に、
コンデンサＣ１１，Ｃ１２，Ｃ１３およびＣ１４は互い
に同じ容量値Ｃ１を、コンデンサＣ２１，Ｃ２２，Ｃ２
３およびＣ２４は互いに同じ容量値Ｃ２を、コンデンサ
Ｃ３１，Ｃ３２，Ｃ３３およびＣ３４は互いに同じ容量
値Ｃ３を、コンデンサＣ４１，Ｃ４２，Ｃ４３およびＣ
４４は互いに同じ容量値Ｃ４を、コンデンサＣｐ１，Ｃ
ｐ２，Ｃｐ３およびＣｐ４は互いに同じ容量値Ｃｐを有
する。Here, the resistance value of the resistor and the capacitance value of the capacitor are set to the same value for each column. That is, the resistors R11, R12, R13 and R14 have the same resistance value R1, the resistors R21, R22, R23 and R24 have the same resistance value R2, and the resistor R31,
R32, R33 and R34 have the same resistance value R3
The resistors R41, R42, R43 and R44 have the same resistance value R4, and the resistors Rp1, Rp2, Rp3 and Rp4 have the same resistance value Rp. Similarly,
The capacitors C11, C12, C13 and C14 have the same capacitance value C1 as each other, and the capacitors C21, C22 and C2 have the same capacitance value C1.
3 and C24 have the same capacitance value C2, capacitors C31, C32, C33 and C34 have the same capacitance value C3, and capacitors C41, C42, C43 and C34.
Reference numeral 44 designates capacitors Cp1, C having the same capacitance value C4 as each other.
p2, Cp3 and Cp4 have the same capacitance value Cp.
【０００８】図１の回路図において、信号発生器２は、
位相推移器１の入力端子１１および１２に電圧Ｅで且つ
同位相の音声周波数信号ｅ１１およびｅ１２を信号発生
回路３および４からそれぞれ供給し、入力端子１３およ
び１４に電圧−Ｅで且つ同位相の音声周波数信号ｅ１３
およびｅ１４を信号発生回路５および６からそれぞれ供
給する。なお、信号発生回路３と５と，および信号発生
回路４と６とは、一つの音声周波数信号から１８０度位
相の異なる二つの信号を広帯域に亘って取出すことので
きる一つのプッシュプル増幅器でそれぞれ同時に構成で
きる。In the circuit diagram of FIG. 1, the signal generator 2 is
The phase shifter 1 supplies the input terminals 11 and 12 of the audio frequency signals e11 and e12 having the voltage E of the same phase from the signal generating circuits 3 and 4, respectively, and supplies the input terminals 13 and 14 of the voltage -E and of the same phase. Voice frequency signal e13
And e14 are supplied from the signal generating circuits 5 and 6, respectively. Each of the signal generating circuits 3 and 5 and the signal generating circuits 4 and 6 is a push-pull amplifier capable of extracting two signals having a phase difference of 180 degrees from one voice frequency signal over a wide band. Can be configured at the same time.
【０００９】位相推移器１において、第Ｉ（Ｉは１ない
しＰの整数）列のポール周波数ｆｉを（１）式で定義す
る。In the phase shifter 1, the pole frequency fi of the I-th column (I is an integer of 1 to P) is defined by the equation (1).
【００１０】 ｆｉ＝１／（２πＣｉ・Ｒｉ） …（１） ポール周波数ｆｉにおいては、例え、この位相推移器１
の列数Ｐが１であっても、出力端子ｑ１，ｑ２，ｑ３お
よびｑ４に生じる音声周波数信号ｅｑ１，ｅｑ２，ｅｑ
３およびｅｑ４は、互いに９０度異なる位相関係とな
る。しかし、列数Ｐが少ないと、上記出力端子間の位相
差を正確に９０度を保つ帯域が狭いので、一般には、９
０度位相差が得られる帯域を確保できるように列数Ｐを
増加する。また、ポール周波数ｆｉを各列Ｉごとに変え
ることにより、少い列数Ｐで広帯域性を確保することも
ある。Fi = 1 / (2πCi · Ri) (1) At the pole frequency fi, for example, this phase shifter 1
Even if the number of columns P of 1 is 1, the audio frequency signals eq1, eq2, eq generated at the output terminals q1, q2, q3 and q4
3 and eq4 have a phase relationship different from each other by 90 degrees. However, when the number of columns P is small, the band for keeping the phase difference between the output terminals at 90 degrees is narrow, so that in general, 9
The number of columns P is increased so as to secure a band in which a 0 degree phase difference can be obtained. Also, by changing the pole frequency fi for each column I, a wide number of columns may be secured with a small number P of columns.
【００１１】文献２では、図１の構成による４行・１０
列位相推移器のシミュレーション結果を紹介している。
この位相推移器は、第１列のポール周波数ｆ１を２３４
Ｈｚとし、第２列以降のポール周波数ｆｉを約１．２１
の等比級数で増加させ、第１０列のポール周波数ｆ１０
を２．８４２ＫＨｚに設定している（第１表）。また、
各列の抵抗値Ｒｉは、第１列の抵抗値Ｒ１を６８ｋΩと
し、第２列以降の抵抗値Ｒｉを約１／１．２１の等比級
数で減少させ、第１０列の抵抗値Ｒ１０を５．６ｋΩに
設定している。この場合、全てのコンデンサの容量値は
ほぼ同一になる。文献２の位相推移器は、この条件にお
いて、周波数１４０Ｈｚ程度から５ＫＨｚ程度まで２度
以下の出力信号間位相差を得ることができる（第１１
図）。しかし、この位相推移器の出力電圧は、その中心
周波数付近である８００Ｈｚ付近に約３０ｄＢの減衰ピ
ークを持つとともに上記の周波数範囲において５ｄＢ程
度の振幅偏差を持っている（第１０図）。なお、この減
衰ピークおよび振幅偏差は列数とともに大きくなること
が知られている。In Reference 2, 4 lines × 10 with the configuration of FIG.
The simulation results of the column phase shifter are introduced.
This phase shifter sets the pole frequency f1 of the first column to 234
Hz and the pole frequency fi of the second and subsequent columns is about 1.21.
The pole frequency f10 of the 10th column.
Is set to 2.842 KHz (Table 1). Also,
With respect to the resistance value Ri of each column, the resistance value R1 of the first column is set to 68 kΩ, the resistance value Ri of the second and subsequent columns is reduced by a geometric progression of about 1 / 1.21, and the resistance value R10 of the tenth column is It is set to 5.6 kΩ. In this case, the capacitance values of all capacitors are almost the same. Under this condition, the phase shifter of Document 2 can obtain a phase difference between output signals of 2 degrees or less from a frequency of about 140 Hz to about 5 KHz (11th phase).
Figure). However, the output voltage of this phase shifter has an attenuation peak of about 30 dB near 800 Hz, which is near its center frequency, and an amplitude deviation of about 5 dB in the above frequency range (Fig. 10). It is known that the attenuation peak and the amplitude deviation increase with the number of columns.
【００１２】[0012]
【発明が解決しようとする課題】上述した従来技術によ
るポリフェーズ回路型位相推移器は、列数を増大させる
ことにより、出力信号間の位相差を広帯域に亘って９０
度近傍に保つことができるが、出力信号に大きな減衰量
および振幅偏差を生じ、これら減衰量および振幅偏差は
列数とともに増大するという欠点があった。The polyphase circuit type phase shifter according to the prior art described above increases the number of columns so that the phase difference between the output signals is 90 over a wide band.
However, there is a drawback in that a large amount of attenuation and amplitude deviation occur in the output signal, and these attenuation and amplitude deviation increase with the number of rows.
【００１３】上述のＳＳＢ変調方式においては、不要側
帯波抑圧比を４０ｄＢ以上にするには、音声周波数信号
の信号間位相差を１．０度以下と規定すると、振幅偏差
を０．１７ｄＢ以下にしなければならない（文献２，ｐ
ｐ１１０）。この振幅偏差は等化増幅することによって
補償することは原理的には可能であるが、もとの信号の
振幅偏差が大きくなるほど、規格を保つための偏差補償
が困難になってくるという問題が残る。また、等化増幅
器を用いることにより構成が複雑になるばかりでなく製
造費用が増大するという欠点がある。In the above-mentioned SSB modulation method, in order to set the unwanted sideband suppression ratio to 40 dB or more, if the signal phase difference between voice frequency signals is specified to be 1.0 degree or less, the amplitude deviation is set to 0.17 dB or less. Must be (Reference 2, p.
p110). Although it is possible in principle to compensate for this amplitude deviation by equalizing and amplifying it, the larger the amplitude deviation of the original signal, the more difficult it becomes to compensate the deviation for maintaining the standard. Remain. Further, the use of the equalizing amplifier not only complicates the configuration but also increases the manufacturing cost.
【００１４】[0014]
【課題を解決するための手段】本発明のポリフェーズ回
路型位相推移器は、４行をなす第１，第２，第３および
第４の入力端子と、前記第１，第２，第３および第４の
入力端子にそれぞれ行対応する第１，第２，第３および
第４の出力端子と、前記各行の入力端子と出力端子との
間にそれぞれ縦続接続されてＰ（Ｐは２以上の整数）列
をなすとともに各列ごとには抵抗値をほぼ等しくした抵
抗器と、第Ｊ（Ｊは１ないし４の整数）行・第Ｉ（Ｉは
１ないしＰの整数）列の前記抵抗器の入力端と第（Ｊ−
１，但しＪ−１＝０は４と読み替える）行・第Ｉ列の前
記抵抗器の出力端とを循環的に接続するとともに各列ご
とには容量値をほぼ等しくしたコンデンサとを備えるポ
リフェーズ回路型位相推移器において、第Ｉ列の前記抵
抗器の抵抗値Ｒｉと第（Ｉ＋１）列の前記抵抗器の抵抗
値Ｒ（ｉ＋１）との比であるｎ＝Ｒ（ｉ＋１）／Ｒｉを
１．３から４の範囲とする。A polyphase circuit type phase shifter according to the present invention comprises four rows of first, second, third and fourth input terminals and the first, second, third. And the first, second, third and fourth output terminals respectively corresponding to the first and fourth input terminals in a row, and P (P is 2 or more) connected in cascade between the input terminal and the output terminal of each row. Resistors having a substantially equal resistance value in each column, and the resistors in the Jth row (J is an integer of 1 to 4) and the Ith row (I is an integer of 1 to P). Input end of the container and the (J-
1, provided that J-1 = 0 is read as 4) Polyphase provided with a capacitor connected to the output terminal of the resistor in the row / Ith column in a cyclic manner and having a substantially equal capacitance value in each column In the circuit type phase shifter, n = R (i + 1) / Ri, which is the ratio of the resistance value Ri of the resistor in the I-th column to the resistance value R (i + 1) of the resistor in the (I + 1) -th column is set to 1 The range is from 3 to 4.
【００１５】前記ポリフェーズ回路型位相推移器の一つ
は、第Ｉ列の前記抵抗器の抵抗値Ｒｉと第Ｉ列の前記コ
ンデンサの容量値Ｃｉの積と第（Ｉ＋１）列の前記抵抗
器の抵抗値Ｒ（Ｉ＋１）と第（Ｉ＋１）列の前記コンデ
ンサの容量値Ｃ（ｉ＋１）と積との比Ｒｉ・Ｃｉ／｛Ｒ
（ｉ＋１）・Ｃ（ｉ＋１）｝をｍとした場合、前記ｎを
｛（ｍ＋１）＋（ｍ2 ＋６ｍ＋１）1/2 ｝／（２ｍ）の
０．８倍から２倍までの範囲とする構成をとることがで
きる。One of the polyphase circuit type phase shifters is a product of a resistance value Ri of the resistor in the I-th column and a capacitance value Ci of the capacitor in the I-th column and the resistor in the (I + 1) -th column. Ratio of the resistance value R (I + 1) of the capacitor and the capacitance value C (i + 1) of the capacitor in the (I + 1) th column to the product Ri · Ci / {R
When (i + 1) · C (i + 1)} is m, n is in the range of 0.8 to 2 times {(m + 1) + (m 2 + 6m + 1) 1/2 } / (2m) Can be taken.
【００１６】前記ポリフェーズ回路型位相推移器の別の
一つは、前記ｎおよび前記ｍを各列ともほぼ等しくした
構成をとることができる。Another one of the polyphase circuit type phase shifters may have a configuration in which the n and the m are substantially equal in each column.
【００１７】[0017]
【実施例】次に本発明について図面を参照して説明す
る。The present invention will be described below with reference to the drawings.
【００１８】図１は本発明の一実施例の回路図であり、
図２は図１の実施例においてＰ＝２列とした場合の位相
推移器１Ａの回路図である。また、図３は位相推移器１
Ａの特性を示す図であり、（ａ）は出力振幅特性、
（ｂ）は隣接する出力端子間の位相差特性である。FIG. 1 is a circuit diagram of an embodiment of the present invention.
FIG. 2 is a circuit diagram of the phase shifter 1A when P = 2 columns in the embodiment of FIG. Further, FIG. 3 shows the phase shifter 1
It is a figure which shows the characteristic of A, (a) is an output amplitude characteristic,
(B) is a phase difference characteristic between adjacent output terminals.
【００１９】本発明においても、ポリフェーズ回路型の
位相推移器１の回路図自体は、図１を参照して説明した
従来例と変るところはない。しかしながら、本発明の特
徴は、位相推移器１の出力端子ｑ１ないしｑ４における
音声周波数信号ｅｑ１ないしｅｑ４の帯域内振幅偏差お
よび減衰量を減少させるために発明努力し、第２列以降
に配置する抵抗器の抵抗値Ｒｉの適正値を見い出したと
ころにある。まず、Ｐ＝２列の位相推移器１Ａにおける
適正な抵抗値Ｒｉを計算によって求める。Also in the present invention, the circuit diagram itself of the polyphase circuit type phase shifter 1 is not different from the conventional example described with reference to FIG. However, the feature of the present invention is that the efforts are made to reduce the in-band amplitude deviation and the attenuation amount of the audio frequency signals eq1 to eq4 at the output terminals q1 to q4 of the phase shifter 1, and the resistors arranged in the second and subsequent columns are used. The optimum value of the resistance value Ri of the container is found. First, an appropriate resistance value Ri in the phase shifter 1A of P = 2 columns is calculated.
【００２０】図２を参照すると、この位相推移器１Ａ
は、Ｐ＝２列であり、図１の位相推移器１の第３列以降
を省いたものである。従って、出力端子ｑ１，ｑ２，ｑ
３およびｑ４はそれぞれ出力端子３１，３２，３３およ
び３４と具体的に示し、音声周波数信号ｅｑ１，ｅｑ
２，ｅｑ３およびｅｑ４も音声周波数信号ｅ３１，ｅ３
２，ｅ３３およびｅ３４と具体的に示している。Referring to FIG. 2, this phase shifter 1A
Shows P = 2 columns and omits the third and subsequent columns of the phase shifter 1 of FIG. Therefore, the output terminals q1, q2, q
3 and q4 are specifically shown as output terminals 31, 32, 33, and 34, respectively, and audio frequency signals eq1, eq
2, eq3 and eq4 are also voice frequency signals e31, e3
2, e33 and e34.
【００２１】この位相推移器１Ａにキルヒホッフの法則
を適用して（２）式を得る。Kirchhoff's law is applied to the phase shifter 1A to obtain the equation (2).
【００２２】 Ｒ１（ｉ１１−ｉ２４）−ｊＸＣ１（ｉ１１−ｉ２１）＝０ Ｒ１（ｉ１２−ｉ２１）−ｊＸＣ１（ｉ１２−ｉ２２）＝２Ｅ Ｒ１（ｉ１３−ｉ２２）−ｊＸＣ１（ｉ１３−ｉ２３）＝０ Ｒ１（ｉ１４−ｉ２３）−ｊＸＣ１（ｉ１４−ｉ２４）＝−２Ｅ Ｒ１（ｉ２１−ｉ１２）−ｊＸＣ１（ｉ２１−ｉ１１） ＋（Ｒ２−ｊＸＣ２）・ｉ２１＝０ Ｒ１（ｉ２２−ｉ１３）−ｊＸＣ１（ｉ２２−ｉ１２） ＋（Ｒ２−ｊＸＣ２）・ｉ２２＝０ Ｒ１（ｉ２３−ｉ１４）−ｊＸＣ１（ｉ２３−ｉ１３） ＋（Ｒ２−ｊＸＣ２）・ｉ２３＝０ Ｒ１（ｉ２４−ｉ１１）−ｊＸＣ１（ｉ２４−ｉ１４） ＋（Ｒ２−ｊＸＣ２）・ｉ２４＝０ ＸＣ１＝１／（ω・Ｃ１），ＸＣ２＝１／（ω・Ｃ２） …（２） なお、ｉ１１は第１行および第２行において信号発生回
路３，抵抗器Ｒ１１，コンデンサＣ１２および信号発生
回路４を循環するメッシュ電流であり、ｉ２１は同じ行
において抵抗器２１，コンデンサＣ２２，抵抗器Ｒ１２
およびコンデンサＣ１２を循環するメッシュ電流であ
る。以下、各行間を循環するメッシュ電流ｉ１２ないし
ｉ１４およびｉ２２ないしｉ２４を図示の如く定義して
いる。R1 (i11-i24) -jXC1 (i11-i21) = 0 R1 (i12-i21) -jXC1 (i12-i22) = 2E R1 (i13-i22) -jXC1 (i13-i23) = 0 R1 ( i14-i23) -jXC1 (i14-i24) =-2E R1 (i21-i12) -jXC1 (i21-i11) + (R2-jXC2) .i21 = 0 R1 (i22-i13) -jXC1 (i22-i12) + (R2-jXC2) · i22 = 0 R1 (i23-i14) -jXC1 (i23-i13) + (R2-jXC2) · i23 = 0 R1 (i24-i11) -jXC1 (i24-i14) + (R2- jXC2) · i24 = 0 XC1 = 1 / (ω · C1), XC2 = 1 / (ω · C2) (2) Note that i11 is in the first and second rows. Is a mesh current circulating in the signal generating circuit 3, the resistor R11, the capacitor C12, and the signal generating circuit 4, and i21 is the resistor 21, the capacitor C22, and the resistor R12 in the same row.
And the mesh current circulating in the capacitor C12. Hereinafter, the mesh currents i12 to i14 and i22 to i24 circulating between the rows are defined as shown.
【００２３】ここで、ω１＝１／（Ｃ１・Ｒ１），つま
り第１列のポール周波数ｆ１＝１／（２π・Ｃ１・Ｒ
１），ω２＝１／（Ｃ２・Ｒ２），つまり第２列のポー
ル周波数ｆ２＝１／（２π・Ｃ２・Ｒ２）、ｎ＝Ｒ２／
Ｒ１として（２）式を解くと、（３）式が得られる。Here, ω1 = 1 / (C1 · R1), that is, the pole frequency f1 = 1 / (2π · C1 · R) of the first column.
1), ω2 = 1 / (C2 · R2), that is, the pole frequency f2 = 1 / (2π · C2 · R2) of the second column, n = R2 /
When equation (2) is solved for R1, equation (3) is obtained.
【００２４】 ｅ３１／Ｅ＝｛Ａ（ω）＋ｊＢ（ω）｝／｛Ｃ（ω）＋ｊＤ（ω）｝ ｅ３２／Ｅ＝｛Ａ（ω）−ｊＢ（ω）｝／｛Ｃ（ω）＋ｊＤ（ω）｝ ｅ３３／Ｅ＝−｛Ａ（ω）＋ｊＢ（ω）｝／｛Ｃ（ω）＋ｊＤ（ω）｝ ｅ３４／Ｅ＝−｛Ａ（ω）−ｊＢ（ω）｝／｛Ｃ（ω）＋ｊＤ（ω）｝ Ａ（ω）＝１＋ω2 ／（ω１・ω２） Ｂ（ω）＝ω／ω１＋ω／ω２ Ｃ（ω）＝１−ω2 ／（ω１・ω２） Ｄ（ω）＝２ω／｛（ｎ・ω２）＋ω／ω１＋ω／ω２｝ …（３） 図３（ａ）はｎをパラメータとした出力端子３１等にお
ける音声周波数信号ｅ３１の振幅特性，例えばｅ３１／
Ｅの絶対値（入出力電圧比）をｄＢ表示で示し、図３
（ｂ）は隣接する出力端子間（例えば端子３１と３２と
の間）の位相差，例えば（ｅ３１／Ｅ）と（ｅ３２／
Ｅ）との位相差を示している。なお、第１列と第２列の
ポール周波数ｆ１とｆ２とは同一としている。E31 / E = {A (ω) + jB (ω)} / {C (ω) + jD (ω)} e32 / E = {A (ω) −jB (ω)} / {C (ω) + jD (Ω)} e33 / E = − {A (ω) + jB (ω)} / {C (ω) + jD (ω)} e34 / E = − {A (ω) −jB (ω)} / {C ( ω) + jD (ω)} A (ω) = 1 + ω 2 / (ω1 · ω2) B (ω) = ω / ω1 + ω / ω2 C (ω) = 1−ω 2 / (ω1 · ω2) D (ω) = 2ω / {(n · ω2) + ω / ω1 + ω / ω2} (3) FIG. 3A shows an amplitude characteristic of the audio frequency signal e31 at the output terminal 31 or the like with n as a parameter, for example, e31 /
The absolute value of E (input / output voltage ratio) is shown in dB and is shown in FIG.
(B) is a phase difference between adjacent output terminals (for example, between terminals 31 and 32), for example, (e31 / E) and (e32 /
It shows the phase difference with E). The pole frequencies f1 and f2 in the first and second columns are the same.
【００２５】この位相推移器１Ａは、わずか２列の構成
であるため、９０度の出力端子間位相差が得られる帯域
はポール周波数ｆｉの近傍しかない。一方、振幅特性
は、ｎ＝２．４１とすると、信号ｅ３１（ｅ３２，ｅ３
３，ｅ３４も同じ）は全周波数に亘って平坦でしかも減
衰のない振幅特性を生じる。この振幅特性はｎを１以下
で構成していた従来の位相推移器がポール周波数ｆ１付
近で３ｄＢの減衰量とこれと同程度の振幅偏差を有する
のに比べて明らかに優れている。また、ｎ＝１００程度
に大きくすると、逆にポール周波数ｆ１付近で３ｄＢの
振幅増加を示し、振幅偏差もｎ＝１の場合と同程度にな
る。Since the phase shifter 1A has a structure of only two columns, the band in which the phase difference between the output terminals of 90 degrees can be obtained is only near the pole frequency fi. On the other hand, if the amplitude characteristic is n = 2.41, the signal e31 (e32, e3)
3 and e34 are also the same), an amplitude characteristic that is flat over all frequencies and has no attenuation is generated. This amplitude characteristic is clearly superior to that of the conventional phase shifter in which n is set to 1 or less, which has an attenuation amount of 3 dB near the pole frequency f1 and an amplitude deviation of the same degree. On the other hand, when the value is increased to about n = 100, on the contrary, the amplitude increases by 3 dB near the pole frequency f1, and the amplitude deviation becomes about the same as when n = 1.
【００２６】図２の如き列数Ｐが２程度の位相推移器
は、音声周波数信号のように広い帯域の信号に対しては
出力信号間に正確な９０度位相差を生じる帯域が狭すぎ
るという問題がある。そこで、図１の位相推移器１は、
例えば信号ｅｑ１とｅｑ２間に９０度位相差を生じる帯
域を増加させるために、列数Ｐを増やし、また各列のポ
ール周波数ｆｉを変化させる必要がある。The phase shifter with the number of columns P of about 2 as shown in FIG. 2 is said to have a too narrow band for producing an accurate 90-degree phase difference between output signals for a wide band signal such as a voice frequency signal. There's a problem. Therefore, the phase shifter 1 of FIG.
For example, it is necessary to increase the number of columns P and change the pole frequency fi of each column in order to increase the band in which a 90-degree phase difference occurs between the signals eq1 and eq2.
【００２７】いま、図１の位相推移器１において、第Ｉ
列の抵抗器の抵抗値Ｒｉと第Ｉ列のコンデンサの容量値
Ｃｉの積にと第（Ｉ＋１）列の抵抗器の抵抗値Ｒ（Ｉ＋
１）と第（Ｉ＋１）列のコンデンサの容量値Ｃ（ｉ＋
１）の積との比Ｒｉ・Ｃｉ／｛Ｒ（ｉ＋１）・Ｃ（ｉ＋
１）｝をｍとする。このｍは第Ｉ列のポール周波数と第
（Ｉ＋１）列のポール周波数との比ｆ（ｉ＋１）／ｆ
ｉ）でもある。このｍを変える場合には、各列の抵抗器
を（４）式に示す抵抗比ｎで構成すると、この位相推移
器１は全信号帯域に亘って振幅偏差のない音声周波数信
号ｅｑ１等を生じる。Now, in the phase shifter 1 of FIG.
The product of the resistance value Ri of the resistor of the column and the capacitance value Ci of the capacitor of the I-th column and the resistance value R (I + of the resistor of the (I + 1) -th column
1) and the capacitance value C (i +) of the capacitors in the (I + 1) th column
1) ratio of product Ri * Ci / {R (i + 1) * C (i +
1)} is m. This m is the ratio f (i + 1) / f of the pole frequency of the I-th column and the pole frequency of the (I + 1) -th column.
It is also i). In the case of changing this m, if the resistors in each column are constituted by the resistance ratio n shown in the equation (4), this phase shifter 1 produces the audio frequency signal eq1 and the like having no amplitude deviation over the entire signal band. .
【００２８】 ｎ＝｛（ｍ＋１）＋（ｍ2 ＋６ｍ＋１）1/2 ｝／（２ｍ） …（４） 即ち、抵抗比ｎ，ポール周波数比ｍ＝ω２／ω１，正規
化周波数ｘ＝ω／ω１を用いて（３）式を解くと、音声
周波数信号ｅ３１，ｅ３２，ｅ３３およびｅ３４の振幅
の絶対値は全て等しいので、（５）式が得られる。N = {(m + 1) + (m 2 + 6m + 1) 1/2 } / (2m) (4) That is, the resistance ratio n, the pole frequency ratio m = ω2 / ω1, the normalized frequency x = ω / ω1 When the equation (3) is solved by using, the absolute values of the amplitudes of the voice frequency signals e31, e32, e33, and e34 are all equal, and thus the equation (5) is obtained.
【００２９】 （１＋ｘ2 ／ｍ）2 ＋（１＋１／ｍ）2 ×ｘ2 ＝（１−ｘ2 ／ｍ）2 ＋｛１＋１／ｍ＋２／（ｍ・ｎ）｝2 ×ｘ2 …（５） （５）式を解くと、正規化周波数ｘに関係のない上記
（４）式が得られる。この（５）式，従って（４）式は
位相推移器１の列数をＰ＝２列より増しても成立する。(1 + x 2 / m) 2 + (1 + 1 / m) 2 × x 2 = (1-x 2 / m) 2 + {1 + 1 / m + 2 / (m · n)} 2 × x 2 (5) By solving the equation (5), the above equation (4) irrelevant to the normalized frequency x is obtained. The equation (5), and therefore the equation (4), holds even if the number of columns of the phase shifter 1 is increased from P = 2 columns.
【００３０】図４は図１の位相推移器１におけるポール
周波数比ｍに対する適正抵抗比ｎ０を（４）式に従って
示す図である。FIG. 4 is a diagram showing the proper resistance ratio n0 with respect to the pole frequency ratio m in the phase shifter 1 of FIG. 1 according to the equation (4).
【００３１】ポール周波数比ｍが１の場合は、図２の位
相推移器１Ａの場合と同じであり、抵抗比ｎ＝２．４１
において出力端子ｑ１等における信号ｅｑ１等の減衰お
よび振幅偏差をなくすることができる。ｍが増大するに
つれて，つまりｆ（ｉ＋１）／ｆｉが大きくなるにつれ
て適正なｎ値は減少するが、ｎ値は１より常に大きい。When the pole frequency ratio m is 1, it is the same as the case of the phase shifter 1A of FIG. 2, and the resistance ratio n = 2.41.
At, the attenuation and amplitude deviation of the signal eq1 and the like at the output terminal q1 and the like can be eliminated. The proper n value decreases as m increases, that is, as f (i + 1) / fi increases, but the n value is always greater than 1.
【００３２】図５は列数Ｐ＝４の位相推移器１の実験デ
ータを示す図であり、（ａ）は出力振幅特性、（ｂ）は
出力位相差特性である。FIG. 5 is a diagram showing experimental data of the phase shifter 1 having the number of columns P = 4, where (a) is the output amplitude characteristic and (b) is the output phase difference characteristic.
【００３３】位相推移器１は、上述したＳＳＢ変調方式
の音声周波数信号用として用いるには、少くとも３００
Ｈｚから３ｋＨｚまでの音声周波数帯域で９０度の位相
差を隣接する出力端子間に確保する必要がある。そこ
で、位相推移器１の実験回路は、列数をＰ＝４，使用帯
域を最小周波数ｆｍｉｎ．＝３００Ｈｚ，最高周波数ｆ
ｍａｘ．＝３ｋＨｚ（ｆｍａｘ．／ｆｍｉｎ．＝１０）
として諸パラメータを選定した。ここで、ポール周波数
比ｍを等比級数的に設定すると、少ない列数Ｐでも出力
信号間位相差，例えば信号ｅｑ１とｅｑ２との間の位相
差を少なくできることが従来から知られている。上述の
とおりに設定すると、ポール周波数比ｍ＝２．１５にな
り、各列のポール周波数ｆｉは、ｆ１＝３００Ｈｚ，ｆ
２＝６４３．３Ｈｚ，ｆ３＝１．３９２ｋＨｚ，ｆ４＝
３．０ｋＨｚとなる。また、（４）式に従うと、ポール
周波数比ｍを各列とも同じにする場合には、抵抗比ｎも
各列とも同一にすべきであり、適正抵抗比ｎ０はｎ０＝
１．５となる。The phase shifter 1 is at least 300 in order to be used for the voice frequency signal of the above-mentioned SSB modulation system.
It is necessary to secure a phase difference of 90 degrees between adjacent output terminals in the audio frequency band from Hz to 3 kHz. Therefore, the experimental circuit of the phase shifter 1 has the number of columns P = 4 and the used band of the minimum frequency fmin. = 300Hz, maximum frequency f
max. = 3 kHz (fmax./fmin.=10)
Various parameters were selected as Here, it is conventionally known that setting the pole frequency ratio m in a geometric progression can reduce the phase difference between output signals, for example, the phase difference between the signals eq1 and eq2, even with a small number of columns P. When set as described above, the pole frequency ratio m = 2.15, and the pole frequency fi of each column is f1 = 300 Hz, f
2 = 643.3 Hz, f3 = 1.392 kHz, f4 =
It becomes 3.0 kHz. Further, according to the equation (4), when the pole frequency ratio m is the same in each column, the resistance ratio n should be the same in each column, and the proper resistance ratio n0 is n0 =
It becomes 1.5.
【００３４】図５の実験データは、実験例Ａが抵抗比ｎ
をポール周波数比ｍ＝２．１５における適正値に近いｎ
≒１．７（ｎ／ｎ０＝１．１３），実験例Ｂがやや高い
値であるｎ≒２．２（ｎ／ｎ０＝１．４７），実験例Ｃ
が従来例と変らない値であるｎ＝１（ｎ／ｎ０＝０．６
７）の３種類を示している。なお、実験に用いた位相推
移器１は、市販でしかも安く手に入る部品で構成してい
る。このため、ポール周波数比ｍおよび抵抗値ｎは、上
記設定値とは必らずしも一致せず多少のずれがある。表
１に実験した位相推移器１の諸パラメータを示す。In the experimental data of FIG. 5, the resistance ratio n in the experimental example A is n.
Is close to an appropriate value when the pole frequency ratio m = 2.15
≈1.7 (n / n0 = 1.13), Experimental Example B has a slightly higher value n≈2.2 (n / n0 = 1.47), Experimental Example C
Is the same value as the conventional example, n = 1 (n / n0 = 0.6
3) of 7) is shown. The phase shifter 1 used in the experiment is composed of commercially available and inexpensive parts. For this reason, the pole frequency ratio m and the resistance value n do not always match the above-mentioned set values, and there is some deviation. Table 1 shows various parameters of the experimental phase shifter 1.
【００３５】[0035]
【表１】 [Table 1]
【００３６】図５に示した実験例Ａは、周波数３００Ｈ
ｚないし３ｋＨｚの帯域において、９０度＋０．１９度
−０．１２度以内の出力端子間の信号位相差を保ってい
る。また、ｎ値の異なる実験例ＢおよびＣも実験例Ａに
ほぼ等しい位相差特性を持っている。The experimental example A shown in FIG. 5 has a frequency of 300H.
In the band of z to 3 kHz, the signal phase difference between the output terminals is maintained within 90 ° + 0.19 ° −0.12 °. Further, Experimental Examples B and C having different n values also have a phase difference characteristic substantially equal to that of Experimental Example A.
【００３７】一方、実験例Ａの出力振幅特性は、上記帯
域内において０．３ｄＢの振幅偏差でありしかも出力信
号に減衰がない。On the other hand, the output amplitude characteristic of Experimental Example A has an amplitude deviation of 0.3 dB within the above band, and there is no attenuation in the output signal.
【００３８】また、実験例Ｂは、抵抗比ｎを適正値抵抗
比ｎ０＝１．５の１．５倍程度であるｎ≒２．２に設定
したため、図３（ａ）から予想されるごとく帯域の中央
付近で入出力電圧比が増大し、最大＋１．２ｄＢの振幅
偏差を有する。しかし、この実験例Ｂは、上記帯域内で
最大６ｄＢの出力電圧の減衰および約２ｄＢの振幅偏差
を有する抵抗比ｎ＝１の実験例Ｃと比べると、はるかに
よい出力振幅特性を有する。Further, in Experimental Example B, the resistance ratio n was set to n≈2.2 which is about 1.5 times the appropriate value resistance ratio n0 = 1.5, and as expected from FIG. 3 (a). The input / output voltage ratio increases near the center of the band, and the maximum amplitude deviation is +1.2 dB. However, this experimental example B has a much better output amplitude characteristic than the experimental example C in which the resistance ratio n = 1 has the maximum output voltage attenuation of 6 dB and the amplitude deviation of about 2 dB in the band.
【００３９】位相推移器１の好ましい抵抗比ｎを考察す
ると、図３，図４および図５から、１．３〈ｎ〈４程度
がよいことが分かる。また、好ましい抵抗比ｎは、式
（４）および図６から分かるとおり、ポール周波数比ｍ
によって変わるので、上記条件のもとで、抵抗比ｎをポ
ール周波数比ｍの関数として｛（ｍ＋１）＋（ｍ2 ＋６
ｍ＋１）1/2 ｝／（２ｍ）の０．８倍から２倍までの範
囲とするのがさらに好ましい。Considering the preferable resistance ratio n of the phase shifter 1, it can be seen from FIG. 3, FIG. 4 and FIG. 5 that 1.3 <n <4 is preferable. In addition, the preferable resistance ratio n is, as can be seen from the equation (4) and FIG. 6, the pole frequency ratio m.
As a function of the pole frequency ratio m, the resistance ratio n is {(m + 1) + (m 2 +6) under the above conditions.
It is more preferable that the range is 0.8 times to 2 times m + 1) 1/2 } / (2m).
【００４０】[0040]
【発明の効果】以上説明したように本発明は、第Ｉ列の
抵抗器の抵抗値Ｒｉと第（Ｉ＋１）列の抵抗器の抵抗値
Ｒ（ｉ＋１）との比であるｎ＝Ｒ（ｉ＋１）／Ｒｉを
１．３から４の範囲とするので、列数を増加させること
により出力信号間の位相差を広帯域に亘って９０度近傍
に保つことができるとともに、上記列数を増加させても
出力信号の減衰量および振幅偏差を少なくすることがで
きるという効果がある。As described above, according to the present invention, the ratio of the resistance value Ri of the resistor in the I-th column to the resistance value R (i + 1) of the resistor in the (I + 1) -th column is n = R (i + 1). ) / Ri is in the range of 1.3 to 4, the phase difference between the output signals can be kept close to 90 degrees over a wide band by increasing the number of columns, and the number of columns can be increased. Also has the effect of reducing the amount of attenuation and amplitude deviation of the output signal.
【００４１】従って本発明の位相推移器をＳＳＢ変調回
路の音声周波数信号供給回路に用いると、等化増幅器を
不要とするか、必要であっても簡単な構成とすることが
できるので、上記ＳＳＢ変調回路は構成を簡単にできる
だけでなく製造費用を軽減できるという効果もある。Therefore, when the phase shifter of the present invention is used in the audio frequency signal supply circuit of the SSB modulation circuit, the equalization amplifier is not necessary, or even if it is necessary, the structure can be simplified. The modulation circuit not only has a simple structure, but also has the effect of reducing manufacturing costs.
【図１】本発明によるポリフェーズ回路型位相推移器の
一実施例の回路図である。FIG. 1 is a circuit diagram of an embodiment of a polyphase circuit type phase shifter according to the present invention.
【図２】本実施例においてＰ＝２列とした場合の位相推
移器１Ａの回路図である。FIG. 2 is a circuit diagram of a phase shifter 1A when P = 2 columns in this embodiment.
【図３】図２の位相推移器１Ａの出力信号特性を示す図
であり、（ａ）は振幅特性、（ｂ）は出力位相差特性で
ある。3A and 3B are diagrams showing output signal characteristics of the phase shifter 1A of FIG. 2, where FIG. 3A is an amplitude characteristic and FIG. 3B is an output phase difference characteristic.
【図４】本実施例の位相推移器１におけるポール周波数
比ｍに対する適正抵抗比ｎ０を示す図である。FIG. 4 is a diagram showing an appropriate resistance ratio n0 with respect to a pole frequency ratio m in the phase shifter 1 of the present embodiment.
【図５】列数Ｐ＝４の場合の位相推移器１の実験データ
を示す図であり、（ａ）は出力振幅特性、（ｂ）は出力
位相差特性である。FIG. 5 is a diagram showing experimental data of the phase shifter 1 when the number of columns P = 4, where (a) is an output amplitude characteristic and (b) is an output phase difference characteristic.
１，１Ａ 位相推移器 ２ 信号発生器 ３〜６ 信号発生回路 １１〜１４ 入力端子 ３１〜３４，ｑ１〜ｑ４ 出力端子 Ｃ１１〜Ｃ１４，Ｃ２１〜Ｃ２４，Ｃ３１〜Ｃ３４，Ｃ
４１〜Ｃ４４，Ｃｐ１〜Ｃｐ４ コンデンサ Ｒ１１〜Ｒ１４，Ｒ２１〜Ｒ２４，Ｒ３１〜Ｒ３４，Ｒ
４１〜Ｒ４４，Ｒｐ１〜Ｒｐ４ 抵抗器1, 1A Phase shifter 2 Signal generator 3-6 Signal generation circuit 11-14 Input terminal 31-34, q1-q4 output terminal C11-C14, C21-C24, C31-C34, C
41-C44, Cp1-Cp4 capacitors R11-R14, R21-R24, R31-R34, R
41-R44, Rp1-Rp4 resistors
Claims (3)
の入力端子と、前記第１，第２，第３および第４の入力
端子にそれぞれ行対応する第１，第２，第３および第４
の出力端子と、前記各行の入力端子と出力端子との間に
それぞれ縦続接続されてＰ（Ｐは２以上の整数）列をな
すとともに各列ごとには抵抗値をほぼ等しくした抵抗器
と、第Ｊ（Ｊは１ないし４の整数）行・第Ｉ（Ｉは１な
いしＰの整数）列の前記抵抗器の入力端と第（Ｊ−１，
但しＪ−１＝０は４と読み替える）行・第Ｉ列の前記抵
抗器の出力端とを循環的に接続するとともに各列ごとに
は容量値をほぼ等しくしたコンデンサとを備えるポリフ
ェーズ回路型位相推移器において、 第Ｉ列の前記抵抗器の抵抗値Ｒｉと第（Ｉ＋１）列の前
記抵抗器の抵抗値Ｒ（ｉ＋１）との比であるｎ＝Ｒ（ｉ
＋１）／Ｒｉを１．３から４の範囲とすることを特徴と
するポリフェーズ回路型位相推移器。1. A first row, a second row, a third row and a fourth row which form four rows.
Input terminals and the first, second, third and fourth corresponding to the first, second, third and fourth input terminals respectively.
And a resistor having P (P is an integer of 2 or more) columns connected in series between the input terminal and the output terminal of each row and having substantially the same resistance value for each column, The input terminal of the resistor in the Jth row (J is an integer of 1 to 4) and the Ith row (I is an integer of 1 to P) and the (J-1,
However, J-1 = 0 is read as 4) Polyphase circuit type in which the output terminals of the resistors in the row / Ith column are cyclically connected and each column is provided with a capacitor having a substantially equal capacitance value. In the phase shifter, n = R (i, which is the ratio of the resistance value Ri of the resistor in the I-th column and the resistance value R (i + 1) of the resistor in the (I + 1) -th column.
A polyphase circuit type phase shifter characterized in that +1) / Ri is in the range of 1.3 to 4.
列の前記コンデンサの容量値Ｃｉの積と第（Ｉ＋１）列
の前記抵抗器の抵抗値Ｒ（Ｉ＋１）と第（Ｉ＋１）列の
前記コンデンサの容量値Ｃ（ｉ＋１）の積との比Ｒｉ・
Ｃｉ／｛Ｒ（ｉ＋１）・Ｃ（ｉ＋１）｝をｍとした場
合、前記ｎを｛（ｍ＋１）＋（ｍ2 ＋６ｍ＋１）1/2 ｝
／（２ｍ）の０．８倍から２倍までの範囲とすることを
特徴とする請求項１記載のポリフェーズ回路型位相推移
器。2. The resistance value Ri of the resistor in the I-th column and the I-th value
The ratio Ri of the product of the capacitance value Ci of the capacitors in the column and the product of the resistance value R (I + 1) of the resistors in the (I + 1) th column and the capacitance value C (i + 1) of the capacitors in the (I + 1) th column Ri.
When Ci / {R (i + 1) · C (i + 1)} is m, the n is {(m + 1) + (m 2 + 6m + 1) 1/2 }
2. The polyphase circuit type phase shifter according to claim 1, wherein the range is 0.8 times to 2 times of / (2m).
くしたことを特徴とする請求項１記載のポリフェーズ回
路型位相推移器。3. The polyphase circuit type phase shifter according to claim 1, wherein the n and the m are substantially equal in each column.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP19525094A JP2729150B2 (en) | 1994-08-19 | 1994-08-19 | Polyphase circuit type phase shifter |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP19525094A JP2729150B2 (en) | 1994-08-19 | 1994-08-19 | Polyphase circuit type phase shifter |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPH0865083A true JPH0865083A (en) | 1996-03-08 |
| JP2729150B2 JP2729150B2 (en) | 1998-03-18 |
Family
ID=16338008
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP19525094A Expired - Lifetime JP2729150B2 (en) | 1994-08-19 | 1994-08-19 | Polyphase circuit type phase shifter |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JP2729150B2 (en) |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2004524716A (en) * | 2000-09-18 | 2004-08-12 | スカイワークス ソリューションズ，インコーポレイテッド | 8 phase 45 ° polyphase filter system with amplitude matching |
| WO2006054364A1 (en) * | 2004-11-19 | 2006-05-26 | Hitachi Communication Technologies, Ltd. | Method of designing passive rc complex filter of hartley radio receiver |
-
1994
- 1994-08-19 JP JP19525094A patent/JP2729150B2/en not_active Expired - Lifetime
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2004524716A (en) * | 2000-09-18 | 2004-08-12 | スカイワークス ソリューションズ，インコーポレイテッド | 8 phase 45 ° polyphase filter system with amplitude matching |
| JP4740408B2 (en) * | 2000-09-18 | 2011-08-03 | スカイワークス ソリューションズ，インコーポレイテッド | 8-phase 45 ° polyphase filter system with amplitude matching |
| WO2006054364A1 (en) * | 2004-11-19 | 2006-05-26 | Hitachi Communication Technologies, Ltd. | Method of designing passive rc complex filter of hartley radio receiver |
| US7941476B2 (en) | 2004-11-19 | 2011-05-10 | Hitachi, Ltd. | Method of designing passive RC complex filter of Hartley radio receiver |
Also Published As
| Publication number | Publication date |
|---|---|
| JP2729150B2 (en) | 1998-03-18 |
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