JPH0637831A - Linear transmission circuit - Google Patents
Linear transmission circuitInfo
- Publication number
- JPH0637831A JPH0637831A JP19117092A JP19117092A JPH0637831A JP H0637831 A JPH0637831 A JP H0637831A JP 19117092 A JP19117092 A JP 19117092A JP 19117092 A JP19117092 A JP 19117092A JP H0637831 A JPH0637831 A JP H0637831A
- Authority
- JP
- Japan
- Prior art keywords
- phase
- signal
- phase difference
- burst signal
- phase shift
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
Abstract
Description
【0001】[0001]
【産業上の利用分野】本発明は線形送信回路に係り、特
に高出力電力増幅器の非線形歪補償回路を備えるディジ
タル無線方式における線形送信回路に関する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a linear transmission circuit, and more particularly to a linear transmission circuit in a digital radio system including a non-linear distortion compensation circuit for a high output power amplifier.
【0002】移動通信などのディジタル無線通信の分野
では、特に周波数の有効利用の観点より線形変調方式が
用いられる。一方、自動車電話、携帯電話等移動通信端
末は、低消費電力化への要求が益々強くなってきてお
り、そのため移動通信端末で最も消費電力の大きな電力
増幅器の出力効率を向上させることが重要となる。In the field of digital radio communication such as mobile communication, a linear modulation system is used particularly from the viewpoint of effective use of frequencies. On the other hand, mobile communication terminals such as car phones and mobile phones are increasingly required to reduce power consumption. Therefore, it is important to improve the output efficiency of the power amplifier, which consumes the most power in mobile communication terminals. Become.
【0003】しかし、電力増幅器を高効率増幅点で使用
すると非線形歪が発生するので、前記線形変調方式を利
用してディジタル無線送信するためには、非線形歪補償
回路を備える線形送信回路が必要とされる。However, since nonlinear distortion occurs when the power amplifier is used at a high efficiency amplification point, a linear transmission circuit having a nonlinear distortion compensation circuit is required for digital radio transmission using the linear modulation method. To be done.
【0004】[0004]
【従来の技術】図15は従来の線形送信回路の一例のブ
ロック図を示す。同図中、送信すべきディジタル入力信
号は信号処理部1に入力されて、ベースバンドのアナロ
グ信号である2つの直交信号(Q信号)と同相信号(I
信号)とに変換された後、歪補償回路2に供給される。
歪補償回路2は直交復調器7により、送信信号を直交復
調して得たQ’信号とI’信号と、信号処理回路1より
のQ信号とI信号とから、非線形歪みの振幅と位相の両
方をベースバンドのベクトル平面で補償した、Q信号と
I信号とを生成し、直交変調器3へ入力する。2. Description of the Related Art FIG. 15 shows a block diagram of an example of a conventional linear transmission circuit. In the figure, a digital input signal to be transmitted is input to the signal processing unit 1, and two quadrature signals (Q signals) which are baseband analog signals and an in-phase signal (I
Signal) and then supplied to the distortion compensation circuit 2.
The distortion compensating circuit 2 uses the quadrature demodulator 7 to quadrature demodulate the transmission signal and obtains the amplitude and phase of the nonlinear distortion from the Q ′ signal and the I ′ signal obtained by the signal processing circuit 1 and the Q signal and the I signal. A Q signal and an I signal, both of which are compensated by the vector plane of the baseband, are generated and input to the quadrature modulator 3.
【0005】直交変調器3は搬送波発振器4よりの搬送
波でQ信号とI信号を直交変調してQPSK変調波を生
成し、そのQPSK変調波を高出力増幅器5に供給す
る。高出力増幅器5の高効率増幅点で増幅されたQPS
K変調波は方向性結合器6で2分岐され、一方は送信さ
れ、他方は直交復調器7に供給され、ここで搬送波発振
器4より移相器8を通して入力される復調用搬送波に基
づいて直交復調される。ここで、上記の歪補償回路21
はカルテシアン型歪補償方式による歪補償を行なう回路
で、例えば図16に示す如き回路構成とされている。同
図中、信号処理回路1よりのI信号とQ信号が、直交復
調器7よりのI’信号,Q’信号と減算器11,12で
減算されて夫々の歪成分を低減した後、増幅器13,1
4で増幅して出力する。この歪補償回路は負帰還により
歪みを低減する。The quadrature modulator 3 quadrature modulates the Q signal and the I signal with the carrier wave from the carrier wave oscillator 4 to generate a QPSK modulated wave, and supplies the QPSK modulated wave to the high output amplifier 5. QPS amplified at high efficiency amplification point of high-power amplifier 5
The K-modulated wave is branched into two by the directional coupler 6, one of which is transmitted and the other is supplied to the quadrature demodulator 7, where the quadrature is quadrature based on the demodulation carrier input from the carrier oscillator 4 through the phase shifter 8. Demodulated. Here, the above distortion compensation circuit 21
Is a circuit that performs distortion compensation by the Cartesian distortion compensation method, and has a circuit configuration as shown in FIG. 16, for example. In the figure, the I signal and the Q signal from the signal processing circuit 1 are subtracted by the I ′ signal and the Q ′ signal from the quadrature demodulator 7 and the subtracters 11 and 12 to reduce the respective distortion components, and then the amplifier 13, 1
Amplified by 4 and output. This distortion compensation circuit reduces distortion by negative feedback.
【0006】また、歪補償回路2の他の例として図17
に示す如き回路構成が知られている。同図中、減算器1
6,17でI信号とQ信号がI’信号とQ’信号と減算
されて歪み成分が取り出され、増幅器18,19及びス
イッチSW1 ,SW2 を通して加算器20,21に入力
され、ここで、I信号とQ信号と加算される。これによ
り、加算器20,21からはI−I’とQ−Q’(歪成
分)が増幅器18,19のゲイン分の一に圧縮されたI
信号、Q信号が取り出される。FIG. 17 shows another example of the distortion compensation circuit 2.
A circuit configuration as shown in is known. Subtractor 1 in FIG.
In 6 and 17, the I signal and the Q signal are subtracted from the I ′ signal and the Q ′ signal to extract the distortion component, which are input to the adders 20 and 21 through the amplifiers 18 and 19 and the switches SW 1 and SW 2 , respectively. , I signal and Q signal are added. As a result, I-I 'and Q-Q' (distortion components) from the adders 20 and 21 are compressed to the gain of the amplifiers 18 and 19 divided by I.
The signal and the Q signal are taken out.
【0007】[0007]
【発明が解決しようとする課題】しかるに、上記の従来
回路では、種々の要因(例えば、温度、電力レベル、周
波数、経時変化など)に起因して直交復調器7の出力復
調位相が信号処理部1からの送信信号位相と異なると、
ベースバンド信号(I信号、Q信号)が正しく出力され
ず、歪補償が正しくできない。However, in the above-mentioned conventional circuit, the output demodulation phase of the quadrature demodulator 7 is caused by various factors (for example, temperature, power level, frequency, aging, etc.). If it is different from the transmission signal phase from 1,
The baseband signal (I signal, Q signal) is not output correctly, and distortion compensation cannot be performed correctly.
【0008】本発明は上記の点に鑑みなされたもので、
バースト信号の立ち上がり時間中に位相差を測定し、そ
の測定結果に基づいて復調位相を調整することにより、
上記の課題を解決した線形送信回路を提供することを目
的とする。The present invention has been made in view of the above points,
By measuring the phase difference during the rise time of the burst signal and adjusting the demodulation phase based on the measurement result,
It is an object of the present invention to provide a linear transmission circuit that solves the above problems.
【0009】[0009]
【課題を解決するための手段】図1は本発明の原理構成
図を示す。本発明は、送信ベースバンド信号を歪補償回
路31を通して直交変調器32へ入力し、直交変調器3
2の出力直交変調波を電力増幅器33で増幅した後、時
分割多重方式のバースト信号として出力すると共に直交
復調器34で復調し、その復調信号を前記歪補償回路3
1に帰還入力してベースバンドのベクトル座標で歪みを
補償する線形送信回路において、位相差測定手段35と
移相手段36を有する。FIG. 1 is a block diagram showing the principle of the present invention. According to the present invention, the transmission baseband signal is input to the quadrature modulator 32 through the distortion compensation circuit 31, and the quadrature modulator 3
After the output quadrature modulated wave of No. 2 is amplified by the power amplifier 33, it is output as a burst signal of the time division multiplexing system and demodulated by the quadrature demodulator 34, and the demodulated signal is the distortion compensation circuit 3
The linear transmission circuit which feeds back to 1 and compensates the distortion by the vector coordinates of the baseband has a phase difference measuring means 35 and a phase shift means 36.
【0010】上記位相差測定手段35は前記バースト信
号の立ち上がり時間中に歪補償回路31に入力されるベ
ースバンド信号の位相と前記復調信号の位相との位相差
を測定する。移相手段36は位相差測定手段35による
位相差測定後に該測定位相差をゼロにするように、直交
変調器34の復調搬送波と直交変調器32の変調搬送波
との相対位相を移相調整する。The phase difference measuring means 35 measures the phase difference between the phase of the baseband signal input to the distortion compensation circuit 31 and the phase of the demodulated signal during the rising time of the burst signal. The phase shift means 36 performs phase shift adjustment of the relative phase between the demodulation carrier wave of the quadrature modulator 34 and the modulation carrier wave of the quadrature modulator 32 so that the measured phase difference becomes zero after the phase difference measurement means 35 measures the phase difference. .
【0011】[0011]
【作用】本発明では、時分割多重(TDMA)方式のバ
ースト信号を送信するが、バースト信号の立ち上がりの
時間(ランプタイム)はスペクトラムの拡がりを抑える
ために滑らかに立ち上がり、また電力増幅器33より出
力される電力は小さいので、電力増幅器33の非線形歪
みは小さい。しかも、このバースト信号のランプタイム
は伝送データに依存せず、一定の符号が送出されるの
で、このランプタイムは非線形歪補償を必要としない。In the present invention, a burst signal of the time division multiplexing (TDMA) system is transmitted, but the rising time (ramp time) of the burst signal rises smoothly to suppress the spread of the spectrum, and is output from the power amplifier 33. Since the generated power is small, the nonlinear distortion of the power amplifier 33 is small. Moreover, since the ramp time of this burst signal does not depend on the transmission data and a constant code is transmitted, this ramp time does not require nonlinear distortion compensation.
【0012】そこで、本発明はバースト信号のランプタ
イム中に歪補償回路31の動作を実質的に停止させて位
相差測定手段35により位相差の測定を行なうことによ
り、位相差を最も位相測定誤差が少ない状態で測定する
ことができる。そして、この位相測定誤差の少ない位相
差測定結果に基づいて、復調搬送波と変調搬送波の相対
位相を上記位相差がゼロとなるように変化させ、その後
に歪補償の制御を開始する。Therefore, according to the present invention, the operation of the distortion compensating circuit 31 is substantially stopped during the ramp time of the burst signal, and the phase difference is measured by the phase difference measuring means 35. Can be measured in a state where there is little. Then, based on the phase difference measurement result with less phase measurement error, the relative phase between the demodulated carrier wave and the modulated carrier wave is changed so that the phase difference becomes zero, and then the distortion compensation control is started.
【0013】[0013]
【実施例】本実施例の線形送信回路は、ディジタル方式
自動車電話システムの移動局に組込まれ、基地局に対し
て情報を送信するので、まずディジタル方式自動車電話
システムの概要について説明する。この自動車電話シス
テムでは移動局は3チャネル多重TDMA方式を使用
し、例えば図2に示す如きスロット配置とされている。
同図中、(A)は基地局から移動局へ送信される信号の
スロット配置を示し、第3チャネルの移動局への送信ス
ロットTB-M3,第1チャネルの移動局への送信スロット
TB-M1,第2チャネルの移動局への送信スロットTB-M2
が順次時系列的に合成されてなる。DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The linear transmission circuit of this embodiment is incorporated in a mobile station of a digital car telephone system and transmits information to a base station. Therefore, the outline of the digital car telephone system will be described first. In this car telephone system, the mobile station uses the three-channel multiplex TDMA system and has the slot arrangement as shown in FIG. 2, for example.
In the figure, (A) shows the slot arrangement of the signal transmitted from the base station to the mobile station. The transmission slot T B-M3 to the mobile station of the third channel and the transmission slot T to the mobile station of the first channel are shown. B-M1 , transmission slot T B-M2 to mobile station of second channel
Are sequentially combined in time series.
【0014】また、図2(B)は移動局の送信信号を受
信する基地局の受信信号のスロット配置を示し、第1,
第2及び第3チャネルの順で各移動局からの送信スロッ
トが時系列的に合成されている。図2(C)は第1チャ
ネルの移動局M1のスロット配置を示し、送信スロット
TM1-B,アンテナ切替ダイバーシチ用レベル測定LM,
基地局からの送信信号の受信スロットRB-M1,アイドル
Iの各スロットが時系列的に合成されている。FIG. 2 (B) shows the slot arrangement of the received signal of the base station which receives the transmitted signal of the mobile station.
The transmission slots from each mobile station are combined in time series in the order of the second and third channels. FIG. 2C shows the slot arrangement of the mobile station M1 of the first channel, including the transmission slot T M1-B , the antenna switching diversity level measurement LM,
The reception slot RB -M1 of the transmission signal from the base station and each slot of the idle I are combined in time series.
【0015】同様にして、第2チャネルの移動局M2の
スロット配置は図2(D)に、また第3チャネルの移動
局M3のスロット配置は図2(E)に夫々示す如くにな
り、同図(A)に示す基地局からの送信スロットに対応
して受信スロットが配置されている。これにより、送受
信で二つのキャリアで3つの移動局とのTDMA通信が
可能になる。Similarly, the slot arrangement of the mobile station M2 of the second channel is as shown in FIG. 2 (D), and the slot arrangement of the mobile station M3 of the third channel is as shown in FIG. 2 (E). Reception slots are arranged corresponding to the transmission slots from the base station shown in FIG. This enables TDMA communication with three mobile stations using two carriers for transmission and reception.
【0016】各移動局は自己に割当てられたタイムスロ
ットで基地局との間で、無線通信を行なう。各スロット
毎に独立した物理チャネルとして制御チャネルと通信チ
ャネルがある。移動局から基地局へ送信される信号の制
御チャネルは図3(A)に示す如き信号フォーマットと
され、通信チャネルは同図(B)に示す如き信号フォー
マットとされている。Each mobile station performs radio communication with the base station in a time slot assigned to itself. There are a control channel and a communication channel as an independent physical channel for each slot. The control channel of the signal transmitted from the mobile station to the base station has a signal format as shown in FIG. 3 (A), and the communication channel has a signal format as shown in FIG. 3 (B).
【0017】同図(A),(B)中、Rは4ビットのバ
ースト過渡応答用ガード時間、Pは2ビットのプリアン
ブル、CACは制御信号、SWは20ビットの同期ワー
ド、CCは8ビットのカラーコード、Gはガード時間、
TCHはユーザ情報転送用チャネル、SFは1ビットの
スチールフラグ、SACCHは低速付随制御チャネルで
ある。In FIGS. 1A and 1B, R is a 4-bit burst transient response guard time, P is a 2-bit preamble, CAC is a control signal, SW is a 20-bit synchronization word, and CC is 8-bit. Color code, G is guard time,
TCH is a user information transfer channel, SF is a 1-bit steal flag, and SACCH is a low speed associated control channel.
【0018】そして、送信回路からは図3(C)に示す
如く、上記のバースト過渡応答用ガード時間Rの間に立
ち上がり、ガード時間Gで立ち下がり、かつ、その立ち
上がりと立ち下がりの間は例えば搬送波を情報でπ/4
シフトQPSK変調した変調波が存在するバースト信号
を自己に割当てられたタイムスロット毎に送信する。Then, as shown in FIG. 3C, the transmitter circuit rises during the burst transient response guard time R and falls during the guard time G, and between the rise and the fall, for example. Carrier wave with information π / 4
A burst signal in which a modulated wave subjected to shift QPSK modulation is present is transmitted for each time slot assigned to itself.
【0019】次に上記のディジタル方式自動車電話シス
テムの移動局に組込まれ、図3(C)に示すTDMA方
式のバースト信号を送信する送信回路の各実施例につい
て説明する。図4は本発明になる線形送信回路の第1実
施例のブロック図を示す。同図中、図1と同一構成部分
には同一符号を付してある。Next, each embodiment of the transmission circuit incorporated in the mobile station of the above-mentioned digital type automobile telephone system and transmitting the burst signal of the TDMA type shown in FIG. 3C will be described. FIG. 4 shows a block diagram of a first embodiment of a linear transmission circuit according to the present invention. In the figure, the same components as those in FIG. 1 are designated by the same reference numerals.
【0020】図4において、送信する情報データは信号
処理部30に供給され、ここでベースバンドのアナログ
信号であるI信号とQ信号に変換された後、図17に示
した構成の歪補償回路31を通して直交変調器32及び
移相制御器42に夫々入力される。直交変調器32は搬
送波発振器40よりの変調搬送波を上記2つのI信号及
びQ信号で例えばπ/4シフトQPSK変調する。In FIG. 4, the information data to be transmitted is supplied to the signal processing unit 30, where it is converted into I and Q signals which are baseband analog signals, and then the distortion compensating circuit having the configuration shown in FIG. It is inputted to the quadrature modulator 32 and the phase shift controller 42 through 31 respectively. The quadrature modulator 32 subjects the modulated carrier wave from the carrier wave oscillator 40 to, for example, π / 4 shift QPSK modulation with the above two I signals and Q signals.
【0021】このπ/4シフトQPSK変調波は電力増
幅器33で電力増幅された後、方向性結合器41で2分
岐され、一方は基地局へバースト信号として送信される
一方、他方は直交復調器34に入力され、ここで移相器
回転手段43を通して入力される搬送波発振器40より
の復調搬送波を用いて直交復調されて復調I信号及び復
調Q信号とされる。この復調I信号及び復調Q信号は移
相制御器42に入力される一方、歪補償回路31に入力
される。This π / 4 shift QPSK modulated wave is power-amplified by a power amplifier 33 and then branched into two by a directional coupler 41, one of which is transmitted to a base station as a burst signal and the other of which is a quadrature demodulator. A demodulation carrier signal 40 is input to the signal generator 34 and is orthogonally demodulated using the demodulation carrier wave from the carrier wave oscillator 40 input through the phase shifter rotating means 43 to obtain a demodulation I signal and a demodulation Q signal. The demodulated I signal and the demodulated Q signal are input to the phase shift controller 42, while being input to the distortion compensation circuit 31.
【0022】本実施例は移相制御器42と移相器回転手
段43を有する点に特徴がある。移相制御器42は前記
位相差測定手段35に相当し、例えば図5に示す如き構
成とされている。同図中、AD変換器(ADC)421 及
び422 には直交変調器32に入力されるI信号MDI及
びQ信号MDQが夫々入力されてディジタル信号に変換
される。また、ADC423 及び424 には直交復調器34
より出力される復調I信号DEI及び復調Q信号DEQ
が夫々入力されてディジタル信号に変換される。The present embodiment is characterized in having a phase shift controller 42 and a phase shifter rotating means 43. The phase shift controller 42 corresponds to the phase difference measuring means 35 and has a structure as shown in FIG. 5, for example. In the figure, the AD converters (ADCs) 421 and 422 respectively receive the I signal MDI and the Q signal MDQ input to the quadrature modulator 32 and are converted into digital signals. In addition, the quadrature demodulator 34 is provided in the ADCs 423 and 424.
Demodulated I signal DEI and demodulated Q signal DEQ output from
Are input and converted into digital signals.
【0023】ADC421 及び422 の各出力信号はtan -1
MDQ/MDIの演算を425で行い、送信位相が検出
される。他方、ADC423 及び424 の各出力信号はtan
-1DEQ/DEIの演算を426で行い、復調位相が検
出される。回路425,426はROM等を用いれば簡
単に構成出来る。The output signals of the ADCs 421 and 422 are tan -1.
The MDQ / MDI calculation is performed at 425, and the transmission phase is detected. On the other hand, the output signals of ADC423 and 424 are tan.
-1 DEQ / DEI calculation is performed at 426, and the demodulation phase is detected. The circuits 425 and 426 can be easily constructed by using a ROM or the like.
【0024】演算部429 は上記の送信位相と復調位相と
の差を位相差として測定し、その位相差が零となるよう
な制御信号を生成して図4の移相器回転手段43へ供給
する。なお、演算器429 はTDMA制御部(図示せず)
よりの制御信号に同期してタイミング信号を発生するタ
イミング発生部420 よりのタイミング信号に基づいて位
相差測定の演算を行なう。The arithmetic unit 429 measures the difference between the transmission phase and the demodulation phase as a phase difference, generates a control signal that makes the phase difference zero, and supplies it to the phase shifter rotating means 43 in FIG. To do. The arithmetic unit 429 is a TDMA control unit (not shown).
Phase difference measurement is performed based on the timing signal from the timing generation unit 420 that generates the timing signal in synchronization with the control signal.
【0025】移相器回転手段43はEPS(エンドレス
・フェーズ・シフタ)回路と称される回路で、前記移相
手段36の一例を示しており、例えば図6に示す如き構
成とされている。同図中、端子430 は移相制御器42の
出力制御信号の入力端子、端子431 は搬送波発振器40
よりの搬送波入力端子である。入力端子430 よりの制御
信号はcos θ演算器432 及びsin θ演算器433 に夫々入
力されてcos θ,sinθの演算結果を出力させる。The phase shifter rotating means 43 is a circuit called an EPS (endless phase shifter) circuit, and shows an example of the phase shifting means 36, and has a structure as shown in FIG. 6, for example. In the figure, a terminal 430 is an input terminal for the output control signal of the phase shift controller 42, and a terminal 431 is a carrier wave oscillator 40.
This is a carrier wave input terminal. The control signal from the input terminal 430 is input to the cos θ calculator 432 and the sin θ calculator 433, respectively, and outputs the calculation results of cos θ and sin θ.
【0026】cos θ,sin θの各演算結果は夫々DA変
換器434 ,435 でアナログ信号に変換された後、乗算器
436 ,437 に供給され、ここで90°移相器438 よりの
互いに位相が90°異なるが周波数は同一の搬送波と乗
算される。乗算器436 ,437の各出力信号は加算器439
に供給されて加算され、前記入力端子430 よりの制御信
号の値に応じた移相量の復調搬送波とされて図4の直交
復調器34へ出力される。The calculation results of cos θ and sin θ are converted into analog signals by DA converters 434 and 435, respectively, and then multiplied by multipliers.
436, 437, where the frequencies are multiplied by the same carrier by a 90 ° phase shifter 438, but 90 ° out of phase with each other. The output signals of the multipliers 436 and 437 are added by the adder 439.
Is added to the quadrature demodulator 34 and is added to the quadrature demodulator 34 of FIG. 4 to obtain a demodulated carrier having a phase shift amount corresponding to the value of the control signal from the input terminal 430.
【0027】上記の構成の第1実施例において、移相制
御器42での位相差測定は図5に示すタイミング発生部
420 よりのタイミング信号が、図7に示す電力増幅器
(又は方向性結合器)の出力バースト信号の立ち上がり
時間(ランプタイム)中に発生出力されるために、ラン
プタイム中に行なわれることとなる。In the first embodiment having the above structure, the phase difference measurement by the phase shift controller 42 is performed by the timing generator shown in FIG.
Since the timing signal from 420 is generated and output during the rising time (ramp time) of the output burst signal of the power amplifier (or directional coupler) shown in FIG. 7, it is performed during the ramp time.
【0028】図7からわかるように、バースト信号のラ
ンプタイムは電力増幅器33の出力電力が小さいために
非線形歪が少なく、またランプタイム中のデータは送信
データに関係のない一定の符号であるため、このランプ
タイム中の位相差測定によりかなり正確な位相差測定が
できる。As can be seen from FIG. 7, the ramp time of the burst signal has a small non-linear distortion because the output power of the power amplifier 33 is small, and the data during the ramp time is a constant code irrelevant to the transmission data. By the phase difference measurement during this ramp time, a fairly accurate phase difference measurement can be performed.
【0029】ただし、位相差が大きいとフィードバック
系は不安定となり発振を生じることもあり得るので、図
4の信号処理部30の出力信号により歪補償回路31内
のスイッチ(図17のSW1 ,SW2 )を図7に示す如
くランプタイム中はオフとし、フィードバック系をオー
プンとして帰還を遮断しておく。ランプタイム経過後は
図7に示す如く歪補償回路31内のスイッチSW1 及び
SW2 をオンとし、フィードバックループを形成して歪
補償回路31によりベースバンドのベクトル座標で歪み
を補償するカルテシアン型歪補償を行なう。このカルテ
シアン型歪補償は測定位相差を零とするように移相器回
転手段43により変調移相量が制御され、常に送信位相
と復調位相とが合わせられているため、歪補償の制御を
安定に行なえる。However, if the phase difference is large, the feedback system may become unstable and oscillation may occur. Therefore, the output signal of the signal processing unit 30 in FIG. 4 causes a switch in the distortion compensating circuit 31 (SW 1 in FIG. 17, SW 2 ) is turned off during the ramp time as shown in FIG. 7, and the feedback system is opened to interrupt the feedback. After the ramp time has elapsed, as shown in FIG. 7, the switches SW 1 and SW 2 in the distortion compensating circuit 31 are turned on to form a feedback loop and the distortion compensating circuit 31 compensates the distortion at the vector coordinates of the base band. Perform distortion compensation. In this Cartesian distortion compensation, the modulation phase shift amount is controlled by the phase shifter rotating means 43 so that the measured phase difference becomes zero, and the transmission phase and the demodulation phase are always matched, so that the distortion compensation is controlled. You can do it stably.
【0030】図8は本発明になる線形送信回路の第2実
施例のブロック図を示す。同図中、図4と同一構成部分
には同一符号を付し、その説明を省略する。本実施例は
図8に示すように、搬送波発振器40の出力搬送波を復
調搬送波として直交復調器34に供給する一方、移相器
回転手段46を通して変調搬送波として直交変調器32
に供給するようにした点が図4の第1実施例と異なる。
位相差測定タイミングは第1実施例と同じである。FIG. 8 shows a block diagram of a second embodiment of the linear transmission circuit according to the present invention. In the figure, parts that are the same as the parts shown in FIG. 4 are given the same reference numerals, and descriptions thereof will be omitted. In this embodiment, as shown in FIG. 8, the output carrier of the carrier oscillator 40 is supplied to the quadrature demodulator 34 as a demodulation carrier, while the quadrature modulator 32 is supplied as a modulation carrier through the phase shifter rotating means 46.
Is different from the first embodiment in FIG.
The phase difference measurement timing is the same as in the first embodiment.
【0031】移相器回転手段46は搬送波発振器40と
共に前記移相手段36を構成しており、図6と同じ構成
とされている。本実施例では位相差測定結果に応じて位
相差が零となるよう、変調搬送波の方を移相するように
したものであり、第1実施例と同じ効果が得られる。The phase shifter rotating means 46 constitutes the phase shifting means 36 together with the carrier wave oscillator 40, and has the same configuration as in FIG. In this embodiment, the modulated carrier wave is phase-shifted so that the phase difference becomes zero according to the phase difference measurement result, and the same effect as that of the first embodiment can be obtained.
【0032】次に本発明の第3実施例について説明す
る。本実施例はブロック構成は図4と図8のどちらでも
よく、歪補償回路31を図9の回路構成とすると共に、
歪補償回路の動作と移相制御器42による位相差測定を
図10に示す如きタイミングで行なうようにしたもので
ある。Next, a third embodiment of the present invention will be described. In this embodiment, the block configuration may be either of FIG. 4 or FIG. 8, and the distortion compensating circuit 31 has the circuit configuration of FIG.
The operation of the distortion compensation circuit and the phase difference measurement by the phase shift controller 42 are performed at the timing shown in FIG.
【0033】図9に示す歪補償回路31は直交変調器3
2の入力I信号MDI,Q信号MDQを直交復調器34
の出力復調I信号DEI及び復調Q信号DEQとの誤差
信号を減算器51及び52で夫々生成し、その誤差信号
を可変利得増幅器53,54を通して加算器55,56
に入力し、入力信号MDI,MDQと加算する構成であ
る。The distortion compensation circuit 31 shown in FIG.
The quadrature demodulator 34 receives the two input I signals MDI and Q signals MDQ.
Of the output demodulated I signal DEI and the demodulated Q signal DEQ are generated by subtractors 51 and 52, respectively, and the error signals are added through variable gain amplifiers 53 and 54 to adders 55 and 56.
To the input signals MDI and MDQ.
【0034】ここで、可変利得増幅器53及び54の利
得制御は信号処理部30の出力制御信号により、図10
に示す如く、電力増幅器33の出力バースト信号の立ち
上がり時間(ランプタイム)中に移相制御器42による
位相差測定が行なわれているときは利得最小(従って歪
補償のフィードバック量最小)とし、位相差測定後のラ
ンプタイム期間は利得を徐々に大に可変し、ランプタイ
ム後は利得を所定値にセットするように行なわれる。Here, the gain control of the variable gain amplifiers 53 and 54 is performed by the output control signal of the signal processing unit 30 as shown in FIG.
As shown in, when the phase difference is measured by the phase shift controller 42 during the rise time (ramp time) of the output burst signal of the power amplifier 33, the gain is set to the minimum (therefore, the feedback amount for distortion compensation is set to the minimum), and During the ramp time period after the phase difference measurement, the gain is gradually changed to a large value, and after the ramp time, the gain is set to a predetermined value.
【0035】これにより、本実施例によれば、位相差測
定が終了した後は歪補償のフィードバック量が徐々に大
きくされることとなり、歪補償開始時に過渡現象が生じ
て信号が歪むことを防止することができる。As a result, according to the present embodiment, the feedback amount for distortion compensation is gradually increased after the phase difference measurement is completed, and a transient phenomenon is prevented at the beginning of distortion compensation to prevent the signal from being distorted. can do.
【0036】次に本発明の第4実施例について説明す
る。本実施例は図4に示した線形送信回路において、位
相差の測定と歪補償制御を図11に示す如きタイミング
で行なうようにしたものである。なお、本実施例の歪補
償回路31は図9及び図17のどちらの構成でもよい。Next, a fourth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4, the measurement of the phase difference and the distortion compensation control are performed at the timings shown in FIG. The distortion compensation circuit 31 of this embodiment may have either of the configurations shown in FIG. 9 and FIG.
【0037】すなわち、本実施例は図11に示す如く送
信開始より第1バースト信号B1 ,第2バースト信号B
2 ,第3バースト信号B3 ,第4バースト信号B4 ,…
というように送信するに際し、移相制御器42による位
相差測定は各バースト信号のランプタイム中に行なうと
共に、その測定位相差を零とするような移相量を移相器
回転手段43に指示する制御信号の送出タイミングを図
11に示すように1バースト後の移相量調整に用いるこ
とができるタイミングとする。That is, in this embodiment, as shown in FIG. 11, the first burst signal B 1 and the second burst signal B 1 are transmitted from the start of transmission.
2 , third burst signal B 3 , fourth burst signal B 4 , ...
In transmitting, the phase difference is measured by the phase shift controller 42 during the ramp time of each burst signal, and the phase shift amount is set to the phase shifter rotating means 43 so that the measured phase difference becomes zero. The transmission timing of the control signal is set to a timing that can be used for adjusting the amount of phase shift after one burst as shown in FIG.
【0038】従って、歪補償回路31の歪補償動作は図
11の「歪補償制御」の欄に示すように、第1バースト
信号B1 については行なわず、第2バースト信号B2 以
降の各バースト信号について行なわれる。Therefore, the distortion compensating operation of the distortion compensating circuit 31 is not performed for the first burst signal B 1 as shown in the “Distortion Compensation Control” column of FIG. 11, but for each burst after the second burst signal B 2. Performed on signals.
【0039】次に本発明の第5実施例について説明す
る。本実施例は図8に示した線形送信回路において、位
相差の測定と歪補償制御とを図11に示す如きタイミン
グで行なうようにしたものである。Next, a fifth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 8, the measurement of the phase difference and the distortion compensation control are performed at the timings shown in FIG.
【0040】次に本発明の第6実施例について説明す
る。本実施例は図4又は図8に示した線形送信回路にお
いて、位相差測定と移相量設定及び歪補償制御を図12
に模式的に示すタイミングで行なうようにしたものであ
る。なお、歪補償回路31は図9及び図17に示した回
路のいずれでもよい。Next, a sixth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4 or 8, phase difference measurement, phase shift amount setting, and distortion compensation control are performed in FIG.
The timing is schematically shown in FIG. The distortion compensation circuit 31 may be any of the circuits shown in FIGS. 9 and 17.
【0041】本実施例は図12の「位相測定」の欄に模
式的に示す如く、移相制御器42は第1バースト信号B
1 のランプタイムで前記位相差測定を1回行ない、それ
により得られた測定結果に基づく移相量の設定を、同図
に「移相設定」の欄で示すように第1バースト信号B1
のランプタイム終了直前に移相器回転手段43に対して
行なう。この移相設定により歪補償回路31は送信位相
と復調位相とが一致されたI信号MDI及びDEI,Q
信号MDQ及びDEQが入力されるから第1バースト信
号B1 より歪み補償の制御を行なうようにするが、第
4,第5実施例のように第1バースト信号B1 について
は歪み補償の制御は行なわなくてもよい。ただし、第2
バースト信号B2 に対しては図12に「歪補償制御」の
欄で示す如く歪補償を第1バースト信号B1 の位相差測
定結果に基づいて行なう。In the present embodiment, as schematically shown in the column "Phase measurement" in FIG. 12, the phase shift controller 42 sets the first burst signal B
Performed once the phase difference measured in the first ramp time, it phase shift setting based on the measurement results obtained by the first burst signal, as shown in the column of "phase shift setting" in Fig. B 1
Immediately before the end of the ramp time, the phase shifter rotating means 43 is performed. With this phase shift setting, the distortion compensating circuit 31 causes the I signal MDI and DEI, Q whose transmission phase and demodulation phase are matched.
Since the signals MDQ and DEQ are input, the distortion compensation control is performed from the first burst signal B 1, but the distortion compensation control is not performed on the first burst signal B 1 as in the fourth and fifth embodiments. You don't have to do it. However, the second
Performs distortion compensation as shown in the section of "distortion compensation control" in FIG. 12 on the basis of the first burst signal B 1 of the phase difference measurements at burst signal B 2.
【0042】続いて、移相制御器42は第2バースト信
号B2 以降の各バースト信号が入力される毎に、ランプ
タイムであるか否かに関係なく、複数回ずつ位相差測定
を行ない、その複数回の測定値の平均値を算出して次の
バースト信号の歪補償を行なうべく、次のバースト信号
の入力直前に上記平均値に基づく移相量の設定を移相器
回転手段43に対して行なう。複数回の位相差測定は図
5のタイミング発生部420 の出力タイミング信号周波
数、ADC421 〜424 のクロック周波数等を高くするこ
とにより行なえる。Subsequently, the phase shift controller 42 performs the phase difference measurement a plurality of times each time each burst signal after the second burst signal B 2 is input, regardless of whether it is the ramp time or not. Immediately before the input of the next burst signal, the phase shift amount setting based on the average value is set to the phase shifter rotating means 43 in order to calculate the average value of the measured values of the plurality of times and perform the distortion compensation of the next burst signal. To do. The phase difference measurement can be performed a plurality of times by increasing the output timing signal frequency of the timing generator 420 and the clock frequencies of the ADCs 421 to 424 shown in FIG.
【0043】従って、本実施例によれば、第3バースト
信号以降のバースト信号送信時には、前回のバースト信
号の複数回の位相差測定結果の平均値に基づく移相量設
定が行なわれ、歪補償回路31による歪補償動作が行な
われるため、瞬間的な位相変動に追従することなく安定
な歪補償制御ができる。Therefore, according to this embodiment, when transmitting the burst signal after the third burst signal, the phase shift amount is set based on the average value of the phase difference measurement results of the previous burst signal a plurality of times, and the distortion compensation is performed. Since the distortion compensation operation is performed by the circuit 31, stable distortion compensation control can be performed without following an instantaneous phase fluctuation.
【0044】次に本発明の第7実施例について説明す
る。本実施例は図4又は図8に示した線形送信回路にお
いて、位相差測定と移相量設定及び歪補償制御を図13
に模式的に示すタイミングで行なうようにしたものであ
る。なお、歪補償回路31は図9及び図17に示した回
路のいずれでもよい。Next, a seventh embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4 or 8, phase difference measurement, phase shift amount setting, and distortion compensation control are performed in FIG.
The timing is schematically shown in FIG. The distortion compensation circuit 31 may be any of the circuits shown in FIGS. 9 and 17.
【0045】本実施例は図13の「位相測定」の欄に模
式的に示す如く、移相制御器42は各バースト信号のラ
ンプタイムで位相差測定が1回行なわれる点は第1実施
例と同様であるが、移相量設定が第1実施例と異なる。In this embodiment, as schematically shown in the column "Phase measurement" in FIG. 13, the phase shift controller 42 performs the phase difference measurement once at the ramp time of each burst signal. However, the phase shift amount setting is different from that of the first embodiment.
【0046】すなわち、本実施例では第1バースト信号
B1 送信時は図13に示すように、移相制御器42が第
1バースト中に位相差測定を行ない、その測定結果に基
づく移相量の設定を第2バースト信号B2 入力前に移相
器回転手段43に対して行なう。また、この第1バース
ト信号B1 送信時は歪補償回路31の動作を停止(オ
フ)させる。That is, in this embodiment, when transmitting the first burst signal B 1 , the phase shift controller 42 measures the phase difference during the first burst as shown in FIG. 13, and the phase shift amount based on the measurement result. Is set to the phase shifter rotating means 43 before inputting the second burst signal B 2 . Further, the operation of the distortion compensation circuit 31 is stopped (turned off) when the first burst signal B 1 is transmitted.
【0047】続いて、第2バースト信号B2 から第n−
1バースト信号Bn-1 送信時は、図13に示すように、
移相制御器42が各バースト信号のランプタイムで1回
ずつ位相差測定を行ない、それら(n−2)個の位相差
測定結果の平均値を算出し、その平均値に基づく移相量
の設定を第nバースト信号Bn 入力前に移相器回転手段
43に対して行なう。Then, from the second burst signal B 2 to the n-th
At the time of transmitting one burst signal B n−1 , as shown in FIG.
The phase shift controller 42 performs the phase difference measurement once at the ramp time of each burst signal, calculates the average value of the (n-2) phase difference measurement results, and calculates the phase shift amount based on the average value. The setting is performed on the phase shifter rotating means 43 before inputting the nth burst signal B n .
【0048】従って、本実施例では、第2バースト信号
B2 から第n−1バースト信号Bn- 1 送信までは第1バ
ースト信号B1 のランプタイムで測定した位相差に基づ
く移相量が設定され、歪補償が行なわれ、また第nバー
スト信号Bn 以降は第2バースト信号B2 から第n−1
バースト信号Bn-1 の各ランプタイムで測定した各位相
差の平均値に基づく移相量が設定され、歪補償が行なわ
れる。Therefore, in this embodiment, the amount of phase shift based on the phase difference measured by the ramp time of the first burst signal B 1 is from the second burst signal B 2 to the transmission of the n−1 th burst signal B n− 1. Is set, distortion compensation is performed, and after the nth burst signal B n, the second burst signal B 2 to the n−1th burst signal B 2 are set.
A phase shift amount based on the average value of the phase differences measured at each ramp time of the burst signal B n-1 is set, and distortion compensation is performed.
【0049】次に本発明の第8実施例について説明す
る。本実施例は図4又は図8に示した線形送信回路にお
いて、電力増幅器33の線形性と図9又は図17に示し
た歪補償回路31の歪補償制御を図14に模式的に示す
如く行なうようにしたものである。Next, an eighth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4 or 8, the linearity of the power amplifier 33 and the distortion compensation control of the distortion compensation circuit 31 shown in FIG. 9 or 17 are performed as schematically shown in FIG. It was done like this.
【0050】すなわち、図14の「歪補償制御」の欄に
示すように第1バースト信号B1 送信時のみ歪補償回路
31の動作を停止(オフ)し、第2バースト信号B2 以
降の送信時は歪補償回路31を動作させる。このため、
本実施例では歪補償を行なっていない第1バースト信号
B1 送信時は図14の「増幅器線形性」の欄に丸を付け
て示したように、電力増幅器33の線形性を良い状態に
し、第2バースト信号B2 以降の各バースト信号送信時
は歪補償動作を行なっているので、図14にXで示すよ
うに電力増幅器33の線形性を悪くし、その分電力増幅
器33の効率を良くする。That is, as shown in the "Distortion compensation control" column of FIG. 14, the operation of the distortion compensation circuit 31 is stopped (turned off) only when the first burst signal B 1 is transmitted, and the second burst signal B 2 and the subsequent transmissions are transmitted. At this time, the distortion compensation circuit 31 is operated. For this reason,
In the present embodiment, when transmitting the first burst signal B 1 for which distortion compensation has not been performed, the linearity of the power amplifier 33 is set to a good state as indicated by a circle in the "Amplifier linearity" column of FIG. Since distortion compensation operation is performed during transmission of each burst signal after the second burst signal B 2, the linearity of the power amplifier 33 is deteriorated as shown by X in FIG. 14, and the efficiency of the power amplifier 33 is improved correspondingly. To do.
【0051】電力増幅器33の線形性を向上する方法と
しては、例えば増幅用トランジスタが電界効果トランジ
スタ(FET)の場合は、 ゲートのバイアスを浅く
する、 ドレインの電圧を高くする、 負荷条件を
変えるなどがある。As a method of improving the linearity of the power amplifier 33, for example, when the amplifying transistor is a field effect transistor (FET), the bias of the gate is made shallow, the voltage of the drain is made high, the load condition is changed, etc. There is.
【0052】[0052]
【発明の効果】上述の如く、請求項1及び2記載の発明
によれば、送信バースト信号のランプタイム中に送信位
相と復調位相との位相差を測定して位相差をゼロとする
ように搬送波の移相量を設定した後、歪補償を開始する
ようにしたので、位相差を最も位相測定誤差の少ない状
態で測定できることから歪補償を正常に動作させること
ができ、これにより歪みの無い線形なバースト信号の送
信ができる。As described above, according to the first and second aspects of the invention, the phase difference between the transmission phase and the demodulation phase is measured during the ramp time of the transmission burst signal so that the phase difference becomes zero. Since the distortion compensation is started after setting the phase shift amount of the carrier wave, it is possible to operate the distortion compensation normally because the phase difference can be measured in the state in which the phase measurement error is the smallest, and thus, there is no distortion. A linear burst signal can be transmitted.
【0053】また請求項3記載の発明によれば、歪補償
制御開始後徐々に制御のゲインを上げていくため、歪補
償制御開始時の過渡現象による信号の歪みを防止するこ
とができる。According to the third aspect of the invention, since the control gain is gradually increased after the distortion compensation control is started, it is possible to prevent signal distortion due to a transient phenomenon at the time of starting the distortion compensation control.
【0054】請求項4及び5記載の発明によれば、1バ
ースト前の位相差測定結果をもとにバースト信号の非送
信期間に移相量の設定を行なうため、安定に歪補償を行
なうことができ、また請求項6及び7記載の発明によれ
ば、位相差の瞬時変動に追従することなく安定に歪補償
ができる。更に請求項8記載の発明によれば、第1バー
スト信号送信時の線形性を向上することができる等の特
長を有するものである。According to the fourth and fifth aspects of the invention, since the phase shift amount is set in the non-transmission period of the burst signal based on the phase difference measurement result of one burst before, the distortion compensation can be stably performed. According to the sixth and seventh aspects of the present invention, the distortion compensation can be stably performed without following the instantaneous fluctuation of the phase difference. Further, according to the invention described in claim 8, there is a feature that the linearity at the time of transmitting the first burst signal can be improved.
【図1】本発明の原理構成図である。FIG. 1 is a principle configuration diagram of the present invention.
【図2】スロット配置の一例を示す図である。FIG. 2 is a diagram showing an example of slot arrangement.
【図3】送信出力の説明図である。FIG. 3 is an explanatory diagram of transmission output.
【図4】本発明の第1実施例のブロック図である。FIG. 4 is a block diagram of a first embodiment of the present invention.
【図5】移相制御器の構成図である。FIG. 5 is a configuration diagram of a phase shift controller.
【図6】移相器回転手段の構成図である。FIG. 6 is a configuration diagram of a phase shifter rotating means.
【図7】本発明の第1,第2実施例のタイミング関係図
である。FIG. 7 is a timing relationship diagram of the first and second embodiments of the present invention.
【図8】本発明の第2実施例のブロック図である。FIG. 8 is a block diagram of a second embodiment of the present invention.
【図9】歪補償回路の一実施例の構成図である。FIG. 9 is a configuration diagram of an embodiment of a distortion compensation circuit.
【図10】本発明の第3実施例のタイミング関係図であ
る。FIG. 10 is a timing relationship diagram of the third embodiment of the present invention.
【図11】本発明の第4,第5実施例のタイミング関係
図である。FIG. 11 is a timing relationship diagram of the fourth and fifth embodiments of the present invention.
【図12】本発明の第6実施例のタイミング関係図であ
る。FIG. 12 is a timing diagram of the sixth embodiment of the present invention.
【図13】本発明の第7実施例のタイミング関係図であ
る。FIG. 13 is a timing diagram of the seventh embodiment of the present invention.
【図14】本発明の第8実施例のタイミング関係図であ
る。FIG. 14 is a timing diagram of an eighth embodiment of the present invention.
【図15】従来回路の一例のブロック図である。FIG. 15 is a block diagram of an example of a conventional circuit.
【図16】歪補償回路の一例の構成図である。FIG. 16 is a configuration diagram of an example of a distortion compensation circuit.
【図17】歪補償回路の他の例の構成図である。FIG. 17 is a configuration diagram of another example of the distortion compensation circuit.
30 信号処理部 31 歪補償回路 32 直交変調器 33 電力増幅器 34 直交復調器 35 位相差測定手段 36 移相手段 40 搬送波発振器 42 移相制御器 43,46 移相器回転手段 51,52 減算器 53,54 可変利得増幅器 55,56 加算器 30 signal processor 31 distortion compensation circuit 32 quadrature modulator 33 power amplifier 34 quadrature demodulator 35 phase difference measuring means 36 phase shift means 40 carrier wave oscillator 42 phase shift controller 43,46 phase shifter rotation means 51,52 subtractor 53 , 54 Variable gain amplifier 55, 56 Adder
───────────────────────────────────────────────────── フロントページの続き (72)発明者 久保 徳郎 神奈川県川崎市中原区上小田中1015番地 富士通株式会社内 (72)発明者 高野 健 神奈川県川崎市中原区上小田中1015番地 富士通株式会社内 ─────────────────────────────────────────────────── ─── Continued Front Page (72) Inventor Tokuro Kubo 1015 Kamiodanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture, Fujitsu Limited (72) Inventor Ken Ken Takano, 1015, Kamedotachu, Nakahara-ku, Kawasaki City, Kanagawa Prefecture, Fujitsu Limited
Claims (8)
1)を通して直交変調器(32)へ入力し、該直交変調
器(32)の出力直交変調波を電力増幅器(33)で増
幅した後、時分割多重方式のバースト信号として出力す
ると共に直交復調器(34)で復調し、その復調信号を
前記歪補償回路(31)に帰還入力してベースバンドの
ベクトル座標で歪みを補償する線形送信回路において、 前記バースト信号の立ち上がり時間中に前記歪補償回路
(31)に入力されるベースバンド信号の位相と前記復
調信号の位相との位相差を測定する位相差測定手段(3
5)と、 該位相差測定手段(35)による位相差測定後に該測定
位相差をゼロにするように、前記直交復調器(34)の
復調搬送波と前記直交変調器(32)の変調搬送波との
相対位相を移相調整する移相手段(36)とを有するこ
とを特徴とする線形送信回路。1. A distortion compensating circuit (3) for transmitting a baseband signal.
1) is input to a quadrature modulator (32), the quadrature modulated wave output from the quadrature modulator (32) is amplified by a power amplifier (33), and then output as a burst signal of a time division multiplexing system and a quadrature demodulator. A linear transmission circuit that demodulates at (34) and feeds back the demodulated signal to the distortion compensation circuit (31) to compensate for distortion at baseband vector coordinates, wherein the distortion compensation circuit is provided during the rising time of the burst signal. Phase difference measuring means (3) for measuring the phase difference between the phase of the baseband signal input to (31) and the phase of the demodulated signal.
5) and a demodulation carrier of the quadrature demodulator (34) and a modulation carrier of the quadrature modulator (32) so that the measured phase difference becomes zero after the phase difference measurement means (35) measures the phase difference. And a phase shift means (36) for performing phase shift adjustment of the relative phase of the linear transmission circuit.
波又は前記変調搬送波を移相することを特徴とする請求
項1記載の線形送信回路。2. The linear transmission circuit according to claim 1, wherein the phase shift means shifts the demodulated carrier wave or the modulated carrier wave.
測定手段(35)による位相差測定中は歪補償のフィー
ドバック量を小とし、該位相差測定終了後に該フィード
バック量を徐々に大とする手段(53,54,30)を
有することを特徴とする請求項2記載の線形送信回路。3. The distortion compensating circuit (31) reduces the feedback amount of distortion compensation during the phase difference measurement by the phase difference measuring means (35), and gradually increases the feedback amount after the phase difference measurement is completed. 3. The linear transmission circuit according to claim 2, further comprising means (53, 54, 30).
最初に出力される第1バースト信号送信中は歪補償動作
を行なわず、第2バースト信号以降歪補償動作を行な
い、前記移相手段(36)は1バースト信号期間前に前
記位相差測定手段(35)により測定された位相差に基
づく移相量が設定されることを特徴とする請求項1記載
の線形送信回路。4. The distortion compensating circuit (31) does not perform the distortion compensating operation during the transmission of the first burst signal that is first output after the start of transmission, but performs the distortion compensating operation after the second burst signal, and the phase shift is performed. The linear transmission circuit according to claim 1, wherein the means (36) sets a phase shift amount based on the phase difference measured by the phase difference measuring means (35) one burst signal period before.
波又は前記変調搬送波を前記設定された移相量だけ移相
することを特徴とする請求項4記載の線形送信回路。5. The linear transmission circuit according to claim 4, wherein the phase shift means (36) shifts the demodulation carrier wave or the modulation carrier wave by the set phase shift amount.
ースト信号以降の各バースト信号毎に、前記位相差を複
数回測定し、その複数回の測定値の平均値に応じた移相
量を次のバースト信号入力前に前記移相手段(36)に
設定することを特徴とする請求項1乃至5のうちいずれ
か一項記載の線形送信回路。6. The phase difference measuring means (35) measures the phase difference a plurality of times for each burst signal after the second burst signal, and shifts the phase according to an average value of the plurality of measured values. 6. A linear transmission circuit as claimed in any one of claims 1 to 5, characterized in that the quantity is set in the phase shifting means (36) before the next burst signal is input.
ースト信号以降の前記移相量の設定を、それ以前の複数
のバースト信号の各々で測定した位相差の平均値に基づ
いて行なうことを特徴とする請求項4又は5記載の線形
送信回路。7. The phase difference measuring means (35) sets the phase shift amount after the third burst signal based on the average value of the phase differences measured for each of the plurality of burst signals before that. The linear transmission circuit according to claim 4 or 5, characterized in that:
1バースト信号送信時は第2バースト信号以降の送信時
に比し良好にすることを特徴とする請求項1記載の線形
送信回路。8. The linear transmission circuit according to claim 1, wherein the linearity of the power amplifier (33) is improved during transmission of the first burst signal compared to during transmission of the second burst signal and thereafter.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP19117092A JPH0637831A (en) | 1992-07-17 | 1992-07-17 | Linear transmission circuit |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP19117092A JPH0637831A (en) | 1992-07-17 | 1992-07-17 | Linear transmission circuit |
Publications (1)
Publication Number | Publication Date |
---|---|
JPH0637831A true JPH0637831A (en) | 1994-02-10 |
Family
ID=16270068
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP19117092A Withdrawn JPH0637831A (en) | 1992-07-17 | 1992-07-17 | Linear transmission circuit |
Country Status (1)
Country | Link |
---|---|
JP (1) | JPH0637831A (en) |
Cited By (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP2002057733A (en) * | 2000-08-01 | 2002-02-22 | Samsung Electronics Co Ltd | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
US7289776B2 (en) | 2002-10-01 | 2007-10-30 | Matsushita Electric Industrial Co., Ltd. | Nonlinear distortion compensation circuit with feedback and baseband reference signal that are phase and amplitude controllable |
US7305472B2 (en) | 1996-06-03 | 2007-12-04 | Microsoft Corporation | Method for downloading a web page to a client for efficient display on a television screen |
US7523399B2 (en) | 1996-06-03 | 2009-04-21 | Microsoft Corporation | Downloading software from a server to a client |
US7688914B2 (en) | 2005-08-17 | 2010-03-30 | Fujitsu Limited | Distortion compensation in wireless digital communication |
JP4637331B2 (en) * | 2000-08-01 | 2011-02-23 | 三星電子株式会社 | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
JP4703874B2 (en) * | 2001-03-22 | 2011-06-15 | 三星電子株式会社 | Nonlinear distortion compensation method and nonlinear distortion compensation circuit |
-
1992
- 1992-07-17 JP JP19117092A patent/JPH0637831A/en not_active Withdrawn
Cited By (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7305472B2 (en) | 1996-06-03 | 2007-12-04 | Microsoft Corporation | Method for downloading a web page to a client for efficient display on a television screen |
US7523399B2 (en) | 1996-06-03 | 2009-04-21 | Microsoft Corporation | Downloading software from a server to a client |
JP2002057733A (en) * | 2000-08-01 | 2002-02-22 | Samsung Electronics Co Ltd | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
JP4637331B2 (en) * | 2000-08-01 | 2011-02-23 | 三星電子株式会社 | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
JP4703874B2 (en) * | 2001-03-22 | 2011-06-15 | 三星電子株式会社 | Nonlinear distortion compensation method and nonlinear distortion compensation circuit |
US7289776B2 (en) | 2002-10-01 | 2007-10-30 | Matsushita Electric Industrial Co., Ltd. | Nonlinear distortion compensation circuit with feedback and baseband reference signal that are phase and amplitude controllable |
US7688914B2 (en) | 2005-08-17 | 2010-03-30 | Fujitsu Limited | Distortion compensation in wireless digital communication |
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Free format text: JAPANESE INTERMEDIATE CODE: A300 Effective date: 19991005 |