JPH0637831A - Linear transmission circuit - Google Patents
Linear transmission circuitInfo
- Publication number
- JPH0637831A JPH0637831A JP19117092A JP19117092A JPH0637831A JP H0637831 A JPH0637831 A JP H0637831A JP 19117092 A JP19117092 A JP 19117092A JP 19117092 A JP19117092 A JP 19117092A JP H0637831 A JPH0637831 A JP H0637831A
- Authority
- JP
- Japan
- Prior art keywords
- phase
- signal
- phase difference
- burst signal
- phase shift
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
Abstract
Description
【０００１】[0001]
【産業上の利用分野】本発明は線形送信回路に係り、特
に高出力電力増幅器の非線形歪補償回路を備えるディジ
タル無線方式における線形送信回路に関する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a linear transmission circuit, and more particularly to a linear transmission circuit in a digital radio system including a non-linear distortion compensation circuit for a high output power amplifier.
【０００２】移動通信などのディジタル無線通信の分野
では、特に周波数の有効利用の観点より線形変調方式が
用いられる。一方、自動車電話、携帯電話等移動通信端
末は、低消費電力化への要求が益々強くなってきてお
り、そのため移動通信端末で最も消費電力の大きな電力
増幅器の出力効率を向上させることが重要となる。In the field of digital radio communication such as mobile communication, a linear modulation system is used particularly from the viewpoint of effective use of frequencies. On the other hand, mobile communication terminals such as car phones and mobile phones are increasingly required to reduce power consumption. Therefore, it is important to improve the output efficiency of the power amplifier, which consumes the most power in mobile communication terminals. Become.
【０００３】しかし、電力増幅器を高効率増幅点で使用
すると非線形歪が発生するので、前記線形変調方式を利
用してディジタル無線送信するためには、非線形歪補償
回路を備える線形送信回路が必要とされる。However, since nonlinear distortion occurs when the power amplifier is used at a high efficiency amplification point, a linear transmission circuit having a nonlinear distortion compensation circuit is required for digital radio transmission using the linear modulation method. To be done.
【０００４】[0004]
【従来の技術】図１５は従来の線形送信回路の一例のブ
ロック図を示す。同図中、送信すべきディジタル入力信
号は信号処理部１に入力されて、ベースバンドのアナロ
グ信号である２つの直交信号（Ｑ信号）と同相信号（Ｉ
信号）とに変換された後、歪補償回路２に供給される。
歪補償回路２は直交復調器７により、送信信号を直交復
調して得たＱ’信号とＩ’信号と、信号処理回路１より
のＱ信号とＩ信号とから、非線形歪みの振幅と位相の両
方をベースバンドのベクトル平面で補償した、Ｑ信号と
Ｉ信号とを生成し、直交変調器３へ入力する。2. Description of the Related Art FIG. 15 shows a block diagram of an example of a conventional linear transmission circuit. In the figure, a digital input signal to be transmitted is input to the signal processing unit 1, and two quadrature signals (Q signals) which are baseband analog signals and an in-phase signal (I
Signal) and then supplied to the distortion compensation circuit 2.
The distortion compensating circuit 2 uses the quadrature demodulator 7 to quadrature demodulate the transmission signal and obtains the amplitude and phase of the nonlinear distortion from the Q ′ signal and the I ′ signal obtained by the signal processing circuit 1 and the Q signal and the I signal. A Q signal and an I signal, both of which are compensated by the vector plane of the baseband, are generated and input to the quadrature modulator 3.
【０００５】直交変調器３は搬送波発振器４よりの搬送
波でＱ信号とＩ信号を直交変調してＱＰＳＫ変調波を生
成し、そのＱＰＳＫ変調波を高出力増幅器５に供給す
る。高出力増幅器５の高効率増幅点で増幅されたＱＰＳ
Ｋ変調波は方向性結合器６で２分岐され、一方は送信さ
れ、他方は直交復調器７に供給され、ここで搬送波発振
器４より移相器８を通して入力される復調用搬送波に基
づいて直交復調される。ここで、上記の歪補償回路２１
はカルテシアン型歪補償方式による歪補償を行なう回路
で、例えば図１６に示す如き回路構成とされている。同
図中、信号処理回路１よりのＩ信号とＱ信号が、直交復
調器７よりのＩ’信号，Ｑ’信号と減算器１１，１２で
減算されて夫々の歪成分を低減した後、増幅器１３，１
４で増幅して出力する。この歪補償回路は負帰還により
歪みを低減する。The quadrature modulator 3 quadrature modulates the Q signal and the I signal with the carrier wave from the carrier wave oscillator 4 to generate a QPSK modulated wave, and supplies the QPSK modulated wave to the high output amplifier 5. QPS amplified at high efficiency amplification point of high-power amplifier 5
The K-modulated wave is branched into two by the directional coupler 6, one of which is transmitted and the other is supplied to the quadrature demodulator 7, where the quadrature is quadrature based on the demodulation carrier input from the carrier oscillator 4 through the phase shifter 8. Demodulated. Here, the above distortion compensation circuit 21
Is a circuit that performs distortion compensation by the Cartesian distortion compensation method, and has a circuit configuration as shown in FIG. 16, for example. In the figure, the I signal and the Q signal from the signal processing circuit 1 are subtracted by the I ′ signal and the Q ′ signal from the quadrature demodulator 7 and the subtracters 11 and 12 to reduce the respective distortion components, and then the amplifier 13, 1
Amplified by 4 and output. This distortion compensation circuit reduces distortion by negative feedback.
【０００６】また、歪補償回路２の他の例として図１７
に示す如き回路構成が知られている。同図中、減算器１
６，１７でＩ信号とＱ信号がＩ’信号とＱ’信号と減算
されて歪み成分が取り出され、増幅器１８，１９及びス
イッチＳＷ1 ，ＳＷ2 を通して加算器２０，２１に入力
され、ここで、Ｉ信号とＱ信号と加算される。これによ
り、加算器２０，２１からはＩ−Ｉ’とＱ−Ｑ’（歪成
分）が増幅器１８，１９のゲイン分の一に圧縮されたＩ
信号、Ｑ信号が取り出される。FIG. 17 shows another example of the distortion compensation circuit 2.
A circuit configuration as shown in is known. Subtractor 1 in FIG.
In 6 and 17, the I signal and the Q signal are subtracted from the I ′ signal and the Q ′ signal to extract the distortion component, which are input to the adders 20 and 21 through the amplifiers 18 and 19 and the switches SW 1 and SW 2 , respectively. , I signal and Q signal are added. As a result, I-I 'and Q-Q' (distortion components) from the adders 20 and 21 are compressed to the gain of the amplifiers 18 and 19 divided by I.
The signal and the Q signal are taken out.
【０００７】[0007]
【発明が解決しようとする課題】しかるに、上記の従来
回路では、種々の要因（例えば、温度、電力レベル、周
波数、経時変化など）に起因して直交復調器７の出力復
調位相が信号処理部１からの送信信号位相と異なると、
ベースバンド信号（Ｉ信号、Ｑ信号）が正しく出力され
ず、歪補償が正しくできない。However, in the above-mentioned conventional circuit, the output demodulation phase of the quadrature demodulator 7 is caused by various factors (for example, temperature, power level, frequency, aging, etc.). If it is different from the transmission signal phase from 1,
The baseband signal (I signal, Q signal) is not output correctly, and distortion compensation cannot be performed correctly.
【０００８】本発明は上記の点に鑑みなされたもので、
バースト信号の立ち上がり時間中に位相差を測定し、そ
の測定結果に基づいて復調位相を調整することにより、
上記の課題を解決した線形送信回路を提供することを目
的とする。The present invention has been made in view of the above points,
By measuring the phase difference during the rise time of the burst signal and adjusting the demodulation phase based on the measurement result,
It is an object of the present invention to provide a linear transmission circuit that solves the above problems.
【０００９】[0009]
【課題を解決するための手段】図１は本発明の原理構成
図を示す。本発明は、送信ベースバンド信号を歪補償回
路３１を通して直交変調器３２へ入力し、直交変調器３
２の出力直交変調波を電力増幅器３３で増幅した後、時
分割多重方式のバースト信号として出力すると共に直交
復調器３４で復調し、その復調信号を前記歪補償回路３
１に帰還入力してベースバンドのベクトル座標で歪みを
補償する線形送信回路において、位相差測定手段３５と
移相手段３６を有する。FIG. 1 is a block diagram showing the principle of the present invention. According to the present invention, the transmission baseband signal is input to the quadrature modulator 32 through the distortion compensation circuit 31, and the quadrature modulator 3
After the output quadrature modulated wave of No. 2 is amplified by the power amplifier 33, it is output as a burst signal of the time division multiplexing system and demodulated by the quadrature demodulator 34, and the demodulated signal is the distortion compensation circuit 3
The linear transmission circuit which feeds back to 1 and compensates the distortion by the vector coordinates of the baseband has a phase difference measuring means 35 and a phase shift means 36.
【００１０】上記位相差測定手段３５は前記バースト信
号の立ち上がり時間中に歪補償回路３１に入力されるベ
ースバンド信号の位相と前記復調信号の位相との位相差
を測定する。移相手段３６は位相差測定手段３５による
位相差測定後に該測定位相差をゼロにするように、直交
変調器３４の復調搬送波と直交変調器３２の変調搬送波
との相対位相を移相調整する。The phase difference measuring means 35 measures the phase difference between the phase of the baseband signal input to the distortion compensation circuit 31 and the phase of the demodulated signal during the rising time of the burst signal. The phase shift means 36 performs phase shift adjustment of the relative phase between the demodulation carrier wave of the quadrature modulator 34 and the modulation carrier wave of the quadrature modulator 32 so that the measured phase difference becomes zero after the phase difference measurement means 35 measures the phase difference. .
【００１１】[0011]
【作用】本発明では、時分割多重（ＴＤＭＡ）方式のバ
ースト信号を送信するが、バースト信号の立ち上がりの
時間（ランプタイム）はスペクトラムの拡がりを抑える
ために滑らかに立ち上がり、また電力増幅器３３より出
力される電力は小さいので、電力増幅器３３の非線形歪
みは小さい。しかも、このバースト信号のランプタイム
は伝送データに依存せず、一定の符号が送出されるの
で、このランプタイムは非線形歪補償を必要としない。In the present invention, a burst signal of the time division multiplexing (TDMA) system is transmitted, but the rising time (ramp time) of the burst signal rises smoothly to suppress the spread of the spectrum, and is output from the power amplifier 33. Since the generated power is small, the nonlinear distortion of the power amplifier 33 is small. Moreover, since the ramp time of this burst signal does not depend on the transmission data and a constant code is transmitted, this ramp time does not require nonlinear distortion compensation.
【００１２】そこで、本発明はバースト信号のランプタ
イム中に歪補償回路３１の動作を実質的に停止させて位
相差測定手段３５により位相差の測定を行なうことによ
り、位相差を最も位相測定誤差が少ない状態で測定する
ことができる。そして、この位相測定誤差の少ない位相
差測定結果に基づいて、復調搬送波と変調搬送波の相対
位相を上記位相差がゼロとなるように変化させ、その後
に歪補償の制御を開始する。Therefore, according to the present invention, the operation of the distortion compensating circuit 31 is substantially stopped during the ramp time of the burst signal, and the phase difference is measured by the phase difference measuring means 35. Can be measured in a state where there is little. Then, based on the phase difference measurement result with less phase measurement error, the relative phase between the demodulated carrier wave and the modulated carrier wave is changed so that the phase difference becomes zero, and then the distortion compensation control is started.
【００１３】[0013]
【実施例】本実施例の線形送信回路は、ディジタル方式
自動車電話システムの移動局に組込まれ、基地局に対し
て情報を送信するので、まずディジタル方式自動車電話
システムの概要について説明する。この自動車電話シス
テムでは移動局は３チャネル多重ＴＤＭＡ方式を使用
し、例えば図２に示す如きスロット配置とされている。
同図中、（Ａ）は基地局から移動局へ送信される信号の
スロット配置を示し、第３チャネルの移動局への送信ス
ロットＴB-M3，第１チャネルの移動局への送信スロット
ＴB-M1，第２チャネルの移動局への送信スロットＴB-M2
が順次時系列的に合成されてなる。DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The linear transmission circuit of this embodiment is incorporated in a mobile station of a digital car telephone system and transmits information to a base station. Therefore, the outline of the digital car telephone system will be described first. In this car telephone system, the mobile station uses the three-channel multiplex TDMA system and has the slot arrangement as shown in FIG. 2, for example.
In the figure, (A) shows the slot arrangement of the signal transmitted from the base station to the mobile station. The transmission slot T B-M3 to the mobile station of the third channel and the transmission slot T to the mobile station of the first channel are shown. B-M1 , transmission slot T B-M2 to mobile station of second channel
Are sequentially combined in time series.
【００１４】また、図２（Ｂ）は移動局の送信信号を受
信する基地局の受信信号のスロット配置を示し、第１，
第２及び第３チャネルの順で各移動局からの送信スロッ
トが時系列的に合成されている。図２（Ｃ）は第１チャ
ネルの移動局Ｍ１のスロット配置を示し、送信スロット
ＴM1-B，アンテナ切替ダイバーシチ用レベル測定ＬＭ，
基地局からの送信信号の受信スロットＲB-M1，アイドル
Ｉの各スロットが時系列的に合成されている。FIG. 2 (B) shows the slot arrangement of the received signal of the base station which receives the transmitted signal of the mobile station.
The transmission slots from each mobile station are combined in time series in the order of the second and third channels. FIG. 2C shows the slot arrangement of the mobile station M1 of the first channel, including the transmission slot T M1-B , the antenna switching diversity level measurement LM,
The reception slot RB -M1 of the transmission signal from the base station and each slot of the idle I are combined in time series.
【００１５】同様にして、第２チャネルの移動局Ｍ２の
スロット配置は図２（Ｄ）に、また第３チャネルの移動
局Ｍ３のスロット配置は図２（Ｅ）に夫々示す如くにな
り、同図（Ａ）に示す基地局からの送信スロットに対応
して受信スロットが配置されている。これにより、送受
信で二つのキャリアで３つの移動局とのＴＤＭＡ通信が
可能になる。Similarly, the slot arrangement of the mobile station M2 of the second channel is as shown in FIG. 2 (D), and the slot arrangement of the mobile station M3 of the third channel is as shown in FIG. 2 (E). Reception slots are arranged corresponding to the transmission slots from the base station shown in FIG. This enables TDMA communication with three mobile stations using two carriers for transmission and reception.
【００１６】各移動局は自己に割当てられたタイムスロ
ットで基地局との間で、無線通信を行なう。各スロット
毎に独立した物理チャネルとして制御チャネルと通信チ
ャネルがある。移動局から基地局へ送信される信号の制
御チャネルは図３（Ａ）に示す如き信号フォーマットと
され、通信チャネルは同図（Ｂ）に示す如き信号フォー
マットとされている。Each mobile station performs radio communication with the base station in a time slot assigned to itself. There are a control channel and a communication channel as an independent physical channel for each slot. The control channel of the signal transmitted from the mobile station to the base station has a signal format as shown in FIG. 3 (A), and the communication channel has a signal format as shown in FIG. 3 (B).
【００１７】同図（Ａ），（Ｂ）中、Ｒは４ビットのバ
ースト過渡応答用ガード時間、Ｐは２ビットのプリアン
ブル、ＣＡＣは制御信号、ＳＷは２０ビットの同期ワー
ド、ＣＣは８ビットのカラーコード、Ｇはガード時間、
ＴＣＨはユーザ情報転送用チャネル、ＳＦは１ビットの
スチールフラグ、ＳＡＣＣＨは低速付随制御チャネルで
ある。In FIGS. 1A and 1B, R is a 4-bit burst transient response guard time, P is a 2-bit preamble, CAC is a control signal, SW is a 20-bit synchronization word, and CC is 8-bit. Color code, G is guard time,
TCH is a user information transfer channel, SF is a 1-bit steal flag, and SACCH is a low speed associated control channel.
【００１８】そして、送信回路からは図３（Ｃ）に示す
如く、上記のバースト過渡応答用ガード時間Ｒの間に立
ち上がり、ガード時間Ｇで立ち下がり、かつ、その立ち
上がりと立ち下がりの間は例えば搬送波を情報でπ／４
シフトＱＰＳＫ変調した変調波が存在するバースト信号
を自己に割当てられたタイムスロット毎に送信する。Then, as shown in FIG. 3C, the transmitter circuit rises during the burst transient response guard time R and falls during the guard time G, and between the rise and the fall, for example. Carrier wave with information π / 4
A burst signal in which a modulated wave subjected to shift QPSK modulation is present is transmitted for each time slot assigned to itself.
【００１９】次に上記のディジタル方式自動車電話シス
テムの移動局に組込まれ、図３（Ｃ）に示すＴＤＭＡ方
式のバースト信号を送信する送信回路の各実施例につい
て説明する。図４は本発明になる線形送信回路の第１実
施例のブロック図を示す。同図中、図１と同一構成部分
には同一符号を付してある。Next, each embodiment of the transmission circuit incorporated in the mobile station of the above-mentioned digital type automobile telephone system and transmitting the burst signal of the TDMA type shown in FIG. 3C will be described. FIG. 4 shows a block diagram of a first embodiment of a linear transmission circuit according to the present invention. In the figure, the same components as those in FIG. 1 are designated by the same reference numerals.
【００２０】図４において、送信する情報データは信号
処理部３０に供給され、ここでベースバンドのアナログ
信号であるＩ信号とＱ信号に変換された後、図１７に示
した構成の歪補償回路３１を通して直交変調器３２及び
移相制御器４２に夫々入力される。直交変調器３２は搬
送波発振器４０よりの変調搬送波を上記２つのＩ信号及
びＱ信号で例えばπ／４シフトＱＰＳＫ変調する。In FIG. 4, the information data to be transmitted is supplied to the signal processing unit 30, where it is converted into I and Q signals which are baseband analog signals, and then the distortion compensating circuit having the configuration shown in FIG. It is inputted to the quadrature modulator 32 and the phase shift controller 42 through 31 respectively. The quadrature modulator 32 subjects the modulated carrier wave from the carrier wave oscillator 40 to, for example, π / 4 shift QPSK modulation with the above two I signals and Q signals.
【００２１】このπ／４シフトＱＰＳＫ変調波は電力増
幅器３３で電力増幅された後、方向性結合器４１で２分
岐され、一方は基地局へバースト信号として送信される
一方、他方は直交復調器３４に入力され、ここで移相器
回転手段４３を通して入力される搬送波発振器４０より
の復調搬送波を用いて直交復調されて復調Ｉ信号及び復
調Ｑ信号とされる。この復調Ｉ信号及び復調Ｑ信号は移
相制御器４２に入力される一方、歪補償回路３１に入力
される。This π / 4 shift QPSK modulated wave is power-amplified by a power amplifier 33 and then branched into two by a directional coupler 41, one of which is transmitted to a base station as a burst signal and the other of which is a quadrature demodulator. A demodulation carrier signal 40 is input to the signal generator 34 and is orthogonally demodulated using the demodulation carrier wave from the carrier wave oscillator 40 input through the phase shifter rotating means 43 to obtain a demodulation I signal and a demodulation Q signal. The demodulated I signal and the demodulated Q signal are input to the phase shift controller 42, while being input to the distortion compensation circuit 31.
【００２２】本実施例は移相制御器４２と移相器回転手
段４３を有する点に特徴がある。移相制御器４２は前記
位相差測定手段３５に相当し、例えば図５に示す如き構
成とされている。同図中、ＡＤ変換器（ＡＤＣ）421 及
び422 には直交変調器３２に入力されるＩ信号ＭＤＩ及
びＱ信号ＭＤＱが夫々入力されてディジタル信号に変換
される。また、ＡＤＣ423 及び424 には直交復調器３４
より出力される復調Ｉ信号ＤＥＩ及び復調Ｑ信号ＤＥＱ
が夫々入力されてディジタル信号に変換される。The present embodiment is characterized in having a phase shift controller 42 and a phase shifter rotating means 43. The phase shift controller 42 corresponds to the phase difference measuring means 35 and has a structure as shown in FIG. 5, for example. In the figure, the AD converters (ADCs) 421 and 422 respectively receive the I signal MDI and the Q signal MDQ input to the quadrature modulator 32 and are converted into digital signals. In addition, the quadrature demodulator 34 is provided in the ADCs 423 and 424.
Demodulated I signal DEI and demodulated Q signal DEQ output from
Are input and converted into digital signals.
【００２３】ＡＤＣ421 及び422 の各出力信号はtan -1
ＭＤＱ／ＭＤＩの演算を４２５で行い、送信位相が検出
される。他方、ＡＤＣ423 及び424 の各出力信号はtan
-1ＤＥＱ／ＤＥＩの演算を４２６で行い、復調位相が検
出される。回路４２５，４２６はＲＯＭ等を用いれば簡
単に構成出来る。The output signals of the ADCs 421 and 422 are tan -1.
The MDQ / MDI calculation is performed at 425, and the transmission phase is detected. On the other hand, the output signals of ADC423 and 424 are tan.
-1 DEQ / DEI calculation is performed at 426, and the demodulation phase is detected. The circuits 425 and 426 can be easily constructed by using a ROM or the like.
【００２４】演算部429 は上記の送信位相と復調位相と
の差を位相差として測定し、その位相差が零となるよう
な制御信号を生成して図４の移相器回転手段４３へ供給
する。なお、演算器429 はＴＤＭＡ制御部（図示せず）
よりの制御信号に同期してタイミング信号を発生するタ
イミング発生部420 よりのタイミング信号に基づいて位
相差測定の演算を行なう。The arithmetic unit 429 measures the difference between the transmission phase and the demodulation phase as a phase difference, generates a control signal that makes the phase difference zero, and supplies it to the phase shifter rotating means 43 in FIG. To do. The arithmetic unit 429 is a TDMA control unit (not shown).
Phase difference measurement is performed based on the timing signal from the timing generation unit 420 that generates the timing signal in synchronization with the control signal.
【００２５】移相器回転手段４３はＥＰＳ（エンドレス
・フェーズ・シフタ）回路と称される回路で、前記移相
手段３６の一例を示しており、例えば図６に示す如き構
成とされている。同図中、端子430 は移相制御器４２の
出力制御信号の入力端子、端子431 は搬送波発振器４０
よりの搬送波入力端子である。入力端子430 よりの制御
信号はcos θ演算器432 及びsin θ演算器433 に夫々入
力されてcos θ，sinθの演算結果を出力させる。The phase shifter rotating means 43 is a circuit called an EPS (endless phase shifter) circuit, and shows an example of the phase shifting means 36, and has a structure as shown in FIG. 6, for example. In the figure, a terminal 430 is an input terminal for the output control signal of the phase shift controller 42, and a terminal 431 is a carrier wave oscillator 40.
This is a carrier wave input terminal. The control signal from the input terminal 430 is input to the cos θ calculator 432 and the sin θ calculator 433, respectively, and outputs the calculation results of cos θ and sin θ.
【００２６】cos θ，sin θの各演算結果は夫々ＤＡ変
換器434 ，435 でアナログ信号に変換された後、乗算器
436 ，437 に供給され、ここで９０°移相器438 よりの
互いに位相が９０°異なるが周波数は同一の搬送波と乗
算される。乗算器436 ，437の各出力信号は加算器439
に供給されて加算され、前記入力端子430 よりの制御信
号の値に応じた移相量の復調搬送波とされて図４の直交
復調器３４へ出力される。The calculation results of cos θ and sin θ are converted into analog signals by DA converters 434 and 435, respectively, and then multiplied by multipliers.
436, 437, where the frequencies are multiplied by the same carrier by a 90 ° phase shifter 438, but 90 ° out of phase with each other. The output signals of the multipliers 436 and 437 are added by the adder 439.
Is added to the quadrature demodulator 34 and is added to the quadrature demodulator 34 of FIG. 4 to obtain a demodulated carrier having a phase shift amount corresponding to the value of the control signal from the input terminal 430.
【００２７】上記の構成の第１実施例において、移相制
御器４２での位相差測定は図５に示すタイミング発生部
420 よりのタイミング信号が、図７に示す電力増幅器
（又は方向性結合器）の出力バースト信号の立ち上がり
時間（ランプタイム）中に発生出力されるために、ラン
プタイム中に行なわれることとなる。In the first embodiment having the above structure, the phase difference measurement by the phase shift controller 42 is performed by the timing generator shown in FIG.
Since the timing signal from 420 is generated and output during the rising time (ramp time) of the output burst signal of the power amplifier (or directional coupler) shown in FIG. 7, it is performed during the ramp time.
【００２８】図７からわかるように、バースト信号のラ
ンプタイムは電力増幅器３３の出力電力が小さいために
非線形歪が少なく、またランプタイム中のデータは送信
データに関係のない一定の符号であるため、このランプ
タイム中の位相差測定によりかなり正確な位相差測定が
できる。As can be seen from FIG. 7, the ramp time of the burst signal has a small non-linear distortion because the output power of the power amplifier 33 is small, and the data during the ramp time is a constant code irrelevant to the transmission data. By the phase difference measurement during this ramp time, a fairly accurate phase difference measurement can be performed.
【００２９】ただし、位相差が大きいとフィードバック
系は不安定となり発振を生じることもあり得るので、図
４の信号処理部３０の出力信号により歪補償回路３１内
のスイッチ（図１７のＳＷ1 ，ＳＷ2 ）を図７に示す如
くランプタイム中はオフとし、フィードバック系をオー
プンとして帰還を遮断しておく。ランプタイム経過後は
図７に示す如く歪補償回路３１内のスイッチＳＷ1 及び
ＳＷ2 をオンとし、フィードバックループを形成して歪
補償回路３１によりベースバンドのベクトル座標で歪み
を補償するカルテシアン型歪補償を行なう。このカルテ
シアン型歪補償は測定位相差を零とするように移相器回
転手段４３により変調移相量が制御され、常に送信位相
と復調位相とが合わせられているため、歪補償の制御を
安定に行なえる。However, if the phase difference is large, the feedback system may become unstable and oscillation may occur. Therefore, the output signal of the signal processing unit 30 in FIG. 4 causes a switch in the distortion compensating circuit 31 (SW 1 in FIG. 17, SW 2 ) is turned off during the ramp time as shown in FIG. 7, and the feedback system is opened to interrupt the feedback. After the ramp time has elapsed, as shown in FIG. 7, the switches SW 1 and SW 2 in the distortion compensating circuit 31 are turned on to form a feedback loop and the distortion compensating circuit 31 compensates the distortion at the vector coordinates of the base band. Perform distortion compensation. In this Cartesian distortion compensation, the modulation phase shift amount is controlled by the phase shifter rotating means 43 so that the measured phase difference becomes zero, and the transmission phase and the demodulation phase are always matched, so that the distortion compensation is controlled. You can do it stably.
【００３０】図８は本発明になる線形送信回路の第２実
施例のブロック図を示す。同図中、図４と同一構成部分
には同一符号を付し、その説明を省略する。本実施例は
図８に示すように、搬送波発振器４０の出力搬送波を復
調搬送波として直交復調器３４に供給する一方、移相器
回転手段４６を通して変調搬送波として直交変調器３２
に供給するようにした点が図４の第１実施例と異なる。
位相差測定タイミングは第１実施例と同じである。FIG. 8 shows a block diagram of a second embodiment of the linear transmission circuit according to the present invention. In the figure, parts that are the same as the parts shown in FIG. 4 are given the same reference numerals, and descriptions thereof will be omitted. In this embodiment, as shown in FIG. 8, the output carrier of the carrier oscillator 40 is supplied to the quadrature demodulator 34 as a demodulation carrier, while the quadrature modulator 32 is supplied as a modulation carrier through the phase shifter rotating means 46.
Is different from the first embodiment in FIG.
The phase difference measurement timing is the same as in the first embodiment.
【００３１】移相器回転手段４６は搬送波発振器４０と
共に前記移相手段３６を構成しており、図６と同じ構成
とされている。本実施例では位相差測定結果に応じて位
相差が零となるよう、変調搬送波の方を移相するように
したものであり、第１実施例と同じ効果が得られる。The phase shifter rotating means 46 constitutes the phase shifting means 36 together with the carrier wave oscillator 40, and has the same configuration as in FIG. In this embodiment, the modulated carrier wave is phase-shifted so that the phase difference becomes zero according to the phase difference measurement result, and the same effect as that of the first embodiment can be obtained.
【００３２】次に本発明の第３実施例について説明す
る。本実施例はブロック構成は図４と図８のどちらでも
よく、歪補償回路３１を図９の回路構成とすると共に、
歪補償回路の動作と移相制御器４２による位相差測定を
図１０に示す如きタイミングで行なうようにしたもので
ある。Next, a third embodiment of the present invention will be described. In this embodiment, the block configuration may be either of FIG. 4 or FIG. 8, and the distortion compensating circuit 31 has the circuit configuration of FIG.
The operation of the distortion compensation circuit and the phase difference measurement by the phase shift controller 42 are performed at the timing shown in FIG.
【００３３】図９に示す歪補償回路３１は直交変調器３
２の入力Ｉ信号ＭＤＩ，Ｑ信号ＭＤＱを直交復調器３４
の出力復調Ｉ信号ＤＥＩ及び復調Ｑ信号ＤＥＱとの誤差
信号を減算器５１及び５２で夫々生成し、その誤差信号
を可変利得増幅器５３，５４を通して加算器５５，５６
に入力し、入力信号ＭＤＩ，ＭＤＱと加算する構成であ
る。The distortion compensation circuit 31 shown in FIG.
The quadrature demodulator 34 receives the two input I signals MDI and Q signals MDQ.
Of the output demodulated I signal DEI and the demodulated Q signal DEQ are generated by subtractors 51 and 52, respectively, and the error signals are added through variable gain amplifiers 53 and 54 to adders 55 and 56.
To the input signals MDI and MDQ.
【００３４】ここで、可変利得増幅器５３及び５４の利
得制御は信号処理部３０の出力制御信号により、図１０
に示す如く、電力増幅器３３の出力バースト信号の立ち
上がり時間（ランプタイム）中に移相制御器４２による
位相差測定が行なわれているときは利得最小（従って歪
補償のフィードバック量最小）とし、位相差測定後のラ
ンプタイム期間は利得を徐々に大に可変し、ランプタイ
ム後は利得を所定値にセットするように行なわれる。Here, the gain control of the variable gain amplifiers 53 and 54 is performed by the output control signal of the signal processing unit 30 as shown in FIG.
As shown in, when the phase difference is measured by the phase shift controller 42 during the rise time (ramp time) of the output burst signal of the power amplifier 33, the gain is set to the minimum (therefore, the feedback amount for distortion compensation is set to the minimum), and During the ramp time period after the phase difference measurement, the gain is gradually changed to a large value, and after the ramp time, the gain is set to a predetermined value.
【００３５】これにより、本実施例によれば、位相差測
定が終了した後は歪補償のフィードバック量が徐々に大
きくされることとなり、歪補償開始時に過渡現象が生じ
て信号が歪むことを防止することができる。As a result, according to the present embodiment, the feedback amount for distortion compensation is gradually increased after the phase difference measurement is completed, and a transient phenomenon is prevented at the beginning of distortion compensation to prevent the signal from being distorted. can do.
【００３６】次に本発明の第４実施例について説明す
る。本実施例は図４に示した線形送信回路において、位
相差の測定と歪補償制御を図１１に示す如きタイミング
で行なうようにしたものである。なお、本実施例の歪補
償回路３１は図９及び図１７のどちらの構成でもよい。Next, a fourth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4, the measurement of the phase difference and the distortion compensation control are performed at the timings shown in FIG. The distortion compensation circuit 31 of this embodiment may have either of the configurations shown in FIG. 9 and FIG.
【００３７】すなわち、本実施例は図１１に示す如く送
信開始より第１バースト信号Ｂ1 ，第２バースト信号Ｂ
2 ，第３バースト信号Ｂ3 ，第４バースト信号Ｂ4 ，…
というように送信するに際し、移相制御器４２による位
相差測定は各バースト信号のランプタイム中に行なうと
共に、その測定位相差を零とするような移相量を移相器
回転手段４３に指示する制御信号の送出タイミングを図
１１に示すように１バースト後の移相量調整に用いるこ
とができるタイミングとする。That is, in this embodiment, as shown in FIG. 11, the first burst signal B 1 and the second burst signal B 1 are transmitted from the start of transmission.
2 , third burst signal B 3 , fourth burst signal B 4 , ...
In transmitting, the phase difference is measured by the phase shift controller 42 during the ramp time of each burst signal, and the phase shift amount is set to the phase shifter rotating means 43 so that the measured phase difference becomes zero. The transmission timing of the control signal is set to a timing that can be used for adjusting the amount of phase shift after one burst as shown in FIG.
【００３８】従って、歪補償回路３１の歪補償動作は図
１１の「歪補償制御」の欄に示すように、第１バースト
信号Ｂ1 については行なわず、第２バースト信号Ｂ2 以
降の各バースト信号について行なわれる。Therefore, the distortion compensating operation of the distortion compensating circuit 31 is not performed for the first burst signal B 1 as shown in the “Distortion Compensation Control” column of FIG. 11, but for each burst after the second burst signal B 2. Performed on signals.
【００３９】次に本発明の第５実施例について説明す
る。本実施例は図８に示した線形送信回路において、位
相差の測定と歪補償制御とを図１１に示す如きタイミン
グで行なうようにしたものである。Next, a fifth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 8, the measurement of the phase difference and the distortion compensation control are performed at the timings shown in FIG.
【００４０】次に本発明の第６実施例について説明す
る。本実施例は図４又は図８に示した線形送信回路にお
いて、位相差測定と移相量設定及び歪補償制御を図１２
に模式的に示すタイミングで行なうようにしたものであ
る。なお、歪補償回路３１は図９及び図１７に示した回
路のいずれでもよい。Next, a sixth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4 or 8, phase difference measurement, phase shift amount setting, and distortion compensation control are performed in FIG.
The timing is schematically shown in FIG. The distortion compensation circuit 31 may be any of the circuits shown in FIGS. 9 and 17.
【００４１】本実施例は図１２の「位相測定」の欄に模
式的に示す如く、移相制御器４２は第１バースト信号Ｂ
1 のランプタイムで前記位相差測定を１回行ない、それ
により得られた測定結果に基づく移相量の設定を、同図
に「移相設定」の欄で示すように第１バースト信号Ｂ1
のランプタイム終了直前に移相器回転手段４３に対して
行なう。この移相設定により歪補償回路３１は送信位相
と復調位相とが一致されたＩ信号ＭＤＩ及びＤＥＩ，Ｑ
信号ＭＤＱ及びＤＥＱが入力されるから第１バースト信
号Ｂ1 より歪み補償の制御を行なうようにするが、第
４，第５実施例のように第１バースト信号Ｂ1 について
は歪み補償の制御は行なわなくてもよい。ただし、第２
バースト信号Ｂ2 に対しては図１２に「歪補償制御」の
欄で示す如く歪補償を第１バースト信号Ｂ1 の位相差測
定結果に基づいて行なう。In the present embodiment, as schematically shown in the column "Phase measurement" in FIG. 12, the phase shift controller 42 sets the first burst signal B
Performed once the phase difference measured in the first ramp time, it phase shift setting based on the measurement results obtained by the first burst signal, as shown in the column of "phase shift setting" in Fig. B 1
Immediately before the end of the ramp time, the phase shifter rotating means 43 is performed. With this phase shift setting, the distortion compensating circuit 31 causes the I signal MDI and DEI, Q whose transmission phase and demodulation phase are matched.
Since the signals MDQ and DEQ are input, the distortion compensation control is performed from the first burst signal B 1, but the distortion compensation control is not performed on the first burst signal B 1 as in the fourth and fifth embodiments. You don't have to do it. However, the second
Performs distortion compensation as shown in the section of "distortion compensation control" in FIG. 12 on the basis of the first burst signal B 1 of the phase difference measurements at burst signal B 2.
【００４２】続いて、移相制御器４２は第２バースト信
号Ｂ2 以降の各バースト信号が入力される毎に、ランプ
タイムであるか否かに関係なく、複数回ずつ位相差測定
を行ない、その複数回の測定値の平均値を算出して次の
バースト信号の歪補償を行なうべく、次のバースト信号
の入力直前に上記平均値に基づく移相量の設定を移相器
回転手段４３に対して行なう。複数回の位相差測定は図
５のタイミング発生部420 の出力タイミング信号周波
数、ＡＤＣ421 〜424 のクロック周波数等を高くするこ
とにより行なえる。Subsequently, the phase shift controller 42 performs the phase difference measurement a plurality of times each time each burst signal after the second burst signal B 2 is input, regardless of whether it is the ramp time or not. Immediately before the input of the next burst signal, the phase shift amount setting based on the average value is set to the phase shifter rotating means 43 in order to calculate the average value of the measured values of the plurality of times and perform the distortion compensation of the next burst signal. To do. The phase difference measurement can be performed a plurality of times by increasing the output timing signal frequency of the timing generator 420 and the clock frequencies of the ADCs 421 to 424 shown in FIG.
【００４３】従って、本実施例によれば、第３バースト
信号以降のバースト信号送信時には、前回のバースト信
号の複数回の位相差測定結果の平均値に基づく移相量設
定が行なわれ、歪補償回路３１による歪補償動作が行な
われるため、瞬間的な位相変動に追従することなく安定
な歪補償制御ができる。Therefore, according to this embodiment, when transmitting the burst signal after the third burst signal, the phase shift amount is set based on the average value of the phase difference measurement results of the previous burst signal a plurality of times, and the distortion compensation is performed. Since the distortion compensation operation is performed by the circuit 31, stable distortion compensation control can be performed without following an instantaneous phase fluctuation.
【００４４】次に本発明の第７実施例について説明す
る。本実施例は図４又は図８に示した線形送信回路にお
いて、位相差測定と移相量設定及び歪補償制御を図１３
に模式的に示すタイミングで行なうようにしたものであ
る。なお、歪補償回路３１は図９及び図１７に示した回
路のいずれでもよい。Next, a seventh embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4 or 8, phase difference measurement, phase shift amount setting, and distortion compensation control are performed in FIG.
The timing is schematically shown in FIG. The distortion compensation circuit 31 may be any of the circuits shown in FIGS. 9 and 17.
【００４５】本実施例は図１３の「位相測定」の欄に模
式的に示す如く、移相制御器４２は各バースト信号のラ
ンプタイムで位相差測定が１回行なわれる点は第１実施
例と同様であるが、移相量設定が第１実施例と異なる。In this embodiment, as schematically shown in the column "Phase measurement" in FIG. 13, the phase shift controller 42 performs the phase difference measurement once at the ramp time of each burst signal. However, the phase shift amount setting is different from that of the first embodiment.
【００４６】すなわち、本実施例では第１バースト信号
Ｂ1 送信時は図１３に示すように、移相制御器４２が第
１バースト中に位相差測定を行ない、その測定結果に基
づく移相量の設定を第２バースト信号Ｂ2 入力前に移相
器回転手段４３に対して行なう。また、この第１バース
ト信号Ｂ1 送信時は歪補償回路３１の動作を停止（オ
フ）させる。That is, in this embodiment, when transmitting the first burst signal B 1 , the phase shift controller 42 measures the phase difference during the first burst as shown in FIG. 13, and the phase shift amount based on the measurement result. Is set to the phase shifter rotating means 43 before inputting the second burst signal B 2 . Further, the operation of the distortion compensation circuit 31 is stopped (turned off) when the first burst signal B 1 is transmitted.
【００４７】続いて、第２バースト信号Ｂ2 から第ｎ−
１バースト信号Ｂn-1 送信時は、図１３に示すように、
移相制御器４２が各バースト信号のランプタイムで１回
ずつ位相差測定を行ない、それら（ｎ−２）個の位相差
測定結果の平均値を算出し、その平均値に基づく移相量
の設定を第ｎバースト信号Ｂn 入力前に移相器回転手段
４３に対して行なう。Then, from the second burst signal B 2 to the n-th
At the time of transmitting one burst signal B n−1 , as shown in FIG.
The phase shift controller 42 performs the phase difference measurement once at the ramp time of each burst signal, calculates the average value of the (n-2) phase difference measurement results, and calculates the phase shift amount based on the average value. The setting is performed on the phase shifter rotating means 43 before inputting the nth burst signal B n .
【００４８】従って、本実施例では、第２バースト信号
Ｂ2 から第ｎ−１バースト信号Ｂn- 1 送信までは第１バ
ースト信号Ｂ1 のランプタイムで測定した位相差に基づ
く移相量が設定され、歪補償が行なわれ、また第ｎバー
スト信号Ｂn 以降は第２バースト信号Ｂ2 から第ｎ−１
バースト信号Ｂn-1 の各ランプタイムで測定した各位相
差の平均値に基づく移相量が設定され、歪補償が行なわ
れる。Therefore, in this embodiment, the amount of phase shift based on the phase difference measured by the ramp time of the first burst signal B 1 is from the second burst signal B 2 to the transmission of the n−1 th burst signal B n− 1. Is set, distortion compensation is performed, and after the nth burst signal B n, the second burst signal B 2 to the n−1th burst signal B 2 are set.
A phase shift amount based on the average value of the phase differences measured at each ramp time of the burst signal B n-1 is set, and distortion compensation is performed.
【００４９】次に本発明の第８実施例について説明す
る。本実施例は図４又は図８に示した線形送信回路にお
いて、電力増幅器３３の線形性と図９又は図１７に示し
た歪補償回路３１の歪補償制御を図１４に模式的に示す
如く行なうようにしたものである。Next, an eighth embodiment of the present invention will be described. In this embodiment, in the linear transmission circuit shown in FIG. 4 or 8, the linearity of the power amplifier 33 and the distortion compensation control of the distortion compensation circuit 31 shown in FIG. 9 or 17 are performed as schematically shown in FIG. It was done like this.
【００５０】すなわち、図１４の「歪補償制御」の欄に
示すように第１バースト信号Ｂ1 送信時のみ歪補償回路
３１の動作を停止（オフ）し、第２バースト信号Ｂ2 以
降の送信時は歪補償回路３１を動作させる。このため、
本実施例では歪補償を行なっていない第１バースト信号
Ｂ1 送信時は図１４の「増幅器線形性」の欄に丸を付け
て示したように、電力増幅器３３の線形性を良い状態に
し、第２バースト信号Ｂ2 以降の各バースト信号送信時
は歪補償動作を行なっているので、図１４にＸで示すよ
うに電力増幅器３３の線形性を悪くし、その分電力増幅
器３３の効率を良くする。That is, as shown in the "Distortion compensation control" column of FIG. 14, the operation of the distortion compensation circuit 31 is stopped (turned off) only when the first burst signal B 1 is transmitted, and the second burst signal B 2 and the subsequent transmissions are transmitted. At this time, the distortion compensation circuit 31 is operated. For this reason,
In the present embodiment, when transmitting the first burst signal B 1 for which distortion compensation has not been performed, the linearity of the power amplifier 33 is set to a good state as indicated by a circle in the "Amplifier linearity" column of FIG. Since distortion compensation operation is performed during transmission of each burst signal after the second burst signal B 2, the linearity of the power amplifier 33 is deteriorated as shown by X in FIG. 14, and the efficiency of the power amplifier 33 is improved correspondingly. To do.
【００５１】電力増幅器３３の線形性を向上する方法と
しては、例えば増幅用トランジスタが電界効果トランジ
スタ（ＦＥＴ）の場合は、 ゲートのバイアスを浅く
する、 ドレインの電圧を高くする、 負荷条件を
変えるなどがある。As a method of improving the linearity of the power amplifier 33, for example, when the amplifying transistor is a field effect transistor (FET), the bias of the gate is made shallow, the voltage of the drain is made high, the load condition is changed, etc. There is.
【００５２】[0052]
【発明の効果】上述の如く、請求項１及び２記載の発明
によれば、送信バースト信号のランプタイム中に送信位
相と復調位相との位相差を測定して位相差をゼロとする
ように搬送波の移相量を設定した後、歪補償を開始する
ようにしたので、位相差を最も位相測定誤差の少ない状
態で測定できることから歪補償を正常に動作させること
ができ、これにより歪みの無い線形なバースト信号の送
信ができる。As described above, according to the first and second aspects of the invention, the phase difference between the transmission phase and the demodulation phase is measured during the ramp time of the transmission burst signal so that the phase difference becomes zero. Since the distortion compensation is started after setting the phase shift amount of the carrier wave, it is possible to operate the distortion compensation normally because the phase difference can be measured in the state in which the phase measurement error is the smallest, and thus, there is no distortion. A linear burst signal can be transmitted.
【００５３】また請求項３記載の発明によれば、歪補償
制御開始後徐々に制御のゲインを上げていくため、歪補
償制御開始時の過渡現象による信号の歪みを防止するこ
とができる。According to the third aspect of the invention, since the control gain is gradually increased after the distortion compensation control is started, it is possible to prevent signal distortion due to a transient phenomenon at the time of starting the distortion compensation control.
【００５４】請求項４及び５記載の発明によれば、１バ
ースト前の位相差測定結果をもとにバースト信号の非送
信期間に移相量の設定を行なうため、安定に歪補償を行
なうことができ、また請求項６及び７記載の発明によれ
ば、位相差の瞬時変動に追従することなく安定に歪補償
ができる。更に請求項８記載の発明によれば、第１バー
スト信号送信時の線形性を向上することができる等の特
長を有するものである。According to the fourth and fifth aspects of the invention, since the phase shift amount is set in the non-transmission period of the burst signal based on the phase difference measurement result of one burst before, the distortion compensation can be stably performed. According to the sixth and seventh aspects of the present invention, the distortion compensation can be stably performed without following the instantaneous fluctuation of the phase difference. Further, according to the invention described in claim 8, there is a feature that the linearity at the time of transmitting the first burst signal can be improved.
【図１】本発明の原理構成図である。FIG. 1 is a principle configuration diagram of the present invention.
【図２】スロット配置の一例を示す図である。FIG. 2 is a diagram showing an example of slot arrangement.
【図３】送信出力の説明図である。FIG. 3 is an explanatory diagram of transmission output.
【図４】本発明の第１実施例のブロック図である。FIG. 4 is a block diagram of a first embodiment of the present invention.
【図５】移相制御器の構成図である。FIG. 5 is a configuration diagram of a phase shift controller.
【図６】移相器回転手段の構成図である。FIG. 6 is a configuration diagram of a phase shifter rotating means.
【図７】本発明の第１，第２実施例のタイミング関係図
である。FIG. 7 is a timing relationship diagram of the first and second embodiments of the present invention.
【図８】本発明の第２実施例のブロック図である。FIG. 8 is a block diagram of a second embodiment of the present invention.
【図９】歪補償回路の一実施例の構成図である。FIG. 9 is a configuration diagram of an embodiment of a distortion compensation circuit.
【図１０】本発明の第３実施例のタイミング関係図であ
る。FIG. 10 is a timing relationship diagram of the third embodiment of the present invention.
【図１１】本発明の第４，第５実施例のタイミング関係
図である。FIG. 11 is a timing relationship diagram of the fourth and fifth embodiments of the present invention.
【図１２】本発明の第６実施例のタイミング関係図であ
る。FIG. 12 is a timing diagram of the sixth embodiment of the present invention.
【図１３】本発明の第７実施例のタイミング関係図であ
る。FIG. 13 is a timing diagram of the seventh embodiment of the present invention.
【図１４】本発明の第８実施例のタイミング関係図であ
る。FIG. 14 is a timing diagram of an eighth embodiment of the present invention.
【図１５】従来回路の一例のブロック図である。FIG. 15 is a block diagram of an example of a conventional circuit.
【図１６】歪補償回路の一例の構成図である。FIG. 16 is a configuration diagram of an example of a distortion compensation circuit.
【図１７】歪補償回路の他の例の構成図である。FIG. 17 is a configuration diagram of another example of the distortion compensation circuit.
３０ 信号処理部 ３１ 歪補償回路 ３２ 直交変調器 ３３ 電力増幅器 ３４ 直交復調器 ３５ 位相差測定手段 ３６ 移相手段 ４０ 搬送波発振器 ４２ 移相制御器 ４３，４６ 移相器回転手段 ５１，５２ 減算器 ５３，５４ 可変利得増幅器 ５５，５６ 加算器 30 signal processor 31 distortion compensation circuit 32 quadrature modulator 33 power amplifier 34 quadrature demodulator 35 phase difference measuring means 36 phase shift means 40 carrier wave oscillator 42 phase shift controller 43,46 phase shifter rotation means 51,52 subtractor 53 , 54 Variable gain amplifier 55, 56 Adder
───────────────────────────────────────────────────── フロントページの続き (72)発明者 久保 徳郎 神奈川県川崎市中原区上小田中1015番地 富士通株式会社内 (72)発明者 高野 健 神奈川県川崎市中原区上小田中1015番地 富士通株式会社内 ─────────────────────────────────────────────────── ─── Continued Front Page (72) Inventor Tokuro Kubo 1015 Kamiodanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture, Fujitsu Limited (72) Inventor Ken Ken Takano, 1015, Kamedotachu, Nakahara-ku, Kawasaki City, Kanagawa Prefecture, Fujitsu Limited
Claims (8)
１）を通して直交変調器（３２）へ入力し、該直交変調
器（３２）の出力直交変調波を電力増幅器（３３）で増
幅した後、時分割多重方式のバースト信号として出力す
ると共に直交復調器（３４）で復調し、その復調信号を
前記歪補償回路（３１）に帰還入力してベースバンドの
ベクトル座標で歪みを補償する線形送信回路において、 前記バースト信号の立ち上がり時間中に前記歪補償回路
（３１）に入力されるベースバンド信号の位相と前記復
調信号の位相との位相差を測定する位相差測定手段（３
５）と、 該位相差測定手段（３５）による位相差測定後に該測定
位相差をゼロにするように、前記直交復調器（３４）の
復調搬送波と前記直交変調器（３２）の変調搬送波との
相対位相を移相調整する移相手段（３６）とを有するこ
とを特徴とする線形送信回路。1. A distortion compensating circuit (3) for transmitting a baseband signal.
1) is input to a quadrature modulator (32), the quadrature modulated wave output from the quadrature modulator (32) is amplified by a power amplifier (33), and then output as a burst signal of a time division multiplexing system and a quadrature demodulator. A linear transmission circuit that demodulates at (34) and feeds back the demodulated signal to the distortion compensation circuit (31) to compensate for distortion at baseband vector coordinates, wherein the distortion compensation circuit is provided during the rising time of the burst signal. Phase difference measuring means (3) for measuring the phase difference between the phase of the baseband signal input to (31) and the phase of the demodulated signal.
5) and a demodulation carrier of the quadrature demodulator (34) and a modulation carrier of the quadrature modulator (32) so that the measured phase difference becomes zero after the phase difference measurement means (35) measures the phase difference. And a phase shift means (36) for performing phase shift adjustment of the relative phase of the linear transmission circuit.
波又は前記変調搬送波を移相することを特徴とする請求
項１記載の線形送信回路。2. The linear transmission circuit according to claim 1, wherein the phase shift means shifts the demodulated carrier wave or the modulated carrier wave.
測定手段（３５）による位相差測定中は歪補償のフィー
ドバック量を小とし、該位相差測定終了後に該フィード
バック量を徐々に大とする手段（５３，５４，３０）を
有することを特徴とする請求項２記載の線形送信回路。3. The distortion compensating circuit (31) reduces the feedback amount of distortion compensation during the phase difference measurement by the phase difference measuring means (35), and gradually increases the feedback amount after the phase difference measurement is completed. 3. The linear transmission circuit according to claim 2, further comprising means (53, 54, 30).
最初に出力される第１バースト信号送信中は歪補償動作
を行なわず、第２バースト信号以降歪補償動作を行な
い、前記移相手段（３６）は１バースト信号期間前に前
記位相差測定手段（３５）により測定された位相差に基
づく移相量が設定されることを特徴とする請求項１記載
の線形送信回路。4. The distortion compensating circuit (31) does not perform the distortion compensating operation during the transmission of the first burst signal that is first output after the start of transmission, but performs the distortion compensating operation after the second burst signal, and the phase shift is performed. The linear transmission circuit according to claim 1, wherein the means (36) sets a phase shift amount based on the phase difference measured by the phase difference measuring means (35) one burst signal period before.
波又は前記変調搬送波を前記設定された移相量だけ移相
することを特徴とする請求項４記載の線形送信回路。5. The linear transmission circuit according to claim 4, wherein the phase shift means (36) shifts the demodulation carrier wave or the modulation carrier wave by the set phase shift amount.
ースト信号以降の各バースト信号毎に、前記位相差を複
数回測定し、その複数回の測定値の平均値に応じた移相
量を次のバースト信号入力前に前記移相手段（３６）に
設定することを特徴とする請求項１乃至５のうちいずれ
か一項記載の線形送信回路。6. The phase difference measuring means (35) measures the phase difference a plurality of times for each burst signal after the second burst signal, and shifts the phase according to an average value of the plurality of measured values. 6. A linear transmission circuit as claimed in any one of claims 1 to 5, characterized in that the quantity is set in the phase shifting means (36) before the next burst signal is input.
ースト信号以降の前記移相量の設定を、それ以前の複数
のバースト信号の各々で測定した位相差の平均値に基づ
いて行なうことを特徴とする請求項４又は５記載の線形
送信回路。7. The phase difference measuring means (35) sets the phase shift amount after the third burst signal based on the average value of the phase differences measured for each of the plurality of burst signals before that. The linear transmission circuit according to claim 4 or 5, characterized in that:
１バースト信号送信時は第２バースト信号以降の送信時
に比し良好にすることを特徴とする請求項１記載の線形
送信回路。8. The linear transmission circuit according to claim 1, wherein the linearity of the power amplifier (33) is improved during transmission of the first burst signal compared to during transmission of the second burst signal and thereafter.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP19117092A JPH0637831A (en) | 1992-07-17 | 1992-07-17 | Linear transmission circuit |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP19117092A JPH0637831A (en) | 1992-07-17 | 1992-07-17 | Linear transmission circuit |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| JPH0637831A true JPH0637831A (en) | 1994-02-10 |
Family
ID=16270068
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP19117092A Withdrawn JPH0637831A (en) | 1992-07-17 | 1992-07-17 | Linear transmission circuit |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPH0637831A (en) |
Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2002057733A (en) * | 2000-08-01 | 2002-02-22 | Samsung Electronics Co Ltd | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
| US7289776B2 (en) | 2002-10-01 | 2007-10-30 | Matsushita Electric Industrial Co., Ltd. | Nonlinear distortion compensation circuit with feedback and baseband reference signal that are phase and amplitude controllable |
| US7305472B2 (en) | 1996-06-03 | 2007-12-04 | Microsoft Corporation | Method for downloading a web page to a client for efficient display on a television screen |
| US7523399B2 (en) | 1996-06-03 | 2009-04-21 | Microsoft Corporation | Downloading software from a server to a client |
| US7688914B2 (en) | 2005-08-17 | 2010-03-30 | Fujitsu Limited | Distortion compensation in wireless digital communication |
| JP4637331B2 (en) * | 2000-08-01 | 2011-02-23 | 三星電子株式会社 | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
| JP4703874B2 (en) * | 2001-03-22 | 2011-06-15 | 三星電子株式会社 | Nonlinear distortion compensation method and nonlinear distortion compensation circuit |
-
1992
- 1992-07-17 JP JP19117092A patent/JPH0637831A/en not_active Withdrawn
Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US7305472B2 (en) | 1996-06-03 | 2007-12-04 | Microsoft Corporation | Method for downloading a web page to a client for efficient display on a television screen |
| US7523399B2 (en) | 1996-06-03 | 2009-04-21 | Microsoft Corporation | Downloading software from a server to a client |
| JP2002057733A (en) * | 2000-08-01 | 2002-02-22 | Samsung Electronics Co Ltd | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
| JP4637331B2 (en) * | 2000-08-01 | 2011-02-23 | 三星電子株式会社 | Nonlinear distortion compensation circuit and nonlinear distortion compensation method |
| JP4703874B2 (en) * | 2001-03-22 | 2011-06-15 | 三星電子株式会社 | Nonlinear distortion compensation method and nonlinear distortion compensation circuit |
| US7289776B2 (en) | 2002-10-01 | 2007-10-30 | Matsushita Electric Industrial Co., Ltd. | Nonlinear distortion compensation circuit with feedback and baseband reference signal that are phase and amplitude controllable |
| US7688914B2 (en) | 2005-08-17 | 2010-03-30 | Fujitsu Limited | Distortion compensation in wireless digital communication |
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Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| A300 | Withdrawal of application because of no request for examination |
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