JPH04322191A - Controller for synchronous motor - Google Patents
Controller for synchronous motorInfo
- Publication number
- JPH04322191A JPH04322191A JP3088148A JP8814891A JPH04322191A JP H04322191 A JPH04322191 A JP H04322191A JP 3088148 A JP3088148 A JP 3088148A JP 8814891 A JP8814891 A JP 8814891A JP H04322191 A JPH04322191 A JP H04322191A
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- Prior art keywords
- field current
- current
- field
- current command
- magnetic flux
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- Granted
Links
- 230000001360 synchronised effect Effects 0.000 title claims abstract description 27
- 230000004907 flux Effects 0.000 claims abstract description 35
- 230000003313 weakening effect Effects 0.000 claims description 11
- 238000004804 winding Methods 0.000 claims description 5
- 230000005284 excitation Effects 0.000 claims description 4
- 238000010586 diagram Methods 0.000 description 9
- 238000001514 detection method Methods 0.000 description 5
- 230000000694 effects Effects 0.000 description 5
- 238000000034 method Methods 0.000 description 5
- 239000002131 composite material Substances 0.000 description 1
- 230000007423 decrease Effects 0.000 description 1
- 239000004065 semiconductor Substances 0.000 description 1
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- Control Of Ac Motors In General (AREA)
Abstract
Description
［発明の目的］ [Purpose of the invention]
【０００１】0001
【産業上の利用分野】本発明は同期電動機をサイクロコ
ンバータやインバータ等の可変電圧、可変周波数電源で
駆動する際の制御装置に関する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for driving a synchronous motor with a variable voltage, variable frequency power source such as a cycloconverter or an inverter.
【０００２】0002
【従来の技術】同期電動機を可変電圧、可変周波数で駆
動する方式には他励転流を用いた無整流子電動機やベク
トル制御を用いた駆動方式がある。このうちベクトル制
御は高性能な駆動が可能で、速い応答や精密な制御を要
する分野に適用されている。従来のベクトル制御方式の
例を図４に示す。2. Description of the Related Art Methods for driving a synchronous motor with variable voltage and variable frequency include a commutatorless motor using separately excited commutation and a drive method using vector control. Among these, vector control allows for high-performance driving and is applied to fields that require fast response and precise control. An example of a conventional vector control method is shown in FIG.
【０００３】図４は、同期電動機を自動界磁弱め方式を
用いて速度制御を行う場合の例で、同期電動機２の界磁
巻線２１は界磁用の電力変換器４により直流励磁され、
電力変換器１から駆動電力が供給される。速度制御部４
０は速度基準ω＊ と同期電動機２の速度検出信号ωを
比較してトルク基準Ｔ＊ を出力する。速度検出信号ω
は同期電動機２の回転角度を検出する位置検出器３の信
号θを微分器４１で微分して得られる。関数器４２は速
度検出信号ωに応じて磁束基準φ＊ を出力し自動界磁
弱めを可能にし、関数器４３は同期電動機の特性からφ
＊ に応じた界磁電流基準Ｉｆ ＊ を出力する。また
、除算器４４はＴ＊ をφ＊ で除算して電流基準ｉＴ
＊ を出力する。FIG. 4 shows an example in which the speed of a synchronous motor is controlled using an automatic field weakening method, in which the field winding 21 of the synchronous motor 2 is excited with direct current by a field power converter 4,
Drive power is supplied from the power converter 1. Speed control section 4
0 compares the speed reference ω* with the speed detection signal ω of the synchronous motor 2 and outputs the torque reference T*. Speed detection signal ω
is obtained by differentiating the signal θ from the position detector 3, which detects the rotation angle of the synchronous motor 2, using a differentiator 41. The function unit 42 outputs the magnetic flux reference φ* according to the speed detection signal ω to enable automatic field weakening, and the function unit 43 outputs the magnetic flux reference φ* based on the characteristics of the synchronous motor.
* Outputs the field current reference If * according to the field current reference If *. Further, the divider 44 divides T* by φ* to obtain the current reference iT.
Output *.
【０００４】同期電動機２の電機子電流と界磁電流は電
流検出器１１，１２で検出される。一方、位置検出器３
と位置演算器５では同期電動機の回転子磁極の位置を演
算する。電流検出器１１で検出した三相電機子電流は、
ｄｑ軸電流演算器１３で位置演算器５からの位置信号と
ともにｄｑ軸上での電流検出値ｉｄ ，ｉｑ として演
算される。この値と界磁電流検出値ｉｆ とから磁束演
算器６により電機子磁束が演算される。磁束演算器６で
は、検出電流と磁極位置から回転子磁極に並行なｄ軸磁
束φｄと、これに直行するｑ軸磁束φｑ 及び合成磁束
ベクトルの角度δを演算する。電流基準演算器７ではト
ルク電流指令ｉＴ ＊ と角度δとから磁束方向に直行
する電機子電流基準値ｉｄ ＊ ，ｉｑ ＊ を演算す
る。磁束演算器６は、図５に示すようにｄｑ軸上で表し
た同期電動機の電圧方程式から出発し、ｄｑ軸上の磁束
を演算する演算式をブロック図で表したものである。ｄ
軸磁束φｄ とｑ軸磁束φｑ は次式で示される。
φｄ ＝Ｌａｄ（１＋Ｔｄ２Ｓ）／（１＋
Ｔｄ１Ｓ）×（ｉｄ ＋ｉｆ ）
＋１ａ ×（ｉｄ ＋ｉｆ ）
（１）
φｑ ＝Ｌａｑ（１＋Ｔｑ２Ｓ）／（１＋Ｔｑ１
Ｓ）×ｉｑ ＋１ａ ×ｉｑ
（
２）The armature current and field current of the synchronous motor 2 are detected by current detectors 11 and 12. On the other hand, position detector 3
The position calculator 5 calculates the position of the rotor magnetic pole of the synchronous motor. The three-phase armature current detected by the current detector 11 is
The dq-axis current calculator 13 calculates current detection values id and iq on the dq-axes together with the position signal from the position calculator 5. The armature magnetic flux is calculated by the magnetic flux calculator 6 from this value and the field current detection value if. The magnetic flux calculator 6 calculates the d-axis magnetic flux φd parallel to the rotor magnetic poles, the q-axis magnetic flux φq orthogonal thereto, and the angle δ of the composite magnetic flux vector from the detected current and the magnetic pole position. The current reference calculator 7 calculates armature current reference values id*, iq* that are perpendicular to the magnetic flux direction from the torque current command iT* and the angle δ. As shown in FIG. 5, the magnetic flux calculator 6 starts from the voltage equation of the synchronous motor expressed on the dq axes, and represents in a block diagram an arithmetic expression for calculating the magnetic flux on the dq axes. d
Axial magnetic flux φd and q-axis magnetic flux φq are expressed by the following equations. φd = Lad(1+Td2S)/(1+
Td1S)×(id+if)
+1a × (id +if)
(1)
φq = Laq (1+Tq2S)/(1+Tq1
S)×iq +1a×iq
(
2)
【０００５】ここでＳはラプラス演算子、Ｌａｄ，
Ｌａｑはｄｑ軸の相互インダクタンス、１ａは電機子も
れインダクタンス、Ｔａ１，Ｔａ１はそれぞれｄｑ軸の
ダンパ抵抗と同期インダクタンスにかかわる時定数、Ｔ
ｄ２，Ｔｑ２はそれぞれｄｑ軸のダンパ抵抗とダンパも
れインダクタンスにかかわる時定数である。ｄ軸磁束φ
ｄ は、界磁電流ｉｆ とｄ軸電機子電流ｉｄ の和か
ら演算され、ｑ軸磁束φｑ はｑ軸電流ｉｑ から演算
される。δは磁束のｄ軸方向に対する角度で、
δ＝ｔａｎ−１（Φｑ ／Φｄ ）
（３）で演算される。図４の電流基準演算器７は、
トルク分電流基準ｉＴ ＊ と角度δから、ｄｑ軸の電
流基準ｉｄ ＊ ，ｉｑ ＊ を次式により演算する。
ｉｄ ＊ ＝−ｉＴ ＊ ・ｓｉｎδ
（４） ｉｑ ＊ ＝ ｉＴ ＊
・ｃｏｓδ
（５）[0005] Here, S is the Laplace operator, Lad,
Laq is the mutual inductance of the dq-axes, 1a is the armature leakage inductance, Ta1 and Ta1 are the time constants related to the damper resistance and synchronous inductance of the dq-axes, respectively, and T
d2 and Tq2 are time constants related to the dq-axis damper resistance and damper leakage inductance, respectively. d-axis magnetic flux φ
d is calculated from the sum of the field current if and the d-axis armature current id, and the q-axis magnetic flux φq is calculated from the q-axis current iq. δ is the angle of magnetic flux with respect to the d-axis direction, δ=tan-1(Φq /Φd)
Calculated in (3). The current reference calculator 7 in FIG.
From the torque component current reference iT* and the angle δ, the dq-axis current references id* and iq* are calculated by the following equation. id*=−iT*・sinδ
(4) iq * = iT *
・cos δ
(5)
【０００６】なわち、トル
ク分電流は演算された磁束と直行するように指令する。
演算された電流基準値は検出値と比較され、増幅器９に
よってｄｑ軸の電圧基準値ｖｄ＊ ，ｖｑ ＊ として
出力される。増幅器９は通常比例積分器により構成され
る。この電圧基準値は位置演算器５の出力との合成で三
相巻線電圧指令ｖｕ ＊ ，ｖｖ ＊，ｖｗ ＊ を得
る。この値は回転電気角周波数で変化する交流量となり
、電力変換器１を介して同期電動機２に加えられる。That is, the torque component current is instructed to run orthogonally to the calculated magnetic flux. The calculated current reference value is compared with the detected value and outputted by the amplifier 9 as dq-axis voltage reference values vd*, vq*. Amplifier 9 is usually constituted by a proportional integrator. This voltage reference value is combined with the output of the position calculator 5 to obtain three-phase winding voltage commands vu*, vv*, vw*. This value becomes an alternating current amount that changes with the rotational electrical angular frequency, and is applied to the synchronous motor 2 via the power converter 1.
【０００７】一方同期電動機の界磁電流ｉｆ は基準値
ｉｆ ＊ と比較され、その偏差に応じて増幅器１０に
より界磁電圧基準ｖｆ ＊ を得て、界磁用電力変換器
４を介して制御される。On the other hand, the field current if of the synchronous motor is compared with a reference value if*, and according to the deviation, a field voltage reference vf* is obtained by an amplifier 10, which is controlled via a field power converter 4. Ru.
【０００８】このように、電機子電流と界磁電流とから
ｄｑ軸上の磁束を演算し、電機子のトルク分電流を、演
算した磁束と直行するように流すことにより、同一の電
機子電流でも高いトルクが得られる。なお、ベクトル制
御方式については、Ｂ．Ｋ．Ｂｏｓｅ著Ｐｏｗｅｒ
Ｅｌｅｃｔｒｏｎｉｃｓ ａｎｄ ＡＣ Ｄｒｉ
ｖｅｓにも記載されている。In this way, by calculating the magnetic flux on the d and q axes from the armature current and the field current, and by flowing the armature torque current perpendicular to the calculated magnetic flux, the same armature current However, high torque can be obtained. Regarding the vector control method, see B. K. Power by Bose
Electronics and AC Dri
It is also described in ves.
【０００９】[0009]
【発明が解決しようとする課題】弱め界磁範囲での速度
制御では、電機子電圧を一定に保ち界磁電流により速度
制御を行うので、弱め界磁による速度制御範囲が広い場
合には界磁電流が非常に小さくなる。例えば弱め界磁範
囲が１：５の場合は、定格時の電流に対して１／５以下
の界磁電流となる。このような領域で大きな負荷変動が
発生すると電機子電流が大きく変化し電機子反作用によ
り界磁電流に大きな影響が生じる。この場合、図６に示
すように界磁電流ｉｆ が減少する方向に電機子反作用
が生じたとき、電動機速度が不安定になる現象が生じる
。[Problem to be solved by the invention] In speed control in the field weakening range, the armature voltage is kept constant and speed control is performed using the field current. The current becomes very small. For example, if the field weakening range is 1:5, the field current will be 1/5 or less of the rated current. When a large load fluctuation occurs in such a region, the armature current changes greatly, and the armature reaction has a large effect on the field current. In this case, as shown in FIG. 6, when an armature reaction occurs in a direction in which the field current if decreases, a phenomenon occurs in which the motor speed becomes unstable.
【００１０】一般に、界磁回路は一方向の電流を流すよ
うに構成されているため、上記の電機子反作用が大きく
なり、界磁電流を零以下に抑えこむと、界磁電流は負の
電流を流せないため、界磁制御が不能となる。このため
磁束が基準通り確立できなくなり急峻な負荷変動に対し
、所定のトルクを出せないことにもなる。従って界磁範
囲の広い高速度では安定な制御ができない。[0010] In general, the field circuit is constructed to allow current to flow in one direction, so the armature reaction described above becomes large, and when the field current is suppressed below zero, the field current becomes a negative current. Since the current cannot flow, field control becomes impossible. As a result, the magnetic flux cannot be established according to the standard, and a predetermined torque cannot be produced in response to sudden load fluctuations. Therefore, stable control cannot be achieved at high speeds with a wide field range.
【００１１】また、そのエネルギーが大きい場合は、界
磁電流が負側に流れないため蓄積されて界磁過電圧とな
り界磁回路の半導体素子を破壊してしまう恐れがある。
従って弱め界磁範囲の広い高速度のシステムには適用で
きなかった。Furthermore, if the energy is large, the field current will not flow to the negative side and will accumulate, resulting in a field overvoltage that may destroy the semiconductor element of the field circuit. Therefore, it could not be applied to high-speed systems with a wide field weakening range.
【００１２】本発明は上記欠点を改良するためになされ
たもので、弱め界磁の広い制御範囲において大きな負荷
変動に対しても制御不能やトルク不足となることなく安
定に運転することのできる高性能の同期電動機の制御装
置を提供することを目的としている。
［発明の構成］The present invention has been made in order to improve the above-mentioned drawbacks, and is capable of stable operation without loss of control or torque shortage even with large load fluctuations in a wide control range of field weakening. The purpose is to provide a high performance synchronous motor control device. [Structure of the invention]
【００１３】[0013]
【課題を解決するための手段】本発明は、上記目的を達
成すめため、可変電圧可変周波数の電力を同期電動機に
供給する制御電源部と、該同期電動機の界磁巻線に可変
の界磁電流（ｉｆ ）を供給する励磁電源部と、該同期
電動機の電機子電流と回転子の位置から該回転子の磁極
と同方向（ｄ軸）及び直交方向（ｑ軸）のｄｑ軸電流成
分（ｉｄ ，ｉｑ ）を得る電流演算部と、該ｄｑ軸電
流成分（ｉｄ ，ｉｑ ）と該界磁電流（ｉｆ ）から
該同期電動機の磁束を模擬する磁束演算部と、速度指令
と該同期電動機の検出速度を比較して電流指令を得る速
度制御部と、該電流指令と該磁束演算部の出力からｄｑ
軸電流指令（ｉｄ ＊ ，ｉｑ＊ ）を得る電流指令演
算部と、該ｄｑ軸電流指令と該ｄｑ軸電流成分（ｉｄ
，ｉｑ ）の偏差に応じて該制御電源部に対する電圧指
令を得る電圧指令演算部と、該検出速度に応じて該励磁
電源部に対する界磁電流指令（ｉｆ ＊ ）を得る界磁
電流指令部を備え、自動弱め界磁により速度を制御する
装置において、該界磁電流指令（ｉｆ ＊ ）に対し部
分的あるいは全体的に増加した界磁電流指令（ｉｆ２＊
）を得る補償手段を設け、該界磁電流指令（ｉｆ２＊
）を実際の界磁電流指令とするように構成する。[Means for Solving the Problems] In order to achieve the above object, the present invention provides a control power supply unit that supplies variable voltage and variable frequency power to a synchronous motor, and a variable field power supply to a field winding of the synchronous motor. An excitation power supply section that supplies current (if), and dq-axis current components (d-axis) in the same direction (d-axis) and orthogonal direction (q-axis) to the magnetic poles of the rotor from the armature current of the synchronous motor and the rotor position. id, iq); a magnetic flux calculation unit that simulates the magnetic flux of the synchronous motor from the dq-axis current components (id, iq) and the field current (if); A speed control section that obtains a current command by comparing the detected speeds, and dq from the current command and the output of the magnetic flux calculation section.
a current command calculation unit that obtains the axis current command (id*, iq*);
, iq ), and a field current command unit that obtains a field current command (if *) for the excitation power supply according to the detected speed. In a device that controls speed by automatic field weakening, a field current command (if2*) that is partially or totally increased with respect to the field current command (if*) is used.
) is provided, and the field current command (if2*
) is the actual field current command.
【００１４】[0014]
【作用】上記補償手段により最小界磁電流を所定値に制
限し電機子反作用が生じたときにも界磁電流が零になら
ないようにする。[Operation] The compensation means limits the minimum field current to a predetermined value so that the field current does not become zero even when armature reaction occurs.
【００１５】[0015]
【実施例】図１は本発明の一実施例を示す構成図である
。DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a block diagram showing an embodiment of the present invention.
【００１６】図１において、界磁電流補償器３０が新に
設けられた要素である他の要素は図４で示したものと同
一である。界磁電流補償器３０は界磁電流指令ｉｆ ＊
を補償して実際の界磁電流指令ｉｆ２＊ を得るもの
で、その補償の例を図２に示す。この補償特性Ｃ１ ，
Ｃ２ は界磁電流補償器３０の入力である界磁電流基準
ｉｆ ＊を横軸にとり、出力ｉｆ２＊ を縦軸にした関
数を示している。補償特性Ｃ１ の場合In FIG. 1, the field current compensator 30 is a new element, but the other elements are the same as those shown in FIG. The field current compensator 30 receives the field current command if *
The actual field current command if2* is obtained by compensating for this, and an example of this compensation is shown in FIG. This compensation characteristic C1,
C2 represents a function in which the horizontal axis is the field current reference if*, which is the input of the field current compensator 30, and the vertical axis is the output if2*. In case of compensation characteristic C1
【００１７】界磁電流基準ｉｆ ＊ が最低界磁電流制
限値ｉｆＭＩＮより小さい場合は、ｉｆ２＊ をｉｆＭ
ＩＮの値に制限する。ｉｆ ＊ がｉｆＭＩＮより大き
い場合はｉｆ２＊ ＝ｉｆ ＊ となるように関数を決
定する。If the field current reference if* is smaller than the minimum field current limit value ifMIN, if2* is changed to ifM
Restrict to the value of IN. If if* is larger than ifMIN, a function is determined so that if2*=if*.
【００１８】この最低界磁電流制限値ｉｆＭＩＮは弱め
界磁領域において大きな負荷がかかっても電機子反作用
で、界磁電流が零以下とならないように設定するもので
ある。The minimum field current limit value ifMIN is set so that even if a large load is applied in the field weakening region, the field current will not fall below zero due to armature reaction.
【００１９】このように界磁電流ｉｆ ＊ が小さい領
域で最低界磁電流制限値ｉｆＭＩＮにより増加した界磁
電流ｉｆ は、磁束演算器６に入力され、図５のブロッ
ク図により内部相差角δが演算され、界磁電流の増加分
だけｄ軸磁束φｄ が増大し、内部相差角が開くことに
なり、電流基準演算器７を介してｄ軸電流ｉｄ ＊ が
増加させられる。The field current if increased by the minimum field current limit value ifMIN in the region where the field current if* is small as described above is input to the magnetic flux calculator 6, and the internal phase difference angle δ is calculated according to the block diagram of FIG. The d-axis magnetic flux φd increases by the increase in field current, the internal phase difference angle opens, and the d-axis current id* is increased via the current reference calculator 7.
【００２０】このｉｄ ＊ の増加分が減磁作用として
界磁電流ｉｆ ＊ の増加による磁束と相殺されるため
、磁束を増加することなく制御できる。このことにより
、出力電圧は一定に保たれるため電圧過大となることは
ない。Since this increase in id* cancels out the magnetic flux due to the increase in field current if* as a demagnetizing effect, it is possible to control the magnetic flux without increasing it. As a result, the output voltage is kept constant, so that the voltage does not become excessive.
【００２１】最低界磁電流制限値ｉｆＭＩＮを適当に設
定することにより、弱め界磁範囲の広い高速領域におい
ても電機子反作用により界磁電流が零となってならない
ため界磁制御が安定に行なえる。従って大きな負荷変動
に対しても安定な制御特性が実現できる。この様子を図
３に示す。実線は本実施例による特性で点線は従来の特
性を示す。また、補償特性Ｃ２ のように界磁電流ｉｆ
＊ に対し、最低界磁電流ｉｆＭＩＮをバイアスする
だけでも同様の効果が得られる。By appropriately setting the minimum field current limit value ifMIN, field current can be stably controlled even in a high-speed region with a wide field weakening range because the field current does not become zero due to armature reaction. Therefore, stable control characteristics can be achieved even with large load fluctuations. This situation is shown in FIG. The solid line shows the characteristics according to this embodiment, and the dotted line shows the conventional characteristics. Also, as shown in the compensation characteristic C2, the field current if
*A similar effect can be obtained by simply biasing the lowest field current ifMIN.
【００２２】[0022]
【発明の効果】本発明によれば、弱め界磁範囲の広い高
速運転において急峻な負荷変動が生じても制御不能やト
ルク不足となることなく安定した速度制御を行うことの
できる同期電動機の制御装置が得られる。[Effects of the Invention] According to the present invention, the synchronous motor can be controlled to perform stable speed control without becoming uncontrollable or lacking torque even if sudden load fluctuations occur during high-speed operation with a wide field weakening range. A device is obtained.
【図１】本発明の一実施例を示す構成図。FIG. 1 is a configuration diagram showing an embodiment of the present invention.
【図２】上記実施例の補償特性図。FIG. 2 is a compensation characteristic diagram of the above embodiment.
【図３】上記実施例の効果を説明するための特性図。FIG. 3 is a characteristic diagram for explaining the effects of the above embodiment.
【図４】従来装置の構成図。FIG. 4 is a configuration diagram of a conventional device.
【図５】磁束演算器のブロック図。FIG. 5 is a block diagram of a magnetic flux calculator.
【図６】従来装置の問題を説明するための特性図。FIG. 6 is a characteristic diagram for explaining a problem with a conventional device.
１…電力変換器 ２…同期電動機３
…位置検出器 ４…界磁用電力変換
器５…位置演算器 ６…磁束演算器
７…電流基準演算器 ９，１０…増幅器１１
，１２…電流検出器 １３…ｄｑ軸電流演算
器１４…座標変換器 ２１…界磁巻
線３０…界磁電流補償器 ４０…速度制御部
４１…微分器 ４２，４３
…関数器４４…除算器1...Power converter 2...Synchronous motor 3
...Position detector 4...Field power converter 5...Position calculator 6...Magnetic flux calculator 7...Current reference calculator 9, 10...Amplifier 11
, 12...Current detector 13...DQ-axis current calculator 14...Coordinate converter 21...Field winding 30...Field current compensator 40...Speed control section 41...Differentiator 42, 43
...Function unit 44...Divider
Claims (1)
機に供給する制御電源部と、該同期電動機の界磁巻線に
可変の界磁電流（ｉｆ ）を供給する励磁電源部と、該
同期電動機の電機子電流と回転子の位置から該回転子の
磁極と同方向（ｄ軸）及び直交方向（ｑ軸）のｄｑ軸電
流成分（ｉｄ ，ｉｑ）を得る電流演算部と、該ｄｑ軸
電流成分（ｉｄ ，ｉｑ ）と該界磁電流（ｉｆ ）か
ら該同期電動機の磁束を模擬する磁束演算部と、速度指
令と該同期電動機の検出速度を比較して電流指令を得る
速度制御部と、該電流指令と該磁束演算部の出力からｄ
ｑ軸電流指令（ｉｄ ＊ ，ｉｑ ＊ ）を得る電流指
令演算部と、該ｄｑ軸電流指令と該ｄｑ軸電流成分（ｉ
ｄ ，ｉｑ ）の偏差に応じて該制御電源部に対する電
圧指令を得る電圧指令演算部と、該検出速度に応じて該
励磁電源部に対する界磁電流指令（ｉｆ ＊ ）を得る
界磁電流指令部を備え、自動弱め界磁により速度を制御
する装置において、該界磁電流指令（ｉｆ ＊ ）に対
し部分的あるいは全体的に増加した界磁電流指令（ｉｆ
２＊ ）を得る補償手段を設け、該界磁電流指令（ｉｆ
２＊ ）を実際の界磁電流指令とすることを特徴とする
同期電動機の制御装置。1. A control power supply unit that supplies variable voltage and variable frequency power to a synchronous motor, an excitation power supply unit that supplies a variable field current (if) to a field winding of the synchronous motor, and the synchronous motor. a current calculation unit that obtains dq-axis current components (id, iq) in the same direction (d-axis) and orthogonal direction (q-axis) to the magnetic poles of the rotor from the armature current and the rotor position; a magnetic flux calculation unit that simulates the magnetic flux of the synchronous motor from the components (id, iq) and the field current (if); a speed control unit that obtains a current command by comparing the speed command and the detected speed of the synchronous motor; d from the current command and the output of the magnetic flux calculation section
A current command calculation unit that obtains the q-axis current command (id*, iq*), and a current command calculation unit that obtains the dq-axis current command and the dq-axis current component (i
a voltage command calculation unit that obtains a voltage command for the control power supply unit according to the deviation of d, iq), and a field current command unit that obtains a field current command (if*) for the excitation power supply unit according to the detected speed. In a device that controls speed by automatic field weakening, a field current command (if *) that is partially or totally increased with respect to the field current command (if
2*) is provided, and the field current command (if
2*) as an actual field current command.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP3088148A JP2856564B2 (en) | 1991-04-19 | 1991-04-19 | Control device for synchronous motor |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP3088148A JP2856564B2 (en) | 1991-04-19 | 1991-04-19 | Control device for synchronous motor |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPH04322191A true JPH04322191A (en) | 1992-11-12 |
| JP2856564B2 JP2856564B2 (en) | 1999-02-10 |
Family
ID=13934851
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP3088148A Expired - Lifetime JP2856564B2 (en) | 1991-04-19 | 1991-04-19 | Control device for synchronous motor |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JP2856564B2 (en) |
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US7455133B2 (en) | 2005-03-14 | 2008-11-25 | Hitachi, Ltd. | Electric four-wheel drive vehicle and control unit for same |
| JP2009183051A (en) * | 2008-01-30 | 2009-08-13 | Mitsubishi Electric Corp | Controller of synchronous machine |
| CN103988420A (en) * | 2011-11-30 | 2014-08-13 | 三菱电机株式会社 | Inverter device for electric vehicle |
| CN108322125A (en) * | 2018-01-25 | 2018-07-24 | 上海电气富士电机电气技术有限公司 | A kind of torque response control method of synchronous motor |
-
1991
- 1991-04-19 JP JP3088148A patent/JP2856564B2/en not_active Expired - Lifetime
Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US7455133B2 (en) | 2005-03-14 | 2008-11-25 | Hitachi, Ltd. | Electric four-wheel drive vehicle and control unit for same |
| JP2009183051A (en) * | 2008-01-30 | 2009-08-13 | Mitsubishi Electric Corp | Controller of synchronous machine |
| CN103988420A (en) * | 2011-11-30 | 2014-08-13 | 三菱电机株式会社 | Inverter device for electric vehicle |
| KR20140101386A (en) * | 2011-11-30 | 2014-08-19 | 미쓰비시덴키 가부시키가이샤 | Inverter device for electric vehicle |
| EP2787623A4 (en) * | 2011-11-30 | 2015-09-30 | Mitsubishi Electric Corp | Inverter device for electric vehicle |
| US9350283B2 (en) | 2011-11-30 | 2016-05-24 | Mitsubishi Electric Corporation | Inverter device for electric vehicle |
| CN108322125A (en) * | 2018-01-25 | 2018-07-24 | 上海电气富士电机电气技术有限公司 | A kind of torque response control method of synchronous motor |
Also Published As
| Publication number | Publication date |
|---|---|
| JP2856564B2 (en) | 1999-02-10 |
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