JPH04121061A - High frequency power supply circuit for two load circuits - Google Patents
High frequency power supply circuit for two load circuitsInfo
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- JPH04121061A JPH04121061A JP2237089A JP23708990A JPH04121061A JP H04121061 A JPH04121061 A JP H04121061A JP 2237089 A JP2237089 A JP 2237089A JP 23708990 A JP23708990 A JP 23708990A JP H04121061 A JPH04121061 A JP H04121061A
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- load
- power
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- impedance
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- 230000005540 biological transmission Effects 0.000 claims abstract description 16
- 239000003990 capacitor Substances 0.000 description 4
- 238000010586 diagram Methods 0.000 description 4
- 230000000694 effects Effects 0.000 description 3
- 230000001939 inductive effect Effects 0.000 description 2
- 230000007423 decrease Effects 0.000 description 1
- 238000010438 heat treatment Methods 0.000 description 1
- 230000006698 induction Effects 0.000 description 1
- 230000002265 prevention Effects 0.000 description 1
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- Inverter Devices (AREA)
- Supply And Distribution Of Alternating Current (AREA)
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Abstract
Description
【発明の詳細な説明】
〔産業上の利用分野〕
この発明は、互いに異なる固有の共振周波数を持つ２つ
の誘導加熱負荷回路やプラズマ発生用共振回路などに対
し、１台のインバータから個別に高周波電力を供給する
電力供給回路に関する。[Detailed Description of the Invention] [Field of Industrial Application] The present invention provides high-frequency transmission from one inverter to two induction heating load circuits, plasma generation resonant circuits, etc. that have different unique resonance frequencies. The present invention relates to a power supply circuit that supplies electric power.
第２図は１合の高周波インバータから２つの共振回路に
高周波電力を供給する従来例を示す回路図である。同図
において、１は高周波（数ＭＨｚのオーダ）の交流電力
を出力する電子形インバータ、６は交流電力を所定の負
荷回路に供給するための切換スイッチで、この切換スイ
ッチ６には配線８Ａを介して並列共振回路２、および配
線８Ｂを介して直列共振回路３が接続され、スイッチ６
の切り換えにより各負荷にそれぞれ高周波電力が供給さ
れる。FIG. 2 is a circuit diagram showing a conventional example in which high-frequency power is supplied from one high-frequency inverter to two resonant circuits. In the figure, 1 is an electronic inverter that outputs high frequency (on the order of several MHz) AC power, 6 is a changeover switch for supplying AC power to a predetermined load circuit, and 8A wiring is connected to this changeover switch 6. The parallel resonant circuit 2 is connected through the switch 6, and the series resonant circuit 3 is connected through the wiring 8B.
High frequency power is supplied to each load by switching.
電圧形インバータ１から高周波の交流電力が供給される
負荷は一般に誘導性であることから、図示のように負荷
リアクタンス２１と負荷抵抗２２の直列回路、および負
荷リアクタンス３１と負荷抵抗３２の直列回路で構成さ
れるが、負荷回路２ではこの直列回路に共振コンデンサ
２３を並列に接続することで並列共振回路が形成され、
負荷回路３ではその直列回路に共振コンデンサ３３を直
列に接続することで直列共振回路が形成されている。そ
して、これら共振負荷回路に対して高周波電力を高い効
率で供給するため、電圧形インバータ１は一般に、スイ
ッチ６を介して接続される共振回路の共振周波数または
その近傍の周波数で運転される。Since the load to which high-frequency AC power is supplied from the voltage source inverter 1 is generally inductive, a series circuit of a load reactance 21 and a load resistance 22, and a series circuit of a load reactance 31 and a load resistance 32 are used as shown in the figure. However, in the load circuit 2, a parallel resonant circuit is formed by connecting the resonant capacitor 23 in parallel to this series circuit,
In the load circuit 3, a series resonant circuit is formed by connecting a resonant capacitor 33 in series to the series circuit. In order to supply high-frequency power to these resonant load circuits with high efficiency, the voltage source inverter 1 is generally operated at the resonant frequency of the resonant circuit connected via the switch 6 or a frequency near the resonant frequency.
第２図の如き回路では、並列共振回路２はその共振周波
数におけるインピーダンスをＺＴい高調波に対するイン
ピーダンスをＺＮＩとすれば、−ｉにＺＴ！＞＞ＺＮＩ
である。一方、電圧形インバータｌから出力される電圧
には多くの高調波成分が含まれているので、この電圧形
インバータ１の出力側を配線８Ａを介して直接並列共振
回路２に接続すると、高調波に対するインピーダンスＺ
８□が小であることから、多くの高調波電流が並列共振
回路２に流れ、電圧形インバータ１が過電流となる。In the circuit shown in FIG. 2, the parallel resonant circuit 2 has an impedance at the resonant frequency of ZT and an impedance of harmonics of ZNI, then -i has ZT! >>ZNI
It is. On the other hand, since the voltage output from voltage source inverter 1 contains many harmonic components, if the output side of voltage source inverter 1 is directly connected to parallel resonant circuit 2 via wiring 8A, harmonics will be generated. Impedance Z for
Since 8□ is small, many harmonic currents flow into the parallel resonant circuit 2, causing an overcurrent in the voltage source inverter 1.
そこで、このような高調波電流が流れるのを抑制する目
的で、配線８Ａには直列リアクトル７Ａが挿入される。Therefore, in order to suppress the flow of such harmonic current, a series reactor 7A is inserted into the wiring 8A.
しかしながら、この直列リアクトル７Ａの挿入により、
運転時の電圧変動率が大となり、並列共振回路２に印加
される電圧が低下するという問題が生じる。さらに、配
線８Ａには配線インダクタンス７Ｂが存在しているので
、電圧形インバータ１と並列共振回路２とを接続するた
めの距離が長い場合、またはこの配線を流れる交流の周
波数が特に高い領域にある場合は、配線インダクタンス
７Ｂの影響が大となり、挿入された直列リアクトル７Ａ
を取り除いても全体のインダクタンス値が過大となって
並列共振回路２に印加される電圧が所定値以下になり、
電圧形インバータ１から充分な電力が供給できないとい
う問題が生じる。However, by inserting this series reactor 7A,
A problem arises in that the voltage fluctuation rate during operation increases, and the voltage applied to the parallel resonant circuit 2 decreases. Furthermore, since the wiring inductance 7B exists in the wiring 8A, when the distance for connecting the voltage source inverter 1 and the parallel resonant circuit 2 is long, or when the frequency of the alternating current flowing through this wiring is particularly high. In this case, the influence of the wiring inductance 7B becomes large, and the inserted series reactor 7A
Even if it is removed, the overall inductance value becomes excessive and the voltage applied to the parallel resonant circuit 2 falls below a predetermined value.
A problem arises in that sufficient power cannot be supplied from the voltage source inverter 1.
一方、配ｍ８Ｂにも配線インダクタンス７Ｃが存在して
いるので、電圧形インバータｌと直列共振回路３とを接
続するための距離が長い場合、またはこの配線を流れる
交流の周波数が特に高い領域にある場合は、配線インダ
クタンス値が増加し、このインダクタンス値が負荷リア
クタンスのインダクタンス値に加算されることになるの
で、インバータ出力端子から直列共振負荷側を見た回路
の共振周波数が、直列共振回路３の共振周波数と異なっ
てしまうという問題や、直列共振回路３に印加される電
子が所定値以下になり、電圧形インバータ１から充分な
電力が供給できないという問題が生じる。On the other hand, wiring inductance 7C also exists in wiring m8B, so when the distance for connecting voltage source inverter l and series resonant circuit 3 is long, or when the frequency of alternating current flowing through this wiring is particularly high. In this case, the wiring inductance value increases and this inductance value is added to the inductance value of the load reactance, so the resonant frequency of the circuit viewed from the inverter output terminal to the series resonant load side becomes the same as that of the series resonant circuit 3. There arises a problem that the frequency differs from the resonant frequency, and that the number of electrons applied to the series resonant circuit 3 becomes less than a predetermined value, so that sufficient power cannot be supplied from the voltage source inverter 1.
また、電圧形インバータ１から供給する電力を２つの負
荷回路のうちの一方から他方に切り換える場合、スイッ
チ６による切り換えが必要になること、さらには電圧形
インバータｌから各負荷回路に高周波電力を供給するた
めの配線が個別に必要で、そのためのスペースを確保し
なければならない、などの問題がある。In addition, when switching the power supplied from the voltage source inverter 1 from one of the two load circuits to the other, switching using the switch 6 is required, and furthermore, high frequency power is supplied from the voltage source inverter 1 to each load circuit. There are problems such as the need for separate wiring for this and the need to secure space for it.
したがって、この発明の課題はこのような様々な問題を
解消し、１つのインバータ回路から、２つの共振負荷回
路に高周波電力を高効率に供給し得る回路を提供するこ
とにある。Therefore, an object of the present invention is to provide a circuit that can solve these various problems and supply high-frequency power to two resonant load circuits from one inverter circuit with high efficiency.
〔課題を解決するための手段〕
高周波の交流電力を出力する電圧形インバータから、第
１の固有周波数では特定のインピーダンスを持ち、それ
以外の周波数ではこの特定インピーダンスよりも小さな
値のインピーダンスを持つ第１の負荷と、前記第１の固
有周波数とは異なる第２の固有周波数では特定のインピ
ーダンスを持ち、それ以外の周波数ではこの特定インピ
ーダンスよりも大きな値のインピーダンスを持つ第２の
負荷とにそれぞれ高周波電力を供給するため、前記電圧
形インバータから前記第１の負荷へ電力を供給する電力
伝送線路には、この線路内における第１の固有周波数の
伝搬波長の１／４またはその奇数倍に相当する線路長を
持つ第１の分布定数線路を、また前記電圧形インバータ
から前記第２の負荷へ電力を供給する電力伝送線路には
、前記第１の負荷と直列に接続され、かつ前記第１の分
布定数線路長との和がこれらの線路内における第２の固
有周波数の伝搬波長の１／２波長またはその整数倍に相
当する線路長を持つ第２の分布定数線路を設け、前記第
１の負荷への電力供給は前記電圧形インバータの動作周
波数を前言己第１の固有周波数またはその近傍の周波数
にして行ない、また前記第２の負荷への電力供給は前記
電圧形インバータの動作周波数を前記第２の固有周波数
またはその近傍の周波数にしてそれぞれ行なう。[Means for solving the problem] A voltage source inverter that outputs high-frequency AC power has a specific impedance at the first natural frequency and a smaller impedance than this specific impedance at other frequencies. 1 and a second load that has a specific impedance at a second natural frequency different from the first natural frequency and has an impedance larger than this specific impedance at other frequencies. In order to supply electric power, a power transmission line for supplying electric power from the voltage source inverter to the first load is provided with a transmission line having a wavelength corresponding to 1/4 of the propagation wavelength of the first natural frequency in the line or an odd multiple thereof. A first distributed constant line having a line length and a power transmission line for supplying power from the voltage source inverter to the second load are connected in series with the first load and have the first A second distributed constant line having a line length whose sum with the distributed constant line length corresponds to 1/2 wavelength of the propagation wavelength of the second natural frequency in these lines or an integral multiple thereof, Power is supplied to the load by setting the operating frequency of the voltage source inverter to the first natural frequency or a frequency close to the first natural frequency, and supplying power to the second load by setting the operating frequency of the voltage source inverter to the first natural frequency or a frequency close to the first natural frequency. This is carried out at the second natural frequency or at a frequency near the second natural frequency.
伝搬波長の１／４またはその整数倍の長さを持つ第１の
分布定数線路において、その終端にインピーダンスＺＬ
を接続し、分布定数線路の特性インピーダンスをＺ。と
すると、送電端インピーダンスＺ３は、
Ｚ ｓ −Ｚ。”／ＺＬ
となり、終端が短絡状態の場合は終端に適当な大きさの
インピーダンス負荷を接続した場合に比べて極端に大き
な値を持つことが一般に良く知られている。そのため、
前記第１の負荷回路を接続すると、その固有周波数にお
ける送電端インピーダンスは適切な値となる一方、高調
波周波数成分における送電端インピーダンスは極端に大
きくなるので、電圧形インバータの出力電圧に含まれる
高調波電圧成分に伴うその出力高調波電流は極小値に抑
制され、本来伝送すべき基本周波数電圧成分に伴う基本
波電流成分のみを第１の負荷回路に供給することができ
る。しかもこのとき、第１の負荷回路と直列に接続され
る第２の分布定数線路も第１の伝搬波長の１／４の奇数
倍になり、かつ第２の負荷回路のインピーダンスＺＬ２
は大きな値になるので、第２の送電端インピーダンスＺ
ｓ２は第１の固有周波数およびその高調波周波数成分に
おいて非常に低インピーダンスになる。第１の負荷回路
のインピーダンスＺＬと第２の送電端インピーダンス７
５２は直列に接続されるので、第２の負荷回路には電力
の供給は殆ど行なわれない。In the first distributed constant line having a length of 1/4 of the propagation wavelength or an integral multiple thereof, an impedance ZL is placed at the end of the first distributed constant line.
Connect the characteristic impedance of the distributed constant line to Z. Then, the transmission end impedance Z3 is Z s −Z. ”/ZL, and it is generally well known that when the termination is short-circuited, the value is extremely large compared to when an appropriately sized impedance load is connected to the termination. Therefore,
When the first load circuit is connected, the transmission end impedance at its natural frequency becomes an appropriate value, but the transmission end impedance at harmonic frequency components becomes extremely large. The output harmonic current accompanying the wave voltage component is suppressed to a minimum value, and only the fundamental wave current component accompanying the fundamental frequency voltage component to be transmitted can be supplied to the first load circuit. Moreover, at this time, the second distributed constant line connected in series with the first load circuit also has an odd number multiple of 1/4 of the first propagation wavelength, and the impedance ZL2 of the second load circuit
has a large value, so the second transmission end impedance Z
s2 has a very low impedance at the first natural frequency and its harmonic frequency components. Impedance ZL of the first load circuit and second transmission end impedance 7
52 are connected in series, almost no power is supplied to the second load circuit.
一方、伝搬波長の１／２またはその整数倍の長さを持つ
第１．第２の分布定数線路の直列線路において、その終
端にインピーダンスＺＬを接続し、分布定数線路の特性
インピーダンスを７８とすると、送電端インピーダンス
Ｚ、は、
ＺＳ＝ＺＬ
となることが一般に知られている。そのため、前記第２
の負荷回路を終端に接続すると、これはあたかも送電端
すなわちインバータの出力端に直接接続したのと同様と
なる。つまり、第２の固有周波数における送電端インピ
ーダンスは適切な値となる一方、その高調波周波数成分
における送電端インピーダンスは極端に大きくなるので
、電圧形インバータの出力電圧に含まれる高調波電圧成
分に伴う出力高調波電流は極小値に抑制され、本来伝送
すべき基本周波数電圧成分に伴う基本波電流成分を第２
の負荷回路に供給することができる。On the other hand, the first . It is generally known that if an impedance ZL is connected to the end of the series line of the second distributed constant line and the characteristic impedance of the distributed constant line is 78, the transmission end impedance Z is ZS=ZL. . Therefore, the second
When the load circuit of In other words, while the transmitting end impedance at the second natural frequency is an appropriate value, the transmitting end impedance at its harmonic frequency component becomes extremely large. The output harmonic current is suppressed to a minimum value, and the fundamental wave current component accompanying the fundamental frequency voltage component that should originally be transmitted is
load circuit.
しかもこのとき、第１の分布定数線路と第２の分布定数
線路との間に接続される前記第１の負荷回路のインピー
ダンスは充分に小さな値となるので、第１の負荷回路に
は電力の供給は殆ど行なわれない。Moreover, at this time, the impedance of the first load circuit connected between the first distributed constant line and the second distributed constant line becomes a sufficiently small value, so that the first load circuit has a sufficiently low impedance. There is almost no supply.
第１図はこの発明の実施例を示す構成図でζ １は電圧
形インバータ、２は並列共振負荷回路、３は直列共振負
荷回路、４は第１の分布定数形量軸線路、５は第２の分
布定数形量軸線路である。FIG. 1 is a configuration diagram showing an embodiment of the present invention. ζ 1 is a voltage source inverter, 2 is a parallel resonant load circuit, 3 is a series resonant load circuit, 4 is a first distributed constant type quantity axis line, and 5 is a first 2 distributed constant shape quantity axis line.
電圧形インバータ１はスイッチング素子、還流ダイオー
ドおよび逆流防止用ダイオード等から構成され、その高
周波交流出力端子ｕ、ｖは第１の分布定数形量軸線路４
の一方の中心ＭＡ＠子および被覆線端子に接続され、第
１の分布定数形同軸線路４の他方の中心線端子は並列共
振負荷回路２の一方の端子に接続される。また、第１の
分布定数形量軸線路４の他方の被覆線端子は第２の分布
定数形量軸線路５の被覆線端子に、第２の分布定数形量
軸線路５の一方の中心線端子は並列共振負荷回路２の他
方の端子に接続される。さらに、第２の分布定数形量軸
線路５の他方の中心線端子および被覆線端子は直列共振
負荷回路３の２つの端子にそれぞれ接続される。しかし
て、並列共振負荷回路２は第１の負荷リアクタンス２１
と第１の負荷抵抗２２からなる直列回路と第１の共振コ
ンデンサ２３との並列回路で構成されて第１の共振周波
数ＦＴ、を持ち、直列共振負荷回路３は第２の負荷リア
クタンス３１．第２の負荷抵抗３２および第２の共振コ
ンデンサ３３の直列回路で構成されて第２の共振周波数
ＦＴｚを持つ。The voltage source inverter 1 is composed of a switching element, a free wheel diode, a backflow prevention diode, etc., and its high frequency AC output terminals u and v are connected to a first distributed constant variable axis line 4.
The other center line terminal of the first distributed constant type coaxial line 4 is connected to one terminal of the parallel resonant load circuit 2. Further, the other covered wire terminal of the first distributed constant quantity axis line 4 is connected to the coated wire terminal of the second distributed constant quantity axis line 5, and one center line of the second distributed constant quantity axis line 5 is connected to the covered wire terminal of the second distributed constant quantity axis line 5. The terminal is connected to the other terminal of the parallel resonant load circuit 2. Further, the other center line terminal and coated wire terminal of the second distributed constant quantity axis line 5 are connected to two terminals of the series resonant load circuit 3, respectively. Therefore, the parallel resonant load circuit 2 has a first load reactance 21
The series resonant load circuit 3 has a first resonant frequency FT, and has a first resonant frequency FT. It is composed of a series circuit of a second load resistor 32 and a second resonant capacitor 33, and has a second resonant frequency FTz.
そして、第１の共振周波数Ｆア、は第２の共振周波数Ｆ
ｔ□の整数倍、例えば２倍の周波数とし、第１の分布定
数線路の長さＸｌは周波数ＦＴＩの伝搬波長λ１の１／
４またはその奇数倍、また第２の分布定数線路の長さＸ
２と第１の分布定数線路の長さＸｌとの和は、周波数Ｆ
ＴＺの伝搬波長λ２の１／２またはその整数倍とする。The first resonant frequency F is the second resonant frequency F.
The frequency is an integral multiple of t□, for example, twice, and the length Xl of the first distributed constant line is 1/1 of the propagation wavelength λ1 of the frequency FTI.
4 or an odd multiple thereof, and the length of the second distributed constant line
2 and the length Xl of the first distributed constant line is the frequency F
It is assumed to be 1/2 of the TZ propagation wavelength λ2 or an integral multiple thereof.
これらの関係を数式で示すと、次のようになる。These relationships can be expressed mathematically as follows.
Ｘ ｌ ＝ （２ｎ ＋ １） ・λ１／４Ｘ１＋Ｘ
２＝ｎ・λ２／２
（ｎ＝１．２・・・）
なお、第１．第２分布定数線路の特性インピーダンスは
、いずれも等しくＺｌ、ｌとする。X l = (2n + 1) ・λ1/4X1+X
2=n・λ2/2 (n=1.2...) Note that the first. The characteristic impedances of the second distributed constant lines are both Zl and l.
このような構成において、インハークｌの出力周波数が
第１の共振周波数Ｆア、のとき並列共振負荷回路２は共
振状態となるので、その基本波インピーダンスは線路の
特性インピーダンスＺ。とほぼ等しいか多少大きく、第
１の共振周波数ＦＴＩの高調波周波数成分に対するイン
ピーダンスは充分に小さくなり、また直列共振負荷回路
３のインピーダンスにおいてはその共振周波数ＦＴ□よ
りも高いので、常に誘導性でその絶対値も大きな値とな
る。このとき、第２の分布定数線路の長さは、第１の共
振周波数の１／４伝搬波長の奇数倍になる（ＸＩをλ１
／４、λ２−２λ１とすると、ｘ２−３／４・ｊｌとな
る）から、第２の分布定数線路の送電端におけるインピ
ーダンスは、共振状態にある並列共振負荷回路２のイン
ピーダンスに比べて充分小さく、したがって第１の分布
定数線路受電端には、あたかも並列共振負荷回路２だけ
が接続されたのと同様となる。第１の分布定数線路の長
さは周波数ＦＴＩの波長λ１の１／４またはその整数倍
であるから、その結果として、第１の分布定数線路の送
電端インピーダンスＺ、は周波数Ｆ丁、においでは、
Ｚ３吋Ｚｗ”／Ｚｔ＋
となり、その高調波成分に対しては充分に大きな値とな
る。そのため、インバータ１の出力電流は周波数ＦＴＩ
の正弦波形になり、その電力の殆ど全てが並列共振負荷
回路２に供給される。In such a configuration, when the output frequency of the inverter I is the first resonant frequency F, the parallel resonant load circuit 2 is in a resonant state, so that its fundamental wave impedance is the characteristic impedance Z of the line. is approximately equal to or somewhat larger than the first resonant frequency FTI, and the impedance to the harmonic frequency component of the first resonant frequency FTI is sufficiently small, and the impedance of the series resonant load circuit 3 is higher than its resonant frequency FT□, so it is always inductive. Its absolute value also becomes a large value. At this time, the length of the second distributed constant line is an odd multiple of the 1/4 propagation wavelength of the first resonant frequency (XI is λ1
/4, λ2-2λ1, then x2-3/4・jl), the impedance at the transmission end of the second distributed constant line is sufficiently small compared to the impedance of the parallel resonant load circuit 2 in the resonant state. , Therefore, it is as if only the parallel resonant load circuit 2 were connected to the power receiving end of the first distributed constant line. Since the length of the first distributed constant line is 1/4 of the wavelength λ1 of the frequency FTI or an integral multiple thereof, as a result, the transmitting end impedance Z of the first distributed constant line is equal to the frequency FTI, and the odor is , Z3''Zw''/Zt+, which is a sufficiently large value for its harmonic components.Therefore, the output current of inverter 1 is equal to the frequency FTI
almost all of the power is supplied to the parallel resonant load circuit 2.
次に、インバータ１の出力周波数が第２の共振周波数Ｆ
ＴＺのとき直列共振負荷回路３は共振状態となるので、
その基本波インピーダンスは線路の特性インピーダンス
Ｚ１．ｌに比べ一般に比較的小さく、高調波に対するイ
ンピーダンスは充分に大きい。しかも、並列共振負荷回
路２は上記いずれの周波数においても充分に低いインピ
ーダンスになるので、送電端インピーダンスは見かけ上
直列共振負荷回路３の基本波インピーダンス値とほぼ等
しくなる。そのため、インバータ１の出力電流は周波数
Ｆア２の正弦波形になり、その電力の殆ど全てが直列共
振負荷回路３に供給される。Next, the output frequency of the inverter 1 becomes the second resonant frequency F
At TZ, the series resonant load circuit 3 is in a resonant state, so
The fundamental wave impedance is the line characteristic impedance Z1. It is generally relatively small compared to l, and the impedance to harmonics is sufficiently large. Moreover, since the parallel resonant load circuit 2 has a sufficiently low impedance at any of the above frequencies, the transmission end impedance appears to be approximately equal to the fundamental wave impedance value of the series resonant load circuit 3. Therefore, the output current of the inverter 1 has a sinusoidal waveform with a frequency FA2, and almost all of its power is supplied to the series resonant load circuit 3.
この発明によれば、各々が固有の共振周波数を持つ２つ
の負荷回路を、１本の分布定数線路上において各共振周
波数の伝搬波長がら与えられる長さの位置に接続し、こ
れを１台のインバータ回路に直接接続することにより、
以下のような効果を期待することができる。According to this invention, two load circuits each having a unique resonant frequency are connected on one distributed constant line at a length given by the propagation wavelength of each resonant frequency, and these are connected to one unit. By connecting directly to the inverter circuit,
The following effects can be expected.
イ）インバータ回路と各負荷回路とは分布定数線路によ
り直接接続されるため、配線のインダクタンス成分に伴
う電圧の低下を防止することができる。b) Since the inverter circuit and each load circuit are directly connected by distributed constant lines, it is possible to prevent a drop in voltage due to the inductance component of the wiring.
口）インバータの動作周波数を負荷回路の共振周波数ま
たはその近傍で運転することにより、その共振周波数を
持つ負荷回路にだけその周波数成分の電力を供給でき、
しかもインバータの出力電流波形をほぼ正弦波状にする
ことができる。By operating the inverter at or near the resonant frequency of the load circuit, power of that frequency component can be supplied only to the load circuit that has that resonant frequency.
Moreover, the output current waveform of the inverter can be made almost sinusoidal.
第１図はこの発明の実施例を示す構成図、第２図は従来
例を示す構成図である。
１・・・電圧形インバータ、２・・・並列共振負荷回路
、２１．３１・・・負荷リアクタンス、２２．３２・・
・負荷抵抗、２３．３３・・・共振コンデンサ、４・・
・第１の分布定数線路、５・・・第２の分布定数線路、
６・・・切換スイッチ、７Ａ・・・直列リアクトル、７
Ｂ、７Ｃ・・・配線インダクタンス、８Ａ、８Ｂ・・・
配線。FIG. 1 is a block diagram showing an embodiment of the present invention, and FIG. 2 is a block diagram showing a conventional example. 1... Voltage source inverter, 2... Parallel resonant load circuit, 21.31... Load reactance, 22.32...
・Load resistance, 23.33... Resonance capacitor, 4...
- first distributed constant line, 5... second distributed constant line,
6...Selector switch, 7A...Series reactor, 7
B, 7C... Wiring inductance, 8A, 8B...
wiring.
Claims (1)
、第１の固有周波数では特定のインピーダンスを持ち、
それ以外の周波数ではこの特定インピーダンスよりも小
さな値のインピーダンスを持つ第１の負荷と、前記第１
の固有周波数とは異なる第２の固有周波数では特定のイ
ンピーダンスを持ち、それ以外の周波数ではこの特定イ
ンピーダンスよりも大きな値のインピーダンスを持つ第
２の負荷とにそれぞれ高周波電力を供給する回路であっ
て、 前記電圧形インバータから前記第１の負荷へ電力を供給
する電力伝送線路には、この線路内における第１の固有
周波数の伝搬波長の１／４またはその奇数倍に相当する
線路長を持つ第１の分布定数線路を、また前記電圧形イ
ンバータから前記第２の負荷へ電力を供給する電力伝送
線路には、前記第１の負荷と直列に接続され、かつ前記
第１の分布定数線路長との和がこれらの線路内における
第２の固有周波数の伝搬波長の１／２波長またはその整
数倍に相当する線路長を持つ第２の分布定数線路を設け
、 前記第１の負荷への電力供給は前記電圧形インバータの
動作周波数を前記第１の固有周波数またはその近傍の周
波数にして行ない、また前記第２の負荷への電力供給は
前記電圧形インバータの動作周波数を前記第２の固有周
波数またはその近傍の周波数にしてそれぞれ行なうこと
を特徴とする２つの負荷回路への高周波電力供給回路。[Claims] 1) A voltage source inverter that outputs high-frequency AC power has a specific impedance at a first natural frequency,
a first load having an impedance smaller than this specific impedance at other frequencies;
A circuit that supplies high-frequency power to a second load that has a specific impedance at a second natural frequency different from the natural frequency of the load, and a second load that has an impedance larger than the specific impedance at other frequencies. , a power transmission line for supplying power from the voltage source inverter to the first load has a line length corresponding to 1/4 of the propagation wavelength of the first natural frequency in the line or an odd multiple thereof. A power transmission line that supplies power from the voltage source inverter to the second load has a distributed constant line connected in series with the first load, and has a length of the first distributed constant line. A second distributed constant line having a line length whose sum corresponds to 1/2 wavelength of the propagation wavelength of the second natural frequency in these lines or an integral multiple thereof is provided, and power is supplied to the first load. The operation frequency of the voltage source inverter is set to the first natural frequency or a frequency close to the first natural frequency, and the power supply to the second load is performed by changing the operating frequency of the voltage source inverter to the second natural frequency or a frequency close to the first natural frequency. A high-frequency power supply circuit to two load circuits, characterized in that the high-frequency power supply circuit operates at a frequency in the vicinity of the two load circuits.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2237089A JPH04121061A (en) | 1990-09-10 | 1990-09-10 | High frequency power supply circuit for two load circuits |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2237089A JPH04121061A (en) | 1990-09-10 | 1990-09-10 | High frequency power supply circuit for two load circuits |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| JPH04121061A true JPH04121061A (en) | 1992-04-22 |
Family
ID=17010248
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP2237089A Pending JPH04121061A (en) | 1990-09-10 | 1990-09-10 | High frequency power supply circuit for two load circuits |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPH04121061A (en) |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2010279214A (en) * | 2009-06-01 | 2010-12-09 | Toshiba Mitsubishi-Electric Industrial System Corp | Power conversion apparatus |
| WO2017037815A1 (en) * | 2015-08-31 | 2017-03-09 | 株式会社 東芝 | Voltage conversion device and wireless power transmission device |
-
1990
- 1990-09-10 JP JP2237089A patent/JPH04121061A/en active Pending
Cited By (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2010279214A (en) * | 2009-06-01 | 2010-12-09 | Toshiba Mitsubishi-Electric Industrial System Corp | Power conversion apparatus |
| WO2017037815A1 (en) * | 2015-08-31 | 2017-03-09 | 株式会社 東芝 | Voltage conversion device and wireless power transmission device |
| JPWO2017037815A1 (en) * | 2015-08-31 | 2018-01-18 | 株式会社東芝 | Voltage conversion device and wireless power transmission device |
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