JPH0374014B2 - - Google Patents
Info
- Publication number
- JPH0374014B2 JPH0374014B2 JP58138441A JP13844183A JPH0374014B2 JP H0374014 B2 JPH0374014 B2 JP H0374014B2 JP 58138441 A JP58138441 A JP 58138441A JP 13844183 A JP13844183 A JP 13844183A JP H0374014 B2 JPH0374014 B2 JP H0374014B2
- Authority
- JP
- Japan
- Prior art keywords
- winding
- primary
- windings
- leakage inductance
- diameter
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired - Lifetime
Links
- 238000004804 winding Methods 0.000 claims description 60
- 239000004020 conductor Substances 0.000 claims description 8
- 238000010586 diagram Methods 0.000 description 5
- 239000004065 semiconductor Substances 0.000 description 3
- XEEYBQQBJWHFJM-UHFFFAOYSA-N Iron Chemical group [Fe] XEEYBQQBJWHFJM-UHFFFAOYSA-N 0.000 description 2
- 230000004907 flux Effects 0.000 description 2
- 230000001174 ascending effect Effects 0.000 description 1
- 230000005284 excitation Effects 0.000 description 1
- 238000010438 heat treatment Methods 0.000 description 1
- 230000006698 induction Effects 0.000 description 1
- WABPQHHGFIMREM-UHFFFAOYSA-N lead(0) Chemical compound [Pb] WABPQHHGFIMREM-UHFFFAOYSA-N 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
- H01F19/00—Fixed transformers or mutual inductances of the signal type
- H01F19/04—Transformers or mutual inductances suitable for handling frequencies considerably beyond the audio range
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Multimedia (AREA)
- Coils Or Transformers For Communication (AREA)
Description
【発明の詳細な説明】
この発明は漏れインダクタンスの低減化を図つ
た高周波変圧器の巻線構造に関する。DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a winding structure for a high frequency transformer that reduces leakage inductance.
近年、誘導加熱装置等、大容量の高周波電流を
使用する装置の発達にともない、それらの負荷整
合に用いる高周波変圧器の漏れインダクタンスを
下げることが、皮相電力の低減の観点から極めて
重要になつている。 In recent years, with the development of devices that use large-capacity high-frequency currents, such as induction heating devices, reducing the leakage inductance of the high-frequency transformers used for load matching has become extremely important from the perspective of reducing apparent power. There is.
例えば、１次側からみた漏れインダクタンスが
10μHであつたとすると 50KHz、1000Aの電流
を流した場合、この漏れインダクタンスによる電
圧降下Vdは、
Vd＝1000A×2π×50×103Hz×10×10-6H
＝1000π≒3140V ……(1)
となり、漏れインダクタンスによる無効電力は
3140KVarである。従つて、負荷電力を1000KW
とするとこの時の皮相電力P1は、
P1＝√10002＋31402＝3295KVA ……(2)
となる。 For example, the leakage inductance seen from the primary side is
Assuming that the current is 10 μH and a current of 50 KHz and 1000 A flows, the voltage drop Vd due to this leakage inductance is Vd = 1000 A × 2 π × 50 × 10 3 Hz × 10 × 10 -6 H = 1000 π ≒ 3140 V ……(1 ), and the reactive power due to leakage inductance is
It is 3140KVar. Therefore, the load power is 1000KW
Then, the apparent power P 1 at this time is P 1 =√1000 2 +3140 2 =3295KVA...(2).
一方、１次側からみた漏れインダクタンスが
2μHの場合、50KHz、1000Aの電流を流したとき
の無効電力は628KVarであり、負荷電力1000KW
のときの皮相電力P2は、
P2＝√10002＋6282＝1181KVA ……(3)
となり、極めて有利である。しかも近年、上記１
次側に電力を供給する高周波電源は半導体を利用
したものが普及し始め、前記皮相電力を低減する
ことは、これら使用半導体の規格を市場部品の定
格値内に収める意味でも、また直並列数を低減す
る上でも極めて重要である。 On the other hand, the leakage inductance seen from the primary side is
For 2μH, the reactive power is 628KVar when a current of 50KHz and 1000A flows, and the load power is 1000KW.
The apparent power P 2 at this time is P 2 =√1000 2 +628 2 =1181KVA (3), which is extremely advantageous. Moreover, in recent years,
High-frequency power supplies that use semiconductors to supply power to the next side have begun to spread, and reducing the apparent power is important in keeping the standards of these semiconductors within the rated values of market components, and also in series and parallel numbers. It is also extremely important to reduce the
そして、上記漏れインダクタンスの絶対値を小
さくするために、鉄心を全く持たない空心変圧器
を適用するケースが増加してきた。これによれ
ば、漏れの割合は増加するものの漏れインダクタ
ンスの値そのものは大幅に下げうるからである。
また、励磁電流は相当に増加するけれども、これ
は通常、負荷電流と比べて僅かであるから、結局
全体としては皮相電力を極めて小さな値に抑える
ことができる。（例えば上記皮相電力P1を鉄心を
有する変圧器に、P2を空心変圧器に各々対応さ
せて考えれば、その事情が理解される。）
ところで、上述した従来の空心変圧器において
は、第１図に示すように、１次巻線１のリード線
１ａ，１ｂおよび２次巻線２のリード線２ａ，２
ｂが各々巻線部の反対側に位置し、リード線１ａ
−１ｂ間、２ａ−２ｂ間を通る磁束による配線イ
ンダクタンスが相当に大きくなり、大型の変圧器
においては2μH程度にもなつて、これを除去する
ことが大きな問題となつていた。 In order to reduce the absolute value of the leakage inductance, air-core transformers having no iron core are increasingly being used. According to this, although the leakage rate increases, the value of the leakage inductance itself can be significantly lowered.
Further, although the excitation current increases considerably, this is usually small compared to the load current, so the overall apparent power can be suppressed to an extremely small value. (For example, if we consider the above-mentioned apparent power P 1 to correspond to a transformer with an iron core and P 2 to an air-core transformer, the situation can be understood.) By the way, in the conventional air-core transformer described above, As shown in Figure 1, lead wires 1a, 1b of the primary winding 1 and lead wires 2a, 2 of the secondary winding 2.
b are each located on the opposite side of the winding part, and the lead wire 1a
The wiring inductance due to the magnetic flux passing between -1b and 2a and 2b becomes considerably large, reaching about 2 μH in large transformers, and eliminating this has become a major problem.
この発明は上記の事情に鑑み、漏れインダクタ
ンスの低減を図つた高周波変圧器の巻線構造を提
供するもので、１次側および２次側の巻線を平帯
導体により形成するとともに、前記各巻線の巻始
めおよび巻終りを巻線部の同一側に形成し、かつ
前記１次２次巻線間および各巻線の往路と復路間
の間隔を内側巻線の直径に対して十分に小さくし
たことを特徴とする。 In view of the above-mentioned circumstances, the present invention provides a winding structure for a high frequency transformer that aims to reduce leakage inductance. The winding start and winding end of the wire are formed on the same side of the winding part, and the intervals between the primary and secondary windings and between the forward and backward paths of each winding are sufficiently small with respect to the diameter of the inner winding. It is characterized by
さらに詳述するとこの発明は、次の２点に着眼
してなされたものである。第１に、各巻線を断面
長方形状の平帯導体によつて形成するとともに、
第２図に示す２次巻線（内側巻線）１２の直径ｄ
を１次巻線１１と２次巻線１２の間隔Δdに比べ
て十分大きくとり、両巻線間の面積ΔSを磁束通
過断面積Ｓに比較して十分に小さくした点であ
る。 More specifically, this invention has been made with attention to the following two points. First, each winding is formed of a flat conductor with a rectangular cross section, and
Diameter d of the secondary winding (inner winding) 12 shown in Figure 2
is made sufficiently larger than the interval Δd between the primary winding 11 and the secondary winding 12, and the area ΔS between the two windings is made sufficiently smaller than the magnetic flux passage cross-sectional area S.
以下、図面を参照して本発明の実施例を説明す
る。 Embodiments of the present invention will be described below with reference to the drawings.
第３図は本発明の一実施例に係る１次巻線１１
（または２次巻線１２）の構成を示す図で、同図
イはその斜視図、同図ロは等価回路図である。こ
れらの図において、１３は１次巻線１１を形成す
るテープ状の平帯導体で、この平帯導体１３は１
次巻線１１の上端に形成された巻始めターミナル
１１ａから下端に向う往路を平巻き状に（すなわ
ち、平帯導体１３の長方形断面の長辺を軸方向に
して）巻回されつつ下行する。そして、下端に達
すると今度は下端から上端に向う復路と往路と同
方向に巻回されつつ上行して上端に至り、巻始め
ターミナル１１ａと最短間隔で対向する巻終りタ
ーミナル１１ｂに終端する。ここで、往路、復路
２層間の間隔も極めて小さくする。 FIG. 3 shows a primary winding 11 according to an embodiment of the present invention.
(or the secondary winding 12), in which A is a perspective view thereof and B is an equivalent circuit diagram. In these figures, 13 is a tape-shaped flat conductor that forms the primary winding 11;
The next winding 11 moves downward while being wound in a flat winding (with the long side of the rectangular cross section of the flat band conductor 13 in the axial direction) on the outward path from the winding start terminal 11a formed at the upper end to the lower end. When it reaches the lower end, it is wound in the same direction as the backward and forward passes from the lower end to the upper end, ascending to the upper end, and terminating at the winding end terminal 11b facing the winding start terminal 11a at the shortest interval. Here, the interval between the two layers on the outbound and return trips is also made extremely small.
次に、２次巻線１２も平帯導体１３で同様に構
成し、１次巻線１１の中に緊密に挿入する。この
とき、両巻線１１，１２間の距離Δdは第２図に
示すように２次巻線１２の直径ｄよりも遥かに小
さくする。 Next, the secondary winding 12 is similarly constructed of a flat band conductor 13 and inserted tightly into the primary winding 11. At this time, the distance Δd between both windings 11 and 12 is made much smaller than the diameter d of the secondary winding 12, as shown in FIG.
このような構成によれば、巻始めターミナル１
１ａ（１２ａ）と巻終りターミナル１１ｂ（１２
ｂ）間のインダクタンスを0.1μH程度の極めて小
さな値を抑えることが可能である。また、１次巻
線１１と２次巻線１２の漏れインダクタンスも最
小化でき、１次側電圧300V、２次側電圧1000V、
容量250KVAの空心変圧器の例では、１次側から
みた漏れインダクタンスを1μH程度にすることが
できる。 According to such a configuration, the winding start terminal 1
1a (12a) and winding end terminal 11b (12
It is possible to suppress the inductance between b) to an extremely small value of about 0.1 μH. In addition, the leakage inductance of the primary winding 11 and the secondary winding 12 can be minimized, and the primary side voltage is 300V, the secondary side voltage is 1000V,
In the example of an air-core transformer with a capacity of 250 KVA, the leakage inductance seen from the primary side can be reduced to about 1 μH.
なお、上記実施例においては、平帯導体１３を
１往復させた例について説明したが、ｎ往復（ｎ
は任意の自然数）にしても問題はない。 In the above embodiment, an example was explained in which the flat band conductor 13 was made to reciprocate once, but n reciprocations (n
is any natural number).
以上説明したようにこの発明は１次側および２
次側巻線を平帯導体により形成するとともに、前
記各巻線の巻始めおよび巻終りを巻線部の同一側
に形成し、かつ前記１次２次巻線間および各巻線
の往路と復路間の間隔を内側巻線の直径に対して
十分に小さくしたので、高周波変圧器の漏れイン
ダクタンスを極めて小さくすることができる。こ
の結果、前記高周波変圧器の皮相電力を低減し得
て、電源回路に使用する半導体の許容値や数を減
らしうる利点が得られる。 As explained above, this invention is applicable to the primary side and the secondary side.
The next winding is formed of a flat band conductor, and the winding start and winding end of each of the windings are formed on the same side of the winding part, and between the primary and secondary windings and between the forward and return paths of each winding. Since the spacing is made sufficiently small with respect to the diameter of the inner winding, the leakage inductance of the high frequency transformer can be made extremely small. As a result, it is possible to reduce the apparent power of the high-frequency transformer, and there is an advantage that the permissible value and number of semiconductors used in the power supply circuit can be reduced.
第１図は従来の空心変圧器において、巻線のリ
ード線間に生じる配線インダクタンスを説明する
ための図、第２図は本発明の一実施例の１次巻線
と２次巻線の間隔Δdと２次巻線の内径ｄとの関
係を示す概略平面図、第３図は同実施例に係る巻
線の構成を示す図で、同図イはその斜視図、同図
ロは等価回路図である。
１１……１次巻線、１１ａ，１２ａ……巻始め
ターミナル、（巻始め）、１１ｂ，１２ｂ……巻終
りターミナル（巻終り）、１２……２次巻線、１
３……平帯導体、ｄ……内側巻線の直径、Δd…
…１次２次巻線間の間隔。
Fig. 1 is a diagram for explaining the wiring inductance that occurs between the lead wires of the windings in a conventional air-core transformer, and Fig. 2 is a diagram illustrating the spacing between the primary and secondary windings in an embodiment of the present invention. FIG. 3 is a schematic plan view showing the relationship between Δd and the inner diameter d of the secondary winding, and FIG. 3 is a diagram showing the configuration of the winding according to the same embodiment, and FIG. It is a diagram. 11... Primary winding, 11a, 12a... Winding start terminal, (winding start), 11b, 12b... Winding end terminal (winding end), 12... Secondary winding, 1
3...Flat band conductor, d...Diameter of inner winding, Δd...
...The spacing between the primary and secondary windings.
Claims (1)
おいて、１次側および２次側の巻線が平帯導体か
らなり、前記各巻線の巻始めおよび巻終りが巻線
部の同一側にあり、かつ前記１次２次巻線間およ
び前記各巻線の往路と復路間の間隔を内側巻線の
直径に対して十分小さくしたことを特徴とする高
周波変圧器の巻線構造。1. In an air-core transformer used for high frequencies of 10 KHz or higher, the primary and secondary windings are made of flat band conductors, and the beginning and end of each winding are on the same side of the winding section, and A winding structure for a high frequency transformer, characterized in that the interval between the primary and secondary windings and between the forward and backward paths of each of the windings is made sufficiently small with respect to the diameter of the inner winding.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP58138441A JPS6030106A (en) | 1983-07-28 | 1983-07-28 | Winding composition of high frequency transformer |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP58138441A JPS6030106A (en) | 1983-07-28 | 1983-07-28 | Winding composition of high frequency transformer |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS6030106A JPS6030106A (en) | 1985-02-15 |
| JPH0374014B2 true JPH0374014B2 (en) | 1991-11-25 |
Family
ID=15222062
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP58138441A Granted JPS6030106A (en) | 1983-07-28 | 1983-07-28 | Winding composition of high frequency transformer |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPS6030106A (en) |
Families Citing this family (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US7573000B2 (en) | 2003-07-11 | 2009-08-11 | Lincoln Global, Inc. | Power source for plasma device |
| PL233867B1 (en) * | 2017-12-04 | 2019-12-31 | Siec Badawcza Lukasiewicz – Instytut Tele I Radiotechniczny | Coreless current to voltage converter |
-
1983
- 1983-07-28 JP JP58138441A patent/JPS6030106A/en active Granted
Also Published As
| Publication number | Publication date |
|---|---|
| JPS6030106A (en) | 1985-02-15 |
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