JPH0336327B2 - - Google Patents

Info

Publication number
JPH0336327B2
JPH0336327B2 JP14645983A JP14645983A JPH0336327B2 JP H0336327 B2 JPH0336327 B2 JP H0336327B2 JP 14645983 A JP14645983 A JP 14645983A JP 14645983 A JP14645983 A JP 14645983A JP H0336327 B2 JPH0336327 B2 JP H0336327B2
Authority
JP
Japan
Prior art keywords
electrode
intersecting
conductance
surface acoustic
electrodes
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired
Application number
JP14645983A
Other languages
Japanese (ja)
Other versions
JPS6038912A (en
Inventor
Akitsuna Yuhara
Takashi Shiba
Jun Yamada
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Hitachi Ltd
Original Assignee
Hitachi Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Hitachi Ltd filed Critical Hitachi Ltd
Priority to JP14645983A priority Critical patent/JPS6038912A/en
Publication of JPS6038912A publication Critical patent/JPS6038912A/en
Publication of JPH0336327B2 publication Critical patent/JPH0336327B2/ja
Granted legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/125—Driving means, e.g. electrodes, coils
    • H03H9/145—Driving means, e.g. electrodes, coils for networks using surface acoustic waves
    • H03H9/14544—Transducers of particular shape or position
    • H03H9/14552—Transducers of particular shape or position comprising split fingers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02535—Details of surface acoustic wave devices
    • H03H9/02818—Means for compensation or elimination of undesirable effects
    • H03H9/02842—Means for compensation or elimination of undesirable effects of reflections
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02535—Details of surface acoustic wave devices
    • H03H9/02637—Details concerning reflective or coupling arrays
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/125—Driving means, e.g. electrodes, coils
    • H03H9/145—Driving means, e.g. electrodes, coils for networks using surface acoustic waves
    • H03H9/14517—Means for weighting

Landscapes

  • Physics & Mathematics (AREA)
  • Acoustics & Sound (AREA)
  • Surface Acoustic Wave Elements And Circuit Networks Thereof (AREA)

Description

【発明の詳細な説明】 〔発明の利用分野〕 本発明は電極反射が抑圧された弾性表面波装置
に係り、特にUHF等の高周波における使用に際
し好適なるよう、インピーダンス設計の自由度を
広げることができると共に外部回路との整合を簡
易化することができる弾性表面波フイルタに関す
るものである。[Detailed Description of the Invention] [Field of Application of the Invention] The present invention relates to a surface acoustic wave device in which electrode reflection is suppressed, and it is possible to widen the degree of freedom in impedance design so that it is particularly suitable for use at high frequencies such as UHF. The present invention relates to a surface acoustic wave filter that can be easily matched with an external circuit.

〔発明の背景〕[Background of the invention]

弾性表面波装置では弾性表面波基板上に１個以
上のすだれ状電極が設けられている。このすだれ
状電極の各電極指の交差幅、ピツチに重みをつけ
ることにより、振幅、位相の伝達周波数特性を
種々設計できる。この弾性表面波装置のすだれ状
電極は弾性表面波を反射する。そのため、電極間
の多重反射による遅延信号が生じて、振幅、位相
の伝達周波数特性にリツプルが発生するので、い
くつかの対策が施されている。 In a surface acoustic wave device, one or more interdigital electrodes are provided on a surface acoustic wave substrate. By weighting the intersecting width and pitch of each electrode finger of this interdigital electrode, various transmission frequency characteristics of amplitude and phase can be designed. The interdigital electrodes of this surface acoustic wave device reflect surface acoustic waves. As a result, a delayed signal is generated due to multiple reflections between the electrodes, causing ripples in the transmission frequency characteristics of amplitude and phase, so several countermeasures have been taken.

上記のすだれ状電極による弾性表面波の反射に
は、電気的な再放射と、音響的不連続による反射
の二種類に分類できる。後者の音響的不連続によ
る反射はUSP3699364に示されたような電極幅と
空白部が1/8波長で、同一極性の電極指が２本１
組となつているスプリツト電極指により、一般に
対策されている。 The reflection of surface acoustic waves by the interdigital electrodes described above can be classified into two types: electrical re-radiation and reflection due to acoustic discontinuity. The latter reflection due to acoustic discontinuity occurs when the electrode width and blank space are 1/8 wavelength and two electrode fingers of the same polarity are used as shown in USP 3699364.
This is generally countered by using pairs of split electrode fingers.

一方、上記のうち前者の電気的再放射による電
極反射は、送受波電極の放射コンダクタンスGa
と外部回路のコンダクタンスGLとの相対関係で
定まる。 On the other hand, the former electrode reflection due to electrical re-radiation is caused by the radiation conductance G a of the transmitting and receiving electrodes.
It is determined by the relative relationship between and the conductance G L of the external circuit.

電極反射の尺度として、 −10log10（反射波弾性表面波エネルギ／単
位時間）／（入射弾性表面波エネルギ／単位時間） をとり、反射損失と呼んでいる。電極反射による
特性劣化を避けるため、反射損失の基準として
は、入力電極と出力電極の二回の反射を合せた値
としてTV画像の主観評価から得られた40dB以上
が多く採用されている。これは入力と出力の各電
極に等しく分担させた場合、一電極当りの反射波
損失として20dBであつて、これを達成するため
従来は第２図に準拠して、GL／Ga＞９なる如く
送受波電極放射コンダクタンスGaを与えていた。
この際フイルタ通過損失は実用上約20dB程度で
用いられている。 As a measure of electrode reflection, −10log 10 (reflected surface acoustic wave energy/unit time)/(incident surface acoustic wave energy/unit time) is taken and is called reflection loss. In order to avoid characteristic deterioration due to electrode reflection, the standard for reflection loss is often 40 dB or more, which is the sum of two reflections from the input and output electrodes, obtained from subjective evaluation of TV images. This is a reflected wave loss of 20 dB per electrode when shared equally between the input and output electrodes, and to achieve this, conventionally, G L /G a > 9 according to Figure 2. The transmitting/receiving electrode radiation conductance G a was given as follows.
In this case, the filter passing loss is practically used at about 20 dB.

一方、外部回路系のコンダクタンスは一定の
値、多くは20mm（インピーダンスで50Ω）と定
められているので、上記従来法では送受波電極の
放射コンダクタンスを2.2m以下と小さくする
必要があつた。このため、VHF高域ないしUHF
帯の高周波領域あるいは比帯域幅の狭い場合に、
通常の二電極構成では次の２つの欠点が発生し
た。 On the other hand, the conductance of the external circuit system is set at a constant value, often 20 mm (50 Ω impedance), so in the conventional method described above, it was necessary to reduce the radiation conductance of the transmitting and receiving electrodes to 2.2 m or less. For this reason, VHF high range or UHF
In the high frequency region of the band or when the fractional bandwidth is narrow,
The following two drawbacks occurred in the conventional two-electrode configuration.

周波数が高く、あるいは対数が大きく（狭帯
域）定められた条件の下でGaを従来法による値
（2.2m）とするためには電極の開口長Ｗを小さ
く定めねばならない。そのため、特に重み付け電
極の交差幅の小さい部分のため、回折による帯域
通過特性の裾部の乱れ、帯域内のうねり等の特性
劣化が発生する。例えば、結合定数の大きな
（0.055）のLiNbO3単結晶−128゜回転Ｙ軸カツト、
Ｘ軸方向伝搬基板を用いた中心周波数400MHz、
3dB帯域幅30MHzの入力正規型、出力重み付けの
弾性表面波フイルタでは開口長Ｗは80μmすなわ
ち約８波長と小さくなり、第１図に示した上記例
の特性に見られるように上記した劣化は顕著であ
る。 In order to set G a to the value of the conventional method (2.2 m) under defined conditions of high frequency or large logarithm (narrow band), the aperture length W of the electrode must be set small. Therefore, especially in the portion where the cross width of the weighting electrodes is small, characteristic deterioration such as disturbance of the tail of the bandpass characteristic due to diffraction and waviness within the band occurs. For example, LiNbO 3 single crystal with a large coupling constant (0.055) - 128° rotation Y-axis cut,
Center frequency 400MHz using X-axis direction propagation board,
In a surface acoustic wave filter with a normal input type and output weighting with a 3 dB bandwidth of 30 MHz, the aperture length W is as small as 80 μm, or approximately 8 wavelengths, and the above-mentioned deterioration is noticeable as seen in the characteristics of the above example shown in Figure 1. It is.

次に、高周波になると、電極の容量だけでなく
パツケージや外部回路系の容量分を含むサセプタ
ンス分が、送受波電極の放射コンダクタンス分に
比べて、かなり大きな値となるので、サセプタン
ス分を打消すインダクタンスを含む結合回路が必
要であり、部品点数が増えるだけでなく、インダ
クタンスの調整が必要となる欠点が生ずる。 Next, at high frequencies, the susceptance, which includes not only the capacitance of the electrode but also the capacitance of the package and external circuit system, becomes a much larger value than the radiation conductance of the transmitting and receiving electrodes, so the susceptance is canceled out. A coupling circuit including an inductance is required, which not only increases the number of parts but also has the drawback of requiring inductance adjustment.

〔発明の目的〕[Purpose of the invention]

本発明の目的は、VHF高域、あるいはUHF帯
域の高周波領域、あるいは、帯域幅が比較的に狭
い場合に、電極反射による特性劣化を無くし、し
かも回折等による特性劣化も無く、かつ外部回路
と整合しやすく、調整工程の少い弾性表面波装置
を提供するにある。 The purpose of the present invention is to eliminate characteristic deterioration due to electrode reflection in the high frequency range of VHF or UHF band, or when the bandwidth is relatively narrow, and also to eliminate characteristic deterioration due to diffraction etc. An object of the present invention is to provide a surface acoustic wave device that is easy to match and requires few adjustment steps.

〔発明の概要〕[Summary of the invention]

本発明では、圧電体基板と、圧電体基板上に設
けられた互に交差するくし歯状電極から各なる入
力および出力変換器とからなり、各くし歯状電極
は交差部分と非交差部分とからなる電極指と電極
指の非交差部分側端部を結合するバスバーとから
なる弾性表面波装置において、各変換器の一方の
くし歯状電極のバスバーと他方のくし歯状電極の
電極指先端との間隔が互いの接触を防ぐのに必要
な間隔よりも充分大きな間隔をもつように一方の
くし歯状電極の全電極指の非交差部分を長くする
ことにより、電極反射、回折による特性劣化を防
ぐとともに外部回路との結合を容易に行なえるよ
うにしたものである。 The present invention comprises a piezoelectric substrate and an input and output transducer formed of mutually intersecting comb-like electrodes provided on the piezoelectric substrate, and each comb-like electrode has an intersecting portion and a non-intersecting portion. In a surface acoustic wave device comprising electrode fingers consisting of electrode fingers and a bus bar connecting the non-intersecting side ends of the electrode fingers, the bus bar of one comb-like electrode and the tip of the electrode finger of the other comb-like electrode of each transducer By lengthening the non-intersecting portions of all the electrode fingers of one comb-shaped electrode so that the distance between the two electrodes is sufficiently larger than the distance required to prevent contact with each other, characteristic deterioration due to electrode reflection and diffraction can be prevented. This structure prevents this from occurring and also allows for easy connection with external circuits.

〔発明の実施例〕[Embodiments of the invention]

以下、本発明を実施例とともに説明する。 The present invention will be explained below along with examples.

第２図は本発明の基本原理を説明するための電
極形状を示す電極パタン図である。図から明らか
のように、電極指８の交差部分１８の長さＷは非
交差部分１９，１９′の長さW′，W″の和よりも
短くされ、非交差部分１９，１９′抵抗分が考慮
される。関係するパラメータは放射コンダクタン
スGa、電極全体にわたる電極指コンダクタンス
Gf、外部回路のコンダクタンスGLである。ここ
で上記電極指コンダクタンスGfは、対数Ｎと一
対当りの電極指抵抗rfにより次の第(1)式により与
えられる（スプリツト電極指の場合は並列接続し
た値で与える）。 FIG. 2 is an electrode pattern diagram showing electrode shapes for explaining the basic principle of the present invention. As is clear from the figure, the length W of the intersecting portion 18 of the electrode finger 8 is made shorter than the sum of the lengths W', W'' of the non-intersecting portions 19, 19', and the resistance of the non-intersecting portions 19, 19' is The relevant parameters are the radiation conductance G a , the electrode finger conductance across the electrode
G f is the conductance G L of the external circuit. Here, the electrode finger conductance G f is given by the following equation (1) using the logarithm N and the electrode finger resistance r f per pair (in the case of split electrode fingers, it is given as a value connected in parallel).

Gf＝Ｎ／rf (1) また、一対当りの電極指抵抗rfは次式で与え
る。 G f =N/r f (1) Also, the electrode finger resistance r f per pair is given by the following formula.

rf＝（一方の極性の電極指の有効開口部と非交
差部における抵抗）＋（他の極性の電極指の非交差
部における抵抗） (2) 上述の反射損失は１電極当り、次に与えられ
る。 r f = (resistance at the effective opening and non-intersecting part of the electrode fingers of one polarity) + (resistance at the non-intersecting part of the electrode fingers of the other polarity) (2) The above reflection loss is per electrode, then Given.

−10log10｛c2／（ｂ＋１）c2＋（2b＋１）
ｃ＋ｂ・abc／１＋abc｝2(3) ここで、ｂ＝GL／Ga，ｃ＝Gf／Gaであり、係
数ａは実験的に0.0415と定めた。 −10log 10 {c 2 /(b+1)c 2 +(2b+1)
c+b·abc/1+abc} 2 (3) Here, b=G L /G a , c=G f /G a , and the coefficient a was experimentally determined to be 0.0415.

第(3)式を20dBと置いた第(4)式が、コンダクタ
ンスGLの外部回路に接続して、一電極当りの反
射損失を基準の20dBとするために必要なGfを定
める条件式である。 Equation (4), where Equation (3) is set at 20 dB, is a conditional expression that determines G f required to set the return loss per electrode to the standard 20 dB when connected to an external circuit with conductance G L. It is.

−10log10｛c2／（ｂ＋１）c2＋（2b＋１）
ｃ＋ｂ×0.0415bc／１＋0.0415bc｝2≧20（dB）(4) 先ず、電極指コンダクタンスGfの電極反射低
減効果（反射損失の増大）について説明する。前
記の如く、反射損失はｂ＝GL／Ga，ｃ＝Gf／Ga
は次の様に与えられる。 −10log 10 {c 2 /(b+1)c 2 +(2b+1)
c+b×0.0415bc/1+0.0415bc} 2 ≧20 (dB) (4) First, the electrode reflection reduction effect (increase in reflection loss) of the electrode finger conductance G f will be explained. As mentioned above, the reflection loss is b=G L /G a , c=G f /G a
is given as follows.

−10log10｛c2／（ｂ＋１）c2＋（2b＋１）
ｃ＋ｂ・0.0415bc／１＋0.0415bc｝2(5) 実験例として、外部回路のコンダクタンスGL
＝20mで、基板にLiNbO3単結晶、128゜回転Ｙ
軸カツト、Ｘ軸方向伝搬を用い、送受波電極は入
力出力共に、中心周波数400MHz、対数14の正規
型電極で、電極材料は純Alで膜厚0.1μm、有効開
口長は200μm、放射コンダクタンスGaは5.6m
であつて、入出力共に等しく非交差部の延長によ
る電極指コンダクタンスGfの変化が行なわれて
いる場合の反射損失と非交差部長さの関係を第３
図に示す。第３図で、縦軸は１電極当りの反射損
失、横軸は電極指抵抗rf、電極指コンダクタンス
Gf（＝Ｎ／rf）、全電極指長（Ｗ＋W′＋W″）を同
時に示す。第３図では電極指非交差部による電極
指抵抗の増大で反射損失が増加することおよび実
線で示す第(5)式の計算値と、実験値を示すＡ（非
交差部延長せず、Gf＝622.2m）、Ｂ（非交差部
長さW′＋W″＝500μm、Gf＝177.8mΩ）、Ｃ（非交
差部長さW′＋W″＝1000μm、Gf＝103.7m）、Ｄ
（非交差部長さW′＋W″＝1600μm、Gf＝69.1m
）、Ｅ（非交差部長さW′＋W″＝2400μm、Gf＝
47.9ｍ）の各点の値が一致することが示され
る。 −10log 10 {c 2 /(b+1)c 2 +(2b+1)
c+b・0.0415bc/1+0.0415bc} 2 (5) As an experimental example, conductance G L of the external circuit
= 20m, LiNbO 3 single crystal substrate, 128° rotation Y
Using axial cut and propagation in the X-axis direction, the transmitting and receiving electrodes are regular type electrodes with a center frequency of 400 MHz and a logarithm of 14 for both input and output.The electrode material is pure Al with a film thickness of 0.1 μm, an effective aperture length of 200 μm, and a radiation conductance. G a is 5.6m
The relationship between the reflection loss and the length of the non-crossing section when the electrode finger conductance G f is changed equally by the extension of the non-crossing section for input and output is expressed as
As shown in the figure. In Figure 3, the vertical axis is the reflection loss per electrode, and the horizontal axis is the electrode finger resistance r f and the electrode finger conductance.
G f (=N/r f ) and the total electrode finger length (W+W′+W″) are shown at the same time. In Figure 3, the reflection loss increases due to the increase in electrode finger resistance due to the non-crossing of the electrode fingers, and this is shown by the solid line. A (non-intersecting part not extended, G f = 622.2 m), B (non-intersecting part length W' + W'' = 500 μm, G f = 177.8 mΩ), showing the calculated value of equation (5) and the experimental value, C (non-intersecting length W′+W″=1000μm, G f =103.7m), D
(Non-intersecting length W′+W″=1600μm, G f =69.1m
), E (non-intersecting length W′+W″=2400 μm, G f =
47.9m) are shown to match.

次に本発明の第１の実施例について説明する。
第４図は本発明の第１の実施例を示す電極パタン
図である。第４図に示す電極パタンは第３図の実
験例と同じLiNbO3基板上に形成される。第２図
と同様な長さＷ＝200μmの交差部分18、長さ
W′＝W″＝1200μmの非交差部分１９，１９′を持
つ正規型入力電極２と、該正規型入力電極と同じ
有効開口部、非交差部を持つ交差幅重み付けを持
つ出力電極２′が、電極材料に純Alを用い膜厚
0.1μm、電極線幅、空白部が1.2μmで形成され、
中心周波数は400MHz、入力正規型電極の対数は
14対、出力重み付け電極の対数は60対で3dB帯域
幅は30MHzである。中心周波数における放射コン
ダクタンスはいずれも5.6m、電極指コンダク
タンスGfは正規型電極２で、69.1mであり、重
み付け電極では電極指コンダクタンスGfにはそ
の主要交差部（メインローブ）が支配的であつ
て、等価的には正規型と同様である。 Next, a first embodiment of the present invention will be described.
FIG. 4 is an electrode pattern diagram showing the first embodiment of the present invention. The electrode pattern shown in FIG. 4 is formed on the same LiNbO 3 substrate as in the experimental example shown in FIG. Intersection part 18 with length W = 200 μm similar to Fig. 2, length
A normal type input electrode 2 with non-intersecting parts 19, 19' of W'=W''=1200 μm, and an output electrode 2' having crossing width weighting with the same effective opening and non-intersecting part as the normal type input electrode. , using pure Al as the electrode material and reducing the film thickness.
The electrode line width is 0.1μm, the blank area is 1.2μm,
The center frequency is 400MHz, and the logarithm of the input normal electrode is
14 pairs, the number of pairs of output weighting electrodes is 60 pairs, and the 3 dB bandwidth is 30 MHz. The radiation conductance at the center frequency is 5.6 m in both cases, and the electrode finger conductance G f is 69.1 m for normal electrode 2. In the weighted electrode, the electrode finger conductance G f is dominated by its main intersection (main lobe). It is equivalently the same as the normal type.

第１の実施例における上記の非交差部分１８の
長さはコンダクタンスGL＝20mの外部回路を
考えた場合における第(5)式で与えられる反射損失
計算値を20dBとすることにより定められる。（入
出力合せた反射損失は40dBとなる。） 一方、電極指非交差部の長さが、例えば上記実
施例ではW′＋W″＝2400μmと長くなるため、補
助手段として、第５図の如くコンダクタンスGL
を有する外部回路１６，１７と弾性表面波装置１
４との結合回路１３，１３′として並列抵抗（コ
ンダクタンスGp）を用いて、一電極当りの反射
損失を20dB（もしくは、入出力合せて40dB）と
するのが電極指非交差部を短くできるので、実際
的である。この際、GLをGL＋GPと置き換えて、
第(4)式を適用することで、GfとGPが決定される。 The length of the non-intersecting portion 18 in the first embodiment is determined by setting the return loss calculation value given by equation (5) to 20 dB when considering an external circuit with conductance G L =20 m. (The total reflection loss for input and output is 40 dB.) On the other hand, since the length of the non-intersecting part of the electrode fingers is, for example, W′+W″=2400 μm in the above embodiment, as an auxiliary measure, as shown in FIG. Conductance G L
External circuits 16 and 17 and surface acoustic wave device 1 having
Using a parallel resistance (conductance G p ) as the coupling circuit 13, 13' with 4 and setting the reflection loss per electrode to 20 dB (or 40 dB in total for input and output) can shorten the non-crossing part of the electrode fingers. So it's practical. At this time, replace G L with G L + G P ,
By applying equation (4), G f and G P are determined.

上記の考えに基づいた実施例を第２の実施例と
して示す。本実施例の平面構成その他は、入出力
共に非交差部長さW′＝W″＝500μmとし、並列コ
ンダクタンスGP＝21.3mとした以外は前記した
第１の実施例と同じである。本実施例の実験結果
を第６図（記号C1）に示す。第６図で、コンダ
クタンスGL＝20mを有する外部回路に、並列
コンダクタンスGPを接続した等価外部コンダク
タンス（GL＋GP）を横軸に、通過損失、反射損
失（いずれも入出力２電極分）を縦軸にとり示
す。第６図でGL＝20mとGP＝21.3mの和、
41.3mで、入出力２電極の反射損失は40dBと目
標に達した。一方、通過損失は25dBに止まり、
極端に大きくはならない。 An example based on the above idea will be shown as a second example. The planar configuration and other aspects of this embodiment are the same as those of the first embodiment described above, except that the non-intersecting portion length W' = W'' = 500 μm for both input and output, and the parallel conductance G P = 21.3 m. The experimental results of the example are shown in Fig. 6 (symbol C 1 ). In Fig. 6, the equivalent external conductance (G L + G P ) in which a parallel conductance G P is connected to an external circuit having a conductance G L = 20 m is plotted horizontally. The vertical axis shows the transmission loss and reflection loss (both for input and output 2 electrodes) on the axis. In Figure 6, the sum of G L = 20 m and G P = 21.3 m,
At 41.3m, the reflection loss of the two input and output electrodes reached the target of 40dB. On the other hand, the passing loss remains at 25dB,
It will not become extremely large.

第３の実施例として、第２の実施例の非交差部
長さから、W′＝W″＝750μmと変えた場合を示
す。本実施例の実験結果を第２の実施例と同様に
第６図（記号D1）に示すが、コンダクタンスGL
＝20mの外部回路で、並列コンダクタンス10m
を接続することで、入出力２電極の反射損失は
40dBと目標に達し、通過損失は22dBであつた。 As a third example, we will show a case where the length of the non-intersecting part in the second example is changed to W' = W'' = 750 μm. As shown in the figure (symbol D 1 ), the conductance G L
= 20m external circuit, parallel conductance 10m
By connecting, the reflection loss of the two input and output electrodes is
The target was reached at 40dB, and the passing loss was 22dB.

本発明の第４の実施例として、第３の実施例に
て出力の重み付け電極に非交差部の延長を行なつ
ていない場合を示す。この実施例の電極パタンを
第７図に示す。本実施例では、非交差部延長を行
なつていない出力重み付け電極の反射損失が小さ
く、GL＝20mの外部回路に対しては14dB、並
列コンダクタンスGP＝21.3mを接続して18dB
であるが、非交差部延長を行なつた正規型電極で
はGL＝20mの外部回路に対しては反射損失が
18.2dB、並列コンダクタンスGP＝21.3mを接続
して、反射損失が22dBとなるので、入出力合せ
た反射損失が40dBの目標値に達した。 As a fourth embodiment of the present invention, a case will be shown in which the non-intersecting portions of the output weighting electrodes are not extended in the third embodiment. The electrode pattern of this example is shown in FIG. In this example, the reflection loss of the output weighting electrode without any non-intersection extension is small; it is 14 dB for an external circuit of G L = 20 m, and 18 dB for connecting a parallel conductance G P = 21.3 m.
However, for a regular electrode with non-intersecting extension, the reflection loss will be low for an external circuit of G L = 20 m.
By connecting 18.2 dB and parallel conductance G P = 21.3 m, the return loss was 22 dB, so the combined input and output return loss reached the target value of 40 dB.

次に本発明の第５の実施例として、非交差部の
電極構造に改善を施した場合を第８図に示す。第
８図の電極構造では、有効開口部１８ではスプリ
ツト電極指であるが、非交差部１９，１９′では
同じ極性の隣接した２本の電極指が１本化され、
かつ電極幅が有効開口部の１本分の幅と同じであ
り、非交差部の抵抗値が２倍となるため、第２図
の場合に比べて非交差部の長さが1/2で済み、装
置が小型となる利点が有る。逆に、上記第５の実
施例では膜厚を厚くしても良いので、ワイヤボン
デイングが容易となる利点が有る。 Next, as a fifth embodiment of the present invention, FIG. 8 shows a case in which the electrode structure at non-intersecting portions is improved. In the electrode structure shown in FIG. 8, split electrode fingers are used in the effective opening 18, but two adjacent electrode fingers of the same polarity are combined into one in the non-intersecting parts 19 and 19'.
In addition, the electrode width is the same as the width of one effective opening, and the resistance value of the non-intersecting part is doubled, so the length of the non-intersecting part is 1/2 compared to the case in Figure 2. This has the advantage of making the device smaller. On the other hand, in the fifth embodiment, the film thickness may be increased, which has the advantage of facilitating wire bonding.

また、非交差部の延長に代えて、電極指膜厚を
薄くして電極指抵抗を増大させることが考えられ
るが、例えば第１の実施例に対応させると膜厚が
0.008μmと薄く、ばらつき、信頼性、抵抗による
重み付けの変動等が問題となるので、上記問題を
避ける上から、本発明の非交差部延長が有効であ
る。 Furthermore, instead of extending the non-intersecting portion, it is possible to reduce the electrode finger film thickness to increase the electrode finger resistance.
Since the thickness is as thin as 0.008 μm, there are problems with variations, reliability, and fluctuations in weighting due to resistance, so the non-intersecting portion extension of the present invention is effective in avoiding the above problems.

〔発明の効果〕〔Effect of the invention〕

本発明により、VHF高域以上（約200MHz）以
上の高周波、あるいは比較的に狭い帯域幅（比帯
域10％以下）で用いる弾性表面波装置において、
電極多重反射を40dB以下に押えて振幅、位相
（群遅延時間）の乱れを無くし、かつ、従来法よ
り大きな放射コンダクタンス、有効開口が得られ
るので、上記第３の実施例での周波数特性を示し
た第９図の如く、回折による帯域内うねり、裾部
の乱れ等の無い良好な周波数特性が得られると共
に、調整を要するインダクタンスを結合回路に用
いることなく、第５図の如く単に固定の並列抵抗
を用いることができるので、部品点数を低減し、
無調整化を進めることができた。 According to the present invention, in a surface acoustic wave device used at high frequencies above the VHF high range (approximately 200 MHz) or above, or at a relatively narrow bandwidth (fractional bandwidth 10% or less),
The frequency characteristics of the third embodiment above are shown because it suppresses multiple electrode reflections to 40 dB or less, eliminates disturbances in amplitude and phase (group delay time), and provides larger radiation conductance and effective aperture than the conventional method. As shown in Fig. 9, good frequency characteristics without in-band waviness or tail disturbance due to diffraction can be obtained, and there is no need to use an inductance that requires adjustment in the coupling circuit, and simply a fixed parallel connection as shown in Fig. 5. Since a resistor can be used, the number of parts can be reduced,
We were able to move forward with no adjustments.

【図面の簡単な説明】[Brief explanation of the drawing]

第１図は従来例の弾性表面波装置の周波数特性
を示す特性図、第２図は本発明による電極指の非
交差部を延長した電極構造の平面図、第３図は反
射損失と非交差部長さの関係を示す特性図、第４
図は本発明の一実施例を示す電極パタン図、第５
図は並列にコンダクタンスを接続した結合回路を
説明する回路図、第６図は、本発明の第２、第３
の実施例における反射損失、通過損失と、並列コ
ンダクタンスとの関係を示す特性図、第７図は本
発明の第４の実施例を模式的に示す電極パタン
図、第８図は本発明の非交差部延長の他の構造を
示す電極パタン図、第９図は本発明により改善さ
れた周波数特性を示す特性図である。 符号の説明、２，２′；弾性表面波の送受波電
極、３；ボンデイングパツド、４；母線電極、
５；電源、６；電源の内部コンダクタンス、
６′；負荷コンダクタンス、８；電極指、S1，S2，
…，SN；１，２，…，Ｎ番目の電極、１３，１
３′；並列抵抗（コンダクタンスGP）、１４；弾
性表面波装置、１５，１５′；結合回路部、１
６；信号源、１７；負荷、１８；有効開口、１
９；非交差部電極指、１９′；１９とは極性の異
なる非交差部電極指、Ａ；Gf＝622.2m（非交
差部延長なし）の場合、Ｂ；Gf＝177.8mの場
合、Ｃ；Gf＝103.7mの場合、Ｄ；Gf＝69.1m
の場合、Ｅ；Gf＝47.9mの場合、C1；第２の実
施例を示す測定点、D1；第３の実施例を示す測
定点。 Fig. 1 is a characteristic diagram showing the frequency characteristics of a conventional surface acoustic wave device, Fig. 2 is a plan view of the electrode structure in which the non-intersecting parts of the electrode fingers according to the present invention are extended, and Fig. 3 is a reflection loss and non-intersecting part of the electrode fingers. Characteristic diagram showing the relationship between lengths, No. 4
The figure is an electrode pattern diagram showing one embodiment of the present invention.
The figure is a circuit diagram explaining a coupling circuit in which conductances are connected in parallel, and FIG.
FIG. 7 is an electrode pattern diagram schematically showing the fourth embodiment of the present invention, and FIG. FIG. 9 is an electrode pattern diagram showing another structure of the intersection extension, and FIG. 9 is a characteristic diagram showing frequency characteristics improved by the present invention. Explanation of symbols, 2, 2'; Surface acoustic wave transmitting and receiving electrode, 3; Bonding pad, 4; Bus electrode,
5; power supply, 6; internal conductance of power supply,
6'; Load conductance, 8; Electrode fingers, S 1 , S 2 ,
..., S N ; 1, 2, ..., Nth electrode, 13, 1
3'; Parallel resistance (conductance G P ), 14; Surface acoustic wave device, 15, 15'; Coupling circuit section, 1
6; Signal source, 17; Load, 18; Effective aperture, 1
9; Non-intersecting electrode finger, 19'; Non-intersecting electrode finger with a different polarity from 19; A: In the case of G f = 622.2 m (no extension of the non-intersecting part); B; In the case of G f = 177.8 m, When C; G f = 103.7 m, D; G f = 69.1 m
In the case of E; in the case of G f =47.9m, C 1 ; measurement point indicating the second embodiment; D 1 ; measurement point indicating the third embodiment.

Claims (1)

【特許請求の範囲】 １ 圧電性基板と、圧電性基板上に設けられた互
いに交差するくし歯状電極から各々なる入力およ
び出力変換器とからなり、各くし歯状電極は交差
部分と非交差部分とからなる電極指と電極指の非
交差部分側端部を結合するバスバーとからなり、
少なくとも一方の変換器において、異なる極性の
電極指の非交差部分の長さの和が交差部分の長さ
より長く構成したことを特徴とする弾性表面波装
置。 ２ 送受波電極の放射コンダクタンスGaの大き
さを接続される外部回路コンダクタンスG1の1/9
以上とし、かつ、送受波電極の対数Ｎと一対の電
極指抵抗rfからＮ／rfと定義される電極指コンダ
クタンスGfが非交差部分の延長が行われて、ｂ
＝G1／Ga，ｃ＝Gf／Gaで定義されるｂ，ｃによ
る次の関係を満たすことを特徴とする特許請求の
範囲第１項記載の弾性表面波装置。 −10log10｛c2／（ｂ＋１）c2＋（2b＋１）
ｃ＋ｂ・0.0415bc／１＋0.0415bc｝2≧20dB[Claims] 1. Consists of a piezoelectric substrate and input and output transducers each consisting of intersecting comb-like electrodes provided on the piezoelectric substrate, each of the comb-like electrodes having an intersecting portion and a non-intersecting portion. and a bus bar connecting the ends of the non-intersecting parts of the electrode fingers,
A surface acoustic wave device characterized in that, in at least one transducer, the sum of the lengths of non-intersecting portions of electrode fingers of different polarities is longer than the length of the intersecting portions. 2 The size of the radiation conductance G a of the wave transmitting/receiving electrode is 1/9 of the connected external circuit conductance G1
With the above, the electrode finger conductance G f defined as N/r f is extended from the number N of the wave transmitting/receiving electrodes and the electrode finger resistance r f of a pair, and the non-intersecting portion is extended b
The surface acoustic wave device according to claim 1, wherein the surface acoustic wave device satisfies the following relationship by b and c defined by =G1/G a and c=G f /G a . −10log 10 {c 2 /(b+1)c 2 +(2b+1)
c+b・0.0415bc/1+0.0415bc} 2 ≧20dB
JP14645983A 1983-08-12 1983-08-12 Elastic surface wave device Granted JPS6038912A (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
JP14645983A JPS6038912A (en) 1983-08-12 1983-08-12 Elastic surface wave device

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
JP14645983A JPS6038912A (en) 1983-08-12 1983-08-12 Elastic surface wave device

Publications (2)

Publication Number Publication Date
JPS6038912A JPS6038912A (en) 1985-02-28
JPH0336327B2 true JPH0336327B2 (en) 1991-05-31

Family

ID=15408112

Family Applications (1)

Application Number Title Priority Date Filing Date
JP14645983A Granted JPS6038912A (en) 1983-08-12 1983-08-12 Elastic surface wave device

Country Status (1)

Country Link
JP (1) JPS6038912A (en)

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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1969438B1 (en) 2005-12-02 2009-09-09 iRobot Corporation Modular robot

Also Published As

Publication number Publication date
JPS6038912A (en) 1985-02-28

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