JPH0210989A - Color difference signal demodulation circuit - Google Patents
Color difference signal demodulation circuitInfo
- Publication number
- JPH0210989A JPH0210989A JP15912688A JP15912688A JPH0210989A JP H0210989 A JPH0210989 A JP H0210989A JP 15912688 A JP15912688 A JP 15912688A JP 15912688 A JP15912688 A JP 15912688A JP H0210989 A JPH0210989 A JP H0210989A
- Authority
- JP
- Japan
- Prior art keywords
- signal
- color difference
- color
- demodulation circuit
- difference signal
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
- 238000005070 sampling Methods 0.000 claims description 6
- 238000000034 method Methods 0.000 abstract description 8
- 238000001228 spectrum Methods 0.000 abstract description 5
- 238000007796 conventional method Methods 0.000 abstract description 4
- 238000001514 detection method Methods 0.000 abstract 1
- 239000000284 extract Substances 0.000 abstract 1
- 238000000926 separation method Methods 0.000 abstract 1
- 238000010586 diagram Methods 0.000 description 5
- 230000000694 effects Effects 0.000 description 3
- 150000001875 compounds Chemical class 0.000 description 2
- 238000006243 chemical reaction Methods 0.000 description 1
Landscapes
- Processing Of Color Television Signals (AREA)
- Filters That Use Time-Delay Elements (AREA)
Abstract
Description
【発明の詳細な説明】
〔産業上の利用分野〕
本発明は直交位相変調波の復調に係り、特にカラーテレ
ビジョン(T V)受像機における色差信号の復調に好
適な色差信号復調回路に関する。DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to the demodulation of quadrature-phase modulated waves, and particularly to a color difference signal demodulation circuit suitable for demodulating color difference signals in a color television (TV) receiver.
カラーTVにおける色差信号は直交位相変調により伝送
されている。直交位相変調波の復調には従来から直交搬
送波を掛けて変調信号を低域通過フィルタで取り出す方
法や、ヒルベルト(Hilbert)変換器を用いる方
法などが知られている。後者の方法により、色差信号を
変復調した例として、特開昭62−298295号があ
げられる。Color difference signals in color TVs are transmitted using quadrature phase modulation. Conventionally known methods for demodulating orthogonal phase modulated waves include a method of multiplying by an orthogonal carrier wave and extracting the modulated signal with a low-pass filter, and a method of using a Hilbert converter. An example of modulating and demodulating a color difference signal using the latter method is disclosed in Japanese Patent Laid-Open No. 62-298295.
上記従来技術は、直交搬送波を掛ける方法では、周波数
の和と差の成分が出力され、どちらか不要な信号成分を
取り除くフィルタが必要である。また、ヒルベルト変換
器を用いる方法では、不要成分を発生しないので、フィ
ルタは不要であるが、実信号を複素信号に変換するため
のヒルベルト変換器を実現するためのハードウェアが大
規模で複雑となり、従来の実信号による方法と較べて、
利点はなく、ともにハードウェアが多くなるという欠点
を有していた。In the above-mentioned conventional technology, in the method of multiplying orthogonal carrier waves, the sum and difference components of frequencies are output, and a filter is required to remove either unnecessary signal component. In addition, the method using a Hilbert transformer does not generate unnecessary components and does not require a filter, but the hardware required to implement the Hilbert transformer for converting a real signal into a complex signal is large-scale and complicated. , compared to the conventional method using real signals,
Both methods had the disadvantage of requiring more hardware.
本発明の目的は、複索係数フィルタを用いて、色信号の
ような直交変調された信号を復調する装置を提供するこ
とにあり、さらに従来の方法よりハードウェア量の少な
い復調回路を実現することにある。An object of the present invention is to provide a device for demodulating orthogonally modulated signals such as color signals using a multi-coefficient filter, and further to realize a demodulation circuit with less hardware than conventional methods. There is a particular thing.
CAMを解決するための手段〕
上記目的は、実信号の複素化に複素係数フィルタを用い
、さらに、標本化周波数を色副搬送波周波数f scの
4倍に選ぶことにより、複索係数フィルタの構成および
複索正弦波信号掛算器の構成を簡易化することにより、
達成される。Means for Solving CAM] The above purpose is to construct a complex coefficient filter by using a complex coefficient filter to complexize the real signal and further selecting the sampling frequency to be four times the color subcarrier frequency fsc. And by simplifying the configuration of the multi-line sine wave signal multiplier,
achieved.
複素係数フィルタは、実係数フィルタの周波数特性を周
波数シフトすることにより設計することができる。N次
の実係数のFIRフィルタの伝達関数をH(z)=ao
+azz−’+azz−”+−+a N Z−Nとする
。Z=ejcvTである。ここで周波数特性をfαだけ
シフトすると、2は
、 jhc(f−/a)T −j2πfaT=
z(z 、−1a)z=s
=zeとなる。これをH(z )に代入すると。A complex coefficient filter can be designed by frequency shifting the frequency characteristics of a real coefficient filter. The transfer function of the FIR filter with N-th real coefficients is H(z)=ao
+azz-'+azz-''+-+a N Z-N.Z=ejcvT.Here, if the frequency characteristic is shifted by fα, 2 becomes jhc(f-/a)T −j2πfaT=
z(z, -1a)z=s
=ze. Substituting this into H(z):
H(z’ )=ao+a+α−”z−’+aza−”z
−”+−・+aNcx″′r4z” =(2)となる、
すなわち、係数が複素数となったフィルタが得られる。H(z')=ao+a+α-"z-'+aza-"z
-"+-・+aNcx"'r4z" = (2),
In other words, a filter whose coefficients are complex numbers is obtained.
このフィルタに実信号を入力すると複素信号が得られ、
これに複索正弦波信号を掛けることにより直交復調が行
なわれる。When a real signal is input to this filter, a complex signal is obtained,
Orthogonal demodulation is performed by multiplying this by a compound sine wave signal.
入力直交変調波をa cosωot + bsinωO
tとすると。The input quadrature modulated wave is a cosωot + bsinωO
Let it be t.
これを複索化した信号は
x (t )=acosωot+bsinωot+j(
asinωat−bcosωot)となる。これに複素
正弦波 −jcvo″=cosωot −jsinωo
tを掛けると、出力y(t)はy(t)=x(t)・e
−”0t=a+j b −(3)となり、実部にa
、虚部にbが出力され、復調されることがわかる。さら
に、実信号処理におけるような、高調波の発生がない(
2O2成分)ので、これを除去するフィルタは不要であ
る。The signal obtained by converting this into a compound signal is x (t) = acosωot+bsinωot+j(
asinωat-bcosωot). Add to this the complex sine wave −jcvo″=cosωot −jsinωo
When multiplied by t, the output y(t) is y(t)=x(t)・e
−”0t=a+j b −(3), and the real part is a
, b is output to the imaginary part and demodulated. Furthermore, unlike in real signal processing, there is no generation of harmonics (
2O2 component), so a filter to remove this is not necessary.
以上の説明では、複素係数フィルタの周波数シフト量お
よび、複素正弦波の周波数は任意であった。ここで、標
本化周波数を色副搬送波周波数(f sc)の4倍に選
び1周波数シフト量をf sc とすると、複素係数フ
ィルタの係数および複素正弦波掛算が簡単になる。(1
)式において、T=4fsc=
Jα=fscとすると。In the above explanation, the frequency shift amount of the complex coefficient filter and the frequency of the complex sine wave were arbitrary. Here, if the sampling frequency is chosen to be four times the color subcarrier frequency (f sc ) and the amount of one frequency shift is f sc , the coefficients of the complex coefficient filter and complex sine wave multiplication become simple. (1
), if T=4fsc=Jα=fsc.
すると、
H(z’ )= ao −a2z−2+ atz
一番一・)+j(alz−1aaz−’+asz−’−
−)となり、係数は実数あるいは純虚数となるので。Then, H(z') = ao -a2z-2+ atz
Ichibanichi・)+j(alz-1aaz-'+asz-'-
-), and the coefficients are real numbers or pure imaginary numbers.
係数の個数は元の実係数フィルタの個数と変わらない、
従って実信号処理における低域通過フィルタ1個分のハ
ードウェアで実現でき、フィルタ1個分を縮減できる。The number of coefficients is the same as the number of original real coefficient filters,
Therefore, it can be realized with the hardware equivalent to one low-pass filter in actual signal processing, and the number of filters can be reduced.
(実信号処理ではフィルタが2個必要である。)
次に、複素正弦波掛算は(3)式をハードウェアで行な
うが、ωo=2πfsc、T= であるかfsc
”1+ J+ 1−+ j+1・・・ら、と
なる、すなわち、標本化周波数を4fscに選ぶと複索
正弦波の掛算は符号変換と、実部、虚部の入れ換えのみ
で実行でき、掛算器は不要となる。(Two filters are required in real signal processing.) Next, for complex sine wave multiplication, equation (3) is performed by hardware, but it is determined whether ωo=2πfsc, T= or fsc ”1+ J+ 1−+ j+1 . . . In other words, if the sampling frequency is selected to be 4 fsc, multiplication of the multi-line sine wave can be performed only by code conversion and exchanging the real and imaginary parts, and no multiplier is required.
以下、本発明の一実施例を図面により説明する。 An embodiment of the present invention will be described below with reference to the drawings.
第1図は本発明の一実施例を示す構成図である。FIG. 1 is a block diagram showing an embodiment of the present invention.
図において、1,2は複索係数フィルタ、3,4は複素
正弦波掛算器、5は加減算器、6,7は係数掛算器であ
る。カラー受像機で受信されたTV信号は、映像検波さ
れてYC分離され、搬送色差信号Cが第1図の入力端子
に印加される。ここで複素係数フィルタ1,2によって
各々複索信号化され、複素正弦波掛算器3,4によって
複索正弦波cosωsct+1sinωsctが掛けら
れて、信号を周波数シフトし、Izおよび2Iz、2Q
をそれぞれ出力する。2I2,2Qは係数掛算器6,7
により1/2を掛けられ、Izはさらに加減算器5によ
り、Izから減算されて工信号となる。信号スペクトル
図を第2図に示す。搬送色信号は第2−8図に示すよう
な帯域の異なる信号I、Qでf scを直交変調したも
のである。複素フィルタ2は3 、1 M Hzから4
、2 M Hz の信号を抜き出し、複素信号とす
る。複素フィルタ1は2 、1 M Hzから4 、2
M Hz の信号を抜き出し、複素化する。In the figure, 1 and 2 are complex coefficient filters, 3 and 4 are complex sine wave multipliers, 5 is an adder/subtractor, and 6 and 7 are coefficient multipliers. A TV signal received by a color receiver is image-detected and separated into Y and C signals, and a carrier color difference signal C is applied to the input terminal of FIG. Here, complex coefficient filters 1 and 2 convert them into complex signals, and complex sine wave multipliers 3 and 4 multiply them by a complex sine wave cosωsct+1sinωsct to shift the frequency of the signals, resulting in Iz, 2Iz, 2Q
Output each. 2I2, 2Q are coefficient multipliers 6, 7
is multiplied by 1/2, and Iz is further subtracted from Iz by an adder/subtractor 5 to obtain an engineering signal. A signal spectrum diagram is shown in FIG. The carrier color signal is obtained by orthogonally modulating fsc with signals I and Q having different bands as shown in FIGS. 2-8. Complex filter 2 is 3, 1 MHz to 4
, 2 MHz signals are extracted and made into complex signals. Complex filter 1 is 2.1 MHz to 4.2
The MHz signal is extracted and complexized.
複素正弦波掛算器3,4で周波数シフトすると第2−b
図のような信号が得られる。複素正弦波掛算器3から得
られる色信号工1は残留側波帯変調されているため2−
b図のようなスペクトルとなる。これを工2を減算する
ことにより、平坦なスペクトルをもつ広帯域工信号が復
調できる。When the frequency is shifted by complex sine wave multipliers 3 and 4, the second -b
The signal shown in the figure is obtained. Since the color signal signal 1 obtained from the complex sine wave multiplier 3 is vestigial sideband modulated, 2-
The spectrum will be as shown in figure b. By subtracting step 2 from this, a broadband signal with a flat spectrum can be demodulated.
第1図と同機能を有する従来回路を第3図に示す、第3
図で31.32は周波数混合器、33゜34.35は高
調波成分を除く低域通過フィルタ、36.37は係数掛
算器、38は加減算器である。A conventional circuit having the same function as that shown in Fig. 1 is shown in Fig. 3.
In the figure, 31.32 is a frequency mixer, 33.degree. 34.35 is a low-pass filter for removing harmonic components, 36.37 is a coefficient multiplier, and 38 is an adder/subtractor.
第3図で判るように、工信号を復調するために2つのL
PFが必要となり、しかもLPF34はLPF35とま
ったく同じ構成のものである。本発明は第1図の実施例
では従来例に較ベフィルタの1個少なくなっており、構
成ハードウェアの縮減が図れていることがわかる。なお
、第1図のフィルタ1と掛算器3は従来例のような方法
に置き換えて構成することもできる。As can be seen in Figure 3, two L
A PF is required, and the LPF 34 has exactly the same configuration as the LPF 35. In the embodiment of FIG. 1, the present invention has one less filter than the conventional example, and it can be seen that the configuration hardware can be reduced. Note that the filter 1 and multiplier 3 shown in FIG. 1 can also be constructed by replacing them with a conventional method.
本発明によれば、複素信号処理により復調を行なうので
、実信号処理におけるような不要成分の発生(高調波の
発生)がなく、これを除去するためのフィルタが不要と
なる。また従来のヒルベルト変換器による複素処理と異
なり、複索係数フィルタを用いるため、ハードウェアは
ごく小さく、実係数フィルタと変わらない。さらに、動
作標本化周波数を色副搬送波周波数の4倍に選ぶことに
より、複素正弦波掛算もスイッチと極性切換のみで実現
でき、色差信号復調回路の構成を簡単にできる効果があ
る。According to the present invention, since demodulation is performed by complex signal processing, there is no generation of unnecessary components (generation of harmonics) unlike in real signal processing, and a filter for removing these components is not required. Also, unlike complex processing using a conventional Hilbert transformer, since a multi-coefficient filter is used, the hardware is extremely small and is no different from a real coefficient filter. Furthermore, by selecting the operation sampling frequency to be four times the color subcarrier frequency, complex sine wave multiplication can be realized using only switches and polarity switching, which has the effect of simplifying the configuration of the color difference signal demodulation circuit.
第1図は本発明による色差信号復調回路の一実施例の構
成図、第2図は第1図の構成の動作を説明するスペクト
ル図、第3図は従来の色差信号復調回路図である。
1.2・・・複索係数フィルタ、3,4・・・複素正弦
波掛算器、5,38・・・加減算器、6,7,36゜3
7・・・係数掛算器、31.32・・・周波数混合器、
第 1
口
第 22FIG. 1 is a block diagram of an embodiment of a color difference signal demodulation circuit according to the present invention, FIG. 2 is a spectrum diagram illustrating the operation of the structure of FIG. 1, and FIG. 3 is a diagram of a conventional color difference signal demodulation circuit. 1.2... Multiple coefficient filter, 3, 4... Complex sine wave multiplier, 5, 38... Addition/subtraction device, 6, 7, 36°3
7... Coefficient multiplier, 31.32... Frequency mixer,
Part 1 Part 22
Claims (1)
二つの色差信号I、Qを再生する色差信号復調回路にお
いて、複素係数フィルタにより入力搬送色差信号を複素
信号化し、これに色副搬送波周波数の複素正弦波信号を
掛け、出力複素信号の実数部をI、虚数部をQとするこ
とを特徴とする色差信号復調回路。 2、請求項第1記載の色差信号復調回路において、上記
複素係数フィルタに入る入力信号を標本化するとき上記
標本化周波数を色副搬送波周波数の4倍とすることを特
徴とする色差信号復調回路。 3、請求項第1記載の色差信号復調回路において、入力
搬送色差信号から、第2のI信号のみを復調する回路を
設け、上記第2のI信号から、上記I信号の1/2を引
算した信号を得ることにより、広帯域なI信号を復調す
ることを特徴とする色差信号復調回路。[Claims] 1. Demodulating the carrier color difference signal in a color TV receiver,
In a color difference signal demodulation circuit that reproduces two color difference signals I and Q, the input carrier color difference signal is converted into a complex signal by a complex coefficient filter, and this is multiplied by a complex sine wave signal of the color subcarrier frequency to obtain the real part of the output complex signal. A color difference signal demodulation circuit characterized in that I and an imaginary part are Q. 2. The color difference signal demodulation circuit according to claim 1, wherein the sampling frequency is set to four times the color subcarrier frequency when sampling the input signal input to the complex coefficient filter. . 3. The color difference signal demodulation circuit according to claim 1, further comprising a circuit for demodulating only the second I signal from the input carrier color difference signal, and subtracting 1/2 of the I signal from the second I signal. A color difference signal demodulation circuit is characterized in that it demodulates a wideband I signal by obtaining a calculated signal.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP15912688A JPH0210989A (en) | 1988-06-29 | 1988-06-29 | Color difference signal demodulation circuit |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP15912688A JPH0210989A (en) | 1988-06-29 | 1988-06-29 | Color difference signal demodulation circuit |
Publications (1)
Publication Number | Publication Date |
---|---|
JPH0210989A true JPH0210989A (en) | 1990-01-16 |
Family
ID=15686807
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP15912688A Pending JPH0210989A (en) | 1988-06-29 | 1988-06-29 | Color difference signal demodulation circuit |
Country Status (1)
Country | Link |
---|---|
JP (1) | JPH0210989A (en) |
Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
KR100807990B1 (en) * | 2006-08-23 | 2008-03-07 | (주) 넥스트칩 | Frequency modulator for recovering secam chrominance signal and method thereof |
KR100818279B1 (en) * | 2005-12-16 | 2008-04-01 | 삼성전자주식회사 | Complex number transversal filter and complex frequency converter |
JP2013102670A (en) * | 2011-10-21 | 2013-05-23 | Daihen Corp | Control apparatus for single-phase power conversion apparatus |
-
1988
- 1988-06-29 JP JP15912688A patent/JPH0210989A/en active Pending
Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
KR100818279B1 (en) * | 2005-12-16 | 2008-04-01 | 삼성전자주식회사 | Complex number transversal filter and complex frequency converter |
KR100807990B1 (en) * | 2006-08-23 | 2008-03-07 | (주) 넥스트칩 | Frequency modulator for recovering secam chrominance signal and method thereof |
JP2013102670A (en) * | 2011-10-21 | 2013-05-23 | Daihen Corp | Control apparatus for single-phase power conversion apparatus |
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