JP6410681B2 - Power converter - Google Patents
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- JP6410681B2 JP6410681B2 JP2015137249A JP2015137249A JP6410681B2 JP 6410681 B2 JP6410681 B2 JP 6410681B2 JP 2015137249 A JP2015137249 A JP 2015137249A JP 2015137249 A JP2015137249 A JP 2015137249A JP 6410681 B2 JP6410681 B2 JP 6410681B2
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この発明は、誘導電動機の２次抵抗の温度変化を補正するようにした電力変換装置に関する。 The present invention relates to a power converter configured to correct a temperature change of a secondary resistance of an induction motor.
誘導電動機のベクトル制御において、出力トルクを適切に制御するためには、励磁電流及びモータの２次時定数を適正に制御系に与える必要がある。２次時定数は２次インダクタンスと２次抵抗で構成されるが、２次抵抗は温度によって大きく変化するので注意が必要である。このため、誘導電動機のロータ温度を温度センサによって検出し、２次抵抗を補正することが従来実施されてきた。また、温度センサの取り付けが困難な用途に対しては、例えば、トルク電流指令値とトルク電流実際値の偏差に従って２次抵抗を補正することが行われている。（例えば特許文献１参照。）。 In the vector control of the induction motor, in order to appropriately control the output torque, it is necessary to appropriately give the excitation current and the secondary time constant of the motor to the control system. The secondary time constant is composed of a secondary inductance and a secondary resistance, but care must be taken because the secondary resistance varies greatly with temperature. For this reason, it has been practiced to detect the rotor temperature of the induction motor by a temperature sensor and correct the secondary resistance. For applications where it is difficult to attach the temperature sensor, for example, the secondary resistance is corrected according to the deviation between the torque current command value and the actual torque current value. (For example, refer to Patent Document 1).
特許文献１に示された手法は、トルク電流を制御する通常のベクトル制御には用いることができない。通常のベクトル制御においては、トルク電流指令値とトルク電流実際値の偏差が零になるように誘導電動機の電圧を制御しているからである。 The technique disclosed in Patent Document 1 cannot be used for normal vector control for controlling torque current. This is because in normal vector control, the voltage of the induction motor is controlled so that the deviation between the torque current command value and the actual torque current value becomes zero.
この発明は上記問題点に鑑みてなされたもので、温度センサなしで通常のベクトル制御に使用することができる２次抵抗補正手段を有する電力変換装置を提供することを目的とする。 The present invention has been made in view of the above problems, and an object thereof is to provide a power converter having a secondary resistance correction unit that can be used for normal vector control without a temperature sensor.
上記目的を達成するために、本発明の電力変換装置は、交流電圧を直流電圧に変換するコンバータと、前記コンバータの出力電圧を交流に変換して誘導電動機を駆動するインバータと、前記インバータの出力電流を検出する電流検出手段と、前記交流電動機の速度を実質的に検出する速度検出手段と、前記インバータの出力を制御するインバータ制御部とを具備し、前記インバータ制御部は、前記速度検出手段によって得られる速度帰還が所定の速度基準となるように制御してトルク基準及びトルク電流基準を出力する速度制御手段と、前記電流検出手段の検出電流を３相−２相変換してトルク帰還電流及び磁束帰還電流を得る３相−２相変換手段と、前記トルク電流基準と、前記トルク帰還電流とを比較してトルク軸電圧基準を出力するトルク軸電流制御手段と、与えられた励磁電流基準と、前記磁束帰還電流とを比較して磁束軸電圧基準を出力する磁束軸電流制御手段と、前記トルク軸電圧基準及び前記磁束軸電圧基準を２相−３相変換して前記インバータの各相の電圧基準を得る２相−３相変換手段と、前記インバータの各相の電圧基準をＰＷＭ制御して前記インバータを構成するスイッチング素子へのゲート信号を出力するＰＷＭ制御手段と、前記トルク電流基準と前記励磁電流基準を用いてスベリ周波数を求めるスベリ演算手段と、前記スベリ周波数と、前記速度帰還から得られる回転周波数を加算して得られるインバータ周波数を積分して前記３相−２相変換手段及び前記２相−３相変換手段の基準位相を得る積分手段と、前記速度帰還と前記トルク基準を乗算して演算上の電動機出力を得る乗算手段と、前記電流検出手段の各相の検出電流と前記インバータの各相の電圧基準を相ごとに掛け合わせて３相分加算する瞬時電力演算手段と、前記瞬時電力演算手段の出力から、電動機損失設定手段で設定された電動機損失を減算する減算手段と、前記減算手段の出力から前記乗算手段の出力を減算した偏差が最小となるように調節制御して２次抵抗補正値を出力する２次抵抗制御手段とを具備し、前記２次抵抗補正値に基準となる２次抵抗値を加算した２次抵抗を前記スベリ演算手段の演算式に用いる２次抵抗としたことを特徴としている。 In order to achieve the above object, a power converter according to the present invention includes a converter that converts an AC voltage into a DC voltage, an inverter that converts an output voltage of the converter into an AC and drives an induction motor, and an output of the inverter Current detection means for detecting current, speed detection means for substantially detecting the speed of the AC motor, and an inverter control unit for controlling the output of the inverter, wherein the inverter control unit is provided with the speed detection means. A speed control means for outputting the torque reference and the torque current reference by controlling so that the speed feedback obtained by the above becomes a predetermined speed reference, and a torque feedback current by converting the detection current of the current detection means by three-phase to two-phase. And a three-phase to two-phase conversion means for obtaining a magnetic flux feedback current, the torque current reference, and the torque feedback current are compared to output a torque shaft voltage reference. The magnetic axis current control means, the given excitation current reference, the magnetic flux axis current control means for comparing the magnetic flux feedback current and outputting the magnetic flux axis voltage reference, the torque axis voltage reference and the magnetic flux axis voltage reference. Two-phase to three-phase conversion means for obtaining a voltage reference for each phase of the inverter by performing two-phase to three-phase conversion, and a gate to a switching element constituting the inverter by PWM control of the voltage reference for each phase of the inverter PWM control means for outputting a signal, slip calculation means for obtaining a slip frequency using the torque current reference and the excitation current reference, an inverter obtained by adding the slip frequency and the rotation frequency obtained from the speed feedback Starring by multiplying an integration means for obtaining a reference phase of the three-phase two-phase conversion means and the two-phase three-phase conversion means by integrating the frequency, the torque reference and the speed feedback Multiplication means for obtaining the above motor output; instantaneous power calculation means for multiplying the detected current of each phase of the current detection means by the voltage reference of each phase of the inverter for each phase and adding three phases; and the instantaneous power Subtracting means for subtracting the motor loss set by the motor loss setting means from the output of the calculating means, and adjusting and controlling so that the deviation obtained by subtracting the output of the multiplying means from the output of the subtracting means is minimized. Secondary resistance control means for outputting a resistance correction value, and using a secondary resistance obtained by adding a secondary resistance value as a reference to the secondary resistance correction value as an arithmetic expression of the slip calculation means; It is characterized by that.
この発明によれば、温度センサなしで通常のベクトル制御に使用することができる２次抵抗補正手段を有する電力変換装置を提供することが可能となる。 According to the present invention, it is possible to provide a power conversion device having secondary resistance correction means that can be used for normal vector control without a temperature sensor.
以下、図面を参照して本発明の実施例について説明する。 Embodiments of the present invention will be described below with reference to the drawings.
以下、本発明の実施例１に係る電力変換装置を、図１を参照して説明する。 Hereinafter, the power converter concerning Example 1 of the present invention is explained with reference to FIG.
図１は、本発明に係る電力変換装置の回路構成図である。交流電源１から電力変換器２のコンバータ２１に交流が給電され、これを所望の電圧の直流に変換し、平滑コンデンサ２２Ｐ、２２Ｎを介してインバータ２３に与える。インバータ２３は直流を交流電圧に変換して誘導電動機３を駆動する。図１においてはコンバータ２１及びインバータ２３は３レベル回路としているが、２レベルであっても４レベル以上の多レベル回路であっても良い。インバータ２３を構成するパワーデバイスはインバータ制御部６から与えられるゲート信号ＧＡＴＥによってオンオフ制御されている。交流電動機３には速度検出器４が取り付けられており、この出力は速度帰還信号ＳＰ＿Ｆとしてインバータ制御部６に与えられる。また、インバータ２３の出力側には電流検出器５が設けられ、この出力も電流帰還信号ＩＵ＿Ｆ、ＩＶ＿Ｆ、ＩＷ＿Ｆとしてインバータ制御部６に与えられる。尚、電力変換器２と図示しないコンバータ制御部とインバータ制御部６とで電力変換装置は構成されている。 FIG. 1 is a circuit configuration diagram of a power converter according to the present invention. An alternating current is fed from the alternating current power source 1 to the converter 21 of the power converter 2, converted into a direct current of a desired voltage, and supplied to the inverter 23 via the smoothing capacitors 22 </ b> P and 22 </ b> N. The inverter 23 converts the direct current into an alternating voltage and drives the induction motor 3. In FIG. 1, the converter 21 and the inverter 23 are three-level circuits, but may be two-level circuits or multi-level circuits of four levels or more. The power device constituting the inverter 23 is on / off controlled by a gate signal GATE given from the inverter control unit 6. A speed detector 4 is attached to the AC motor 3, and this output is given to the inverter controller 6 as a speed feedback signal SP_F. Further, the current detector 5 is provided on the output side of the inverter 23, and this output is also given to the inverter control unit 6 as current feedback signals IU_F, IV_F, and IW_F. The power converter 2, the converter controller (not shown), and the inverter controller 6 constitute a power converter.
次にインバータ制御部６の内部構成について説明する。 Next, the internal configuration of the inverter control unit 6 will be described.
電流検出器５の出力は３相／２相変換器６１に与えられ、３相の電流帰還信号ＩＵ＿Ｆ、ＩＶ＿Ｆ、ＩＷ＿Ｆを２軸の電流帰還信号ＩＤ＿Ｆ及びＩＱ＿Ｆに変換する。ここでＩＤ＿Ｆ及びＩＱ＿Ｆは直流量である。 The output of the current detector 5 is supplied to a three-phase / two-phase converter 61 to convert the three-phase current feedback signals IU_F, IV_F, and IW_F into biaxial current feedback signals ID_F and IQ_F. Here, ID_F and IQ_F are DC amounts.
外部から与えられた速度基準ＳＰ＿Ｒは、速度検出器４からの速度帰還信号ＳＰ＿Ｆと比較され、両者の偏差が速度制御器６２の入力となる。速度制御器６２においてはこの偏差が最小となるように調節制御し、トルク基準Ｔ＿Ｒを出力する。トルク基準Ｔ＿Ｒは除算器６４で磁束演算器６３の出力である磁束基準ＦＬ＿Ｒによって除算され、トルク電流基準ＩＱ＿Ｒとなる。このトルク電流基準ＩＱ＿Ｒは、３相−２相変換器６１で変換して得られたトルク電流帰還ＩＱ＿Ｆと比較され、両者の偏差が電流制御器６５の入力となる。 The speed reference SP_R given from the outside is compared with the speed feedback signal SP_F from the speed detector 4, and the deviation between them is an input to the speed controller 62. The speed controller 62 performs adjustment control so that this deviation is minimized, and outputs a torque reference T_R. The torque reference T_R is divided by the divider 64 by the magnetic flux reference FL_R that is the output of the magnetic flux calculator 63 to become the torque current reference IQ_R. This torque current reference IQ_R is compared with the torque current feedback IQ_F obtained by conversion by the three-phase to two-phase converter 61, and the deviation between the two becomes the input to the current controller 65.
速度帰還信号ＳＰ＿Ｆは磁束演算器６３に与えられる。磁束演算器６３は基底速度までは一定で、それを超えると速度に反比例して減少するような弱め界磁特性を有する磁束基準ＦＬ＿Ｒを出力する。磁束基準ＦＬ＿Ｒは上述のように除算器６４に与えられると共に磁束電流演算器６８に与えられる。磁束電流演算器６８は磁束基準ＦＬ＿Ｒに応じて磁束電流基準ＩＤ＿Ｒを出力する。この磁束電流基準ＩＤ＿Ｒは、３相−２相変換器６１で変換して得られた磁束電流帰還ＩＤ＿Ｆと比較され、両者の偏差が電流制御器６５の他方の入力となる。電流制御器６５はこの磁束電流の偏差と前述のトルク電流の偏差の夫々が最小となるように調節制御し、磁束電圧基準ＥＤ＿Ｒとトルク電圧基準ＥＱ＿Ｒを夫々出力する。すなわち、電流制御器６５はトルク電流制御と磁束電流制御を独立して行う。磁束電圧基準ＥＤ＿Ｒ及びトルク電圧基準ＥＱ＿Ｒは２相−３相変換器６６に与えられ、２相−３相変換器６６は３相電圧基準ＥＵ＿Ｒ、ＥＶ＿Ｒ、ＥＷ＿Ｒを出力する。この３相電圧基準ＥＵ＿Ｒ、ＥＶ＿Ｒ、ＥＷ＿ＲはＰＷＭ制御器６７に与えられる。ＰＷＭ制御器６７はインバータ２３の各相の出力電圧が３相電圧基準ＥＵ＿Ｒ、ＥＶ＿Ｒ、ＥＷ＿Ｒとなるようにインバータ２３の各パワーデバイスに対して、ＰＷＭ変調されたゲート信号ＧＡＴＥを供給する。上記説明における３相−２相変換器６１及び２相−３相変換器６６の相変換のための基準位相の生成方法について以下説明する。 The speed feedback signal SP_F is given to the magnetic flux calculator 63. The magnetic flux calculator 63 outputs a magnetic flux reference FL_R having a field weakening characteristic that is constant up to the base velocity and decreases in inverse proportion to the velocity beyond that. The magnetic flux reference FL_R is supplied to the divider 64 and the magnetic flux current calculator 68 as described above. The magnetic flux current calculator 68 outputs the magnetic flux current reference ID_R according to the magnetic flux reference FL_R. This magnetic flux current reference ID_R is compared with the magnetic flux current feedback ID_F obtained by conversion by the three-phase to two-phase converter 61, and the deviation between the two becomes the other input of the current controller 65. The current controller 65 adjusts and controls the deviation of the magnetic flux current and the deviation of the torque current described above to minimize the magnetic flux voltage reference ED_R and the torque voltage reference EQ_R. That is, the current controller 65 performs torque current control and magnetic flux current control independently. The magnetic flux voltage reference ED_R and the torque voltage reference EQ_R are supplied to the two-phase / three-phase converter 66, and the two-phase / three-phase converter 66 outputs the three-phase voltage references EU_R, EV_R, and EW_R. The three-phase voltage references EU_R, EV_R, and EW_R are given to the PWM controller 67. The PWM controller 67 supplies a PWM-modulated gate signal GATE to each power device of the inverter 23 so that the output voltage of each phase of the inverter 23 becomes the three-phase voltage reference EU_R, EV_R, EW_R. A reference phase generation method for phase conversion of the three-phase to two-phase converter 61 and the two-phase to three-phase converter 66 in the above description will be described below.
トルク電流基準ＩＱ＿Ｒと励磁電流基準ＩＤ＿Ｒを入力とするスベリ演算器６９は通常以下の（１）式に従って誘導電動機３のスベリ周波数Ｓを求める。 The slip calculator 69 that receives the torque current reference IQ_R and the excitation current reference ID_R normally obtains the slip frequency S of the induction motor 3 according to the following equation (1).
Ｓ＝（Ｒ２／Ｌ２）×ＩＱ＿Ｒ／ＩＤ＿Ｒ・・・（１）
ここで、Ｒ２とＬ２は夫々誘導電動機３の二次抵抗とインダクタンスである。（１）式から得られるスベリ周波数Ｓに、速度帰還信号ＳＰ＿Ｆから得られる回転周波数を加算器７０によって加算することによってインバータ２３の出力周波数が得られ、この出力周波数を積分器７１によって積分することによって基準位相θが得られる。この基準位相θに基づいて３相−２相変換器６１及び２相−３相変換器６６の相変換を行う。
S = (R2 / L2) × IQ_R / ID_R (1)
Here, R2 and L2 are the secondary resistance and inductance of the induction motor 3, respectively. The output frequency of the inverter 23 is obtained by adding the rotational frequency obtained from the speed feedback signal SP_F by the adder 70 to the smooth frequency S obtained from the equation (1), and integrating the output frequency by the integrator 71. Gives the reference phase θ. The three-phase to two-phase converter 61 and the two-phase to three-phase converter 66 perform phase conversion based on the reference phase θ.
次に、２次抵抗Ｒ２の補正方法について説明する。 Next, a method for correcting the secondary resistance R2 will be described.
速度帰還信号ＳＰ＿Ｆとトルク基準Ｔ＿Ｒを乗算器７２で乗算することにより、演算による誘導電動機３の出力パワーＭＯＴ＿ＰＯＷＥＲが得られる。一方、電流帰還信号ＩＵ＿Ｆ、ＩＶ＿Ｆ、ＩＷ＿Ｆと３相電圧基準ＥＵ＿Ｒ、ＥＶ＿Ｒ、ＥＷ＿Ｒを瞬時電力演算器７３に与えることによって誘導電動機３の実際の入力パワーＩ＿ＰＯＷＥＲが得られる。ここで、瞬時電力演算器７３は各相の電流帰還信号と電圧基準を乗算して各々の乗算結果を３相分加算する構成とする。そしてこの入力パワーＩ＿ＰＯＷＥＲから電動機損失設定器７４によって設定された電動機損失ＭＬを減算して実際の電動機の出力パワーＯ＿ＰＯＷＥＲが得られる。このＯ＿ＰＯＷＥＲからＭＯＴ＿ＰＯＷＥＲを減算した偏差を２次抵抗制御器７５に与える。２次抵抗制御器７５は入力された偏差が最小となるように調節制御して２次抵抗補正値ΔＲ２を出力する。そして、基準温度における２次抵抗の値Ｒ２＊にこの２次抵抗補正値ΔＲ２を加算器７６で加算し、得られた２次抵抗を上記（１）式の２次抵抗Ｒ２とする。尚、２次抵抗制御器７５は積分制御型の増幅器を使用するが、その時定数は２次抵抗の温度に作用する熱時定数を考慮して十分長くしておくことが好ましい。 By multiplying the speed feedback signal SP_F and the torque reference T_R by the multiplier 72, the output power MOT_POWER of the induction motor 3 by calculation is obtained. On the other hand, the actual input power I_POWER of the induction motor 3 is obtained by supplying the current feedback signals IU_F, IV_F, IW_F and the three-phase voltage references EU_R, EV_R, EW_R to the instantaneous power calculator 73. Here, the instantaneous power calculator 73 is configured to multiply the current feedback signal of each phase by the voltage reference and add the multiplication results for three phases. The actual motor output power O_POWER is obtained by subtracting the motor loss ML set by the motor loss setter 74 from the input power I_POWER. A deviation obtained by subtracting MOT_POWER from O_POWER is given to the secondary resistance controller 75. The secondary resistance controller 75 adjusts and controls the input deviation to be minimum, and outputs a secondary resistance correction value ΔR2. Then, this secondary resistance correction value ΔR2 is added to the value R2 * of the secondary resistance at the reference temperature by the adder 76, and the obtained secondary resistance is defined as the secondary resistance R2 in the above equation (1). Although the secondary resistance controller 75 uses an integral control type amplifier, it is preferable that the time constant be sufficiently long in consideration of the thermal time constant acting on the temperature of the secondary resistance.
ここで、２次抵抗補正値ΔＲ２の符号について考察する。図１で、実際の電動機の出力パワーＯ＿ＰＯＷＥＲが演算の出力パワーＭＯＴ＿ＰＯＷＥＲより大きい場合を考える。スベリ周波数Ｓが小さい通常の運転領域では誘導電動機３の出力トルクＴはスベリ周波数Ｓに比例し、そして（１）式より２次抵抗Ｒ２に比例する。従ってこの場合は、演算上用いた２次抵抗Ｒ２が実際の２次抵抗より小さかったためであるので、この場合は、２次抵抗補正値ΔＲ２を加算する補正を行ってスベリ演算器６９の２次抵抗とすれば良いことになる。 Here, the sign of the secondary resistance correction value ΔR2 will be considered. Consider the case where the actual output power O_POWER of the electric motor is larger than the calculated output power MOT_POWER in FIG. In a normal operation region where the slip frequency S is small, the output torque T of the induction motor 3 is proportional to the slip frequency S, and is proportional to the secondary resistance R2 from the equation (1). Therefore, in this case, since the secondary resistance R2 used in the calculation is smaller than the actual secondary resistance, in this case, the secondary resistance correction value ΔR2 is corrected to perform a correction to add the secondary resistance of the slip calculator 69. It would be good to use resistance.
磁束電流演算器６８は基本的には磁束の飽和を考慮した入出力特性とするが、磁束の変化に対して２次抵抗Ｒ２を含む演算を行う場合もある。その場合はスベリ演算器６９で用いた２次抵抗Ｒ２を使用すれば良い。 The magnetic flux current calculator 68 basically has an input / output characteristic considering the saturation of the magnetic flux, but there is a case where a calculation including the secondary resistance R2 is performed on the change of the magnetic flux. In that case, the secondary resistor R2 used in the slip calculator 69 may be used.
このように、本願の実施例１によれば、誘導電動機３の出力パワーについて、２次抵抗が温度によって変化したときに変化する演算上のＭＯＴ＿ＰＯＷＥＲと、実際の出力パワーであるＯ＿ＰＯＷＥＲの二つを比較することによって２次抵抗の温度変化を補正することが可能となる。この比較を誘導電動機３の出力パワーではなく、インバータ２３の入力パワーについて行うことも原理的には可能であるが、その場合インバータ２３の実際の入力パワーを求めるための直流電流検出器が必要となる。このような直流電流検出器は通常は設ける必要がないので、本願の手法の方が、構成がより簡略化され経済的である。 As described above, according to the first embodiment of the present application, the output power of the induction motor 3 is divided into two in terms of the operation MOT_POWER that changes when the secondary resistance changes with temperature, and O_POWER that is the actual output power. By comparing, it becomes possible to correct the temperature change of the secondary resistance. In principle, this comparison can be performed not on the output power of the induction motor 3 but on the input power of the inverter 23. In this case, however, a DC current detector for obtaining the actual input power of the inverter 23 is required. Become. Since such a DC current detector does not normally need to be provided, the method of the present application is simpler and more economical in configuration.
尚、電動機損失設定器７４は、誘導電動機３の運転が定格運転状態に限定されるような場合は、その状態における一定値を与えることが簡明である。より細かく制御する場合については実施例２以下に述べる。 When the operation of the induction motor 3 is limited to the rated operation state, the motor loss setting device 74 is easy to give a constant value in that state. More detailed control will be described below in the second embodiment.
図２は本発明の実施例２に係る電力変換装置の電動機損失設定器７４Ａのブロック構成図である。 FIG. 2 is a block diagram of a motor loss setting unit 74A of the power conversion apparatus according to the second embodiment of the present invention.
誘導電動機３の損失ＭＬは、与えられた電圧とそのときの電流によって決まる。この場合の電圧と電流は、正規化された値であっても良い。例えば、図２の電動機損失設定器７４Ａに示すように、電流に関しては、トルク電流基準ＩＱ＿Ｒと励磁電流基準ＩＤ＿ＲのＲＭＳ値Ｉ（二乗平均平方根）をＲＭＳ演算器７４１で演算して求める。すなわち、Ｉ＝{１／２・（ＩＱ＿Ｒ）２＋１／２・（ＩＤ＿Ｒ）２}１／２である。同様にＲＭＳ演算器７４２によって、トルク電圧基準ＥＱ＿Ｒと励磁電流基準ＥＤ＿ＲのＲＭＳ値Ｅを求める。そしてこれらＩとＥをパラメータとした電動機損失ＭＬの損失テーブル７４３を予め電動機の試験データ等を参照して準備しておき、この損失テーブル７４３に基づいて電動機損失ＭＬを決定する。このようにすれば、誘導電動機３の運転速度に見合う電動機電圧、あるいは誘導電動機３の負荷の大きさに見合う電動機電流が変化した場合であっても電動機損失ＭＬを精度良く求めることが可能となる。 The loss ML of the induction motor 3 is determined by a given voltage and a current at that time. The voltage and current in this case may be normalized values. For example, as shown in the motor loss setting unit 74A in FIG. 2, the RMS value I (root mean square) of the torque current reference IQ_R and the excitation current reference ID_R is calculated by the RMS calculator 741 as to the current. That is, I = {1/2 · (IQ_R) 2 + 1/2 · (ID_R) 2 } 1/2 . Similarly, the RMS calculator 742 calculates the RMS value E of the torque voltage reference EQ_R and the excitation current reference ED_R. Then, a motor loss ML loss table 743 using these I and E as parameters is prepared in advance by referring to motor test data and the like, and the motor loss ML is determined based on the loss table 743. In this way, it is possible to accurately obtain the motor loss ML even when the motor voltage corresponding to the operating speed of the induction motor 3 or the motor current corresponding to the load of the induction motor 3 changes. .
図３は本発明の実施例３に係る電力変換装置の電動機損失設定器７４Ｂのブロック構成図である。この電動機損失設定器７４Ｂが図２の電動機損失設定器７４Ａと異なる点は、損失テーブル７４３に代えて損失演算器７４４を設けた点である。電動機損失ＭＬは基本的に銅損と鉄損と固定損（機械損＋漂遊損）とから成り、銅損は電流の２乗に比例し、鉄損は電圧に比例し、固定損はほぼ一定である。従って、実施例２と同様に、規格化された電流Ｉ及び電圧Ｅを入力とし、損失演算器７４４の演算式をＭＬ＝Ｋ１・Ｉ２＋Ｋ２・Ｅ＋Ｃとする。Ｋ１、Ｋ２及びＣは誘導電動機３の固有の定数である。 FIG. 3 is a block diagram of the motor loss setting unit 74B of the power conversion apparatus according to the third embodiment of the present invention. The motor loss setting unit 74B is different from the motor loss setting unit 74A of FIG. 2 in that a loss calculator 744 is provided instead of the loss table 743. The motor loss ML basically consists of copper loss, iron loss, and fixed loss (mechanical loss + stray loss). Copper loss is proportional to the square of current, iron loss is proportional to voltage, and fixed loss is almost constant. It is. Accordingly, as in the second embodiment, the standardized current I and voltage E are input, and the calculation formula of the loss calculator 744 is ML = K1 · I 2 + K2 · E + C. K1, K2 and C are intrinsic constants of the induction motor 3.
この実施例３によれば、誘導電動機３の固有の定数による演算を電動機損失設定器７４Ｂで行うことによって、比較的簡単な手法で精度良く電動機損失ＭＬを求めることができる。 According to the third embodiment, the motor loss ML can be obtained with high accuracy by a relatively simple method by performing the calculation based on the inherent constant of the induction motor 3 by the motor loss setting unit 74B.
以上、いくつかの実施例について説明したが、これらの実施例は例として提示したものであり、発明の範囲を限定することは意図していない。これら新規な実施例は、その他の様々な形態で実施されることが可能であり、発明の要旨を逸脱しない範囲で、種々の省略、置き換え、変更を行うことができる。これら実施例やその変形は、発明の範囲や要旨に含まれるとともに、特許請求の範囲に記載された発明とその均等の範囲に含まれる。 Although several embodiments have been described above, these embodiments are presented as examples, and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.
例えば、磁束電流基準ＩＤ＿Ｒは、通常の弱め界磁を行わない、あるいは磁束の飽和を考慮しない、また磁束の変化の追従を考慮しない用途であれば一定値であっても良い。 For example, the magnetic flux current reference ID_R may be a constant value as long as the field weakening is not performed, the saturation of the magnetic flux is not considered, and the follow-up of the change in the magnetic flux is not considered.
また、実施例２及び３では正規化された電動機電流Ｉ及び電動機電圧Ｅを用いたが、特に正規化された値を用いなくても良い。例えば相電圧ＩＵ＿Ｆの実効値を電動機電圧としても良く、また相電流ＩＵ＿Ｆの実効値を電動機電流としても良い。また、必ずしも実効値とする必要はなく、電動機電圧と電動機電流の大きさを表す量であれば良い。 In the second and third embodiments, the normalized motor current I and motor voltage E are used. However, normalized values may not be used. For example, the effective value of the phase voltage IU_F may be the motor voltage, and the effective value of the phase current IU_F may be the motor current. Moreover, it is not necessarily required to be an effective value, and any amount that represents the magnitude of the motor voltage and the motor current may be used.
１ 交流電源
２ 電力変換器
２１ コンバータ
２２Ｐ、２２Ｎ 平滑コンデンサ
３ 誘導電動機
４ 速度検出器
５ 電流検出器
６ インバータ制御部
７ 電流検出器
６１ ３相２相変換器
６２ 速度制御器
６３ 磁束演算器
６４ 除算器
６５ 電流制御器
６６ ２相３相変換器
６７ ＰＷＭ制御器
６８ 励磁電流演算器
６９ スベリ演算器
７０ 加算器
７１ 積分器
７２ 乗算器
７３ 電力演算器
７４、７４Ａ、７４Ｂ 電動機損失設定器
７５ ２次抵抗制御器
７６ 加算器
７４１、７４２ ＲＭＳ演算器
７４３ 損失テーブル
７４４ 損失演算器
DESCRIPTION OF SYMBOLS 1 AC power supply 2 Power converter 21 Converter 22P, 22N Smoothing capacitor 3 Induction motor 4 Speed detector 5 Current detector 6 Inverter control part 7 Current detector 61 Three-phase two-phase converter 62 Speed controller 63 Magnetic flux calculator 64 Division 65 Current controller 66 Two-phase three-phase converter 67 PWM controller 68 Excitation current calculator 69 Subtractor calculator 70 Adder 71 Integrator 72 Multiplier 73 Power calculators 74, 74A, 74B Motor loss setting unit 75 Secondary Resistance controller 76 Adders 741 and 742 RMS calculator 743 Loss table 744 Loss calculator
Claims (5)
前記コンバータの出力電圧を交流に変換して誘導電動機を駆動するインバータと、
前記インバータの出力電流を検出する電流検出手段と、
前記交流電動機の速度を実質的に検出する速度検出手段と、
前記インバータの出力を制御するインバータ制御部と
を具備し、
前記インバータ制御部は、
前記速度検出手段によって得られる速度帰還が所定の速度基準となるように制御してトルク基準及びトルク電流基準を出力する速度制御手段と、
前記電流検出手段の検出電流を３相−２相変換してトルク帰還電流及び磁束帰還電流を得る３相−２相変換手段と、
前記トルク電流基準と、前記トルク帰還電流とを比較してトルク軸電圧基準を出力するトルク軸電流制御手段と、
与えられた励磁電流基準と、前記磁束帰還電流とを比較して磁束軸電圧基準を出力する磁束軸電流制御手段と、
前記トルク軸電圧基準及び前記磁束軸電圧基準を２相−３相変換して前記インバータの各相の電圧基準を得る２相−３相変換手段と、
前記インバータの各相の電圧基準をＰＷＭ制御して前記インバータを構成するスイッチング素子へのゲート信号を出力するＰＷＭ制御手段と、
前記トルク電流基準と前記励磁電流基準を用いてスベリ周波数を求めるスベリ演算手段と、
前記スベリ周波数と、前記速度帰還から得られる回転周波数を加算して得られるインバータ周波数を積分して前記３相−２相変換手段及び前記２相−３相変換手段の基準位相を得る積分手段と、
前記速度帰還と前記トルク基準を乗算して演算上の電動機出力を得る乗算手段と、
前記電流検出手段の各相の検出電流と前記インバータの各相の電圧基準を相ごとに掛け合わせて３相分加算する瞬時電力演算手段と、
前記瞬時電力演算手段の出力から、電動機損失設定手段で設定された電動機損失を減算する減算手段と、
前記減算手段の出力から前記乗算手段の出力を減算した偏差が最小となるように調節制御して２次抵抗補正値を出力する２次抵抗制御手段と
を具備し、
前記２次抵抗補正値に基準となる２次抵抗値を加算した２次抵抗を前記スベリ演算手段の演算式に用いる２次抵抗としたことを特徴とする電力変換装置。
A converter that converts AC voltage to DC voltage;
An inverter that drives the induction motor by converting the output voltage of the converter into alternating current;
Current detection means for detecting an output current of the inverter;
Speed detecting means for substantially detecting the speed of the AC motor;
An inverter control unit for controlling the output of the inverter;
The inverter control unit
Speed control means for controlling the speed feedback obtained by the speed detection means to be a predetermined speed reference and outputting a torque reference and a torque current reference;
Three-phase to two-phase conversion means for obtaining torque feedback current and magnetic flux feedback current by three-phase to two-phase conversion of the detection current of the current detection means;
Torque axis current control means for comparing the torque current reference and the torque feedback current to output a torque axis voltage reference;
A magnetic flux axis current control means for comparing a given excitation current reference with the magnetic flux feedback current and outputting a magnetic flux axis voltage reference;
Two-phase to three-phase conversion means for obtaining a voltage reference for each phase of the inverter by performing two-phase to three-phase conversion on the torque axis voltage reference and the magnetic flux axis voltage reference;
PWM control means for PWM-controlling the voltage reference of each phase of the inverter and outputting a gate signal to a switching element constituting the inverter;
A slip calculation means for determining a slip frequency using the torque current reference and the excitation current reference;
Integrating means for integrating the slip frequency and the inverter frequency obtained by adding the rotational frequency obtained from the speed feedback to obtain a reference phase of the three-phase to two-phase converting means and the two-phase to three-phase converting means; ,
Multiplication means for multiplying the speed feedback and the torque reference to obtain an operational motor output;
Instantaneous power calculation means for multiplying the detected current of each phase of the current detection means by the voltage reference of each phase of the inverter for each phase and adding three phases;
Subtracting means for subtracting the motor loss set by the motor loss setting means from the output of the instantaneous power calculation means,
Secondary resistance control means for adjusting and controlling so that a deviation obtained by subtracting the output of the multiplication means from the output of the subtraction means is minimized, and outputting a secondary resistance correction value;
2. A power converter according to claim 1, wherein a secondary resistance obtained by adding a secondary resistance value as a reference to the secondary resistance correction value is used as a secondary resistance used in an arithmetic expression of the slip calculation means.
基底速度以上で弱め界磁となるような磁束演算手段の出力を入力とする励磁電流演算手段の出力から得るようにしたことを特徴とする請求項１に記載の電力変換装置。 The excitation current reference is
2. The power conversion device according to claim 1, wherein the power conversion device is obtained from an output of an excitation current calculation means that receives an output of a magnetic flux calculation means that becomes a field weakening at a base speed or higher.
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