JP2017146267A - Non-contact type voltage detector - Google Patents

Non-contact type voltage detector Download PDF

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JP2017146267A
JP2017146267A JP2016030049A JP2016030049A JP2017146267A JP 2017146267 A JP2017146267 A JP 2017146267A JP 2016030049 A JP2016030049 A JP 2016030049A JP 2016030049 A JP2016030049 A JP 2016030049A JP 2017146267 A JP2017146267 A JP 2017146267A
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博男 今村
Hiroo Imamura
博男 今村
裕司 田中
Yuji Tanaka
裕司 田中
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East Japan Railway Co
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Abstract

PROBLEM TO BE SOLVED: To provide a non-contact type voltage detector capable of detecting the charged state of a charging unit to which a low voltage such as DC 100 V is applied in a non-contact manner and at a high sensibility.SOLUTION: A non-contact type voltage detector (100) detects the charged state of a charging unit charged with a DC voltage in a non-contact manner. The non-contact type voltage detector (100) includes: an electrostatic antenna (120) charged by electrostatic induction; a detection circuit (110) for detecting the charged state of the electrostatic antenna; and a determination unit (130) for determining the charged state of the charging unit to which the electrostatic antenna is made close on the basis of a detection result of the detection circuit. The electrostatic antenna (120) includes: a first electrode (121A) for detection to be connected to an input part of the detection circuit; and a second electrode (121B) for detection to be connected to a ground potential of the detection circuit.SELECTED DRAWING: Figure 1

Description

本発明は、直流電圧により充電される充電部の充電状態を非接触に検出する非接触型検電器に関する。   The present invention relates to a non-contact type voltage detector that detects a charging state of a charging unit charged by a DC voltage in a non-contact manner.

以前より、電線路に電気が供給されているか否かを判別するために検電器が用いられている。一般に交流の電線路を対象とする検電器には接触型と非接触型の二種類がある。一方、直流の電線路を非接触で検出することは難しく、直流用の検電器は接触型が一般的である。
本発明者らは、先に、直流式電気鉄道の「き電回路」を検出対象とする非接触型の検電器の開発を行った（非特許文献１を参照）。「き電回路」には１５００Ｖなど高い直流電圧が印加される。この非接触型検電器は、静電誘導により帯電する検出用電極と、検出用電極に伝送路を介して接続された可変容量手段と、可変容量手段の容量を周期的に変化させる発振器とを有する。そして、検出用電極を充電部（例えば架線）に向け、可変容量手段の容量を周期的に変化させることで、検出用電極の帯電状態に応じて振幅が変化する交流信号を生成し、この交流信号の振幅に基づいて充電部の充電状態を判定する。 For some time, a voltage detector has been used to determine whether or not electricity has been supplied to the electrical line. Generally, there are two types of voltage detectors for AC electric lines, contact type and non-contact type. On the other hand, it is difficult to detect a direct current electric line in a non-contact manner, and a direct current detector is generally a contact type.
The present inventors have previously developed a non-contact type voltage detector that detects a “feed circuit” of a DC electric railway (see Non-Patent Document 1). A high DC voltage such as 1500 V is applied to the “feed circuit”. This non-contact type voltage detector includes a detection electrode that is charged by electrostatic induction, variable capacitance means connected to the detection electrode via a transmission line, and an oscillator that periodically changes the capacitance of the variable capacitance means. Have. Then, by directing the detection electrode toward the charging part (for example, an overhead line) and periodically changing the capacity of the variable capacitance means, an AC signal whose amplitude changes according to the charging state of the detection electrode is generated. The charging state of the charging unit is determined based on the amplitude of the signal.

一般に、検出用電極に生じた静的な帯電状態を検出しようとすると、検出用電極から検出回路へわずかに電荷が移動することにより、検出用電極の帯電状態が時間の経過に伴って薄まってしまい、安定した検出が阻害される。一方、上記の非接触型検電器によれば、充電部の電圧に変化がなくても、検電器内で動的な処理を行って検出用電極と可変容量手段とが接続された伝送路に交流信号を生成する。これにより、検出用電極の帯電状態が時間の経過に伴って薄まってしまうことが回避され、検出用電極の帯電状態に応じた振幅を有する交流信号により充電部の充電状態を安定的且つ高感度で検出することが可能となる。   In general, when trying to detect a static charged state generated in the detection electrode, the charge moves slightly from the detection electrode to the detection circuit, so that the charged state of the detection electrode fades with time. As a result, stable detection is hindered. On the other hand, according to the above non-contact type voltage detector, even if there is no change in the voltage of the charging unit, a dynamic process is performed in the voltage detector and the transmission line is connected to the detection electrode and the variable capacitance means. Generate an AC signal. As a result, it is avoided that the charged state of the detection electrode fades with time, and the charged state of the charging unit is stably and highly sensitive by an AC signal having an amplitude corresponding to the charged state of the detection electrode. Can be detected.

田中裕司 福原卓 下原光幸 西村一実 中島等 今村博男、「直流電圧を非接触で確認する方法の開発」、電気学会研究会資料、一般社団法人電気学会、２０１５年５月２０日、ｐ．７−１０Yuji Tanaka Taku Fukuhara Mitsuyuki Shimohara Kazumi Nishimura, etc. Hiroo Imamura, “Development of a method for checking DC voltage in a non-contact manner”, IEEJ Technical Report, The Institute of Electrical Engineers of Japan, May 20, 2015, p . 7-10

近年、「き電回路」の電圧と比べて低圧の直流電圧、例えば直流１００Ｖのバッテリー電源などによって動作する設備が増えている。従来、このような設備においては、メンテナンスの際などに接触型の検電器を用いて充電部（例えば電線路）の充電状態を検出することが行われている。接触型の検電器を用いた場合、プローブが充電部の電極と意図しない金属部分とに触れて充電部をショートさせてしまうなど、電気事故を引き起こす潜在的な危険性がある。
そこで、本発明者らは、先に開発した「き電回路」を検出対象とする直流用の非接触型検電器を用いて、直流１００Ｖのような低圧の直流電圧が印加される充電部の充電状態を、非接触で検出できないか検証を行った。しかしながら、このような充電部では、思ったような感度で充電状態を検出することは困難であった。 In recent years, there is an increasing number of facilities that operate with a low-voltage DC voltage, for example, a DC 100V battery power supply, compared to the voltage of the “feed circuit”. Conventionally, in such equipment, a charging state of a charging unit (for example, an electric line) is detected using a contact-type voltage detector during maintenance or the like. When a contact-type voltage detector is used, there is a potential danger of causing an electrical accident, such as the probe touching the electrode of the charging unit and an unintended metal part to short-circuit the charging unit.
Therefore, the present inventors use a non-contact type voltage detector for DC, which is a previously developed “feeding circuit”, to detect a charging unit to which a low-voltage DC voltage such as 100 V DC is applied. It was verified whether the state of charge could not be detected without contact. However, in such a charging unit, it has been difficult to detect the state of charge with the desired sensitivity.

検証実験においては、検出用電極（静電アンテナとも呼ぶ）を充電部（陽極側の電線路など）に近づけ、非接触型検電器の筐体を基準電位点（陰極側の電線路など）に近づけたときに、検電器に電圧検出の反応が生じることが分かった。また、検証実験においては、検出対象の充電部に誤って検出用電極が接触した場合に、両者が絶縁被覆されていた場合でも、検出用電極の帯電状態や内部回路の状態に変化が生じ、その後、回路を初期化しないと、正常な回路動作が得られなくなることが分かった。これは絶縁被覆のわずかな衝突により、ピエゾ効果或いは摩擦によって僅かな帯電が生じることに起因すると推定される。
本発明は、例えば直流１００Ｖのような低圧の電圧が印加される充電部の充電状態を非接触に且つ感度良く検出することができる非接触型検電器を提供することを目的とする。 In the verification experiment, the detection electrode (also referred to as an electrostatic antenna) is brought close to the charging part (such as the electric wire on the anode side), and the housing of the non-contact type voltage detector is placed at the reference potential point (such as the electric wire on the cathode side). It was found that a voltage detection reaction occurred in the voltage detector when approaching. In addition, in the verification experiment, when the detection electrode is accidentally contacted with the charging part to be detected, even when both are covered with insulation, the charging state of the detection electrode and the state of the internal circuit change, Thereafter, it was found that normal circuit operation cannot be obtained unless the circuit is initialized. This is presumed to be caused by slight charging caused by the piezoelectric effect or friction due to slight collision of the insulating coating.
An object of the present invention is to provide a non-contact type voltage detector capable of detecting a charging state of a charging unit to which a low voltage such as DC 100V is applied in a non-contact manner and with high sensitivity.

本発明は、上記目的を達成するため、
直流電圧によって充電される充電部の充電状態を非接触で検出する非接触型検電器であって、
静電誘導により帯電する静電アンテナと、
前記静電アンテナの帯電状態を検出する検出回路と、
前記検出回路の検出結果に基づいて前記静電アンテナが近づけられた前記充電部の充電状態を判定する判定部と、
を備え、
前記静電アンテナは、前記検出回路の入力部に導通される第１検出用電極と、前記検出回路の接地電位と導通される第２検出用電極とを有することを特徴としている。 In order to achieve the above object, the present invention
A non-contact type voltage detector for detecting a charging state of a charging unit charged by a DC voltage in a non-contact manner,
An electrostatic antenna charged by electrostatic induction;
A detection circuit for detecting a charged state of the electrostatic antenna;
A determination unit that determines a charging state of the charging unit to which the electrostatic antenna is brought close based on a detection result of the detection circuit;
With
The electrostatic antenna includes a first detection electrode that is electrically connected to an input portion of the detection circuit, and a second detection electrode that is electrically connected to a ground potential of the detection circuit.

ここで、「充電部」とは、例えば直流電圧が印加される電線路や電極のようなもののことである。
この構成によれば、静電アンテナに、検出回路の入力部に導通される第１検出用電極に加えて、検出回路の接地電位と導通される第２検出用電極が含まれる。よって、検出対象の充電部とその陰極側の配線又は電極に対して、第１検出用電極と第２検出用電極とを適宜に配置することで、静電アンテナに充電部の充電状態を検出可能な大きさの帯電を発生させることができる。これにより、充電部の充電状態を非接触に且つ高感度に検出することができる。また、高感度に検出できることから、検出時に充電部と静電アンテナとの距離を大きくとることができ、両者が接触して正常な回路動作が阻害されるという課題を回避できる。 Here, the “charging unit” is, for example, an electric line or an electrode to which a DC voltage is applied.
According to this configuration, the electrostatic antenna includes the second detection electrode that is electrically connected to the ground potential of the detection circuit, in addition to the first detection electrode that is electrically connected to the input portion of the detection circuit. Therefore, the charging state of the charging unit is detected by the electrostatic antenna by appropriately arranging the first detection electrode and the second detection electrode with respect to the detection target charging unit and the wiring or electrode on the cathode side thereof. A possible amount of charge can be generated. Thereby, the charging state of the charging unit can be detected in a non-contact and highly sensitive manner. Moreover, since it can detect with high sensitivity, the distance of a charging part and an electrostatic antenna can be taken at the time of detection, and the subject that both contact and normal circuit operation | movement is inhibited can be avoided.

ここで、前記第１検出用電極および前記第２検出用電極は共に一方に長い平面形状部を含み、
前記第１検出用電極の前記平面形状部と前記第２検出用電極の前記平面形状部とは、長手方向に見てＶ字形状となるように、互いの長辺部が近接し、互いのもう一方の長辺部が離間しているとよい。
この構成によれば、検出対象の充電部として陽極側の電線路があり、この電線路と陰極側の電線路とが離間して配置されている場合に、これら２つの電線路の間に第１検出用電極と第２検出用電極とを挿入するように配置することができる。このような配置により、充電部に直流電圧が印加されている場合に、静電アンテナに検出可能な大きさの帯電を発生させて、その充電状態を高感度に検出することができる。 Here, both the first detection electrode and the second detection electrode include a long planar shape portion on one side,
The long side portions of the first detection electrode and the flat shape portion of the second detection electrode are close to each other so that they are V-shaped when viewed in the longitudinal direction. The other long side portion is preferably separated.
According to this configuration, when there is an anode-side electric wire as the charging unit to be detected and the electric wire and the cathode-side electric wire are arranged apart from each other, the second electric wire is between the two electric wires. The first detection electrode and the second detection electrode can be arranged to be inserted. With such an arrangement, when a DC voltage is applied to the charging unit, it is possible to generate a charge having a magnitude that can be detected by the electrostatic antenna and to detect the charged state with high sensitivity.

また、前記第１検出用電極および前記第２検出用電極は共に一方に長い平面形状部を含み、
前記第１検出用電極の前記平面形状部と前記第２検出用電極の前記平面形状部とは、前記充電部を双方の間へ挿入可能なように離間しているとよい。
この構成によれば、検出対象の充電部として陽極側の電線路があり、この電線路と陰極側の電線路とが近接して配置されている場合に、これら２つの電線路の両側を囲うように第１検出用電極と第２検出用電極とを配置することができる。このような配置により、充電部に直流電圧が印加されている場合に、静電アンテナに検出可能な大きさの帯電を発生させて、その充電状態を高感度に検出することができる。 The first detection electrode and the second detection electrode both include a long planar shape portion on one side,
The planar shape portion of the first detection electrode and the planar shape portion of the second detection electrode may be separated so that the charging portion can be inserted between the two.
According to this configuration, when there is an anode-side electric wire as a charging unit to be detected and the electric wire and the cathode-side electric wire are arranged close to each other, both sides of these two electric wires are enclosed. Thus, the first detection electrode and the second detection electrode can be arranged. With such an arrangement, when a DC voltage is applied to the charging unit, it is possible to generate a charge having a magnitude that can be detected by the electrostatic antenna and to detect the charged state with high sensitivity.

また、前記第１検出用電極および前記第２検出用電極は共に一方に長い１つ又は複数の平面形状部を含み、
前記静電アンテナは、
前記第１検出用電極の何れかの前記平面形状部と前記第２検出用電極の何れかの前記平面形状部とが、長手方向に見てＶ字形状となるように、互いの長辺部が近接し、互いのもう一方の長辺部が離間した第１形態と、
前記第１検出用電極の何れかの前記平面形状部と前記第２検出用電極の何れかの前記平面形状部とが、前記充電部を双方の間へ挿入可能なように離間した第２形態と、
に変形可能に構成されているとよい。
この構成によれば、検出対象の形態が異なる場合（例えば、接近した平行線と離間した平行線）に、対象の形態に応じて、静電アンテナの形態を変えることで、充電部の充電状態を高感度に検出することができる。 The first detection electrode and the second detection electrode both include one or a plurality of planar shape portions that are long on one side,
The electrostatic antenna is
The long side portions of the first detection electrode and the second detection electrode so that the planar shape portion of the first detection electrode and the planar shape portion of the second detection electrode are V-shaped when viewed in the longitudinal direction. Are close to each other, and the other long side of each other is separated,
A second configuration in which the planar shape portion of any of the first detection electrodes and the planar shape portion of any of the second detection electrodes are separated so that the charging portion can be inserted between the two. When,
It may be configured to be deformable.
According to this configuration, when the form of the detection target is different (for example, an approaching parallel line and a spaced parallel line), the charging state of the charging unit is changed by changing the form of the electrostatic antenna according to the form of the target. Can be detected with high sensitivity.

さらに、前記検出回路は、
前記入力部から電荷を導く第１伝送路と、
前記第１伝送路に接続された第１可変容量手段と、
前記第１可変容量手段の容量を周期的に変化させる発振器と、
を有し、
前記第１可変容量手段の容量の変化により前記静電アンテナの帯電状態に応じて振幅が変化する交流信号を前記第１伝送路に生成し、
前記判定部は、前記交流信号の振幅変化が位相変化に拡大変換された信号の位相に基づいて前記充電部の充電状態を判定するように構成するとよい。 Furthermore, the detection circuit includes:
A first transmission line for guiding charges from the input unit;
First variable capacitance means connected to the first transmission line;
An oscillator that periodically changes the capacitance of the first variable capacitance means;
Have
An AC signal whose amplitude changes according to the charging state of the electrostatic antenna due to a change in the capacity of the first variable capacitance means is generated in the first transmission line;
The determination unit may be configured to determine a charging state of the charging unit based on a phase of a signal obtained by enlarging and converting an amplitude change of the AC signal into a phase change.

一般に、静電誘導により生じた微小な帯電状態をそのまま検出しようとすると、静電アンテナから検出回路への電荷の移動によって、検出用電極の帯電状態が時間の経過に伴って薄まってしまい、安定した検出が阻害される。しかしながら、上記構成によれば、発振器と第１可変容量手段との作用によって交流信号を生成して静電アンテナの帯電状態が薄まることを回避できる。さらに、第１可変容量手段の容量を周期的に変化させて生成した交流信号は、静電アンテナの帯電状態に応じて振幅と位相が変化するが、これらは微小な変化である。そこで、上記構成によれば、判定部は交流信号の振幅変化が位相変化に拡大変換された信号の位相に基づいて充電部の充電状態を判定する。これにより、微小帯電領域すなわち弱電界領域において、充電部の充電状態を正しく判定することができる。また、振幅変化を直接検出する場合と異なり、位相変化に拡大変換された信号の位相を検出するために、高い分解能のＡＤ変換器とフーリエ変換などのディジタル信号処理のための高速処理能力を持つＣＰＵ（中央演算処理装置）が不要であり、部品コストの低減を図れる。   In general, if it is attempted to detect a minute charged state caused by electrostatic induction as it is, the charged state of the detection electrode fades with time due to the movement of charge from the electrostatic antenna to the detection circuit, and is stable. Detection is inhibited. However, according to the above-described configuration, it is possible to avoid that the AC antenna is generated by the action of the oscillator and the first variable capacitance means and the charged state of the electrostatic antenna is reduced. Further, the AC signal generated by periodically changing the capacitance of the first variable capacitance means changes in amplitude and phase according to the charged state of the electrostatic antenna, but these are minute changes. So, according to the said structure, a determination part determines the charge condition of a charging part based on the phase of the signal by which the amplitude change of the alternating current signal was enlarged and converted into the phase change. Thereby, the charging state of the charging unit can be correctly determined in the minute charging region, that is, the weak electric field region. In addition, unlike the case of directly detecting the amplitude change, it has a high resolution AD converter and a high-speed processing capability for digital signal processing such as Fourier transform in order to detect the phase of the signal expanded and converted into the phase change. A CPU (central processing unit) is not required, and the cost of parts can be reduced.

さらに、前記検出回路は、
前記第１伝送路と並列に接続される第２伝送路と、
前記第２伝送路に接続されて前記発振器により容量が周期的に変化する第２可変容量手段と、
前記第２伝送路上に接続されて直流成分の信号が伝わるのを阻止する直流阻止フィルタと、
前記第１伝送路に出力される交流信号と前記第２伝送路に出力される交流信号との差を出力する差動増幅器と、
を有し、
前記判定部は、前記差動増幅器の出力と前記第２伝送路に出力される交流信号との位相差に基づいて、前記静電アンテナが近づけられた前記充電部の充電状態を判定するとよい。 Furthermore, the detection circuit includes:
A second transmission line connected in parallel with the first transmission line;
A second variable capacitance means connected to the second transmission line and having a capacitance periodically changed by the oscillator;
A DC blocking filter that is connected to the second transmission line and blocks transmission of a DC component signal;
A differential amplifier that outputs a difference between an AC signal output to the first transmission path and an AC signal output to the second transmission path;
Have
The determination unit may determine a charging state of the charging unit to which the electrostatic antenna is brought close based on a phase difference between an output of the differential amplifier and an AC signal output to the second transmission line.

この構成によれば、第２伝送路により出力される交流信号として、静電アンテナの帯電状態の影響がなく、回路の温度特性や手振れなどによる外的要因の影響が現れた交流信号が生成される。よって、第１伝送路の交流信号と第２伝送路の交流信号とに基づき判定を行うことで、外的要因による位相の変化をキャンセルして正確な判定を行うことができる。   According to this configuration, an AC signal that is not affected by the charging state of the electrostatic antenna and that is influenced by external factors such as circuit temperature characteristics and camera shake is generated as the AC signal output from the second transmission line. The Therefore, by performing the determination based on the AC signal of the first transmission path and the AC signal of the second transmission path, it is possible to cancel the phase change due to an external factor and perform an accurate determination.

本発明によれば、例えば直流１００Ｖのような低圧の電圧が印加される充電部の充電状態を非接触に且つ感度良く検出することができる。   According to the present invention, it is possible to detect a charging state of a charging unit to which a low voltage such as DC 100V is applied, for example, in a non-contact manner and with high sensitivity.

本発明の第１実施の形態の非接触型検電器を示す構成図である。It is a lineblock diagram showing the non-contact type electric detector of a 1st embodiment of the present invention. 図１の検出回路の具体的な一例を示す回路図である。FIG. 2 is a circuit diagram showing a specific example of the detection circuit of FIG. 1. 図１の検出回路の動作原理を説明する図である。It is a figure explaining the principle of operation of the detection circuit of FIG. 図１のノッチフィルタの好適な回路例を示す回路構成図である。It is a circuit block diagram which shows the suitable circuit example of the notch filter of FIG. 図２の抵抗分圧器の一例を示す回路図である。FIG. 3 is a circuit diagram illustrating an example of a resistance voltage divider in FIG. 2. 図２の抵抗分圧器兼位相差ゼロ点調整要素の一例を示す回路図である。FIG. 3 is a circuit diagram showing an example of a resistance voltage divider and phase difference zero point adjustment element of FIG. 2. 位相差検出回路の動作を説明するもので位相差が小さいときの状態を示す波形図である。FIG. 6 is a waveform diagram for explaining the operation of the phase difference detection circuit and showing a state when the phase difference is small. 位相差検出回路の動作を説明するもので位相差が大きいときの状態を示す波形図である。FIG. 6 is a waveform diagram for explaining the operation of the phase difference detection circuit and showing a state when the phase difference is large. 図２の回路のシミュレーション結果と理論式である式（１ｂ）を用いた数値計算結果の第１の比較を表わすグラフである。FIG. 3 is a graph showing a first comparison between a simulation result of the circuit of FIG. 2 and a numerical calculation result using a theoretical formula (1b). FIG. 図２の回路のシミュレーション結果と理論式である式（１ｅ）を用いた数値計算結果の第２の比較を表わすグラフである。FIG. 3 is a graph showing a second comparison between a simulation result of the circuit of FIG. 2 and a numerical calculation result using a theoretical formula (1e). FIG. 図１の静電アンテナの構造を示す斜視図（Ａ）と正面図（Ｂ）である。It is the perspective view (A) and front view (B) which show the structure of the electrostatic antenna of FIG. 第１変形例の静電アンテナを示す斜視図（Ａ）と正面図（Ｂ）である。It is the perspective view (A) and front view (B) which show the electrostatic antenna of a 1st modification. 第２変形例の静電アンテナの第１形態を示す正面図（Ａ）と第２形態を示す正面図（Ｂ）である。It is the front view (A) which shows the 1st form of the electrostatic antenna of a 2nd modification, and the front view (B) which shows a 2nd form. 本発明の第２実施の形態の検出回路を示す構成図である。It is a block diagram which shows the detection circuit of 2nd Embodiment of this invention.

以下、本発明の各実施の形態について図面を参照して詳細に説明する。
（第１実施の形態）
図１は、本発明の第１実施の形態の非接触型検電器を示す構成図である。
本発明の第１実施の形態の非接触型検電器１００は、例えば直流１００Ｖのような低圧の直流電圧が印加される充電部（例えばバッテリ電源の電線路など）の充電状態を非接触に検出する装置である。非接触型検電器１００は、静電アンテナ１２０、検出回路１１０、判定部１３０、出力部１４０、および各部に動作電圧を供給する電源部１５０を備える。 Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.
(First embodiment)
FIG. 1 is a configuration diagram showing the non-contact type voltage detector according to the first embodiment of the present invention.
The non-contact type voltage detector 100 according to the first embodiment of the present invention detects a charging state of a charging unit (for example, a battery power supply line) to which a low-voltage DC voltage such as DC 100V is applied in a non-contact manner. It is a device to do. The non-contact type voltage detector 100 includes an electrostatic antenna 120, a detection circuit 110, a determination unit 130, an output unit 140, and a power supply unit 150 that supplies an operating voltage to each unit.

静電アンテナ１２０は、検出対象の充電部に近接したときに、充電部からの静電誘導によって帯電するアンテナである。静電アンテナ１２０は、検出回路１１０の入力部に導通される第１検出用電極１２１Ａと、検出回路１１０の接地電位に導通される第２検出用電極１２１Ｂとを有する。静電アンテナ１２０の具体的な構造については後述する。
検出回路１１０は、帯域阻止フィルタとしてのノッチフィルタ１１５と、ノッチフィルタ１１５の出力端子側に設けられた一対の可変容量素子１１６Ａ，１１６Ｂと、可変容量素子１１６Ａ，１１６Ｂの容量値を周期的に変化させる発振器１１７と、可変容量素子１１６Ａ，１１６Ｂの出力電圧（交流信号）の位相差を検出する位相差検出部１１８とを備える。また、ノッチフィルタ１１５と可変容量素子１１６Ｂとの間には、直流成分を遮断し交流成分のみを通過させるＡＣ結合用の蓄電器（コンデンサ）１１９Ｃが設けられている。蓄電器（コンデンサ）１１９Ｃと可変容量素子１１６Ｂとの接続ノードと、検出回路１１０の接地電位ＧＮＤ１との間には抵抗器Ｒ５が接続され、この抵抗器Ｒ５とコンデンサＣとによって直流阻止フィルタが構成されている。 The electrostatic antenna 120 is an antenna that is charged by electrostatic induction from the charging unit when approaching the charging unit to be detected. The electrostatic antenna 120 includes a first detection electrode 121A that is conducted to the input portion of the detection circuit 110, and a second detection electrode 121B that is conducted to the ground potential of the detection circuit 110. A specific structure of the electrostatic antenna 120 will be described later.
The detection circuit 110 periodically changes the capacitance values of the notch filter 115 as a band rejection filter, the pair of variable capacitance elements 116A and 116B provided on the output terminal side of the notch filter 115, and the variable capacitance elements 116A and 116B. And a phase difference detector 118 that detects a phase difference between output voltages (AC signals) of the variable capacitors 116A and 116B. In addition, an AC coupling capacitor (capacitor) 119C is provided between the notch filter 115 and the variable capacitance element 116B to block the DC component and allow only the AC component to pass. A resistor R5 is connected between a connection node between the capacitor (capacitor) 119C and the variable capacitance element 116B and the ground potential GND1 of the detection circuit 110. The resistor R5 and the capacitor C constitute a DC blocking filter. ing.

さらに、ノッチフィルタ１１５と可変容量素子１１６Ａ，１１６Ｂとの間には、それぞれ高抵抗値（例えば１０ＭΩ）を有する抵抗器Ｒ４ａ，Ｒ４ｂが接続されている。この抵抗器Ｒ４ａ，Ｒ４ｂは、ノッチフィルタ１１５の出力と可変容量素子１１６Ａ，１１６Ｂとの間、さらには可変容量素子１１６Ａ，１１６Ｂ相互間を、検出対象の信号の周波数を含む高い周波数領域において交流的に分離するためのものである。
さらに、可変容量素子１１６Ａ，１１６Ｂと位相差検出部１１８の２つの入力端子の間には、それぞれ交流結合用の蓄電器１１９Ａ，１１９Ｂが接続されている。 Further, resistors R4a and R4b each having a high resistance value (for example, 10 MΩ) are connected between the notch filter 115 and the variable capacitance elements 116A and 116B. The resistors R4a and R4b are AC-shaped between the output of the notch filter 115 and the variable capacitors 116A and 116B, and between the variable capacitors 116A and 116B in a high frequency region including the frequency of the signal to be detected. It is for separating.
Further, AC coupling capacitors 119A and 119B are connected between the two input terminals of the variable capacitance elements 116A and 116B and the phase difference detection unit 118, respectively.

図２は、図１の検出回路の具体的な一例を示す回路図である。
図２に示すように、可変容量素子１１６Ａ，１１６Ｂとしては、例えばＭＯＳ電界効果トランジスタのゲート容量（ゲート電極−半導体基板間の容量）を適用できる。ＭＯＳ電界効果トランジスタのソース端子を基準電位点に接続し、ドレイン端子に発振器１１７で生成された電圧（サイン波）を印加することで、ゲート容量の容量値を変化させることができる。なお、ＭＯＳ電界効果トランジスタの代わりにバラクタダイオードなど他の素子や回路を使用しても良い。
発振器１１７は、所定の周波数のサイン波（正弦波）を生成する。所定の周波数としては、商用交流電源の周波数（５０Ｈｚまたは６０Ｈｚ）やその整数倍の周波数の影響を受けにくい周波数を選択するのが望ましい。 FIG. 2 is a circuit diagram showing a specific example of the detection circuit of FIG.
As shown in FIG. 2, for example, a gate capacitance (capacitance between a gate electrode and a semiconductor substrate) of a MOS field effect transistor can be applied as the variable capacitance elements 116A and 116B. The capacitance value of the gate capacitance can be changed by connecting the source terminal of the MOS field effect transistor to the reference potential point and applying the voltage (sine wave) generated by the oscillator 117 to the drain terminal. Other elements and circuits such as varactor diodes may be used instead of the MOS field effect transistors.
The oscillator 117 generates a sine wave (sine wave) having a predetermined frequency. As the predetermined frequency, it is desirable to select a frequency that is not easily affected by the frequency of commercial AC power (50 Hz or 60 Hz) or an integer multiple thereof.

ここで、可変容量素子１１６Ａ，１１６Ｂを設け、発振器１１７によって容量値を変化させる構成を採用した理由を説明する。
図３は、図１の検出回路の動作原理を説明する図を示す。
先ず、検出用電極に静電誘導された静的な電荷の量を検出することで充電部の充電状態（例えば電線路の停電／活線）を検出する電界検出方式の検電器について説明する。この方式では、検出用電極を充電部に近接した直後は検出用電極に電界の強さに応じた電荷が静電誘導される。しかしながら、検出用電極に接続される実際のアンプの入力インピーダンスは無限大にすることはできず有限の値を持つこととなる。そのため、時間が経過するとリークにより検出用電極から電荷が抜けてしまい、電界の強さを正確に検出することができない。 Here, the reason why the variable capacitance elements 116A and 116B are provided and the capacitance value is changed by the oscillator 117 will be described.
FIG. 3 is a diagram for explaining the operation principle of the detection circuit of FIG.
First, a description will be given of an electric field detection type voltage detector that detects a charged state of a charging unit (for example, a power failure / hot line of a power line) by detecting the amount of static charge electrostatically induced in the detection electrode. In this method, immediately after the detection electrode is brought close to the charging unit, a charge corresponding to the strength of the electric field is electrostatically induced in the detection electrode. However, the input impedance of the actual amplifier connected to the detection electrode cannot be infinite and has a finite value. For this reason, when time elapses, the charge escapes from the detection electrode due to leakage, and the strength of the electric field cannot be accurately detected.

一方、本実施の形態の非接触型検電器１００では、図３に示すように、第１検出用電極１２１Ａに接続された可変容量素子１１６Ａ，１１６Ｂを設け、その容量値が周期的に変化する。かかる構成によれば、可変容量素子１１６Ａ，１１６Ｂの容量値が変化すると、第１検出用電極１２１Ａに誘導された電荷が第１検出用電極１２１Ａと可変容量素子１１６Ａ，１１６Ｂとの間を往復移動することとなり、それによって可変容量素子１１６Ａ，１１６Ｂとその駆動手段（発振器１１７）とが一種の静電式発電機として動作し、静電アンテナ１２０に生じた静電誘導の大きさに応じて振幅や位相が変化する交流信号が生成される。このような動作過程では、第１検出用電極１２１Ａの帯電量が電荷の抜けによって次第に減少したりしないため、時間が経過しても静電誘導の大きさに応じた振幅や位相の交流信号が出力される。そして、このような交流信号によって、充電部の充電状態を正確に検出することができる。   On the other hand, in the non-contact type voltage detector 100 of the present embodiment, as shown in FIG. 3, the variable capacitance elements 116A and 116B connected to the first detection electrode 121A are provided, and the capacitance values thereof periodically change. . According to this configuration, when the capacitance values of the variable capacitance elements 116A and 116B change, the charges induced in the first detection electrode 121A reciprocate between the first detection electrode 121A and the variable capacitance elements 116A and 116B. As a result, the variable capacitance elements 116A and 116B and their driving means (oscillator 117) operate as a kind of electrostatic generator, and the amplitude depends on the magnitude of electrostatic induction generated in the electrostatic antenna 120. Or an alternating signal whose phase changes. In such an operation process, since the charge amount of the first detection electrode 121A does not gradually decrease due to the loss of electric charge, an AC signal having an amplitude or phase corresponding to the magnitude of electrostatic induction is generated even after a lapse of time. Is output. And the charging state of a charging part can be correctly detected by such an AC signal.

図４に、図１のノッチフィルタ１１５として好適な２−Ｔ型フィルタの回路例を示す。
帯電状態に応じて変化する交流信号を生成して充電部の充電状態を正確に判定するには、ノイズとなるその他の交流成分が検出回路１１０に混入することを抑制する必要がある。一方、充電部に印加される直流電圧は、例えば商用交流電源などから生成していることが多く、それ故、検出回路１１０に商用交流電源の周波数成分又はその高調波成分がノイズとして混入する状況が生じやすい。 FIG. 4 shows a circuit example of a 2-T filter suitable as the notch filter 115 in FIG.
In order to generate an AC signal that changes in accordance with the charging state and accurately determine the charging state of the charging unit, it is necessary to prevent other AC components that cause noise from entering the detection circuit 110. On the other hand, the DC voltage applied to the charging unit is often generated from, for example, a commercial AC power supply. Therefore, the frequency component of the commercial AC power supply or its harmonic component is mixed in the detection circuit 110 as noise. Is likely to occur.

ノッチフィルタ１１５は、このようなノイズの混入を防ぐために、商用交流電源の周波数成分とその高調波成分を除去もしくは低減するために設けられている。ノッチフィルタ１１５は、具体的には、図４（Ａ）に示すように３個の抵抗素子Ｒ１〜Ｒ３と３個の容量素子Ｃ１〜Ｃ３により構成された単段ノッチフィルタＮＦを、図４（Ｂ）に示すように、３個縦続接続して構成したものである。図４（Ａ）の単段ノッチフィルタＮＦを構成する素子は、１０ＭΩ１％品と１００ｐＦ１％品の２種類のみであり、これらを直列あるいは並列に接続してフィルタを構成する。このように部品の種類を２種類に限定することで組立を容易し、なおかつ品質にやや不安の残る１０ＭΩを超える抵抗器を用いずに済むという利点がある。また、かかる構成のノッチフィルタ１１５は、２−Ｔ型フィルタとしての正規の設計値から意図的にずらすことで中心周波数での減衰が劣る代わりに減衰帯域を広げることができるので、商用交流電源基本波の５０、６０Ｈｚから第二次高調波の１００、１２０Ｈｚまでの領域で、妨害波を１／３０００〜１／１００００に大きく減衰させることができる。加えて、素子の誤差の影響を受けにくくなるという効果もある。   The notch filter 115 is provided to remove or reduce the frequency component of the commercial AC power supply and its harmonic component in order to prevent such noise from being mixed. Specifically, as shown in FIG. 4A, the notch filter 115 is a single-stage notch filter NF including three resistance elements R1 to R3 and three capacitance elements C1 to C3, as shown in FIG. As shown in B), three are connected in cascade. There are only two types of elements constituting the single-stage notch filter NF of FIG. 4A, a 10 MΩ 1% product and a 100 pF 1% product, and these are connected in series or in parallel to constitute a filter. Thus, by limiting the number of parts to two types, there is an advantage that assembly is easy and there is no need to use a resistor exceeding 10 MΩ, which is somewhat uneasy about quality. Further, the notch filter 115 having such a configuration can broaden the attenuation band instead of inferior in attenuation at the center frequency by intentionally shifting from the normal design value as the 2-T filter, so that the commercial AC power supply basics The interference wave can be greatly attenuated to 1 / 3,000 to 1/10000 in the region from 50, 60 Hz of the wave to 100, 120 Hz of the second harmonic. In addition, there is an effect that it is difficult to be influenced by the error of the element.

位相差検出部１１８は、図２に示すように、一方の可変容量素子１１６Ａが接続される第１伝送路から交流信号を入力する抵抗分圧器１５と、抵抗分圧器１５の出力を増幅するアンプ１７Ａと、他方の可変容量素子１１６Ｂが接続される第２伝送路から交流信号を入力する抵抗分圧器兼位相差ゼロ点調整要素１６と、抵抗分圧器兼位相差ゼロ点調整要素１６の出力を増幅するアンプ１７Ｂと、アンプ１７Ａの出力とアンプ１７Ｂの出力の差動電圧を抽出する差動増幅器１７Ｃと、差動増幅器１７Ｃの出力を入力とするコンパレータ１８Ａと、アンプ１７Ｂの出力を入力とするコンパレータ１８Ｂと、ロジック回路１９と、抵抗Ｒ２１とコンデンサＣ２１とからなるＲＣ一次低域通過フィルタ２１を備える。   As shown in FIG. 2, the phase difference detection unit 118 includes a resistance voltage divider 15 that inputs an AC signal from the first transmission line to which one variable capacitance element 116 </ b> A is connected, and an amplifier that amplifies the output of the resistance voltage divider 15. 17A and the output of the resistance voltage divider / phase difference zero point adjustment element 16 for inputting an AC signal from the second transmission line to which the other variable capacitance element 116B is connected, and the resistance voltage divider / phase difference zero point adjustment element 16 The amplifier 17B to be amplified, the differential amplifier 17C that extracts the differential voltage between the output of the amplifier 17A and the output of the amplifier 17B, the comparator 18A that receives the output of the differential amplifier 17C, and the output of the amplifier 17B are input. An RC primary low-pass filter 21 including a comparator 18B, a logic circuit 19, and a resistor R21 and a capacitor C21 is provided.

図５は、図２の抵抗分圧器の一例を示す回路図を示す。図６は、図２の抵抗分圧器兼位相差ゼロ点調整要素の一例を示す回路図を示す。
抵抗分圧器１５は、第１伝送路に直列に接続された抵抗Ｒ１５と、第１伝送路と接地電位線との間に接続された抵抗Ｒ１６とにより、第１伝送路に出力される交流信号を所定比で分圧する。
抵抗分圧器兼位相差ゼロ点調整要素１６は、第２伝送路に直列に接続された抵抗Ｒ１７と第２伝送路と接地電位線との間に接続された抵抗Ｒ１８とにより、第２伝送路に出力される交流信号を所定比で分圧する。また、抵抗分圧器兼位相差ゼロ点調整要素１６は、第２伝送路と接地電位線との間に直列に接続されたコンデンサＣ１７と可変抵抗Ｒ１９により、第２伝送路に出力される交流信号に遅延を与えて位相を調整する。 FIG. 5 is a circuit diagram showing an example of the resistance voltage divider shown in FIG. FIG. 6 is a circuit diagram showing an example of the resistive voltage divider / phase difference zero point adjusting element of FIG.
The resistive voltage divider 15 includes an AC signal output to the first transmission line by a resistor R15 connected in series to the first transmission line and a resistor R16 connected between the first transmission line and the ground potential line. Is divided at a predetermined ratio.
The resistor voltage divider / phase difference zero adjustment element 16 includes a resistor R17 connected in series to the second transmission line and a resistor R18 connected between the second transmission line and the ground potential line. The AC signal output to is divided by a predetermined ratio. In addition, the resistor voltage divider / phase difference zero adjustment element 16 includes an AC signal output to the second transmission line by the capacitor C17 and the variable resistor R19 connected in series between the second transmission line and the ground potential line. To adjust the phase.

ここで、抵抗分圧器兼位相差ゼロ点調整要素１６による位相の調整方法の一例を説明する。
第２伝送路に出力される交流信号の位相は、第１検出用電極１２１Ａおよび第２検出用電極１２１Ｂを短絡し誘導電荷がゼロのときに、第１伝送路に出力される交流信号と位相差φがゼロでないゼロ近傍の値になるように調整する。
調整の際には、先ず、第１検出用電極１２１Ａおよび第２検出用電極１２１Ｂを短絡し誘導電荷をゼロに保つ。さらに、位相差検出部１１８の２つの入力に同一振幅同一位相の交流電圧を加える。図２の蓄電器１１９Ａ，１１９Ｂの両方の出力端子を短絡すればよい。ここで、アンプ１７Ａ，１７Ｂの一方の電圧利得を調整し差動増幅器１７Ｃの出力Ｖｄｉｆがゼロになるように調整する。利得を調整する代わりに抵抗分圧器１５あるいは抵抗分圧器兼位相差ゼロ点調整要素１６を構成する抵抗Ｒ１６、Ｒ１８の値を調整しても良い。 Here, an example of a phase adjustment method by the resistor voltage divider / phase difference zero point adjustment element 16 will be described.
The phase of the AC signal output to the second transmission path is the same as that of the AC signal output to the first transmission path when the first detection electrode 121A and the second detection electrode 121B are short-circuited and the induced charge is zero. Adjustment is made so that the phase difference φ is a non-zero value near zero.
In the adjustment, first, the first detection electrode 121A and the second detection electrode 121B are short-circuited to keep the induced charge at zero. Further, AC voltages having the same amplitude and the same phase are applied to the two inputs of the phase difference detection unit 118. It is only necessary to short-circuit both output terminals of the capacitors 119A and 119B in FIG. Here, the voltage gain of one of the amplifiers 17A and 17B is adjusted so that the output Vdif of the differential amplifier 17C becomes zero. Instead of adjusting the gain, the values of the resistors R16 and R18 constituting the resistor divider 15 or the resistor divider / phase difference zero adjustment element 16 may be adjusted.

次に、蓄電器１１９Ａ，１１９Ｂの出力端子の短絡を開放し、抵抗分圧器兼位相差ゼロ点調整要素１６の可変抵抗Ｒ１９の値を調整し、差動増幅器１７Ｃの出力Ｖｄｉｆの振幅（ピーク間電圧）が例えば１０ｍＶｐ−ｐなどの小さい電圧になるように調整する。出力Ｖｄｉｆの振幅を小さく設定するほど位相差φが小さくなり振幅差Δに対する感度が向上するがゼロは望ましくない。位相差φおよび振幅差Δとは、第１伝送路に出力される交流信号と、第２伝送路に出力される交流信号との位相差と振幅差を示す。
このような調整方法により、目的の位相差φの調整が達成される。なお、位相の調整は、典型的には、例えば工場出荷前の調整段階で行われるが、検電前の回路リセット時、或いは較正処理の際に行うようにしてもよい。また、位相の調整は、調整装置によって自動的に行われるようにしてもよいし、人手により行ってもよい。 Next, the short circuit of the output terminals of the capacitors 119A and 119B is opened, the value of the variable resistor R19 of the resistor voltage divider / phase difference zero adjustment element 16 is adjusted, and the amplitude (peak-to-peak voltage) of the output Vdif of the differential amplifier 17C ) Is adjusted to a small voltage such as 10 mVp-p. The smaller the amplitude of the output Vdif is, the smaller the phase difference φ is and the sensitivity to the amplitude difference Δ is improved, but zero is not desirable. The phase difference φ and the amplitude difference Δ indicate the phase difference and the amplitude difference between the AC signal output to the first transmission path and the AC signal output to the second transmission path.
With such an adjustment method, the target phase difference φ can be adjusted. The phase adjustment is typically performed, for example, at an adjustment stage before factory shipment, but may be performed at the time of circuit reset before power detection or at the time of calibration processing. The phase adjustment may be performed automatically by an adjustment device or may be performed manually.

なお、位相の調整は、次のような点に注意して行うのがよい。可変容量素子１１６Ａが電界効果トランジスタの場合はドレイン・ソース間電圧およびゲート・ソース間電圧の増大とともにドレイン・ソース間容量およびゲート・ソース容量が減少する。このため帯電していない状態よりも帯電している状態において可変容量素子１１６Ａの静電容量が減少するため検出側の交流信号Ｖｄ１（図２を参照）が補正側の交流信号Ｖｄ２（図２を参照）に比較して進み位相に変化する。この場合は、補正側の位相差調整器（図６の可変抵抗Ｒ１９）で交流信号Ｖｄ２を検出側の交流信号Ｖｄ１に対して遅れ位相側に、たとえば差動増幅器１７Ｃの電圧利得が１００倍の場合にはその出力である差電圧Ｖｄｉｆをたとえば１０ｍＶに設定する。可変容量素子１１６Ａが帯電増加で静電容量が増加するなら補正側の交流信号Ｖｄ２を検出側の交流信号Ｖｄ１に対して進み位相側とし、差動増幅器１７Ｃの出力（差電圧Ｖｄｉｆ）の大きさを設定する。補正側の交流信号Ｖｄ２を遅れ位相に設定すると交流信号Ｖｄ１、Ｖｄ２の振幅は｜Ｖｄ１｜＞｜Ｖｄ２｜となるが、差電圧Ｖｄｉｆ＝１０ｍＶ程度では｜Ｖｄ２｜＝｜Ｖｄ１｜に再調整する必要はない。なお検出側の交流信号Ｖｄ１に対して補正側の交流信号Ｖｄ２側を進み位相にするには検出側の抵抗分圧器１５を構成する抵抗Ｒ１６と並列にコンデンサＣ１７の半分程度の静電容量を追加するなどしておく。   It should be noted that the phase adjustment should be performed while paying attention to the following points. When the variable capacitance element 116A is a field effect transistor, the drain-source capacitance and the gate-source capacitance decrease as the drain-source voltage and the gate-source voltage increase. For this reason, since the capacitance of the variable capacitance element 116A decreases in the charged state rather than in the uncharged state, the detection-side AC signal Vd1 (see FIG. 2) becomes the correction-side AC signal Vd2 (see FIG. 2). Compared to the reference), the lead phase changes. In this case, the AC signal Vd2 is delayed from the detection-side AC signal Vd1 by the correction-side phase difference adjuster (variable resistor R19 in FIG. 6), for example, the voltage gain of the differential amplifier 17C is 100 times. In this case, the output differential voltage Vdif is set to 10 mV, for example. If the capacitance of the variable capacitance element 116A increases due to an increase in charging, the correction-side AC signal Vd2 is set to a phase advance with respect to the detection-side AC signal Vd1, and the output (differential voltage Vdif) of the differential amplifier 17C is increased. Set. When the AC signal Vd2 on the correction side is set to a delayed phase, the amplitudes of the AC signals Vd1 and Vd2 are | Vd1 |> | Vd2 |. There is no. In order to advance the phase of the AC signal Vd2 on the correction side relative to the AC signal Vd1 on the detection side, an electrostatic capacity approximately half that of the capacitor C17 is added in parallel with the resistor R16 constituting the resistance voltage divider 15 on the detection side. Do it.

ここでは、分圧比と増幅率との好適な設定例を示す。例えば、発振器１１７の出力電圧が０．３Ｖｐ−ｐ、および抵抗分圧器１５と抵抗分圧器兼位相差ゼロ点調整要素１６との分圧比が共に１１：１である場合、アンプ１１７Ａおよび１１７Ｂの電圧利得は３３倍、差動増幅器１７Ｃの差動電圧利得は１００倍に設定するとよい。差動電圧利得を１００倍としているのは、アンプ１７Ａ，１７Ｂの出力である交流信号Ｖｄ１、Ｖｄ２の差電圧Ｖｄｉｆが非常に小さくなるため、その振幅を大きくしてコンパレータ１８Ａを正常動作させるためである。ただし、上記の設定値は、誘導電荷を検出するセンサー素子として動作する可変容量素子１１６Ａ、１１６Ｂに所定の電界効果トランジスタを用いた場合に適した設定であり、選定するセンサー素子によって適宜変更する必要がある。   Here, a preferred setting example of the voltage division ratio and the amplification factor is shown. For example, when the output voltage of the oscillator 117 is 0.3 Vp-p and the voltage dividing ratio between the resistor voltage divider 15 and the resistor voltage divider / phase difference zero adjustment element 16 is 11: 1, the voltages of the amplifiers 117A and 117B The gain is preferably set to 33 times, and the differential voltage gain of the differential amplifier 17C is preferably set to 100 times. The reason why the differential voltage gain is set to 100 times is that the differential voltage Vdif between the AC signals Vd1 and Vd2, which are the outputs of the amplifiers 17A and 17B, becomes very small, so that the amplitude is increased and the comparator 18A operates normally. is there. However, the above set value is a setting suitable when a predetermined field effect transistor is used for the variable capacitance elements 116A and 116B that operate as sensor elements for detecting the induced charge, and needs to be changed as appropriate depending on the sensor element to be selected. There is.

図７および図８は、位相差検出部１１８の動作を説明する波形図を示す。図７および図８の（Ａ）〜（Ｆ）には、図２に示した位相差検出部１１８の各ノードの出力電圧の波形を示している。図７および図８の（Ｇ）には、交流信号Ｖｄ１の波形図の一部分Ｘを拡大して示している。ここでは、コンパレータ１８Ａ、１８Ｂの極性は逆転している。
これらの波形図に示すように、一対のアンプ１７Ａ，１７Ｂは、一方の可変容量素子１１６Ａの容量の周期的な変化によって生成された交流信号と、他方の可変容量素子１１６Ｂの容量の周期的な変化によって生成された交流信号とを増幅して出力する（図７、図８の交流信号Ｖｄ１，Ｖｄ２）。差動増幅器１７Ｃはアンプ１７Ａ，１７Ｂから交流信号Ｖｄ１、Ｖｄ２を入力して差電圧Ｖｄｉｆを出力する。差電圧Ｖｄｉｆは、交流信号Ｖｄ１、Ｖｄ２の振幅と位相が完全に一致するとゼロになる。一方、交流信号Ｖｄ１、Ｖｄ２の振幅と位相の一方又は両方が異なれば差電圧Ｖｄｉｆは交流信号Ｖｄ１、Ｖｄ２と同じ周波数の交流信号となる。 7 and 8 are waveform diagrams for explaining the operation of the phase difference detection unit 118. FIG. FIGS. 7A and 7F show waveforms of output voltages at the respective nodes of the phase difference detection unit 118 shown in FIG. FIG. 7 and FIG. 8G show an enlarged portion X of the waveform diagram of the AC signal Vd1. Here, the polarities of the comparators 18A and 18B are reversed.
As shown in these waveform diagrams, the pair of amplifiers 17A and 17B includes an AC signal generated by a periodic change of the capacitance of one variable capacitance element 116A and a periodicity of the capacitance of the other variable capacitance element 116B. The AC signal generated by the change is amplified and output (AC signals Vd1 and Vd2 in FIGS. 7 and 8). The differential amplifier 17C receives the AC signals Vd1 and Vd2 from the amplifiers 17A and 17B and outputs a differential voltage Vdif. The difference voltage Vdif becomes zero when the amplitudes and phases of the AC signals Vd1 and Vd2 completely coincide. On the other hand, if one or both of the amplitude and phase of the AC signals Vd1, Vd2 are different, the differential voltage Vdif becomes an AC signal having the same frequency as the AC signals Vd1, Vd2.

交流信号Ｖｄ１、Ｖｄ２の位相差φが小さくほぼ固定である場合、交流信号Ｖｄ１、Ｖｄ２の微小な振幅差Δの変化は、差電圧Ｖｄｉｆと交流信号Ｖｄ２との位相差θの大きな変化となって表れる。すなわち、差電圧Ｖｄｉｆは、交流信号Ｖｄ１、Ｖｄ２の微小な振幅差Δの変化が、位相の変化に拡大変換された信号となる。図７では振幅差Δが小さいときを表わし、図８では振幅差Δが大きいときを表わしている。振幅差Δの変化は、図７および図８のグラフ上で目視できない程度に非常に小さいが、図７の振幅差Δが小さいときには位相差θは９０度に近くなり、図８の振幅差Δが大きいときには位相差θは０度に近くなる。この動作原理については後で詳細に説明する。   When the phase difference φ between the AC signals Vd1 and Vd2 is small and substantially fixed, the change in the minute amplitude difference Δ between the AC signals Vd1 and Vd2 is a large change in the phase difference θ between the difference voltage Vdif and the AC signal Vd2. appear. That is, the difference voltage Vdif is a signal obtained by enlarging and converting a minute change in amplitude difference Δ between the AC signals Vd1 and Vd2 into a change in phase. FIG. 7 shows the case where the amplitude difference Δ is small, and FIG. 8 shows the case where the amplitude difference Δ is large. The change in the amplitude difference Δ is so small that it cannot be seen on the graphs of FIGS. 7 and 8. However, when the amplitude difference Δ in FIG. 7 is small, the phase difference θ is close to 90 degrees, and the amplitude difference Δ in FIG. When is large, the phase difference θ is close to 0 degrees. This operating principle will be described in detail later.

コンパレータ１８Ａは差動増幅器１７Ｃから出力された差電圧Ｖｄｉｆを基準電圧と比較して二値レベルの信号Ｖｃｄｉｆを出力する。コンパレータ１８Ｂは、交流信号Ｖｄ２を基準電圧と比較して二値レベルの信号Ｖｃ２を出力する。ロジック回路１９は、これらの信号Ｖｃｄｉｆ，Ｖｃ２を入力とし、図２の真理値表に示す論理演算を行って、その結果を示す２値レベルの信号Ｖｘｌｇを出力する。   The comparator 18A compares the difference voltage Vdif output from the differential amplifier 17C with a reference voltage and outputs a binary level signal Vcdif. The comparator 18B compares the AC signal Vd2 with the reference voltage and outputs a binary level signal Vc2. The logic circuit 19 receives these signals Vcdif and Vc2, performs a logical operation shown in the truth table of FIG. 2, and outputs a binary level signal Vxlg indicating the result.

ここで得られた出力信号Ｖｘｌｇのデューティ比は、差電圧Ｖｄｉｆと交流信号Ｖｄ２との位相差θを反映した値となる。図７および図８の例では、出力信号Ｖｘｌｇのデューティ比は位相差θに反比例した値となるが、出力信号Ｖｘｌｇの極性を変えることでデューティ比を位相差θに比例した値とすることもできる。
図２の回路は、第１検出用電極１２１Ａの誘導電荷が位相差φよりも振幅差Δに反映されるような電荷量の範囲において使用される。さらに、素子定数の最適化により、位相差φはゼロより大きいが、非常に小さい値に設定される。このような使用範囲と初期設定により、第１検出用電極１２１Ａの誘導電荷の変化に応じて振幅差Δが僅かに変化すると、この微小な振幅差Δの変化が位相差θの大きな変化に拡大変換されて、出力信号Ｖｘｌｇのデューティ比を連続的に変える。出力信号Ｖｘｌｇは、ＲＣ一次低域通過フィルタ２１によりデューティ比に比例した直流電圧Ｖｏｕｔに変換されて、判定部１３０に出力される。 The duty ratio of the output signal Vxlg obtained here is a value reflecting the phase difference θ between the differential voltage Vdif and the AC signal Vd2. In the example of FIGS. 7 and 8, the duty ratio of the output signal Vxlg is a value inversely proportional to the phase difference θ, but the duty ratio can be set to a value proportional to the phase difference θ by changing the polarity of the output signal Vxlg. it can.
The circuit in FIG. 2 is used in a charge amount range in which the induced charge of the first detection electrode 121A is reflected in the amplitude difference Δ rather than the phase difference φ. Further, by optimizing the element constant, the phase difference φ is set to a very small value although it is larger than zero. With such a use range and initial setting, when the amplitude difference Δ slightly changes according to the change in the induced charge of the first detection electrode 121A, the minute change in the amplitude difference Δ expands to a large change in the phase difference θ. As a result of the conversion, the duty ratio of the output signal Vxlg is continuously changed. The output signal Vxlg is converted to a DC voltage Vout proportional to the duty ratio by the RC primary low-pass filter 21 and output to the determination unit 130.

なお、交流信号Ｖｄ１、Ｖｄ２の振幅差Δが、大きく変化する領域を使えるのであれば、差電圧Ｖｄｉｆの振幅を検出して、第１検出用電極１２１Ａの誘導電荷量を検出すればよい。これにより、充電部の充電状態（例えば電線路の停電／活線）を検出することができる。しかしながら、直流電圧を非接触で検電する場合、交流信号Ｖｄ１、Ｖｄ２の振幅差Δの変化が小さく、振幅差Δの変化を直接に検出しにくい状況になるため、上述のように振幅差Δの変化を位相差θの大きな変化に拡大変換している。   Note that if an area in which the amplitude difference Δ between the AC signals Vd1 and Vd2 varies greatly can be used, the amplitude of the differential voltage Vdif may be detected to detect the induced charge amount of the first detection electrode 121A. Thereby, the charge state (for example, the power failure / hot line of an electric line) of a charge part is detectable. However, when the DC voltage is detected without contact, since the change in the amplitude difference Δ between the AC signals Vd1 and Vd2 is small and it is difficult to directly detect the change in the amplitude difference Δ, the amplitude difference Δ as described above. Is enlarged and converted into a large change in the phase difference θ.

＜回路の動作原理の説明＞
ここで、図２の回路の動作のうち、交流信号Ｖｄ１、Ｖｄ２の位相差φが小さくほぼ固定に設定されている場合に、交流信号Ｖｄ１、Ｖｄ２の振幅差Δの微小な変化が、差電圧Ｖｄｉｆと交流信号Ｖｄ２との位相差θの大きな変化に拡大変換される動作原理について説明する。この説明は、静電容量変化型センサーを用いて物理量の変化を位相変化として出力する回路の性質についての説明でもあり、本原理は、ほぼ全ての静電容量変化型センサーを用いる回路に適用できるといってよい。以下では、可変容量素子１１６Ａ、１１６Ｂのことを静電容量変化型センサーと呼んでいる。 <Description of circuit operation principle>
Here, in the operation of the circuit of FIG. 2, when the phase difference φ between the AC signals Vd1 and Vd2 is set to be small and substantially fixed, a minute change in the amplitude difference Δ between the AC signals Vd1 and Vd2 An operation principle that is enlarged and converted into a large change in the phase difference θ between Vdif and the AC signal Vd2 will be described. This explanation is also about the nature of a circuit that outputs a change in physical quantity as a phase change using a capacitance change sensor, and this principle can be applied to circuits using almost all capacitance change sensors. It can be said. Hereinafter, the variable capacitance elements 116A and 116B are referred to as capacitance change type sensors.

本実施形態では電界Ｅにより静電容量変化型センサー（１１６Ａ、１１６Ｂ）が静電誘導され電荷ｑを蓄積する。この状態で、静電容量変化型センサー（１１６Ａ、１１６Ｂ）を能動的に一定周波数ｆ［Ｈｚ］で駆動することで交流信号Ｖｄ１、Ｖｄ２を得る。そして、交流信号Ｖｄ１、Ｖｄ２の振幅差Δの変化を、差動増幅器１７Ｃの出力である差電圧Ｖｄｉｆ（＝Ｖｄ１−Ｖｄ２）から拡大（増幅）された位相差θの変化として得る。
ここで、電界Ｅは検出対象の物理量である。また、静電容量変化型センサー（１１６Ａ、１１６Ｂ）は一定周波数ｆ［Ｈｚ］の交流電圧で能動的に駆動しているのでアクチュエータ、或いは静電式発電機と見なすこともできる。 In the present embodiment, the electrostatic capacitance change type sensors (116A, 116B) are electrostatically induced by the electric field E and accumulate the charge q. In this state, the AC signals Vd1 and Vd2 are obtained by actively driving the capacitance change sensors (116A and 116B) at a constant frequency f [Hz]. Then, a change in the amplitude difference Δ between the AC signals Vd1 and Vd2 is obtained as a change in the phase difference θ expanded (amplified) from the difference voltage Vdif (= Vd1−Vd2) that is the output of the differential amplifier 17C.
Here, the electric field E is a physical quantity to be detected. Further, since the capacitance change type sensors (116A, 116B) are actively driven with an AC voltage having a constant frequency f [Hz], they can be regarded as actuators or electrostatic generators.

差動増幅器１７Ｃで振幅差Δの変化が位相差θの変化に「拡大変換」されることは次のように説明される。
先ず、差動増幅器１７Ｃの出力である差電圧Ｖｄｉｆは、次式（１）、（１ｂ）のように表わすことができる。
The fact that the change in the amplitude difference Δ is “enlarged and converted” into the change in the phase difference θ by the differential amplifier 17C is explained as follows.
First, the differential voltage Vdif, which is the output of the differential amplifier 17C, can be expressed as the following equations (1) and (1b).

ここで、
ｆ：静電容量変化型センサーの駆動周波数［Ｈｚ］
π：円周率
ｔ：時間［ｓｅｃ］
Ｖｄ２：補正側の交流信号［電圧］であり、振幅｜Ｖｄ２｜＝１とする。
Ｖｄ１：検出側の交流信号［電圧］であり、振幅｜Ｖｄ１｜＝１＋Δとする。なお、Δは正の値にも負の値にもなるが、Δの増大といった場合には、絶対値｜Δ｜の増大を意味する。
Δ：検出側の交流信号Ｖｄ１の補正側の交流信号Ｖｄ２に対する振幅差
φ：検出側の交流信号Ｖｄ１の補正側の交流信号Ｖｄ２に対する位相差［ｒａｄ］
Ｖｄｉｆ：差動増幅器１７Ｃの出力（差電圧）
Ａ：差動増幅器１７Ｃの電圧利得
α＝２・π・ｆ・ｔ
β＝φ
”・”（中黒）：掛算記号
としている。
また、β＝φ＜＜πであり、ｃｏｓφ≒１、ｓｉｎφ≒φと近似できるものとする。 here,
f: Driving frequency [Hz] of the capacitance change type sensor
π: Pi ratio t: Time [sec]
Vd2: AC signal [voltage] on the correction side, and amplitude | Vd2 | = 1.
Vd1: AC signal [voltage] on the detection side, and amplitude | Vd1 | = 1 + Δ. Δ can be either a positive value or a negative value, but in the case of an increase in Δ, it means an increase in absolute value | Δ |.
Δ: Amplitude difference between detection-side AC signal Vd1 and correction-side AC signal Vd2 φ: Phase difference between detection-side AC signal Vd1 and correction-side AC signal Vd2 [rad]
Vdif: output (differential voltage) of the differential amplifier 17C
A: Voltage gain of the differential amplifier 17C α = 2 · π · f · t
β = φ
“・” (Nakakuro): Multiplication symbol.
It is assumed that β = φ << π and can be approximated as cos φ≈1 and sin φ≈φ.

式（１ｂ）の第２項”−sinφcos（2・π・f・t）”は、”ｓｉｎ（２・π・ｆ・ｔ）”に対してφ＞０かφ＜０で９０度遅れ位相か進み位相の正弦波となることを表す。
回路はφ＜＜π［ｒａｄ］の領域で動作するので、１＋Δ−ｃｏｓφ≒Δと見なしてもよく、これから差電圧Ｖｄｉｆは、次式（１ｃ）のように表わすことができる。 The second term “−sinφcos (2 · π · f · t)” in the equation (1b) is 90 ° delayed phase when φ> 0 or φ <0 with respect to “sin (2 · π · f · t)”. It represents a sine wave with a leading phase.
Since the circuit operates in the region of φ << π [rad], it may be considered that 1 + Δ−cos φ≈Δ, and the difference voltage Vdif can be expressed as the following equation (1c).

φ＝０つまり検出側の交流信号Ｖｄ１と補正側の交流信号Ｖｄ２の位相が完全に一致する場合において回路は機能しない。ただし、本実施形態のように静電容量変化型センサー（１１６Ａ、１１６Ｂ）を用いる回路においては静電容量変化型センサーと抵抗との組み合わせが信号に遅延を及ぼして各信号経路の位相特性に影響を及ぼす。このため、物理量Ｅのゼロ点で完全に交流信号Ｖｄ１、Ｖｄ２が同一位相であっても物理量Ｅの変化によって静電容量変化型センサー（１１６Ａ、１１６Ｂ）の静電容量に変化が起これば交流信号Ｖｄ１、Ｖｄ２の位相ずれは発生する。よって位相差φが完全にゼロになることは多く生じない。 When φ = 0, that is, when the detection-side AC signal Vd1 and the correction-side AC signal Vd2 are completely in phase, the circuit does not function. However, in the circuit using the capacitance change type sensors (116A, 116B) as in the present embodiment, the combination of the capacitance change type sensor and the resistor delays the signal and affects the phase characteristics of each signal path. Effect. Therefore, even if the AC signals Vd1 and Vd2 are completely in the same phase at the zero point of the physical quantity E, if the capacitance of the capacitance change type sensors (116A, 116B) changes due to the change of the physical quantity E, the AC signal A phase shift occurs between the signals Vd1 and Vd2. Therefore, the phase difference φ does not often become completely zero.

式（１ｂ）、（１ｃ）を見ると、検出側の交流信号Ｖｄ１と補正側の交流信号Ｖｄ２との振幅差Δがゼロであれば、差電圧Ｖｄｉｆは、”ｃｏｓ（２・π・ｆ・ｔ）”成分が優勢となることが分かる。よって、差電圧Ｖｄｉｆの位相は、交流信号Ｖｄ１、Ｖｄ２から９０度異なる位相に近づく。なお、φ＜＜π［ｒａｄ］の場合、ｓｉｎφ≒φと見なしてもよいため、振幅差Δがゼロであれば、差電圧Ｖｄｉｆは位相差φに比例した小さな振幅となる。
一方、振幅差Δの増大に従い、次式（３）のように、式（１ｂ）の各項の大小が決まる。
この場合、式（１ｂ）、（１ｃ）は、sin（2・π・f・t）成分が優勢となるので、差電圧Ｖｄｉｆの位相は、交流信号Ｖｄ１、Ｖｄ２の位相に近づく。 Looking at the equations (1b) and (1c), if the amplitude difference Δ between the detection-side AC signal Vd1 and the correction-side AC signal Vd2 is zero, the difference voltage Vdif is “cos (2 · π · f · t) It can be seen that the “component” is dominant. Therefore, the phase of the difference voltage Vdif approaches 90 degrees different from the AC signals Vd1 and Vd2. In the case of φ << π [rad], sin φ≈φ may be considered. Therefore, if the amplitude difference Δ is zero, the difference voltage Vdif has a small amplitude proportional to the phase difference φ.
On the other hand, as the amplitude difference Δ increases, the magnitude of each term in the equation (1b) is determined as in the following equation (3).
In this case, in the equations (1b) and (1c), since the sin (2 · π · f · t) component is dominant, the phase of the difference voltage Vdif approaches the phase of the AC signals Vd1 and Vd2.

つまり差動増幅器１７Ｃの出力（差電圧Ｖｄｉｆ）の位相は、振幅差Δ＝０のときから振幅差Δの増大にともない、＋９０度から０度の範囲、または０度から９０度の範囲で変化する。このような回路の動作原理によって、交流信号Ｖｄ１、Ｖｄ２の振幅差Δの微小な変化が、差電圧Ｖｄｉｆと交流信号Ｖｄ２との位相差θの大きな変化に拡大変換される。   That is, the phase of the output (difference voltage Vdif) of the differential amplifier 17C changes in the range of +90 degrees to 0 degrees or in the range of 0 degrees to 90 degrees as the amplitude difference Δ increases from when the amplitude difference Δ = 0. To do. By such an operation principle of the circuit, a minute change in the amplitude difference Δ between the AC signals Vd1 and Vd2 is enlarged and converted into a large change in the phase difference θ between the difference voltage Vdif and the AC signal Vd2.

続いて、この回路の特性について説明する。
先ず、上記の式（１ｂ）から、差電圧Ｖｄｉｆの位相差θと、検出側の交流信号Ｖｄ１の補正側の交流信号Ｖｄ２に対する振幅差Δとの関係式θ（Δ）を求める。
式（４−１）のように、同じ周波数ωのサイン波とコサイン波とが加算された信号Ｕは、同じ周波数ωの信号となり、その振幅｜Ｕ｜と位相δは、サイン波の係数Ｖとコサイン波の係数Ｗとから、次式（４−２）、（４−３）のように計算できる。
Next, the characteristics of this circuit will be described.
First, a relational expression θ (Δ) between the phase difference θ of the difference voltage Vdif and the amplitude difference Δ of the detection-side AC signal Vd1 with respect to the correction-side AC signal Vd2 is obtained from the above equation (1b).
As in Expression (4-1), a signal U obtained by adding a sine wave and a cosine wave having the same frequency ω becomes a signal having the same frequency ω, and its amplitude | U | and phase δ are coefficients V of the sine wave. And the coefficient C of the cosine wave can be calculated as in the following equations (4-2) and (4-3).

これを差電圧Ｖｄｉｆの式（１ｂ）に当てはめれば、差電圧Ｖｄｉｆの振幅|Ｖｄｉｆ|と位相δは、次式（５−１）、（５−２）のように求められる。
If this is applied to the equation (1b) of the difference voltage Vdif, the amplitude | Vdif | and the phase δ of the difference voltage Vdif are obtained as in the following equations (5-1) and (5-2).

ここで、位相δは、コサイン波を基準波形とする極座標形式で表わした値を示している。本実施の形態では、信号波形のゼロクロス点を検出するため、位相差θ（Δ）は位相δからπ／２だけずらしてサイン波を基準波形とした表現とし、さらに、位相差の変化方向も図７、８の波形図に合わせて逆向きに調整する。すると、位相差θ（Δ）は、次式（5-3）のように表わすことができ、差電圧Ｖｄｉｆは、次式（１ｄ）のように表わすことができる。
式（１ｄ）、式（５−３）では、振幅差Δ＝０のときに位相差θ（Δ）は９０度となり、振幅差Δの増大に伴い位相差θ（Δ）は０度となり、図７および図８の波形と合致する。 Here, the phase δ indicates a value expressed in a polar coordinate format using a cosine wave as a reference waveform. In the present embodiment, in order to detect the zero cross point of the signal waveform, the phase difference θ (Δ) is shifted from the phase δ by π / 2, and the sine wave is used as a reference waveform, and the change direction of the phase difference is also indicated. Adjustments are made in the reverse direction according to the waveform diagrams of FIGS. Then, the phase difference θ (Δ) can be expressed as the following equation (5-3), and the difference voltage Vdif can be expressed as the following equation (1d).
In the equations (1d) and (5-3), the phase difference θ (Δ) is 90 degrees when the amplitude difference Δ = 0, and the phase difference θ (Δ) is 0 degrees as the amplitude difference Δ increases. It matches the waveforms in FIGS.

式（１ｂ）からも分かるが式（５−３）から差動増幅器１７Ｃの電圧利得Ａは検出する位相差θに影響しないことがわかる。また、交流信号Ｖｄ１、Ｖｄ２の位相差φは一度設定したら固定したままなので定数となり、検出する位相差θは検出側電圧Ｖｄ１の振幅差Δだけの関数となる。
式（５−３）のアークタンジェントの括弧内は、式（１ｂ）のサイン波成分と初期位相差φに依存するコサイン波成分との比を表している。式（５−３）から分かるように、初期設定の位相差φは小さいほど振幅差Δの変化が位相差θに大きく影響することが分かる。 As can be seen from the equation (1b), it can be seen from the equation (5-3) that the voltage gain A of the differential amplifier 17C does not affect the detected phase difference θ. In addition, the phase difference φ between the AC signals Vd1 and Vd2 remains fixed once set, and thus becomes a constant, and the detected phase difference θ is a function of only the amplitude difference Δ of the detection-side voltage Vd1.
The parentheses in the arctangent of the equation (5-3) represent the ratio between the sine wave component of the equation (1b) and the cosine wave component depending on the initial phase difference φ. As can be seen from the equation (5-3), it can be seen that the smaller the initial phase difference φ is, the more the change in the amplitude difference Δ greatly affects the phase difference θ.

また、本回路は交流信号Ｖｄ１、Ｖｄ２の振幅差Δと位相差φとが非常に小さな値となる領域で機能させるので、式（５−１）から分かるように、差動増幅器の出力（差電圧Ｖｄｉｆ）の振幅｜Ｖｄｉｆ｜も非常に小さくなる。このため、次段のコンパレータ１８Ａが所望の動作をしない場合も考慮すれば差動増幅器１７Ｃの電圧利得Ａは十分に大きな値に設定すべきと想定される。差動増幅器１７Ｃの電圧利得Ａは交流信号Ｖｄ１、Ｖｄ２の振幅差Δおよび位相差φに合わせて適宜に設定するとよい。特に静電容量変化型センサー（１１６Ａ、１１６Ｂ）のゼロ点においては振幅差Δ＝０に設定するので差動増幅器１７Ｃの出力電圧の振幅｜Ｖｄｉｆ｜がさらに小さくなることを考慮しなければならない。   In addition, since this circuit functions in a region where the amplitude difference Δ and the phase difference φ between the AC signals Vd1 and Vd2 are very small values, as can be seen from the equation (5-1), the output (difference) of the differential amplifier. The amplitude | Vdif | of the voltage Vdif) is also very small. Therefore, it is assumed that the voltage gain A of the differential amplifier 17C should be set to a sufficiently large value in consideration of the case where the comparator 18A at the next stage does not perform a desired operation. The voltage gain A of the differential amplifier 17C may be appropriately set according to the amplitude difference Δ and the phase difference φ between the AC signals Vd1 and Vd2. In particular, since the amplitude difference Δ = 0 is set at the zero point of the capacitance change type sensors (116A, 116B), it must be considered that the amplitude | Vdif | of the output voltage of the differential amplifier 17C is further reduced.

続いて、回路シミュレーションの結果と、理論式に基づく数値計算結果との比較を示す。
図９と図１０では、理論式に基づく結果として、差電圧Ｖｄｉｆ（ｔ，Δ）＝０（ゼロクロス点）となる位相角θを、位相差φをパラメータとして振幅差Δの関数として描いている。回路シミュレーションの結果は、いわば実際に回路を組み立てての実験結果そのものに近く多くの場合ほぼ実験結果そのものになる。回路シミュレーションの振幅差Δはセンサーに加えた直流電圧であり、検出側の交流電圧Ｖｄ１の振幅差Δの変化を間接的に増大させる作用を持つ。一方、理論式の振幅差Δは検出側の交流電圧Ｖｄ１の振幅を直接増大させている点が異なる。
そこで、図９と図１０の数値計算結果には、式（１ｂ）、（１ｄ）から得られる式（５−３）のθ（Δ）の代わりに、実際の回路に近づけた、補正式（１ｅ）の計算結果を用いている。
補正の詳細は次のごとくである。回路シミュレーションでは、差動増幅器１７Ｃより前段の回路の影響と思われるが、位相差φが完全に固定とはならない。具体的には、交流信号Ｖｄ１の振幅差Δが０〜０．１の範囲で変化したときに、初期の位相差φ＝０．５ｄｅｇが０．４６ｄｅｇに減少（初期の８％減）していた。式（１ｅ）では、このような位相差φの変化が反映されるように、位相差φを“φ＋Δσ”と補正した。σは補正定数で、例えば−０．４ｄｅｇである。この補正によれば、振幅差Δが０〜０．１と変化すると、位相差φは”０．５＋０×０．４”〜”０．５−０．１×０．４”となり、回路シミュレーションの実際の動作に近づく。なお、θ（Δ）は、式（１ｅ）から前述の通り求めることができるので、詳細は省略する。 Next, a comparison between the result of circuit simulation and the result of numerical calculation based on a theoretical formula is shown.
9 and 10, as a result based on the theoretical formula, the phase angle θ at which the difference voltage Vdif (t, Δ) = 0 (zero cross point) is drawn as a function of the amplitude difference Δ using the phase difference φ as a parameter. . The result of the circuit simulation is close to the experimental result itself when the circuit is actually assembled, and in many cases almost the experimental result itself. The amplitude difference Δ in the circuit simulation is a DC voltage applied to the sensor, and has an effect of indirectly increasing the change in the amplitude difference Δ of the AC voltage Vd1 on the detection side. On the other hand, the amplitude difference Δ in the theoretical formula is different in that the amplitude of the AC voltage Vd1 on the detection side is directly increased.
Therefore, in the numerical calculation results of FIGS. 9 and 10, the correction formula (closed to the actual circuit) instead of θ (Δ) in the formula (5-3) obtained from the formulas (1b) and (1d) ( The calculation result of 1e) is used.
Details of the correction are as follows. In the circuit simulation, the phase difference φ is not completely fixed, although it is considered that it is an influence of the circuit before the differential amplifier 17C. Specifically, when the amplitude difference Δ of the AC signal Vd1 changes in the range of 0 to 0.1, the initial phase difference φ = 0.5 deg is reduced to 0.46 deg (down 8% in the initial period). It was. In the equation (1e), the phase difference φ is corrected to “φ + Δσ” so that such a change in the phase difference φ is reflected. σ is a correction constant, for example, −0.4 deg. According to this correction, when the amplitude difference Δ changes from 0 to 0.1, the phase difference φ becomes “0.5 + 0 × 0.4” to “0.5−0.1 × 0.4”, which is a circuit simulation. Approach the actual operation of the. Since θ (Δ) can be obtained from the equation (1e) as described above, the details are omitted.

先ず、図９のグラフの説明を行う。
横軸Δは回路シミュレーションにおいて電極１２Ａに加えた電圧である。
図９のＶｘｌｇＮ［φ０．１ｄｅｇ］とＶｘｌｇＮ［φ０．５ｄｅｇ］のプロット線は、回路シミューレータで位相差φ＝０．１度と０．５度の場合（パラメータ）をシミュレーションして得た結果であり、ロジック回路１９の出力Ｖｘｌｇの最大値（ロジックレベルＨ）を１に最小値（ロジックレベルＬ）を０に規格化して平均値をとった値である。この値は出力Ｖｘｌｇの最大値の７５％から約１００％の間で変化している。これは出力Ｖｘｌｇのデューティ比が７５％から約１００％に変化したことを表しているとともに位相差θ（Δ）が９０度から０度までの間で変化したことを表している。 First, the graph of FIG. 9 will be described.
The horizontal axis Δ is the voltage applied to the electrode 12A in the circuit simulation.
The plot lines of VxlgN [φ0.1 deg] and VxlgN [φ0.5 deg] in FIG. 9 are the results obtained by simulating the case where the phase difference φ = 0.1 degrees and 0.5 degrees (parameters) with a circuit simulator. In other words, the maximum value (logic level H) of the output Vxlg of the logic circuit 19 is normalized to 1 and the minimum value (logic level L) is normalized to 0 to obtain an average value. This value varies between 75% and about 100% of the maximum value of the output Vxlg. This indicates that the duty ratio of the output Vxlg has changed from 75% to about 100%, and that the phase difference θ (Δ) has changed between 90 degrees and 0 degrees.

θ（Δ）［φ０．１ｄｅｇ］とθ（Δ）［φ０．５ｄｅｇ］のプロット線は、それぞれ電圧利得Ａ＝１とし、φ＝０．１度と０．５度とした場合に、式（１ｅ）を用いて差電圧Ｖｄｉｆがゼロとなる点（ゼロクロス）の位相角をプロットしたシミュレーション結果であり、振幅差Δの変化に対する差動増幅器１７Ｃの出力電圧の位相角の変化（位相差θ（Δ）の変化）を表す。
ＶｄｉｆＮ［φ０．５ｄｅｇ］のプロット線は、初期（あるいは設定）位相差φ＝０．５度の場合で、振幅差Δに対する差電圧Ｖｄｉｆの変化Ｖｄｉｆ（Δ）をＶｘｌｇＮと同じく０〜１の範囲に規格化してプロットして示している。このプロット線は、比較のため振幅差Δをそのまま検出した場合の特性を表わしている。 The plot lines of θ (Δ) [φ0.1 deg] and θ (Δ) [φ0.5 deg] are obtained when the voltage gain A = 1 and φ = 0.1 degrees and 0.5 degrees, respectively. 1e) is a simulation result in which the phase angle at the point where the differential voltage Vdif becomes zero (zero cross) is plotted, and the change in the phase angle of the output voltage of the differential amplifier 17C with respect to the change in the amplitude difference Δ (phase difference θ ( Δ)).
The plot line of VdifN [φ0.5 deg] is the initial (or set) phase difference φ = 0.5 degrees, and the change Vdif (Δ) of the difference voltage Vdif with respect to the amplitude difference Δ is in the range of 0 to 1 like VxlgN. Is normalized and plotted. This plot line represents the characteristic when the amplitude difference Δ is detected as it is for comparison.

振幅差Δを直接検出する場合の特性線（ＶｄｉｆＮ［φ０．５ｄｅｇ］のプロット線）を見ると、振幅差Δ＝０．００３以下ではほとんど感度がなく、振幅差Δ＝０．００３〜０．０１の領域で低い感度（勾配が緩い）が生じることがわかる。
それに対して、本実施の形態の回路の特性線であるＶｘｌｇＮ［φ０．１又は０．５ｄｅｇ］のプロット線を見ると、振幅差Δ＝０．００３以下の領域でも感度を有し（勾配を有する）、振幅差Δ＝０．００３〜０．０１の領域では、ＶｄｉｆＮのプロット線よりも高い感度が得られることが分かる。
本実施の形態の非接触型検電器のように、直流電圧を非接触で検電する場合、振幅差Δが０．００３以下〜０．０１（３ｍＶ以下〜１０ｍＶに相当）のような小さな値となる広い領域で感度が求められる。図９から分かるように、本実施の形態の回路では、このような振幅差Δにおいて必要な感度が得られている。 Looking at the characteristic line (VdifN [φ0.5 deg] plot line) in the case of directly detecting the amplitude difference Δ, there is almost no sensitivity when the amplitude difference Δ = 0.003 or less, and the amplitude difference Δ = 0.003-0. It can be seen that low sensitivity (gradient slope) occurs in the 01 region.
On the other hand, when looking at the plot line of VxlgN [φ0.1 or 0.5 deg], which is the characteristic line of the circuit of the present embodiment, it has sensitivity even in a region where the amplitude difference Δ = 0.003 or less (the slope is It can be seen that in the region where the amplitude difference Δ = 0.003 to 0.01, higher sensitivity than the plot line of VdifN is obtained.
When the DC voltage is detected in a non-contact manner as in the non-contact type voltage detector of the present embodiment, the amplitude difference Δ is a small value such as 0.003 or less to 0.01 (corresponding to 3 mV or less to 10 mV). Sensitivity is required in a wide area. As can be seen from FIG. 9, in the circuit of the present embodiment, necessary sensitivity is obtained for such an amplitude difference Δ.

なお、図９では、ＶｘｌｇＮのプロット線とθ（Δ）のプロット線とが一致していないように見える。これは、ＶｘｌｇＮのスケールを示す左縦軸と、θ（Δ）のスケールを示す右縦軸とが対応していないことに起因する。図９では、回路シミュレーションのＶｘｌｇＮの感度と、ＶｄｉｆＮの感度との比較を行うため、ＶｘｌｇＮのプロット線を左縦軸のスケールで示していた。   In FIG. 9, it appears that the plot line of VxlgN and the plot line of θ (Δ) do not match. This is because the left vertical axis indicating the VxlgN scale does not correspond to the right vertical axis indicating the θ (Δ) scale. In FIG. 9, in order to compare the sensitivity of VxlgN and the sensitivity of VdifN in the circuit simulation, the plot line of VxlgN is shown on the scale of the left vertical axis.

一方、図１０では、ＶｘｌｇＮとθ（Δ）とを比較できるように、ＶｘｌｇＮを右横軸の位相差のスケールで示している。先に述べたように、ＶｘｌｇＮの平均値が７５％は位相差９０度に対応し、１００％は位相差０度に対応する。よって、ＶｘｌｇＮを位相角のスケールに変換すると、図１０に示されるように変換される。図１０のグラフから、位相差θ（Δ）の変化が、回路シミュレーションと理論式（１ｅ）を用いた数値計算シミュレーションの特性において一致していて、理論式と回路の特性が一致していることが分かる。
本発明者が作成した理論式である式（１ｅ）による数値計算は原理の確認程度になる。それは回路の全てを記述していないからであるが、図１０の比較結果を見る限り差動増幅器の動作を理論式に記述できている。 On the other hand, in FIG. 10, VxlgN is shown as a phase difference scale on the right horizontal axis so that VxlgN and θ (Δ) can be compared. As described above, an average value of VxlgN of 75% corresponds to a phase difference of 90 degrees, and 100% corresponds to a phase difference of 0 degrees. Therefore, when VxlgN is converted to a phase angle scale, it is converted as shown in FIG. From the graph of FIG. 10, the change in the phase difference θ (Δ) is consistent in the characteristics of the circuit simulation and the numerical calculation simulation using the theoretical formula (1e), and the theoretical formula and the characteristics of the circuit are consistent. I understand.
The numerical calculation by the formula (1e), which is a theoretical formula created by the present inventor, is about the confirmation of the principle. This is because not all of the circuit is described, but the operation of the differential amplifier can be described in a theoretical formula as far as the comparison result of FIG. 10 is seen.

続いて、図１に戻って判定部１３０と出力部１４０について説明する。
判定部１３０は、ＡＤ変換器１３１と、判別器１３２とを備える。ＡＤ変換器１３１は、位相差検出部１１８の出力Ｖｏｕｔを周期的にサンプリングして取り込む。判別器１３２は、位相差検出部１１８の出力信号Ｖｘｌｇのデューティ比に基づく直流出力電圧Ｖｏｕｔの大きさを判別し、静電アンテナ１２０の帯電の大きさを表わす判別信号（判定結果）を出力部１４０へ出力する。このような機能を有する判定部１３０は、ＡＤ変換回路を内蔵したシングルチップマイコンのような１個の半導体集積回路、あるいはＡＤ変換ＩＣや演算機能を有するマイクロプロセッサ、半導体メモリ（ＲＯＭやＲＡＭ）など複数の半導体集積回路によって構成することができる。本実施の形態においては出力電圧Ｖｏｕｔは直流となる。よって、処理能力が要求される比較的高価なＣＰＵ（中央演算処理装置）が不要となり、例えば低価格な８〜１２ｂｉｔＡＤ変換回路を内蔵したシングルチップマイコンを用いることが可能となる。よって、非接触型検電器１００の部品コストを大幅に削減できる。 Next, returning to FIG. 1, the determination unit 130 and the output unit 140 will be described.
The determination unit 130 includes an AD converter 131 and a discriminator 132. The AD converter 131 periodically samples and captures the output Vout of the phase difference detection unit 118. The discriminator 132 discriminates the magnitude of the DC output voltage Vout based on the duty ratio of the output signal Vxlg of the phase difference detection unit 118, and outputs a discrimination signal (determination result) representing the magnitude of charging of the electrostatic antenna 120. Output to 140. The determination unit 130 having such a function is a single semiconductor integrated circuit such as a single-chip microcomputer incorporating an AD conversion circuit, a microprocessor having an AD conversion IC or an arithmetic function, a semiconductor memory (ROM or RAM), or the like. It can be constituted by a plurality of semiconductor integrated circuits. In the present embodiment, the output voltage Vout is a direct current. Therefore, a relatively expensive CPU (central processing unit) requiring processing capability is not required, and it is possible to use, for example, a single-chip microcomputer incorporating a low-priced 8 to 12 bit AD conversion circuit. Therefore, the component cost of the non-contact type voltage detector 100 can be greatly reduced.

出力部１４０は、複数のＬＥＤ（発光ダイオード）ランプを用いて、判定部１３０の判定結果を複数レベルで表示する表示器１４１と、スピーカやブザーのような発音装置１４２と、表示器１４１と発音装置１４２とを駆動する駆動回路１４３等を備える。駆動回路１４３は、判別器１３２の判別信号により、静電アンテナ１２０の帯電の大きさに応じて表示レベルが変化するように表示器１４１を駆動する。また、駆動回路１４３は、判別信号が異常な値を示しているときに、アラーム音を出力するように発音装置１４２を駆動する。計測者は、表示器１４１の表示レベルを見て、充電部の充電状態を判断することができる。   The output unit 140 uses a plurality of LED (light emitting diode) lamps to display the determination result of the determination unit 130 at a plurality of levels, a sounding device 142 such as a speaker or a buzzer, and the display 141. A drive circuit 143 for driving the device 142 and the like are provided. The drive circuit 143 drives the display device 141 so that the display level changes according to the charge level of the electrostatic antenna 120 according to the determination signal of the determination device 132. Further, the drive circuit 143 drives the sounding device 142 so as to output an alarm sound when the determination signal indicates an abnormal value. The measurer can determine the state of charge of the charging unit by looking at the display level of the display 141.

＜静電アンテナ＞
図１１は、図１の静電アンテナの構造を示す斜視図（Ａ）と正面図（Ｂ）である。
静電アンテナ１２０は、２つの基板１２３Ａ，１２３Ｂと、第１検出用電極１２１Ａおよび第２検出用電極１２１Ｂと、検出回路１１０と電気的に接続されるコネクタ１２２とを備える。
基板１２３Ａ，１２３Ｂは、共に長方形状の板面を有し、長手方向に見てＶ字状になるように、互いの長辺部が近接し、もう一方の長辺部がある他端側にかけて互いに離間するよう、固定具１２４によって連結されている。
第１検出用電極１２１Ａと第２検出用電極１２１Ｂとは、例えば長方形状の電極であり、基板１２３Ａ，１２３Ｂ上にそれぞれ設けられている。 <Electrostatic antenna>
FIG. 11 is a perspective view (A) and a front view (B) showing the structure of the electrostatic antenna of FIG.
The electrostatic antenna 120 includes two substrates 123A and 123B, a first detection electrode 121A and a second detection electrode 121B, and a connector 122 that is electrically connected to the detection circuit 110.
Each of the substrates 123A and 123B has a rectangular plate surface, and its long sides are close to each other so as to be V-shaped when viewed in the longitudinal direction, and the other long side is on the other end side. The fixtures 124 are connected so as to be separated from each other.
The first detection electrode 121A and the second detection electrode 121B are, for example, rectangular electrodes and are provided on the substrates 123A and 123B, respectively.

コネクタ１２２は、基板１２３Ａ，１２３Ｂの何れかの箇所に固定され、検出回路１１０に電気的に接続される。コネクタ１２２は、検出回路１１０の入力部（図１のノッチフィルタ１１５の入力端子）を、導線１２２ａを介して第１検出用電極１２１Ａに導通させ、検出回路１１０の接地電位ＧＮＤ１を第２検出用電極１２１Ｂに導通させる。
ここで、検出対象の充電部として、陽極と陰極の一対の電線路３１Ａ，３１Ｂが互いに離間して配置されていたとする（図１１（Ｂ）を参照）。このような場合、静電アンテナ１２０によれば、２本の電線路３１Ａ，３１Ｂの間に挟まれるように第１検出用電極１２１Ａと第２検出用電極１２１Ｂとを配置することができる。電線路３１Ａ，３１Ｂに直流電圧が印加されている場合、２本の電線路３１Ａ，３１Ｂの間に比較的に大きな電界が生じるので、図１１（Ｂ）のような配置により、静電アンテナ１２０に大きな帯電を発生させることができる。よって、低圧の直流電圧が印加される電線路３１Ａ，３１Ｂであっても、この直流電圧を高い感度で検出することができる。 The connector 122 is fixed to one of the boards 123A and 123B and is electrically connected to the detection circuit 110. The connector 122 conducts the input part of the detection circuit 110 (the input terminal of the notch filter 115 in FIG. 1) to the first detection electrode 121A via the conductive wire 122a, and the ground potential GND1 of the detection circuit 110 is used for the second detection. Conduction is made to the electrode 121B.
Here, it is assumed that a pair of electric wire paths 31A and 31B of an anode and a cathode are arranged apart from each other as a charging unit to be detected (see FIG. 11B). In such a case, according to the electrostatic antenna 120, the first detection electrode 121A and the second detection electrode 121B can be disposed so as to be sandwiched between the two electric wire paths 31A and 31B. When a direct current voltage is applied to the electric lines 31A and 31B, a relatively large electric field is generated between the two electric lines 31A and 31B. Therefore, the electrostatic antenna 120 is arranged as shown in FIG. A large charge can be generated. Therefore, even in the electric lines 31A and 31B to which a low-voltage DC voltage is applied, this DC voltage can be detected with high sensitivity.

＜第１変形例の静電アンテナ＞
図１２は、第１変形例の静電アンテナを示す斜視図（Ａ）と正面図（Ｂ）である。
実施の形態の非接触型検電器１００は、第１変形例の静電アンテナ２２０を備えていてもよい。静電アンテナ２２０は、２つの基板２２３Ａ，２２３Ｂと、第１検出用電極２２１Ａおよび第２検出用電極２２１Ｂと、検出回路１１０と電気的な接続を行うためのコネクタ２２２とを備える。
基板２２３Ａ，２２３Ｂは、共に一部が屈曲した板形状をしており、一方の基板２２３Ａの平板状の部分と、他方の基板２２３Ｂの平板状の部分とが、互いに離間して対向配置されるように固定具２２４によって連結されている。 <Electrostatic Antenna of First Modification>
FIG. 12 is a perspective view (A) and a front view (B) showing the electrostatic antenna of the first modification.
The non-contact type voltage detector 100 according to the embodiment may include the electrostatic antenna 220 of the first modification. The electrostatic antenna 220 includes two substrates 223A and 223B, a first detection electrode 221A and a second detection electrode 221B, and a connector 222 for electrical connection with the detection circuit 110.
Each of the substrates 223A and 223B has a plate shape in which a part thereof is bent, and a flat plate portion of one substrate 223A and a flat plate portion of the other substrate 223B are disposed to be opposed to each other. As shown in FIG.

第１検出用電極２２１Ａと第２検出用電極２２１Ｂとは、共に長方形状の電極であり、基板２２３Ａ，２２３Ｂの互いに対向配置された平板状の部分にそれぞれ設けられている。
コネクタ２２２は、検出回路１１０の入力部（図１のノッチフィルタ１１５の入力端子）を、第１検出用電極２２１Ａに導通させ、検出回路１１０の接地電位ＧＮＤ１を第２検出用電極２２１Ｂに導通させる。 The first detection electrode 221A and the second detection electrode 221B are both rectangular electrodes, and are respectively provided on the flat plate-like portions of the substrates 223A and 223B that are arranged to face each other.
The connector 222 causes the input part of the detection circuit 110 (the input terminal of the notch filter 115 in FIG. 1) to be conducted to the first detection electrode 221A, and causes the ground potential GND1 of the detection circuit 110 to be conducted to the second detection electrode 221B. .

ここで、検出対象の充電部として、陽極と陰極の一対の電線路３１Ａ，３１Ｂが近接して配置されていたとする（図１２（Ｂ）を参照）。このような場合、静電アンテナ２２０によれば、第１検出用電極１２１Ａと第２検出用電極１２１Ｂとの間に２本の電線路３１Ａ，３１Ｂを挟み込むように静電アンテナ２２０を配置することができる。このとき、２本の電線路３１Ａ，３１Ｂは、第１検出用電極１２１Ａおよび第２検出用電極１２１Ｂの長手方向に沿って延びる配置にできる。このような配置によって、２本の電線路３１Ａ，３１Ｂの電界が第１検出用電極１２１Ａと第２検出用電極１２１Ｂとに作用して、第１検出用電極１２１Ａと第２検出用電極１２１Ｂとに検出可能な大きさの帯電を発生させることができる。これにより、低圧の直流電圧が印加される電線路３１Ａ，３１Ｂであっても、この直流電圧を高い感度で検出することが可能となる。   Here, it is assumed that a pair of electric wire paths 31A and 31B of an anode and a cathode are arranged close to each other as a charging unit to be detected (see FIG. 12B). In such a case, according to the electrostatic antenna 220, the electrostatic antenna 220 is disposed so that the two electric wire paths 31A and 31B are sandwiched between the first detection electrode 121A and the second detection electrode 121B. Can do. At this time, the two electric lines 31A and 31B can be arranged to extend along the longitudinal direction of the first detection electrode 121A and the second detection electrode 121B. With such an arrangement, the electric fields of the two electric lines 31A and 31B act on the first detection electrode 121A and the second detection electrode 121B, and the first detection electrode 121A and the second detection electrode 121B It is possible to generate a charge having a detectable size. Thereby, even if it is electric wire way 31A and 31B to which a low voltage direct-current voltage is applied, it becomes possible to detect this direct-current voltage with high sensitivity.

＜第２変形例の静電アンテナ＞
図１３は、第２変形例の静電アンテナの第１形態を示す正面図（Ａ）と第２形態を示す正面図（Ｂ）である。図１３（Ａ），（Ｂ）は、静電アンテナ３２０を長手方向に見たときの図を示している。
実施の形態の非接触型検電器１００は、第２変形例の静電アンテナ３２０を備えていてもよい。第２変形例の静電アンテナ３２０は、２つの基板３２３Ａ，３２３Ｂと、第１検出用電極３２１Ａおよび第２検出用電極３２１Ｂと、２つの基板３２３Ａ，３２３Ｂを角度変更可能に連結するヒンジ３２４と、検出回路１１０と電気的に接続されるコネクタ３２２とを備える。 <Electrostatic Antenna of Second Modification>
FIG. 13: is the front view (A) which shows the 1st form of the electrostatic antenna of a 2nd modification, and the front view (B) which shows a 2nd form. FIGS. 13A and 13B show views when the electrostatic antenna 320 is viewed in the longitudinal direction.
The non-contact type voltage detector 100 according to the embodiment may include the electrostatic antenna 320 of the second modification. The electrostatic antenna 320 of the second modified example includes two substrates 323A and 323B, a first detection electrode 321A and a second detection electrode 321B, and a hinge 324 that couples the two substrates 323A and 323B so that the angle can be changed. And a connector 322 electrically connected to the detection circuit 110.

２つの基板３２３Ａ，３２３Ｂは、共に一部が屈曲した一方に長い板形状をしており、互いの長辺部がヒンジ３２４を介して連結されている。ヒンジ３２４を回動することで、図１３（Ａ）に示す第１形態と、図１３（Ｂ）に示す第２形態とに変形可能である。
コネクタ３２２は、検出回路１１０の入力部（図１のノッチフィルタ１１５の入力端子）を、第１検出用電極３２１Ａに導通させ、検出回路１１０の接地電位ＧＮＤ１を第２検出用電極３２１Ｂに導通させる。
第１検出用電極３２１Ａと第２検出用電極３２１Ｂとは、基板３２３Ａ，３２３Ｂの屈曲した板面に沿って設けられ、共に第１平面形状部Ｗ１と第２平面形状部Ｗ２とを有する。
第１形態では、図１３（Ａ）に示すように、第１検出用電極３２１Ａの第１平面形状部Ｗ１と、第２検出用電極３２１Ｂの第１平面形状部Ｗ１とが、長手方向に見てＶ字状の配置となる。このような形態では、２本の電線路３１Ａ，３１Ｂが離間して配置されている場合に、これらの間に第１検出用電極３２１Ａと第２検出用電極３２１Ｂとを挿入するように配置できる。これにより、図１１（Ｂ）の場合と同様に、電線路３１Ａ，３１Ｂの直流電圧の充電状態を高感度に検出することができる。 The two substrates 323 </ b> A and 323 </ b> B each have a long plate shape with one part being bent, and the long sides of the two substrates are connected via a hinge 324. By rotating the hinge 324, the first form shown in FIG. 13 (A) and the second form shown in FIG. 13 (B) can be transformed.
The connector 322 makes the input part of the detection circuit 110 (the input terminal of the notch filter 115 in FIG. 1) conductive to the first detection electrode 321A, and makes the ground potential GND1 of the detection circuit 110 conductive to the second detection electrode 321B. .
The first detection electrode 321A and the second detection electrode 321B are provided along the bent plate surfaces of the substrates 323A and 323B, and both have a first planar shape portion W1 and a second planar shape portion W2.
In the first embodiment, as shown in FIG. 13A, the first planar shape portion W1 of the first detection electrode 321A and the first planar shape portion W1 of the second detection electrode 321B are seen in the longitudinal direction. This is a V-shaped arrangement. In such a form, when the two electric wire paths 31A and 31B are arranged apart from each other, the first detection electrode 321A and the second detection electrode 321B can be inserted therebetween. . Thereby, similarly to the case of FIG. 11 (B), the charging state of the DC voltage of the electric lines 31A and 31B can be detected with high sensitivity.

第２形態では、図１３（Ｂ）に示すように、第１検出用電極３２１Ａの第２平面形状部Ｗ２と第２検出用電極３２１Ｂの第２平面形状部Ｗ２とが互いに離間して対向した配置となる。このような形態では、２本の電線路３１Ａ，３１Ｂが近接して配置されている場合に、第１検出用電極３２１Ａの第２平面形状部Ｗ２と第２検出用電極３２１Ｂの第２平面形状部Ｗ２との間に電線路３１Ａ，３１Ｂを挟み込むように静電アンテナ３２０を配置することができる。これにより、図１２（Ｂ）の場合と同様に、電線路３１Ａ，３１Ｂの直流電圧の充電状態を高感度に検出することができる。   In the second embodiment, as shown in FIG. 13B, the second planar shape portion W2 of the first detection electrode 321A and the second planar shape portion W2 of the second detection electrode 321B are opposed to each other. Arrangement. In such a form, when the two electric lines 31A and 31B are arranged close to each other, the second planar shape portion W2 of the first detection electrode 321A and the second planar shape of the second detection electrode 321B are used. The electrostatic antenna 320 can be arranged so that the electric wire paths 31A and 31B are sandwiched between the part W2. Thereby, similarly to the case of FIG. 12B, the charging state of the DC voltage of the electric lines 31A and 31B can be detected with high sensitivity.

以上のように、本実施の形態の非接触型検電器１００によれば、静電アンテナ１２０に、検出回路１１０の入力部に導通される第１検出用電極１２１Ａと、検出回路１１０の接地電位ＧＮＤ１と導通される第２検出用電極１２１Ｂとが設けられている。よって、検出対象の充電部（電線路など）とその陰極側の配線又は電極に対して、第１検出用電極１２１Ａと第２検出用電極１２１Ｂとを適宜に配置することで、充電部に低圧の直流電圧が印加されていても、静電アンテナ１２０に比較的に大きな帯電を発生させることができる。よって、このような充電部を検出対象として、その充電状態を非接触に且つ高感度に検出することができる。   As described above, according to the non-contact type voltage detector 100 of the present embodiment, the first detection electrode 121A that is conducted to the input part of the detection circuit 110 and the ground potential of the detection circuit 110 are connected to the electrostatic antenna 120. A second detection electrode 121B that is electrically connected to GND1 is provided. Therefore, by appropriately arranging the first detection electrode 121A and the second detection electrode 121B with respect to the detection target charging unit (electric line or the like) and the wiring or electrode on the cathode side, a low voltage is applied to the charging unit. Even if a direct current voltage is applied, a relatively large charge can be generated in the electrostatic antenna 120. Therefore, such a charging unit can be detected, and the state of charge can be detected in a non-contact and highly sensitive manner.

また、本実施の形態の検出回路１１０によれば、可変容量素子１１６Ａの容量を周期的に変更して、静電アンテナ１２０と可変容量素子１１６Ａとが接続される第１伝送路に交流信号を生成するので、静電アンテナ１２０に生じた帯電状態が検出過程で薄まることを回避できる。よって、低圧の直流電圧が印加される充電部の充電状態を高感度に検出することができる。
さらに、直流阻止フィルタと可変容量素子１１６Ｂとが接続される第２伝送路の交流信号を基準に、第１伝送路の交流信号の位相差を判定に用いているので、回路の温度特性や手振れなどの外的要因による位相の変化をキャンセルして正確な判定を行うことができる。 Further, according to the detection circuit 110 of the present embodiment, the capacitance of the variable capacitance element 116A is periodically changed, and an AC signal is sent to the first transmission line to which the electrostatic antenna 120 and the variable capacitance element 116A are connected. Since it is generated, it is possible to prevent the charged state generated in the electrostatic antenna 120 from fading during the detection process. Therefore, the charging state of the charging unit to which the low-voltage DC voltage is applied can be detected with high sensitivity.
Further, since the phase difference of the AC signal of the first transmission path is used for determination with reference to the AC signal of the second transmission path to which the DC blocking filter and the variable capacitance element 116B are connected, the circuit temperature characteristics and camera shake It is possible to cancel the phase change due to external factors such as the above and make an accurate determination.

（第２実施の形態）
図１４は、第２実施の形態の非接触型検電器の検出回路を示す構成である。
第２実施の形態は、図１４に示すように、可変容量素子１１６Ａ，１１６Ｂの前段にそれぞれ別個にノッチフィルタ１１５Ａ，１１５Ｂを設けるとともに、ノッチフィルタ１１５Ａ，１１５Ｂと可変容量素子１１６Ａ，１１６Ｂとの間にそれぞれ高抵抗値を有する抵抗器Ｒ４ａ，Ｒ４ｂを接続したものである。また、交流結合手段としての蓄電器（コンデンサ）１１９Ｃを、可変容量素子１１６Ｂの前段ではなく、ノッチフィルタ１１５Ｂの前段に設けている。また、ノッチフィルタ１１５Ａ，１１５Ｂとして、図６に示すような３個接続のものを使用する。それ以外は、図１の検電器と同様である。蓄電器（コンデンサ）１１９Ｃの容量値としては、例えば２２００ｐＦのような値、また抵抗器Ｒ５の抵抗値としては、例えば３０ＭΩのような値を採用できる。 (Second Embodiment)
FIG. 14 shows a configuration of a detection circuit of the non-contact type voltage detector according to the second embodiment.
In the second embodiment, as shown in FIG. 14, notch filters 115A and 115B are separately provided in front of variable capacitance elements 116A and 116B, respectively, and between notch filters 115A and 115B and variable capacitance elements 116A and 116B. Are connected to resistors R4a and R4b each having a high resistance value. In addition, a capacitor (capacitor) 119C as an AC coupling means is provided not in the preceding stage of the variable capacitance element 116B but in the preceding stage of the notch filter 115B. Further, as the notch filters 115A and 115B, those having three connections as shown in FIG. 6 are used. Other than that, it is the same as the voltage detector of FIG. As the capacitance value of the capacitor (capacitor) 119C, a value such as 2200 pF, for example, and as the resistance value of the resistor R5, a value such as 30 MΩ can be employed.

図１４に示す検電器は、可変容量素子１１６Ａおよび１１６Ｂの入力とノッチフィルタ１１５Ａ，１１５Ｂとの間に、１０ＭΩの抵抗器Ｒ４ａ，Ｒ４ｂを入れることで、ノッチフィルタ１１５Ａ，１１５Ｂの３段目のフィルタの出力と可変容量素子１１６Ａ，１１６Ｂ、さらには可変容量素子１１６Ａ，１１６Ｂ相互間を交流的に分離している。そして、この抵抗器Ｒ４ａ，Ｒ４ｂと可変容量素子１１６Ａ，１１６ＢとでＲＣ一次低域通過濾波器を構成することで、変電所での６相および１２相整流の脈流周波数や中波放送波の妨害を抑制する効果も持たせることができる。
また、２個のノッチフィルタ１１５Ａ，１１５Ｂを用いることで可変容量素子１１６Ａおよび１１６Ｂがノッチフィルタに及ぼす影響を軽減できる利点がある。 The voltage detector shown in FIG. 14 includes third-stage filters of notch filters 115A and 115B by inserting 10 MΩ resistors R4a and R4b between the inputs of the variable capacitors 116A and 116B and the notch filters 115A and 115B. And the capacitive elements 116A and 116B, and the variable capacitive elements 116A and 116B are separated in an alternating manner. The resistors R4a and R4b and the variable capacitors 116A and 116B constitute an RC primary low-pass filter, so that the pulsating frequency of 6-phase and 12-phase rectification at the substation and the medium wave broadcast wave It is also possible to have an effect of suppressing interference.
Further, the use of the two notch filters 115A and 115B has an advantage that the influence of the variable capacitance elements 116A and 116B on the notch filter can be reduced.

さらに、手持ち揺らぎの影響を検出する可変容量素子１１６Ｂ側で直流に対する感度があると検電器としての感度が落ちるので、可変容量素子１１６Ｂ側の直流感度をなくさなければならない。図１４の例であれば蓄電器１１９Ｃと可変容量素子１１６Ｂとの接続点と回路内部の基準電位点との間に抵抗器Ｒ５が設けられ、直流感度をほぼゼロにする直流阻止フィルタ（ＲＣ一次高域通過フィルタ）が構成されている。   Furthermore, if the sensitivity to direct current is present on the variable capacitance element 116B side that detects the influence of hand-held fluctuation, the sensitivity as a voltage detector decreases, so the direct current sensitivity on the variable capacitance element 116B side must be eliminated. In the example of FIG. 14, a resistor R5 is provided between a connection point between the capacitor 119C and the variable capacitance element 116B and a reference potential point inside the circuit, and a DC blocking filter (RC primary high-frequency) that makes DC sensitivity almost zero. A band-pass filter is configured.

以上、本発明の各実施の形態について説明したが、本発明は上記の実施の形態に限られるものではない。例えば、上記実施の形態では、検出対象の充電部として陽極と陰極の一対の電線路を示し、このような電線路に適した静電アンテナの形態を示した。しかしながら、検出対象の充電部の形態が異なれば、それに適応させて静電アンテナの形態も様々に変更可能である。例えば、第１検出用電極と第２検出用電極とを互いに異なる大きさとしたり、非対称に配置してもよい。
また、上記実施の形態では、判定部が静電アンテナの帯電の大きさを表わす判別信号を出力する構成としているが、判定部は位相差検出部の検出結果を閾値と比較することで、充電部が直流電圧で充電されているか否かを判定する構成としてもよい。 As mentioned above, although each embodiment of this invention was described, this invention is not limited to said embodiment. For example, in the said embodiment, a pair of electric wire of an anode and a cathode was shown as a charging part of a detection target, and the form of the electrostatic antenna suitable for such an electric wire was shown. However, if the form of the charging unit to be detected is different, the form of the electrostatic antenna can be variously changed in accordance with it. For example, the first detection electrode and the second detection electrode may have different sizes or may be asymmetrically arranged.
In the above-described embodiment, the determination unit outputs a determination signal indicating the magnitude of the electrostatic antenna charge. However, the determination unit compares the detection result of the phase difference detection unit with a threshold value, thereby charging. It is good also as a structure which determines whether a part is charged with the DC voltage.

１５ 抵抗分圧器
１６ 抵抗分圧器兼位相差ゼロ点調整要素
１７Ａ，１７Ｂ アンプ
１７Ｃ 差動増幅器
１８Ａ，１８Ｂ コンパレータ
１９ ロジック回路
２１ ＲＣ一次低域通過フィルタ
１００ 非接触型検電器
１１０ 検出回路
１１６Ａ，１１６Ｂ 可変容量素子
１１７ 発振器
１１８ 位相差検出部
１１９Ｃ 蓄電器（直流阻止フィルタ）
Ｒ５ 抵抗器（直流阻止フィルタ）
１２０、２２０、３２０ 静電アンテナ
１２１Ａ，２２１Ａ，３２１Ａ 第１検出用電極
１２１Ｂ，２２１Ｂ，３２１Ｂ 第２検出用電極
１２２，２２２，３２２ コネクタ
１２３Ａ，１２３Ｂ，２２３Ａ，２２３Ｂ，３２３Ａ，３２３Ｂ 基板
１３０ 判定部
１４０ 出力部
３２４ ヒンジ DESCRIPTION OF SYMBOLS 15 Resistance voltage divider 16 Resistance voltage divider and phase difference zero point adjustment element 17A, 17B Amplifier 17C Differential amplifier 18A, 18B Comparator 19 Logic circuit 21 RC primary low-pass filter 100 Non-contact type voltage detector 110 Detection circuit 116A, 116B Variable Capacitance element 117 Oscillator 118 Phase difference detection unit 119C Capacitor (DC blocking filter)
R5 resistor (DC blocking filter)
120, 220, 320 Electrostatic antennas 121A, 221A, 321A First detection electrode 121B, 221B, 321B Second detection electrode 122, 222, 322 Connector 123A, 123B, 223A, 223B, 323A, 323B Substrate 130 Determination unit 140 Output unit 324 Hinge

Claims (6)

直流電圧によって充電される充電部の充電状態を非接触で検出する非接触型検電器であって、
静電誘導により帯電する静電アンテナと、
前記静電アンテナの帯電状態を検出する検出回路と、
前記検出回路の検出結果に基づいて前記静電アンテナが近づけられた前記充電部の充電状態を判定する判定部と、
を備え、
前記静電アンテナは、前記検出回路の入力部に導通される第１検出用電極と、前記検出回路の接地電位と導通される第２検出用電極とを有することを特徴とする非接触型検電器。 A non-contact type voltage detector for detecting a charging state of a charging unit charged by a DC voltage in a non-contact manner,
An electrostatic antenna charged by electrostatic induction;
A detection circuit for detecting a charged state of the electrostatic antenna;
A determination unit that determines a charging state of the charging unit to which the electrostatic antenna is brought close based on a detection result of the detection circuit;
With
The electrostatic antenna includes a first detection electrode that is electrically connected to an input portion of the detection circuit, and a second detection electrode that is electrically connected to a ground potential of the detection circuit. Electricity. 前記第１検出用電極および前記第２検出用電極は共に一方に長い平面形状部を含み、
前記第１検出用電極の前記平面形状部と前記第２検出用電極の前記平面形状部とは、長手方向に見てＶ字形状となるように、互いの長辺部が近接し、互いのもう一方の長辺部が離間していることを特徴とする請求項１記載の非接触型検電器。 The first detection electrode and the second detection electrode both include a long planar shape portion on one side,
The long side portions of the first detection electrode and the flat shape portion of the second detection electrode are close to each other so that they are V-shaped when viewed in the longitudinal direction. The non-contact type voltage detector according to claim 1, wherein the other long side portion is separated. 前記第１検出用電極および前記第２検出用電極は共に一方に長い平面形状部を含み、
前記第１検出用電極の前記平面形状部と前記第２検出用電極の前記平面形状部とは、前記充電部を双方の間へ挿入可能なように離間していることを特徴とする請求項１記載の非接触型検電器。 The first detection electrode and the second detection electrode both include a long planar shape portion on one side,
The planar shape portion of the first detection electrode and the planar shape portion of the second detection electrode are separated so that the charging portion can be inserted between both. The non-contact type voltage detector according to 1. 前記第１検出用電極および前記第２検出用電極は共に一方に長い１つ又は複数の平面形状部を含み、
前記静電アンテナは、
前記第１検出用電極の何れかの前記平面形状部と前記第２検出用電極の何れかの前記平面形状部とが、長手方向に見てＶ字形状となるように、互いの長辺部が近接し、互いのもう一方の長辺部が離間した第１形態と、
前記第１検出用電極の何れかの前記平面形状部と前記第２検出用電極の何れかの前記平面形状部とが、前記充電部を双方の間へ挿入可能なように離間した第２形態と、
に変形可能に構成されていることを特徴とする請求項１記載の非接触型検電器。 The first detection electrode and the second detection electrode both include one or more planar shape portions that are long on one side,
The electrostatic antenna is
The long side portions of the first detection electrode and the second detection electrode so that the planar shape portion of the first detection electrode and the planar shape portion of the second detection electrode are V-shaped when viewed in the longitudinal direction. Are close to each other, and the other long side of each other is separated,
A second configuration in which the planar shape portion of any of the first detection electrodes and the planar shape portion of any of the second detection electrodes are separated so that the charging portion can be inserted between the two. When,
The non-contact type voltage detector according to claim 1, wherein the non-contact type voltage detector is configured to be deformable. 前記検出回路は、
前記入力部から電荷を導く第１伝送路と、
前記第１伝送路に接続された第１可変容量手段と、
前記第１可変容量手段の容量を周期的に変化させる発振器と、
を有し、
前記第１可変容量手段の容量の変化により前記静電アンテナの帯電状態に応じて振幅が変化する交流信号を前記第１伝送路に生成し、
前記判定部は、前記交流信号の振幅変化が位相変化に拡大変換された信号の位相に基づいて前記充電部の充電状態を判定することを特徴とする請求項１〜請求項４の何れか一項に記載の非接触型検電器。 The detection circuit includes:
A first transmission line for guiding charges from the input unit;
First variable capacitance means connected to the first transmission line;
An oscillator that periodically changes the capacitance of the first variable capacitance means;
Have
An AC signal whose amplitude changes according to the charging state of the electrostatic antenna due to a change in the capacity of the first variable capacitance means is generated in the first transmission line;
The said determination part determines the charge state of the said charging part based on the phase of the signal by which the amplitude change of the said alternating current signal was expanded-converted into the phase change. A non-contact type voltage detector according to item. 前記検出回路は、
前記第１伝送路と並列に接続される第２伝送路と、
前記第２伝送路に接続されて前記発振器により容量が周期的に変化する第２可変容量手段と、
前記第２伝送路上に接続されて直流成分の信号が伝わるのを阻止する直流阻止フィルタと、
前記第１伝送路に出力される交流信号と前記第２伝送路に出力される交流信号との差を出力する差動増幅器と、
を有し、
前記判定部は、前記差動増幅器の出力と前記第２伝送路に出力される交流信号との位相差に基づいて、前記静電アンテナが近づけられた前記充電部の充電状態を判定することを特徴とする請求項５記載の非接触型検電器。 The detection circuit includes:
A second transmission line connected in parallel with the first transmission line;
A second variable capacitance means connected to the second transmission line and having a capacitance periodically changed by the oscillator;
A DC blocking filter that is connected to the second transmission line and blocks transmission of a DC component signal;
A differential amplifier that outputs a difference between an AC signal output to the first transmission path and an AC signal output to the second transmission path;
Have
The determination unit determines a charging state of the charging unit to which the electrostatic antenna is brought close based on a phase difference between an output of the differential amplifier and an AC signal output to the second transmission line. The non-contact type voltage detector according to claim 5, wherein:
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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN113794260A (en) * 2021-09-28 2021-12-14 长春捷翼汽车零部件有限公司 Charging guide signal acquisition circuit, new energy vehicle-mounted charging seat and charging pile
KR20230086131A (en) * 2021-12-08 2023-06-15 송종환 Movable Non-contact DC High Voltage Detector

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN113794260A (en) * 2021-09-28 2021-12-14 长春捷翼汽车零部件有限公司 Charging guide signal acquisition circuit, new energy vehicle-mounted charging seat and charging pile
KR20230086131A (en) * 2021-12-08 2023-06-15 송종환 Movable Non-contact DC High Voltage Detector
KR102617040B1 (en) * 2021-12-08 2023-12-21 송종환 Movable Non-contact DC High Voltage Detector

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