JP2015144502A - Controller of permanent magnet synchronous motor - Google Patents

Controller of permanent magnet synchronous motor Download PDF

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JP2015144502A
JP2015144502A JP2014016395A JP2014016395A JP2015144502A JP 2015144502 A JP2015144502 A JP 2015144502A JP 2014016395 A JP2014016395 A JP 2014016395A JP 2014016395 A JP2014016395 A JP 2014016395A JP 2015144502 A JP2015144502 A JP 2015144502A
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野村　尚史
Naofumi Nomura
尚史 野村
暁峰 伍
Wu Xiaofeng
暁峰 伍
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Fuji Electric Co Ltd
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Abstract

PROBLEM TO BE SOLVED: To facilitate auto-tuning of the parameters of a flux model, where the magnetic saturation characteristics including interference between d and q axes of the controller for a permanent magnet synchronous motor are taken into account, and to allow for highly accurate torque control of the permanent magnet synchronous motor based on this flux model.SOLUTION: The controller for a permanent magnet synchronous motor includes means for controlling the d-axis current DC component to a command value, means for applying a sine wave AC voltage to the d-axis, Fourier coefficients calculation means for of the d-axis current, means for calculating the Fourier coefficients of the d-axis flux fundamental wave component from Fourier coefficients calculation means of the q-axis voltage and the angular frequency of the AC voltage, means for operating the Fourier coefficients of the d-axis DC component, and means for determining first through fourth parameters from the q-axis current fundamental wave component, the Fourier coefficients of the third harmonic component, the DC component of the d-axis current, the Fourier coefficients of the second harmonic component, the Fourier coefficients of the q-axis flux fundamental wave component, and the Fourier coefficients of the d-axis flux DC component. A flux model is constituted using these parameters.

Description

本発明は、永久磁石形同期電動機の制御装置に関し、詳しくは、永久磁石形同期電動機の電気定数を自動測定する、いわゆるオートチューニング技術に関するものである。   The present invention relates to a control device for a permanent magnet type synchronous motor, and more particularly to a so-called auto-tuning technique for automatically measuring the electrical constant of a permanent magnet type synchronous motor.

永久磁石形同期電動機のトルクを高精度に制御するためには、電動機鉄芯の磁気飽和特性を考慮した磁束モデルを求め、これに基づいて電流制御を行うことが望ましい。磁気飽和特性を考慮した磁束モデルの代表的なものとしては、非特許文献１に記載されたモデルが知られている。なお、磁気飽和特性とは、電流の増加に伴う電動機鉄芯の磁気飽和により、ｄ，ｑ軸磁束とこれらに対応する各軸電流との線形性が崩れる特性をいう。   In order to control the torque of the permanent magnet type synchronous motor with high accuracy, it is desirable to obtain a magnetic flux model in consideration of the magnetic saturation characteristics of the electric motor core and to control the current based on this. As a representative magnetic flux model considering magnetic saturation characteristics, a model described in Non-Patent Document 1 is known. The magnetic saturation characteristic refers to a characteristic in which the linearity between the d and q axis magnetic fluxes and the corresponding axial currents is lost due to the magnetic saturation of the electric motor core as the current increases.

ここで、図３は、非特許文献１に記載された永久磁石形同期電動機のモデルであり、ｄ，ｑ軸間の干渉を含む磁気飽和特性を考慮して構成されている。なお、ｄ，ｑ軸間の干渉とは、他軸電流の影響により自軸磁束が変化する特性をいう。
図３において、φｄ，φｑはｄ，ｑ軸磁束、ω１は角周波数、ｉｄ，ｉｑはｄ，ｑ軸電流、τは出力トルク、Ｒは巻線抵抗、φｍは永久磁石磁束、Ｐｎは極対数である。また、ｄ，ｑ軸磁束φｄ，φｑからｄ，ｑ軸電流ｉｄ，ｉｑを求める数式Ａ，Ｂは、磁気飽和特性を考慮した磁束モデル（後述する数式１６）を逆関数化したものであり、以下に示すとおりである。

Figure 2015144502
Figure 2015144502
 Here, FIG. 3 is a model of the permanent magnet type synchronous motor described in Non-Patent Document 1, and is configured in consideration of magnetic saturation characteristics including interference between the d and q axes. Incidentally, the interference between the d and q axes refers to the characteristic that the self-axis magnetic flux changes due to the influence of the other-axis current.
In FIG. 3, φ d and φ q are d and q axis magnetic fluxes, ω 1 is an angular frequency, i d and i q are d and q axis currents, τ is an output torque, R is a winding resistance, and φ m is a permanent magnet. Magnetic flux, Pn, is the number of pole pairs. Also, d, q-axis magnetic flux phi d, d from phi q, q-axis current i d, Equation A for obtaining the i q, B was the inverse function of the magnetic flux model considering magnetic saturation characteristics (formulas 16 to be described later) As shown below.
Figure 2015144502
Figure 2015144502

数式Ａ，Ｂでは、ｄ，ｑ軸電流ｉｄ，ｉｑとｄ，ｑ軸磁束φｄ，φｑとの関係を８つのパラメータＫＬｄ,ＫＳｄ，ＫＳｄｑ，ＫＬｑ，ＫＳｑ，ＫＳｑｄ，Ｉ０，φ０を用いて表しており、ｄ，ｑ軸間の干渉を含む磁気飽和特性を考慮した電動機モデルとなっている。 In the formulas A and B, the relationship between the d and q axis currents i d and i q and the d and q axis magnetic fluxes φ d and φ q is expressed by eight parameters K Ld , K Sd , K Sdq , K Lq , K Sq , K This is expressed using Sqd , I 0 , and φ 0 , and is an electric motor model that takes into account magnetic saturation characteristics including interference between the d and q axes.

ここで、ＫＬｄはｄ軸電流ｉｄに対するｄ軸磁束φｄの傾きの最大値に相当するパラメータ、ＫＬｑはｑ軸電流ｉｑに対するｑ軸磁束φｑの傾きの最大値に相当するパラメータ、ＫＳｄ，ＫＳｑは磁気飽和の度合いを示すパラメータ、ＫＳｄｑ，ＫＳｑｄはｄ，ｑ軸間の干渉の度合いを示すパラメータ、Ｉ０は等価磁化電流、φ０は磁束オフセットである。なお、Ｉ０，φ０は、ｑ軸電流ｉｑの大きさに関わらずｄ軸磁束φｄがほぼ一定値をとる時のｄ軸電流を−Ｉ０とし、これに対応するｄ軸磁束をφ０としている。 Here, K Ld is a parameter corresponding to the maximum value of the gradient of the d-axis magnetic flux φ d with respect to the d-axis current i d , and K Lq is a parameter corresponding to the maximum value of the gradient of the q-axis magnetic flux φ q with respect to the q-axis current i q . , K Sd and K Sq are parameters indicating the degree of magnetic saturation, K Sdq and K Sqd are parameters indicating the degree of interference between the d and q axes, I 0 is an equivalent magnetization current, and φ 0 is a magnetic flux offset. Incidentally, I 0, phi 0 is the d-axis magnetic flux of the d-axis current when taking almost constant value d-axis magnetic flux phi d regardless of the magnitude of the q-axis current i q and -I 0, corresponding to the φ 0 .

中津川潤之介，岩崎則久，名倉寛和，岩路善なお，「磁気飽和及びｄｑ軸間干渉を考慮した永久磁石同期モータの数式モデルの提案」，電気学会論文誌Ｄ，Ｖｏｌ．１３０，Ｎｏ．１１，ｐ．１２１２−ｐ．１２２０（２０１０年）Nakatsugawa Junnosuke, Iwasaki Norihisa, Nakura Hirokazu, Iwaji Yoshinao, “Proposal of Mathematical Model of Permanent Magnet Synchronous Motor Considering Magnetic Saturation and Interference between dq Axes”, IEEJ Transactions D, Vol. 130, no. 11, p. 1212-p. 1220 (2010)

非特許文献１に記載された磁束モデルを利用するためには、パラメータＫＬｄ,ＫＳｄ，ＫＳｄｑ，ＫＬｑ，ＫＳｑ，ＫＳｑｄ，Ｉ０，φ０の値を求める必要がある。非特許文献１では、ｄ，ｑ軸電流ｉｄ，ｉｑとｄ，ｑ軸磁束φｄ，φｑとの関係を測定し、これに基づいて各パラメータの値を求めている。しかし、ｄ，ｑ軸電流ｉｄ，ｉｑとｄ，ｑ軸磁束φｄ，φｑとの関係を測定するのは煩雑であり、また、非特許文献１では、これらの測定データから各パラメータを計算する方法が明確に開示されていない。更に、ｄ，ｑ軸電流ｉｄ，ｉｑとｄ，ｑ軸磁束φｄ，φｑとの関係を測定するためには、電動機に負荷をかける必要があり、負荷試験を行うための設備が必要になる。 In order to use the magnetic flux model described in Non-Patent Document 1, it is necessary to obtain values of parameters K Ld , K Sd , K Sdq , K Lq , K Sq , K Sqd , I 0 , and φ 0 . In Non-Patent Document 1, the relationship between d and q-axis currents i d and i q and d and q-axis magnetic fluxes φ d and φ q is measured, and the value of each parameter is obtained based on this. However, it is complicated to measure the relationship between the d and q axis currents i d and i q and the d and q axis magnetic fluxes φ d and φ q . In Non-Patent Document 1, each parameter is determined from these measurement data. The method of calculating is not clearly disclosed. Furthermore, in order to measure the relationship between the d and q axis currents i d and i q and the d and q axis magnetic fluxes φ d and φ q , it is necessary to apply a load to the motor, and there is no facility for performing a load test. I need it.

一方、電動機を駆動するインバータを使用して電動機の電気定数を自動測定する、いわゆるオートチューニング技術が開発されており、これにより、電動機の高性能な制御を容易に実現することが可能である。しかしながら、非特許文献１では、前述した各パラメータをオートチューニングする技術についても、特に開示されていない。   On the other hand, a so-called auto-tuning technique has been developed in which an electric constant of an electric motor is automatically measured using an inverter that drives the electric motor, whereby high-performance control of the electric motor can be easily realized. However, Non-Patent Document 1 does not particularly disclose a technique for auto-tuning each parameter described above.

そこで、本発明の解決課題は、ｄ，ｑ軸間の干渉を含む磁気飽和特性を考慮した磁束モデルのパラメータのオートチューニングを容易化し、この磁束モデルに基づいて、永久磁石形同期電動機の高精度なトルク制御を可能にした制御装置を提供することにある。   Therefore, the problem to be solved by the present invention is to facilitate the automatic tuning of the parameters of the magnetic flux model considering the magnetic saturation characteristics including the interference between the d and q axes, and based on this magnetic flux model, the high accuracy of the permanent magnet synchronous motor It is an object of the present invention to provide a control device that makes possible torque control.

上記課題を解決するため、請求項１に記載した発明は、電力変換器により永久磁石形同期電動機に供給する電流及び電圧を、前記電動機の回転子磁極方向に平行なｄ軸とこのｄ軸に直交するｑ軸とからなるｄ，ｑ直交回転座標上で制御するための制御装置であって、電動機鉄芯の磁気飽和特性を少なくとも考慮した磁束モデルに基づいて構成される制御装置において、
前記電動機のｄ軸電流の直流成分を指令値に制御する第１の手段と、
前記ｑ軸に正弦波の交番電圧を印加する第２の手段と、
前記ｑ軸電流の基本波成分のフーリエ係数を演算する第３の手段と、
前記ｑ軸電流の３倍高調波成分のフーリエ係数を演算する第４の手段と、
前記ｄ軸電流の直流成分のフーリエ係数を演算する第５の手段と、
前記ｄ軸電流の２倍高調波成分のフーリエ係数を演算する第６の手段と、
前記ｑ軸電圧の基本波成分のフーリエ係数を演算する第７の手段と、
前記ｑ軸電圧の基本波成分のフーリエ係数、及び、交番電圧の角周波数から、前記電動機のｑ軸磁束の基本波成分のフーリエ係数を演算する第８の手段と、
前記ｄ軸電流の直流成分のフーリエ係数から、前記ｄ軸磁束の直流成分のフーリエ係数を演算する第９の手段と、
前記ｑ軸電流の基本波成分のフーリエ係数、前記ｑ軸電流の３倍高調波成分のフーリエ係数、前記ｄ軸電流の直流成分のフーリエ係数、前記ｄ軸電流の２倍高調波成分のフーリエ係数、前記ｑ軸磁束の基本波成分のフーリエ係数、及び、前記ｄ軸磁束の直流成分のフーリエ係数から、前記ｑ軸電流に対するｑ軸磁束の傾きの最大値に相当する第１のパラメータ、ｑ軸上の磁気飽和の度合いを示す第２のパラメータ、及び、ｄ，ｑ軸間の干渉の度合いを示す第３，第４のパラメータを求める第１０の手段と、を備え、
前記第１〜第４のパラメータを用いて前記磁束モデルを構成するものである。
これにより、磁気飽和特性を考慮した磁束モデルのパラメータのオートチューニングが可能になる。 In order to solve the above-mentioned problems, the invention described in claim 1 is configured such that a current and a voltage supplied to a permanent magnet type synchronous motor by a power converter are applied to a d-axis parallel to the rotor magnetic pole direction of the motor and the d-axis. In a control device for controlling on d, q orthogonal rotation coordinates composed of orthogonal q axes, and configured based on a magnetic flux model at least considering the magnetic saturation characteristics of the electric motor core,
First means for controlling the DC component of the d-axis current of the motor to a command value;
A second means for applying a sinusoidal alternating voltage to the q-axis;
A third means for calculating a Fourier coefficient of the fundamental wave component of the q-axis current;
A fourth means for calculating a Fourier coefficient of a third harmonic component of the q-axis current;
A fifth means for calculating a Fourier coefficient of a DC component of the d-axis current;
A sixth means for calculating a Fourier coefficient of a second harmonic component of the d-axis current;
A seventh means for calculating a Fourier coefficient of the fundamental wave component of the q-axis voltage;
An eighth means for calculating a Fourier coefficient of a fundamental wave component of the q-axis magnetic flux of the motor from a Fourier coefficient of the fundamental wave component of the q-axis voltage and an angular frequency of the alternating voltage;
A ninth means for calculating a Fourier coefficient of a DC component of the d-axis magnetic flux from a Fourier coefficient of a DC component of the d-axis current;
Fourier coefficient of fundamental wave component of q-axis current, Fourier coefficient of third harmonic component of q-axis current, Fourier coefficient of DC component of d-axis current, Fourier coefficient of second harmonic component of d-axis current A first parameter corresponding to the maximum value of the gradient of the q-axis magnetic flux with respect to the q-axis current from the Fourier coefficient of the fundamental wave component of the q-axis magnetic flux and the Fourier coefficient of the DC component of the d-axis magnetic flux, A second parameter indicating the degree of magnetic saturation above, and tenth means for determining third and fourth parameters indicating the degree of interference between the d and q axes,
The magnetic flux model is configured using the first to fourth parameters.
Thereby, it is possible to automatically tune the parameters of the magnetic flux model in consideration of the magnetic saturation characteristics.

請求項２に記載した発明は、請求項１に記載した永久磁石形同期電動機の制御装置において、
前記第１０の手段は、
前記ｑ軸電流の基本波成分のフーリエ係数から、第２のｑ軸電流の基本波成分のフーリエ正弦係数を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｑ軸電流の３倍高調波成分のフーリエ係数とから、第２のｑ軸電流の３倍高調波成分のフーリエ正弦係数を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｄ軸電流の２倍高調波成分のフーリエ係数とから、第２のｄ軸電流の２倍高調波成分のフーリエ余弦係数を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｑ軸電圧の基本波成分のフーリエ係数とから、第２のｑ軸磁束の基本波成分のフーリエ正弦係数を演算する手段と、
前記第２のｑ軸電流の基本波成分のフーリエ正弦係数、前記第２のｑ軸電流の３倍高調波成分のフーリエ正弦係数、前記ｄ軸電流の直流成分のフーリエ係数、前記第２のｄ軸電流の２倍高調波成分のフーリエ余弦係数、前記第２のｑ軸磁束の基本波成分のフーリエ正弦係数、及び、前記ｄ軸磁束の直流成分のフーリエ係数から、前記第１〜第４のパラメータを演算する手段と、を備えたものである。
これにより、電機子抵抗による電圧降下や電流の制御誤差の影響を低減でき、磁束モデルのパラメータを正確に演算することができる。 According to a second aspect of the present invention, there is provided the permanent magnet synchronous motor control device according to the first aspect,
The tenth means includes
Means for calculating a Fourier sine coefficient of the fundamental wave component of the second q-axis current from a Fourier coefficient of the fundamental wave component of the q-axis current;
Means for calculating the Fourier sine coefficient of the third harmonic component of the second q-axis current from the Fourier coefficient of the fundamental component of the q-axis current and the Fourier coefficient of the third harmonic component of the q-axis current;
Means for calculating the Fourier cosine coefficient of the second harmonic component of the second d-axis current from the Fourier coefficient of the fundamental component of the q-axis current and the Fourier coefficient of the second harmonic component of the d-axis current;
Means for calculating a Fourier sine coefficient of the fundamental wave component of the second q-axis magnetic flux from a Fourier coefficient of the fundamental wave component of the q-axis current and a Fourier coefficient of the fundamental wave component of the q-axis voltage;
A Fourier sine coefficient of a fundamental wave component of the second q-axis current, a Fourier sine coefficient of a third harmonic component of the second q-axis current, a Fourier coefficient of a DC component of the d-axis current, the second d From the Fourier cosine coefficient of the second harmonic component of the axial current, the Fourier sine coefficient of the fundamental wave component of the second q-axis magnetic flux, and the Fourier coefficient of the DC component of the d-axis magnetic flux, the first to fourth Means for calculating a parameter.
Thereby, the influence of the voltage drop by the armature resistance and the current control error can be reduced, and the parameters of the magnetic flux model can be accurately calculated.

請求項３に記載した発明は、請求項１または２に記載した永久磁石形同期電動機の制御装置において、
前記第１の手段は、
前記ｄ軸電流の直流成分の指令値とｄ軸電流検出値との偏差からｄ軸電圧指令値を演算する手段を備え、
前記第２の手段は、
ｑ軸電流指令値を正弦波に制御する手段と、
前記ｑ軸電流指令値とｑ軸電流検出値との偏差からｑ軸電圧フィードバック制御値を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｑ軸電圧の基本波成分のフーリエ係数とからｑ軸リアクタンス及びｑ軸電機子抵抗を推定する手段と、
前記ｑ軸電流指令値、前記ｑ軸リアクタンスの推定値、及び、前記ｑ軸電機子抵抗の推定値からｑ軸電圧フィードフォワード補償値を演算する手段と、
前記ｑ軸電圧フィードバック制御値と前記ｑ軸電圧フィードフォワード補償値とからｑ軸電圧指令値を演算する手段と、を備えたものである。
これにより、ｄ軸電流の直流成分及びｑ軸の交番電流を各指令値にそれぞれ正確に制御することができる。 According to a third aspect of the present invention, there is provided the control device for the permanent magnet type synchronous motor according to the first or second aspect.
The first means includes
Means for calculating a d-axis voltage command value from a deviation between a command value of a DC component of the d-axis current and a detected d-axis current value;
The second means includes
means for controlling the q-axis current command value to a sine wave;
Means for calculating a q-axis voltage feedback control value from a deviation between the q-axis current command value and the q-axis current detection value;
Means for estimating the q-axis reactance and the q-axis armature resistance from the Fourier coefficient of the fundamental wave component of the q-axis current and the Fourier coefficient of the fundamental wave component of the q-axis voltage;
Means for calculating a q-axis voltage feedforward compensation value from the q-axis current command value, the estimated value of the q-axis reactance, and the estimated value of the q-axis armature resistance;
Means for calculating a q-axis voltage command value from the q-axis voltage feedback control value and the q-axis voltage feedforward compensation value.
As a result, the DC component of the d-axis current and the alternating current of the q-axis can be accurately controlled to each command value.

請求項４に記載した発明は、請求項１〜３の何れか１項に記載した永久磁石形同期電動機の制御装置において、
前記ｄ軸電流の２倍高調波成分のフーリエ係数と電機子抵抗とから、前記ｄ軸電流の２倍高調波成分に起因する前記電機子抵抗の電圧降下を演算する手段と、
前記ｑ軸電流の３倍高調波成分のフーリエ係数と電機子抵抗とから、前記ｑ軸電流の３倍高調波成分に起因する前記電機子抵抗の電圧降下を演算する手段と、
前記ｄ軸電流の２倍高調波成分に起因する前記電機子抵抗の電圧降下からｄ軸電圧指令値を演算する手段と、
前記ｑ軸電流の３倍高調波成分に起因する前記電機子抵抗の電圧降下からｑ軸電圧指令値を演算する手段と、を備えたものである。
これにより、電機子抵抗による電圧降下の影響を受けることなく、磁束モデルのパラメータを正確に演算することができる。 According to a fourth aspect of the present invention, in the control device for a permanent magnet type synchronous motor according to any one of the first to third aspects,
Means for calculating a voltage drop of the armature resistance caused by a second harmonic component of the d-axis current from a Fourier coefficient of the second harmonic component of the d-axis current and an armature resistance;
Means for calculating a voltage drop of the armature resistance caused by a third harmonic component of the q-axis current from a Fourier coefficient of the third harmonic component of the q-axis current and an armature resistance;
Means for calculating a d-axis voltage command value from a voltage drop of the armature resistance caused by a second harmonic component of the d-axis current;
Means for calculating a q-axis voltage command value from a voltage drop of the armature resistance caused by a third harmonic component of the q-axis current.
Thereby, the parameters of the magnetic flux model can be accurately calculated without being affected by the voltage drop due to the armature resistance.

本発明によれば、鉄芯の磁極飽和特性を考慮した磁束モデルのパラメータのオートチューニングにより、永久磁石形同期電動機のトルク制御を高精度に実現することができる。   According to the present invention, torque control of a permanent magnet synchronous motor can be realized with high accuracy by automatic tuning of parameters of a magnetic flux model in consideration of magnetic pole saturation characteristics of an iron core.

本発明の実施形態の構成を示すブロック図である。It is a block diagram which shows the structure of embodiment of this invention. 図１におけるアドミタンス推定部のブロック図である。It is a block diagram of the admittance estimation part in FIG. 非特許文献１に記載された電動機モデルを示す図である。It is a figure which shows the electric motor model described in the nonpatent literature 1.

以下、図に沿って本発明の実施形態を説明する。この実施形態に係る制御装置では、永久磁石形同期電動機の電流をｄ，ｑ軸成分に分解し、ｄ軸電流の直流成分を指令値に制御してｑ軸に正弦波の交番電圧を印加したときの電流及び電圧から磁束モデルのパラメータを演算する。そして、この磁束モデルに基づいて永久磁石形同期電動機における磁束と電流との間の非線形性を解析し、制御装置の設計等を行うものである。   Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the control device according to this embodiment, the current of the permanent magnet type synchronous motor is decomposed into d and q axis components, the DC component of the d axis current is controlled to a command value, and a sinusoidal alternating voltage is applied to the q axis. The magnetic flux model parameters are calculated from the current and voltage. Based on this magnetic flux model, the nonlinearity between the magnetic flux and the current in the permanent magnet type synchronous motor is analyzed, and the control device is designed.

まず、図１は、この実施形態に係る制御装置を主回路と共に示したブロック図であり、以下では、永久磁石形同期電動機（以下、単に電動機ともいう）の電圧及び電流の制御方法を制御装置の構成と共に説明する。なお、電圧及び電流の制御演算は、ｄ，ｑ軸直交回転座標上で行うこととし、電動機の回転子の磁極（Ｎ極）方向をｄ軸、ｄ軸から９０°進み方向をｑ軸と定義する。   First, FIG. 1 is a block diagram showing a control device according to this embodiment together with a main circuit. Hereinafter, a control method of voltage and current control methods for a permanent magnet type synchronous motor (hereinafter also simply referred to as an electric motor) will be described. This will be described together with the configuration. The voltage and current control calculations are performed on the d and q axis orthogonal rotation coordinates, and the magnetic pole (N pole) direction of the rotor of the motor is defined as the d axis, and the 90 ° advance direction from the d axis is defined as the q axis. To do.

図１において、積分器３０は、交番電圧の角周波数ωｈを積分してｑ軸交番電流指令値の角度θｈを演算する。
ｑ軸交番電流指令演算器３１は、ｑ軸交番電流指令値ｉｑｈ ＊を数式１により演算する。

Figure 2015144502
また、ｄ軸電流指令値Ｉｄ ＊はｄ軸電流の直流成分の指令値Ｉｄ（ａ０） ＊に制御し、ｑ軸電流指令値ｉｑ ＊はｑ軸交番電流指令値ｉｑｈ ＊に制御する。 In FIG. 1, the integrator 30 integrates the angular frequency ω h of the alternating voltage to calculate the angle θ h of the q-axis alternating current command value.
The q-axis alternating current command calculator 31 calculates the q-axis alternating current command value i qh * using Equation 1.
Figure 2015144502
 Further, the d-axis current command value I d * is controlled to the command value I d (a0) * of the DC component of the d-axis current, and the q-axis current command value i q * is controlled to the q-axis alternating current command value i qh * . To do.

座標変換器１４は、電気角θ１（ｕ相巻線を基準としたｄ軸の角度）を用いて、ｕ相電流検出器１１ｕ、ｗ相電流検出器１１ｗによりそれぞれ検出した相電流検出値ｉｕ，ｉｗをｄ，ｑ軸電流検出値ｉｄ，ｉｑに座標変換する。なお、上記電気角θ１は、例えば、特開２００１−１９００９３号公報や特許第３３１２４７２号公報に記載されている技術により、事前に求めた磁極位置演算値θ１０に制御する。 The coordinate converter 14 uses the electrical angle θ 1 (d-axis angle with respect to the u-phase winding) to detect the phase current detection value i detected by the u-phase current detector 11u and the w-phase current detector 11w, respectively. u, the i w d, q-axis current detection value i d, a coordinate conversion of i q. The electrical angle θ 1 is controlled to a magnetic pole position calculation value θ 10 obtained in advance by a technique described in, for example, Japanese Patent Application Laid-Open No. 2001-190093 and Japanese Patent No. 3321472.

ローパスフィルタ１８ａ，１８ｂは、ｄ，ｑ軸電流検出値ｉｄ，ｉｑの高周波成分を除去してｄ，ｑ軸電流検出値ｉｄｆ，ｉｑｆを演算する。
ｄ軸電流調節器２０ａは、減算器１９ａにより演算したｄ軸電流指令値Ｉｄ ＊とｄ軸電流検出値ｉｄｆとの偏差が零になるように動作してｄ軸電圧フィードバック制御値ｖｄＡＣＲを求める。また、ｑ軸電流調節器２０ｂは、減算器１９ｂにより演算したｑ軸電流指令値ｉｑ ＊とｑ軸電流検出値ｉｑｆとの偏差が零になるように動作してｑ軸電圧フィードバック制御値ｖｑＡＣＲを求める。 The low-pass filters 18a and 18b calculate d and q-axis current detection values i df and i qf by removing high-frequency components of the d and q-axis current detection values i d and i q .
The d-axis current adjuster 20a operates so that the deviation between the d-axis current command value I d * calculated by the subtracter 19a and the detected d-axis current value i df becomes zero, and the d-axis voltage feedback control value v dACR. Ask for. The q-axis current regulator 20b operates so that the deviation between the q-axis current command value i q * calculated by the subtracter 19b and the q-axis current detection value i qf becomes zero, and the q-axis voltage feedback control value is obtained. v Find qACR .

後述するように、磁束モデルのパラメータは、電動機鉄芯の磁気飽和に起因して流れる高調波電流を利用して推定する。このため、ｄ軸電流調節器２０ａ及びｑ軸電流調節器２０ｂが高調波電流に作用しないようにするため、各調節器２０ａ，２０ｂの応答周波数、及び、ローパスフィルタ１８ａ，１８ｂのカットオフ周波数は、交番電圧の角周波数ωｈの２倍よりも小さく設計する。 As will be described later, the parameters of the magnetic flux model are estimated using the harmonic current that flows due to the magnetic saturation of the motor core. Therefore, in order to prevent the d-axis current regulator 20a and the q-axis current regulator 20b from acting on the harmonic current, the response frequency of each of the regulators 20a and 20b and the cutoff frequency of the low-pass filters 18a and 18b are Designed to be smaller than twice the angular frequency ω h of the alternating voltage.

直流電機子抵抗補償器２１は、ｄ軸電機子抵抗フィードフォワード補償値ｖｄｒａを数式２により演算する。

Figure 2015144502
The DC armature resistance compensator 21 calculates the d-axis armature resistance feed-forward compensation value v dra according to Equation 2.
Figure 2015144502

電圧補償値演算器３２は、ｑ軸電圧フィードフォワード補償値ｖｑｈＦＦを数式３により演算する。

Figure 2015144502
The voltage compensation value calculator 32 calculates the q-axis voltage feedforward compensation value v qhFF by Equation 3.
Figure 2015144502

電機子抵抗補償器３３は、ｄ軸電流の２倍高調波成分のフーリエ余弦係数Ｉｄ（ａ２）、ｄ軸電流の２倍高調波成分のフーリエ正弦係数Ｉｄ（ｂ２）、ｑ軸電流の３倍高調波成分のフーリエ余弦係数Ｉｑ（ａ３）、ｑ軸電流の３倍高調波成分のフーリエ正弦係数Ｉｑ（ｂ３）から、数式４によりｄ軸高調波電機子抵抗補償値ｖｄｈｒａ、ｑ軸高調波電機子抵抗補償値ｖｑｈｒａを演算する。

Figure 2015144502
The armature resistance compensator 33 includes a Fourier cosine coefficient I d (a2) of the second harmonic component of the d-axis current, a Fourier sine coefficient I d (b2) of the second harmonic component of the d- axis current, and the q-axis current. From the Fourier cosine coefficient I q (a3) of the third harmonic component and the Fourier sine coefficient I q (b3) of the third harmonic component of the q-axis current, the d-axis harmonic armature resistance compensation value v dra , The q-axis harmonic armature resistance compensation value v qra is calculated.
Figure 2015144502

ｄ軸電圧指令値ｖｄ ＊は、加算器２２ａにより、ｄ軸電圧フィードバック制御値ｖｄＡＣＲ、ｄ軸電機子抵抗フィードフォワード補償値ｖｄｒａ、ｄ軸高調波電機子抵抗補償値ｖｄｈｒａを加算して求める。一方、ｑ軸電圧指令値ｖｑ ＊は、加算器２２ｂ，３４により、ｑ軸電圧フィードバック制御値ｖｑＡＣＲ、ｑ軸電圧フィードフォワード補償値ｖｑｈＦＦ、ｑ軸高調波電機子抵抗補償値ｖｑｈｒａを加算して求める。 The d-axis voltage command value v d * is added by the adder 22a to the d-axis voltage feedback control value v dACR , the d-axis armature resistance feedforward compensation value v dra , and the d-axis harmonic armature resistance compensation value v dhra. Ask. On the other hand, the q-axis voltage command value v q * is added to the q-axis voltage feedback control value v qACR , the q-axis voltage feedforward compensation value v qhFF , and the q-axis harmonic armature resistance compensation value v qra by the adders 22 b and 34. Add to find.

座標変換器１５は、ｄ軸電圧指令値ｖｄ ＊及びｑ軸電圧指令値ｖｑ ＊を電気角θ１に基づいて相電圧指令値ｖｕ ＊，ｖｖ ＊，ｖｗ ＊に座標変換する。
整流回路６０は、三相交流電源５０の三相交流電圧を整流して得た直流電圧をインバータ等の電力変換器７０に供給する。
ＰＷＭ回路１３は、相電圧指令値ｖｕ ＊，ｖｖ ＊，ｖｗ ＊から、電力変換器７０の出力電圧を前記相電圧指令値に制御するためのゲート信号を生成する。電力変換器７０は、ゲート信号に基づいて内部の半導体スイッチング素子を制御することにより、永久磁石形同期電動機８０の端子電圧を相電圧指令値ｖｕ ＊，ｖｖ ＊，ｖｗ ＊に制御する。
以上に述べた制御により、ｄ軸電流ｉｄを直流成分の指令値Ｉｄ（ａ０） ＊に制御し、また、ｑ軸電流ｉｑを交番電流の指令値ｉｑｈ ＊に制御することができる。 The coordinate converter 15 converts the d-axis voltage command value v d * and the q-axis voltage command value v q * into a phase voltage command value v u * , v v * , v w * based on the electrical angle θ 1. .
The rectifier circuit 60 supplies a DC voltage obtained by rectifying the three-phase AC voltage of the three-phase AC power supply 50 to a power converter 70 such as an inverter.
The PWM circuit 13 generates a gate signal for controlling the output voltage of the power converter 70 to the phase voltage command value from the phase voltage command values v u * , v v * , and v w * . The power converter 70 controls the internal semiconductor switching element based on the gate signal, thereby controlling the terminal voltage of the permanent magnet type synchronous motor 80 to the phase voltage command values v u * , v v * , v w * . .
The control mentioned above, it is possible to control the d-axis current i d to the command value I d of the DC component (a0) *, and controls the q-axis current i q to the command value of the alternating current i qh * .

次に、θｈ，ｖｑ ＊，ｉｄ，ｉｑに基づいて、ｑ軸リアクタンス推定値Ｘｑｈｅｓｔ及びｑ軸電機子抵抗推定値Ｒｑｈｅｓｔを求める方法について説明する。
図１のフーリエ係数演算器３５は、θｈ，ｖｑ ＊，ｉｄ，ｉｑから、ｑ軸電圧、ｑ軸電流及びｄ軸電流のフーリエ係数を数式５により演算する。

Figure 2015144502
なお、ｑ軸電圧の基本波成分のフーリエ係数Ｖｑ（ａ１），Ｖｑ（ｂ１）の演算は、ｑ軸電圧指令値ｖｑ ＊の代わりにｑ軸電圧検出値ｖｑを用いて行ってもよい。 Next, a method for obtaining the q-axis reactance estimated value X qhest and the q-axis armature resistance estimated value R qhest based on θ h , v q * , i d , i q will be described.
The Fourier coefficient calculator 35 in FIG. 1 calculates the Fourier coefficients of the q-axis voltage, the q-axis current, and the d-axis current from θ h , v q * , i d , i q using Equation 5.
Figure 2015144502
 The calculation of the Fourier coefficients V q (a1) and V q (b1) of the fundamental component of the q-axis voltage is performed using the q-axis voltage detection value v q instead of the q-axis voltage command value v q *. Also good.

アドミタンス推定部３６は、ｑ軸電圧の基本波成分のフーリエ余弦係数Ｖｑ（ａ１）、ｑ軸電圧の基本波成分のフーリエ正弦係数Ｖｑ（ｂ１）、ｑ軸電流の基本波成分のフーリエ余弦係数Ｉｑ（ａ１）、ｑ軸電流の基本波成分のフーリエ正弦係数Ｉｑ（ｂ１）から、ｑ軸コンダクタンス推定値Ｇｑｈｅｓｔ及びｑ軸サセプタンス推定値（−Ｂｑｈｅｓｔ）を演算する。 The admittance estimation unit 36 includes a Fourier cosine coefficient V q (a1) of the fundamental wave component of the q-axis voltage, a Fourier sine coefficient V q (b1) of the fundamental wave component of the q-axis voltage, and a Fourier cosine of the fundamental wave component of the q-axis current. From the coefficient I q (a1) and the Fourier sine coefficient I q (b1) of the fundamental wave component of the q-axis current, the q-axis conductance estimated value G qhest and the q-axis susceptance estimated value (−B qhest ) are calculated.

図２は、アドミタンス推定部３６の構成を示すブロック図である。
図２において、振幅演算器１０１は、ｑ軸電圧の基本波成分の振幅Ｖｑ（ｃ１）を数式６により演算する。

Figure 2015144502
FIG. 2 is a block diagram illustrating a configuration of the admittance estimation unit 36.
In FIG. 2, the amplitude calculator 101 calculates the amplitude V q (c1) of the fundamental wave component of the q-axis voltage using Equation 6.
Figure 2015144502

有効電流・無効電流演算器１０２は、ｑ軸の交番電圧と同位相のｑ軸有効電流Ｉｑｐｏｗｄｅｔと、ｑ軸の交番電圧と９０度の位相差を持つｑ軸無効電流Ｉｑｖａｒｄｅｔとを、数式７により演算する。

Figure 2015144502
The active current / reactive current calculator 102 calculates a q-axis active current I qpowdet having the same phase as the q-axis alternating voltage and a q-axis reactive current I qvardet having a phase difference of 90 degrees from the q-axis alternating voltage. 7 to calculate.
Figure 2015144502

乗算器１０３ａは、ｑ軸有効電流推定値Ｉｑｐｏｗｅｓｔを数式８により演算し、乗算器１０３ｂは、ｑ軸無効電流推定値ｉｑｖａｒｅｓｔを数式９により演算する。

Figure 2015144502
Figure 2015144502
 The multiplier 103a calculates the q-axis effective current estimated value I qpowest by Equation 8, and the multiplier 103b calculates the q-axis reactive current estimated value i qvarest by Equation 9.
Figure 2015144502
Figure 2015144502

減算器１０４ａは、ｑ軸有効電流推定値Ｉｑｐｏｗｅｓｔとｑ軸有効電流演算値Ｉｑｐｏｗｄｅｔとから、ｑ軸有効電流推定誤差Ｉｑｐｏｗｅｒｒを数式１０により演算する。

Figure 2015144502
The subtractor 104a calculates the q-axis effective current estimation error I qpowerr from the q-axis effective current estimated value I qpower and the q-axis effective current calculated value I qpowdet according to Equation 10.
Figure 2015144502

パラメータ推定器１０５ａは、ｑ軸有効電流推定誤差Ｉｑｐｏｗｅｒｒを積分制御してｑ軸コンダクタンス推定値Ｇｑｈｅｓｔを演算する。具体的には、数式１１の演算を行う。

Figure 2015144502
数式１１による演算の結果、ｑ軸有効電流推定誤差Ｉｑｐｏｗｅｒｒが零になるようにｑ軸コンダクタンス推定値Ｇｑｈｅｓｔが演算され、ｑ軸コンダクタンス推定値Ｇｑｈｅｓｔは真値に収束する。 The parameter estimator 105a calculates the q-axis conductance estimated value G qhest by integrating and controlling the q-axis effective current estimation error I qpowerr . Specifically, the calculation of Expression 11 is performed.
Figure 2015144502
 As a result of the calculation according to Expression 11, the q-axis conductance estimated value G qhest is calculated so that the q-axis effective current estimation error I qpowerr becomes zero, and the q-axis conductance estimated value G qhest converges to a true value.

減算器１０４ｂは、ｑ軸無効電流推定値Ｉｑｖａｒｅｓｔとｑ軸無効電流演算値Ｉｑｖａｒｄｅｔとから、ｑ軸無効電流推定誤差Ｉｑｖａｒｅｒｒを数式１２により演算する。

Figure 2015144502
The subtractor 104b calculates the q-axis reactive current estimation error I qvarerr from the q-axis reactive current estimated value I qvarest and the q-axis reactive current calculated value I qvardet according to Equation 12.
Figure 2015144502

パラメータ推定器１０５ｂは、ｑ軸無効電流推定誤差Ｉｑｖａｒｅｒｒを積分制御してｑ軸サセプタンス推定値（−Ｂｑｈｅｓｔ）を演算する。具体的には、数式１３の演算を行う。

Figure 2015144502
数式１３による演算の結果、ｑ軸無効電流推定誤差Ｉｑｖａｒｅｒｒが零になるようにｑ軸サセプタンス推定値（−Ｂｑｈｅｓｔ）が演算され、ｑ軸サセプタンス推定値（−Ｂｑｈｅｓｔ）は真値に収束する。 The parameter estimator 105b integrates and controls the q-axis reactive current estimation error I qvarerr to calculate the q-axis susceptance estimated value (−B qhest ). Specifically, the calculation of Expression 13 is performed.
Figure 2015144502
 As a result of the calculation according to Expression 13, the q-axis susceptance estimated value (−B qhest ) is calculated so that the q-axis reactive current estimation error I qvarerr becomes zero, and the q-axis susceptance estimated value (−B qhest ) converges to a true value. To do.

図１におけるインピーダンス演算器３７は、数式１４により、ｑ軸リアクタンス推定値Ｘｑｈｅｓｔ及びｑ軸電機子抵抗推定値Ｒｑｈｅｓｔを演算する。
これらのｑ軸リアクタンス推定値Ｘｑｈｅｓｔ及びｑ軸電機子抵抗推定値Ｒｑｈｅｓｔは、電圧補償値演算器３２にも入力される。

Figure 2015144502
The impedance calculator 37 in FIG. 1 calculates a q-axis reactance estimated value X qhest and a q-axis armature resistance estimated value R qhest using Equation 14.
These q-axis reactance estimated value X qhest and q-axis armature resistance estimated value R qhest are also input to the voltage compensation value calculator 32.
Figure 2015144502

次に、パラメータ推定部４０において、磁束モデルのパラメータを求める方法について説明する。
まず、前述した非特許文献１と同様に、電流と磁束との関係を関数化した数式１５の磁束モデルを構成する。

Figure 2015144502
Next, a method for obtaining the parameters of the magnetic flux model in the parameter estimation unit 40 will be described.
First, similarly to Non-Patent Document 1 described above, a magnetic flux model of Formula 15 in which the relationship between current and magnetic flux is functionalized is configured.
Figure 2015144502

パラメータ推定部４０は、前述したフーリエ係数とパラメータＫＬｄ，ＫＳｄ，Ｉ０，φ０とを用いて、数式１５に示した磁束モデルのパラメータのうち、ＫＬｑ，ＫＳｑ，ＫＳｑｄ，ＫＳｄｑを推定する。
なお、ｄ軸電流ｉｄに対するｄ軸磁束Ψｄの傾きの最大値に相当するパラメータＫＬｄ、ｄ軸磁束Ψｄにおけるｄ軸電流反比例係数に相当するパラメータＫＳｄ、等価磁化電流に相当するパラメータＩ０、及び、磁束オフセットに相当するパラメータφ０は、後述するように既知の値とする。 The parameter estimation unit 40 uses the above-described Fourier coefficients and the parameters K Ld , K Sd , I 0 , φ 0, and among the parameters of the magnetic flux model shown in Equation 15, K Lq , K Sq , K Sqd , K Estimate Sdq .
A parameter K Ld corresponding to the maximum value of the inclination of the d-axis magnetic flux Ψ d with respect to the d-axis current i d , a parameter K Sd corresponding to the d-axis current inverse proportionality coefficient in the d-axis magnetic flux Ψ d, and a parameter corresponding to the equivalent magnetization current The parameter φ 0 corresponding to I 0 and the magnetic flux offset is a known value as will be described later.

次に、パラメータの推定の原理及び方法について説明する。
説明を簡単にするため、最初に、電動機の電機子抵抗を零に近似でき、電流を指令値に正確に制御できる場合について説明する。
前述したように、ｑ軸交番電流指令値ｉｑｈ ＊を数式１のように制御する場合、ｄ，ｑ軸電圧ｖｄ，ｖｑ及びｄ，ｑ軸磁束Ψｄ，Ψｑは数式１６，数式１７のように表される。

Figure 2015144502
Figure 2015144502
 Next, the principle and method of parameter estimation will be described.
In order to simplify the explanation, first, a case where the armature resistance of the motor can be approximated to zero and the current can be accurately controlled to the command value will be described.
As described above, when the q-axis alternating current command value i qh * is controlled as shown in Equation 1, the d and q-axis voltages v d and v q and the d and q-axis magnetic fluxes Ψ d and Ψ q are expressed by Equation 16 and Equation It is expressed as 17.
Figure 2015144502
Figure 2015144502

数式１５によって表される磁束モデルにおいて、ｄ，ｑ軸磁束を数式１７のように制御した場合、ｄ軸電流ｉｄには直流成分と２倍高調波成分とが発生し、ｑ軸電流ｉｑには基本波成分と３倍高調波成分とが発生する。これを数式で表すと、数式１８となる。

Figure 2015144502
In the magnetic flux model represented by Equation 15, d, when controlled as the q-axis magnetic flux formulas 17, a DC component and the second harmonic component is generated in the d-axis current i d, q-axis current i q Generates a fundamental wave component and a triple harmonic component. This can be expressed as Equation 18.
Figure 2015144502

数式１７，数式１８を数式１５に代入して整理すると、数式１９を導出することができる。但し、数式１９における係数Ａ〜Ｅは、数式２０のとおりである。

Figure 2015144502
Figure 2015144502
 By substituting Equations 17 and 18 into Equation 15 and rearranging, Equation 19 can be derived. However, the coefficients A to E in Equation 19 are as in Equation 20.
Figure 2015144502
Figure 2015144502

ここで、数式２１の近似を行う。

Figure 2015144502
Here, approximation of Formula 21 is performed.
Figure 2015144502

数式１９に数式２１の近似を適用し、両辺の直流成分、ｃｏｓ２θｈ成分，ｓｉｎθｈ成分，ｓｉｎ３θｈ成分を比較することにより、数式２２を導出する。

Figure 2015144502
Applying the approximation of equation 21 to equation 19, the DC component of both sides, cos h component, sin [theta h component, by comparing the Sin3shita h component derives a formula 22.
Figure 2015144502

磁束モデルのパラメータ推定は、数式２２の関係を利用して実現する。
まず、数式２０，数式２２を整理し、パラメータＫＬｑ，ＫＳｑｄの関係式を数式２３のように導出する。但し、数式２３における係数ａ，ｂ，ｃは、数式２４に示すとおりである。

Figure 2015144502
Figure 2015144502
 The parameter estimation of the magnetic flux model is realized using the relationship of Equation 22.
First, Formula 20 and Formula 22 are arranged, and a relational expression of parameters K Lq and K Sqd is derived as Formula 23. However, the coefficients a, b, and c in Expression 23 are as shown in Expression 24.
Figure 2015144502
Figure 2015144502

数式２４において、フーリエ係数Ψｑ（ｂ１），Ｉｑ（ｂ１），Ｉｑ（ｂ３）、及び、パラメータＫＬｄ，ＫＳｄは既知の値である。
ｄ軸磁束の直流成分のフーリエ余弦係数Ψｄ（ａ０）は、この時点では未知であるパラメータＫＳｄｑを零に近似し、数式１５に基づいて数式２５により近似計算する。

Figure 2015144502
この結果、係数ａ，ｂ，ｃは、既知の値から計算することができる。 In Formula 24, Fourier coefficients Ψ q (b1) , I q (b1) , I q (b3) , and parameters K Ld and K Sd are known values.
The Fourier cosine coefficient Ψ d (a0) of the DC component of the d-axis magnetic flux is approximated by Equation 25 based on Equation 15 by approximating the parameter K Sdq that is unknown at this time to zero.
Figure 2015144502
 As a result, the coefficients a, b, and c can be calculated from known values.

数式２４，数式２５より、係数ａ，ｂ，ｃは、ｄ軸電流の直流成分のフーリエ余弦係数Ｉｄ（ａ０）の関数となる。このことから、ｄ軸電流の直流成分のフーリエ余弦係数Ｉｄ（ａ０）を異なる２種類の値に制御すると、数式２３から、数式２６に示す連立方程式を得ることができる。

Figure 2015144502
From Equations 24 and 25, the coefficients a, b, and c are functions of the Fourier cosine coefficient I d (a0) of the DC component of the d-axis current. From this, when the Fourier cosine coefficient I d (a0) of the DC component of the d-axis current is controlled to two different values, the simultaneous equations shown in Expression 26 can be obtained from Expression 23.
Figure 2015144502

数式２６の連立方程式をパラメータＫＬｑ，ＫＳｑｄについて解くと、パラメータＫＬｑ，ＫＳｑｄは、数式２７のように、既知の係数ａ（１），ｂ（１），ｃ（１），ａ（２），ｂ（２），ｃ（２）から演算することができる。

Figure 2015144502
Solving the simultaneous equations of Equation 26 parameter K Lq, the K SQD, parameter K Lq, K SQD, like the equation 27, a known coefficient a (1), b (1 ), c (1), a ( 2) , b (2) , c (2) can be calculated.
Figure 2015144502

以上のことから、パラメータＫＬｑ，ＫＳｑｄは、下記の手順により演算する。
（１）ｄ軸電流を、直流成分の指令値Ｉｄ（ａ０） ＊が異なる２種類の値に制御し、ｑ軸に交番電圧を印加する。そして、２つのケースについて、電圧及び電流のフーリエ係数を測定する。
（２）前記２つのケースについて、数式１７により、ｑ軸電圧の基本波成分のフーリエ余弦係数Ｖｑ（ａ１）と交番電圧の角周波数ωｈとから、ｑ軸磁束の基本波成分のフーリエ正弦係数Ψｑ（ｂ１）を演算する。
（３）前記２つのケースについて、数式２４，数式２５により、電流及び磁束のフーリエ係数から、係数ａ，ｂ，ｃを演算する。
（４）パラメータＫＬｑ，ＫＳｑｄを、数式２７により、係数ａ（１），ｂ（１），ｃ（１），ａ（２），ｂ（２），ｃ（２）から演算する。 From the above, the parameters K Lq and K Sqd are calculated according to the following procedure.
(1) The d-axis current is controlled to two types of values having different DC component command values I d (a0) * , and an alternating voltage is applied to the q-axis. And the Fourier coefficient of a voltage and an electric current is measured about two cases.
(2) For the above two cases, the Fourier sine coefficient of the fundamental wave component of the q-axis magnetic flux is calculated from the Fourier cosine coefficient Vq (a1) of the fundamental wave component of the q-axis voltage and the angular frequency ω h of the alternating voltage. Ψ q (b1) is calculated.
(3) In the above two cases, the coefficients a, b, and c are calculated from the Fourier coefficients of the current and the magnetic flux according to the expressions 24 and 25.
(4) The parameters K Lq and K Sqd are calculated from the coefficients a (1) , b (1) , c (1) , a (2) , b (2) , c (2) according to Equation 27.

数式２２より、係数Ｂ，Ｄは、数式２８の関係にある。

Figure 2015144502
数式２０より、係数Ａは既知の値から演算可能であるため、係数Ｂ，Ｄは数式２８により演算可能である。 From Equation 22, the coefficients B and D are in the relationship of Equation 28.
Figure 2015144502
 From Equation 20, since the coefficient A can be calculated from a known value, the coefficients B and D can be calculated by Equation 28.

また、数式２０をパラメータＫＳｑについて整理することで、数式２９を導出することができる。

Figure 2015144502
数式２９より、パラメータＫＳｑを既知の値から演算することができる。
更に、数式２０をパラメータＫＳｄｑについて整理することで、数式３０の導出が可能である。
Figure 2015144502
 よって、数式３０により、パラメータＫＳｄｑを既知の値から演算することができる。 Further, the numerical formula 29 can be derived by arranging the numerical formula 20 with respect to the parameter K Sq .
Figure 2015144502
 From Equation 29, the parameter K Sq can be calculated from a known value.
Furthermore, the numerical formula 30 can be derived by arranging the numerical formula 20 with respect to the parameter K Sdq .
Figure 2015144502
 Therefore, the parameter K Sdq can be calculated from a known value by Equation 30.

以上のことから、パラメータＫＳｑ，ＫＳｄｑは、下記の手順で演算する。
（１）数式２０により、係数Ａを演算する。
（２）数式２８により、係数Ｂ，Ｄを演算する。
（３）数式２９により、パラメータＫＳｑを演算する。
（４）数式３０により、パラメータＫＳｄｑを演算する。 From the above, the parameters K Sq and K Sdq are calculated according to the following procedure.
(1) The coefficient A is calculated by Equation 20.
(2) The coefficients B and D are calculated using Equation 28.
(3) The parameter K Sq is calculated by Equation 29.
(4) The parameter K Sdq is calculated by Equation 30.

次に、電機子抵抗、及び、電流の制御誤差を考慮した場合のパラメータ推定の原理と方法について説明する。
電機子抵抗、及び、電流の制御誤差を考慮した場合、ｑ軸交番電流指令値ｉｑｈ ＊を前述した数式１のように制御すると、ｄ，ｑ軸電圧ｖｄ，ｖｑは数式３１により表される。但し、電機子抵抗推定値ｒａＭは、電機子抵抗の真値ｒａに等しいものとする。

Figure 2015144502
Next, the principle and method of parameter estimation in consideration of armature resistance and current control error will be described.
When the armature resistance and the current control error are taken into account, when the q-axis alternating current command value i qh * is controlled as shown in Equation 1 above, the d and q-axis voltages v d and v q are expressed by Equation 31. Is done. However, armature resistance estimate r aM is assumed to be equal to the true value r a of the armature resistance.
Figure 2015144502

数式３１より、ｄ軸電流の２倍高調波に起因する電機子抵抗の電圧降下、及び、ｑ軸電流の３倍高調波に起因する電機子抵抗の電圧降下は、図１の電機子抵抗補償器３３によって補償されるので、ｄ軸磁束Ψｄは直流成分のみ、ｑ軸磁束Ψｑは基本波成分のみとなる。
これを数式で表すと、数式３２となる。

Figure 2015144502
From Equation 31, the voltage drop of the armature resistance caused by the second harmonic of the d-axis current and the voltage drop of the armature resistance caused by the third harmonic of the q-axis current are shown in FIG. Therefore, the d-axis magnetic flux Ψ d has only a DC component, and the q-axis magnetic flux Ψ q has only a fundamental wave component.
When this is expressed by a mathematical formula, a mathematical formula 32 is obtained.
Figure 2015144502

数式１５の磁束モデルにおいて、ｄ，ｑ軸磁束Ψｄ，Ψｑを数式３２のように制御した場合、ｄ，ｑ軸電流ｉｄ，ｉｑは数式３３のようになる。

Figure 2015144502
In the magnetic flux model of Equation 15, when the d and q axis magnetic fluxes Ψ d and Ψ q are controlled as in Expression 32, the d and q axis currents i d and i q are as in Expression 33.
Figure 2015144502

数式３２，数式３３より、電機子抵抗、及び、電流の制御誤差を考慮する場合のパラメータ推定は、先に述べた、電機子抵抗を零に近似でき、電流を指令値に正確に制御できる場合のパラメータ推定において、数式３４に示すように、フーリエ係数Ψｑ（ｂ１），Ｉｑ（ｂ１），Ｉｑ（ｂ３），Ｉｄ（ａ２）を第２のフーリエ係数Ψｑ（ｂ１）’，Ｉｑ（ｂ１）’，Ｉｑ（ｂ３）’，Ｉｄ（ａ２）’にそれぞれ置き換えることで実現可能である。

Figure 2015144502
According to Equations 32 and 33, the parameter estimation when the armature resistance and the current control error are taken into account can be approximated to the armature resistance as described above, and the current can be accurately controlled to the command value. In the parameter estimation, the Fourier coefficients Ψ q (b1) , I q (b1) , I q (b3) , I d (a2) are converted into the second Fourier coefficients Ψ q (b1) ′, This can be realized by replacing with I q (b1) ′, I q (b3) ′, and I d (a2) ′.
Figure 2015144502

次に、第２のフーリエ係数Ψｑ（ｂ１）’，Ｉｑ（ｂ１）’，Ｉｑ（ｂ３）’，Ｉｄ（ａ２）’の演算方法について説明する。
第２のｑ軸電流の基本波成分のフーリエ正弦係数Ｉｑ（ｂ１）’は、数式３５により演算する。

Figure 2015144502
Next, a method of calculating the second Fourier coefficients Ψ q (b1) ′, I q (b1) ′, I q (b3) ′, I d (a2) ′ will be described.
The Fourier sine coefficient I q (b1) ′ of the fundamental wave component of the second q-axis current is calculated by Equation 35.
Figure 2015144502

第２のｑ軸電流の３倍高調波成分のフーリエ正弦係数Ｉｑ（ｂ３）’と第２のｄ軸電流の２倍高調波成分のフーリエ余弦係数Ｉｄ（ａ２）’とは、数式３６により演算する。

Figure 2015144502
The Fourier sine coefficient I q (b3) ′ of the third harmonic component of the second q-axis current and the Fourier cosine coefficient I d (a2) ′ of the second harmonic component of the second d-axis current are expressed by Equation 36. It calculates by.
Figure 2015144502

磁束は電流と同位相であり、磁束に起因する電圧降下は磁束から９０度進みであることから、第２のｄ軸磁束の基本波成分のフーリエ正弦係数Ψｑ（ｂ１）’は、数式３７により演算する。

Figure 2015144502
Since the magnetic flux is in phase with the current and the voltage drop caused by the magnetic flux is 90 degrees ahead of the magnetic flux, the Fourier sine coefficient Ψ q (b1) ′ of the fundamental wave component of the second d-axis magnetic flux is expressed by Equation 37. It calculates by.
Figure 2015144502

このようにして、数式１５における磁束モデルのパラメータのうち、ＫＬｑ，ＫＳｑ，ＫＳｑｄ，ＫＳｄｑを推定することができる。なお、前述したように、他のパラメータＫＬｄ，ＫＳｄ，Ｉ０，φ０は既知の値として与えられる。
これにより、数式１５に示した磁束モデルのパラメータのオートチューニングを実現することができる。 In this way, K Lq , K Sq , K Sqd , and K Sdq among the parameters of the magnetic flux model in Expression 15 can be estimated. As described above, the other parameters K Ld , K Sd , I 0 , and φ 0 are given as known values.
Thereby, auto-tuning of the parameters of the magnetic flux model shown in Formula 15 can be realized.

１１ｕ ｕ相電流検出器
１１ｗ ｗ相電流検出器
１３ ＰＷＭ回路
１４ 座標変換器
１５ 座標変換器
１８ａ ローパスフィルタ
１８ｂ ローパスフィルタ
１９ａ 減算器
１９ｂ 減算器
２０ａ ｄ軸電流調節器
２０ｂ ｑ軸電流調節器
２１ 直流電機子抵抗補償器
２２ａ 加算器
２２ｂ 加算器
３０ 積分器
３１ ｑ軸交番電流指令演算器
３２ 電圧補償値演算器
３３ 高調波電機子抵抗補償器
３４ 加算器
３５ フーリエ係数演算器
３６ アドミタンス推定部
３７ インピーダンス演算器
４０ パラメータ推定部
５０ 三相交流電源
６０ 整流回路
７０ 電力変換器
８０ 永久磁石形同期電動機
１０１ 振幅演算器
１０２ 有効電流・無効電流演算器
１０３ａ 乗算器
１０３ｂ 乗算器
１０４ａ 減算器
１０４ｂ 減算器
１０５ａ 推定器
１０５ｂ 推定器 11u u-phase current detector 11w w-phase current detector 13 PWM circuit 14 coordinate converter 15 coordinate converter 18a low-pass filter 18b low-pass filter 19a subtractor 19b subtractor 20a d-axis current regulator 20b q-axis current regulator 21 Child resistance compensator 22a Adder 22b Adder 30 Integrator 31 q-axis alternating current command calculator 32 Voltage compensation value calculator 33 Harmonic armature resistance compensator 34 Adder 35 Fourier coefficient calculator 36 Admittance estimator 37 Impedance calculation 40 Parameter Estimator 50 Three-phase AC Power Supply 60 Rectifier Circuit 70 Power Converter 80 Permanent Magnet Synchronous Motor 101 Amplitude Calculator 102 Active Current / Reactive Current Calculator 103a Multiplier 103b Multiplier 104a Subtractor 104b Subtractor 105a Estimator 105b Estimator

Claims (4)

電力変換器により永久磁石形同期電動機に供給する電流及び電圧を、前記電動機の回転子磁極方向に平行なｄ軸とこのｄ軸に直交するｑ軸とからなるｄ，ｑ直交回転座標上で制御するための制御装置であって、電動機鉄芯の磁気飽和特性を少なくとも考慮した磁束モデルに基づいて構成される制御装置において、
前記電動機のｄ軸電流の直流成分を指令値に制御する第１の手段と、
前記ｑ軸に正弦波の交番電圧を印加する第２の手段と、
前記ｑ軸電流の基本波成分のフーリエ係数を演算する第３の手段と、
前記ｑ軸電流の３倍高調波成分のフーリエ係数を演算する第４の手段と、
前記ｄ軸電流の直流成分のフーリエ係数を演算する第５の手段と、
前記ｄ軸電流の２倍高調波成分のフーリエ係数を演算する第６の手段と、
前記ｑ軸電圧の基本波成分のフーリエ係数を演算する第７の手段と、
前記ｑ軸電圧の基本波成分のフーリエ係数、及び、交番電圧の角周波数から、前記電動機のｑ軸磁束の基本波成分のフーリエ係数を演算する第８の手段と、
前記ｄ軸電流の直流成分のフーリエ係数から、前記ｄ軸磁束の直流成分のフーリエ係数を演算する第９の手段と、
前記ｑ軸電流の基本波成分のフーリエ係数、前記ｑ軸電流の３倍高調波成分のフーリエ係数、前記ｄ軸電流の直流成分のフーリエ係数、前記ｄ軸電流の２倍高調波成分のフーリエ係数、前記ｑ軸磁束の基本波成分のフーリエ係数、及び、前記ｄ軸磁束の直流成分のフーリエ係数から、前記ｑ軸電流に対するｑ軸磁束の傾きの最大値に相当する第１のパラメータ、ｑ軸上の磁気飽和の度合いを示す第２のパラメータ、及び、ｄ，ｑ軸間の干渉の度合いを示す第３，第４のパラメータを求める第１０の手段と、
を備え、
前記第１〜第４のパラメータを用いて前記磁束モデルを構成することを特徴とする永久磁石形同期電動機の制御装置。 The current and voltage supplied to the permanent magnet type synchronous motor by the power converter are controlled on the d and q orthogonal rotation coordinates composed of the d axis parallel to the rotor magnetic pole direction of the motor and the q axis orthogonal to the d axis. In a control device configured to be based on a magnetic flux model that takes into account at least the magnetic saturation characteristics of the motor iron core,
First means for controlling the DC component of the d-axis current of the motor to a command value;
A second means for applying a sinusoidal alternating voltage to the q-axis;
A third means for calculating a Fourier coefficient of the fundamental wave component of the q-axis current;
A fourth means for calculating a Fourier coefficient of a third harmonic component of the q-axis current;
A fifth means for calculating a Fourier coefficient of a DC component of the d-axis current;
A sixth means for calculating a Fourier coefficient of a second harmonic component of the d-axis current;
A seventh means for calculating a Fourier coefficient of the fundamental wave component of the q-axis voltage;
An eighth means for calculating a Fourier coefficient of a fundamental wave component of the q-axis magnetic flux of the motor from a Fourier coefficient of the fundamental wave component of the q-axis voltage and an angular frequency of the alternating voltage;
A ninth means for calculating a Fourier coefficient of a DC component of the d-axis magnetic flux from a Fourier coefficient of a DC component of the d-axis current;
Fourier coefficient of fundamental wave component of q-axis current, Fourier coefficient of third harmonic component of q-axis current, Fourier coefficient of DC component of d-axis current, Fourier coefficient of second harmonic component of d-axis current A first parameter corresponding to the maximum value of the gradient of the q-axis magnetic flux with respect to the q-axis current from the Fourier coefficient of the fundamental wave component of the q-axis magnetic flux and the Fourier coefficient of the DC component of the d-axis magnetic flux, A tenth means for obtaining a second parameter indicating the degree of magnetic saturation and third and fourth parameters indicating the degree of interference between the d and q axes;
With
A control apparatus for a permanent magnet synchronous motor, wherein the magnetic flux model is configured using the first to fourth parameters. 請求項１に記載した永久磁石形同期電動機の制御装置において、
前記第１０の手段は、
前記ｑ軸電流の基本波成分のフーリエ係数から、第２のｑ軸電流の基本波成分のフーリエ正弦係数を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｑ軸電流の３倍高調波成分のフーリエ係数とから、第２のｑ軸電流の３倍高調波成分のフーリエ正弦係数を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｄ軸電流の２倍高調波成分のフーリエ係数とから、第２のｄ軸電流の２倍高調波成分のフーリエ余弦係数を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｑ軸電圧の基本波成分のフーリエ係数とから、第２のｑ軸磁束の基本波成分のフーリエ正弦係数を演算する手段と、
前記第２のｑ軸電流の基本波成分のフーリエ正弦係数、前記第２のｑ軸電流の３倍高調波成分のフーリエ正弦係数、前記ｄ軸電流の直流成分のフーリエ係数、前記第２のｄ軸電流の２倍高調波成分のフーリエ余弦係数、前記第２のｑ軸磁束の基本波成分のフーリエ正弦係数、及び、前記ｄ軸磁束の直流成分のフーリエ係数から、前記第１〜第４のパラメータを演算する手段と、
を備えたことを特徴とする永久磁石形同期電動機の制御装置。 In the control device for the permanent magnet type synchronous motor according to claim 1,
The tenth means includes
Means for calculating a Fourier sine coefficient of the fundamental wave component of the second q-axis current from a Fourier coefficient of the fundamental wave component of the q-axis current;
Means for calculating the Fourier sine coefficient of the third harmonic component of the second q-axis current from the Fourier coefficient of the fundamental component of the q-axis current and the Fourier coefficient of the third harmonic component of the q-axis current;
Means for calculating the Fourier cosine coefficient of the second harmonic component of the second d-axis current from the Fourier coefficient of the fundamental component of the q-axis current and the Fourier coefficient of the second harmonic component of the d-axis current;
Means for calculating a Fourier sine coefficient of the fundamental wave component of the second q-axis magnetic flux from a Fourier coefficient of the fundamental wave component of the q-axis current and a Fourier coefficient of the fundamental wave component of the q-axis voltage;
A Fourier sine coefficient of a fundamental wave component of the second q-axis current, a Fourier sine coefficient of a third harmonic component of the second q-axis current, a Fourier coefficient of a DC component of the d-axis current, the second d From the Fourier cosine coefficient of the second harmonic component of the axial current, the Fourier sine coefficient of the fundamental wave component of the second q-axis magnetic flux, and the Fourier coefficient of the DC component of the d-axis magnetic flux, the first to fourth Means for calculating parameters;
A control device for a permanent magnet type synchronous motor. 請求項１または２に記載した永久磁石形同期電動機の制御装置において、
前記第１の手段は、
前記ｄ軸電流の直流成分の指令値とｄ軸電流検出値との偏差からｄ軸電圧指令値を演算する手段を備え、
前記第２の手段は、
ｑ軸電流指令値を正弦波に制御する手段と、
前記ｑ軸電流指令値とｑ軸電流検出値との偏差からｑ軸電圧フィードバック制御値を演算する手段と、
前記ｑ軸電流の基本波成分のフーリエ係数と前記ｑ軸電圧の基本波成分のフーリエ係数とからｑ軸リアクタンス及びｑ軸電機子抵抗を推定する手段と、
前記ｑ軸電流指令値、前記ｑ軸リアクタンスの推定値、及び、前記ｑ軸電機子抵抗の推定値からｑ軸電圧フィードフォワード補償値を演算する手段と、
前記ｑ軸電圧フィードバック制御値と前記ｑ軸電圧フィードフォワード補償値とからｑ軸電圧指令値を演算する手段と、
を備えたことを特徴とする永久磁石形同期電動機の制御装置。 In the control device for a permanent magnet type synchronous motor according to claim 1 or 2,
The first means includes
Means for calculating a d-axis voltage command value from a deviation between a command value of a DC component of the d-axis current and a detected d-axis current value;
The second means includes
means for controlling the q-axis current command value to a sine wave;
Means for calculating a q-axis voltage feedback control value from a deviation between the q-axis current command value and the q-axis current detection value;
Means for estimating the q-axis reactance and the q-axis armature resistance from the Fourier coefficient of the fundamental wave component of the q-axis current and the Fourier coefficient of the fundamental wave component of the q-axis voltage;
Means for calculating a q-axis voltage feedforward compensation value from the q-axis current command value, the estimated value of the q-axis reactance, and the estimated value of the q-axis armature resistance;
Means for calculating a q-axis voltage command value from the q-axis voltage feedback control value and the q-axis voltage feedforward compensation value;
A control device for a permanent magnet type synchronous motor. 請求項１〜３の何れか１項に記載した永久磁石形同期電動機の制御装置において、
前記ｄ軸電流の２倍高調波成分のフーリエ係数と電機子抵抗とから、前記ｄ軸電流の２倍高調波成分に起因する前記電機子抵抗の電圧降下を演算する手段と、
前記ｑ軸電流の３倍高調波成分のフーリエ係数と電機子抵抗とから、前記ｑ軸電流の３倍高調波成分に起因する前記電機子抵抗の電圧降下を演算する手段と、
前記ｄ軸電流の２倍高調波成分に起因する前記電機子抵抗の電圧降下からｄ軸電圧指令値を演算する手段と、
前記ｑ軸電流の３倍高調波成分に起因する前記電機子抵抗の電圧降下からｑ軸電圧指令値を演算する手段と、
を備えたことを特徴とする永久磁石形同期電動機の制御装置。 In the control apparatus for the permanent magnet type synchronous motor according to any one of claims 1 to 3,
Means for calculating a voltage drop of the armature resistance caused by a second harmonic component of the d-axis current from a Fourier coefficient of the second harmonic component of the d-axis current and an armature resistance;
Means for calculating a voltage drop of the armature resistance caused by a third harmonic component of the q-axis current from a Fourier coefficient of the third harmonic component of the q-axis current and an armature resistance;
Means for calculating a d-axis voltage command value from a voltage drop of the armature resistance caused by a second harmonic component of the d-axis current;
Means for calculating a q-axis voltage command value from a voltage drop of the armature resistance caused by a third harmonic component of the q-axis current;
A control device for a permanent magnet type synchronous motor.
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