JP2013005573A - Control device for ac motor, and refrigeration and air conditioning device using the same - Google Patents
Control device for ac motor, and refrigeration and air conditioning device using the same Download PDFInfo
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- JP2013005573A JP2013005573A JP2011133796A JP2011133796A JP2013005573A JP 2013005573 A JP2013005573 A JP 2013005573A JP 2011133796 A JP2011133796 A JP 2011133796A JP 2011133796 A JP2011133796 A JP 2011133796A JP 2013005573 A JP2013005573 A JP 2013005573A
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/14—Arrangements for reducing ripples from dc input or output
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
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Abstract
Description
本発明は、交流モータの制御装置に関するものであり、特に、インバータの直流母線電流に基づいてモータ電流を推定する方法に関する。 The present invention relates to an AC motor control device, and more particularly, to a method for estimating a motor current based on a DC bus current of an inverter.
交流モータに流れるモータ電流を正確に推定し、これを交流モータの制御に用いることができれば、交流モータをより高性能に駆動することができる。 If the motor current flowing through the AC motor can be accurately estimated and used for controlling the AC motor, the AC motor can be driven with higher performance.
特許文献1では、インバータの直流母線電流から2相分の電流を検出し、この検出値に基づいてモータ電流を推定する。このためには、キャリア1周期内で2回電流検出する必要があり、高速なA/D変換器を必要とする。 In Patent Document 1, a current for two phases is detected from a DC bus current of an inverter, and a motor current is estimated based on the detected value. For this purpose, it is necessary to detect current twice within one carrier cycle, and a high-speed A / D converter is required.
特許文献2では、2相分の電流を検出しなければならないという特許文献1とは異なり、1相分の検出値のみでモータ電流を推定できる。ただし、モータ電流を1回推定するには、所定の期間において、直流母線電流を複数回検出する必要があり、高速なA/D変換器を必要とする点は改善されていない。 In Patent Document 2, unlike in Patent Document 1 in which the current for two phases must be detected, the motor current can be estimated using only the detected value for one phase. However, in order to estimate the motor current once, it is necessary to detect the DC bus current a plurality of times in a predetermined period, and the point that a high-speed A / D converter is required is not improved.
交流モータを低負荷あるいは低速域で駆動する場合、変調率を低くするため、インバータの直流母線電流の通電期間を短くする。直流母線電流の通電期間が短いと、通電期間内に直流母線電流を複数回検出することが困難なため、従来の技術では、直流母線電流の検出に誤差が生じ、モータ電流を正確に推定できず、交流モータを高性能に駆動できないという問題が生じる。 When the AC motor is driven in a low load or low speed region, the energization period of the DC bus current of the inverter is shortened in order to reduce the modulation rate. If the energization period of the DC bus current is short, it is difficult to detect the DC bus current multiple times within the energization period, so the conventional technology has an error in the detection of the DC bus current and can accurately estimate the motor current. Therefore, there arises a problem that the AC motor cannot be driven with high performance.
本発明の目的は、インバータの通電期間が短い場合でも直線母線電流を正確に検出することで、モータ電流を正確に推定し、交流モータを高性能に駆動する手法を提供することである。 An object of the present invention is to provide a technique for accurately estimating a motor current and accurately driving an AC motor by accurately detecting a linear bus current even when an inverter energization period is short.
直流電源と、該直流電源より供給される直流電力を交流電力に変換するインバータと、該インバータに備わるスイッチ素子を制御するインバータ制御回路と、前記インバータに流れる直流母線電流を検出する直流母線電流検出器と、該直流母線電流検出器で検出した直流母線電流を平滑化するローパスフィルタと、該ローパスフィルタで平滑された直流母線電流の減衰分を補正する補正器と、を備える交流モータの制御装置。 DC power supply, inverter for converting DC power supplied from the DC power supply to AC power, inverter control circuit for controlling a switch element provided in the inverter, and DC bus current detection for detecting DC bus current flowing in the inverter AC motor control device comprising: a power supply; a low pass filter that smoothes the DC bus current detected by the DC bus current detector; and a corrector that corrects the attenuation of the DC bus current smoothed by the low pass filter .
本発明により、変調率が低い場合においても、直線母線電流を正確に検出できる。このため、低負荷あるいは低速域においても交流モータを高性能に駆動できる。 According to the present invention, even when the modulation rate is low, the straight-line bus current can be accurately detected. For this reason, the AC motor can be driven with high performance even in a low load or low speed range.
以下、図面を用いて本発明の各実施例を説明する。 Embodiments of the present invention will be described below with reference to the drawings.
図1〜図6を用いて実施例1の制御装置を説明する。図1において、交流モータ1は、インバータ2から印加される三相交流電流Iu,Iv,Iwに応じたトルクを出力する。インバータ2は、スイッチ素子Sup,Sun,Svp,Svn,Swp,Swnを備え、交流モータ1へ三相交流電圧Vu,Vv,Vwを印加し、交流電力を供給する。直流電源3は、直流電圧VDCをインバータ2へ印加し、直流電力を供給する。直流母線電流検出器4は、インバータ2の直流母線電流IDCを検出する。ローパスフィルタ5は、直流母線電流検出器4の検出値を平滑化する。補正器6は、ローパスフィルタ5による検出値の減衰分を補正する。インバータ制御回路7は、スイッチ素子Sup,Sun,Svp,Svn,Swp,Swnのオン・オフを制御する。なお、インバータ制御回路7内のPWM信号発生部7a,ベクトル制御部7b,電流推定部7cの詳細については、実施例2で説明することとする。 The control apparatus of Example 1 is demonstrated using FIGS. In FIG. 1, an AC motor 1 outputs torque according to three-phase AC currents Iu, Iv, Iw applied from an inverter 2. The inverter 2 includes switch elements Sup, Sun, Svp, Svn, Swp, Swn, applies three-phase AC voltages Vu, Vv, Vw to the AC motor 1 and supplies AC power. DC power supply 3 applies DC voltage VDC to inverter 2 to supply DC power. The DC bus current detector 4 detects the DC bus current IDC of the inverter 2. The low-pass filter 5 smoothes the detection value of the DC bus current detector 4. The corrector 6 corrects the attenuation of the detection value by the low-pass filter 5. The inverter control circuit 7 controls ON / OFF of the switch elements Sup, Sun, Svp, Svn, Swp, Swn. The details of the PWM signal generation unit 7a, the vector control unit 7b, and the current estimation unit 7c in the inverter control circuit 7 will be described in the second embodiment.
インバータ2は、(数1)の三相交流電圧Vu,Vv,Vwを交流モータ1に印加し、交流モータ1には、(数2)の三相交流電流Iu,Iv,Iwが流れる。(数1)の三相交流電圧は、図2(a)に示され、(数2)の三相交流電流は図2(b)に示される。 The inverter 2 applies the three-phase AC voltages Vu, Vv, and Vw of (Equation 1) to the AC motor 1, and the three-phase AC currents Iu, Iv, and Iw of (Equation 2) flow through the AC motor 1. The three-phase AC voltage of (Equation 1) is shown in FIG. 2 (a), and the three-phase AC current of (Equation 2) is shown in FIG. 2 (b).
なお、数1,数2において、V1はモータ電圧、I1はモータ電流、θvはU軸を基準とする電圧位相、ψは電圧/電流位相差 である。 In Equations 1 and 2, V1 is a motor voltage, I1 is a motor current, θv is a voltage phase based on the U axis, and ψ is a voltage / current phase difference.
次に、図3を用いて、電圧・電流の各成分を説明する。図3において、U軸は交流モータ1の固定子のU相コイル方向を表す。モータ電圧V1,モータ電流I1のU軸方向の成分を、それぞれ、U相電圧Vu,U相電流Iuとする。同様に、図3では省略するが、V軸方向の成分を、V相電圧Vv,V相電流Ivとし、W軸方向の成分を、W相電圧Vw,W相電流Iwとする。また、モータ電流I1のモータ電圧V1方向の成分を有効電流Iaとし、それに直交する成分を無効電流Irとする。 Next, voltage and current components will be described with reference to FIG. In FIG. 3, the U axis represents the U phase coil direction of the stator of the AC motor 1. The components of the motor voltage V1 and the motor current I1 in the U-axis direction are respectively a U-phase voltage Vu and a U-phase current Iu. Similarly, although omitted in FIG. 3, the components in the V-axis direction are V-phase voltage Vv and V-phase current Iv, and the components in the W-axis direction are W-phase voltage Vw and W-phase current Iw. A component of the motor current I1 in the direction of the motor voltage V1 is defined as an effective current Ia, and a component orthogonal thereto is defined as a reactive current Ir.
直流母線電流検出器4は、三相交流電流Iu,Iv,Iwのいずれかを直流母線電流IDCとして検出する。三相交流電流のうち何れが検出されるかは、スイッチ素子Sup,Sun,Svp,Svn,Swp,Swnのオン・オフの組み合わせに依存する。図4を用いて、スイッチ素子のオン・オフの組み合わせと直流母線電流IDCの理論上の波形の関係を詳細に説明する。 The DC bus current detector 4 detects any of the three-phase AC currents Iu, Iv, Iw as the DC bus current IDC. Which of the three-phase alternating currents is detected depends on the combination of ON / OFF of the switch elements Sup, Sun, Svp, Svn, Swp, and Swn. With reference to FIG. 4, the relationship between the ON / OFF combination of the switch elements and the theoretical waveform of the DC bus current IDC will be described in detail.
図4(a)は、キャリア周期Tcのキャリア信号である。ここでは、キャリア信号として三角波を用いた例を図示するが、鋸波を用いても良い。 FIG. 4A shows a carrier signal having a carrier period Tc. Here, an example using a triangular wave as a carrier signal is shown, but a sawtooth wave may be used.
インバータ制御回路7は、キャリア信号と三相交流電圧の指令値Vu*,Vv*,Vw*を比較して、スイッチ素子Sup,Sun,Svp,Svn,Swp,Swnのオン・オフを決定する。 The inverter control circuit 7 compares the carrier signal and the command values Vu * , Vv * , Vw * of the three-phase AC voltage, and determines on / off of the switch elements Sup, Sun, Svp, Svn, Swp, Swn.
図4(b)のSuのハッチングは、キャリア信号がVu*以下となる期間に相当し、スイッチ素子Supをオンにする期間を表す。同様に、図4(c)のSvのハッチングは、キャリア信号がVv*以下となる期間に相当し、スイッチ素子Svpをオンにする期間を表し、図4(d)のSwのハッチングは、キャリア信号がVw*以下となる期間に相当し、スイッチ素子Swpをオンにする期間を表す。 The hatching of Su in FIG. 4B corresponds to a period during which the carrier signal is equal to or lower than Vu * and represents a period during which the switch element Sup is turned on. Similarly, the hatching of Sv in FIG. 4C corresponds to a period in which the carrier signal is equal to or lower than Vv * , and represents a period in which the switch element Svp is turned on. The hatching of Sw in FIG. This corresponds to a period during which the signal is equal to or lower than Vw * and represents a period during which the switch element Swp is turned on.
図1に示すスイッチ素子SupとSunは、それぞれ相補的に動作し、一方がオンであるとき、他方はオフである。同様に、スイッチ素子SvpとSvn,SwpとSwnも相補的に動作し、一方がオンであるとき、他方はオフである。ただし、インバータ2の短絡防止のため、両者がオフとなるデッドタイムを設けても良い。 The switch elements Sup and Sun shown in FIG. 1 operate in a complementary manner, and when one is on, the other is off. Similarly, the switch elements Svp and Svn, Swp and Swn operate in a complementary manner, and when one is on, the other is off. However, in order to prevent a short circuit of the inverter 2, a dead time during which both are turned off may be provided.
図4に示すように、Sup,Svp,Swpの全てがオンのときと全てがオフのときには、直流母線電流IDCは流れない。一方、Supのみオンとなる期間Tuでは、U相電流Iuが順方向に流れ、SupとSvpがオンとなる期間Twでは、W相電流Iwが逆方向に流れる。この結果、論理的には、図4(e)に示す直流母線電流IDCが得られる。 As shown in FIG. 4, the DC bus current IDC does not flow when all of Sup, Svp, and Swp are on and when all are off. On the other hand, in the period Tu in which only Sup is on, the U-phase current Iu flows in the forward direction, and in the period Tw in which Sup and Svp are on, the W-phase current Iw flows in the reverse direction. As a result, logically, the DC bus current IDC shown in FIG.
次に、図4(e)を用いて、直流母線電流検出器4による直流母線電流IDCの検出方法を具体的に説明する。直流母線電流検出器4は、キャリア信号下り時のSupオンからTs後のタイミングAにおいて、U相電流Iuを検出する。Tsを設ける理由については後述する。なお、ここで示した例に代え、キャリア信号上り時のタイミングBにおいてU相電流を検出しても良いし、W相電流Iwを検出しても良い。 Next, a method for detecting the DC bus current IDC by the DC bus current detector 4 will be specifically described with reference to FIG. The DC bus current detector 4 detects the U-phase current Iu at the timing A Ts after Sup on at the time of falling of the carrier signal. The reason for providing Ts will be described later. Instead of the example shown here, the U-phase current may be detected at the timing B when the carrier signal goes up, or the W-phase current Iw may be detected.
次に、図5,図6を用いて、実際に観測される直流母線電流IDCを検出する例を説明する。なお、図5は、変調率が低く期間Tuが短い場合の直流母線電流IDCの波形図であり、図6は、変調率が高く期間Tuが長い場合の直流母線電流IDCの波形図である。 Next, an example of detecting the actually observed DC bus current IDC will be described with reference to FIGS. FIG. 5 is a waveform diagram of the DC bus current IDC when the modulation rate is low and the period Tu is short. FIG. 6 is a waveform diagram of the DC bus current IDC when the modulation rate is high and the period Tu is long.
まず、変調率が低い場合について、図5を用いて説明する。スイッチ素子に遅れや非線形特性がある実際の回路では、直流母線電流IDCの波形は、図4(e)で示した理想的な形状とはならず、図5(b)の実線で示すように脈動する。このため、U相電流Iuを正確に検出するには、脈動が収まってから電流検出する必要がある。しかし、変調率が低い場合、三相電圧Vu,Vv,Vwがゼロ近傍にあることから期間Tuが短く、リンギングが収まる前に次のスイッチングが行われ、U相電流Iuを正確に検出できない。 First, the case where the modulation rate is low will be described with reference to FIG. In an actual circuit in which the switching element has a delay or non-linear characteristic, the waveform of the DC bus current IDC does not have the ideal shape shown in FIG. 4 (e), but as shown by the solid line in FIG. 5 (b). It pulsates. For this reason, in order to accurately detect the U-phase current Iu, it is necessary to detect the current after the pulsation has subsided. However, when the modulation rate is low, the three-phase voltages Vu, Vv, and Vw are near zero, so the period Tu is short, and the next switching is performed before ringing is settled, and the U-phase current Iu cannot be accurately detected.
そこで、本実施例では、図1に示すように、直流母線電流検出器4の後段に、(数3)に示す特性を持ったローパスフィルタ5を設けた。ここで、(数3)において、Tfは時定数である。(数3)の特性のローパスフィルタ5を用いることで、実線で示した直流母線電流IDCは点線で示すフィルタ値IDC′に平滑化される。なお、(数3)に示すローパスフィルタ5の特性は一例であり、より高次のローパスフィルタでもよい。 Therefore, in this embodiment, as shown in FIG. 1, a low-pass filter 5 having the characteristic shown in (Equation 3) is provided at the subsequent stage of the DC bus current detector 4. Here, in (Equation 3), Tf is a time constant. By using the low pass filter 5 having the characteristic of (Equation 3), the DC bus current IDC indicated by the solid line is smoothed to the filter value IDC ′ indicated by the dotted line. The characteristic of the low-pass filter 5 shown in (Equation 3) is an example, and a higher-order low-pass filter may be used.
図5のように変調率が低い場合、キャリア周期Tcに対して期間Tuが短いため、脈動が早期に収束し、直流母線電流IDCの波形の立ち上がるタイミングであるタイミングCにおけるフィルタ値IDC′をゼロと近似できる。なお、タイミングCは、理想的には、図4における、キャリア信号立下り時のSupのオン時である。このとき、U相電流Iuの検出値Iu′は(数4)で求めることができる。 When the modulation rate is low as shown in FIG. 5, since the period Tu is short with respect to the carrier period Tc, the pulsation converges early, and the filter value IDC ′ at timing C, which is the timing when the waveform of the DC bus current IDC rises, is zero. Can be approximated. Note that the timing C is ideally when Sup is turned on when the carrier signal falls in FIG. At this time, the detected value Iu ′ of the U-phase current Iu can be obtained by (Equation 4).
次に、変調率が高い場合について図6を用いて説明する。図5では、タイミングCにおけるフィルタ値IDC′をゼロと近似できたが、図6では、キャリア周期Tcに対して期間Tuが長いため、タイミングCにおいても脈動は収束せず、フィルタ値IDC′もゼロに収束しない。この場合、タイミングCにおけるフィルタ値IDC′をゼロと近似することなく正確に演算するのが望ましい。 Next, the case where the modulation rate is high will be described with reference to FIG. In FIG. 5, the filter value IDC ′ at the timing C can be approximated to zero. However, in FIG. 6, the pulsation does not converge at the timing C because the period Tu is longer than the carrier cycle Tc, and the filter value IDC ′ is also Does not converge to zero. In this case, it is desirable to accurately calculate the filter value IDC ′ at timing C without approximating it to zero.
そこで、本実施例では、図1に示すように、ローパスフィルタ5の後段に補正器6を設けた。補正器6は、(数4)に補正係数εを代入し、U相電流検出値Iu′からU相電流Iuを逆算する。これにより、ローパスフィルタ5の減衰分を補正することができる。インバータ制御回路7は、このようにして得られた三相交流電流Iu,Iv,Iwに基づいてスイッチ素子Sup,Sun,Svp,Svn,Swp,Swnのオン・オフを制御するので、交流モータ1を安定駆動することができる。 Therefore, in this embodiment, as shown in FIG. 1, the corrector 6 is provided after the low-pass filter 5. The corrector 6 substitutes the correction coefficient ε into (Equation 4), and reversely calculates the U-phase current Iu from the U-phase current detection value Iu ′. Thereby, the attenuation of the low-pass filter 5 can be corrected. The inverter control circuit 7 controls on / off of the switch elements Sup, Sun, Svp, Svn, Swp, Swn based on the three-phase AC currents Iu, Iv, Iw thus obtained. Can be driven stably.
以上で説明したように、ローパスフィルタ5および補正器6を用いた本実施例の構成により、変調率が低い場合にも高い場合にもともに正しく電流を検出することができる。このため、低負荷あるいは低速域においても交流モータ1を高性能に駆動でき、駆動範囲をワイドレンジ化できる。 As described above, the configuration of the present embodiment using the low-pass filter 5 and the corrector 6 can correctly detect the current both when the modulation rate is low and when the modulation rate is high. For this reason, the AC motor 1 can be driven with high performance even in a low load or low speed range, and the drive range can be widened.
図1を用いて実施例2を説明する。なお、実施例1と同等の点については説明を省略することとする。 Example 2 will be described with reference to FIG. Note that the description of the same points as in the first embodiment will be omitted.
本実施例は、図1に示すように、インバータ制御回路7内に、PWM信号発生部7a,ベクトル制御部7b,電流推定部7cを備え、これらにより、モータ電流I1を推定し、交流モータ1をベクトル制御するものである。以下、各々につき詳細に説明する。 In this embodiment, as shown in FIG. 1, the inverter control circuit 7 includes a PWM signal generation unit 7a, a vector control unit 7b, and a current estimation unit 7c. Is vector controlled. Each will be described in detail below.
電流推定部7cは、補正器6からの出力に基づき、図3で説明した無効電流Irおよび有効電流Iaを推定し、モータ電流I1の振幅および位相を推定するものである。この推定原理を以下に述べる。 The current estimation unit 7c estimates the reactive current Ir and the effective current Ia described with reference to FIG. 3 based on the output from the corrector 6, and estimates the amplitude and phase of the motor current I1. This estimation principle is described below.
図3から明らかなように、無効電流Irおよび有効電流Iaは、それぞれ(数6),(数7)で表される。 As is apparent from FIG. 3, the reactive current Ir and the effective current Ia are expressed by (Equation 6) and (Equation 7), respectively.
(数2)に(数6),(数7)を代入すると、(数8)を得る。 Substituting (Equation 6) and (Equation 7) into (Equation 2) yields (Equation 8).
また、(数4)で表されるU相電流検出値Iu′を、図2に示す区間Q1(θv1≦θv≦θv2)で積分し、(数9)の積分値S1を求める。同様に、Q2(θv2≦θv≦θv3)で積分し、(数10)の積分値S2を求める。 Further, the U-phase current detection value Iu ′ represented by (Equation 4) is integrated in the section Q1 (θv1 ≦ θv ≦ θv2) shown in FIG. 2 to obtain the integrated value S1 of (Equation 9). Similarly, integration is performed with Q2 (θv2 ≦ θv ≦ θv3) to obtain an integral value S2 of (Equation 10).
(数4),(数8)を(数9),(数10)に代入すると、(数11),(数12)を得る。 When (Equation 4) and (Equation 8) are substituted into (Equation 9) and (Equation 10), (Equation 11) and (Equation 12) are obtained.
なお、(数11)中のkr1,Δkr1,ka1,Δka1は(数13)で定義され、(数12)中のkr2,Δkr2,ka2,Δka2は(数14)で定義される。 Note that k r1, Δk r1, k a1 , Δk a1 of (11) in defined in equation (13), is k r2, Δk r2, k a2 , Δk a2 (Equation 12) in equation (14) Defined by
そして、(数11),(数12)より(数15)を得る。 Then, (Equation 15) is obtained from (Equation 11) and (Equation 12).
電流推定部7cは、以上で求めた(数15)より、無効電流Irおよび有効電流Iaを推定できる。なお、(数15)の係数Δkr1,Δkr2,Δka1,Δka2は、ローパスフィルタ5による直流母線電流IDCの減衰分に相当するものであり、補正器6は、減衰の影響を補正するため、予め記憶された、或いは、(数13),(数14)を用いて演算した係数Δkr1,Δkr2,Δka1,Δka2を用いて、直流母線電流IDCを補正する。 The current estimation unit 7c can estimate the reactive current Ir and the effective current Ia from (Equation 15) obtained above. Note that the coefficients Δk r1 , Δk r2 , Δk a1 , Δk a2 in (Equation 15) correspond to the amount of attenuation of the DC bus current IDC by the low-pass filter 5, and the corrector 6 corrects the influence of attenuation. Therefore, the DC bus current IDC is corrected using the coefficients Δk r1 , Δk r2 , Δk a1 , Δk a2 stored in advance or calculated using (Equation 13) and (Equation 14).
なお、以上では連続する区間Q1,Q2のU相電流検出値Iu′積分値を用いる例を説明したが、連続しない区間Q1,Q2の積分値を求めても良く、このようにして電流推定を行う場合には、演算処理を行うタイミングを分散させることができ、演算負荷の集中を防ぐことができる。また、3つ以上の区間Q1,Q2,…,Qnの積分値を求めても良く、より多くの積分値を用いて電流推定を行うことで、ノイズが混入した場合であっても好適な電流推定を行うことができる。これらの方法で電流推定を行う場合は、数13,数14の積分区間を実際に電流検出した区間に変更するだけで対応できる。 In the above description, the U-phase current detection value Iu ′ integral value in the continuous sections Q1 and Q2 has been described. However, the integral value in the non-continuous sections Q1 and Q2 may be obtained. In the case of performing the calculation, it is possible to distribute the timing of performing the calculation processing, and to prevent the calculation load from being concentrated. Further, the integral values of three or more sections Q1, Q2,..., Qn may be obtained, and the current is estimated using more integral values, so that a suitable current can be obtained even when noise is mixed. Estimation can be performed. When current estimation is performed by these methods, it is possible to cope with the problem by simply changing the integration intervals of Equations 13 and 14 to the actual current detection intervals.
電流推定部7cの後段に設けられるベクトル制御部7bは、ベクトル制御に基づいて、モータ電流I1の振幅および位相より電圧指令Vu*,Vv*,Vw*を演算する。PWM信号発生部7a,ベクトル制御部7bからの電圧指令Vu*,Vv*,Vw*に基づいて、スイッチング素子Sup,Sun,Svp,Svn,Swp,Swnの制御信号を出力する。これにより、インバータ2がPWM制御され、交流モータ1が駆動される。 A vector control unit 7b provided in the subsequent stage of the current estimation unit 7c calculates voltage commands Vu * , Vv * , Vw * from the amplitude and phase of the motor current I1 based on vector control. Based on the voltage commands Vu * , Vv * , Vw * from the PWM signal generator 7a and the vector controller 7b, control signals for the switching elements Sup, Sun, Svp, Svn, Swp, Swn are output. Thereby, the inverter 2 is PWM-controlled and the AC motor 1 is driven.
以上で説明したように、本実施例の構成によれば、ローパスフィルタ5による直流母線電流IDCの減衰分を補正し、無効電流Irおよび有効電流Iaを推定することができる。これにより、ローパスフィルタ5による減衰を補正しない構成に比べ、交流モータ1を高性能に駆動することができる。 As described above, according to the configuration of this embodiment, the reactive current Ir and the effective current Ia can be estimated by correcting the attenuation of the DC bus current IDC by the low-pass filter 5. Thereby, compared with the structure which does not correct | amend attenuation by the low-pass filter 5, the AC motor 1 can be driven with high performance.
図7を用いて実施例3を説明する。なお、実施例2と同等の点については説明を省略することとする。 Example 3 will be described with reference to FIG. Note that the description of the same points as in the second embodiment will be omitted.
(数13),(数14)から分かるように、係数Δkr1,Δkr2,Δka1,Δka2は、図2で示した電圧位相θv1,θv2,θv3に依存する。電圧位相θvは交流モータ1の駆動とともに進むため、電流推定毎に(数13),(数14)を用いて係数Δkr1,Δkr2,Δka1,Δka2を再計算する場合には、補正器6に大きな演算負荷がかかることになる。また、係数Δkr1,Δkr2,Δka1,Δka2を予め記憶しておく場合には、補正器6は全ての係数を記憶した大容量のメモリを備えなければならない。 (Number 13), as can be seen from equation (14), the coefficient Δk r1, Δk r2, Δk a1 , Δk a2 is voltage phase θv1 shown in FIG. 2, Shitabui2, it depends on Shitabui3. Since the voltage phase θv advances as the AC motor 1 is driven, correction is required when the coefficients Δk r1 , Δk r2 , Δk a1 , and Δk a2 are recalculated using (Equation 13) and (Equation 14) for each current estimation. A large calculation load is applied to the device 6. Further, when the coefficients Δk r1 , Δk r2 , Δk a1 , Δk a2 are stored in advance, the corrector 6 must have a large capacity memory storing all the coefficients.
そこで、本実施例は、(数15)の係数Δkr1,Δkr2,Δka1,Δka2を統一し、補正器6の演算負荷あるいは記憶容量を低減させるものである。 Therefore, in this embodiment, the coefficients Δk r1 , Δk r2 , Δk a1 , and Δka 2 of (Equation 15) are unified to reduce the calculation load or the storage capacity of the corrector 6.
係数Δkr1,Δkr2,Δka1,Δka2を統一するには、電流検出値が周期性を有し、これに同期して電流推定すればよい。電圧位相θvに応じて検出する電流相を適切に切り換えれば、電流検出値は周期性を有する。 In order to unify the coefficients Δk r1 , Δk r2 , Δka 1 , and Δka 2 , the current detection value has periodicity, and the current may be estimated in synchronization with this. If the current phase to be detected is appropriately switched according to the voltage phase θv, the current detection value has periodicity.
表1に示すように電流検出相を切り換える場合について説明する。ここで、−Uは、U相電流検出値Iu′の正負を逆にした−Iu′を検出することを示す。V相およびW相に関しても同様である。なお、表1は本実施例の一態様であり、電流検出値が周期性を有し、これに同期して電流推定できる場合は、表1に示す例に代えて、種々の区間,電圧位相,検出相の組み合わせを用いることができる。 A case where the current detection phase is switched as shown in Table 1 will be described. Here, −U indicates that −Iu ′ is detected by reversing the positive / negative of the U-phase current detection value Iu ′. The same applies to the V phase and the W phase. Table 1 shows one aspect of the present embodiment. When the current detection value has periodicity and current estimation can be performed in synchronization therewith, instead of the example shown in Table 1, various sections and voltage phases are used. , A combination of detection phases can be used.
このときの電流検出値を図7に実線で示す。電流検出値は周期性を有し、例えば、区間DにおいてU相を検出することは、区間Cにおいて−V相を検出することと等しい。他の区間においても同様であり、係数Δkr1,Δkr2,Δka1,Δka2は、いずれか一つの区間に基づいて演算すればよい。 The current detection value at this time is shown by a solid line in FIG. The detected current value has periodicity. For example, detecting the U phase in the section D is equivalent to detecting the −V phase in the section C. The same applies to the other sections, and the coefficients Δk r1 , Δk r2 , Δk a1 , and Δka 2 may be calculated based on any one of the sections.
以上で説明した本実施例では、係数Δkr1,Δkr2,Δka1,Δka2を統一できるため、電流推定部7cの演算負荷が小さくなる効果も得られる。 In the present embodiment described above, the coefficients Δk r1 , Δk r2 , Δk a1 , and Δk a2 can be unified, so that the calculation load of the current estimation unit 7 c can be reduced.
図8を用いて実施例4を説明する。なお、既に説明した実施例と同等の点については説明を省略することとする。 Example 4 will be described with reference to FIG. It should be noted that description of points that are the same as in the previously described embodiments will be omitted.
一般に、時定数Tfが大きいほど、直線母線電流IDCに対するフィルタ値IDC′の応答は低下する。このとき、実際の電流に対する電流推定値の応答が低下し、交流モータ1の制御も劣化することになる。 In general, the greater the time constant Tf, the lower the response of the filter value IDC ′ to the linear bus current IDC. At this time, the response of the estimated current value to the actual current is reduced, and the control of the AC motor 1 is also deteriorated.
そこで、本実施例では、フィルタ値IDC′が直線母線電流IDCを十分な応答速度で追従するように、(数3)の時定数Tfをキャリア周期Tcとの関係で規定することにより、制御応答性の低下を防止する。 Therefore, in the present embodiment, the control response is obtained by defining the time constant Tf of (Equation 3) in relation to the carrier cycle Tc so that the filter value IDC ′ follows the linear bus current IDC at a sufficient response speed. To prevent deterioration of sex.
図8に直流母線電流IDCおよびIDC′の波形図を示す。フィルタ値IDC′はキャリア周期Tc毎に1回検出されると仮定する。ここで、IDC0は直流母線電流IDCの定格値を表し、直流母線電流IDCは定格値IDC0以下とする。また、ある通電期間でのフィルタ値IDC′の検出値をIDC1、次の通電期間直前のフィルタ値IDC′をIDC2とする。IDCmは直流母線電流検出器4の最小分解能を表し、フィルタ値IDC′がIDCm以下の場合、ゼロが検出されるとする。 FIG. 8 shows waveform diagrams of the DC bus currents IDC and IDC ′. It is assumed that the filter value IDC ′ is detected once every carrier period Tc. Here, IDC0 represents the rated value of the DC bus current IDC, and the DC bus current IDC is not more than the rated value IDC0. Further, the detection value of the filter value IDC ′ in a certain energization period is IDC1, and the filter value IDC ′ immediately before the next energization period is IDC2. IDCm represents the minimum resolution of the DC bus current detector 4, and when the filter value IDC 'is equal to or less than IDCm, zero is detected.
このとき、キャリア周期Tc以内にフィルタ値IDC′が定格値IDC0から最小分解能IDCm以下まで変動できれば、ローパスフィルタ5の遅れは無視できる。この条件は、(数16)で表される。 At this time, if the filter value IDC ′ can vary from the rated value IDC0 to the minimum resolution IDCm or less within the carrier period Tc, the delay of the low-pass filter 5 can be ignored. This condition is expressed by (Expression 16).
ここで、IDC1とIDC2に関して、(数17)が成り立つ。 Here, with respect to IDC1 and IDC2, (Equation 17) holds.
(数16),(数17)より(数18)を得る。 (Equation 18) is obtained from (Equation 16) and (Equation 17).
本実施例では(数18)を満たす時定数Tfを用いることで、十分な応答速度のフィルタ値IDC′を得ることができ、このフィルタ値IDC′に基づいて交流モータ1を制御するので、交流モータ1の制御応答性の低下を防止することができる。 In this embodiment, by using the time constant Tf that satisfies (Equation 18), a filter value IDC ′ having a sufficient response speed can be obtained, and the AC motor 1 is controlled based on this filter value IDC ′. A decrease in control response of the motor 1 can be prevented.
図9,図10を用いて実施例5を説明する。なお、既に説明した実施例と同等の点については説明を省略することとする。 Embodiment 5 will be described with reference to FIGS. It should be noted that description of points that are the same as in the previously described embodiments will be omitted.
本実施例では、インバータ2の変調率に応じて時定数Tfを調整することにより、電流推定の精度を向上させる。 In this embodiment, the accuracy of current estimation is improved by adjusting the time constant Tf according to the modulation rate of the inverter 2.
図9は、実施例5で用いられるローパスフィルタ5の構成図である。ここに示すように、ローパスフィルタ5は複数のローパスフィルタ5a,5b,5cを内蔵しており、マルチプレクサ5dによって、使用するローパスフィルタを切り換えることができる。ローパスフィルタ5aは、所定の抵抗とコンデンサを組み合わせ所定の時定数のフィルタを構成している。他のローパスフィルタ5b,5cも同様の構成であるが、各々異なる時定数のフィルタを構成するよう、抵抗,コンデンサが選択される。ここでは、ローパスフィルタ5a,5b,5cの順に時定数が小さくなるものとする。 FIG. 9 is a configuration diagram of the low-pass filter 5 used in the fifth embodiment. As shown here, the low-pass filter 5 includes a plurality of low-pass filters 5a, 5b, and 5c, and the low-pass filter to be used can be switched by the multiplexer 5d. The low-pass filter 5a constitutes a filter having a predetermined time constant by combining a predetermined resistor and a capacitor. The other low-pass filters 5b and 5c have the same configuration, but resistors and capacitors are selected so as to form filters with different time constants. Here, it is assumed that the time constant decreases in the order of the low-pass filters 5a, 5b, and 5c.
図4で説明した期間Tuは、変調率が高いほど長くなり、ローパスフィルタ5により直流母線電流IDCを平滑化する必要がなくなる。そこで、図10に示すように、変調率が高いほど時定数Tfを小さくするようにマルチプレクサ5dを切り換える。これにより、変調率が最も高いときにはローパスフィルタ5aを用い、変調率が最も小さいときにはローパスフィルタ5cを用いるなど、変調率に応じた適切なフィルタを用いることで、電流推定の精度を向上させることができる。 The period Tu described in FIG. 4 becomes longer as the modulation rate is higher, and it becomes unnecessary to smooth the DC bus current IDC by the low-pass filter 5. Therefore, as shown in FIG. 10, the multiplexer 5d is switched so that the time constant Tf decreases as the modulation rate increases. Thus, the accuracy of current estimation can be improved by using an appropriate filter corresponding to the modulation rate, such as using the low-pass filter 5a when the modulation rate is the highest and using the low-pass filter 5c when the modulation rate is the lowest. it can.
なお、ここでは常にローパスフィルタを用いる構成としたが、変調率が十分に高く、閾値を超える場合には、ローパスフィルタ5および補正器6を回避し、直接直流母線電流を観測してもよい。また、本実施例は、図9以外の構成を用い、変調率に応じて時定数Tfを変更しても良い。例えば、ローパスフィルタ5のコンデンサと並列にショート回路を設けて、放電量を制御することにより、変調率に応じて連続的に時定数Tfを変更してもよい。 Although the low-pass filter is always used here, when the modulation factor is sufficiently high and exceeds the threshold, the low-pass filter 5 and the corrector 6 may be avoided and the direct current bus current may be observed directly. In the present embodiment, a configuration other than that shown in FIG. 9 may be used, and the time constant Tf may be changed according to the modulation rate. For example, the time constant Tf may be continuously changed according to the modulation rate by providing a short circuit in parallel with the capacitor of the low-pass filter 5 and controlling the discharge amount.
図11を用いて実施例6を説明する。実施例6は、実施例1から実施例5の説明した何れかの制御装置を冷凍空調機の駆動装置に適用した実施例である。 Example 6 will be described with reference to FIG. The sixth embodiment is an embodiment in which any one of the control devices described in the first to fifth embodiments is applied to a driving device for a refrigeration air conditioner.
これらの駆動装置は、しばしば、低速になるほど負荷が小さくなる運転条件で駆動される。低速になるほど交流モータ1の誘起電圧は小さくなり、また、負荷が小さくなるほどモータ電流I1も小さくなる。このため、極低速での変調率は極めて小さい。そこで、ファン駆動装置あるいは圧縮機駆動装置などの、冷凍空調装置の駆動装置へ実施例1から実施例5の何れかで説明した制御装置を適用することにより、駆動範囲をワイドレンジ化することができる。本発明を該当駆動装置に適用することにより、変調率が極めて小さい場合にも電流検出が可能となり、極低速から駆動装置を制御することが可能となる。 These drive devices are often driven under operating conditions where the load decreases as the speed decreases. The induced voltage of AC motor 1 decreases as the speed decreases, and the motor current I1 decreases as the load decreases. For this reason, the modulation rate at extremely low speed is extremely small. Therefore, the drive range can be widened by applying the control device described in any one of the first to fifth embodiments to the drive device of the refrigeration air conditioner such as a fan drive device or a compressor drive device. it can. By applying the present invention to the corresponding driving device, it is possible to detect current even when the modulation rate is extremely small, and it is possible to control the driving device from an extremely low speed.
1 交流モータ
2 インバータ
3 直流電源
4 直流母線電流検出器
5 ローパスフィルタ
5d マルチプレクサ
6 補正器
7 インバータ制御回路
7a PWM信号発生部
7b ベクトル制御部
7c 電流推定部
VDC 直流電圧
IDC 直流母線電流
Sup,Sun,Svp,Svn,Swp,Swn スイッチ素子
Vu,Vv,Vw U相電圧,V相電圧,W相電圧
Vu*,Vv*,Vw* U相電圧指令,V相電圧指令,W相電圧指令
V1 モータ電圧
I1 モータ電流
Ir 無効電流
Ia 有効電流
θv 電圧位相
ψ 電圧/電流位相差
DESCRIPTION OF SYMBOLS 1 AC motor 2 Inverter 3 DC power supply 4 DC bus current detector 5 Low pass filter 5d Multiplexer 6 Corrector 7 Inverter control circuit 7a PWM signal generation unit 7b Vector control unit 7c Current estimation unit VDC DC voltage IDC DC bus current Sup, Sun, Svp, Svn, Swp, Swn Switch elements Vu, Vv, Vw U phase voltage, V phase voltage, W phase voltage Vu * , Vv * , Vw * U phase voltage command, V phase voltage command, W phase voltage command V1 Motor voltage I1 Motor current Ir Reactive current Ia Effective current θv Voltage phase ψ Voltage / current phase difference
Claims (7)
該直流電源より供給される直流電力を交流電力に変換するインバータと、
該インバータに備わるスイッチ素子を制御するインバータ制御回路と、
前記インバータに流れる直流母線電流を検出する直流母線電流検出器と、
該直流母線電流検出器で検出した直流母線電流を平滑化するローパスフィルタと、
該ローパスフィルタで平滑された直流母線電流の減衰分を補正する補正器と
を備えることを特徴とする交流モータの制御装置。 DC power supply,
An inverter that converts DC power supplied from the DC power source into AC power;
An inverter control circuit for controlling a switch element provided in the inverter;
A DC bus current detector for detecting a DC bus current flowing in the inverter;
A low-pass filter for smoothing the DC bus current detected by the DC bus current detector;
A controller for an AC motor, comprising: a corrector that corrects the attenuation of the DC bus current smoothed by the low-pass filter.
前記インバータ制御回路は、
前記直流母線電流検出器で検出した複数の直流母線電流から前記交流モータに流れる交流電流の振幅および位相を推定する電流推定手段と、
該電流推定手段の出力に基づいて前記インバータ制御回路の電圧指令を演算するベクトル制御手段と、
該ベクトル制御手段の出力に基づいて前記スイッチ素子をPWM制御するPWM信号発生手段と、
を備えることを特徴とする交流モータの制御装置。 The control device according to claim 1,
The inverter control circuit is
Current estimation means for estimating the amplitude and phase of an alternating current flowing through the alternating current motor from a plurality of direct current bus currents detected by the direct current bus current detector;
Vector control means for calculating a voltage command of the inverter control circuit based on the output of the current estimation means;
PWM signal generating means for PWM controlling the switch element based on the output of the vector control means;
An AC motor control apparatus comprising:
前記電流推定手段は、前記直流母線電流検出器で検出される直流母線電流の周期性に同期して前記交流モータに流れる交流電流の振幅および位相を推定することを特徴とする交流モータの制御装置。 The control device according to claim 2.
The control apparatus for an AC motor, wherein the current estimation means estimates the amplitude and phase of an AC current flowing through the AC motor in synchronization with the periodicity of the DC bus current detected by the DC bus current detector .
前記ローパスフィルタの時定数Tfが
ただし、Tf:時定数、Tc:インバータ制御回路のキャリア周期
IDC0:直線母線電流の定格値、IDCm直流母線電流検出器の最小分解能 The control device according to claim 2.
The time constant Tf of the low-pass filter is
Where Tf: time constant, Tc: carrier cycle of the inverter control circuit
IDC0: Rated value of linear bus current, minimum resolution of IDCm DC bus current detector
前記インバータの変調率に応じて前記ローパスフィルタの時定数を変更することを特徴とする交流モータの制御装置。 The control device according to claim 2.
An AC motor control device, wherein a time constant of the low-pass filter is changed in accordance with a modulation factor of the inverter.
前記直流母線電流検出器は、前記インバータの変調率が閾値以上の場合、前記ローパスフィルタおよび前記補正器を回避し、前記インバータに流れる直線母線電流を直接検出することを特徴とする交流モータの制御装置。 The control device according to claim 2.
The DC bus current detector detects the linear bus current flowing in the inverter directly, avoiding the low-pass filter and the corrector when the modulation factor of the inverter is greater than or equal to a threshold value. apparatus.
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JP2011133796A JP5492826B2 (en) | 2011-06-16 | 2011-06-16 | AC motor control device and refrigeration air conditioner using the same |
KR1020110088979A KR101272450B1 (en) | 2011-06-16 | 2011-09-02 | Control device of ac motor and refrigerating air conditioner using the same |
CN201110260124.XA CN102832875B (en) | 2011-06-16 | 2011-09-05 | Control apparatus of AC motor and refrigerating and air conditioning apparatus using same |
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JP2014190843A (en) * | 2013-03-27 | 2014-10-06 | Ngk Spark Plug Co Ltd | Sensor control device and gas detection system |
WO2015087437A1 (en) * | 2013-12-13 | 2015-06-18 | 株式会社日立産機システム | Power conversion device |
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JP6317904B2 (en) * | 2013-10-02 | 2018-04-25 | 日立ジョンソンコントロールズ空調株式会社 | Motor control device and air conditioner |
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CN104467505B (en) * | 2014-11-28 | 2017-12-29 | 佛山市众盈电子有限公司 | One kind adjusts wide inverter circuit |
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KR20120139517A (en) | 2012-12-27 |
KR101272450B1 (en) | 2013-06-07 |
CN102832875A (en) | 2012-12-19 |
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