JP2011010486A - Control device for permanent-magnet synchronous machine - Google Patents

Control device for permanent-magnet synchronous machine Download PDF

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JP2011010486A
JP2011010486A JP2009152470A JP2009152470A JP2011010486A JP 2011010486 A JP2011010486 A JP 2011010486A JP 2009152470 A JP2009152470 A JP 2009152470A JP 2009152470 A JP2009152470 A JP 2009152470A JP 2011010486 A JP2011010486 A JP 2011010486A
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synchronous machine
magnetic flux
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Yasuyuki Noto
泰之 能登
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Fuji Electric Co Ltd
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Fuji Electric Systems Co Ltd
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Abstract

PROBLEM TO BE SOLVED: To estimate the magnet magnetic flux of a permanent-magnet synchronous machine, without being affected by a calculation error of the magnetic pole position.SOLUTION: A permanent-magnet synchronous machine includes an observer that supplies a voltage of a variable voltage and frequency to a motor by a power conversion device so as to detect and estimate a motor current. The permanent-magnet synchronous machine estimates the rotational speed and the magnetic pole position of the motor, by using the detected motor current detection value, at a position sensorless control method for the permanent magnet synchronous machine, the magnet-magnetic flux is estimated, on the basis of an inverse matrix of a transfer function matrix in a range from an error of the magnetic pole position and an error of the magnet magnetic flux which are obtained from a deviation between the motor current detection value and a current estimated value estimated by the observer, up to a deviation between the current detection value and the current estimated value.

Description

本発明は、永久磁石同期機の位置センサレス制御時の磁石磁束オンラインチューニングに関する。   The present invention relates to on-line magnet magnetic flux tuning during position sensorless control of a permanent magnet synchronous machine.

永久磁石同期機位置センサレス制御において、温度変化により変動する磁石磁束の設定誤差によりトルク制御精度が悪化する。そのため、磁石磁束のオンラインチューニング技術が提案されている。温度センサを用いない、永久磁石同期機位置センサレス制御時の磁石磁束オンラインチューニングの技術として、特許文献１に記載の技術がある。   In the permanent magnet synchronous machine position sensorless control, the torque control accuracy deteriorates due to the setting error of the magnet magnetic flux that fluctuates due to the temperature change. Therefore, on-line tuning technology for magnet magnetic flux has been proposed. As a technique of magnet magnetic flux online tuning at the time of permanent magnet synchronous machine position sensorless control without using a temperature sensor, there is a technique described in Patent Document 1.

特許文献１に記載の技術を用いて構成した位置センサレス制御磁石磁束オンラインチューニングシステムのシステム構成を図２に示す。
このシステムは、永久磁石同期機（以下PMSM）１と、 負荷２と、電流検出手段３と、 電力変換装置４と、回転二相／三相座標変換手段５と、三相／回転二相座標変換手段６と、電流制御手段７と、電流指令値演算手段８と、速度PID 調節器９と、速度推定器１０と、モータ軸推定器１１と、逆起電圧定数同定器１２と、積分手段１３と、ローパスフィルタ１４と、を備える。
三相／回転二相座標変換手段６は、電流検出手段３が出力する三相の電流値ｉua，ｉva，ｉwaをα，β軸の固定二相変換した後、積分手段１３が出力している角度の回転座標変換を行い、二軸の量であるγ，δ軸電流ｉγa，ｉδaに変換する。
なお、磁石の磁極に平行な方向と推定している軸をγ軸, それに直交する方向をδ軸としている。
また、回転二相／三相座標変換手段５では、積分手段１３が出力している角度θ#を用いて、γ，δ軸電圧指令値ｖγa *，ｖδa *を逆回転変換し、固定二相の値に変換した後、二相三相変換を行って三相電圧指令値ｖua *，ｖva *，ｖwa *を出力する。この電圧指令値ｖua *，ｖva *，ｖwa *は電力変換装置４に入力される。電力変換装置4 は電圧指令値ｖua *，ｖva *，ｖwa *に基づいて、可変周波数・可変電圧の電圧電流をモータに供給する電源装置である。
速度PID 調節器９は、速度指令値(電気角)ω*と速度推定器１０が出力するモータの推定速度(電気角)ω#の推定値の偏差がゼロになるよう、偏差にPID 演算を行った値をトルク指令値Ｔe *として出力する。
電流指令値演算手段８は、速度PID 調節器９から受けたトルク指令値に基づき、γ，δ軸電流指令値ｉγa *，ｉδa *を作成する。
電流制御手段７は、電流指令値演算手段８から受けたγ，δ軸電流指令値ｉγa *，ｉδa *と、三相／回転二相座標変換手段６から受けたγ，δ軸電流ｉγa，ｉδaがそれぞれ一致するようにγ，δ軸電圧指令値ｖγa *，ｖδa *を制御する。
速度推定器１０では、速度指令値(電気角) ω*とδ 軸電流ｉδaとから速度を推定し推定速度(電気角) ω#を得る。回転二相／三相座標変換手段５および三相／回転二相座標変換手段６にて使用する座標変換角度θ#は，推定速度(電気角) ω#を積分手段１３により積分することによって得る。 FIG. 2 shows a system configuration of a position sensorless control magnet magnetic flux online tuning system configured using the technique described in Patent Document 1.
This system includes a permanent magnet synchronous machine (hereinafter referred to as PMSM) 1, a load 2, a current detection means 3, a power conversion device 4, a rotating two-phase / three-phase coordinate converting means 5, and a three-phase / rotating two-phase coordinate. Conversion means 6, current control means 7, current command value calculation means 8, speed PID adjuster 9, speed estimator 10, motor shaft estimator 11, counter electromotive voltage constant identifier 12, and integration means 13 and a low-pass filter 14.
The three-phase / rotational two-phase coordinate conversion means 6 performs the fixed two-phase conversion of the three-phase current values i ua , i va , i wa output from the current detection means 3 on the α and β axes, and then the output from the integration means 13. Rotational coordinate conversion of the angle is performed and converted into γ and δ axis currents iγ a and iδ a which are biaxial quantities.
The axis estimated to be parallel to the magnetic pole of the magnet is the γ-axis, and the direction orthogonal to it is the δ-axis.
Further, the rotating two-phase / three-phase coordinate converting means 5 uses the angle θ # output from the integrating means 13 to reversely convert the γ and δ-axis voltage command values vγ a * and v δ a * to be fixed. After conversion to a two-phase value, two-phase three-phase conversion is performed to output three-phase voltage command values v ua * , v va * , and v wa * . The voltage command values v ua * , v va * and v wa * are input to the power converter 4. The power conversion device 4 is a power supply device that supplies a voltage current of variable frequency / variable voltage to the motor based on the voltage command values v ua * , v va * , v wa * .
The speed PID controller 9 performs PID calculation on the deviation so that the deviation between the speed command value (electrical angle) ω * and the estimated value of the estimated motor speed (electrical angle) ω # output from the speed estimator 10 becomes zero. The performed value is output as a torque command value Te * .
Current command value calculating unit 8, based on the torque command value received from the speed PID controller 9, gamma, [delta] -axis current value i? A *, created a i? A *.
Current control means 7, gamma received from the current command value computing means 8, [delta] -axis current value iγ a *, iδ a * and, gamma received from the three-phase / rotation two-phase coordinate conversion unit 6, [delta] -axis current i? a, gamma as i? a match respectively, [delta] -axis voltage value v? a *, and controls the v? a *.
The speed estimator 10, the speed command value (electric angle) omega * and δ-axis current i? A and the estimated speed from the estimated speed (electrical angle) to obtain the omega #. The coordinate conversion angle θ # used in the rotating two-phase / three-phase coordinate converting means 5 and the three-phase / rotating two-phase coordinate converting means 6 is obtained by integrating the estimated speed (electrical angle) ω # by the integrating means 13. .

推定座標であるγδ座標の回転角速度はω# (電気角)、γ軸はu相軸からθ#(電気角)の位相角上にあり、磁石の磁極に平行な軸（d軸）よりΔθ (電気角)だけ遅れているものとする。γδ座標でのPMSMの電圧方程式は（数１）のように表される。 The rotational angular velocity of the estimated γδ coordinate is ω # (electrical angle), the γ axis is on the phase angle of θ # (electrical angle) from the u phase axis, and Δθ from the axis parallel to the magnetic pole of the magnet (d axis) It is assumed that it is delayed by (electrical angle). The voltage equation of PMSM in γδ coordinates is expressed as (Equation 1).

ここで、 here,

Ｒa:電機子巻線抵抗，Ｌd，Ｌq: d,q軸インダクタンス，φf:磁石磁束，
ｖγa，ｖδa:γ,δ軸電圧，ｉγa，ｉδa:γ,δ軸電流，ω:回転角速度( 電気角) ，
Ｐ:微分演算子

（数１）において、 R a : Armature winding resistance, L d , L q : d, q axis inductance, φ f : Magnet flux,
vγ a, vδ a: γ, δ -axis voltage, iγ a, iδ a: γ , δ -axis current, omega: rotational angular velocity (electrical angle),
P: Differential operator

In (Equation 1),

とすると、PMSMの電圧方程式は（数５）のように表される。 Then, the voltage equation of PMSM is expressed as (Equation 5).

モータ軸推定器１１では、（数５）よりモータの軸誤差を（数６）で推定する。 The motor shaft estimator 11 estimates the shaft error of the motor from (Equation 5) using (Equation 6).

逆起電圧定数同定器１２では、（数６）で求めた（Δθ）#を用いて磁石磁束を（数７）で推定する。 In the counter electromotive voltage constant identifier 12, the magnetic flux is estimated by (Equation 7) using (Δθ) # obtained by (Equation 6).

ここで、 here,

なお、電圧検出器を使用し、（数６）及び（数７）中でγ,δ軸電圧指令値に変えてγ,δ軸電圧検出値を用い、推定精度の向上を図ることも可能である。
（数７）で求めたφ#をローパスフィルタ１４へ入力し、その出力φf#を電流制御手段７ と、電流指令値演算手段８に入力する。 It is also possible to improve the estimation accuracy by using a voltage detector and using γ, δ-axis voltage detection values instead of γ, δ-axis voltage command values in (Equation 6) and (Equation 7). is there.
Φ # obtained in (Equation 7) is input to the low-pass filter 14, and its output φ f # is input to the current control means 7 and the current command value calculation means 8.

特開２００４−７９２４号公報JP 2004-7924 A

しかしながら、この特許文献１に記載の永久磁石同期機位置センサレス制御時の磁石磁束オンラインチューニング技術では、加減速時等磁極位置演算誤差（Δθー（Δθ）#）が０でない場合に、磁石磁束推定誤差が生じるという問題がある。
そのため、磁石磁束推定誤差によりトルク制御精度が悪化する。 However, in the magnetic flux online tuning technique at the time of permanent magnet synchronous machine position sensorless control described in Patent Document 1, when the magnetic pole position calculation error (Δθ− (Δθ) # ) is not zero during acceleration / deceleration, the magnetic flux estimation is performed. There is a problem that an error occurs.
Therefore, torque control accuracy is deteriorated due to a magnet magnetic flux estimation error.

そこで、本発明は、磁極位置演算誤差の影響を受けることなく、永久磁石同期機の磁石磁束を推定することを目的としている。   Accordingly, an object of the present invention is to estimate the magnetic flux of a permanent magnet synchronous machine without being affected by a magnetic pole position calculation error.

上記目的を達成するために、請求項１に係る発明では、電力変換装置によりモータに可変電圧可変周波数の電圧を与え、モータ電流を検出し、モータ電流を推定するオブザーバとを備え、検出したモータ電流検出値を用いて、モータの回転速度および磁極位置を推定する永久磁石同期機の位置センサレス制御方法において、
モータ電流検出値と、オブザーバにて推定される電流推定値との偏差から得られる磁極位置誤差及び磁石磁束誤差から、電流検出値と電流推定値の偏差までの伝達関数行列の逆行列を用いて磁石磁束推定を行うことを特徴とする。 In order to achieve the above object, the invention according to claim 1 includes an observer that applies a variable voltage variable frequency voltage to the motor by the power conversion device, detects the motor current, and estimates the motor current. In the position sensorless control method of the permanent magnet synchronous machine that estimates the rotational speed and magnetic pole position of the motor using the current detection value,
Using the inverse matrix of the transfer function matrix from the magnetic pole position error and magnet magnetic flux error obtained from the deviation between the motor current detection value and the current estimation value estimated by the observer to the deviation between the current detection value and the current estimation value Magnet magnetic flux estimation is performed.

本発明の磁石磁束推定器では, 磁極位置演算誤差と独立に磁石磁束を推定することができる。従って磁極位置演算誤差がある場合でも磁石磁束を高精度に推定することができ、これにより高精度なトルク制御が可能である。   The magnet flux estimator of the present invention can estimate the magnet flux independently of the magnetic pole position calculation error. Therefore, even when there is a magnetic pole position calculation error, the magnetic flux of the magnet can be estimated with high accuracy, thereby enabling highly accurate torque control.

本発明による実施形態を示す構成図である。It is a block diagram which shows embodiment by this invention. 従来技術を示す構成図である。It is a block diagram which shows a prior art.

以下、本発明の実施の形態を図面に基づいて説明する。
図１に、この発明の第１の実施例を示す。
この図において、図２に示した従来例構成と同一機能を有するものには同一符号を付して、ここではその説明を省略する。
図１と図２に示した従来例と異なる点は、速度推定器１０と、モータ軸推定器１１と、 逆起電圧定数同定器１２と、積分手段１３と、ローパスフィルタ１４とに代え、電流オブザーバ２１と、磁石磁束推定器２２と、磁極位置・回転速度推定器２３とを用いていることである。 Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 shows a first embodiment of the present invention.
In this figure, components having the same functions as those in the conventional configuration shown in FIG. 2 are denoted by the same reference numerals, and description thereof is omitted here.
The difference from the conventional example shown in FIGS. 1 and 2 is that the speed estimator 10, the motor shaft estimator 11, the counter electromotive voltage constant identifier 12, the integrating means 13, and the low-pass filter 14 are replaced with a current. That is, an observer 21, a magnetic flux estimator 22, and a magnetic pole position / rotation speed estimator 23 are used.

γ,δ軸電流を推定する電流オブザーバ21 は、（数９）のものを用いる。   The current observer 21 for estimating the γ and δ axis currents is the one of (Equation 9).

なお、#付きの電流及びパラメータは推定値を表す。推定値を採用するパラメータはθ#，ω#，φf#である。また、ｇ11，ｇ12，ｇ21，ｇ22はオブザーバゲインである。なお、電圧検出器を使用し、（数９）中でγ,δ軸電圧指令値に変えてγ,δ軸電圧検出値を用い、推定精度の向上を図ることも可能である。
また、オブザーバゲインは（数１０）のように設定する。 The current and parameters with # represent estimated values. The parameters that employ the estimated values are θ # , ω # , and φf # . G 11 , g 12 , g 21 , and g 22 are observer gains. It is also possible to improve the estimation accuracy by using a voltage detector and using the detected γ and δ-axis voltage values instead of the γ and δ-axis voltage command values in (Equation 9).
Further, the observer gain is set as shown in (Equation 10).

ここで，ｇa，ｇb，ｇc，ｇdはオブザーバゲインの極を決める制御変数である。
電流オブザーバ21 の特性方程式は（数１１）のようになるため、 Here, g a , g b , g c , and g d are control variables that determine the poles of the observer gain.
Since the characteristic equation of the current observer 21 is as shown in (Equation 11),

オブザーバゲイン中の制御変数ｇa，ｇb，ｇc，ｇdは（数１１）が安定条件を満たすように設定すればよい。なお、右下付き添字0は平衡点を表す。
次に、磁石磁束推定器２２の動作を説明する。磁石磁束推定器２２ は（数１２）を演算する。 Control variables g a , g b , g c , and g d in the observer gain may be set so that (Equation 11) satisfies the stability condition. The subscript 0 at the lower right indicates the equilibrium point.
Next, the operation of the magnet magnetic flux estimator 22 will be described. The magnet flux estimator 22 calculates (Equation 12).

ここで、Ｋφ，ｘ1，ｘ0は推定器ゲインであり、 Where Kφ, x 1 , x 0 are estimator gains,

である。
（数１）を変形すると、（数１４）が得られる。 It is.
By transforming (Equation 1), (Equation 14) is obtained.

（数９）、（数１０）、（数１４）より、
ｃｏｓΔθ＝ｃｏｓ２Δθ＝１、ｓｉｎΔθ＝Δθ、ｓｉｎ２Δθ＝２Δθ
とそれぞれ近似し、平衡点近傍で線形化すると次の線形化誤差状態方程式（数１５）が得られる。ここで，Δω＝ｓΔθと置いて整理している。また，磁極位置，回転速度，磁石磁束以外のパラメータの誤差は無視している。また、ｖγa *−ｖγa＝０，ｖδa *ーｖδa＝０としている。なお，sはラプラス演算子である。 From (Equation 9), (Equation 10), and (Equation 14),
cosΔθ = cos2Δθ = 1, sinΔθ = Δθ, sin2Δθ = 2Δθ
And linearizing near the equilibrium point, the following linearization error state equation (Equation 15) is obtained. Here, Δω = sΔθ is put in order. In addition, errors in parameters other than the magnetic pole position, rotational speed, and magnet magnetic flux are ignored. Further, vγ a * −vγ a = 0 and vδ a * −vδ a = 0. Note that s is a Laplace operator.

ここで、 here,

（数１５）より、uを入力，ｅiaを出力とする線形化誤差伝達関数行列Ｐ0（ｓ）は次式（数１７）となる。 From (Equation 15), the linearization error transfer function matrix P 0 (s) with u as input and e ia as output is expressed by the following equation (Equation 17).

このとき、（数１２）及び（数１７）より、 At this time, from (Equation 12) and (Equation 17),

ここで、 here,

が得られる。（数１２）の推定器において、線形化誤差伝達関数行列の逆行列の分子を用いているため、（数１８）右辺の1行2列行列の1行1列目が0となっている。従って、φf#は磁極位置演算誤差Δθの影響を受けずに推定することができる。推定器ゲインＫφ，ｘ1，ｘ0は，（数１８）より磁石磁束推定系の応答に応じて設定する。
磁極位置・回転速度推定器２３ は、例えば以下の推定器を用いる。
磁極位置，回転速度の推定器を次式（数２０）、（数２１）とする。 Is obtained. Since the numerator of the inverse matrix of the linearization error transfer function matrix is used in the estimator of (Equation 12), the 1st row and 1st column of the 1 × 2 matrix on the right side is (0). Therefore, φ f # can be estimated without being affected by the magnetic pole position calculation error Δθ. The estimator gains Kφ, x 1 and x 0 are set according to the response of the magnetic flux estimation system from (Equation 18).
The magnetic pole position / rotation speed estimator 23 uses, for example, the following estimator.
The magnetic pole position and rotational speed estimators are represented by the following equations (Equation 20) and (Equation 21).

ここで、Ｋθ1，Ｋθ2は推定器ゲインである。
オブザーバゲイン中の制御変数ｇb，ｇcを、 Here, Kθ 1 and Kθ 2 are estimator gains.
Control variables g b and g c in the observer gain

のように選ぶことで、θ#とΔθ，Δφfの関係は，（数１２）及び（数２０）より、（数２３）となる。 By so choosing it as, theta # and [Delta] [theta], the relationship of [Delta] [phi f, from (Equation 12) and (Expression 20), the equation (23).

ここで、ＰD0＝ｓ2＋（ｇa0＋ｇb0）ｓ＋（ｇa0ｇb0ーｇc0ｇd0）となる。（数２３）より、磁極位置の推定は、磁石磁束誤差の影響を受けるが、磁石磁束推定が磁極位置誤差、回転速度誤差に独立に行えるため、磁石磁束推定が収束している状態では、磁石磁束誤差の影響を無視することができる。
また、このときω#とΔω，Δφfの関係は，（数１２）、（数２１）より（数２４）のようになる。 Here, P D0 = s 2 + (g a0 + g b0 ) s + (g a0 g b0 −g c0 g d0 ). From (Equation 23), the estimation of the magnetic pole position is affected by the magnet magnetic flux error. However, since the magnet magnetic flux estimation can be performed independently of the magnetic pole position error and the rotation speed error, The influence of magnetic flux error can be ignored.
The relationship between the time omega # and [Delta] [omega, [Delta] [phi f is as (Expression 12), (Expression 21) from equation (24).

（数２４）より、回転速度の推定は，磁石磁束誤差の影響を受けるが、磁石磁束推定が磁極位置誤差、回転速度誤差に独立に行えるため、磁石磁束推定が収束している状態では、磁石磁束誤差の影響を無視することができる。
推定器ゲインＫθ1，Ｋθ2は，（数２３）、（数２４）より磁極位置・回転速度推定系の応答に応じて設定する。
なお、磁極位置及び回転速度の推定は、他の方式を用いてもよい。
また、この位置センサレス制御時の磁石磁束オンラインチューニング技術は、Ｌd≠Ｌqの埋め込み磁石形永久磁石同期機及びＬd＝Ｌqの表面磁石形永久磁石同期機の双方に適用可能である。 From (Equation 24), the estimation of the rotational speed is affected by the magnet magnetic flux error. However, since the magnet magnetic flux estimation can be performed independently of the magnetic pole position error and the rotational speed error, the magnet magnetic flux estimation is converged. The influence of magnetic flux error can be ignored.
The estimator gains Kθ 1 and Kθ 2 are set according to the response of the magnetic pole position / rotation speed estimation system from (Equation 23) and (Equation 24).
In addition, you may use another system for estimation of a magnetic pole position and a rotational speed.
Further, the magnet magnetic flux online tuning technology at the time of position sensorless control can be applied to both an embedded magnet type permanent magnet synchronous machine with L d ≠ L q and a surface magnet type permanent magnet synchronous machine with L d = L q .

１…直流電源、２…電力変換装置、３…正側給電回路、４…負側給電回路、５…基準電圧設定回路、６…電圧比較回路、７…間欠信号発振回路、８…Ｅ／Ｏ変換回路、９…光ケーブル、１０…Ｏ／Ｅ変換回路、１１…制御回路、ＧＲ１…第１の抵抗素子群、ＧＲ２…第２の抵抗素子群、ＧＲ３…第３の抵抗素子群、Ｒ１Ｐ，Ｒ２Ｐ，Ｒ１Ｎ，Ｒ２Ｎ…分圧抵抗、Ｇ１Ｇ，Ｒ２Ｇ…接地抵抗   DESCRIPTION OF SYMBOLS 1 ... DC power supply, 2 ... Power converter, 3 ... Positive side feeding circuit, 4 ... Negative side feeding circuit, 5 ... Reference voltage setting circuit, 6 ... Voltage comparison circuit, 7 ... Intermittent signal oscillation circuit, 8 ... E / O Conversion circuit, 9 ... optical cable, 10 ... O / E conversion circuit, 11 ... control circuit, GR1 ... first resistance element group, GR2 ... second resistance element group, GR3 ... third resistance element group, R1P, R2P , R1N, R2N ... voltage dividing resistors, G1G, R2G ... grounding resistors

Claims (1)

電力変換装置によりモータに可変電圧可変周波数の電圧を与え、
モータ電流を検出し、
モータ電流を推定するオブザーバとを備え、
検出したモータ電流検出値を用いて、モータの回転速度および磁極位置を推定する永久磁石同期機の位置センサレス制御方法において、
モータ電流検出値と、オブザーバにて推定される電流推定値との偏差から得られる磁極位置誤差及び磁石磁束誤差から、電流検出値と電流推定値の偏差までの伝達関数行列の逆行列に基づいて磁石磁束推定を行うことを特徴とする永久磁石同期機の制御方法。 A voltage of variable voltage and variable frequency is given to the motor by the power converter,
Detect motor current,
An observer for estimating the motor current,
In the position sensorless control method of the permanent magnet synchronous machine for estimating the rotation speed and the magnetic pole position of the motor using the detected motor current detection value,
Based on the inverse matrix of the transfer function matrix from the magnetic pole position error and the magnetic flux error obtained from the deviation between the motor current detection value and the current estimation value estimated by the observer to the deviation between the current detection value and the current estimation value A method of controlling a permanent magnet synchronous machine, wherein magnet magnetic flux estimation is performed.
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Publication number Priority date Publication date Assignee Title
CN112217428A (en) * 2020-09-18 2021-01-12 江苏科技大学 Underwater robot propulsion system position-sensor-free control method
CN113328664A (en) * 2020-02-28 2021-08-31 南京理工大学 Strong tracking UKF asynchronous motor rotating speed identification method based on fading factor matrix

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JP2004135458A (en) * 2002-10-11 2004-04-30 Yaskawa Electric Corp Control method for ipm motor and control device
JP2009060688A (en) * 2007-08-30 2009-03-19 Fuji Electric Systems Co Ltd Controller for synchronous motors
JP2009095135A (en) * 2007-10-09 2009-04-30 Fuji Electric Systems Co Ltd Controller of synchronous electric motor

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JP2004135458A (en) * 2002-10-11 2004-04-30 Yaskawa Electric Corp Control method for ipm motor and control device
JP2009060688A (en) * 2007-08-30 2009-03-19 Fuji Electric Systems Co Ltd Controller for synchronous motors
JP2009095135A (en) * 2007-10-09 2009-04-30 Fuji Electric Systems Co Ltd Controller of synchronous electric motor

Cited By (2)

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Publication number Priority date Publication date Assignee Title
CN113328664A (en) * 2020-02-28 2021-08-31 南京理工大学 Strong tracking UKF asynchronous motor rotating speed identification method based on fading factor matrix
CN112217428A (en) * 2020-09-18 2021-01-12 江苏科技大学 Underwater robot propulsion system position-sensor-free control method

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