JP2006332769A - Ofdm receiver - Google Patents

Ofdm receiver Download PDF

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JP2006332769A
JP2006332769A JP2005149658A JP2005149658A JP2006332769A JP 2006332769 A JP2006332769 A JP 2006332769A JP 2005149658 A JP2005149658 A JP 2005149658A JP 2005149658 A JP2005149658 A JP 2005149658A JP 2006332769 A JP2006332769 A JP 2006332769A
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JP4115466B2 (en
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Yasuo Matsunami
靖雄 松波
Jun Ido
純 井戸
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Mitsubishi Electric Corp
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Abstract

<P>PROBLEM TO BE SOLVED: To provide an OFDM receiver for attaining improvement of a C/N level by applying weighting to amplitude components of each element by transmission path estimation and thereafter carrying out frequency diversity composition. <P>SOLUTION: The OFDM receiver respectively includes a plurality of systems of: time domain processing sections 3<SB>1</SB>, 3<SB>k</SB>for obtaining DFT (discrete Fourier transform) window positions received by antenna elements 1<SB>1</SB>, 1<SB>k</SB>by each symbol; DFT means 4<SB>1</SB>, 4<SB>k</SB>for applying DFT to the OFDM signals on the basis of the DFT window positions; SP demodulation means 5<SB>1</SB>, 5<SB>k</SB>for correcting outputs of the DFT means by using a result of the transmission path estimation by each carrier on the basis of SP signals and includes: a weight generating means 8 for generating weight coefficients α<SB>1i</SB>, α<SB>ki</SB>as to each system and each symbol on the basis of signals from the time domain processing sections; and a space diversity composite means 6 for carrying out space diversity composition on the basis of output signals f<SB>1i</SB>, f<SB>ki</SB>, g<SB>1i</SB>, g<SB>ki</SB>from the SP demodulation means and the weight coefficients α<SB>1i</SB>, α<SB>ki</SB>. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

本発明は、複数のアンテナ素子を用いるＯＦＤＭ（直交周波数分割多重）受信装置に関し、特に、空間ダイバーシチを利用することによってキャリア対雑音比（以下「Ｃ／Ｎ比」と言う。）を向上させるダイバーシチ受信技術に関するものである。   The present invention relates to an OFDM (orthogonal frequency division multiplexing) receiver using a plurality of antenna elements, and in particular, diversity that improves a carrier-to-noise ratio (hereinafter referred to as “C / N ratio”) by using spatial diversity. It relates to reception technology.

送信データを無線伝送する場合には、マルチパスやフェージングなどによって受信した信号の復号誤り率が増大し、伝送特性が劣化するという問題がある。このため、一般に、互いに相関のない複数のアンテナ素子を配置し、空間ダイバーシチ合成を行って信号対雑音比（以下「Ｓ／Ｎ比」と言う。）を最大にすることで、受信性能を向上させる対策が講じられている。   When transmitting transmission data by radio, there is a problem that the decoding error rate of a received signal increases due to multipath, fading, etc., and transmission characteristics deteriorate. For this reason, generally, a plurality of antenna elements having no correlation with each other are arranged, and spatial diversity combining is performed to maximize the signal-to-noise ratio (hereinafter referred to as “S / N ratio”), thereby improving the reception performance. Measures to take are taken.

例えば、従来のＯＦＤＭ受信装置においては、複数のアンテナ素子の受信信号のそれぞれに対して重み付けを行って合成し、合成信号のＳ／Ｎ比が最大となるようにしている。ＯＦＤＭ信号の空間ダイバーシチ合成には、離散フーリエ変換（以下「ＤＦＴ」と言う。）前の時間ドメインでダイバーシチ合成を行う方式（以下「Ｐｒｅ−ＤＦＴ型」と言う。）と、ＤＦＴ後の周波数ドメインでのキャリア毎にダイバーシチ合成を行う方式（以下「Ｐｏｓｔ−ＤＦＴ型」と言う。）があり、一般的に、Ｐｏｓｔ−ＤＦＴ型の方が誤り率特性に優れている（例えば、非特許文献１参照）。   For example, in a conventional OFDM receiving apparatus, each received signal of a plurality of antenna elements is weighted and combined so that the S / N ratio of the combined signal is maximized. For spatial diversity combining of OFDM signals, a method of performing diversity combining in the time domain before discrete Fourier transform (hereinafter referred to as “DFT”) (hereinafter referred to as “Pre-DFT type”) and a frequency domain after DFT. There is a method of performing diversity combining for each carrier (hereinafter referred to as “Post-DFT type”). Generally, the Post-DFT type is superior in error rate characteristics (for example, Non-Patent Document 1). reference).

図１１は、一般的なＰｏｓｔ−ＤＦＴ型空間ダイバーシチ合成を行うＯＦＤＭ受信装置の構成を概略的に示すブロック図である。このＯＦＤＭ受信装置は、複数のアンテナ素子１１１，１１ｋ（図には、複数のアンテナ素子１１１，…，１１ｋ，…の内の２つを示す。）と、チューナ部１１，１ｋ（図には、複数のチューナ部１１，…，１ｋ，…の内の２つを示す。）と、アナログデジタル（以下「ＡＤ」と言う。）変換手段２１，２ｋ（図には、複数のＡＤ変換手段２１，…，２ｋ，…の内の２つを示す。）と、時間ドメイン処理部３１，３ｋ（図には、複数の時間ドメイン処理部３１，…，３ｋ，…の内の２つを示す。）と、ＤＦＴ手段４１，４ｋ（図には、複数のＤＦＴ手段４１，…，４ｋ，…の内の２つを示す。）と、スキャッタードパイロット（以下「ＳＰ」と言う。）復調手段５１，５ｋ（図には、複数のＳＰ復調手段５１，…，５ｋ，…の内の２つを示す。）と、空間ダイバーシチ合成手段６ａと、誤り訂正手段７とを有する。ここで、添え字１及びｋは、各系統（アンテナ素子からＳＰ復調手段までの構成）を区別するためのものであり、それぞれ第１系統及び第ｋ系統を示し、ｋは、２以上で且つ任意の整数を表す。 FIG. 11 is a block diagram schematically illustrating a configuration of an OFDM receiver that performs general Post-DFT type spatial diversity combining. This OFDM receiving apparatus includes a plurality of antenna elements 11 1 , 11 k (in the figure, two of the plurality of antenna elements 11 1 ,..., 11 k ,...) And tuner sections 1 1 , 1. k (shown in the figure are two of a plurality of tuner units 1 1 ,..., 1 k ,...) and analog-digital (hereinafter referred to as “AD”) conversion means 2 1 , 2 k (FIG. Shows two of a plurality of AD conversion means 2 1 ,..., 2 k ,..., And time domain processing units 3 1 , 3 k (in the figure, a plurality of time domain processing units 3 1 ,..., 3 k ,..., And DFT means 4 1 , 4 k (in the figure, two of the plurality of DFT means 4 1 ,..., 4 k ,. ) And scattered pilot (hereinafter referred to as “SP”) demodulation means 5 1 , 5 k (in the figure, a plurality of SP demodulation means 5 1 ,..., 5 k ,... Are shown, and space diversity combining means 6a and error correcting means 7 are included. Here, the subscripts 1 and k are for distinguishing each system (configuration from the antenna element to the SP demodulating means), and indicate the first system and the k-th system, respectively, and k is 2 or more and Represents an arbitrary integer.

チューナ部１１，１ｋはそれぞれ、アンテナ素子１１１，１１ｋで受信された信号をＩＦ（中間周波数）信号に変換する。ＡＤ変換手段２１，２ｋはそれぞれ、チューナ部１１，１ｋからのＩＦ信号に対して離散値化の処理を行う。時間ドメイン処理部３１，３ｋはそれぞれ、ＡＤ変換手段２１，２ｋ出力を直交復調して実部データと虚数部データを生成し、ゲイン調整、クロック再生、キャリア再生を行う。ＤＦＴ手段４１，４ｋはそれぞれ、時間ドメイン処理部３１，３ｋからの時間ドメイン信号にＤＦＴ処理を施して周波数ドメイン信号を供給する。ＳＰ復調手段５１，５ｋはそれぞれ、ＤＦＴ手段４１，４ｋからの周波数ドメイン信号におけるＳＰ信号やコンティニュアスパイロット（以下「ＣＰ」と言う。）信号などのサブキャリアに基づいて伝送路推定を行い、キャリアの位相回転及び振幅を補正する。空間ダイバーシチ合成手段６ａは、ＳＰ復調手段５１，５ｋによるＳＰ復調結果のキャリア毎に各素子の出力信号のダイバーシチ合成を行う。誤り訂正手段７は、空間ダイバーシチ合成手段６ａによるダイバーシチ結果又はＳＰ復調結果に対してデインタリーバ、誤り訂正などを行い、トランスポートストリーム（以下「ＴＳ」と言う。）データとして出力する。なお、時間ドメイン処理部３１，３ｋにおけるクロック再生及びキャリア再生、ＤＦＴ手段４１，４ｋによる処理、及び誤り訂正手段７による処理には、既知の技術を用いることができる。 The tuner units 1 1 and 1 k convert the signals received by the antenna elements 11 1 and 11 k into IF (intermediate frequency) signals, respectively. The AD conversion units 2 1 and 2 k perform discrete value processing on the IF signals from the tuner units 1 1 and 1 k , respectively. The time domain processing units 3 1 and 3 k respectively perform quadrature demodulation of the outputs of the AD conversion units 2 1 and 2 k to generate real part data and imaginary part data, and perform gain adjustment, clock recovery, and carrier recovery. The DFT means 4 1 , 4 k respectively perform DFT processing on the time domain signals from the time domain processing units 3 1 , 3 k and supply frequency domain signals. The SP demodulating means 5 1 and 5 k respectively have transmission paths based on subcarriers such as SP signals and continuous pilot (hereinafter referred to as “CP”) signals in the frequency domain signals from the DFT means 4 1 and 4 k . An estimation is performed to correct the phase rotation and amplitude of the carrier. The spatial diversity combining unit 6a performs diversity combining of the output signals of the respective elements for each carrier of the SP demodulation results by the SP demodulating units 5 1 and 5k. The error correction means 7 performs deinterleaver, error correction, etc. on the diversity result or the SP demodulation result by the spatial diversity combining means 6a, and outputs it as transport stream (hereinafter referred to as “TS”) data. A known technique can be used for clock recovery and carrier recovery in the time domain processing units 3 1 and 3 k , processing by the DFT means 4 1 and 4 k , and processing by the error correction means 7.

図１１に示されるように、時間ドメイン処理部３１，３ｋはそれぞれ、ゲイン調整手段３１１，３１ｋと、１シンボル遅延手段３２１，３２ｋと、相関演算手段３３１，３３ｋと、相関強度演算手段３４１，３４ｋと、移動平均演算手段３５１，３５ｋと、ＤＦＴ窓位置検出手段３６１，３６ｋとを有する。また、ＳＰ復調手段５１，５ｋはそれぞれ、伝送路推定手段５１１，５１ｋと、ＳＰ補正手段５２１，５２ｋとを有する。時間ドメイン処理部３１，３ｋ内のゲイン調整手段３１１，３１ｋは、ＡＤ変換手段２１，２ｋの出力に対して電力演算を行い、ＡＤ変換手段２１，２ｋの出力がリファレンス値に収束するように調整する。 As shown in FIG. 11, the time domain processing units 3 1 and 3 k include gain adjustment means 31 1 and 31 k , 1-symbol delay means 32 1 and 32 k , and correlation calculation means 33 1 and 33 k , respectively. , Correlation strength calculating means 34 1 , 34 k , moving average calculating means 35 1 , 35 k , and DFT window position detecting means 36 1 , 36 k . The SP demodulating means 5 1 and 5 k include transmission path estimating means 51 1 and 51 k and SP correcting means 52 1 and 52 k , respectively. Gain adjustment means 31 1, 31 k time-domain processing unit 3 1, 3 in k performs power operation on the output of the AD converter 2 1, 2 k, AD conversion unit 2 1, 2 k output of Adjust to converge to the reference value.

次に、受信信号にＤＦＴを行う際に、ＤＦＴの窓位置を検出する一般的な方法を説明する。例えば、ＩＳＤＢ−Ｔ（地上デジタル放送）で使用するＯＦＤＭ信号は、マルチパスの影響を軽減するために送信側で有効シンボル間にガードインターバル（以下「ＧＩ」と言う。）信号を挿入する。ＧＩは、ｉ番目（ｉは、任意の正の整数）の有効シンボルの後半の一部を、ｉ−１番目とｉ番目の有効シンボル間に挿入する。一般的に、このＧＩ信号と有効シンボルの後半の一部の信号との相関性を利用し、有効シンボルの先頭を推定し、ＤＦＴの窓位置を確定する。   Next, a general method for detecting the DFT window position when performing DFT on a received signal will be described. For example, in an OFDM signal used in ISDB-T (terrestrial digital broadcasting), a guard interval (hereinafter referred to as “GI”) signal is inserted between effective symbols on the transmission side in order to reduce the influence of multipath. The GI inserts a part of the latter half of the i-th (i is an arbitrary positive integer) effective symbol between the (i−1) -th and i-th effective symbols. In general, by utilizing the correlation between the GI signal and a signal in the latter half of the effective symbol, the head of the effective symbol is estimated to determine the DFT window position.

相関性を導くために、１シンボル遅延手段３２１，３２ｋによって受信信号ａ（ｔ）を１有効シンボル長遅延させた信号ａ（ｔーＴ）を生成し、相関演算手段３３１，３３ｋによって１シンボル遅延手段３２１，３２ｋの入力信号ａ（ｔ）と１シンボル遅延手段３２１，３２ｋの出力信号ａ（ｔーＴ）との相関演算を行う。以下に、相関演算式（１）を示す。

Figure 2006332769
ここで、ｔは時刻を示し、添字Ｈを備えたａ（ｔ）Ｈはａ（ｔ）の複素共役を示し、Ｔは１有効シンボルサンプル数を示し、ｂ（ｔ）は相関演算結果を示す。 In order to derive the correlation, the signal a (t−T) obtained by delaying the received signal a (t) by one effective symbol length is generated by the one symbol delay means 32 1 , 32 k , and the correlation calculation means 33 1 , 33 k is generated. performing correlation operation between the 1-symbol delay unit 32 1, 32 k of the input signal a (t) and the one-symbol delay unit 32 1, 32 k of the output signal a (t over T) by. The correlation calculation formula (1) is shown below.
Figure 2006332769
 Here, t indicates time, a (t) H with subscript H indicates the complex conjugate of a (t), T indicates the number of 1 effective symbol samples, and b (t) indicates the correlation calculation result. .

相関強度演算手段３４１，３４ｋは、相関演算手段３３１，３３ｋの相関演算結果ｂ（ｔ）に対して相関強度ｃ（ｔ）を求めるために、相関演算結果ｂ（ｔ）信号の電力を求める。以下に、相関強度の演算式（２）を示す。

Figure 2006332769
ここで、Ｒｅ［ｂ（ｔ）］はｂ（ｔ）の実部を示し、Ｉｍ［ｂ（ｔ）］はｂ（ｔ）の虚数部を示す。なお、相関強度として相関演算結果ｂ（ｔ）の実部及び虚数部の二乗和（式（２））でなく、実部及び虚数部の各絶対値の和を利用してもよい。また、式（２）の相関強度演算結果ｃ（ｔ）に対して正規化処理を施してもよい。 The correlation strength calculation means 34 1 and 34 k obtain the correlation strength c (t) with respect to the correlation calculation result b (t) of the correlation calculation means 33 1 and 33 k . Ask for power. The correlation intensity calculation formula (2) is shown below.
Figure 2006332769
 Here, Re [b (t)] represents the real part of b (t), and Im [b (t)] represents the imaginary part of b (t). In addition, you may utilize the sum of each absolute value of a real part and an imaginary part instead of the square sum (Formula (2)) of the real part and imaginary part of a correlation calculation result b (t) as correlation strength. Moreover, you may perform a normalization process with respect to the correlation intensity | strength calculation result c (t) of Formula (2).

次に、移動平均演算手段３５１，３５ｋは、相関強度演算手段３４１，３４ｋの相関強度演算結果ｃ（ｔ）に対してＧＩ長の移動平均ｄ（ｔ）を求める。以下に、移動平均の演算式（３）を示す。

Figure 2006332769
ここで、ＧＩはガードインターバル長サンプル数を示す。 Next, the moving average calculating means 35 1 , 35 k obtains a GI length moving average d (t) with respect to the correlation strength calculating result c (t) of the correlation strength calculating means 34 1 , 34 k . The moving average calculation formula (3) is shown below.
Figure 2006332769
 Here, GI indicates the number of guard interval length samples.

図１２（ａ）乃至（ｃ）はそれぞれ、到来波、到来波の相関強度ｃ（ｔ）、到来波の相関強度のＧＩ長移動平均結果ｄ（ｔ）の一例を示す図である。１波ＡＷＧＮ（加法的白色ガウス雑音）で考えた場合、熱雑音は１有効シンボル後の熱雑音との相関を持たないので、信号のＧＩ区間で相関のある部分のみ相関強度が大きくなり、移動平均結果ｄ（ｔ）は有効シンボル先頭で最大となる。なお、相関強度を正規化していない場合は、必ずしも有効シンボル先頭で最大とはならないので、一般的には、相関強度を正規化するか、又は、複数シンボルの履歴などに基づいて有効シンボルの先頭を推定して、ＤＦＴ窓位置を確定する。   FIGS. 12A to 12C are diagrams illustrating examples of the incoming wave, the correlation strength c (t) of the incoming wave, and the GI length moving average result d (t) of the correlation strength of the incoming wave. Considering 1-wave AWGN (additive white Gaussian noise), the thermal noise has no correlation with the thermal noise after one effective symbol, so the correlation strength increases only in the correlated part in the GI section of the signal, and the movement The average result d (t) is maximum at the beginning of the effective symbol. Note that if the correlation strength is not normalized, it is not always the maximum at the beginning of the effective symbol, so in general, the correlation strength is normalized, or the beginning of the effective symbol based on the history of multiple symbols, etc. To determine the DFT window position.

図１３（ａ）乃至（ｃ）は、ＧＩ期間内の遅延波を含む受信信号でのＤＦＴ位置を示す図であり、同図（ａ）は２つの到来波、同図（ｂ）は到来波の相関強度ｃ（ｔ）、同図（ｃ）は後述する演算範囲を示す。図１３（ａ）及び（ｂ）に示されるように、遅延がＧＩ期間内である複数の到来波があった場合でも、ＤＦＴ窓先頭位置の決め方によっては、シンボル間干渉成分を含まない周波数ドメインデータを生成することが可能である。   FIGS. 13A to 13C are diagrams showing DFT positions in a received signal including a delayed wave within the GI period. FIG. 13A shows two incoming waves, and FIG. 13B shows an incoming wave. Correlation strength c (t) of FIG. 4C shows the calculation range described later. As shown in FIGS. 13 (a) and 13 (b), even when there are a plurality of incoming waves whose delays are within the GI period, depending on how the DFT window head position is determined, the frequency domain does not include intersymbol interference components. It is possible to generate data.

ＤＦＴ手段４１，４ｋは、検出されたＤＦＴ窓範囲で受信信号をＤＦＴすることにより周波数ドメインデータを生成する。 The DFT means 4 1 and 4 k generate frequency domain data by DFT of the received signal within the detected DFT window range.

ＳＰ復調手段５１，５ｋは、ＯＦＤＭ信号内にあるＳＰキャリアを利用して伝送路推定及び補正を行う。ＳＰキャリアは、既知のＰＲＢＳ（擬似乱数ビット列）信号をＢＰＳＫ変調（２相位相変調）していることより、受信側でも既知信号となる。したがって、受信されたＳＰキャリア位置ＣＡＲｉｍ ＳＰとリファレンスとなる位置ｒｉｍ ＳＰとの関係より、伝送路推定を行う。以下に、伝送路推定手段５１１，５１ｋが行う、第ｉシンボル、第ｍキャリア番号の伝送路推定演算式（４）を示す。

Figure 2006332769
ここで、ＣＡＲｉｍ ＳＰは、該当する第ｉシンボル、第ｍキャリア番号における既知のキャリア位置を示し、ｒｉｍ ＳＰは、該当する第ｉシンボル、第ｍキャリア番号におけるＤＦＴ手段４１，４ｋの出力信号を示し、Ｈｉｍ ＳＰは、該当する第ｉシンボル、第ｍキャリア番号における伝送路推定演算結果を示す。 The SP demodulating means 5 1 and 5 k perform transmission path estimation and correction using SP carriers in the OFDM signal. The SP carrier is a known signal even on the receiving side because the known PRBS (pseudo-random bit string) signal is BPSK-modulated (two-phase phase modulation). Therefore, transmission path estimation is performed based on the relationship between the received SP carrier position CAR im SP and the reference position r im SP . The following is a transmission path estimation formula (4) for the i-th symbol and the m-th carrier number, which is performed by the transmission path estimation means 51 1 and 51 k .
Figure 2006332769
 Here, CAR im SP indicates the known carrier position in the corresponding i-th symbol and m-th carrier number, and r im SP is the DFT means 4 1 , 4 k in the corresponding i-th symbol and m-th carrier number. An output signal, and H im SP indicates a transmission path estimation calculation result for the corresponding i-th symbol and m-th carrier number.

ＳＰ補正手段５２１，５２ｋは、伝送路推定演算式（４）による演算結果Ｈｉｍ ＳＰに対してシンボル方向及びキャリア方向の内挿フィルタにより受信信号の各シンボル番号及び各キャリア番号について伝送路推定を行い、推定結果Ｈｉｍと受信キャリア位置に基づいて補正を行う。以下に、ＳＰ補正手段５２１，５２ｋによる補正演算式（５）を示す。

Figure 2006332769
ここで、ｒｉｍは、第ｉシンボル、第ｍキャリア番号におけるＤＦＴ手段４１，４ｋの出力信号を示し、ｆｉｍは第ｉシンボル、第ｍキャリア番号におけるＳＰ補正手段５２１，５２ｋの出力信号を示す。 The SP correction means 52 1 , 52 k transmits the transmission path for each symbol number and each carrier number of the received signal by the interpolation filter in the symbol direction and the carrier direction with respect to the calculation result H im SP by the transmission path estimation calculation formula (4). Estimation is performed, and correction is performed based on the estimation result H im and the received carrier position. The correction calculation formula (5) by the SP correction means 52 1 and 52 k is shown below.
Figure 2006332769
 Here, r im indicates the output signal of the DFT means 4 1 , 4 k at the i-th symbol and m-th carrier number, and f im is the SP correction means 52 1 , 52 k at the i-th symbol and m-th carrier number. The output signal is shown.

空間ダイバーシチ合成の最大比合成演算は、各素子で同位相にして電力の強さを要素として合成を行う。一方、ＳＰ復調手段５１，５ｋ内の伝送路推定手段５１１，５１ｋによる推定結果Ｈｉｍは、振幅成分と位相成分を持ち、上式により各素子で該当キャリアの同位相化が行われることから、最大比合成の要素として１／Ｈｉｍの振幅成分が使用できると考えられる。このことから、伝送路推定結果の振幅成分を要素とした最大比合成を行うことによって空間ダイバーシチ合成を行うことができる。以下に、空間ダイバーシチ合成手段６ａによる最大比合成ダイバーシチ出力ＤＩＶＯＵＴｉｍの演算式（６）を示す。

Figure 2006332769
ここで、ｋは系統番号を示し、Ｋは素子の総数（ｋの最大値）を示し、添字ｋ、ｉ、及びｍはそれぞれ、第ｋ系統、第ｉシンボル、及び第ｍキャリア番号を示す。 In the maximum ratio combining operation of space diversity combining, each element is combined in the same phase with the power intensity as an element. On the other hand, the estimation result H im by the transmission path estimation means 51 1 and 51 k in the SP demodulation means 5 1 and 5 k has an amplitude component and a phase component. Therefore, it is considered that an amplitude component of 1 / H im can be used as an element of maximum ratio synthesis. From this, it is possible to perform spatial diversity combining by performing maximum ratio combining using the amplitude component of the transmission path estimation result as an element. The following is an arithmetic expression (6) for the maximum ratio combining diversity output DIVOUT im by the space diversity combining means 6a.
Figure 2006332769
 Here, k represents a system number, K represents the total number of elements (maximum value of k), and subscripts k, i, and m represent the k-th system, the i-th symbol, and the m-th carrier number, respectively.

浜住啓之ほか、「広帯域信号移動受信用帯域分散型ダイバーシチ合成受信方式の特性」、電子情報通信学会論文誌、Ｖｏｌ．Ｊ８０−Ｂ−ＩＩ、Ｎｏ．６，ｐｐ．４６６〜４７４（１９９７）Hiroyuki Hamazumi et al., “Characteristics of Band Distributed Diversity Combining System for Wideband Signal Mobile Reception”, IEICE Transactions, Vol. J80-B-II, no. 6, pp. 466-474 (1997)

上記従来の構成では、ＳＰ復調手段５１，５ｋにより各素子のキャリアが同位相になっているものとみなし、各キャリアの伝送路推定結果（１／｜Ｈｉｍ｜＝ｇｉｍ）を要素としてダイバーシチ合成を行う。一方、ゲイン調整手段３１１，３１ｋでは、復調に有効な到来波の信号レベルの大きさにかかわらず、受信信号の電力をリファレンスレベルにゲイン調整することにより、受信信号の変動を補正する。よって、到来波の信号レベルの小さい素子では雑音成分ごと大きく調整され、また、復調に有効でない到来波のＤ／Ｕ（希望波Ｄ対不要波Ｕ比率）が高く含まれる素子では復調に有効な到来波の信号レベルは小さくなり、その結果、ＳＰ復調結果のＣ／Ｎレベルが劣化する。したがって、伝送路推定結果をダイバーシチの合成要素として合成するだけでは伝送特性の劣化を十分に改善できないという問題点があった。 In the above conventional configuration, it is assumed that the carriers of each element are in phase by the SP demodulation means 5 1 and 5 k , and the transmission path estimation result (1 / | H im | = g im ) of each carrier is an element. Diversity synthesis is performed as follows. On the other hand, the gain adjusting means 31 1 , 31 k correct the fluctuation of the received signal by adjusting the gain of the received signal to the reference level regardless of the magnitude of the signal level of the incoming wave effective for demodulation. Therefore, an element having a low signal level of the incoming wave is adjusted to a large degree for each noise component, and an element having a high D / U of the incoming wave that is not effective for demodulation (ratio of desired wave D to unnecessary wave U) is effective for demodulation. The signal level of the incoming wave is reduced, and as a result, the C / N level of the SP demodulation result is deteriorated. Therefore, there is a problem in that deterioration of transmission characteristics cannot be sufficiently improved only by combining transmission path estimation results as diversity combining elements.

そこで、本発明は、上述のような従来技術の課題を解決するためになされたものであり、その目的は、各素子の伝送路推定による振幅成分に重み付けを行ってから周波数ダイバーシチ合成を行うことによってＣ／Ｎレベルを改善させることができるＯＦＤＭ受信装置を提供することにある。   Accordingly, the present invention has been made to solve the above-described problems of the prior art, and its purpose is to perform frequency diversity combining after weighting the amplitude component based on the transmission path estimation of each element. An object of the present invention is to provide an OFDM receiver that can improve the C / N level.

本発明に係るＯＦＤＭ受信装置は、ＯＦＤＭ信号を受信する複数系統の受信手段と、前記受信されたＯＦＤＭ信号の離散フーリエ変換の窓位置をシンボル毎に求める時間ドメイン処理手段、前記窓位置に基づいてＯＦＤＭ信号を離散フーリエ変換する離散フーリエ変換手段、及び前記離散フーリエ変換手段の出力信号を前記受信されたＯＦＤＭ信号に含まれるスキャッタードパイロット信号に基づいてキャリア毎に伝送路推定し、該推定結果を用いて補正するスキャッタードパイロット復調手段を含む複数系統の受信信号演算手段と、前記複数系統の受信信号演算手段の複数の前記時間ドメイン処理手段からの信号に基づいて各系統及び各シンボルについての重み付け係数を生成する重み付け生成手段と、前記複数系統の受信信号演算手段の複数の前記スキャッタードパイロット復調手段からの信号と前記重み付け生成手段から出力される重み付け係数に基づいて空間ダイバーシチ合成を行う空間ダイバーシチ合成手段とを有することを特徴としている。   An OFDM receiver according to the present invention includes a plurality of systems of receiving means for receiving an OFDM signal, time domain processing means for obtaining a discrete Fourier transform window position of the received OFDM signal for each symbol, based on the window position Discrete Fourier transform means for discrete Fourier transform of an OFDM signal, and an output signal of the discrete Fourier transform means for each carrier based on a scattered pilot signal included in the received OFDM signal, and the estimation result For each system and each symbol based on signals from a plurality of received signal calculating means including a scattered pilot demodulating means for correcting using the signal, and a plurality of time domain processing means of the received signal calculating means of the plurality of systems Weighting generating means for generating a weighting coefficient of the plurality of received signal calculating means of the plurality of systems Is characterized by having a spatial diversity synthesizing means based on a weighting factor that is output from the signal and the weighting generator from the scattered pilots demodulation means having performing spatial diversity combining.

本発明によれば、ダイバーシチ合成の合成係数がより適切になるので、空間ダイバーシチ効果を向上させることができるという効果がある。   According to the present invention, since the synthesis coefficient for diversity synthesis becomes more appropriate, there is an effect that the space diversity effect can be improved.

実施の形態１．
図１は、本発明の実施の形態１に係るＯＦＤＭ受信装置の構成を概略的に示すブロック図である。図１において、図１１（従来例）の構成と同一又は対応する構成には、同じ符号を付す。 Embodiment 1 FIG.
FIG. 1 is a block diagram schematically showing a configuration of an OFDM receiving apparatus according to Embodiment 1 of the present invention. In FIG. 1, the same or corresponding components as those in FIG. 11 (conventional example) are denoted by the same reference numerals.

実施の形態１に係るＯＦＤＭ受信装置は、周波数ドメインで空間ダイバーシチ合成を行う装置であり、重み付け生成手段８を備えている点、並びに、空間ダイバーシチ合成手段６が重み付け生成手段８によって生成された重み付け係数α１ｉ，…，αｋｉ，…を用いて空間ダイバーシチ合成を行う点が、図１１に示される従来のＯＦＤＭ受信装置と相違する。 The OFDM receiving apparatus according to the first embodiment is an apparatus that performs spatial diversity combining in the frequency domain, includes a weight generation unit 8, and weights generated by the spatial diversity combination unit 6 by the weight generation unit 8. .. Is different from the conventional OFDM receiver shown in FIG. 11 in that spatial diversity combining is performed using the coefficients α 1i ,..., Α ki,.

図１に示されるように、実施の形態１に係るＯＦＤＭ受信装置は、複数のアンテナ素子１１１，１１ｋ（図には、複数のアンテナ素子１１１，…，１１ｋ，…の内の２つを示す。）と、複数のチューナ部１１，１ｋ（図には、複数のチューナ部１１，…，１ｋ，…の内の２つを示す。）と、複数のＡＤ変換手段２１，２ｋ（図には、複数のＡＤ変換手段２１，…，２ｋ，…の内の２つを示す。）と、複数の時間ドメイン処理部３１，３ｋ（図には、複数の時間ドメイン処理部３１，…，３ｋ，…の内の２つを示す。）と、複数のＤＦＴ手段４１，４ｋ（図には、複数のＤＦＴ手段４１，…，４ｋ，…の内の２つを示す。）と、複数のＳＰ復調手段５１，５ｋ（図には、複数のＳＰ復調手段５１，…，５ｋ，…の内の２つを示す。）と、空間ダイバーシチ合成手段６と、誤り訂正手段７と、時間ドメイン処理部３１，３ｋから出力される信号に基づいて重み付け係数α１ｉ，…，αｋｉ，…を生成する重み付け生成手段８（以下、実施の形態１においては、符号８ａを用いる。）とを有する。ここで、添え字１及びｋは、各系統（アンテナ素子からＳＰ復調手段までの構成）を区別するためのものであり、それぞれ第１系統及び第ｋ系統を示し、ｋは、２以上で且つ任意の整数を表す。 As shown in FIG. 1, the OFDM receiver according to Embodiment 1 includes a plurality of antenna elements 11 1 , 11 k (in the figure, two of the plurality of antenna elements 11 1 ,..., 11 k,. ), A plurality of tuner units 1 1 , 1 k (the figure shows two of the plurality of tuner units 1 1 ,..., 1 k ,...), And a plurality of AD conversion means 2 1 , 2 k (shown in the figure are two of the plurality of AD conversion means 2 1 ,..., 2 k ,...) And a plurality of time domain processing units 3 1 , 3 k (shown in the figure) , Two of the plurality of time domain processing units 3 1 ,..., 3 k ,...) And a plurality of DFT means 4 1 , 4 k (in the figure, a plurality of DFT means 4 1 ,. 4 k, ... and.) shows two of the plurality of SP demodulation means 5 1, the 5 k (Fig., a plurality of SP demodulation means 5 1, ..., 5 k, Showing two of the.), And space diversity combining unit 6, the error correction unit 7, the time domain processing unit 3 1, 3 k based on the signal output from the weighting coefficient α 1i, ..., α ki ,... (Hereinafter, reference numeral 8a is used in the first embodiment). Here, the subscripts 1 and k are for distinguishing each system (configuration from the antenna element to the SP demodulating means), and indicate the first system and the k-th system, respectively, and k is 2 or more and Represents an arbitrary integer.

図１に示されるように、時間ドメイン処理部３１，３ｋはそれぞれ、ゲイン調整手段３１１，３１ｋと、１シンボル遅延手段３２１，３２ｋと、相関演算手段３３１，３３ｋと、相関強度演算手段３４１，３４ｋと、移動平均演算手段３５１，３５ｋと、ＤＦＴ窓位置検出手段３６１，３６ｋとを有する。また、図１に示されるように、ＳＰ復調手段５１，５ｋはそれぞれ、伝送路推定手段５１１，５１ｋと、ＳＰ補正手段５２１，５２ｋとを有する。 As shown in FIG. 1, the time domain processing units 3 1 and 3 k include gain adjustment means 31 1 and 31 k , 1-symbol delay means 32 1 and 32 k , and correlation calculation means 33 1 and 33 k , respectively. , Correlation strength calculating means 34 1 , 34 k , moving average calculating means 35 1 , 35 k , and DFT window position detecting means 36 1 , 36 k . As shown in FIG. 1, the SP demodulation means 5 1 and 5 k include transmission path estimation means 51 1 and 51 k and SP correction means 52 1 and 52 k , respectively.

チューナ部１１，１ｋはそれぞれ、アンテナ素子１１１，１１ｋで受信された信号をＩＦ信号に変換する。ＡＤ変換手段２１，２ｋはそれぞれ、チューナ部１１，１ｋからのＩＦ信号に対して離散値化の処理を行う。時間ドメイン処理部３１，３ｋはそれぞれ、ＡＤ変換手段２１，２ｋ出力を直交復調して実部データと虚数部データを生成し、ゲイン調整、クロック再生、キャリア再生を行う。ＤＦＴ手段４１，４ｋはそれぞれ、時間ドメイン処理部３１，３ｋからの時間ドメイン信号にＤＦＴ処理を施して周波数ドメイン信号を供給する。ＳＰ復調手段５１，５ｋはそれぞれ、ＤＦＴ手段４１，４ｋからの周波数ドメイン信号におけるＳＰ信号やＣＰ信号などのサブキャリアに基づいて伝送路推定を行い、キャリアの位相回転及び振幅を補正する。重み付け生成手段８ａは、時間ドメイン処理部３１，３ｋから出力される信号に基づいて重み付け係数α１ｉ，αｋｉを生成する。空間ダイバーシチ合成手段６は、重み付け係数α１ｉ，αｋｉを用いて、ＳＰ復調手段５１，５ｋによるＳＰ復調結果のキャリア毎に各素子の出力信号のダイバーシチ合成を行う。誤り訂正手段７は、空間ダイバーシチ合成手段６によるダイバーシチ結果又はＳＰ復調結果に対してデインタリーバ、誤り訂正などを行い、ＴＳデータとして出力する。 The tuner units 1 1 and 1 k convert the signals received by the antenna elements 11 1 and 11 k into IF signals, respectively. The AD conversion units 2 1 and 2 k perform discrete value processing on the IF signals from the tuner units 1 1 and 1 k , respectively. The time domain processing units 3 1 and 3 k respectively perform quadrature demodulation of the outputs of the AD conversion units 2 1 and 2 k to generate real part data and imaginary part data, and perform gain adjustment, clock recovery, and carrier recovery. The DFT means 4 1 , 4 k respectively perform DFT processing on the time domain signals from the time domain processing units 3 1 , 3 k and supply frequency domain signals. SP demodulation means 5 1 and 5 k perform transmission path estimation based on subcarriers such as SP signals and CP signals in the frequency domain signals from DFT means 4 1 and 4 k , respectively, and correct the phase rotation and amplitude of the carriers. To do. The weighting generation means 8a generates weighting coefficients α 1i and α ki based on signals output from the time domain processing units 3 1 and 3 k . The spatial diversity combining means 6 performs diversity combining of the output signals of the respective elements for each carrier of the SP demodulation results by the SP demodulating means 5 1 and 5 k using the weighting coefficients α 1i and α ki . The error correction means 7 performs deinterleaver, error correction, etc. on the diversity result or SP demodulation result by the spatial diversity combining means 6 and outputs it as TS data.

図２は、実施の形態１における重み付け生成手段８ａの構成及び空間ダイバーシチ合成手段６を示すブロック図である。図２に示されるように、重み付け生成手段８ａは、複数の相関強度信号処理手段８１ａ１，…，８１ａｋ，…と、重み付け係数生成手段８２と、生成された重み付け係数を遅延させる遅延調整手段８３とを有する。複数の相関強度信号処理手段８１ａ１，…，８１ａｋ，…はそれぞれ、記憶手段８１１と、演算範囲生成手段８１２と、平均演算手段８１３及び８１４と、差分演算手段８１５とを有する。 FIG. 2 is a block diagram showing the configuration of the weight generation means 8a and the spatial diversity combining means 6 in the first embodiment. As shown in FIG. 2, the weight generation unit 8 a includes a plurality of correlation strength signal processing units 81 a 1 ,..., 81 a k ,..., A weighting coefficient generation unit 82, and a delay adjustment unit that delays the generated weighting coefficient. 83. Each of the plurality of correlation strength signal processing means 81a 1 ,..., 81a k ,... Has storage means 811, calculation range generation means 812, average calculation means 813 and 814, and difference calculation means 815.

次に、複数の相関強度信号処理手段８１ａ１，…，８１ａｋ，…の代表例として相関強度信号処理手段８１ａ１を説明する。記憶手段８１１は、相関強度演算手段３４１から出力される相関強度演算結果ｃ１ｉ（ｔ）を、ＯＦＤＭシンボルとＧＩ長サンプル数記憶する。演算範囲生成手段８１２は、ＤＦＴ窓位置検出手段３６１から出力される窓位置検出結果ｅ１ｉ（ｔ）に基づいて、ＤＦＴ窓先頭位置を含む任意の演算範囲（例えば、ＤＦＴ窓先頭位置を中心とした任意の演算範囲）ｈａと、演算範囲ｈａ以外の範囲における任意の幅の演算範囲ｈｂに関する演算範囲情報を生成する。演算範囲ｈａは、例えば、ＤＦＴ窓先頭位置からＧＩ長サンプル数だけ前の位置から、ＤＦＴ窓先頭位置からＧＩ長サンプル数だけ後の位置までの範囲である２ＧＩ長のサンプル範囲である。平均演算手段８１３は、演算範囲ｈａに該当する相関強度演算手段３４１からの相関強度演算結果ｃ１ｉ（ｔ）の平均を計算する。平均演算手段８１４は、演算範囲ｈｂに該当する相関強度演算手段３４１からの相関強度演算結果ｃ１ｉ（ｔ）の平均を計算する。演算範囲ｈａのサンプル数と演算範囲ｈｂのサンプル数が同じである場合には、平均値を演算する代わりに、相関強度演算手段３４１からの相関強度演算結果ｃ１ｉ（ｔ）の単純な総和を演算するものとしてもよい。差分演算手段８１５は、平均演算手段８１３から出力される平均演算結果と平均演算手段８１４から出力される平均演算結果との差分を演算する。 Next, the correlation strength signal processing unit 81a 1 will be described as a representative example of the plurality of correlation strength signal processing units 81a 1 ,..., 81a k ,. The storage unit 811 stores the correlation strength calculation result c 1i (t) output from the correlation strength calculation unit 34 1 and stores the OFDM symbol and the number of GI length samples. Calculating range generation unit 812, a window position detection result output from the DFT window position detection means 36 1 on the basis of the e 1i (t), any calculation range (e.g., about a DFT window start position including the DFT window top position Calculation range information regarding the calculation range hb having an arbitrary width in a range other than the calculation range ha. The calculation range ha is, for example, a sample range of 2 GI length that is a range from a position before the DGI window head position by the number of GI length samples to a position after the DFT window head position by the number of GI length samples. Average computing means 813 computes the average of the correlation strength calculation result c 1i from the correlation intensity calculation unit 34 1 corresponding to the operation range ha (t). Average computing means 814 calculates an average of the correlation strength calculation result c 1i from the correlation intensity calculation unit 34 1 corresponding to the operation range hb (t). When the number of samples in the calculation range ha and the number of samples in the calculation range hb are the same, instead of calculating the average value, a simple sum of the correlation strength calculation results c 1i (t) from the correlation strength calculation means 34 1 It is good also as what calculates. The difference calculation means 815 calculates the difference between the average calculation result output from the average calculation means 813 and the average calculation result output from the average calculation means 814.

一例として、演算範囲ｈａをＤＦＴ窓先頭位置からＧＩ長サンプル数だけ前の位置から、ＤＦＴ窓先頭位置からＧＩ長サンプル数だけ後の位置までの範囲である２ＧＩ長のサンプル範囲とし、演算範囲ｈｂをＤＦＴ窓先頭位置からＧＩ長サンプル数だけ後の位置から、ＤＦＴ窓先頭位置からＧＩ長サンプル数だけ前の位置までの範囲とした場合の演算式（７）を以下に示す。

Figure 2006332769
ここでは、ｃｋｉ（ｔ）はｋ番目の素子を含む第ｋ系統、シンボル番号ｉ、時刻ｔにおける相関強度演算結果を示し、ＧＩは該当するガードインターバル期間を示し、ａｖｅ［・］は括弧内の平均を示し、ｈｋｉはｋ番目の素子を含む第ｋ系統、シンボル番号ｉの平均演算結果の差分を示す。 As an example, the calculation range ha is a sample range of 2 GI length that is a range from the position before the DGI window head position by the number of GI length samples to the position after the DFT window head position by the number of GI length samples, and the calculation range hb Equation (7) below is defined as a range from the position after the number of GI length samples from the DFT window head position to the position before the number of GI length samples from the head position of the DFT window.
Figure 2006332769
 Here, c ki (t) indicates the correlation strength calculation result at the k-th system including the k-th element, symbol number i, and time t, GI indicates the corresponding guard interval period, and ave [·] is in parentheses H ki represents the difference between the average calculation results of the k-th system including the k-th element and the symbol number i.

なお、平均演算手段８１３及び８１４による演算は、演算範囲ｈａ及びｈｂのすべてのサンプルを用いて行ってもよいが、例えば、等間隔にデータを間引くことによって演算に用いるサンプル数を減らしてもよい。   The calculation by the average calculation means 813 and 814 may be performed using all samples in the calculation ranges ha and hb. For example, the number of samples used for the calculation may be reduced by thinning out data at equal intervals. .

また、演算範囲ｈａ及びｈｂの定め方は、上記方法に限定されるものではなく、他の定め方を採用することもできる。例えば、演算範囲ｈｂは、演算範囲ｈａ以外のすべての範囲としてもよいが、演算範囲ｈａ以外の一部の範囲としてもよい。   Further, the method of determining the calculation ranges ha and hb is not limited to the above method, and other methods of determination can be employed. For example, the calculation range hb may be all ranges other than the calculation range ha, or may be a partial range other than the calculation range ha.

さらに、ＤＦＴ窓先頭位置情報として、該当シンボル（シンボル番号ｉ）の１シンボル前のシンボル（シンボル番号（ｉ−１））の結果を使用してもよい。   Furthermore, the result of the symbol (symbol number (i−1)) one symbol before the corresponding symbol (symbol number i) may be used as the DFT window head position information.

さらにまた、式（７）における平均演算結果の差分ｈｋｉについて、予め任意の下限値を決めておき、平均演算結果の差分ｈｋｉがこの下限値より小さくなる場合に、重み付け係数α１ｉ，…，αｋｉ，…を、予め決められた値に制限してもよい。 Furthermore, an arbitrary lower limit value is determined in advance for the difference h ki of the average calculation result in the equation (7), and the weighting coefficient α 1i ,... When the difference h ki of the average calculation result is smaller than the lower limit value. , Α ki ,... May be limited to predetermined values.

従来技術の説明に用いた図１３（ａ）乃至（ｃ）に示されるように、ＤＦＴ窓先頭位置に対して到来波のＧＩ先頭がＤＦＴ窓先頭位置よりＧＩ長以内前にあれば、ＤＦＴ後の周波数ドメイン信号としてＩＳＩ（シンボル間干渉）のない到来波としての扱いが可能となる。また、各到来波のＧＩ相関強度を求めた場合、送信信号で相関を持つＧＩ区間では相関を持ち、それ以外の区間では相関を持たないため平均的に見れば相関強度は小さくなる。   As shown in FIGS. 13A to 13C used in the description of the prior art, after the DFT, if the GI head of the incoming wave is within the GI length before the DFT window head position with respect to the DFT window head position, Can be treated as an incoming wave without ISI (intersymbol interference) as a frequency domain signal. When the GI correlation strength of each incoming wave is obtained, the correlation strength is small in the GI section having the correlation in the transmission signal, and there is no correlation in the other sections.

図３（ａ）乃至（ｃ）は、各素子でのＤＦＴ窓位置と復調に有効となる２つの到来波との関係を示す図である。ここでは、ＤＦＴ窓先頭位置が同じ位置である場合を説明する。例えば、アンテナ素子の指向性等の違いにより、図３（ｂ）及び（ｃ）に示されるように、各アンテナ素子間で到来波の信号成分レベルに違いがある場合、又は、各アンテナ素子間で熱雑音レベルに違いがある場合、又は、妨害波の入力レベルに違いがある場合には、ゲイン調整手段３１１，３１ｋによるゲイン調整よって、各素子で復調に有効な到来波の相関強度の大きさに違いが現れ、また、それ以外の影響にも違いが生じる。したがって、有効な到来波となりうる範囲のＧＩ相関強度と、それ以外の範囲の相関強度との差分を取ることにより得られた値ｈｋｉは、復調に有効な到来波の信号レベルの強さを表す要素と考えることができ、各素子の差分演算結果ｈｋｉからダイバーシチ合成要素の重み付け係数α１ｉ，…，αｋｉ，…を生成して、空間ダイバーシチ合成手段６によるダイバーシチ合成に重み付け係数α１ｉ，…，αｋｉ，…を反映させることは有効である。 FIGS. 3A to 3C are diagrams showing the relationship between the DFT window position in each element and two incoming waves effective for demodulation. Here, a case where the DFT window head position is the same position will be described. For example, when there is a difference in the signal component level of the arriving wave between the respective antenna elements as shown in FIGS. 3B and 3C due to differences in directivity of the antenna elements, or between the antenna elements If there is a difference in the thermal noise level, or if there is a difference in the input level of the disturbing wave, the gain adjustment means 31 1 , 31 k adjusts the gain of the incoming wave effective for demodulation in each element. A difference appears in the size of, and other effects also differ. Therefore, the value h ki obtained by taking the difference between the GI correlation strength in the range that can be an effective incoming wave and the correlation strength in the other range is the signal level strength of the incoming wave that is effective for demodulation. represents can be considered as elements, weighting coefficients alpha 1i diversity combining elements from the difference calculation result h ki of each element, ..., alpha ki, to generate ... weighting factor alpha 1i to diversity combining by the spatial diversity combining means 6 ,..., Α ki ,... Are effective.

各素子の重み付け係数α１ｉ，…，αｋｉ，…は、そのままαｋｉ＝ｈｋｉとしてもよい。また、各素子のｈｋｉに対して共通の任意の係数を掛けてもよい。さらに、各素子の重み付け係数α１ｉ，…，αｋｉ，…を下位にビットシフトさせることによって得られた値を重み付け係数としてダイバーシチ合成に用いてもよい。 The weighting coefficients α 1i ,..., Α ki ,... Of each element may be set as α ki = h ki as they are. Further, a common arbitrary coefficient may be multiplied to h ki of each element. Further, a value obtained by bit shifting the weighting coefficients α 1i ,..., Α ki ,... Of each element to the lower order may be used for diversity combining as a weighting coefficient.

以下に、空間ダイバーシチ合成手段６による合成演算式（８）を示す。

Figure 2006332769
Hereinafter, a synthesis operation formula (8) by the space diversity synthesis means 6 is shown.
Figure 2006332769

以上に説明した処理によって、各素子の復調として有効な到来波の信号成分の大きさに応じて重み付けが可能となり、重み付け係数を用いない場合に比べ、ダイバーシチ効果を向上させることが可能になる。すなわち、各素子の伝送路推定による振幅成分に加えて、各素子のＤＦＴ窓位置より確定する復調に有効となる信号シンボル位置に近接する任意の区間の相関強度レベルの積和結果とこの積和範囲以外の相関強度レベルとの平均との差分を取ったものを重み付け要素として重み付けを行い、周波数ドメインでのダイバーシチ合成を行うことにより、ダイバーシチ合成の合成係数がより最適となるので空間ダイバーシチ効果の向上が期待できるという効果がある。   By the processing described above, weighting can be performed according to the magnitude of the signal component of the incoming wave that is effective for demodulation of each element, and the diversity effect can be improved as compared with the case where no weighting coefficient is used. That is, in addition to the amplitude component obtained by channel estimation of each element, the product-sum result of the correlation strength level in an arbitrary section close to the signal symbol position effective for demodulation determined from the DFT window position of each element and this product-sum By weighting the difference between the average and the correlation intensity level outside the range as a weighting factor, and performing diversity combining in the frequency domain, the combining coefficient of diversity combining becomes more optimal, so the spatial diversity effect There is an effect that improvement can be expected.

なお、上記説明においては、ＯＦＤＭ受信装置がハードウェア（以下「Ｈ／Ｗ」と言う。）として構成された場合を説明したが、デジタルシグナルプロセッサ（以下「ＤＳＰ」と言う。）等によるファームウェアによって重み付け係数生成を行うように構成することもできる。   In the above description, the case where the OFDM receiver is configured as hardware (hereinafter referred to as “H / W”) has been described. However, the firmware is based on firmware such as a digital signal processor (hereinafter referred to as “DSP”). It can also be configured to perform weighting coefficient generation.

実施の形態２．
図４は、本発明の実施の形態２に係るＯＦＤＭ受信装置の重み付け生成手段８ｂの構成及び空間ダイバーシチ合成手段６を示すブロック図である。図４において、図２（実施の形態１）の構成と同一又は対応する構成には、同じ符号を付す。また、実施の形態２に係るＯＦＤＭ受信装置は、図１に示される構成を有する。したがって、実施の形態２の説明においては、図１をも参照する。 Embodiment 2. FIG.
FIG. 4 is a block diagram showing the configuration of the weight generating means 8b and the spatial diversity combining means 6 of the OFDM receiving apparatus according to Embodiment 2 of the present invention. In FIG. 4, the same or corresponding components as those in FIG. 2 (Embodiment 1) are denoted by the same reference numerals. Further, the OFDM receiving apparatus according to Embodiment 2 has the configuration shown in FIG. Therefore, FIG. 1 is also referred to in the description of the second embodiment.

図４に示されるように、実施の形態２における重み付け生成手段８ｂは、時間ドメイン処理部３１，３ｋの情報に基づいて重み付け係数を生成する。重み付け生成手段８ｂは、複数の相関強度信号処理手段８１ｂ１，８１ｂｋ（図には、複数の相関強度信号処理手段８１ｂ１，…，８１ｂｋ，…の内の２つを示す。）と、重み付け係数生成手段８２と、遅延調整手段８３とを有する。相関強度信号処理手段８１ｂ１，８１ｂｋはそれぞれ、記憶手段８１１と、演算範囲生成手段８１２と、フィルタ演算手段８１６と、平均演算手段８１４と、差分演算手段８１５とを有する。 As shown in FIG. 4, the weight generation unit 8b according to the second embodiment generates a weighting coefficient based on information of the time domain processing units 3 1 and 3 k . The weighting generation means 8b includes a plurality of correlation strength signal processing means 81b 1 , 81b k (the figure shows two of the plurality of correlation strength signal processing means 81b 1 ,..., 81b k ,...). Weighting coefficient generating means 82 and delay adjusting means 83 are provided. Each of the correlation strength signal processing units 81b 1 and 81b k includes a storage unit 811, a calculation range generation unit 812, a filter calculation unit 816, an average calculation unit 814, and a difference calculation unit 815.

実施の形態２に係るＯＦＤＭ受信装置は、相関強度信号処理手段８１ｂ１，８１ｂｋのそれぞれにおいて、実施の形態１に係るＯＦＤＭ受信装置の重み付け生成手段８ａの平均演算手段８１３に代えて、フィルタ演算手段８１６を備えた点が、実施の形態１に係るＯＦＤＭ受信装置と相違する。すなわち、実施の形態２における相関強度信号処理手段８１ｂ１，８１ｂｋは、時間ドメイン処理部３１，３ｋにおける相関強度演算結果ｃｋｉ（ｔ），ｃｋｉ（ｔ）を入力とし、ＤＦＴ窓先頭位置に近づくにつれて値が増加し、ＤＦＴ窓先頭位置で値が最大となる係数ｈ（ｔ）を掛けるフィルタ演算手段８１６を備え、差分演算手段８１５によって、フィルタ演算手段８１６からの出力と平均演算手段８１４から出力されるＤＦＴ窓付近以外の相関強度平均との差分を取り、この差分に基づいて重み付け係数を生成する。 In the OFDM receiving apparatus according to the second embodiment, in each of the correlation strength signal processing means 81b 1 and 81b k , a filter calculation is performed instead of the average calculating means 813 of the weight generation means 8a of the OFDM receiving apparatus according to the first embodiment. The point that means 816 is provided is different from the OFDM receiver according to Embodiment 1. That is, the correlation strength signal processing means 81b 1 , 81b k in the second embodiment receives the correlation strength calculation results c ki (t), c ki (t) in the time domain processing units 3 1 , 3 k as input, and the DFT window A filter operation means 816 that multiplies the coefficient h (t) that increases in value as it approaches the head position and that has the maximum value at the head position of the DFT window is provided, and the difference operation means 815 outputs and averages the output from the filter operation means 816. The difference from the average correlation strength other than the vicinity of the DFT window output from the means 814 is taken, and a weighting coefficient is generated based on this difference.

次に、複数の相関強度信号処理手段８１ｂ１，８１ｂｋの代表例として相関強度信号処理手段８１ｂ１を説明する。記憶手段８１１は、相関強度演算手段３４１から出力される相関強度演算結果ｃ１ｉ（ｔ）を、ＯＦＤＭシンボルとＧＩ長サンプル数記憶する。演算範囲生成手段８１２は、ＤＦＴ窓位置検出手段３６１から出力される窓位置検出結果ｅ１ｉ（ｔ）から、ＤＦＴ窓先頭位置を中心とした任意の演算範囲ｈａと、演算範囲ｈａ以外の範囲における任意の幅の演算範囲ｈｂに関する演算範囲情報を生成する。演算範囲ｈａは、例えば、ＤＦＴ窓先頭位置からＧＩ長サンプル数だけ前の位置から、ＤＦＴ窓先頭位置からＧＩ長サンプル数だけ後の位置までの範囲である２ＧＩのサンプル範囲である。フィルタ演算手段８１６は、演算範囲ｈａの範囲に該当する相関強度演算手段３４１からの相関強度演算結果ｃ１ｉ（ｔ）のフィルタ演算を行う。平均演算手段８１４は、演算範囲ｈｂに該当する相関強度演算手段３４１からの相関強度演算結果ｃ１ｉ（ｔ）の平均を計算する。演算範囲ｈａのサンプル数と演算範囲ｈｂのサンプル数が同じである場合には、相関強度演算手段３４１からの相関強度演算結果ｃ１ｉ（ｔ）の単純な総和を演算するものとしてもよい。差分演算手段８１５は、フィルタ演算手段８１６の演算結果と平均演算手段８１４の平均演算結果との差分を演算する。 Next, the correlation strength signal processing unit 81b 1 will be described as a representative example of the plurality of correlation strength signal processing units 81b 1 and 81b k . The storage unit 811 stores the correlation strength calculation result c 1i (t) output from the correlation strength calculation unit 34 1 and stores the OFDM symbol and the number of GI length samples. Calculating range generation unit 812, the window position detection result output from the DFT window position detection means 36 1 e 1i (t), and any calculation range ha around the DFT window top position, the range other than the calculation range ha The calculation range information related to the calculation range hb having an arbitrary width is generated. The calculation range ha is, for example, a 2GI sample range which is a range from a position before the DGI window head position by the number of GI length samples to a position after the DFT window head position by the number of GI length samples. Filtering operation unit 816 performs the filter operation of the correlation intensity calculation result c 1i from the correlation intensity calculation unit 34 1 corresponding to the range of the calculation range ha (t). Average computing means 814 calculates an average of the correlation strength calculation result c 1i from the correlation intensity calculation unit 34 1 corresponding to the operation range hb (t). If the sample number of sample number and calculation range hb calculation range ha are the same, it may be those for calculating a simple sum of the correlation intensity calculation result c 1i from the correlation intensity calculating means 34 1 (t). The difference calculation means 815 calculates the difference between the calculation result of the filter calculation means 816 and the average calculation result of the average calculation means 814.

次に、フィルタ演算手段８１６について説明する。図５（ａ）乃至（ｃ）は、ＤＦＴ窓範囲及びＤＦＴ窓先頭位置、フィルタ係数ｈ（ｔ）、演算範囲ｈａ及びｈｂと到来波との関係を示す図である。図５（ａ）乃至（ｃ）に示されるように、ＤＦＴ窓先頭位置から離れる程ＩＳＩ成分を含む到来波の相関強度を含むことになるため、図５（ｂ）に示されるようなＤＦＴ窓先頭位置を頂点としたフィルタ係数ｈ（ｔ）を生成し、フィルタ係数と相関強度値との掛け算の結果を求める。なお、フィルタ係数は、図５（ｂ）に示されるものに限定されず、任意のサンプル数単位で段階的に係数値を変動させるものであってもよい。   Next, the filter calculation means 816 will be described. FIGS. 5A to 5C are diagrams illustrating the relationship between the DFT window range, the DFT window head position, the filter coefficient h (t), the calculation ranges ha and hb, and the incoming wave. As shown in FIGS. 5A to 5C, since the correlation strength of the incoming wave including the ISI component is included as the distance from the DFT window head position increases, the DFT window as shown in FIG. A filter coefficient h (t) having the head position as a vertex is generated, and a result of multiplication of the filter coefficient and the correlation strength value is obtained. The filter coefficient is not limited to that shown in FIG. 5B, and the coefficient value may be changed step by step in an arbitrary number of samples.

以上に説明した処理によって、有効な到来波よりＧＩサンプル期間内に到達するＩＳＩ成分を持つ到来波の影響を軽減することが可能となり、重み付け係数としてより尤もらしい係数を生成することができ、その結果、更なるダイバーシチ効果の向上が期待できる。すなわち、ＤＦＴ窓先頭位置を最大としてＤＦＴ窓先頭位置より離れるにつれて相関強度の値にゲインを下げたものの累積和を取ったものを重み付け要素として使用することにより、復調に有効な到来波に近接し且つＩＳＩ成分を持つ到来波の成分を軽減した相関強度を生成することが可能となり、より空間ダイバーシチ効果の向上が期待できるという効果がある。   Through the processing described above, it becomes possible to reduce the influence of an incoming wave having an ISI component that arrives within a GI sample period from a valid incoming wave, and a more likely coefficient can be generated as a weighting coefficient. As a result, further improvement of the diversity effect can be expected. That is, by using as a weighting element a value obtained by reducing the gain to the correlation strength value as the distance from the DFT window head position becomes the maximum, and using the cumulative sum as a weighting element, it is possible to approach an incoming wave effective for demodulation. In addition, it is possible to generate a correlation strength in which an incoming wave component having an ISI component is reduced, and there is an effect that an improvement in spatial diversity effect can be expected.

なお、上記説明においては、ＯＦＤＭ受信装置がＨ／Ｗとして構成された場合を説明したが、ＤＳＰ等によるファームウェアによって重み付け係数生成を行うように構成することもできる。   In the above description, the case where the OFDM receiver is configured as H / W has been described. However, weighting coefficient generation may be performed by firmware such as a DSP.

また、実施の形態２において、上記以外の点は、上記実施の形態１の場合と同じである。   In the second embodiment, the points other than the above are the same as those in the first embodiment.

実施の形態３．
図６は、本発明の実施の形態３に係るＯＦＤＭ受信装置の重み付け生成手段８ｃの構成及び空間ダイバーシチ合成手段６を示すブロック図である。図６において、図２（実施の形態１）の構成と同一又は対応する構成には、同じ符号を付す。また、実施の形態３に係るＯＦＤＭ受信装置は、図１に示される構成を有する。したがって、実施の形態３の説明においては、図１をも参照する。 Embodiment 3 FIG.
FIG. 6 is a block diagram showing the configuration of the weight generation means 8c and the spatial diversity combining means 6 of the OFDM receiver according to Embodiment 3 of the present invention. In FIG. 6, the same or corresponding components as those in FIG. 2 (Embodiment 1) are denoted by the same reference numerals. Further, the OFDM receiving apparatus according to Embodiment 3 has the configuration shown in FIG. Therefore, FIG. 1 is also referred to in the description of the third embodiment.

図６に示されるように、重み付け生成手段８ｃは、時間ドメイン処理部３１，３ｋの情報に基づいて重み付け係数を生成する。重み付け生成手段８ｃは、複数の相関強度信号処理手段８１ｃ１，８１ｃｋ（図には、複数の相関強度信号処理手段８１ｃ１，…，８１ｃｋ，…の内の２つを示す。）と、ゲイン調整手段８４と、重み付け係数生成手段８２と、遅延調整手段８３とを有する。複数の相関強度信号処理手段８１ｃ１，８１ｃｋはそれぞれ、記憶手段８１１と、演算範囲生成手段８１２ａと、平均演算手段８１３と、平均演算手段８１４ａとを有する。 As shown in FIG. 6, the weight generation unit 8 c generates a weighting coefficient based on information of the time domain processing units 3 1 and 3 k . The weight generation means 8c includes a plurality of correlation strength signal processing means 81c 1 , 81c k (the figure shows two of the plurality of correlation strength signal processing means 81c 1 ,..., 81c k ,...). Gain adjustment means 84, weighting coefficient generation means 82, and delay adjustment means 83 are provided. Each of the plurality of correlation strength signal processing units 81c 1 and 81c k includes a storage unit 811, a calculation range generation unit 812a, an average calculation unit 813, and an average calculation unit 814a.

実施の形態３に係るＯＦＤＭ受信装置は、相関強度信号処理手段８１ｃ１，８１ｃｋのそれぞれにおいて差分演算手段を備えていない点、及び、ゲイン調整手段８４を備えている点が、実施の形態１に係るＯＦＤＭ受信装置と相違する。すなわち、実施の形態１においては復調として有効な到来波の到達時間範囲における相関強度値平均と、復調として有効でない範囲での相関強度平均との差を求めて重み付け係数を生成したが、実施の形態３においては各素子の復調として有効でない範囲での相関強度平均が共通レベルになるようにゲイン調整するゲイン調整手段８４を備え、ゲイン調整された相関強度平均に基づいて重み付け係数を生成する。 The OFDM receiving apparatus according to the third embodiment is different from the first embodiment in that each of the correlation strength signal processing means 81c 1 and 81c k is not provided with a difference calculating means and a gain adjusting means 84 is provided. This is different from the OFDM receiving apparatus according to FIG. That is, in the first embodiment, the weighting coefficient is generated by obtaining the difference between the correlation strength value average in the arrival time range of the arrival wave effective as the demodulation and the correlation strength average in the range not effective as the demodulation. In the third aspect, gain adjustment means 84 is provided for adjusting the gain so that the correlation intensity average in a range that is not effective as demodulation of each element becomes a common level, and a weighting coefficient is generated based on the gain-adjusted correlation intensity average.

次に、複数の相関強度信号処理手段８１ｃ１，８１ｃｋの代表例として相関強度信号処理手段８１ｃ１を説明する。記憶手段８１１は、相関強度演算手段３４１から出力される相関強度演算結果ｃ１ｉ（ｔ）を、ＯＦＤＭシンボルとＧＩ長サンプル数記憶する。演算範囲生成手段８１２は、ＤＦＴ窓位置検出手段３６１から出力される窓位置検出結果ｅ１ｉ（ｔ）から、ＤＦＴ窓先頭位置を中心とした任意の演算範囲ｈａと、演算範囲ｈａ以外の範囲における任意の幅の演算範囲ｈｂに関する演算範囲情報を生成する。平均演算手段８１４ａは、任意の演算範囲を選択し、その範囲の相関強度平均を演算する。例えば、ＤＦＴ窓先頭位置より１／２ＯＦＤＭシンボルサンプル長後ろから３／４ＯＦＤＭシンボルサンプル長後ろの区間で相関強度平均を計算する。なお、平均演算手段８１４ａが使用する演算範囲として、実施の形態１における平均演算手段８１４で使用する演算範囲ｈｂを用いてもよい。 Next, the correlation strength signal processing unit 81c 1 will be described as a representative example of the plurality of correlation strength signal processing units 81c 1 and 81c k . The storage unit 811 stores the correlation strength calculation result c 1i (t) output from the correlation strength calculation unit 34 1 and stores the OFDM symbol and the number of GI length samples. Calculating range generation unit 812, the window position detection result output from the DFT window position detection means 36 1 e 1i (t), and any calculation range ha around the DFT window top position, the range other than the calculation range ha The calculation range information related to the calculation range hb having an arbitrary width is generated. The average calculation means 814a selects an arbitrary calculation range, and calculates the average correlation strength of the range. For example, the average correlation strength is calculated in a section that is 3/4 OFDM symbol sample length after the 1/2 OFDM symbol sample length after the DFT window head position. Note that the calculation range hb used by the average calculation means 814 in Embodiment 1 may be used as the calculation range used by the average calculation means 814a.

ゲイン調整手段８４は、各素子に対応する各相関強度信号処理手段の平均演算手段８１４ａの演算結果が、同じ値となるようにゲイン係数を生成し、各素子の平均演算手段８１３の出力をゲイン調整する。これは、平均演算手段８１４ａの出力信号が所望でない熱雑音の相関強度に相当するので、各素子の平均演算手段８１４ａのレベルが同一になるように平均演算手段８１３の出力をゲイン調整することで、各素子の復調に有効な到来波の相関強度を得ることが可能になるからである。   The gain adjustment unit 84 generates a gain coefficient so that the calculation result of the average calculation unit 814a of each correlation strength signal processing unit corresponding to each element has the same value, and gains the output of the average calculation unit 813 of each element. adjust. This is because the output signal of the average calculation means 814a corresponds to the correlation strength of an undesired thermal noise, and thus the gain of the output of the average calculation means 813 is adjusted so that the level of the average calculation means 814a of each element becomes the same. This is because it is possible to obtain the correlation strength of the incoming wave effective for the demodulation of each element.

以上に説明した処理によって、各素子の雑音成分による影響を同レベルにできるようになり、復調に有効な到来波の信号強度を示す重み付け要素を生成することが可能となり、重み付け係数としてより尤もらしい係数を生成することができ、その結果、更なるダイバーシチ効果の向上が期待できる。すなわち、ＤＦＴ窓先頭位置から離れた範囲でのＧＩ相関強度の平均値を各素子の雑音による相関強度とみなし、各素子の雑音レベルが同レベルになるようにゲイン係数を生成し各素子のＤＦＴ窓先頭付近の相関強度平均値に掛けたものを重み付け要素として重み付けを行い、周波数ドメインでのダイバーシチ合成を行うことにより、よりＳ／Ｎ比に近い重み付け要素になることから、ダイバーシチ合成の合成係数がより最適となるので空間ダイバーシチ効果の向上が期待できる。   The processing described above enables the influence of the noise component of each element to be at the same level, and it is possible to generate a weighting element indicating the signal strength of an incoming wave effective for demodulation, which is more likely as a weighting coefficient. A coefficient can be generated, and as a result, further improvement of the diversity effect can be expected. That is, the average value of the GI correlation strength in a range away from the DFT window head position is regarded as the correlation strength due to the noise of each element, and a gain coefficient is generated so that the noise level of each element becomes the same level. By multiplying the average value of the correlation strength near the top of the window as a weighting element and performing diversity combining in the frequency domain, it becomes a weighting element closer to the S / N ratio. Can be expected to improve the space diversity effect.

なお、上記説明においては、ＯＦＤＭ受信装置がＨ／Ｗとして構成された場合を説明したが、ＤＳＰ等によるファームウェアによって重み付け係数生成を行うように構成することもできる。   In the above description, the case where the OFDM receiver is configured as H / W has been described. However, weighting coefficient generation may be performed by firmware such as a DSP.

また、実施の形態３において、上記以外の点は、上記実施の形態１又は２の場合と同じである。   Further, in the third embodiment, the points other than the above are the same as those in the first or second embodiment.

実施の形態４．
図７は、本発明の実施の形態４に係るＯＦＤＭ受信装置の重み付け生成手段８ｄの構成及び空間ダイバーシチ合成手段６を示すブロック図である。図７において、図６（実施の形態３）の構成と同一又は対応する構成には、同じ符号を付す。また、実施の形態４に係るＯＦＤＭ受信装置は、図１に示される構成を有する。したがって、実施の形態４の説明においては、図１をも参照する。 Embodiment 4 FIG.
FIG. 7 is a block diagram showing the configuration of the weighting generation means 8d and the spatial diversity combining means 6 of the OFDM receiving apparatus according to Embodiment 4 of the present invention. In FIG. 7, the same or corresponding components as those in FIG. 6 (Embodiment 3) are denoted by the same reference numerals. Further, the OFDM receiving apparatus according to Embodiment 4 has the configuration shown in FIG. Therefore, FIG. 1 is also referred to in the description of the fourth embodiment.

図７に示されるように、重み付け生成手段８ｄは、時間ドメイン処理部３１，３ｋの情報に基づいて重み付け係数を生成する。重み付け生成手段８ｄは、複数の相関強度信号処理手段８１ｄ１，８１ｄｋ（図には、複数の相関強度信号処理手段８１ｄ１，…，８１ｄｋ，…の内の２つを示す。）と、ゲイン調整手段８４と、重み付け係数生成手段８２と、遅延調整手段８３とを有する。複数の相関強度信号処理手段８１ｄ１，８１ｄｋはそれぞれ、記憶手段８１１と、演算範囲生成手段８１２ａと、フィルタ手段８１６と、平均演算手段８１４ａとを有する。 As shown in FIG. 7, the weight generation unit 8 d generates a weighting coefficient based on information of the time domain processing units 3 1 and 3 k . The weighting generation means 8d includes a plurality of correlation strength signal processing means 81d 1 , 81d k (the figure shows two of the plurality of correlation strength signal processing means 81d 1 ,..., 81d k ,...). Gain adjustment means 84, weighting coefficient generation means 82, and delay adjustment means 83 are provided. Each of the plurality of correlation strength signal processing units 81d 1 and 81d k includes a storage unit 811, a calculation range generation unit 812a, a filter unit 816, and an average calculation unit 814a.

実施の形態４に係るＯＦＤＭ受信装置は、相関強度信号処理手段８１ｄ１，８１ｄｋのそれぞれにおいて、実施の形態３に係るＯＦＤＭ受信装置の重み付け生成手段８ｄの平均演算手段８１３に代えて、フィルタ演算手段８１６を備えた点が、実施の形態３に係るＯＦＤＭ受信装置と相違する。すなわち、実施の形態４における相関強度信号処理手段８１ｄ１，８１ｄｋは、時間ドメイン処理部３１，３ｋにおける相関強度演算結果ｃｋｉ（ｔ），ｃｋｉ（ｔ）を入力とし、ＤＦＴ窓先頭位置に近づくにつれて値が増加し、ＤＦＴ窓先頭位置で値が最大となる係数ｈ（ｔ）を掛けるフィルタ演算手段８１６を備え、平均演算手段８１４ａからの出力に基づいてゲイン調整をし、フィルタ演算手段８１６からの出力と平均演算手段８１４ａからの出力に基づいて重み付け係数を生成する。 In the OFDM receiving apparatus according to the fourth embodiment, in each of the correlation strength signal processing means 81d 1 and 81d k , a filter calculation is performed instead of the average calculating means 813 of the weight generation means 8d of the OFDM receiving apparatus according to the third embodiment. The point provided with the means 816 is different from the OFDM receiver according to the third embodiment. That is, the correlation strength signal processing means 81d 1 , 81d k in the fourth embodiment receives the correlation strength calculation results c ki (t), c ki (t) in the time domain processing units 3 1 , 3 k as input, and the DFT window A filter operation unit 816 that multiplies a coefficient h (t) that increases in value as it approaches the head position and that has the maximum value at the head position of the DFT window, adjusts the gain based on the output from the average operation unit 814a, A weighting coefficient is generated based on the output from the calculation means 816 and the output from the average calculation means 814a.

次に、複数の相関強度信号処理手段８１ｄ１，８１ｄｋの代表例として相関強度信号処理手段８１ｄ１を説明する。記憶手段８１１は、相関強度演算手段３４１から出力される相関強度演算結果ｃ１ｉ（ｔ）を、ＯＦＤＭシンボルとＧＩ長サンプル数記憶する。演算範囲生成手段８１２は、ＤＦＴ窓位置検出手段３６１から出力される窓位置検出結果ｅ１ｉ（ｔ）から、ＤＦＴ窓先頭位置を中心とした任意の演算範囲ｈａと、演算範囲ｈａ以外の範囲における任意の幅の演算範囲ｈｂに関する演算範囲情報を生成する。平均演算手段８１４ａは、任意の演算範囲を選択し、その範囲の相関強度平均を演算する。例えば、ＤＦＴ窓先頭位置より１／２ＯＦＤＭシンボルサンプル長後ろから３／４ＯＦＤＭシンボルサンプル長後ろの区間で相関強度平均を計算する。なお、平均演算手段８１４ａが使用する演算範囲として、実施の形態１における平均演算手段８１４で使用する演算範囲ｈｂを用いてもよい。 Next, the correlation strength signal processing unit 81d 1 will be described as a representative example of the plurality of correlation strength signal processing units 81d 1 and 81d k . The storage unit 811 stores the correlation strength calculation result c 1i (t) output from the correlation strength calculation unit 34 1 and stores the OFDM symbol and the number of GI length samples. Calculating range generation unit 812, the window position detection result output from the DFT window position detection means 36 1 e 1i (t), and any calculation range ha around the DFT window top position, the range other than the calculation range ha The calculation range information related to the calculation range hb having an arbitrary width is generated. The average calculation means 814a selects an arbitrary calculation range, and calculates the average correlation strength of the range. For example, the average correlation strength is calculated in a section that is 3/4 OFDM symbol sample length after the 1/2 OFDM symbol sample length after the DFT window head position. Note that the calculation range hb used by the average calculation means 814 in Embodiment 1 may be used as the calculation range used by the average calculation means 814a.

ゲイン調整手段８４は、各素子に対応する各相関強度信号処理手段の平均演算手段８１４ａの結果に対し、同じ平均結果となるようにゲイン係数を生成し、各素子のフィルタ演算手段８１６の出力をゲイン調整する。これは、平均演算手段８１４ａの出力信号が所望でない熱雑音の相関強度に相当するので、各素子の平均演算手段８１４ａのレベルが同一になるようにフィルタ演算手段８１６の出力をゲイン調整することで、各素子の復調に有効な到来波の相関強度を得ることが可能になるからである。   The gain adjusting unit 84 generates a gain coefficient so as to obtain the same average result for the result of the average calculating unit 814a of each correlation strength signal processing unit corresponding to each element, and outputs the output of the filter calculating unit 816 of each element. Adjust the gain. This is because the output signal of the average calculation means 814a corresponds to the correlation strength of an undesired thermal noise, and the gain of the output of the filter calculation means 816 is adjusted so that the level of the average calculation means 814a of each element becomes the same. This is because it is possible to obtain the correlation strength of the incoming wave effective for the demodulation of each element.

以上に説明した処理によって、各素子の雑音成分による差を統一させることでより復調に有効な到来波の信号強度を示す重み付け要素を生成することが可能となり、重み付け係数としてより尤もらしい係数を生成することができ、その結果、更なるダイバーシチ効果の向上が期待できる。すなわち、ＤＦＴ窓先頭位置を最大としてＤＦＴ窓先頭位置より離れるにつれて相関強度の値にゲインを下げたものの累積和を取ったものと、ＤＦＴ窓先頭位置から離れた範囲でのＧＩ相関強度の平均値を各素子の雑音による相関強度とみなし、各素子の雑音レベルが同レベルになるようにゲイン係数を生成し、各素子のＤＦＴ窓先頭付近の相関強度平均値に掛けたものを重み付け要素として重み付けを行い周波数ドメインでのダイバーシチ合成を行うことにより、よりＳ／Ｎ比に近い重み付け要素になることから、ダイバーシチ合成の合成係数がより最適となるので空間ダイバーシチ効果の向上が期待できる。   Through the processing described above, it is possible to generate a weighting element that indicates the signal strength of an incoming wave that is more effective for demodulation by unifying the difference due to the noise component of each element, and to generate a more likely coefficient as a weighting coefficient. As a result, further improvement of the diversity effect can be expected. That is, the maximum sum of the DFT window head position and the gain obtained by reducing the gain as the distance from the DFT window head position increases, and the average value of the GI correlation strengths in the range away from the DFT window head position. Is regarded as the correlation strength due to the noise of each element, a gain coefficient is generated so that the noise level of each element becomes the same level, and weighted by multiplying the correlation intensity average value near the head of the DFT window of each element as a weighting element By performing the diversity combining in the frequency domain, the weighting factor closer to the S / N ratio is obtained, so that the combining coefficient of the diversity combining becomes more optimal, so that an improvement in the spatial diversity effect can be expected.

なお、上記説明においては、ＯＦＤＭ受信装置がＨ／Ｗとして構成された場合を説明したが、ＤＳＰ等によるファームウェアによって重み付け係数生成を行うように構成することもできる。   In the above description, the case where the OFDM receiver is configured as H / W has been described. However, weighting coefficient generation may be performed by firmware such as a DSP.

また、実施の形態４において、上記以外の点は、上記実施の形態３の場合と同じである。   In the fourth embodiment, points other than the above are the same as those in the third embodiment.

実施の形態５．
図８は、本発明の実施の形態５に係るＯＦＤＭ受信装置の構成を概略的に示すブロック図である。図８において、図１（実施の形態１乃至４）の構成と同一又は対応する構成には、同じ符号を付す。 Embodiment 5. FIG.
FIG. 8 is a block diagram schematically showing a configuration of an OFDM receiving apparatus according to Embodiment 5 of the present invention. In FIG. 8, the same or corresponding components as those in FIG. 1 (Embodiments 1 to 4) are denoted by the same reference numerals.

図８に示されるように、実施の形態５に係るＯＦＤＭ受信装置は、複数のアンテナ素子１１１，１１ｋ（図には、複数のアンテナ素子１１１，…，１１ｋ，…の内の２つを示す。）と、複数のチューナ部１１，１ｋ（図には、複数のチューナ部１１，…，１ｋ，…の内の２つを示す。）と、複数のＡＤ変換手段２１，２ｋ（図には、複数のＡＤ変換手段２１，…，２ｋ，…の内の２つを示す。）と、複数の時間ドメイン処理部３１，３ｋ（図には、複数の時間ドメイン処理部３１，…，３ｋ，…の内の２つを示す。）と、複数のＤＦＴ手段４１，４ｋ（図には、複数のＤＦＴ手段４１，…，４ｋ，…の内の２つを示す。）と、複数のＳＰ復調手段５１，５ｋ（図には、複数のＳＰ復調手段５１，…，５ｋ，…の内の２つを示す。）と、空間ダイバーシチ合成手段６と、誤り訂正手段７と、時間ドメイン処理部３１，３ｋから出力される信号に基づいて重み付け係数α１ｉ，…，αｋｉ，…を生成する重み付け生成手段８（以下、実施の形態５においては、符号８ｅを用いる。）とを有する。 As shown in FIG. 8, the OFDM receiver according to the fifth embodiment includes a plurality of antenna elements 11 1 , 11 k (in the figure, two of the plurality of antenna elements 11 1 ,..., 11 k,. ), A plurality of tuner units 1 1 , 1 k (the figure shows two of the plurality of tuner units 1 1 ,..., 1 k ,...), And a plurality of AD conversion means 2 1 , 2 k (shown in the figure are two of the plurality of AD conversion means 2 1 ,..., 2 k ,...) And a plurality of time domain processing units 3 1 , 3 k (shown in the figure) , Two of the plurality of time domain processing units 3 1 ,..., 3 k ,...) And a plurality of DFT means 4 1 , 4 k (in the figure, a plurality of DFT means 4 1 ,. 4 k, ... and.) shows two of the plurality of SP demodulation means 5 1, the 5 k (Fig., a plurality of SP demodulation means 5 1, ..., 5 k, Showing two of the.), And space diversity combining unit 6, the error correction unit 7, the time domain processing unit 3 1, 3 k based on the signal output from the weighting coefficient α 1i, ..., α ki ,... Are generated (hereinafter, reference numeral 8e is used in the fifth embodiment).

図８に示されるように、時間ドメイン処理部３１，３ｋはそれぞれ、ゲイン調整手段３１１，３１ｋと、１シンボル遅延手段３２１，３２ｋと、相関演算手段３３１，３３ｋと、相関強度演算手段３４１，３４ｋと、移動平均演算手段３５１，３５ｋと、ＤＦＴ窓位置検出手段３６１，３６ｋとを有する。また、図１に示されるように、ＳＰ復調手段５１，５ｋはそれぞれ、伝送路推定手段５１１，５１ｋと、ＳＰ補正手段５２１，５２ｋとを有する。 As shown in FIG. 8, the time domain processing units 3 1 and 3 k include gain adjustment means 31 1 and 31 k , 1-symbol delay means 32 1 and 32 k , and correlation calculation means 33 1 and 33 k , respectively. , Correlation strength calculating means 34 1 , 34 k , moving average calculating means 35 1 , 35 k , and DFT window position detecting means 36 1 , 36 k . As shown in FIG. 1, the SP demodulation means 5 1 and 5 k include transmission path estimation means 51 1 and 51 k and SP correction means 52 1 and 52 k , respectively.

図８に示されるように、実施の形態５に係るＯＦＤＭ受信装置は、重み付け生成手段８ｅが移動平均演算手段の出力信号ｄ（ｔ）及びＤＦＴ窓位置検出手段の出力信号ｅ（ｔ）に基づいて重み付け係数を演算する点が、相関強度演算手段の出力信号ｃ（ｔ）及びＤＦＴ窓位置検出手段の出力信号ｅ（ｔ）に基づいて重み付け係数を演算する実施の形態１に係るＯＦＤＭ受信装置と相違する。   As shown in FIG. 8, in the OFDM receiver according to the fifth embodiment, the weight generation means 8e is based on the output signal d (t) of the moving average calculation means and the output signal e (t) of the DFT window position detection means. The OFDM receiver according to the first embodiment calculates the weighting coefficient based on the output signal c (t) of the correlation strength calculating means and the output signal e (t) of the DFT window position detecting means. Is different.

図９は、実施の形態５における重み付け生成手段８ｅの構成及び空間ダイバーシチ合成手段６を示すブロック図である。図９において、図２（実施の形態１）の構成と同一又は対応する構成には、同じ符号を付す。図９に示されるように、重み付け生成手段８ｅは、複数の移動平均信号処理手段８１ｅ１，…，８１ｅｋ，…と、重み付け係数生成手段８２と、生成された重み付け係数を遅延させる遅延調整手段８３とを有する。 FIG. 9 is a block diagram showing the configuration of the weight generation means 8e and the spatial diversity combining means 6 in the fifth embodiment. In FIG. 9, the same reference numerals are given to the same or corresponding components as those in FIG. 2 (Embodiment 1). As shown in FIG. 9, the weighting generation means 8e includes a plurality of moving average signal processing means 81e 1 ,... 81e k ,..., Weighting coefficient generation means 82, and delay adjustment means for delaying the generated weighting coefficients. 83.

次に、複数の移動平均信号処理手段８１ｅ１，…，８１ｅｋ，…の代表例として移動平均信号処理手段８１ｅ１を説明する。記憶手段８３１は、移動平均演算手段３５１から出力されるＧＩ相関強度のＧＩ区間相関移動平均値ｄ１ｉ（ｔ）を記憶する。抽出位置生成手段８３２は、ＤＦＴ窓位置検出手段３６１から出力される検出結果ｅ１ｉ（ｔ）に基づく時刻情報を供給する。加算演算手段８３３及び８３４はそれぞれ、記憶装置８３１から抽出されたＧＩ区間相関移動平均値を加算する。差分演算手段８３５は、加算演算手段８３３の出力と加算演算手段８３４の出力との差分を演算する。 Next, the moving average signal processing means 81e 1 will be described as a representative example of the plurality of moving average signal processing means 81e 1 ,..., 81e k ,. Storage means 831 stores the GI interval correlation moving average value d 1i of the GI correlation strength output from the moving average calculation means 35 1 (t). Extracting position generating means 832 supplies the time information based on the detection result e 1i (t) outputted from the DFT window position detection means 36 1. Each of the addition calculation means 833 and 834 adds the GI interval correlation moving average value extracted from the storage device 831. The difference calculation means 835 calculates the difference between the output of the addition calculation means 833 and the output of the addition calculation means 834.

実施の形態１においては任意のサンプル幅で相関強度を平均していたが、実施の形態５においては、ＧＩ区間単位の幅で移動平均値を加算する。加算演算手段８３３で加算する相関強度の移動平均値は、ＤＦＴ窓先頭位置からＧＩサンプル期間だけ前の位置からＤＦＴ窓先頭位置までの移動平均値と、ＤＦＴ窓先頭位置からＤＦＴ窓先頭位置のＧＩサンプル期間後の位置までの移動平均値とを加算したものである。また、加算演算手段８３４で加算する相関強度の移動平均値は、加算演算手段８３３の演算に用いた前記範囲以外の範囲における移動平均値となる任意の時刻での平均となるように抽出し加算する。例えば、加算演算手段８３４で加算する相関強度の移動平均値は、ＤＦＴ窓先頭位置よりＧＩサンプル期間後からＤＦＴ窓先頭位置より２ＧＩサンプル期間後までの移動平均値と、ＤＦＴ窓先頭位置より２ＧＩサンプル期間後からＤＦＴ窓先頭位置より３ＧＩサンプル期間後までの移動平均値とを加算したものである。加算演算手段８３３の出力と加算演算手段８３４の出力との差分を差分演算手段８３５にて演算することによって、実施の形態１における演算範囲ｈａを（Ｄｉ−ＧＩ）から（Ｄｉ＋ＧＩ）までとし、演算範囲ｈｂを（Ｄｉ＋ＧＩ）から（Ｄｉ＋３ＧＩ）までとした場合の結果と同等となる。また、ＤＦＴ窓先頭位置より２ＧＩサンプル期間だけ前からＤＦＴ窓先頭位置よりＧＩサンプル期間前の移動平均値を使用してもよい。さらに、加算演算手段８３４を備えず、１種類の区画の移動平均値を差分演算手段８３５に直接供給してもよい。   In the first embodiment, the correlation strength is averaged with an arbitrary sample width, but in the fifth embodiment, the moving average value is added with the width of each GI section. The moving average value of the correlation strength to be added by the addition calculating means 833 is the moving average value from the position before the DFT window head position by the GI sample period to the DFT window head position, and the GI of the DFT window head position to the DFT window head position. This is the sum of the moving average up to the position after the sample period. Further, the moving average value of the correlation strength added by the addition calculating means 834 is extracted and added so as to be an average at any time that becomes a moving average value in a range other than the range used for the calculation of the adding calculating means 833. To do. For example, the moving average value of the correlation strength added by the adding operation means 834 is the moving average value from the DFT window head position after the GI sample period to the 2GI sample period after the DFT window head position, and 2 GI samples from the DFT window head position. This is the sum of the moving average value from the DFT window head position to the 3GI sample period after the period. By calculating the difference between the output of the addition calculation means 833 and the output of the addition calculation means 834 by the difference calculation means 835, the calculation range ha in the first embodiment is changed from (Di−GI) to (Di + GI). This is the same as the result when the range hb is (Di + GI) to (Di + 3GI). Alternatively, a moving average value that is two GI sample periods before the DFT window head position and GI sample periods before the DFT window head position may be used. Further, the addition calculation means 834 may not be provided, and the moving average value of one type of section may be directly supplied to the difference calculation means 835.

以上に説明した処理によって、各素子でＧＩ相関強度値を積分し平均する平均演算手段の回路規模を縮小することが可能となり、従来より優れた空間ダイバーシチ効果が期待できる。すなわち、各素子の時間ドメイン処理部におけるＧＩ区間移動平均手段結果を入力としてＤＦＴ窓先頭位置前後ＧＩサンプル長の相関強度平均に相当する値とＤＦＴ窓先頭位置前後ＧＩサンプル長以外のＧＩサンプル長単位の相関強度平均に相当する値との差分値を使用して重み付け係数を生成することで各素子でＧＩ相関強度値を積分し平均する平均演算手段の回路規模を縮小することが可能となり、且つ、従来より空間ダイバーシチ効果が期待できる。   By the processing described above, it is possible to reduce the circuit scale of the average calculation means for integrating and averaging the GI correlation strength values in each element, and a space diversity effect superior to the conventional one can be expected. That is, the value corresponding to the average correlation strength of the GI sample length before and after the DFT window head position and the GI sample length unit other than the GI sample length before and after the DFT window head position using the GI interval moving average means result in the time domain processing unit of each element as an input It is possible to reduce the circuit scale of the average calculation means for integrating and averaging the GI correlation strength values in each element by generating a weighting coefficient using a difference value from the value corresponding to the correlation strength average of In addition, a space diversity effect can be expected from the conventional method.

なお、上記説明においては、ＯＦＤＭ受信装置がＨ／Ｗとして構成された場合を説明したが、ＤＳＰ等によるファームウェアによって重み付け係数生成を行うように構成することもできる。   In the above description, the case where the OFDM receiver is configured as H / W has been described. However, weighting coefficient generation may be performed by firmware such as a DSP.

また、実施の形態５において、上記以外の点は、上記実施の形態１乃至４の場合と同じである。   In the fifth embodiment, points other than those described above are the same as those in the first to fourth embodiments.

実施の形態６．
図１０は、本発明の実施の形態６に係るＯＦＤＭ受信装置の重み付け生成手段８ｆの構成及び空間ダイバーシチ合成手段６を示すブロック図である。図１０において、図９（実施の形態５）の構成と同一又は対応する構成には、同じ符号を付す。また、実施の形態６に係るＯＦＤＭ受信装置は、図１に示される構成を有する。したがって、実施の形態６の説明においては、図１をも参照する。 Embodiment 6 FIG.
FIG. 10 is a block diagram showing the configuration of the weight generation means 8f and the spatial diversity combining means 6 of the OFDM receiving apparatus according to Embodiment 6 of the present invention. 10, components that are the same as or correspond to those in FIG. 9 (Embodiment 5) are assigned the same reference numerals. Further, the OFDM receiving apparatus according to Embodiment 6 has the configuration shown in FIG. Therefore, FIG. 1 is also referred to in the description of the sixth embodiment.

図９に示されるように、重み付け生成手段８ｆは、時間ドメイン処理部３１，３ｋの情報に基づいて重み付け係数を生成する。重み付け生成手段８ｆは、複数の移動平均信号処理手段８１ｆ１，８１ｆｋ（図には、複数の移動平均信号処理手段８１ｆ１，…，８１ｆｋ，…の内の２つを示す。）と、ゲイン調整手段８４と、重み付け係数生成手段８２と、遅延調整手段８３とを有する。複数の移動平均信号処理手段８１ｆ１，８１ｆｋはそれぞれ、記憶手段８３１と、演算範囲生成手段８３２と、加算演算手段８３３と、平均演算手段８３６とを有する。 As shown in FIG. 9, the weight generation unit 8 f generates a weighting coefficient based on information of the time domain processing units 3 1 and 3 k . The weighting generation means 8f includes a plurality of moving average signal processing means 81f 1 , 81f k (the figure shows two of the plurality of moving average signal processing means 81f 1 ,..., 81f k ,...). Gain adjustment means 84, weighting coefficient generation means 82, and delay adjustment means 83 are provided. Each of the plurality of moving average signal processing means 81f 1 and 81f k includes a storage means 831, a calculation range generation means 832, an addition calculation means 833, and an average calculation means 836.

実施の形態３に係るＯＦＤＭ受信装置は、相関強度信号処理手段８１ｃ１，８１ｃｋのそれぞれにおいて差分演算手段を備えていない点、及び、ゲイン調整手段８４を備えている点が、実施の形態１に係るＯＦＤＭ受信装置と相違する。すなわち、実施の形態１においては復調として有効な到来波の到達時間範囲における相関強度値平均と、復調として有効でない範囲での相関強度平均との差を求めて重み付け係数を生成したが、実施の形態３においては各素子の復調として有効でない範囲での相関強度平均が共通レベルになるようにゲイン調整するゲイン調整手段８４を備え、ゲイン調整された相関強度平均に基づいて重み付け係数を生成する。 The OFDM receiving apparatus according to the third embodiment is different from the first embodiment in that each of the correlation strength signal processing means 81c 1 and 81c k is not provided with a difference calculating means and a gain adjusting means 84 is provided. This is different from the OFDM receiving apparatus according to FIG. That is, in the first embodiment, the weighting coefficient is generated by obtaining the difference between the correlation strength value average in the arrival time range of the arrival wave effective as the demodulation and the correlation strength average in the range not effective as the demodulation. In the third aspect, gain adjustment means 84 is provided for adjusting the gain so that the correlation intensity average in a range that is not effective as demodulation of each element becomes a common level, and a weighting coefficient is generated based on the gain-adjusted correlation intensity average.

実施の形態３においては任意のサンプル幅で相関強度を平均していたが、ここでは、ＧＩ区間単位での幅で加算する。加算演算手段８２０で加算する相関強度の移動平均値はＤＦＴ窓先頭位置前ＧＩサンプルからＤＦＴ窓先頭位置までの移動平均値と、ＤＦＴ窓先頭位置からＤＦＴ窓先頭位置後ＧＩサンプルまでの移動平均値との加算とする。加算演算手段８２０で加算する相関強度の移動平均値は前記範囲以外での移動平均値となる任意の時刻での平均となるように抽出し加算する。例えば、ＤＦＴ窓先頭位置後ＧＩサンプルからＤＦＴ窓先頭位置後２ＧＩサンプルまでの移動平均値とＤＦＴ窓先頭位置後２ＧＩサンプルからＤＦＴ窓先頭位置後３ＧＩサンプルまでの移動平均値とを加算する。加算演算手段８２０と加算演算手段８２１との差分を差分演算手段８１４にて演算することで実施の形態３で演算範囲ｈａを（Ｄｉ−ＧＩ）から（Ｄｉ＋ＧＩ）まで、演算範囲ｈｂを（Ｄｉ＋ＧＩ）から（Ｄｉ＋３ＧＩ）までとした場合の結果と同等となる。また、ＤＦＴ窓先頭位置前２ＧＩサンプルからＤＦＴ窓先頭位置前ＧＩサンプルの移動平均値を使用してもよい。また、加算演算手段８２１を備えず１種類の区画の移動平均値を差分演算手段に供給してもよい。   In the third embodiment, the correlation strength is averaged over an arbitrary sample width, but here, the correlation strength is added in units of GI intervals. The moving average value of the correlation strength added by the adding operation means 820 is the moving average value from the GI sample before the DFT window head position to the DFT window head position, and the moving average value from the DFT window head position to the GI sample after the DFT window head position. And the addition. The moving average value of the correlation strength to be added by the addition calculating means 820 is extracted and added so as to be an average at an arbitrary time that becomes a moving average value outside the range. For example, the moving average value from the GI sample after the DFT window head position to the 2 GI sample after the DFT window head position is added to the moving average value from the 2 GI sample after the DFT window head position to the 3 GI sample after the DFT window head position. By calculating the difference between the addition calculation means 820 and the addition calculation means 821 by the difference calculation means 814, the calculation range ha is changed from (Di−GI) to (Di + GI) and the calculation range hb is set to (Di + GI) in the third embodiment. To (Di + 3GI) is equivalent to the result. Also, a moving average value from the 2GI sample before the DFT window head position to the GI sample before the DFT window head position may be used. Moreover, the moving average value of one type of section may be supplied to the difference calculating means without providing the adding calculating means 821.

以上に説明した処理によって、各素子でＧＩ相関強度値を積分し平均する平均演算手段の回路規模を縮小することが可能となり、従来より優れた空間ダイバーシチ効果が期待できる。すなわち、各素子の時間ドメイン処理部におけるＧＩ区間移動平均手段結果を入力として、各素子のＤＦＴ窓先頭位置前後ＧＩサンプル長以外のＧＩサンプル長単位の相関強度平均に相当する値が同一となるようにゲイン係数を生成し、ＤＦＴ窓先頭位置前後ＧＩサンプル長の相関強度平均に相当する値をゲイン調整することで、よりＳ／Ｎ比に近い重み付け要素になることから、ダイバーシチ合成の合成係数がより最適となるので、空間ダイバーシチ効果のより一層の向上が期待できる。   By the processing described above, it is possible to reduce the circuit scale of the average calculation means for integrating and averaging the GI correlation strength values in each element, and a space diversity effect superior to the conventional one can be expected. That is, the result corresponding to the average correlation intensity in units of GI samples other than the GI sample length before and after the DFT window head position of each element is set to be the same by using the GI interval moving average means result in the time domain processing unit of each element as an input. A gain coefficient is generated, and the gain corresponding to the average correlation strength of the GI sample lengths before and after the DFT window head position is adjusted to become a weighting factor closer to the S / N ratio. Since it becomes more optimal, further improvement of the space diversity effect can be expected.

なお、上記説明においては、ＯＦＤＭ受信装置がＨ／Ｗとして構成された場合を説明したが、ＤＳＰ等によるファームウェアによって重み付け係数生成を行うように構成することもできる。   In the above description, the case where the OFDM receiver is configured as H / W has been described. However, weighting coefficient generation may be performed by firmware such as a DSP.

また、実施の形態６において、上記以外の点は、上記実施の形態５の場合と同じである。   In the sixth embodiment, points other than the above are the same as those in the fifth embodiment.

本発明の実施の形態１乃至４に係るＯＦＤＭ受信装置の構成を概略的に示すブロック図である。It is a block diagram which shows roughly the structure of the OFDM receiver which concerns on Embodiment 1 thru | or 4 of this invention. 実施の形態１に係るＯＦＤＭ受信装置の重み付け生成手段の構成及び空間ダイバーシチ合成手段を示すブロック図である。4 is a block diagram showing a configuration of weight generation means and spatial diversity combining means of the OFDM receiving apparatus according to Embodiment 1. FIG. （ａ）乃至（ｃ）は、実施の形態１に係るＯＦＤＭ受信装置において２つの到来波がある場合のＤＦＴ窓範囲、各素子における到来波の相関強度と熱雑音を示す図である。(A) thru | or (c) is a figure which shows the DFT window range in case there exist two incoming waves in the OFDM receiver which concerns on Embodiment 1, the correlation intensity | strength of the incoming wave in each element, and a thermal noise. 本発明の実施の形態２に係るＯＦＤＭ受信装置の重み付け生成手段の構成及び空間ダイバーシチ合成手段を示すブロック図である。It is a block diagram which shows the structure of the weight production | generation means of the OFDM receiver which concerns on Embodiment 2 of this invention, and a space diversity synthetic | combination means. （ａ）乃至（ｃ）は、実施の形態２に係るＯＦＤＭ受信装置において３つの到来波がある場合のＤＦＴ窓範囲、フィルタ係数、各素子における到来波の相関強度と熱雑音を示す図である。(A) thru | or (c) is a figure which shows the DFT window range, filter coefficient, correlation intensity | strength of the incoming wave in each element, and thermal noise in case there exist three incoming waves in the OFDM receiver which concerns on Embodiment 2. FIG. . 本発明の実施の形態３に係るＯＦＤＭ受信装置の重み付け生成手段の構成及び空間ダイバーシチ合成手段を示すブロック図である。It is a block diagram which shows the structure of the weight production | generation means of the OFDM receiver which concerns on Embodiment 3 of this invention, and a space diversity synthetic | combination means. 本発明の実施の形態４に係るＯＦＤＭ受信装置の重み付け生成手段の構成及び空間ダイバーシチ合成手段を示すブロック図である。It is a block diagram which shows the structure of the weight production | generation means of the OFDM receiver which concerns on Embodiment 4 of this invention, and a space diversity synthetic | combination means. 本発明の実施の形態５及び６に係るＯＦＤＭ受信装置の構成を概略的に示すブロック図である。It is a block diagram which shows roughly the structure of the OFDM receiver which concerns on Embodiment 5 and 6 of this invention. 実施の形態５に係るＯＦＤＭ受信装置の重み付け生成手段の構成及び空間ダイバーシチ合成手段を示すブロック図である。FIG. 10 is a block diagram showing a configuration of weight generation means and spatial diversity combining means of an OFDM receiving apparatus according to Embodiment 5. 本発明の実施の形態６に係るＯＦＤＭ受信装置の重み付け生成手段の構成及び空間ダイバーシチ合成手段を示すブロック図である。It is a block diagram which shows the structure of the weight production | generation means of the OFDM receiver which concerns on Embodiment 6 of this invention, and a space diversity synthetic | combination means. 従来のＯＦＤＭ受信装置の構成を概略的に示すブロック図である。It is a block diagram which shows schematically the structure of the conventional OFDM receiver. （ａ）乃至（ｃ）は、ＯＦＤＭ受信装置における到来波、到来波の相関強度、及び相関強度のＧＩ長移動平均結果との関係を示す図である。(A) thru | or (c) is a figure which shows the relationship with the GI length moving average result of the incoming wave in an OFDM receiver, the correlation strength of an incoming wave, and correlation strength. （ａ）乃至（ｃ）は、ＯＦＤＭ受信装置における２つの到来波、相関強度、及び演算範囲の関係を示す図である。(A) thru | or (c) is a figure which shows the relationship between two incoming waves, correlation intensity | strength, and a calculation range in an OFDM receiver.

符号の説明Explanation of symbols

１１，１ｋ チューナ部、 ２１，２ｋ ＡＤ変換手段、 ３１，３ｋ 時間ドメイン処理部、 ４１，４ｋ ＤＦＴ手段、 ５１，５ｋ ＳＰ復調手段、 ６ 空間ダイバーシチ合成手段、 ７ 誤り訂正手段、 ８，８ａ，８ｂ，８ｃ，８ｄ，８ｅ，８ｆ 重み付け生成手段、 １１１，１１ｋ アンテナ素子、 ３１１，３１ｋ ゲイン調整手段、 ３２１，３２ｋ １シンボル遅延手段、 ３３１，３３ｋ 相関演算手段、 ３４１，３４ｋ 相関強度演算手段、 ３５１，３５ｋ 移動平均演算手段、 ３６１，３６ｋ ＤＦＴ窓位置検出手段、 ５１１，５１ｋ 伝送路推定手段、 ５２１，５２ｋ ＳＰ補正手段、 ８１１ 記憶手段、 ８１２ 演算範囲生成手段、 ８１３，８１４ 平均演算手段、 ８１５ 差分演算手段、 ８１６ フィルタ演算手段、 ８１７ 平均演算手段、 ８３１ 記憶手段、 ８３２ 抽出位置生成手段、 ８３３，８３４ 加算演算手段、 ８３５ 差分演算手段、 ８３６ 平均演算手段、 ８１ａ１，８１ａｋ，８１ｂ１，８１ｂｋ，８１ｃ１，８１ｃｋ，８１ｄ１，８１ｄｋ 相関強度信号処理手段、 ８１ｅ１，８１ｅｋ，８１ｆ１，８１ｆｋ 移動平均信号処理手段、 ８２ 重み付け係数生成手段、 ８３ 遅延調整手段、 ８４ ゲイン調整手段、 ｋ 素子番号（又はアンテナ素子からＳＰ復調手段までの構成の系統を示す番号）、 ｉ シンボル番号、 ｍ キャリア番号、 ｔ 時刻、 ａ（ｔ） ゲイン調整された受信信号、 ｂ（ｔ） 相関演算結果、 ｃ（ｔ），ｃ１ｉ（ｔ），ｃｋｉ（ｔ） 相関強度演算結果、 ｄ（ｔ），ｄ１ｉ（ｔ），ｄｋｉ（ｔ） 相関強度のＧＩサンプル長移動平均結果、 ｅ（ｔ），ｅ１ｉ（ｔ），ｅｋｉ（ｔ） ＤＦＴ窓位置検出結果、 ｒｉｍ 該当するキャリアのＤＦＴ結果、 ＣＡＲｉｍ 該当するキャリアの既知情報、 Ｈｉｍ 伝送路推定結果、 ｆ１ｉｍ，ｆｋｉｍ ＳＰ復調結果、 ｇ１ｉｍ，ｇｋｉｍ 伝送路推定振幅情報、 ＤＩＶＯＵＴｉｍ 該当するキャリアのダイバーシチ合成結果、 ｈａ，ｈｂ 演算範囲、 α１ｉ，αｋｉ 重み付け係数。
1 1 , 1 k tuner unit, 2 1 , 2 k AD conversion unit, 3 1 , 3 k time domain processing unit, 4 1 , 4 k DFT unit, 5 1 , 5 k SP demodulation unit, 6 spatial diversity combining unit, 7 error correcting means 8, 8a, 8b, 8c, 8d, 8e, 8f weighting generating means, 11 1 , 11 k antenna elements, 31 1 , 31 k gain adjusting means, 32 1 , 32 k 1 symbol delay means, 33 1 , 33 k correlation calculation means, 34 1 , 34 k correlation strength calculation means, 35 1 , 35 k moving average calculation means, 36 1 , 36 k DFT window position detection means, 51 1 , 51 k transmission path estimation means, 52 1, 52 k SP correction means, 811 storage unit, 812 operation range generation unit, 813 and 814 average calculation unit, 815 differential operation circuit, 816 a filter operation unit, 8 7 average computing unit, 831 memory unit, 832 extraction position generating means, 833, 834 addition operation unit, 835 differential operation circuit, 836 average calculation unit, 81a 1, 81a k, 81b 1, 81b k, 81c 1, 81c k, 81d 1 , 81d k correlation strength signal processing means, 81e 1 , 81e k , 81f 1 , 81f k moving average signal processing means, 82 weighting coefficient generation means, 83 delay adjustment means, 84 gain adjustment means, k element number (or antenna) Number indicating the system of the configuration from the element to the SP demodulation means), i symbol number, m carrier number, t time, a (t) gain adjusted received signal, b (t) correlation calculation result, c (t), c 1i (t), c ki (t) Correlation strength calculation result, d (t), d 1i (t), d ki (t) GI of correlation strength Sample length moving average result, e (t), e 1i (t), e ki (t) DFT window position detection result, r im DFT result of corresponding carrier, CAR im known information of corresponding carrier, H im transmission path Estimation result, f 1im , f kim SP demodulation result, g 1im , g kim transmission path estimation amplitude information, DIVOUT im diversity combining result of corresponding carrier, ha, hb calculation range, α 1i , α ki weighting coefficient.

Claims (10)

ＯＦＤＭ信号を受信する複数系統の受信手段と、
前記受信されたＯＦＤＭ信号の離散フーリエ変換の窓位置をシンボル毎に求める時間ドメイン処理手段、前記窓位置に基づいてＯＦＤＭ信号を離散フーリエ変換する離散フーリエ変換手段、及び前記離散フーリエ変換手段の出力信号を前記受信されたＯＦＤＭ信号に含まれるスキャッタードパイロット信号に基づいてキャリア毎に伝送路推定し、該推定結果を用いて補正するスキャッタードパイロット復調手段を含む複数系統の受信信号演算手段と、
前記複数系統の受信信号演算手段の複数の前記時間ドメイン処理手段からの信号に基づいて各系統及び各シンボルについての重み付け係数を生成する重み付け生成手段と、
前記複数系統の受信信号演算手段の複数の前記スキャッタードパイロット復調手段からの信号と前記重み付け生成手段から出力される重み付け係数に基づいて空間ダイバーシチ合成を行う空間ダイバーシチ合成手段と
を有することを特徴とするＯＦＤＭ受信装置。 A plurality of receiving means for receiving an OFDM signal;
Time domain processing means for obtaining a discrete Fourier transform window position of the received OFDM signal for each symbol, discrete Fourier transform means for performing discrete Fourier transform on the OFDM signal based on the window position, and an output signal of the discrete Fourier transform means A plurality of systems of received signal calculation means including scattered pilot demodulation means for estimating a transmission path for each carrier based on a scattered pilot signal included in the received OFDM signal and correcting using the estimation result ,
Weight generation means for generating weighting coefficients for each system and each symbol based on signals from the plurality of time domain processing means of the plurality of received signal calculation means;
Spatial diversity combining means for performing spatial diversity combining based on signals from the plurality of scattered pilot demodulation means and weighting coefficients output from the weight generation means of the plurality of systems of received signal calculation means. An OFDM receiver. 前記時間ドメイン処理手段は、
前記受信されたＯＦＤＭ信号の振幅を調整する第１のゲイン調整手段と、
前記第１のゲイン調整手段から出力されるＯＦＤＭ信号を１シンボル期間遅延させる遅延手段と、
前記受信されたＯＦＤＭ信号と前記遅延手段により遅延したＯＦＤＭ信号の相関を演算する相関演算手段と、
前記相関演算手段からの出力信号に基づいて相関強度を演算する相関強度演算手段と、
前記相関強度演算手段からの出力信号に基づいて移動平均を演算する移動平均演算手段と、
前記移動平均演算手段からの出力信号に基づいて離散フーリエ変換の前記窓位置を検出する離散フーリエ変換窓位置検出手段と
を有することを特徴とする請求項１に記載のＯＦＤＭ受信装置。 The time domain processing means includes
First gain adjusting means for adjusting the amplitude of the received OFDM signal;
Delay means for delaying the OFDM signal output from the first gain adjustment means by one symbol period;
Correlation calculating means for calculating a correlation between the received OFDM signal and the OFDM signal delayed by the delay means;
Correlation strength calculating means for calculating a correlation strength based on an output signal from the correlation calculating means;
A moving average calculating means for calculating a moving average based on an output signal from the correlation strength calculating means;
2. The OFDM receiver according to claim 1, further comprising: a discrete Fourier transform window position detecting unit that detects the window position of the discrete Fourier transform based on an output signal from the moving average calculating unit. 前記重み付け生成手段に入力される前記時間ドメイン処理手段からの信号は、前記相関強度演算手段からの出力信号と前記離散フーリエ変換窓位置検出手段からの出力信号を含むことを特徴とする請求項２に記載のＯＦＤＭ受信装置。   3. The signal from the time domain processing means input to the weight generation means includes an output signal from the correlation strength calculation means and an output signal from the discrete Fourier transform window position detection means. The OFDM receiver according to 1. 前記重み付け生成手段は、
前記離散フーリエ変換窓位置検出手段から出力される離散フーリエ変換窓先頭位置を含む第１の演算範囲と前記第１の演算範囲以外の範囲内で設定される第２の演算範囲とを生成する演算範囲生成手段、前記第１の演算範囲における前記相関強度の平均を演算する第１の平均演算手段、前記第２の演算範囲における前記相関強度の平均を演算する第２の平均演算手段、及び前記第１の平均演算手段の出力信号と前記第２の平均演算手段の出力信号との差分を演算する差分演算手段を含む複数の相関強度信号処理手段と、
前記複数の相関強度信号処理手段の複数の前記差分演算手段の出力信号に基づいて各系統及び各シンボルについての前記重み付け係数を生成する重み付け係数生成手段と
を有することを特徴とする請求項３に記載のＯＦＤＭ受信装置。 The weight generation means includes:
Calculation for generating a first calculation range including a discrete Fourier transform window head position output from the discrete Fourier transform window position detecting means and a second calculation range set within a range other than the first calculation range. Range generating means, first average calculating means for calculating the average of the correlation strength in the first calculation range, second average calculating means for calculating the average of the correlation strength in the second calculation range, and A plurality of correlation strength signal processing means including difference calculation means for calculating a difference between an output signal of the first average calculation means and an output signal of the second average calculation means;
The weighting coefficient generating means for generating the weighting coefficient for each system and each symbol based on the output signals of the plurality of difference calculating means of the plurality of correlation strength signal processing means. The OFDM receiver according to the description. 前記重み付け生成手段は、
前記離散フーリエ変換窓位置検出手段から出力される離散フーリエ変換窓先頭位置を含む第１の演算範囲と前記第１の演算範囲以外の範囲内で設定される第２の演算範囲とを生成する演算範囲生成手段、前記第１の演算範囲における前記相関強度を前記離散フーリエ変換窓先頭位置で最大となり前記離散フーリエ変換窓先頭位置から離れると小さくなるフィルタ係数を用いて演算するフィルタ演算手段、前記第２の演算範囲における前記相関強度の平均を演算する平均演算手段、及び前記フィルタ演算手段の出力信号と前記平均演算手段の出力信号との差分を演算する差分演算手段を含む複数の相関強度信号処理手段と、
前記複数の相関強度信号処理手段の複数の前記差分演算手段からの出力信号に基づいて各系統及び各シンボルについての前記重み付け係数を生成する重み付け係数生成手段と
を有することを特徴とする請求項３に記載のＯＦＤＭ受信装置。 The weight generation means includes:
Calculation for generating a first calculation range including a discrete Fourier transform window head position output from the discrete Fourier transform window position detecting means and a second calculation range set within a range other than the first calculation range. Range generation means, filter calculation means for calculating the correlation strength in the first calculation range using a filter coefficient that is maximized at the start position of the discrete Fourier transform window and decreases when leaving the start position of the discrete Fourier transform window; A plurality of correlation strength signal processing including an average calculation means for calculating an average of the correlation strengths in two calculation ranges, and a difference calculation means for calculating a difference between an output signal of the filter calculation means and an output signal of the average calculation means Means,
4. Weighting coefficient generation means for generating the weighting coefficient for each system and each symbol based on output signals from the plurality of difference calculation means of the plurality of correlation strength signal processing means. The OFDM receiver according to 1. 前記重み付け生成手段は、
前記離散フーリエ変換窓位置検出手段から出力される離散フーリエ変換窓先頭位置を含む第１の演算範囲と前記第１の演算範囲以外の範囲内で設定される第２の演算範囲とを生成する演算範囲生成手段、前記第１の演算範囲における前記相関強度の平均を演算する第１の平均演算手段、及び前記第２の演算範囲における前記相関強度の平均を演算する第２の平均演算手段を含む複数の相関強度信号処理手段と、
前記複数の相関強度信号処理手段の複数の前記第２の平均演算手段の出力信号のそれぞれを同レベルにするように複数の前記第１の平均演算手段の出力信号をゲイン調整する第２のゲイン調整手段と、
前記第２のゲイン調整手段からの出力信号に基づいて各系統及び各シンボルについての前記重み付け係数を生成する重み付け係数生成手段と
を有することを特徴とする請求項３に記載のＯＦＤＭ受信装置。 The weight generation means includes:
Calculation for generating a first calculation range including a discrete Fourier transform window head position output from the discrete Fourier transform window position detecting means and a second calculation range set within a range other than the first calculation range. A range generation means, a first average calculation means for calculating the average of the correlation strength in the first calculation range, and a second average calculation means for calculating the average of the correlation strength in the second calculation range. A plurality of correlation strength signal processing means;
A second gain for gain adjustment of the output signals of the plurality of first average calculation means so that the output signals of the plurality of second average calculation means of the plurality of correlation strength signal processing means are at the same level. Adjusting means;
The OFDM receiving apparatus according to claim 3, further comprising weighting coefficient generation means for generating the weighting coefficient for each system and each symbol based on an output signal from the second gain adjustment means. 前記重み付け生成手段は、
前記離散フーリエ変換窓位置検出手段から出力される離散フーリエ変換窓先頭位置を含む第１の演算範囲と前記第１の演算範囲以外の範囲内で設定される第２の演算範囲とを生成する演算範囲生成手段、前記第１の演算範囲における前記相関強度を前記離散フーリエ変換窓先頭位置で最大となり前記離散フーリエ変換窓先頭位置から離れると小さくなるフィルタ係数を用いて演算するフィルタ演算手段、及び前記第２の演算範囲における前記相関強度の平均を演算する平均演算手段を含む複数の相関強度信号処理手段と、
前記複数の相関強度信号処理手段の複数の前記平均演算手段の出力信号のそれぞれを同レベルにするように複数の前記フィルタ演算手段の出力信号をゲイン調整する第２のゲイン調整手段と、
前記第２のゲイン調整手段からの出力信号に基づいて各系統及び各シンボルについての前記重み付け係数を生成する重み付け係数生成手段と
を有することを特徴とする請求項３に記載のＯＦＤＭ受信装置。 The weight generation means includes:
Calculation for generating a first calculation range including a discrete Fourier transform window head position output from the discrete Fourier transform window position detecting means and a second calculation range set within a range other than the first calculation range. Range generation means, filter calculation means for calculating the correlation strength in the first calculation range using a filter coefficient that is maximum at the discrete Fourier transform window head position and decreases when the discrete Fourier transform window head position is separated from the discrete Fourier transform window head position; and A plurality of correlation strength signal processing means including an average calculation means for calculating an average of the correlation strengths in a second calculation range;
Second gain adjusting means for adjusting the gain of the output signals of the plurality of filter calculating means so that each of the output signals of the plurality of average calculating means of the plurality of correlation intensity signal processing means has the same level;
The OFDM receiving apparatus according to claim 3, further comprising weighting coefficient generation means for generating the weighting coefficient for each system and each symbol based on an output signal from the second gain adjustment means. 前記重み付け生成手段に入力される前記時間ドメイン処理手段からの信号は、前記移動平均演算手段の出力信号と前記離散フーリエ変換窓位置検出手段の出力信号を含むことを特徴とする請求項２に記載のＯＦＤＭ受信装置。   3. The signal from the time domain processing means input to the weight generation means includes an output signal from the moving average calculation means and an output signal from the discrete Fourier transform window position detection means. OFDM receiver. 前記重み付け生成手段は、
前記離散フーリエ変換窓位置検出手段から出力される離散フーリエ変換窓先頭位置を含む第１の抽出範囲と前記第１の抽出範囲以外の範囲内で設定される第２の抽出範囲とにおいて抽出位置を生成する抽出位置生成手段、前記第１の抽出範囲における各抽出位置における前記移動平均値を加算する第１の加算演算手段、前記第２の抽出範囲における各抽出位置における前記移動平均値を加算する第２の加算演算手段、及び前記第１の加算演算手段の出力信号と前記第２の加算演算手段の出力信号との差分を演算する差分演算手段を含む複数の移動平均信号処理手段と、
前記複数の移動平均処理手段の複数の前記差分演算手段からの出力信号に基づいて各系統及び各シンボルについての前記重み付け係数を生成する重み付け係数生成手段と
を有することを特徴とする請求項８に記載のＯＦＤＭ受信装置。 The weight generation means includes:
Extraction positions are determined in a first extraction range including a discrete Fourier transform window head position output from the discrete Fourier transform window position detection means and a second extraction range set within a range other than the first extraction range. Extraction position generation means for generating, first addition calculation means for adding the moving average values at each extraction position in the first extraction range, and adding the moving average values at each extraction position in the second extraction range A plurality of moving average signal processing means including a second addition calculation means, and a difference calculation means for calculating a difference between an output signal of the first addition calculation means and an output signal of the second addition calculation means;
The weighting coefficient generation means for generating the weighting coefficient for each system and each symbol based on output signals from the plurality of difference calculation means of the plurality of moving average processing means. The OFDM receiver according to the description. 前記重み付け生成手段は、
前記離散フーリエ変換窓位置検出手段から出力される離散フーリエ変換窓先頭位置を含む第１の抽出範囲と前記第１の抽出範囲以外の範囲内で設定される第２の抽出範囲とにおいて抽出位置を生成する抽出位置生成手段、前記第１の抽出範囲における各抽出位置における前記移動平均値を加算する加算演算手段、及び前記第２の抽出範囲における各抽出位置における前記移動平均値の平均値を演算する平均値演算手段を含む複数の移動平均信号処理手段と、
前記複数の移動平均処理手段の複数の前記平均演算手段の出力信号のそれぞれを同レベルにするように前記加算演算手段の出力信号をゲイン調整する第２のゲイン調整手段と、
前記第２のゲイン調整手段からの出力信号に基づいて各系統及び各シンボルについての前記重み付け係数を生成する重み付け係数生成手段と
を有することを特徴とする請求項８に記載のＯＦＤＭ受信装置。 The weight generation means includes:
Extraction positions are determined in a first extraction range including a discrete Fourier transform window head position output from the discrete Fourier transform window position detection means and a second extraction range set within a range other than the first extraction range. An extraction position generating means for generating, an addition calculating means for adding the moving average values at each extraction position in the first extraction range, and calculating an average value of the moving average values at each extraction position in the second extraction range A plurality of moving average signal processing means including an average value calculating means,
A second gain adjusting means for adjusting the gain of the output signal of the addition calculating means so that each of the output signals of the plurality of average calculating means of the plurality of moving average processing means has the same level;
9. The OFDM receiver according to claim 8, further comprising weighting coefficient generation means for generating the weighting coefficient for each system and each symbol based on an output signal from the second gain adjustment means.
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Cited By (4)

* Cited by examiner, † Cited by third party
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JP2007258819A (en) * 2006-03-20 2007-10-04 Casio Comput Co Ltd Ofdm receiver, ofdm reception method, and terrestrial digital receiver
JP2007267198A (en) * 2006-03-29 2007-10-11 Casio Comput Co Ltd Ofdm reception apparatus, ofdm reception method and terrestrial digital reception device
JP2010226696A (en) * 2009-02-25 2010-10-07 Kyocera Corp Communication apparatus and received power measuring method
CN109613334A (en) * 2018-12-04 2019-04-12 上海司南卫星导航技术股份有限公司 A kind of Frequency Estimation device, terminal and computer-readable medium

Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2007258819A (en) * 2006-03-20 2007-10-04 Casio Comput Co Ltd Ofdm receiver, ofdm reception method, and terrestrial digital receiver
JP2007267198A (en) * 2006-03-29 2007-10-11 Casio Comput Co Ltd Ofdm reception apparatus, ofdm reception method and terrestrial digital reception device
JP4692761B2 (en) * 2006-03-29 2011-06-01 カシオ計算機株式会社 OFDM receiving apparatus, OFDM receiving method, and terrestrial digital receiving apparatus
JP2010226696A (en) * 2009-02-25 2010-10-07 Kyocera Corp Communication apparatus and received power measuring method
CN109613334A (en) * 2018-12-04 2019-04-12 上海司南卫星导航技术股份有限公司 A kind of Frequency Estimation device, terminal and computer-readable medium
CN109613334B (en) * 2018-12-04 2022-02-25 上海司南卫星导航技术股份有限公司 Frequency estimation device, terminal and computer readable medium

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