JP2004180368A - Estimate correction method for induction motor magnetic flux estimator - Google Patents
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Abstract
Description
【0001】
【発明の属する技術分野】
本発明は、誘導電動機の速度制御方法に関するものである。
【0002】
【従来の技術】
従来の電動機制御方法における位置・速度などのセンサレス制御方法では、誘導電動機磁束推定器はk・Ts秒(kは制御ステップの順を表す整数で、Tsは制御ステップの周期)の時点で誘導電動機に供給される電流のうち少なくとも2相分のステータ電流(例えば、U相電流isu(k)、V相電流isv(k))、及びステータ側電圧(例えば、U相電圧Vsu(k)、V相電圧Vsv(k))を検出して、3相−2相変換より、通常ベクトル制御で用いられているα−β相座標系に、この電流、電圧を座標変換して、それぞれk・Ts秒時のα相電流isα(k)、β相電流isβ(k)、α相電圧Vα(k)、β相電圧Vβ(k)を算出する。
【0003】
センサレス制御の従来の技術1として挙げる、電学論D、VOL.3.NO.11.p.954−960(1991)に示される誘導電動機の状態方程式から導かれる磁束推定器の式(1)、及び速度推定式(2)、位相推定式(3)は次の通りである。
【数1】
【0004】
【数2】
次に、これらを逐次形に展開した式を示す。
【0005】
【数3】
【0006】
上式に従って動作する誘導電動機制御回路用のオブザーバを用いて、制御ステップkの推定値等により、制御ステップ(k+1)の推定値を算出する。すなわち、制御ステップkの制御ループで推定されたk・Ts秒時のα相電流推定値isαest(k)と、α相電流isα(k)との推定誤差、
isα(k)−isαest(k)、
と、β相電流推定値isβestとβ相電流isβ(k)との推定誤差、
isβ(k)−isβest(k)、
を用いて、(k+1)・Ts秒時の推定値isαest(k+1)、isβest(k+1)、φrαest(k+1)、φrβest(k+1)、を算出する。
【0007】
この従来の技術1の場合は、電圧Vα、Vβを用いるための電圧センサを、なお必要とするが、コストの削減の目的から電圧センサを削減する方法に、従来の技術2として、α相電圧指令値Vα*(k)、β相電圧指令値Vβ*(k)をVα(k)、Vβ(k)の代わりに使用することが考えられる。
例えば、特許文献1の「誘導電動機の速度センサレスベクトル制御方式」は、電圧センサを削除して、補償用の誤差電圧を出力するq相、d相の電流コントローラに、q相電流指令iq *とq相電流iq(及び、d相電流指令id *とd相電流id)の偏差を入力して、出力されるq相の誤差電圧Δv1qをPI演算することにより速度推定を行うものである。
【0008】
【特許文献1】
特開H05−292777号公報([0060]〜[0064]、図1)
【0009】
【発明が解決しようとする課題】
しかしながら、上記従来の技術においては、従来の技術1の場合はコストの削減と言う見地から電圧センサの削減が課題であり、これを実現する方法として、従来の技術2の場合は、指令値と実際値との間に誤差が生ずることがあり、ロータ鎖交磁束推定値、速度推定値に誤差を生ずるという問題があった。
【0010】
そこで、本発明は、速度センサや電圧センサを有していないセンサレスの場合の速度推定を、速度推定値の補正を行って推定誤差を減少させることによって精度の良い速度推定を可能にする誘導電動機磁束推定器の推定値補正方法を提供することを目的としている。
【0011】
【課題を解決するための手段】
上記目的を達成するため、請求項1に記載の発明は、誘導電動機の2次磁束のベクトルの方向をd相とし、それに直交する相をq相とするd−q座標軸に制御系構成の基準座標軸であるγ−δ座標軸を一致するように制御する誘導電動機の制御方法により、誘導電動機の3相の固定子電流のうち少なくとも2相分の固定子電流isu(k)、isv(k)を検出し、α−β相座標系に変換したα相電流のisα(k)、β相電流isβ(k)と、α−β相座標系に変換された電圧指令値Vα*(k)、Vβ*(k)を磁束推定器に入力し電流推定値isαest(k+1)、isβest(k+1)、2次側鎖交磁束推定値φrαest(k+1)、φrβest(k+1)、速度推定値ωrest(k)、位相推定値θ1est(k)を推定する磁束推定方法において、前記位相推定値θ1est(k)と電流検出値isα、isβ(k)よりd−q変換したq相電流iq(k)を計算し、q相電流iq(k)とq相電流指令iqref(k)の差に比例ゲインを乗じた補正量を速度推定値に加えることにより磁束推定器の推定値の精度を向上させることを特徴としている。
また、請求項2に記載の発明は、誘導電動機の2次磁束のベクトルの方向をd相とし、それに直交する相をq相とするd−q座標軸に制御系構成の基準座標軸であるγ−δ座標軸を一致するように制御する誘導電動機の制御方法により、誘導電動機の3相の固定子電流のうち少なくとも2相分の固定子電流isu(k)、isv(k)を検出し、α−β相座標系に変換したα相電流のisα(k)、β相電流isβ(k)と、α−β相座標系に変換された電圧指令値Vα*(k)、Vβ*(k)を磁束推定器に入力し電流推定値isαest(k+1)、isβest(k+1)、2次側鎖交磁束推定値φrαest(k+1)、φrβest(k+1)、速度推定値ωrest(k)、位相推定値θ1est(k)を推定する磁束推定方法において、前記位相推定値θ1est(k)と電流検出値isα、isβ(k)よりd−q変換したq相電流iq(k)を計算し、q相電流iq(k)とq相電流指令iqref(k)の差に比例ゲインを乗じ、積分した補正量を速度推定値に加えることにより磁束推定器の推定値の精度を向上させることを特徴としている。
【0012】
この誘導電動機磁束推定器の推定値補正方法によれば、観測量isα(k)、isβ(k)と、位相推定量θ1est(k)より、下記の(7)式、
【数4】
【0013】
で求めたq相電流iq(k)とq相電流指令iqref(k)よりiq(k)とiqref(k)の差が下記の(8)式、
【数5】
【0014】
のように、実際の速度と速度推定量との誤差に相当することを利用して、下記の速度推定器を構成する。
【数6】
この(9)式の補正を加える前の速度推定値ω´rest(k)(従来技術の速度推定値)と、実際の速度ωr(k)に誤差がある時、(9)式の第2項(比例ゲインによる補正量)と、第3項(積分項による補正量)に補正量が生じ、補正を加えた速度推定値ω´rest(k)を実際の速度ωr(k)に一致させることにより、速度推定精度を向上させることができる。
【0015】
【発明の実施の形態】
以下、本発明の実施の形態について図を参照して説明する。
図1は本発明の誘導電動機磁束推定器の推定値補正方法が適用される誘導電動機の制御システムのブロック図である。
図1において、1は外部からの速度指令ωr*と、後述の速度補正器からの速度推定値ω´restが入力する速度コントローラである。
【0016】
2はq相電圧指令vq *を出力するq相コントローラであり、3はd相電圧指令vd *を出力するd相コントローラである。4は電圧指令vq *、vd *より静止座標系のα相電圧指令vα*、β相電圧指令vβ*に換算して出力するベクトル制御回路である。5はインバータ回路で、6は電動機2相電流isu、isvを入力して静止座標系のα相電流isαとβ相電流isβに相変換を行う相変換器であり、7は磁束推定器でα相、β相電流推定値、ロータ鎖交磁束推定値、速度推定値、位相推定値を演算・出力する。8は速度補正器で速度推定値ωrestを補正してω´restを出力する。9はd−q座標変換器である。10は誘導電動機である。
【0017】
つぎに動作について説明する。
先ず、外部と速度補正器8から、速度指令ωr *と速度推定値ω´restとが、それぞれ速度コントローラ1へ入力し、速度コントローラ1は回転座標系のq相電流指令iqrefを出力する。q相電流コントローラ2はq相電流指令iqrefと、q相電流値iqを入力してq相電圧指令vqrefを出力する。
一方、d相電流指令idrefとd相電流値idがd相電流コントローラ3へ入力して、d相電流コントローラ3はd相電圧指令vdrefを出力する。電圧指令vqrefとvdrefは、ベクトル制御演算回路4に入力して、磁束推定器7より出力される位相θ1estによりα相電圧指令vα*、β相電圧指令vβ*に換算して出力される。
【0018】
相変換器6は誘導電動機10のU相電流isuとV相電流isvとを入力し、静止座標系のα相電流isαとβ相電流isβに変換する。磁束推定器7はこれらのα相電流isα、β相電流isβと、電圧指令vα*、vβ*とを入力し、内蔵している(1)式、(2)式、(3)式により、α相、β相電流推定値isαest、isβest、ロータ鎖交磁束推定値φrαest、φrβest、及び、速度推定値ωrest、位相推定値θ1estを、演算し出力する。
d−q座標変換器9は相変換器6の出力のα相電流isαと、β相電流isβと、位相推定値θ1estを入力として、d相電流idとq相電流iqを出力する。
【0019】
以上は、センサレス・ベクトル制御として公知の部分であるが、本発明では、更に、速度補正器8を設けて、磁束推定器7から出力される従来の速度推定値ωrestを補正して、正確な速度推定値ω´restを速度コントローラ1へ供給する。具体的には、速度補正器8は、d−q座標変換器9で(7)式により求めたq相電流iq(k)と、速度コントローラ1からのq相電流指令iqrefを入力し、その差が(8)式のように、実際の速度と速度推定量との誤差に相当するとして誤差検出を行い、誤差が検出されたら(9)式の第2項による偏差値に比例ゲインkqを乗じた補正量を加算して補正するか、更に、第3項による積分量を加算して正確に補正することによって、速度推定値ωrestの誤差が補正されて実速度と一致した速度推定値ω´restにより速度制御が実施されるので、追従性の高い安定なセンサレス・ベクトル制御が可能になる。
また、本実施の形態は、すべり周波数を用いる誘導電動機のセンサレス速度制御だけではなく、γ−δ座標系などの設定によって、永久磁石形同期電動機のセンサレス速度制御にもそのまま適用できる。
【0020】
【発明の効果】
以上説明したように、本発明によれば、従来センサレス・ベクトル制御に用いられている磁束推定器より演算出力される誘導電動機の速度推定値に対して、更に、速度補正器を設けて補正することによって、速度推定値に誤差が生じても常に正確な速度推定値が得られるので、誘導電動機の速度制御精度、トルク制御精度を高めることができる。
【図面の簡単な説明】
【図1】本発明の誘導電動機磁束推定器の推定値補正方法が適用される制御システムのブロック図である。
【符号の説明】
1 速度コントローラ
2 q相電流コントローラ
3 d相電流コントローラ
4 ベクトル制御回路
5 インバータ回路
6 相変換器
7 磁束推定器
8 速度補正器
9 d−q座標変換器
10 電動機[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a speed control method for an induction motor.
[0002]
[Prior art]
In the sensorless control method such as position and speed in the conventional motor control method, the induction motor magnetic flux estimator uses the induction motor flux estimator at k · Ts seconds (k is an integer representing the order of the control steps, and Ts is the cycle of the control steps). , A stator current (for example, a U-phase current isu (k), a V-phase current isv (k)) and a stator-side voltage (for example, a U-phase voltage Vsu (k) ), V-phase voltage V sv (k)), and the current and voltage are coordinate-transformed from the three-phase-two-phase conversion to the α-β-phase coordinate system normally used in vector control. alpha phase current when each k · Ts sec i s α (k), β-phase current i s β (k), α-phase voltage Vα (k), to calculate the beta phase voltage V? (k).
[0003]
Electron theory D, VOL. 3. NO. 11. p. 954-960 (1991), the magnetic flux estimator equation (1), the speed estimation equation (2), and the phase estimation equation (3) derived from the state equation of the induction motor are as follows.
(Equation 1)
[0004]
(Equation 2)
Next, an expression that expands these into a sequential form is shown.
[0005]
[Equation 3]
[0006]
Using the observer for the induction motor control circuit that operates according to the above equation, the estimated value of the control step (k + 1) is calculated based on the estimated value of the control step k and the like. That is, the control step k alpha phase current estimated value i s alpha est when k · Ts seconds are estimated in the control loop of the (k), the estimated error between the alpha phase current i s α (k),
i s α (k) -i s α est (k),
When the estimation error of the beta phase current estimated value i s beta est and beta-phase current i s β (k),
i s β (k) -i s β est (k),
Using, (k + 1) when · Ts seconds estimate i s α est (k + 1 ), i s β est (k + 1), φ r α est (k + 1), φ r β est (k + 1), is calculated.
[0007]
In the case of the prior art 1, a voltage sensor for using the voltages Vα and Vβ is still required. However, the method of reducing the voltage sensors for the purpose of cost reduction is described in the related art 2 as the α phase voltage. The command value Vα * (k) and the β-phase voltage command value Vβ * (k) may be used instead of Vα (k) and Vβ (k).
For example, the “speed sensorless vector control method of an induction motor” in Patent Document 1 discloses a q-phase current command iq * to a q-phase and d-phase current controller that outputs a compensation error voltage by removing a voltage sensor . that the speed estimated by the q-phase current i q (and, d-phase current command i d * and the d-phase current id) enter the deviation, the error voltage Delta] v 1q q-phase output to PI operation It is.
[0008]
[Patent Document 1]
JP H05-292777 ([0060] to [0064], FIG. 1)
[0009]
[Problems to be solved by the invention]
However, in the above-mentioned conventional technology, the reduction of the voltage sensor is a problem from the viewpoint of cost reduction in the case of the conventional technology 1, and as a method for realizing this, in the case of the conventional technology 2, the command value and the command value are reduced. There is a problem that an error may occur between the actual value and the estimated value of the rotor linkage flux and the estimated speed.
[0010]
Accordingly, the present invention provides an induction motor that enables accurate speed estimation by correcting the speed estimation value and reducing the estimation error by performing speed estimation in the case of a sensorless system having no speed sensor or voltage sensor. It is an object of the present invention to provide a method for correcting an estimated value of a magnetic flux estimator.
[0011]
[Means for Solving the Problems]
In order to achieve the above object, the invention according to claim 1 is based on a d-q coordinate axis in which the direction of a vector of a secondary magnetic flux of an induction motor is a d-phase and a phase orthogonal thereto is a q-phase. the control method for an induction motor is controlled so as to match the gamma-[delta] axes are coordinate axes, at least two phases of the stator current i su of the stator current of the three phase induction motor (k), i sv (k ) detects, alpha-beta of converted alpha phase current phase coordinate system i s α (k), β-phase current i s beta and (k), α-β-phase coordinate converted voltage command value based Vα * (k), Vβ * ( k) the input to the flux estimator current estimated value i s α est (k + 1 ), i s β est (k + 1), 2 primary flux linkage estimate φ r α est (k + 1 ), φ r β est (k + 1), the speed estimated value ω rest (k), phase estimate θ 1est ( ) In flux estimation method of estimating and calculating the current value detected phase estimate θ 1est (k) i s α , i s β (k) from d-q converted q-phase current i q (k), characterized in that to improve the accuracy of the estimated value of the magnetic flux estimator by adding a correction amount obtained by multiplying a proportional gain to the difference between the q-phase current i q (k) and q-phase current command i qref (k) to the speed estimated value And
The invention according to claim 2 is a d-q coordinate axis in which the direction of the vector of the secondary magnetic flux of the induction motor is d-phase and the phase orthogonal thereto is the q-phase, which is the reference coordinate axis of the control system configuration. According to the control method of the induction motor that controls the δ coordinate axes to coincide, the stator currents i su (k) and isv (k) of at least two phases among the three-phase stator currents of the induction motor are detected, alpha-beta phase coordinate system converted alpha phase current i s α (k), and beta-phase current i s β (k), α -β -phase coordinate converted voltage command value based Vα * (k), V? * (k) the input to the flux estimator current estimated value i s α est (k + 1 ), i s β est (k + 1), 2 primary flux linkage estimate φ r α est (k + 1 ), φ r β est (k + 1), the speed estimated value ω rest (k), the magnetic flux to estimate the phase estimate θ 1est (k) In the constant method, wherein the current detection value phase estimate θ 1est (k) i s α , i s β (k) of from d-q converted q-phase current i q (k) calculated, the q-phase current i q It is characterized in that the accuracy of the estimated value of the magnetic flux estimator is improved by multiplying the difference between (k) and the q-phase current command i qref (k) by a proportional gain and adding the integrated correction amount to the speed estimated value.
[0012]
According to estimates the correction method of the induction motor magnetic flux estimator, the observation quantity i s alpha (k), and i s beta (k), the phase estimator θ 1est (k), the following equation (7),
(Equation 4)
[0013]
In the obtained q-phase current i q (k) and the difference is (8) below the q-phase current command i qref (k) from i q (k) and i qref (k),
(Equation 5)
[0014]
The speed estimator described below is configured by utilizing the fact that it corresponds to the error between the actual speed and the speed estimation amount.
(Equation 6)
When there is an error between the speed estimated value ω ′ rest (k) (prior art speed estimated value) before the correction of the expression (9) and the actual speed ω r (k), the expression (9) A correction amount is generated in two terms (a correction amount by a proportional gain) and a third term (a correction amount by an integration term), and the corrected speed estimated value ω ′ rest (k) is converted into an actual speed ω r (k). By making them match, the speed estimation accuracy can be improved.
[0015]
BEST MODE FOR CARRYING OUT THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is a block diagram of an induction motor control system to which an estimation value correction method of an induction motor magnetic flux estimator according to the present invention is applied.
In FIG. 1, reference numeral 1 denotes a speed controller to which a speed command ωr * from the outside and a speed estimate ω ′ rest from a speed corrector described later are input.
[0016]
Reference numeral 2 denotes a q-phase controller that outputs a q-phase voltage command v q *, and
[0017]
Next, the operation will be described.
First, a speed command ω r * and a speed estimated value ω ′ rest are input to the speed controller 1 from the outside and the speed corrector 8, and the speed controller 1 outputs a q-phase current command iqref of the rotating coordinate system. . q-phase current controller 2 and q-phase current command i qref, enter the q-phase current value i q outputs the q-phase voltage command v qref.
On the other hand, d-phase current command i dref and d-phase current value i d is input to the d-phase
[0018]
Phase converter 6 inputs the U-phase current i su and V-phase current i sv of the
and alpha phase current i s alpha of the output of the d-q coordinate
[0019]
The above is a known part of the sensorless vector control. However, in the present invention, a speed corrector 8 is further provided to correct the conventional speed estimated value ω rest output from the magnetic flux estimator 7 to obtain an accurate value. The speed estimation value ω ′ rest is supplied to the speed controller 1. Specifically, the speed corrector 8 includes a d-q coordinate converter 9 (7) was determined by equation q-phase current i q (k), and inputs the q-phase current command i qref from the speed controller 1 The error is detected assuming that the difference corresponds to the error between the actual speed and the estimated speed as shown in the equation (8), and if an error is detected, the proportional gain is added to the deviation value by the second term of the equation (9). By adding a correction amount multiplied by k q to correct the error, or further adding an integral amount according to the third term to correct the error accurately, the error in the speed estimated value ω rest was corrected to match the actual speed. Since the speed control is performed based on the speed estimated value ω ′ rest, stable sensorless vector control with high tracking performance can be performed.
Further, the present embodiment can be applied to not only sensorless speed control of an induction motor using a slip frequency but also sensorless speed control of a permanent magnet synchronous motor by setting a γ-δ coordinate system or the like.
[0020]
【The invention's effect】
As described above, according to the present invention, the speed estimation value of the induction motor calculated and output from the magnetic flux estimator conventionally used for sensorless vector control is further corrected by providing a speed corrector. Thus, even if an error occurs in the speed estimation value, an accurate speed estimation value is always obtained, so that the speed control accuracy and the torque control accuracy of the induction motor can be improved.
[Brief description of the drawings]
FIG. 1 is a block diagram of a control system to which an estimation value correction method of an induction motor magnetic flux estimator of the present invention is applied.
[Explanation of symbols]
Reference Signs List 1 speed controller 2 q-phase current controller 3 d-phase current controller 4 vector control circuit 5 inverter circuit 6 phase converter 7 magnetic flux estimator 8
Claims (2)
前記位相推定値θ1est(k)と電流検出値isα、isβ(k)よりd−q変換したq相電流iq(k)を計算し、q相電流iq(k)とq相電流指令iqref(k)の差に比例ゲインを乗じた補正量を速度推定値に加えることにより磁束推定器の推定値の精度を向上させることを特徴とする誘導電動機磁束推定器の推定値補正方法。An induction motor that controls so that the direction of the vector of the secondary magnetic flux of the induction motor is d phase, and the γ-δ coordinate axis, which is the reference coordinate axis of the control system configuration, is coincident with the dq coordinate axis that has a phase orthogonal to it as the q phase. , The stator currents i su (k) and isv (k) of at least two phases among the three-phase stator currents of the induction motor are detected, and the α phase converted to the α-β phase coordinate system is detected. current i s alpha (k), and beta-phase current i s beta (k), the voltage command value is converted into alpha-beta phase coordinate system V.alpha * (k), the input V? * (k) is the magnetic flux estimator the current estimated value i s α est (k + 1 ), i s β est (k + 1), 2 primary flux linkage estimate φ r α est (k + 1 ), φ r β est (k + 1), the speed estimated value ω rest ( k), in the magnetic flux estimation method for estimating the phase estimation value θ 1est (k),
Wherein the current detection value phase estimate θ 1est (k) i s α , i s β (k) of from d-q converted q-phase current i q (k) is calculated, the q-phase current i q (k) Estimation of an induction motor magnetic flux estimator characterized by improving the accuracy of the magnetic flux estimator estimated value by adding a correction amount obtained by multiplying the difference of the q-phase current command i qref (k) by a proportional gain to the speed estimated value. Value correction method.
前記位相推定値θ1est(k)と電流検出値isα、isβ(k)よりd−q変換したq相電流iq(k)を計算し、q相電流iq(k)とq相電流指令iqref(k)の差に比例ゲインを乗じ、積分した補正量を速度推定値に加えることにより磁束推定器の推定値の精度を向上させることを特徴とする誘導電動機磁束推定器の推定値補正方法。An induction motor that controls so that the direction of the vector of the secondary magnetic flux of the induction motor is d phase, and the γ-δ coordinate axis, which is the reference coordinate axis of the control system configuration, is coincident with the dq coordinate axis that has a phase orthogonal to it as the q phase. , The stator currents i su (k) and isv (k) of at least two phases among the three-phase stator currents of the induction motor are detected, and the α phase converted to the α-β phase coordinate system is detected. current i s alpha (k), and beta-phase current i s beta (k), the voltage command value is converted into alpha-beta phase coordinate system V.alpha * (k), the input V? * (k) is the magnetic flux estimator the current estimated value i s α est (k + 1 ), i s β est (k + 1), 2 primary flux linkage estimate φ r α est (k + 1 ), φ r β est (k + 1), the speed estimated value ω rest ( k), in the magnetic flux estimation method for estimating the phase estimation value θ 1est (k),
Wherein the current detection value phase estimate θ 1est (k) i s α , i s β (k) of from d-q converted q-phase current i q (k) is calculated, the q-phase current i q (k) An induction motor magnetic flux estimator characterized by improving the accuracy of an estimated value of a magnetic flux estimator by multiplying a difference of a q-phase current command i qref (k) by a proportional gain and adding an integrated correction amount to a speed estimated value. Estimated value correction method.
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EP2573934A4 (en) * | 2010-05-20 | 2017-08-30 | Kabushiki Kaisha Toshiba, Inc. | Control device without a rotation sensor |
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EP2573934A4 (en) * | 2010-05-20 | 2017-08-30 | Kabushiki Kaisha Toshiba, Inc. | Control device without a rotation sensor |
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