JP2004173496A - Controller for induction motor - Google Patents

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JP2004173496A
JP2004173496A JP2004010578A JP2004010578A JP2004173496A JP 2004173496 A JP2004173496 A JP 2004173496A JP 2004010578 A JP2004010578 A JP 2004010578A JP 2004010578 A JP2004010578 A JP 2004010578A JP 2004173496 A JP2004173496 A JP 2004173496A
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resistance
induction motor
polyphase
voltage
control device
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JP3771239B2 (en
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Toshiyuki Kaitani
敏之 貝谷
Tetsuaki Nagano
鉄明 長野
Akira Imanaka
晶 今中
Yasuhiro Shiraishi
康裕 白石
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Mitsubishi Electric Corp
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Abstract

<P>PROBLEM TO BE SOLVED: To improve control performance, by reducing an error due to approximating mutual inductance of a circuit to be open. <P>SOLUTION: The inductive motor controller comprises a power arithmetic means for calculating effective power P and reactive power Q corresponding to a test power source of frequency f; a first arithmetic means for obtaining secondary resistance R2d and secondary leakage inductance L2d from an operation expression obtained by an L-type equivalent circuit, from a phase voltage peak value Vp and a primary angular frequency ω corresponding to the frequency f, the effective power P and the reactive power Q, and previously measured primary resistance R1; and a second arithmetic means for obtaining secondary resistance R2 and secondary leakage inductance L by substituting the secondary resistance R2d and the secondary leakage inductance L2d, obtained by the first arithmetic means and previously measured secondary self inductance LL2 for each of operation expressions of secondary resistance R2 and secondary leakage inductance L obtained by a T-type equivalent circuit. <P>COPYRIGHT: (C)2004,JPO

Description

この発明は、多相インバータにより試験電源を供給して多相誘導電動機の１次、２次抵抗及び漏れインダクタンスといった電気的定数を測定し、この測定結果を駆動制御のパラメータに設定して多相誘導電動機を駆動制御する誘導電動機制御装置に関する。   According to the present invention, a test power is supplied by a polyphase inverter to measure electric constants such as primary and secondary resistances and leakage inductances of a polyphase induction motor, and the measurement result is set as a drive control parameter to set the polyphase induction motor. The present invention relates to an induction motor control device that drives and controls an induction motor.

誘導電動機を高精度に制御するには、誘導電動機の電気的定数である１次、２次の巻線抵抗、１次、２次漏れインダクタンス、相互インダクタンスの値が必要になる場合がある。これらの電気的定数を測定し誘導電動機制御装置に設定する機能を有する従来技術の１つとして、図１２に示すものがある。   In order to control the induction motor with high accuracy, the values of the primary and secondary winding resistances, primary and secondary leakage inductances, and mutual inductances, which are the electrical constants of the induction motor, may be required. FIG. 12 shows one of the prior arts having a function of measuring these electric constants and setting them in an induction motor control device.

図１２は特開平７−３２５１３２号公報（特許文献１）に記載された従来の誘導電動機制御装置の要部システム構成を示すものであり、１はインバータ、２は誘導電動機、３は電流検出器、４は電圧検出器、５は磁束トルク制御手段、６は無負荷試験手段、７は直流試験手段、８（１）〜８（ｎ）は単相試験手段、９は定数計算手段、１０は選択器、１１は設定記憶手段である。インバータ１は選択器１０出力のスイッチング信号を入力しそのスイッチング信号に応じて動作させられ、インバータ１により誘導電動機２に電圧を印加する。   FIG. 12 shows the main system configuration of a conventional induction motor control device described in Japanese Patent Application Laid-Open No. 7-325132 (Patent Document 1), wherein 1 is an inverter, 2 is an induction motor, and 3 is a current detector. 4, a voltage detector, 5 a magnetic flux torque control means, 6 a no-load test means, 7 a DC test means, 8 (1) to 8 (n) single-phase test means, 9 a constant calculation means, 10 a The selector 11 is a setting storage means. The inverter 1 receives a switching signal output from the selector 10 and is operated according to the switching signal. The inverter 1 applies a voltage to the induction motor 2.

次に、この誘導電動機制御装置の２次抵抗及び漏れインダクタンスを測定する動作を説明する。選択器１０は複数の単相試験手段８（１）〜８（ｎ）の出力スイッチング信号を順次選択し、インバータ１に出力する。誘導電動機２は停止した状態で３相入力端子のうちの２つの端子間に単相交流電圧が印加される。ここで、ｎ個の単相試験手段８（１）〜８（ｎ）では誘導電動機２に印加される電圧の周波数がそれぞれ異なっているものとする。各々の単相試験手段８（１）〜８（ｎ）は、電流検出器３と電圧検出器４により検出した電流と電圧を入力してそれらの基本波の大きさと位相を求め、それらの関係から１次と２次の漏れインダクタンスの和（Ｌ０１１＋Ｌ０２１）〜（Ｌ０１ｎ＋Ｌ０２ｎ）と、１次と２次の巻線抵抗の和（Ｒ１１＋Ｒ２１）〜（Ｒ１ｎ＋Ｒ２ｎ）とを求め、その電圧の周波数（Ｆ１〜Ｆｎ）とともに定数計算手段９に出力する。ここで、１次漏れインダクタンスと２次漏れインダクタンスは等しいと仮定し、それぞれの単相試験手段８（１）〜８（ｎ）が出力する漏れインダクタンスの和の半分をそれぞれの単相試験手段８（１）〜８（ｎ）の漏れインダクタンス（Ｌ１〜Ｌｎ）とする。   Next, an operation of measuring the secondary resistance and the leakage inductance of the induction motor control device will be described. The selector 10 sequentially selects the output switching signals of the plurality of single-phase test means 8 (1) to 8 (n) and outputs the output switching signals to the inverter 1. With the induction motor 2 stopped, a single-phase AC voltage is applied between two of the three-phase input terminals. Here, it is assumed that the frequency of the voltage applied to the induction motor 2 is different among the n single-phase test means 8 (1) to 8 (n). Each of the single-phase test means 8 (1) to 8 (n) receives the current and the voltage detected by the current detector 3 and the voltage detector 4, obtains the magnitude and phase of the fundamental wave, and determines the relationship between them. From the sum of the primary and secondary leakage inductances (L011 + L021) to (L01n + L02n) and the sum of the primary and secondary winding resistances (R11 + R21) to (R1n + R2n). ) And outputs it to the constant calculation means 9. Here, it is assumed that the primary leakage inductance is equal to the secondary leakage inductance, and half of the sum of the leakage inductances output by the respective single-phase test means 8 (1) to 8 (n) is divided by the respective single-phase test means 8 Let (1) to 8 (n) be the leakage inductances (L1 to Ln).

さらに、印加電圧の周波数を変数として漏れインダクタンスの周波数変化特性を、（ｍ≦ｎ−１）を満足するｍ次多項式を次式とする。   Further, the frequency change characteristic of the leakage inductance using the frequency of the applied voltage as a variable, and an m-th order polynomial satisfying (m ≦ n−1) is expressed by the following equation.

Ｌ＝Ａ・Ｆ（べきｍ）＋Ｂ・Ｆ（べきｍ−１）＋・・・＋Ｚ ・・（１）
ここで、（１）式に周波数（Ｆ１〜Ｆｎ）において測定された漏れインダクタンス（Ｌ１〜Ｌｎ）を代入し、未定係数法により、係数Ａ、Ｂ、Ｚを求めることができる。そして、周波数０Ｈｚの値を採用するため、零次分Ｚのみを算出すればよく計算が非常に簡単である。このｍ次多項式の零次分を漏れインダクタンスの推定真値Ｌとする。同様に、周波数（Ｆ１〜Ｆｎ）にて測定された２次巻線抵抗（Ｒ２１〜Ｒ２ｎ）を用いて、ｍ次多項式の零次分を２次巻線抵抗の真値Ｒ２とする。このように、測定を行うのは誘導電動機２次導体の表皮効果による２次抵抗、漏れインダクタンスの測定誤差を減らすためである。 L = AF (power m) + BF (power m-1) + ... + Z (1)
Here, the coefficients A, B, and Z can be obtained by the undetermined coefficient method by substituting the leakage inductances (L1 to Ln) measured at the frequencies (F1 to Fn) into the equation (1). Since the value of the frequency 0 Hz is adopted, only the zero-order component Z needs to be calculated, and the calculation is very simple. The zero-order part of this m-th order polynomial is defined as an estimated true value L of the leakage inductance. Similarly, using the secondary winding resistances (R21 to R2n) measured at the frequencies (F1 to Fn), the zero-order component of the m-th order polynomial is defined as the true value R2 of the secondary winding resistance. The measurement is performed in order to reduce the measurement error of the secondary resistance and the leakage inductance due to the skin effect of the secondary conductor of the induction motor.

ここで、表皮効果の影響についてもう少し詳細に説明する。図１３は表皮効果により２次抵抗が２次周波数によりどのように変化するかを示す説明図であり、深溝型の２次導体形状を持つ３．７ＫＷ誘導電動機を例として示した。ここで、図１３（ａ）は二次周波数（Ｈｚ）と２次抵抗（Ω）との関係を示す説明図であり、図１３（ｂ）は図１３（ａ）の低周波部分の２次抵抗側レンジを拡大した説明図であり、太実線で示される真値が示すように２次抵抗は２次周波数の増加に伴い表皮効果の影響で大きくなる。図中２次近似、３次近似、４次近似と示してあるのは、多項式近似により求めた曲線であることを示している。図１３（ｃ）は多項式近似により求めた曲線の問題点の説明図である。図１３（ｂ）で判るように多項式近似の次数を上げるほど真値に近く推定できる。しかし、その分多くの周波数での測定が必要となる。この例では４次近似程度でかなり真値からの誤差は少なくなっている。   Here, the effect of the skin effect will be described in more detail. FIG. 13 is an explanatory diagram showing how the secondary resistance changes according to the secondary frequency due to the skin effect, and shows a 3.7 KW induction motor having a deep groove type secondary conductor shape as an example. Here, FIG. 13A is an explanatory diagram showing the relationship between the secondary frequency (Hz) and the secondary resistance (Ω), and FIG. FIG. 4 is an explanatory diagram in which the resistance side range is enlarged, and as indicated by a true value indicated by a thick solid line, the secondary resistance increases due to the skin effect as the secondary frequency increases. In the drawing, the quadratic approximation, the cubic approximation, and the quadratic approximation indicate that the curves are obtained by polynomial approximation. FIG. 13C is an explanatory diagram of a problem of a curve obtained by polynomial approximation. As can be seen from FIG. 13B, the higher the order of the polynomial approximation, the closer to the true value can be estimated. However, measurement at many frequencies is necessary. In this example, the error from the true value is considerably reduced in the fourth order approximation.

また、図１３（ｃ）に示す４次近似#1は４次近似で各周波数で誤差なく測定できた場合を示しており、４次近似#2は１ポイントの周波数（図中２２Ｈｚ）での測定に０．００５Ω誤差を持った場合の推定結果を示している。図から判るように０Ｈｚの２次抵抗値は１ポイントの周波数の測定誤差により大きく誤差が増す。このように、多項式近似によると測定された値のわずかな誤差が測定されていない領域の抵抗推定値に非常に大きな誤差となって影響することが判る。   Further, the fourth-order approximation # 1 shown in FIG. 13 (c) shows a case where the measurement can be performed without error at each frequency by the fourth-order approximation, and the fourth-order approximation # 2 is at one point frequency (22 Hz in the figure). The estimation result when the measurement has a 0.005Ω error is shown. As can be seen from the figure, the error of the secondary resistance at 0 Hz greatly increases due to a measurement error of the frequency at one point. As described above, according to the polynomial approximation, it can be seen that a slight error in the measured value affects the resistance estimation value in an unmeasured region as a very large error.

また、通常運転状態では滑りが小さいので２次周波数はわずか数Ｈｚ程度である。この例で使用した誘導電動機では定格負荷時に４Ｈｚとなる。   In addition, the secondary frequency is only about several Hz in the normal operation state because the slip is small. In the induction motor used in this example, the frequency is 4 Hz at the rated load.

従って通常運転状態で必要とされる２次抵抗は数Ｈｚ程度の値である。   Therefore, the secondary resistance required in the normal operation state is a value of about several Hz.

しかし、誘導電動機の等価回路はよく知られている図１４に示され、数Ｈｚ程度の電圧印加ではほとんど２次側に電流が流れることはなく、相互インダクタンスＭを通って流れてしまい、２次抵抗を精度良く測定することができなくなるために、通常運転状態より高い周波数から必要とされる周波数を推定することが必要となる。   However, the well-known equivalent circuit of the induction motor is shown in FIG. 14, and when a voltage of about several Hz is applied, almost no current flows to the secondary side, but flows through the mutual inductance M. Since the resistance cannot be measured with high accuracy, it is necessary to estimate a required frequency from a frequency higher than the normal operation state.

以上をまとめると、
通常運転時に必要とされる低周波数での２次抵抗はその周波数で直接測定することはできず通常運転時より高い周波数で求めた２次抵抗測定値を用いて通常運転時の２次抵抗を推定しなければならない。 To summarize the above,
The secondary resistance at low frequency required during normal operation cannot be directly measured at that frequency, and the secondary resistance during normal operation is calculated using the measured secondary resistance at a higher frequency than during normal operation. I have to estimate.

多項式近似により通常運転時に必要とされる周波数での２次抵抗値の推定は可能であるが、精度を上げるために近似する次数は４次以上、すなわち周波数５点以上の測定が必要となる。   Although it is possible to estimate the secondary resistance value at the frequency required during normal operation by polynomial approximation, the order of approximation must be fourth or higher, that is, measurement at five or more frequencies is required to improve the accuracy.

またそれぞれの周波数における測定精度は極めて高くなければならず、もし誤差を含むと通常運転周波数での２次抵抗値を推定する際、大きな誤差を生じる。   In addition, the measurement accuracy at each frequency must be extremely high, and if an error is included, a large error occurs when estimating the secondary resistance value at the normal operation frequency.

以上、２次抵抗について述べたが、漏れインダクタンスに関してもまったく同じことがいえる。   Although the secondary resistance has been described above, the same can be said for the leakage inductance.

次に、図１５は、特開平６−９８５９５号公報に記載された従来の誘導電動機制御装置のブロック図であり、この公報には、特に単相印加による１次＋２次の合成抵抗（Ｒ１＋Ｒ２）及び合成漏れインダクタンス（Ｌ１＋Ｌ２）の測定方法が開示されている。   FIG. 15 is a block diagram of a conventional induction motor control device described in Japanese Unexamined Patent Publication No. Hei 6-98595. In this publication, a combined primary-secondary and secondary-order resistance (R1 + R2) by single-phase application is particularly described. And a method of measuring the combined leakage inductance (L1 + L2).

図１５において、２１は交流電源、２２は整流回路、２３は平滑コンデンサ、２４はインバータ、２５は誘導電動機、２６は電流検出器である。   In FIG. 15, 21 is an AC power supply, 22 is a rectifier circuit, 23 is a smoothing capacitor, 24 is an inverter, 25 is an induction motor, and 26 is a current detector.

２７はＰＷＭ信号を発生するゲート回路、２８は通常運転時は速度指令ωｒに追従するよう制御を行う速度センサレスベクトル制御、２９は正弦波変調信号を作り、これによりゲート回路２７を介してインバータ２４を動作させ、交流励磁電圧により誘導電動機２５に交流電流を流す単相交流励磁処理、３０は有効パワー分電流Ｉｑ、及び無効パワー分電流Ｉｄ演算処理、３１は１次、２次合成抵抗及び合成漏れインダクタンス演算処理である。３２は制御回路であり符号２８乃至符号３１を付した構成を含む。ここで、有効パワー分電流Ｉｑ、及び無効パワー分電流Ｉｄ演算処理３０は、Ｕ相の電動機電流ｉｕと励磁電圧ベクトルの固定座標からの回転位相指令をθとすると、有効パワー分電流Ｉｑである１．４１４・ｓｉｎθ・ｉｕを１次周波数の１周期区間で、任意のサンプル周期毎に積算し、積算回数で除算して求め、無効パワー分電流Ｉｄである−１．４１４・ｃｏｓθ・ｉｕを１次周波数の１周期区間で、任意のサンプル周期毎に積算し、積算回数で除算して求める。   27 is a gate circuit for generating a PWM signal, 28 is a speed sensorless vector control for performing control so as to follow the speed command ωr during normal operation, and 29 is a sine wave modulation signal which is generated by the inverter 24 through the gate circuit 27. Is operated, and a single-phase AC excitation process for passing an AC current to the induction motor 25 by an AC excitation voltage, 30 is a process for calculating an effective power component current Iq and a reactive power component current Id, 31 is a primary and secondary combined resistance and a combined This is a leakage inductance calculation process. Reference numeral 32 denotes a control circuit, which includes components denoted by reference numerals 28 to 31. Here, the effective power component current Iq and the reactive power component current Id calculation processing 30 are the effective power component current Iq, where θ is the rotation phase command from the fixed coordinates of the U-phase motor current iu and the excitation voltage vector. 1.414 · sin θ · iu is integrated at an arbitrary sampling period in one cycle section of the primary frequency and is obtained by dividing by the number of times of integration, and the reactive power component current Id, ie, −1.414 · cos θ · iu, is obtained. In one period section of the primary frequency, integration is performed at an arbitrary sampling period, and the result is obtained by dividing by the number of times of integration.

そして、１次、２次合成抵抗及び合成漏れインダクタンス演算処理は、１次、２次合成抵抗（Ｒ１＋Ｒ２）と１次、２次合成漏れインダクタンス（Ｌ１＋Ｌ２）をインバータの１次周波数指令ω１及び１次電圧指令値Ｖｃ１より次式で求める。   Then, the primary and secondary combined resistance and combined leakage inductance calculation processing calculates the primary and secondary combined resistance (R1 + R2) and the primary and secondary combined leakage inductance (L1 + L2) into the primary frequency command ω1 and primary of the inverter. It is obtained from the voltage command value Vc1 by the following equation.

（Ｒ１＋Ｒ２）＝Ｖｃ１・Ｉｑ／１．５（Ｉｄ2＋Ｉｑ2）・・（２）
（Ｌ１＋Ｌ２）＝Ｖｃ１・Ｉｄ／１．５ω１（Ｉｄ2＋Ｉｑ2）・・（３）
このようにして、図１５に示す従来の誘導電動機制御装置は単相印加により１次、２次合成抵抗及び漏れインダクタンスを求める。 (R1 + R2) = Vc1 · Iq / 1.5 (Id 2 + Iq 2) ·· (2)
(L1 + L2) = Vc1 · Id / 1.5ω1 (Id 2 + Iq 2) ·· (3)
In this way, the conventional induction motor control device shown in FIG. 15 obtains the primary and secondary combined resistances and the leakage inductance by single-phase application.

しかし、この演算は相互インダクタンスＭを開放として近似していることによる誤差を生じる。ここでその誤差について説明する。図１４に示された１相分等価回路にて１次側から流入する電流は相互インダクタンスＭと２次の漏れインダクタンス、２次抵抗の直列回路とに分流する。ここで、相互インダクタンスＭを開放として近似するとＭに流れている電流はすべて２次漏れインダクタンスと、２次抵抗の直列回路に流れるとして計算することになる。この相互インダクタンスを開放として近似することによる誤差はＲ１＝０．４Ω、Ｒ２＝０．３Ω、Ｍ＝６２ｍＨ、Ｌ１＝Ｌ２＝１．６ｍＨとすると、２次抵抗値が約５％ほど低く演算されることになる。   However, this calculation causes an error due to approximating the mutual inductance M as being open. Here, the error will be described. In the one-phase equivalent circuit shown in FIG. 14, the current flowing from the primary side is divided into a mutual inductance M, a secondary leakage inductance, and a series circuit of a secondary resistor. Here, when the mutual inductance M is approximated as being open, all the current flowing in M is calculated assuming that the current flows in a series circuit of the secondary leakage inductance and the secondary resistance. The error caused by approximating the mutual inductance as open is R1 = 0.4Ω, R2 = 0.3Ω, M = 62mH, L1 = L2 = 1.6mH, and the secondary resistance is calculated to be about 5% lower. Will be.

図１６は特開平４−３６４３８４号公報（特許文献２）に記載された従来の誘導電動機制御装置を示すブロック図であり、特に、起動時に電動機の１次抵抗Ｒ１と２次抵抗Ｒ２を推定する技術が開示されている。   FIG. 16 is a block diagram showing a conventional induction motor control device described in Japanese Patent Application Laid-Open No. 4-364384 (Patent Document 2). In particular, a primary resistance R1 and a secondary resistance R2 of a motor are estimated at the time of startup. The technology is disclosed.

図１６において、直流電圧信号発生器５１は起動信号ＳＴが零から立ち上がった直後から磁束指令Ψ* が５０％まで立ち上がるまでの一定時間、直流電圧を一時的にスイッチ５２を介してＰＷＭインバータ４１に印加し、ＰＷＭインバータ４１がパルス幅制御で平均的に電動機４２に直流電圧を印加する。次に、起動電流検出手段５７は、ＰＷＭインバータ４１より直流電圧を電動機４２に印加し始めてから磁束指令Ψ* が５０％まで立ち上がるまでの一定時間後電動機４２の１相の電流（以降、起動電流と称す。）を電流検出器４４により検出し、ローパスフィルタ５３及びスイッチ５４を介して抵抗推定演算器５５に出力する。次に、抵抗推定手段５５は、電動機４２の温度が基準温度状態における電動機４２の１次抵抗Ｒ１ｎと２次抵抗Ｒ２ｎ及び電動機４２の温度が基準温度状態における起動電流検出手段５７の出力起動電流Ｉｎを記憶している基準値記憶器５６と、電動機４２の温度が規定されていない状態における起動電流検出手段５７の出力の起動電流Ｉおよび基準値記憶器５６の出力である基準温度における１次抵抗Ｒ１ｎ、２次抵抗Ｒ２ｎおよびＩｎを入力し、
Ｒ１＝｛Ｋｒ１（Ｉｎ−１）／Ｉ＋１｝Ｒ１ｎ ・・・（４）
Ｒ２＝｛Ｋｒ２（Ｉｎ−１）／Ｉ＋１｝Ｒ２ｎ ・・・（５）
（４）、（５）式の演算（Ｋｒ１、Ｋｒ２は補正係数）により電動機４２の１次抵抗Ｒ１と２次抵抗Ｒ２とを推定する。 In FIG. 16, the DC voltage signal generator 51 applies the DC voltage to the PWM inverter 41 via the switch 52 temporarily for a certain period of time from immediately after the start signal ST rises from zero to when the magnetic flux command Ψ * rises to 50%. The PWM inverter 41 applies a DC voltage to the motor 42 on average by pulse width control. Next, the starting current detecting means 57 detects a one-phase current of the motor 42 (hereinafter referred to as starting current) after a certain time from the start of applying the DC voltage from the PWM inverter 41 to the motor 42 until the magnetic flux command Ψ * rises to 50%. ) Is detected by the current detector 44 and output to the resistance estimation calculator 55 via the low-pass filter 53 and the switch 54. Next, the resistance estimating means 55 determines the output starting current In of the starting current detecting means 57 when the temperature of the electric motor 42 is the reference temperature state and the primary resistance R1n and the secondary resistance R2n of the electric motor 42 and the temperature of the electric motor 42 are the reference temperature state. And a primary resistance at a reference temperature which is an output of the starting current detection means 57 and an output of the reference value storage 56 when the temperature of the electric motor 42 is not specified. R1n, a secondary resistor R2n and In,
R1 = {Kr1 (In-1) / I + 1} R1n (4)
R2 = {Kr2 (In-1) / I + 1} R2n (5)
The primary resistance R1 and the secondary resistance R2 of the electric motor 42 are estimated by the calculations of the equations (4) and (5) (Kr1 and Kr2 are correction coefficients).

しかし、この図１６に示す誘導電動機制御装置では、ＰＷＭインバータ４１より直流電圧を電動機４２に印加し始めてから磁束指令Ψ* が５０％まで立ち上がるまでの一定時間後の電動機４２の１相の起動電流を電流検出器４４により検出して抵抗を推定するものである。また、１次抵抗Ｒ１及び２次抵抗Ｒ２は同一であることが前提となっているので、推定された１次抵抗Ｒ１及び２次抵抗Ｒ２は標準温度における１次抵抗Ｒ１ｎ及び２次抵抗Ｒ２ｎよりは実際に近いといった程度であり、また１次抵抗と２次抵抗との温度がいつも一定でなければ誤差を持つといった問題がある。 However, in the induction motor control device shown in FIG. 16, the starting current of one phase of the motor 42 after a certain time from when the DC voltage is applied from the PWM inverter 41 to the motor 42 until the magnetic flux command Ψ * rises to 50%. Is detected by the current detector 44 to estimate the resistance. Further, since it is assumed that the primary resistance R1 and the secondary resistance R2 are the same, the estimated primary resistance R1 and secondary resistance R2 are smaller than the primary resistance R1n and the secondary resistance R2n at the standard temperature. Is close to the actual one, and there is a problem that if the temperatures of the primary resistance and the secondary resistance are not always constant, there is an error.

特開平７−３２５１３２号公報JP-A-7-325132 特開平４−３６４３８４号公報JP-A-4-364384

上述のような従来の誘導電動機制御装置では、次のような問題がある。
（１）２次抵抗、漏れインダクタンスの測定精度を上げるには表皮効果を十分に考慮する必要があるが、従来の考慮方法では誤差が大きく変化する。 The conventional induction motor control device as described above has the following problems.
(1) To increase the measurement accuracy of the secondary resistance and the leakage inductance, it is necessary to sufficiently consider the skin effect, but the error greatly changes in the conventional method.

従来の表皮効果考慮方法は多項式で近似するというものであったが、多項式ではその次数を高くしなければ精度よく推定できなかった。次数を高くすることは、測定する周波数を増やすことにつながるので、測定に時間がかかり多項式近似の演算も複雑になる。   The conventional method for considering the skin effect is to approximate with a polynomial, but in the case of a polynomial, it cannot be accurately estimated unless its order is increased. Since an increase in the order leads to an increase in the frequency to be measured, the measurement takes time and the calculation of the polynomial approximation becomes complicated.

また、多項式近似はもともとその測定ポイントを正しく通る曲線を求めるもので、２次抵抗推定のように例えば１０Ｈｚからの６０Ｈｚまでの測定をして４Ｈｚでの２次抵抗を求めたい場合などのように、測定範囲外の曲線を求めるには不向きである。そのため少しでも各測定ポイントでの測定値に誤差をもつとその誤差を持つ測定値を近似する曲線にのせるために測定範囲外の曲線は大きく変化する。つまり多項式近似を使うためには各測定周波数での測定値は誤差があってはならず、実測定では非常に困難である。
（２）１次、２次の合成抵抗及び合成漏れインダクタンスを求める際に、相互インダクタンスＭを開放と近似しているために、推定誤差を生じる。 Also, the polynomial approximation originally obtains a curve that correctly passes through the measurement point. For example, when measuring from 10 Hz to 60 Hz to obtain the secondary resistance at 4 Hz as in the case of secondary resistance estimation, It is not suitable for obtaining a curve outside the measurement range. Therefore, if there is any error in the measured value at each measurement point, the curve outside the measurement range greatly changes in order to put the measured value having the error on a curve approximating the measured value. That is, in order to use the polynomial approximation, the measured value at each measurement frequency must not have an error, and it is very difficult in actual measurement.
(2) When calculating the primary and secondary combined resistance and the combined leakage inductance, an estimation error occurs because the mutual inductance M is approximated to be open.

従来、１次、２次の合成抵抗及び合成漏れインダクタンスを求める際に相互インダクタンスＭを開放として近似していた。実際に測定を行う場合は測定電流は相互インダクタンスＭへも流れており、その電流は２次側の２次抵抗、２次漏れインダクタンスに流れるとして計算することになる。   Conventionally, when calculating the primary and secondary combined resistance and the combined leakage inductance, the mutual inductance M is approximated as being open. When the measurement is actually performed, the measurement current is also flowing through the mutual inductance M, and the calculation is performed assuming that the current flows through the secondary resistance and the secondary leakage inductance on the secondary side.

この電流値はわずかではあるが、２次抵抗推定値として５％程度の誤差を生じることがあった。しかし、相互インダクタンスＭを考慮して求めるためには収束計算が必要となり構成が困難であった。
（３）起動トルク不足を解消するための、従来の起動時抵抗値推定方式では、１次抵抗と２次抵抗との温度が一定でない場合に誤差が大きくなる。 Although this current value is slight, an error of about 5% may occur as an estimated secondary resistance value. However, a convergence calculation is required to obtain the value in consideration of the mutual inductance M, and the configuration is difficult.
(3) In the conventional startup resistance value estimation method for solving the shortage of the startup torque, an error increases when the temperature of the primary resistance and the secondary resistance is not constant.

従来の起動時抵抗推定方法は起動時に直流電圧を印加しある特定間隔後の電流値をみて抵抗を推定している。通常１次抵抗を測定するために直流を流し、十分定常に達した後の電圧／電流より抵抗を求める方法を過渡状態でもその抵抗値の影響によりその電流値が変化することを利用して比例計算で求めているに過ぎず、その変化は実際には比例的にならない場合も有り誤差を多く含む。また、２次抵抗は同様に比例計算で求めているため、１次、２次の温度が異なっていった場合などはその温度差による抵抗変化分だけ誤差を持つことになる。   In the conventional resistance estimation method at the time of startup, a DC voltage is applied at the time of startup, and the resistance is estimated by looking at the current value after a specific interval. Normally, a DC current is applied to measure the primary resistance, and the method of obtaining the resistance from the voltage / current after reaching a sufficiently steady state is based on the fact that the current value changes under the influence of the resistance value even in a transient state. The change is merely obtained by calculation, and the change may not actually be proportional, and may include many errors. Further, the secondary resistance is similarly obtained by the proportional calculation, so that when the primary and secondary temperatures are different, for example, there is an error by the resistance change due to the temperature difference.

従って、この発明は、かかる問題点を解決するためになされたもので、この発明の第１の目的は、測定に時間がかかることなく、少ない測定周波数で精度よく表皮効果の影響を考慮し、２次抵抗、漏れインダクタンスの測定精度向上することにより制御性能を向上した誘導電動機制御装置を提供すること、この発明の第２の目的は、抵抗及び漏れインダクタンス測定時に相互インダクタンスを考慮することにより、相互インダクタンスを開放と近似することによる誤差を減らすもので、しかも収束計算など使用せずに簡単に計算するようにすることにより制御性能を向上した誘導電動機制御装置を提供すること、この発明の第３の目的は、１次抵抗及び２次抵抗を、比例計算ではなく、直接抵抗値を推定できるようにようにし、１次、２次の温度が異なっている場合にも誤差が少ないようにすることにより制御性能を向上した誘導電動機制御装置を提供することである。   Therefore, the present invention has been made in order to solve such a problem, and a first object of the present invention is to take the effect of the skin effect accurately at a small measurement frequency without taking much time for measurement, A second object of the present invention is to provide an induction motor control device having improved control performance by improving the measurement accuracy of the secondary resistance and the leakage inductance, and by taking the mutual inductance into account when measuring the resistance and the leakage inductance. An object of the present invention is to provide an induction motor control device which reduces errors caused by approximating the mutual inductance as being open and which has an improved control performance by simply calculating without using convergence calculation. The purpose of 3 is to make it possible to estimate the primary resistance and the secondary resistance directly instead of the proportional calculation and to estimate the primary resistance and the secondary resistance. Degrees is that to provide an induction motor controller with improved control performance by also so small error if they differ.

この発明の誘導電動機制御装置は、第１の演算手段が、少なくとも３つの異なる周波数ｆの試験電源による試験条件のそれぞれに対応する２次巻線抵抗Ｒ２と漏れインダクタンスＬとを演算し、第２の演算手段が、２次巻線抵抗Ｒ２と漏れインダクタンスＬのそれぞれを分母と分子に周波数ｆの２次式を含む有理関数を有し、試験条件のそれぞれに対応する２次巻線抵抗Ｒ２と漏れインダクタンスＬとを有理関数に代入してこの有理関数を求め、この求めた有理関数に所要の滑り周波数を代入して２次巻線抵抗Ｒ２と漏れインダクタンスＬとを演算するようにしたので、誘導電動機の滑り周波数に対する表皮効果を考慮した２次抵抗、漏れインダクタンスを少なくとも３つの周波数での測定により、簡単に求めることができるとともに、実運転時に必要とされる誘導電動機の滑り周波数での２次抵抗、漏れインダクタンスを精度よく演算測定することができ、そしてその値を実運転時の制御に使用することができるので、速度やトルクの演算を行う場合においてその演算精度が高くなる効果を奏する。   In the induction motor control device of the present invention, the first calculating means calculates the secondary winding resistance R2 and the leakage inductance L corresponding to each of the test conditions using at least three test power supplies having different frequencies f, and Has a rational function including a quadratic expression of the frequency f in the denominator and the numerator of the secondary winding resistance R2 and the leakage inductance L, respectively. The leakage inductance L is substituted into a rational function to obtain this rational function, and the required slip frequency is substituted into the obtained rational function to calculate the secondary winding resistance R2 and the leakage inductance L. The secondary resistance and leakage inductance in consideration of the skin effect on the slip frequency of the induction motor can be easily obtained by measuring at least three frequencies, The secondary resistance and leakage inductance at the slip frequency of the induction motor, which are sometimes required, can be accurately calculated and measured, and the values can be used for control during actual operation. Is performed, the operation accuracy is improved.

また、この発明の誘導電動機制御装置は、３つの異なる周波数ｆの試験電源による試験条件のそれぞれに対応する２次巻線抵抗Ｒ２と漏れインダクタンスＬとを演算し、次に示す簡単な有理関数、
Ｒ２（ｆ）＝（ａ１＋ａ２・ｆ2）／（１＋ａ３・ｆ2）
Ｌ （ｆ）＝（ｂ１＋ｂ２・ｆ2）／（１＋ｂ３・ｆ2）
により実運転時に必要とされる誘導電動機の滑り周波数での２次抵抗、漏れインダクタンスを精度よく演算測定することができる効果を奏する。 Further, the induction motor control device of the present invention calculates the secondary winding resistance R2 and the leakage inductance L corresponding to each of the test conditions using the test power supply having three different frequencies f, and calculates a simple rational function shown below.
R2 (f) = (a1 + a2 · f 2 ) / (1 + a3 · f 2 )
L (f) = (b1 + b2 · f 2 ) / (1 + b3 · f 2 )
Accordingly, the secondary resistance and the leakage inductance at the slip frequency of the induction motor required at the time of actual operation can be accurately calculated and measured.

また、この発明の誘導電動機制御装置は、第２の演算手段が、係数ａ１及びｂ１のみを求め、低滑り周波数に対応する２次巻線抵抗Ｒ２と漏れインダクタンスＬを演算するので、短時間に実運転時に必要とされる誘導電動機の滑り周波数での２次抵抗、漏れインダクタンスを精度よく演算測定することができる効果を奏する。   Further, in the induction motor control device of the present invention, the second calculating means calculates only the coefficients a1 and b1 and calculates the secondary winding resistance R2 and the leakage inductance L corresponding to the low slip frequency, so that it can be performed in a short time. This has the effect of accurately calculating and measuring the secondary resistance and leakage inductance at the slip frequency of the induction motor required during actual operation.

また、この発明の誘導電動機制御装置は、第１の演算手段が、所定の周波数ｆにおける第１の２次巻線抵抗Ｒ２を演算し、第２の演算手段が、この第２の演算手段が予め求めた有理関数に所定の周波数ｆを代入して第２の２次巻線抵抗Ｒ２を求めるとともに、所要の滑り周波数ｆを代入して第３の２次巻線抵抗Ｒ２を求め、この第３の２次巻線抵抗Ｒ２に第２の２次巻線抵抗Ｒ２と第１の２次巻線抵抗Ｒ２の比を乗じて所要の２次巻線抵抗Ｒ２を求めるので、更に、短時間に実運転時に必要とされる誘導電動機の滑り周波数での２次抵抗、漏れインダクタンスを精度よく演算測定することができる効果を奏する。   Further, in the induction motor control device of the present invention, the first calculating means calculates the first secondary winding resistance R2 at the predetermined frequency f, and the second calculating means calculates the first secondary winding resistance R2. Substituting a predetermined frequency f into a previously obtained rational function to obtain a second secondary winding resistance R2, and substituting a required slip frequency f to obtain a third secondary winding resistance R2. 3 is multiplied by the ratio of the second secondary winding resistance R2 to the first secondary winding resistance R2 to obtain the required secondary winding resistance R2. This has the effect of accurately calculating and measuring the secondary resistance and leakage inductance at the slip frequency of the induction motor required during actual operation.

また、この発明の誘導電動機制御装置は、電力演算手段が、周波数ｆの試験電源に対応する有効電力Ｐと無効電力Ｑとを演算し、第１の演算手段が、周波数ｆに対応する相電圧ピーク値Ｖｐ及び１次角周波数ω、有効電力Ｐと無効電力Ｑ、予め予測された一次巻線抵抗Ｒ１とからＬ型等価回路より求められる演算式により２次抵抗Ｒ２ｄと２次漏れインダクタンスＬ２ｄを求め、第２の演算手段が、Ｔ型等価回路より求められる２次巻線抵抗Ｒ２と２次漏れインダクタンスＬの演算式のおのおのに、第１の演算手段が求めた２次巻線抵抗Ｒ２ｄと２次漏れインダクタンスＬ２ｄと予め測定されている２次自己インダクタンスＬＬ２とを代入して２次巻線抵抗Ｒ２と２次漏れインダクタンスＬを求めるので、有効電力、無効電力を用いて２次巻線抵抗、漏れインダクタンスを計算する際に相互インダクタンスを考慮した演算が収束計算など用いずに求めることができるため、収束計算などの複雑なプログラミングをする手間が省け、更に、相互インダクタンスを考慮しているために２次巻線抵抗、漏れインダクタンスの演算測定誤差が減少できる効果を奏する。   Further, in the induction motor control device according to the present invention, the power calculating means calculates the active power P and the reactive power Q corresponding to the test power supply having the frequency f, and the first calculating means calculates the phase voltage corresponding to the frequency f. The secondary resistance R2d and the secondary leakage inductance L2d are calculated from the peak value Vp, the primary angular frequency ω, the active power P and the reactive power Q, and the primary winding resistance R1 predicted from the L-type equivalent circuit. The second arithmetic means calculates the secondary winding resistance R2d and the secondary winding resistance R2d obtained by the first arithmetic means for each of the arithmetic expressions of the secondary winding resistance R2 and the secondary leakage inductance L obtained from the T-type equivalent circuit. Since the secondary winding resistance R2 and the secondary leakage inductance L are obtained by substituting the secondary leakage inductance L2d and the secondary self inductance LL2 measured in advance, the secondary winding is calculated using active power and reactive power. When calculating the resistance and the leakage inductance, the operation considering the mutual inductance can be obtained without using the convergence calculation or the like, so that complicated programming such as the convergence calculation can be omitted, and the mutual inductance is considered. Therefore, there is an effect that errors in calculation of secondary winding resistance and leakage inductance can be reduced.

また、この発明の誘導電動機制御装置は、Ｌ型等価回路より求められる演算式は、
Ｌ２ｄ＝Ｑ・Ｖｐ2／（４・ω・（Ｐ2＋Ｑ2））
Ｒ２ｄ＝Ｐ・Ｖｐ2／（２・（Ｐ2＋Ｑ2））−Ｒ１
Ｔ型等価回路より求められる演算式は、
Ｌ＝Ｌ２ｄ＋（２・Ｌ２ｄ−ＬＬ２）・Ｒ２ｄ2／（２・ω2・ＬＬ２2）
Ｒ２＝Ｒ２ｄ・ＬＬ２2 ／（ＬＬ２−Ｌ２ｄ）2
であるので、簡単な計算により相互インダクタンスを考慮した演算が収束計算など用いずに求めることができる効果を奏する。 In the induction motor control device of the present invention, the arithmetic expression obtained from the L-type equivalent circuit is:
L2d = Q · Vp 2 / (4 · ω · (P 2 + Q 2 ))
R2d = P · Vp 2 / (2 · (P 2 + Q 2 )) − R1
The arithmetic expression obtained from the T-type equivalent circuit is
L = L2d + (2 · L2d−LL2) · R2d 2 / (2 · ω 2 · LL2 2 )
R2 = R2d · LL2 2 / ( LL2-L2d) 2
Therefore, there is an effect that a calculation in consideration of the mutual inductance can be obtained by a simple calculation without using a convergence calculation or the like.

また、この発明の誘導電動機制御装置は、電力演算手段が、多相インバータが多相誘導電動機に供給する単相交流の印加電圧より瞬時有効電力と瞬時無効電力を求め、瞬時有効電力と瞬時無効電力から高周波成分をフィルタリングするので、有効電力と無効電力を電動機を拘束せずに精度良く測定できる効果を奏する。   Further, in the induction motor control device according to the present invention, the power calculation means obtains the instantaneous active power and the instantaneous reactive power from the applied voltage of the single-phase AC supplied to the polyphase induction motor by the polyphase inverter, and Since the high-frequency component is filtered from the electric power, there is an effect that the active power and the reactive power can be accurately measured without restricting the electric motor.

また、この発明の誘導電動機制御装置は、電力演算手段が、瞬時有効電力と瞬時無効電力から印加電圧の２倍の周波数成分をノッチフィルタにより、２倍の周波数成分を超える高周波成分をローパスフィルタによりフィルタリングするので、より精度良く有効電力と無効電力を測定できる効果を奏する。   Further, in the induction motor control device according to the present invention, the electric power calculation means uses a notch filter to apply a frequency component twice the applied voltage from the instantaneous active power and the instantaneous reactive power using a notch filter, and a high-pass component exceeding the twice frequency component using a low-pass filter. Since the filtering is performed, there is an effect that the active power and the reactive power can be measured more accurately.

また、この発明の誘導電動機制御装置は、試験電源指令手段が、多相インバータに対して多相誘導電動機に指定した実効値電流を供給するように指令電圧を出力するので、実効値電流値により変化する２次巻線抵抗及び漏れインダクタンスに対して、実運転時の実効値電流値に対応する２次巻線抵抗及び漏れインダクタンスを精度良く測定できる効果を奏する。   In the induction motor control device according to the present invention, the test power supply command means outputs the command voltage to the multi-phase inverter so as to supply the specified effective value current to the multi-phase induction motor. With respect to the changing secondary winding resistance and leakage inductance, there is an effect that the secondary winding resistance and leakage inductance corresponding to the effective current value in the actual operation can be accurately measured.

また、この発明の誘導電動機制御装置は、試験電源指令手段が、多相インバータに対する指令電圧を所定の上昇率で上昇し、この指令電圧の上昇に伴って多相インバータが多相誘導電動機へ供給する実効値電流の二乗の時間平均値が指定した実効値電流の１／３と等しいかあるいは大きくなったときに指令電圧の上昇を止めるので、有効電力と無効電力の測定に必要な実効値電流の設定が容易に行える効果を奏する。   In the induction motor control device according to the present invention, the test power supply command means increases a command voltage for the polyphase inverter at a predetermined rate of increase, and supplies the polyphase inverter to the polyphase induction motor with the increase of the command voltage. When the time average of the square of the effective value current becomes equal to or greater than 1/3 of the specified effective value current, the command voltage rise is stopped, so the effective value current required for measuring the active power and the reactive power This makes it possible to easily set.

また、この発明の誘導電動機制御装置は、試験電源指令手段が、多相インバータに指令電圧を出力し、電力演算手段が、測定及び演算による遅れに対応した補正位相により指令電圧の位相を補正して有効電力Ｐと無効電力Ｑとを演算するので、有効電力、無効電力の測定結果に影響を及ぼす指令電圧の位相差を補正できるので、サンプリング周期や演算遅れ時間により有効電力、無効電力の測定結果が変わることなく正確に測定できる効果を奏する。   Further, in the induction motor control device of the present invention, the test power supply command means outputs a command voltage to the polyphase inverter, and the power calculation means corrects the phase of the command voltage with a correction phase corresponding to a delay due to measurement and calculation. Since the active power P and the reactive power Q are calculated by the calculation, the phase difference of the command voltage which affects the measurement results of the active power and the reactive power can be corrected. This has the effect of being able to measure accurately without changing the result.

また、この発明の誘導電動機制御装置は、電流制御手段が、多相インバータを制御して停止中の多相誘導電動機に所定の直流指令電圧を印加して直流電流を供給し、抵抗推定手段が、直流電圧と直流電流とを入力して、電流制御手段の電流供給直後の第１の期間に１次巻線抵抗と２次巻線抵抗との合成抵抗を演算し、第１の期間の次の第２の期間に１次巻線抵抗を演算し、第１の期間に演算した合成抵抗から第２の期間に演算した１次巻線抵抗を減じて２次巻線抵抗を推定するので、起動時に短時間に１次巻線抵抗、２次巻線抵抗を推定可能であり、起動の度に推定することにより、温度変化により１次巻線抵抗、２次巻線抵抗が変化しても起動時にトルク不足になることはない。また１次巻線抵抗、２次巻線抵抗の温度が異なっていてもそれぞれの温度における１次巻線抵抗、２次巻線抵抗を求めることができる効果を奏する。   In the induction motor control device according to the present invention, the current control means controls the polyphase inverter to apply a predetermined DC command voltage to the stopped polyphase induction motor to supply a DC current, and the resistance estimating means , A DC voltage and a DC current, and a combined resistance of the primary winding resistance and the secondary winding resistance is calculated in the first period immediately after the current supply of the current control means, and the next period after the first period is calculated. The primary winding resistance is calculated in the second period, and the secondary winding resistance is estimated by subtracting the primary winding resistance calculated in the second period from the combined resistance calculated in the first period. It is possible to estimate the primary winding resistance and the secondary winding resistance in a short time at the time of starting, and by estimating each time of starting, even if the primary winding resistance and the secondary winding resistance change due to a temperature change. There is no shortage of torque at startup. Further, even when the temperatures of the primary winding resistance and the secondary winding resistance are different, the primary winding resistance and the secondary winding resistance at each temperature can be obtained.

また、この発明の誘導電動機制御装置は、抵抗推定手段に、多相誘導電動機に供給した直流電流と多相誘導電動機の既知の回路定数と帰還された１次巻線抵抗と２次巻線抵抗とから多相誘導電動機に印加した直流電圧値を推定する電圧推定器と、直流電圧値と直流電圧指令値との偏差を積分演算する積分器と、積分器の出力を基に第１の期間に１次巻線と２次巻線との合成抵抗を演算し、合成抵抗を１次巻線抵抗と２次巻線抵抗に分離して電圧推定器に帰還し、第２の期間に積分器出力より１次巻線抵抗を演算し、第１の期間に演算した合成抵抗から第２の期間に測定した１次巻線抵抗を減じて２次巻線抵抗を推定し、演算した１次巻線抵抗と推定した２次巻線抵抗を電圧推定器に帰還する抵抗演算手段とを備え、偏差が所定値以下となった場合に第２の期間に測定及び推定した１次巻線抵抗と２次巻線抵抗を使用して多相誘導電動機を制御するので、抵抗値を精度よく推定できる効果を奏する。   Further, the induction motor control device of the present invention provides the resistance estimating means with a DC current supplied to the multi-phase induction motor, a known circuit constant of the multi-phase induction motor, and the feedback of the primary winding resistance and the secondary winding resistance. , A voltage estimator for estimating a DC voltage value applied to the polyphase induction motor, an integrator for integrating the deviation between the DC voltage value and the DC voltage command value, and a first period based on the output of the integrator Calculates the combined resistance of the primary winding and the secondary winding, separates the combined resistance into a primary winding resistance and a secondary winding resistance, and feeds it back to the voltage estimator. The primary winding resistance is calculated from the output, the secondary winding resistance is estimated by subtracting the primary winding resistance measured in the second period from the combined resistance calculated in the first period, and the calculated primary winding resistance is calculated. A resistance calculating means for feeding back the secondary winding resistance estimated as the line resistance to the voltage estimator; And controls the polyphase induction motor by using the measured and estimated primary winding resistance and the secondary winding resistance in the second period to the case, an effect of the resistance value can be accurately estimated.

また、この発明の誘導電動機制御装置は、抵抗推定手段に、多相誘導電動機に指令した直流指令電圧と多相誘導電動機の既知の回路定数と帰還された１次巻線抵抗と２次巻線抵抗とから多相誘導電動機に供給した直流電流値を推定する電流推定器と、直流電流値と多相誘導電動機に供給した直流電流の検出値との偏差を積分演算する演算器と、この積分器の出力を基に第１の期間に１次巻線と２次巻線との合成抵抗を演算し、合成抵抗を１次巻線抵抗と２次巻線抵抗に分離して前記電圧推定器に帰還し、第２の期間に積分器出力より１次巻線抵抗を演算し、第１の期間に演算した合成抵抗から第２の期間に測定した１次巻線抵抗を減じて２次巻線抵抗を推定し、演算した１次巻線抵抗と推定した２次巻線抵抗を電流推定器に帰還する抵抗演算手段とを備え、偏差が所定値以下となった場合に第２の期間に測定及び推定した１次巻線抵抗と２次巻線抵抗を使用して多相誘導電動機を制御するので、抵抗値を精度よく推定できる効果を奏する。   Further, the induction motor control device of the present invention provides the resistance estimating means with a DC command voltage commanded to the multi-phase induction motor, a known circuit constant of the multi-phase induction motor, and a feedback of the primary winding resistance and the secondary winding. A current estimator for estimating the value of the DC current supplied to the polyphase induction motor from the resistance, a calculator for integrating the deviation between the DC current value and the detected value of the DC current supplied to the polyphase induction motor, Calculating a combined resistance of a primary winding and a secondary winding in a first period based on an output of the voltage estimator, separating the combined resistance into a primary winding resistance and a secondary winding resistance, To calculate the primary winding resistance from the integrator output in the second period, and subtract the primary winding resistance measured in the second period from the combined resistance calculated in the first period to obtain a secondary winding resistance. Resistance calculation for estimating the line resistance and feeding back the calculated primary winding resistance and the estimated secondary winding resistance to the current estimator And a step for controlling the polyphase induction motor using the primary winding resistance and the secondary winding resistance measured and estimated in the second period when the deviation becomes equal to or less than a predetermined value. Has an effect of accurately estimating.

また、この発明の誘導電動機制御装置は、抵抗演算手段が、第１の期間に演算した合成抵抗を１次巻線抵抗と２次巻線抵抗とに所定の比で分離するので、あらかじめ求めた１次巻線抵抗と２次巻線抵抗の比を使うようにすることにより、その比が判らないときに抵抗の推定時間が長くなることを防止できる効果を奏する。   Further, in the induction motor control device of the present invention, since the resistance calculating means separates the combined resistance calculated in the first period into a primary winding resistance and a secondary winding resistance at a predetermined ratio, it is obtained in advance. By using the ratio between the primary winding resistance and the secondary winding resistance, it is possible to prevent the estimation time of the resistance from being lengthened when the ratio is unknown.

また、この発明の誘導電動機制御装置は、予め交流電圧を多相誘導電動機に印加して求めた交流印加による１次巻線抵抗Ｒ１及び２次巻線抵抗Ｒ２と、抵抗推定手段によりほぼ同時に推定した直流電圧印加による１次巻線抵抗Ｒ１及び２次巻線抵抗Ｒ２との比を記憶する記憶手段を備え、抵抗推定手段により推定した１次巻線抵抗Ｒ１及び２次巻線抵抗Ｒ２を記憶手段に記憶した比により補正するので、起動時の抵抗をより精度良く推定することができる効果を奏する。   In addition, the induction motor control device of the present invention estimates the primary winding resistance R1 and the secondary winding resistance R2 by applying an AC voltage obtained in advance by applying an AC voltage to the polyphase induction motor, and the resistance estimation means almost simultaneously. Storage means for storing the ratio between the primary winding resistance R1 and the secondary winding resistance R2 due to the applied DC voltage, and the primary winding resistance R1 and the secondary winding resistance R2 estimated by the resistance estimating means are stored. Since the correction is made based on the ratio stored in the means, there is an effect that the resistance at the time of starting can be more accurately estimated.

また、この発明の誘導電動機制御装置は、運転期間と停止期間を繰り返す多相誘導電動機の停止期間に１次巻線抵抗と２次巻線抵抗を推定して多相誘導電動機を制御するので、運転と停止による巻線の温度変化による抵抗変化を推定できるので起動時の抵抗をより精度良く推定することができる効果を奏する。   Further, the induction motor control device of the present invention controls the polyphase induction motor by estimating the primary winding resistance and the secondary winding resistance during the stop period of the polyphase induction motor in which the operation period and the stop period are repeated. Since the resistance change due to the temperature change of the winding due to the operation and the stop can be estimated, there is an effect that the resistance at the time of starting can be estimated more accurately.

実施の形態１．
図１はこの発明の実施の形態１による誘導電動機制御装置を示すブロック図である。図１において、１０１はインバータ、１０２は誘導電動機、１０３はインバータ１０１から誘導電動機１０２へ供給される電流を検出する電流検出器、１０４はインバータ１０１より誘導電動機１０２へ印加する試験電源を指令する試験電源指令手段であり、３つの異なる周波数の指令電圧を順次出力する。１０５は電流検出器１０３で検出した電流と試験電源指令手段１０４からの指令電圧とを用い有効電力Ｐ及び無効電力Ｑを演算測定する有効電力・無効電力演算測定手段（以降、電力演算手段と称す。）、１０６は電力演算手段１０５より測定された結果を用い２次抵抗Ｒ２と漏れインダクタンスＬを演算測定する２次抵抗・漏れインダクタンス演算測定手段、１０７は２次抵抗・漏れインダクタンス演算測定手段１０６から出力される異なる３つの周波数の指令電圧に対応する２次抵抗・漏れインダクタンスの値を用い表皮効果を考慮した値を演算する表皮効果考慮演算手段である。また、１０８は表皮効果考慮演算手段１０７が出力した値を実際の運転時に使用するため記憶する、２次抵抗・漏れインダクタンス記憶手段である。 Embodiment 1 FIG.
FIG. 1 is a block diagram showing an induction motor control device according to Embodiment 1 of the present invention. In FIG. 1, 101 is an inverter, 102 is an induction motor, 103 is a current detector for detecting a current supplied from the inverter 101 to the induction motor 102, and 104 is a test for instructing a test power supply to be applied from the inverter 101 to the induction motor 102. The power supply command means sequentially outputs command voltages of three different frequencies. Reference numeral 105 denotes an active / reactive power calculation and measurement unit (hereinafter referred to as a power calculation unit) that calculates and measures the active power P and the reactive power Q using the current detected by the current detector 103 and the command voltage from the test power supply command unit 104. ), 106 are secondary resistance / leakage inductance calculation and measurement means for calculating and measuring the secondary resistance R2 and leakage inductance L using the result measured by the power calculation means 105, and 107 is secondary resistance / leakage inductance calculation and measurement means 106 Means for calculating a skin effect taking into account the skin effect using the values of the secondary resistance and the leakage inductance corresponding to the command voltages of three different frequencies output from the controller. Reference numeral 108 denotes a secondary resistance / leakage inductance storage unit that stores the value output by the skin effect consideration calculation unit 107 for use during actual operation.

次に、この誘導電動機制御装置の表皮効果考慮演算手段１０７の動作を説明する。   Next, the operation of the skin effect consideration calculation means 107 of the induction motor control device will be described.

背景技術で説明したように表皮効果により２次抵抗は滑り周波数によって変化する。この変化の様子は籠形回転子の溝形状によって多少違うものの、いろいろな２次導体形状の表皮効果の影響は、その周波数対２次抵抗の実験により、いずれも分母と分子に周波数ｆの２次式を含む有理関数で表せることが判った。この有理関数は次のように示される。   As described in the background art, the secondary resistance changes according to the slip frequency due to the skin effect. Although the state of this change is slightly different depending on the groove shape of the cage rotor, the effect of the skin effect of various secondary conductor shapes can be found in both the denominator and the numerator by the experiment of frequency versus secondary resistance. It was found that it can be expressed by a rational function including the following equation. This rational function is shown as follows.

Ｒ２（ｆ）＝（ｋ１＋ｋ２・ｆ2）／（ｋ３＋ｋ４・ｆ2） ・・・（６）
ここで、ｆは周波数、ｋ１〜ｋ４は定数である。 R2 (f) = (k1 + k2 · f 2 ) / (k3 + k4 · f 2 ) (6)
Here, f is a frequency, and k1 to k4 are constants.

（６）式においてａ１＝ｋ１／ｋ３、ａ２＝ｋ２／ｋ３、ａ３＝ｋ４／ｋ３とおくと
Ｒ２（ｆ）＝（ａ１＋ａ２・ｆ2）／（１＋ａ３・ｆ2） ・・・（７）
となる。ここで例えば籠形回転子の溝形状が２重籠形となっている誘導電動機の等価回路は図２のように示される。ここで、Ｌ２０は２次側共通漏れインダクタンス、Ｒ２ｐは２重籠の上部導体の抵抗値、Ｒ２ｓは２重籠の下部導体の抵抗値、Ｌ２ｐは２重籠の上部導体の漏れインダクタンス、Ｌ２ｓは２重籠の下部導体の漏れインダクタンス、Ｚｒは２次側合成インピーダンスである。この２次側合成インピーダンスＺｒは次式のように示される。 In the equation (6), if a1 = k1 / k3, a2 = k2 / k3, and a3 = k4 / k3, R2 (f) = (a1 + a2 · f 2 ) / (1 + a3 · f 2 ) (7)
It becomes. Here, for example, an equivalent circuit of an induction motor in which the groove shape of a cage rotor is a double cage shape is shown in FIG. Here, L20 is the secondary common leakage inductance, R2p is the resistance value of the upper conductor of the double basket, R2s is the resistance value of the lower conductor of the double basket, L2p is the leakage inductance of the upper conductor of the double basket, L2s is The leakage inductance of the lower conductor of the double basket, Zr, is the secondary-side composite impedance. This secondary-side combined impedance Zr is expressed by the following equation.

Figure 2004173496
Figure 2004173496

（８）式より、その実部と虚部はそれぞれ周波数ｆの２次式を含む有理関数の特性となることが判る。誘導電動機１０２の制御においては２次導体を抵抗と漏れインダクタンスの直列回路として扱うので、２次合成インピーダンスＺｒの実部が２次抵抗、虚部が２次漏れインダクタンスとして扱われることとなる。よって、２重籠では２次側周波数による２次抵抗、漏れインダクタンスは周波数ｆの２次式を含む有理関数で表せることが理論的にも判る。   From equation (8), it can be seen that the real part and the imaginary part have the characteristics of a rational function including a quadratic equation of the frequency f. In the control of the induction motor 102, the secondary conductor is treated as a series circuit of a resistance and a leakage inductance, so that the real part of the secondary combined impedance Zr is treated as a secondary resistance and the imaginary part is treated as a secondary leakage inductance. Therefore, it is theoretically understood that in the double basket, the secondary resistance and the leakage inductance based on the secondary side frequency can be expressed by a rational function including a quadratic expression of the frequency f.

このように、周波数ｆの２次式を含む有理関数で表せると３つの周波数ｆ１、ｆ２、ｆ３に対応するＲ抵抗Ｒ２（ｆ１）、Ｒ２（ｆ２）、Ｒ２（ｆ３）が判ると係数ａ１、ａ２、ａ３は代数計算により求めることができる。このように係数が求まれば測定できないような数Ｈｚといった低周波での２次抵抗も容易に演算することができる。   As described above, when expressed by a rational function including a quadratic expression of the frequency f, the coefficients a1 and R2 (f3) are obtained when the R resistances R2 (f1), R2 (f2) and R2 (f3) corresponding to the three frequencies f1, f2 and f3 are known. a2 and a3 can be obtained by algebraic calculation. Thus, the secondary resistance at a low frequency such as several Hz, which cannot be measured if the coefficient is obtained, can be easily calculated.

上記は２次抵抗について述べたが同様なことが漏れインダクタンスについても言える。   Although the above description has been made on the secondary resistance, the same can be said for the leakage inductance.

また、理論的な説明として２次導体の溝形状が２重籠形の場合について説明したが、深溝形の場合においても上述の有理関数で表せることが判った。   In addition, as a theoretical explanation, the case where the groove shape of the secondary conductor is a double cage shape has been described, but it has been found that the case where the groove shape is the deep groove shape can be represented by the above-mentioned rational function.

また、周波数ｆの２次式と４次式を含む有理関数により２次抵抗及び漏れインダクタンスを表しても良いが、係数の数が５個となり、５つの周波数ｆ１、ｆ２、ｆ３、ｆ４、ｆ５での２次抵抗Ｒ２（ｆ１）、Ｒ２（ｆ２）、Ｒ２（ｆ３）、Ｒ２（ｆ４）、Ｒ２（ｆ５）から５つの係数を求めることが必要となり、演算が複雑となり抵抗の推定に時間がかかることになる。上述の実験では、周波数ｆの２次式の有理関数により十分な精度で２次抵抗及び漏れインダクタンスが推定できることが判った。   The secondary resistance and the leakage inductance may be expressed by a rational function including a quadratic expression and a quartic expression of the frequency f. However, the number of coefficients is five, and the five frequencies f1, f2, f3, f4, f5 It is necessary to obtain five coefficients from the secondary resistances R2 (f1), R2 (f2), R2 (f3), R2 (f4), and R2 (f5), and the calculation becomes complicated, and it takes time to estimate the resistance. That would be. In the above experiment, it was found that the secondary resistance and the leakage inductance can be estimated with sufficient accuracy by the rational function of the quadratic expression of the frequency f.

次に、実施の形態１による誘導電動機制御装置の動作を説明する。   Next, the operation of the induction motor control device according to the first embodiment will be described.

図１において、試験電源指令手段１０４から６０Ｈｚ、３５Ｈｚ、１０Ｈｚの３つの周波数の指令電圧が順次出力される。それぞれの周波数における試験電源指令手段１０４からの指令電圧と電流検出器１０３の出力により電力演算手段１０５にて有効電力・無効電力が求められる。   In FIG. 1, command voltages having three frequencies of 60 Hz, 35 Hz, and 10 Hz are sequentially output from the test power supply command unit 104. Based on the command voltage from the test power supply command means 104 and the output of the current detector 103 at each frequency, the power calculation means 105 calculates active power and reactive power.

この場合試験電源は３相正弦波、あるいは単相正弦波いずれでもよい。   In this case, the test power supply may be a three-phase sine wave or a single-phase sine wave.

但し、３相正弦波の場合、１０Ｈｚの周波数指令を出力すると誘導電動機１０２はすぐに回転を始めてしまい２次抵抗の測定はできない。したがって、３相正弦波の場合にはその周波数での２抵抗及び漏れインダクタンスを測定する間に回転し始めないように、１０Ｈｚより高い周波数を選択する必要がある。この周波数は、負荷の種類、負荷の大きさにより適宜選択することができる。また、使用条件によっては、誘導電動機１０２の回転軸を拘束して測定することができる。単相の場合は１０Ｈｚでも回転することはないので単相を印加する方がよい。また、有効電力・無効電力の演算測定にここでは試験電源指令手段１０４が出力する指令電圧を用いたが、電圧検出器を設け誘導電動機の端子側で検出した電圧を用いてもよい。   However, in the case of a three-phase sine wave, when a frequency command of 10 Hz is output, the induction motor 102 immediately starts rotating, and the secondary resistance cannot be measured. Therefore, in the case of a three-phase sine wave, it is necessary to select a frequency higher than 10 Hz so as not to start rotating while measuring the two resistances and the leakage inductance at that frequency. This frequency can be appropriately selected depending on the type of the load and the size of the load. Further, depending on the use conditions, the measurement can be performed with the rotation axis of the induction motor 102 restricted. In the case of a single phase, since rotation does not occur even at 10 Hz, it is better to apply a single phase. Although the command voltage output from the test power supply command means 104 is used here for calculating and calculating the active power and the reactive power, a voltage detector may be provided and a voltage detected at the terminal side of the induction motor may be used.

このようにして有効電力、無効電力が求まれば２次抵抗・漏れインダクタンスは演算できる。ここで、６０Ｈｚ、３５Ｈｚ、１０Ｈｚでの２次抵抗の値をＲ２（６０）、Ｒ２（３５）、Ｒ２（１０）、漏れインダクタンスの値をＬ（６０）、Ｌ（３５）、Ｌ（１０）とすると、有理関数の係数は次のように求まる。   If the active power and the reactive power are obtained in this way, the secondary resistance and the leakage inductance can be calculated. Here, the values of the secondary resistance at 60 Hz, 35 Hz and 10 Hz are R2 (60), R2 (35) and R2 (10), and the values of the leakage inductance are L (60), L (35) and L (10). Then, the coefficient of the rational function is obtained as follows.

ａ１＝｛−１９・Ｒ２（６０）・Ｒ２（３５）＋３４３・Ｒ２（６０）・Ｒ２（１０）−３２４・Ｒ２（３５）・Ｒ２（１０）｝／
｛３２４・Ｒ２（６０）−３４３・Ｒ２（３５）＋１９・Ｒ２（１０）｝
ａ２＝｛１９・Ｒ２（６０）・Ｒ２（３５）−２８・Ｒ２（６０）・Ｒ２（１０）＋９・Ｒ２（３５）・Ｒ２（１０）｝／
｛１００・〔３２４・Ｒ２（６０）−３４３・Ｒ２（３５）＋１９・Ｒ２（１０）〕｝
ａ３＝｛−９・Ｒ２（６０）＋２８・Ｒ２（３５）−１９・Ｒ２（１０）｝／
｛１００・〔３２４・Ｒ２（６０）−３４３・Ｒ２（３５）＋１９・Ｒ２（１０）〕｝
使用する誘導電動機の定格トルク時の滑り周波数が４Ｈｚとし、その時のＲ２を精度よく設定したい場合は上記の係数を用いて（７）式より４Ｈｚでの２次抵抗Ｒ２求めることができる。 a1 = {− 19 · R2 (60) · R2 (35) + 343 · R2 (60) · R2 (10) −324 · R2 (35) · R2 (10)} /
{324 ・ R2 (60) -343 ・ R2 (35) +19 ・ R2 (10)}
a2 = {19 · R2 (60) · R2 (35) −28 · R2 (60) · R2 (10) + 9 · R2 (35) · R2 (10)} /
{100 • [324 • R2 (60) -343 • R2 (35) + 19 • R2 (10)]}
a3 = {− 9 · R2 (60) + 28 · R2 (35) −19 · R2 (10)} /
{100 • [324 • R2 (60) -343 • R2 (35) + 19 • R2 (10)]}
When the slip frequency at the rated torque of the induction motor to be used is 4 Hz and R2 at that time is desired to be set with high accuracy, the secondary resistance R2 at 4 Hz can be obtained from the equation (7) using the above coefficient.

また、図１３で示した例のように５Ｈｚ以下での２次抵抗はあまり変化しない。このような場合にはｆ＝０の値を使えばよい。その際は（７）式よりわかるようにＲ２（０）＝ａ１となり、ａ２やａ３を求める必要がなく演算の手間を省くことができる。   Further, as in the example shown in FIG. 13, the secondary resistance at 5 Hz or less does not change much. In such a case, a value of f = 0 may be used. In this case, as can be seen from the equation (7), R2 (0) = a1, and it is not necessary to obtain a2 and a3, so that it is possible to save the trouble of the calculation.

同様なことが漏れインダクタンスに関しても言える。   The same can be said for the leakage inductance.

このようにして求めた２次抵抗、漏れインダクタンスは図１の２次抵抗・漏れインダクタンス記憶手段１０８に記憶され実運転時に使用される。
＜相互インダクタンスを考慮したＲ２、Ｌの測定＞
図１に示す２次抵抗・漏れインダクタンス演算測定手段１０６により相互インダクタンスＭを考慮し、２次抵抗、漏れインダクタンスを精度よく演算する誘導電動機制御装置について述べる。図３は相互インダクタンスＭを考慮して２次抵抗、漏れインダクタンスを測定する誘導電動機制御装置を示すブロック図である。図４は誘導電動機１０２の１相分等価回路の説明図であり、図４（ａ）は相互インダクタンスＭを開放としたＬ型等価回路、図４（ｂ）は相互インダクタンスＭを考慮したＴ型等価回路を示す。 The secondary resistance and leakage inductance obtained in this way are stored in the secondary resistance / leakage inductance storage means 108 in FIG. 1 and used during actual operation.
<Measurement of R2 and L considering mutual inductance>
An induction motor control device for calculating the secondary resistance and the leakage inductance with high accuracy by considering the mutual inductance M by the secondary resistance / leakage inductance calculation and measurement means 106 shown in FIG. 1 will be described. FIG. 3 is a block diagram showing an induction motor control device for measuring a secondary resistance and a leakage inductance in consideration of a mutual inductance M. 4A and 4B are explanatory diagrams of a one-phase equivalent circuit of the induction motor 102. FIG. 4A is an L-type equivalent circuit in which the mutual inductance M is opened, and FIG. 4B is a T-type equivalent circuit in which the mutual inductance M is considered. An equivalent circuit is shown.

図３において１０６ａはＬ型等価回路により、所要の周波数における２次抵抗及び漏れインダクタンスを測定する２次抵抗・漏れインダクタンス演算測定手段、１０９は相互インダクタンスＭを考慮して２次抵抗、漏れインダクタンスを演算するＭ考慮２次抵抗・漏れインダクタンス演算測定手段である。   In FIG. 3, reference numeral 106a denotes an L-type equivalent circuit, and a secondary resistance / leakage inductance calculation measuring means for measuring a secondary resistance and a leakage inductance at a required frequency, and 109 a secondary resistance and a leakage inductance in consideration of a mutual inductance M. This is an M-consideration secondary resistance / leakage inductance calculation / measurement means.

次に、この誘導電動機制御装置の動作を説明する。図４（ａ）に示すＬ型等価回路において、相互インダクタンスＭを開放と近似した場合のインピーダンスＺｄは
Ｚｄ＝（Ｒ１＋Ｒ２ｄ）＋ｊω（Ｌ１ｄ＋Ｌ２ｄ） ・・・（９）
となる。ここで１次抵抗Ｒ１は予め正確に求めることができるが、２次抵抗と１次、２次漏れインダクタンスは相互インダクタンスＭを考慮しておらず実際の値とは異なるため添字のｄを付しそれぞれＲ２ｄ、Ｌ１ｄ、Ｌ２ｄと記した。ここで、ωは周波数ｆに対する周波数角速度である。 Next, the operation of the induction motor control device will be described. In the L-type equivalent circuit shown in FIG. 4A, the impedance Zd when the mutual inductance M is approximated to be open is Zd = (R1 + R2d) + jω (L1d + L2d) (9)
It becomes. Here, the primary resistance R1 can be accurately obtained in advance. However, since the secondary resistance and the primary and secondary leakage inductances are different from the actual values without considering the mutual inductance M, the subscript d is added. These are indicated as R2d, L1d, and L2d, respectively. Here, ω is a frequency angular velocity with respect to the frequency f.

これに対し相互インダクタンスＭを考慮した場合のＴ型等価回路は図４（ｂ）となり、インピーダンスＺ０は
Ｚ０＝（Ｒ１＋Ｍ2ω2Ｒ２／（Ｒ２2＋ω2ＬＬ２2））
＋ｊω｛Ｌ１＋Ｍ（Ｒ２2＋ω2Ｌ２ＬＬ２）／（Ｒ２2＋
ω2 ＬＬ２2）｝ ・・・（１０）
となる。ここでＬＬ２は２次自己インダクタンスでＬＬ２＝Ｍ＋Ｌ２であり、周知の方法により予め求められる。 T-type equivalent circuit when the hand considering the mutual inductance M is FIG 4 (b), and the impedance Z0 is Z0 = (R1 + M 2 ω 2 R2 / (R2 2 + ω 2 LL2 2))
+ Jω ｛L1 + M (R2 2 + ω 2 L2LL2) / (R2 2 +
ω 2 LL2 2 )｝ (10)
It becomes. Here, LL2 is a secondary self-inductance, LL2 = M + L2, and is obtained in advance by a known method.

まず、２次抵抗・漏れインダクタンス演算測定手段１０６ａは、相互インダクタンスＭを開放として近似した場合によく用いられる計算方法により、Ｒ２ｄおよびＬ１ｄ＋Ｌ２ｄを次のように求める。   First, the secondary resistance / leakage inductance calculation and measurement means 106a obtains R2d and L1d + L2d as follows using a calculation method often used when the mutual inductance M is approximated as being open.

Ｒ２ｄ ＝Ｐ／Ｉ2−Ｒ１ ・・・（１１）
Ｌ１ｄ＋Ｌ２ｄ＝Ｑ／ωＩ2 ・・・（１２）
ここでＰは有効電力、Ｑは無効電力、Ｉは相電流実効値である。
相電流実効値は瞬時電流から演算して求めることができるが、次のようにして簡単に求めることもできる。 R2d = P / I 2 -R1 ··· (11)
L1d + L2d = Q / ωI 2 (12)
Here, P is active power, Q is reactive power, and I is the phase current effective value.
Although the effective value of the phase current can be calculated from the instantaneous current, it can also be easily obtained as follows.

Ｉ2＝（Ｐ2＋Ｑ2）／（Ｖｐ2／２） ・・・（１３）
ここでＶｐは相電圧ピーク値であり、指令電圧が実電圧に等しければ容易に分かる定数である。１次、２次の漏れインダクタンスは等しいとし上記電流の式を使うと
Ｒ２ｄ＝Ｐ・Ｖｐ2／（２・（Ｐ2＋Ｑ2）） ・・・（１４）
Ｌ２ｄ＝Ｑ・Ｖｐ2／（４・ω・（Ｐ2＋Ｑ2）） ・・・（１５）
となる。 I 2 = (P 2 + Q 2) / (Vp 2/2) ··· (13)
Here, Vp is the peak value of the phase voltage, and is a constant that can be easily understood if the command voltage is equal to the actual voltage. Assuming that the primary and secondary leakage inductances are equal, using the above current equation, R2d = P · Vp 2 / (2 · (P 2 + Q 2 )) (14)
L2d = Q · Vp 2 / (4 · ω · (P 2 + Q 2 )) (15)
It becomes.

相互インダクタンスＭを考慮する場合も１次、２次漏れインダクタンスの値は等しいとおく、すなわちＬ１＝Ｌ２＝Ｌ。Ｍ考慮２次抵抗・漏れインダクタンス演算測定手段１０９は、２次抵抗・漏れインダクタンス演算測定手段１０６ａが相互インダクタンスＭを開放として近似して求めたＲ２ｄ、Ｌ２ｄから相互インダクタンスＭを考慮した２次抵抗Ｒ２、漏れインダクタンスＬを次のようにして求める。上記ＺｄとＺ０の比較より、
Ｒ２ｄ＝Ｍ2ω2Ｒ２／（Ｒ２2＋ω2ＬＬ２2 ） ・・・（１６）
Ｌ２ｄ・２＝Ｌ＋Ｍ（Ｒ２2＋ω2・Ｌ・ＬＬ２）／(Ｒ２2＋ω2ＬＬ２2)
・・（１７）
この２つの式より相互インダクタンスＭを考慮したＲ２、Ｌを求めるためには収束計算が必要となり、実使用においてはプログラミングが大変であるので、誘導電動機１０２の２次時定数Ｒ２／ＬＬ２が角周波数ωより十分小さいとして、（Ｒ２／ＬＬ２）／ω≒０等の近似を行い、相互インダクタンスＭを開放と近似したＲ２ｄ、Ｌ２ｄから相互インダクタンスＭを考慮した２次抵抗Ｒ２、漏れインダクタンスＬを次式により求める。 Even when the mutual inductance M is considered, the values of the primary and secondary leakage inductances are assumed to be equal, that is, L1 = L2 = L. The M-considered secondary resistance / leakage inductance calculation / measurement means 109 is a secondary resistance R2 which considers the mutual inductance M from the R2d and L2d obtained by the secondary resistance / leakage inductance calculation / measurement means 106a approximating the mutual inductance M as being open. , And the leakage inductance L is determined as follows. From the above comparison between Zd and Z0,
R2d = M 2 ω 2 R 2 / (R 2 2 + ω 2 LL 2 2 ) (16)
L2d · 2 = L + M (R2 2 + ω 2 · L·LL2) / (R2 2 + ω 2 LL2 2 )
・ ・ (17)
Convergence calculation is required to obtain R2 and L in consideration of the mutual inductance M from these two equations, and programming is difficult in actual use. Therefore, the secondary time constant R2 / LL2 of the induction motor 102 is determined by the angular frequency. Assuming that the value is sufficiently smaller than ω, an approximation such as (R2 / LL2) / ω ≒ 0 is performed, and a secondary resistance R2 and a leakage inductance L, which take the mutual inductance M into consideration, from R2d and L2d which approximate the mutual inductance M to open are expressed by the following equations. Ask by

Ｒ２＝Ｒ２ｄ・ＬＬ２2／（ＬＬ２−Ｌ２ｄ）2 ・・・（１８）
Ｌ＝Ｌ２ｄ＋（２・Ｌ２ｄ−ＬＬ２）・Ｒ２ｄ2／（２・ω2・ＬＬ２2）
・・・（１９）
（１８）（１９）式を使うことにより、所要の周波数ｆにおける、相互インダクタンスＭを考慮した２次抵抗Ｒ２及び漏れインダクタンスＬを求めることができ、従来のように相互インダクタンスＭを開放と近似していたことによる誤差を無くすことができる。
＜単相電圧印加によるＲ２、Ｌの測定＞
２次抵抗Ｒ２と漏れインダクタンスＬを求める際に必要となる有効電力Ｐ及び無効電力Ｑを単相電圧を印加して演算する誘導電動機制御装置について説明する。図５は単相電圧を印加して有効電力Ｐ及び無効電力Ｑを演算する試験電源指令手段１０４と電力演算手段１０５を示すブロック図である。図５において、１０４は試験用の単相電圧を印加する試験電源指令手段であり、印加する電圧の角周波数ωを積分し位相θを求める積分器２０１、ＶｐＳＩＮ（θ）を出力するＳＩＮ関数発生器２０２、ゲイン２０３を有している。１０５は電力演算手段であり、位相θにπ／２を減算する加算器３０２、ＳＩＮ関数発生器３０３、３０４、乗算器３０５、３０６、フィルタ３０７、３０８を有している。 R2 = R2d · LL2 2 / (LL2-L2d) 2 (18)
L = L2d + (2 · L2d−LL2) · R2d 2 / (2 · ω 2 · LL2 2 )
... (19)
By using the equations (18) and (19), it is possible to obtain the secondary resistance R2 and the leakage inductance L in consideration of the mutual inductance M at the required frequency f. It is possible to eliminate the error caused by the error.
<Measurement of R2 and L by applying single-phase voltage>
An induction motor control device that calculates the active power P and the reactive power Q required for obtaining the secondary resistance R2 and the leakage inductance L by applying a single-phase voltage will be described. FIG. 5 is a block diagram showing the test power supply command means 104 and the power calculation means 105 for calculating the active power P and the reactive power Q by applying a single-phase voltage. In FIG. 5, reference numeral 104 denotes test power supply commanding means for applying a test single-phase voltage, an integrator 201 for integrating the angular frequency ω of the applied voltage to obtain a phase θ, and generating a SIN function for outputting VpSIN (θ). And a gain 203. Reference numeral 105 denotes a power calculation unit, which includes an adder 302 for subtracting π / 2 from the phase θ, SIN function generators 303 and 304, multipliers 305 and 306, and filters 307 and 308.

次に、この誘導電動機制御装置の動作を説明する。Ｕ相、Ｖ相、Ｗ相の３相を持つ誘導電動機１０２に対して、Ｕ相にｖｓ＝Ｖｐ・SIN(θ）、Ｖ相に−ｖｓ、Ｗ相に０を印加する方法と、Ｕ相にｖｓ＝Ｖｐ・SIN(θ）、Ｖ相、Ｗ相に−ｖｓ／２を印加する方法などがある。ここでＶｐは電圧ピーク値で一定の値とする。いずれの方法でも誘導電動機は回転することはない。上述のように試験電源指令手段１０４が単相印加した場合、検出した電流ｉｓはｉｓ＝Ｉｐ・SIN(θ−φ）と表される。（φは電流と電圧の位相差。）この時、瞬時有効電力ｖｓ・ｉｓは
ｖｓ・ｉｓ＝Ｖｐ・SIN(θ）・Ｉｐ・SIN(θ−φ）
＝Ｖｐ・Ｉｐ・COS(φ）／２−Ｖｐ・Ｉｐ・COS(２θ−φ）
・・・（２０）
となり、（２０）式の第１項目は有効電力Ｐで直流量である。２項目は印加された周波数の２倍の周波数で振動する成分である。このように、乗算器３０５より出力される瞬時有効電力ｖｓ・ｉｓには有効電力Ｐと印加された周波数の２倍の周波数成分が含まれる。 Next, the operation of the induction motor control device will be described. For the induction motor 102 having three phases of U phase, V phase and W phase, a method of applying vs = Vp · SIN (θ) to the U phase, applying −vs to the V phase, and applying 0 to the W phase, And the method of applying -vs / 2 to the V phase and the W phase. Here, Vp is a constant voltage peak value. In either case, the induction motor does not rotate. When the test power supply command unit 104 applies a single phase as described above, the detected current is is expressed as is = Ip · SIN (θ−φ). (Φ is the phase difference between current and voltage.) At this time, the instantaneous active power vs. is: vs.is = Vp · SIN (θ) · Ip · SIN (θ−φ)
= Vp · Ip · COS (φ) / 2−Vp · Ip · COS (2θ−φ)
... (20)
The first item in the equation (20) is the active power P and the DC amount. The two items are components that vibrate at twice the frequency of the applied frequency. As described above, the instantaneous active power vs.is output from the multiplier 305 includes the active power P and a frequency component twice as high as the applied frequency.

また、ＳＩＮ関数発生器３０４からはｖｓより９０度（π／２）だけ位相の遅れた電圧ｖｃが出力され、
ｖｃ＝Ｖｐ・SIN(θ−π／２）＝−Ｖｐ・COS(θ）となる。 Further, the SIN function generator 304 outputs a voltage vc delayed in phase by 90 degrees (π / 2) from vs,
vc = Vp · SIN (θ−π / 2) = − Vp · COS (θ)

この時、乗算器３０６より出力される瞬時無効電力ｖｃ・ｉｓは
ｖｃ・ｉｓ＝−Ｖｐ・COS(θ）・Ｉｐ・SIN(θ−φ）
＝Ｖｐ・Ｉｐ・SIN(φ）／２−Ｖｐ・Ｉｐ・SIN(２θ−φ）
・・・（２１）
となり、（２１）式の第１項目は無効電力Ｑで直流量である。２項目は印加された周波数の２倍の周波数で振動する成分である。このように、瞬時無効電力ｖｃ・ｉｓには無効電力Ｑと印加された周波数の２倍の周波数成分が含まれる。 At this time, the instantaneous reactive power vc · is output from the multiplier 306 is vc · is = −Vp · COS (θ) · Ip · SIN (θ−φ)
= Vp · Ip · SIN (φ) / 2−Vp · Ip · SIN (2θ−φ)
... (21)
The first item of the equation (21) is the reactive power Q and the DC amount. The two items are components that vibrate at twice the frequency of the applied frequency. As described above, the instantaneous reactive power vc · is includes the reactive power Q and a frequency component twice as high as the applied frequency.

実際にはこの２倍の周波数成分の他にも単相印加により電流が歪むことによる高調波成分やキャリヤ周波数によるリップルが含まれる。よって有効電力Ｐ、無効電力Ｑは瞬時有効電力ｖｓ・ｉｓと瞬時無効電力ｖｃ・ｉｓで印加された周波数の２倍の周波数成分とそれ以上の周波数成分をフィルタ３０７、３０８によりフィルタリングすることにより求める。   Actually, in addition to the double frequency component, a harmonic component due to current distortion due to single-phase application and a ripple due to a carrier frequency are included. Therefore, the active power P and the reactive power Q are obtained by filtering the frequency component twice as high as the frequency applied by the instantaneous active power vs.is and the instantaneous reactive power vc.is and higher frequency components by the filters 307 and 308. .

このフィルタ３０７、３０８としては印加された周波数の２倍の周波数成分を十分減衰できるようなローパスフィルタとすればよい。しかしその場合は２倍の周波数成分を十分減衰できるようなローパスフィルタとする必要がある。そうした場合、瞬時有効電力ｖｓ・ｉｓおよび瞬時無効電力ｖｃ・ｉｓから有効電力Ｐ、無効電力Ｑを求めるために時間がかからないようにフィルタの次数を大きくする必要がある。   The filters 307 and 308 may be low-pass filters that can sufficiently attenuate the frequency component twice the applied frequency. However, in that case, it is necessary to use a low-pass filter capable of sufficiently attenuating twice the frequency component. In such a case, it is necessary to increase the order of the filter so that it does not take much time to obtain the active power P and the reactive power Q from the instantaneous active power vs.is and the instantaneous reactive power vc.is.

より効果的なフィルタリングの方法としては、２倍の周波数成分をノッチフィルタにより減衰させ、それより高い周波数成分は低次のローパスフィルタにより減衰するようにフィルタ３０７、３０８を構成する。このようにすると、それほど時間をかけずに、有効電力Ｐ、無効電力Ｑを求めることができる。   As a more effective filtering method, the filters 307 and 308 are configured so that a double frequency component is attenuated by a notch filter, and higher frequency components are attenuated by a low-order low-pass filter. In this way, the active power P and the reactive power Q can be obtained without taking much time.

上述のように１０Ｈｚ、３５Ｈｚ、６０Ｈｚと周波数を変えて２次抵抗及びインダクタンスの測定を行う場合はノッチフィルタのゲインとして、１０Ｈｚ印加の時は２０Ｈｚを減衰するようなゲイン、３５Ｈｚ印加の時は７０Ｈｚを減衰するようなゲイン、６０Ｈｚ印加の時は１２０Ｈｚを減衰するようなゲインを使い、低次ローパスフィルタはどの周波数でも同じゲインを使用する。このようにすれば、図１２の例で述べたように平均して２次抵抗インダクタンスを求める場合のように、印加する周波数により平均を取る時間を変える必要はない。このように構成した場合、実験では約１秒程度でその有効電力Ｐ、無効電力Ｑを求めることができた。
＜指定電流でのＲ２、Ｌの測定＞
上述のように有効電力Ｐ、無効電力Ｑを求め、２次抵抗、漏れインダクタンスを計算する際、その電流値によって２次抵抗、漏れインダクタンスは変化する。そこで、定格電流を流してその電流値での２次抵抗、漏れインダクタンスの値を求めるため、指定した電流で測定することが必要となる。図６は、この発明の誘導電動機制御装置における、指定電流で有効電力Ｐ、無効電力Ｑを演算する試験電源指令手段と電力演算手段を示すブロック図であり、図中、図５と同一符号は同一又は相当部分を示す。図６において、２０４はＳＩＮ関数発生器２０２が印加する電圧を一定傾きのスロープをかけて上昇させてその際の検出電流を２乗して時間平均した値が指定した電流実効値ｉｒｍｓの２乗の１／３となったときに電圧の上昇を止める電圧判定手段であり、試験電源指令手段１０４ａに設けられている。電圧判定手段２０４が電圧の上昇を止めた時、電流は指定した電流実効値ｉｒｍｓとなる。 As described above, when the secondary resistance and inductance are measured by changing the frequency to 10 Hz, 35 Hz, and 60 Hz, the gain of the notch filter is set to attenuate 20 Hz when 10 Hz is applied, and 70 Hz when 35 Hz is applied. The gain is used to attenuate the frequency, and when 60 Hz is applied, the gain is attenuated to 120 Hz. The low-order low-pass filter uses the same gain at any frequency. In this case, it is not necessary to change the averaging time depending on the frequency to be applied as in the case of obtaining the secondary resistance inductance by averaging as described in the example of FIG. With such a configuration, the active power P and the reactive power Q could be obtained in about 1 second in an experiment.
<Measurement of R2 and L at specified current>
As described above, when the active power P and the reactive power Q are obtained and the secondary resistance and the leakage inductance are calculated, the secondary resistance and the leakage inductance change depending on the current value. Therefore, in order to determine the value of the secondary resistance and the leakage inductance at the current value by flowing the rated current, it is necessary to perform measurement at the specified current. FIG. 6 is a block diagram showing test power supply command means and power calculation means for calculating active power P and reactive power Q at a specified current in the induction motor control device of the present invention. Indicates the same or corresponding parts. In FIG. 6, reference numeral 204 denotes a value obtained by raising the voltage applied by the SIN function generator 202 with a constant slope, squaring the detected current at that time, and averaging the time, and squaring the specified current effective value irms. Is a voltage judging means for stopping the rise of the voltage when the voltage becomes 1/3 of the value, and is provided in the test power supply command means 104a. When the voltage determination unit 204 stops increasing the voltage, the current becomes the specified current effective value irms.

次に、この誘導電動機制御装置の動作原理を説明する。ＳＩＮ関数発生器２０２は電圧をｖ＝Ｖｐ・SIN(θ）として、このＶｐをＶｐ＝ｋｖ・ｔで表されるように時間ｔにより一定の傾きｋｖで上昇するようにする。そうすると流れる電流ｉもｉ＝ｋｉ・ｔ・SIN(θ）となり一定の傾きｋｉで上昇する。このｋｉ・ｔが√２・ｉｒｍｓと等しくなった時刻ｔで電圧判定手段２０４が電圧の上昇を止めれば良いことが判る。ここで電流ｉを２乗して時間平均した値をＸとすると、
Ｘ＝１／ｔ・∫ｉ2ｄｔ
＝１／ｔ・∫（ｋｉ・ｔ・SIN(θ））2ｄｔ ・・・（２２）
このＸは振動成分を無視すると
Ｘ＝（ｋｉ・ｔ）2／６ ・・・（２３）
となる。ｋｉ・ｔ＝√２・ｉｒｍｓとなる瞬間に電流ｉは指定の実効値ｉｒｍｓになるのでＸ＝ｉｒｍｓ2 ／３となった時に電圧の上昇を止めてやればよいことが判る。このようにすれば、指定した電流実効値で有効電力Ｐ、無効電力Ｑの測定をすることができ、測定する電流値による２次抵抗、漏れインダクタンスの推定ばらつきを無くすことができる。このように、検出電流の２乗の時間平均を用いる他に検出電流のピーク値が指定した電流実効値ｉｒｍｓの√２倍になったとき電圧上昇を止めてもよいが、この方法ではまだ電流が設定値になっていないにもかかわらず電流の誤検出によりスロープを止めてしまう可能性が高く、電流検出時のノイズ等に十分注意をはらわなければならない。
＜電流サンプリング周期、演算遅れの補正＞
有効電力Ｐ、無効電力Ｑを求める際、式（２０）、式（２１）で述べたように検出電流と指令電圧を使用して求める方法では、検出電流と指令電圧の電流サンプリング周期や演算遅れ時間の影響が有効電力Ｐ、無効電力Ｑの演算測定値に影響する。そこでその電流サンプリング周期や演算遅れ時間を補正する誘導電動機制御装置を説明する。 Next, the operation principle of the induction motor control device will be described. The SIN function generator 202 sets the voltage as v = Vp · SIN (θ), and increases the Vp at a constant gradient kv with time t as represented by Vp = kv · t. Then, the flowing current i also becomes i = ki · t · SIN (θ) and rises at a constant gradient ki. At time t when ki · t becomes equal to √2 · irms, it can be seen that voltage determination means 204 should stop increasing the voltage. Here, assuming that a value obtained by averaging the current i squared and time averaged is X,
X = 1 / t∫i 2 dt
= 1 / t · ∫ (ki · t · SIN (θ)) 2 dt (22)
X = This X ignores the vibration component (ki · t) 2/6 ··· (23)
It becomes. current i at the moment which is a ki · t = √2 · irms it can be seen that may do it to stop the rise of the voltage when it becomes the X = irms 2/3 since the designation of the effective value irms. In this way, the active power P and the reactive power Q can be measured at the specified effective current value, and the estimated variation in the secondary resistance and the leakage inductance due to the measured current value can be eliminated. As described above, in addition to using the time average of the square of the detection current, the voltage rise may be stopped when the peak value of the detection current becomes √2 times the specified current effective value irms. It is highly probable that the slope will be stopped due to erroneous detection of the current even though is not the set value, and sufficient attention must be paid to noise and the like at the time of current detection.
<Correction of current sampling cycle and calculation delay>
When calculating the active power P and the reactive power Q by using the detected current and the command voltage as described in the equations (20) and (21), the current sampling period and the calculation delay of the detected current and the command voltage are determined. The influence of time affects the measured values of the active power P and the reactive power Q. Therefore, an induction motor control device for correcting the current sampling period and the calculation delay time will be described.

図７はこの発明の誘導電動機制御装置における試験電源指令手段及び電力演算手段を示すブロック図であり、図中、図５と同一符号は同一又は相当部分を示す。図７において、１０４は試験電源指令手段あり、印加する電圧の角周波数ωを積分器２０１により積分し位相θを求め、ＳＩＮ関数発生器２０２によりＶｐ・ＳＩＮ（θ）の指令を作成する。その指令は上述の＜単相電圧印加によるＲ２、Ｌの測定＞の項で述べたようにＵ相に直接印加され、Ｖ相、Ｗ相にはゲイン２０３により−１／２倍されて印加される。１０５ａは電力演算手段であり、積分器２０１から出力された位相θは加算器３０１によって補正位相θｘと加算される。この補正位相θｘは電流サンプリング周期Ｔｓと演算遅れ時間Ｔｃａｌによって次のように計算される。   FIG. 7 is a block diagram showing test power supply command means and power calculation means in the induction motor control device of the present invention. In the drawing, the same reference numerals as those in FIG. 5 denote the same or corresponding parts. In FIG. 7, reference numeral 104 denotes a test power supply commanding means. The integrator 201 integrates the angular frequency ω of the applied voltage to obtain a phase θ, and the SIN function generator 202 creates a command of Vp · SIN (θ). The command is applied directly to the U-phase as described in the above section <Measurement of R2 and L by applying single-phase voltage>, and applied to the V-phase and W-phase by multiplying -1/2 times by the gain 203. You. Numeral 105a denotes a power calculating means, and the phase θ output from the integrator 201 is added by the adder 301 to the correction phase θx. The correction phase θx is calculated as follows based on the current sampling period Ts and the operation delay time Tcal.

θｘ＝−θ・（Ｔｓ／２＋Ｔｃａｌ） ・・・（２４）
電流検出器１０３に遅れ時間Ｔｉｄがある場合はその分も加算して
θｘ＝−θ・（Ｔｓ／２＋Ｔｃａｌ＋Ｔｉｄ） ・・・（２５）
となる。加算器３０１は位相θと補正位相θｘを加算して位相θ１を出力する。そしてその位相θ１は加算器３０２でπ／２が減算されて位相θ２となる。位相θ１はＳＩＮ関数発生器３０３に入力されＶｐ・SIN(θ１)となる。また位相θ２はＳＩＮ関数発生器３０４に入力されてＶｐ・ SIN（θ２）となる。それぞれは乗算器３０５と３０６により電流ｉｓと乗算されフィルタ３０７と３０８を介して有効電力Ｐ、無効電力Ｑとなる。 θx = −θ · (Ts / 2 + Tcal) (24)
If the current detector 103 has a delay time Tid, the delay time Tid is also added, and θx = −θ · (Ts / 2 + Tcal + Tid) (25)
It becomes. The adder 301 adds the phase θ and the correction phase θx to output a phase θ1. Then, the adder 302 subtracts π / 2 from the phase θ1 to obtain a phase θ2. The phase θ1 is input to the SIN function generator 303 and becomes Vp · SIN (θ1). The phase θ2 is input to the SIN function generator 304 and becomes Vp · SIN (θ2). Each is multiplied by the current is by the multipliers 305 and 306, and becomes the active power P and the reactive power Q via the filters 307 and 308.

このように補正位相θｘを考慮して有効電力Ｐ、無効電力Ｑを演算測定することにより、電流サンプリング周期や演算遅れ時間の影響を補正することができ、サンプリング周期、演算遅れ時間によらず精度よく有効電力Ｐ、無効電力Ｑを求めることができ、結果的に高精度に２次抵抗、漏れインダクタンスを求めることができる。   As described above, by calculating and measuring the active power P and the reactive power Q in consideration of the correction phase θx, it is possible to correct the influence of the current sampling period and the operation delay time, and to obtain the accuracy regardless of the sampling period and the operation delay time. Active power P and reactive power Q can be obtained well, and as a result, secondary resistance and leakage inductance can be obtained with high accuracy.

実施の形態２．
図８は実施の形態２による誘導電動機制御装置を示すブロック図であり、この誘導電動機制御装置は誘導電動機１０２の起動直前（以下、起動時と称す。）の短時間に直流電流を流しその間に１次抵抗、２次抵抗を求めるものである。 Embodiment 2 FIG.
FIG. 8 is a block diagram showing an induction motor control device according to the second embodiment. The induction motor control device supplies a direct current for a short time immediately before starting the induction motor 102 (hereinafter, referred to as starting). The primary resistance and the secondary resistance are determined.

図８おいて、インバータ１０１から誘導電動機１０２へ流れる電流を電流検出器１０３で検出する。４０１は電流を制御する電流制御手段であり、ここでは誘導電動機１０２への電流が直流電流となるように制御する。   In FIG. 8, a current flowing from an inverter 101 to an induction motor 102 is detected by a current detector 103. Reference numeral 401 denotes current control means for controlling current, and here, control is performed such that the current to the induction motor 102 becomes a DC current.

この電流制御手段４０１は次のように構成される。４０２は検出した３相分の電流を励磁分電流ｉｄに変換する電流座標変換器、４０３はその励磁分電流ｉｄと指令電流Ｉｄｃの差を取る減算器、４０４はその差を比例積分演算するＰＩ制御器であり、ＰＩ制御器４０４は励磁電圧指令ｖｄを出力する。４０５は励磁電圧指令ｖｄを３相電圧指令ｖｕ，ｖｖ，ｖｗに変換する電圧座標変換器である。このように電流制御手段４０１は起動直後からわずかな時間だけ励磁分電流のみを位相を固定した直流にて制御するよう構成する。   This current control means 401 is configured as follows. Reference numeral 402 denotes a current coordinate converter for converting the detected currents of the three phases into an excitation current id, 403 a subtractor for obtaining a difference between the excitation current id and the command current Idc, and 404 a PI for performing a proportional integral calculation of the difference. A PI controller 404 outputs an excitation voltage command vd. A voltage coordinate converter 405 converts the excitation voltage command vd into three-phase voltage commands vu, vv, vw. As described above, the current control means 401 is configured to control only the excitation component current with a DC having a fixed phase for a short time immediately after the start.

次にこの直流電流を流している期間に１次抵抗、２次抵抗を推定する抵抗推定器について説明する。   Next, a resistance estimator for estimating the primary resistance and the secondary resistance while the DC current is flowing will be described.

５０１は抵抗推定器であり、次のように構成される。５０２は検出した励磁分電流ｉｄから誘導電動機の電気的定数を使って励磁電圧推定器ｖｄ＾を求める電圧推定器、５０３は励磁電圧指令ｖｄと電圧推定器５０２の出力である励磁電圧推定値ｖｄ＾の偏差を取る減算器、５０４はその偏差を積分する積分器、５０５は切替スイッチで、この切替スイッチ５０５は起動直後からある一定期間はａ側に接続され（以下、期間ａと称す。）、その後の一定期間はｂ側に接続される（以下、期間ｂと称す。）。５０６は期間ａでの推定ゲインである。このとき推定ゲイン５０６の出力は１次抵抗と２次抵抗の和（Ｒ１ａ＋Ｒ２ａ）となる。５０７は（Ｒ１ａ＋Ｒ２ａ）を適当な比ｋ１で分離して１次抵抗Ｒ１ａを求めるゲインである。このゲインｋｌは０以上１以下とする。また、５０８は（Ｒ１ａ＋Ｒ２ａ）から２次抵抗Ｒ２ａを求めるゲインであり、１−ｋ１とする。５０９は期間ａが終了した時点でこの（Ｒ１ａ＋Ｒ２ａ）を記憶する記憶器である。   Reference numeral 501 denotes a resistance estimator, which is configured as follows. Reference numeral 502 denotes a voltage estimator that obtains an excitation voltage estimator vd ＾ from the detected excitation current id by using an electric constant of the induction motor. Reference numeral 503 denotes an excitation voltage command vd and an excitation voltage estimation value vd which is an output of the voltage estimator 502. 504 is an integrator for integrating the deviation, 505 is a changeover switch, and the changeover switch 505 is connected to the a side for a certain period immediately after startup (hereinafter, referred to as period a). , For a certain period thereafter (hereinafter, referred to as period b). 506 is an estimated gain in the period a. At this time, the output of the estimation gain 506 is the sum of the primary resistance and the secondary resistance (R1a + R2a). Reference numeral 507 denotes a gain for separating the primary resistance R1a by separating (R1a + R2a) by an appropriate ratio k1. This gain kl is set to 0 or more and 1 or less. Reference numeral 508 denotes a gain for calculating the secondary resistance R2a from (R1a + R2a), which is 1-k1. A storage unit 509 stores (R1a + R2a) when the period a ends.

つづいて、切替スイッチ５０５がｂ側へ切り替わると推定ゲイン５１０により１次抵抗Ｒ１ｂが推定される。このとき２次抵抗は記憶器５０９で記憶された（Ｒ１ａ＋Ｒ２ａ）から１次抵抗Ｒ１ｂを減算器５１１により減算して求める。このようにして求めた２次抵抗をＲ２ｂとする。前述のように電圧推定器５０２は誘導電動機１０２の電気的定数の内１次、２次漏れインダクタンスＬ１、Ｌ２、相互インダクタンスＭを一定として計算しており、切替スイッチ５１２、５１３から期間ａではＲ１ａ、Ｒ２ａを、期間ｂではＲ１ｂ，Ｒ２ｂをＲ１、Ｒ２としてフィードバックして電圧推定器５０２で励磁電圧推定値ｖｄ＾を出力し、励磁電圧推定値ｖｄ＾と励磁電圧指令ｖｄの偏差が零又は所定の値以下となったときに、抵抗Ｒ１、Ｒ２が真値に近づいたとしている。   Subsequently, when the changeover switch 505 is switched to the b side, the primary resistance R1b is estimated by the estimated gain 510. At this time, the secondary resistance is obtained by subtracting the primary resistance R1b from (R1a + R2a) stored in the storage unit 509 by the subtractor 511. The secondary resistance obtained in this way is defined as R2b. As described above, the voltage estimator 502 calculates the primary and secondary leakage inductances L1 and L2 and the mutual inductance M among the electric constants of the induction motor 102 as constant, and calculates R1a in the period a from the changeover switches 512 and 513. , R2a and R1b and R2b are fed back as R1 and R2 in the period b, and the voltage estimator 502 outputs the excitation voltage estimation value vd ＾. The deviation between the excitation voltage estimation value vd ＾ and the excitation voltage command vd is zero or a predetermined value. It is assumed that the resistances R1 and R2 have approached the true value when the value has become equal to or less than the value of.

そして最終的に記憶装置５１５に期間ａと期間ｂが終了した時点での１次抵抗、２次抵抗Ｒ１、Ｒ２を記憶させて、実際に運転する際の定数として使用する。上述の例では電圧推定器５０２の出力である励磁電圧推定値ｖｄ＾と励磁電圧指令ｖｄの偏差を積分器５０４により積分していたが、積分の代わりに比例積分を行うようにしてもよい。
＜抵抗の推定原理＞
次に、この抵抗の推定原理を説明する。上述のように電流制御系が構成されているので、起動時にステップ状の電流が流れる。期間ａではそのステップ状の電流が加わる過渡的な高周波領域であるため相互インダクタンスＭには電流はほとんど流れず図９（ａ）に示すように２次側に電流が流れる。そのため電流が流れるルートにある抵抗分である１次抵抗と２次抵抗の和が求まる。続いて期間ｂでは電流がほぼ定常となっている直流領域のため相互インダクタンスＭは短絡と等価となり、その電流は図９（ｂ）に示すように１次側のみ流れる。そのため電流が流れるルートにある抵抗分である１次抵抗のみが求まる。期間ｂでも２次側に電流が流れるが、その電流値はわずかであること、ある程度の精度での抵抗値は期間ａで求まっているのでその分がうまく補正できることにより完全に２次側に回り込まなくなるまで推定し続ける必要はなく短時間で本推定は完了できる。 Finally, the primary resistance and the secondary resistances R1 and R2 at the end of the period a and the period b are stored in the storage device 515, and are used as constants for actual operation. In the above-described example, the difference between the excitation voltage estimation value vd # output from the voltage estimator 502 and the excitation voltage command vd is integrated by the integrator 504. However, proportional integration may be performed instead of integration.
<Estimation principle of resistance>
Next, the principle of estimating the resistance will be described. Since the current control system is configured as described above, a step-like current flows at startup. In the period a, since the current is in a transitional high-frequency region in which the step-like current is applied, almost no current flows through the mutual inductance M, and the current flows to the secondary side as shown in FIG. Therefore, the sum of the primary resistance and the secondary resistance, which is the resistance in the route where the current flows, is obtained. Subsequently, in the period b, the mutual inductance M is equivalent to a short circuit due to the DC region in which the current is substantially steady, and the current flows only on the primary side as shown in FIG. 9B. Therefore, only the primary resistance, which is the resistance in the route where the current flows, is obtained. Although the current flows to the secondary side also in the period b, the current value is small, and the resistance value with a certain degree of accuracy is obtained in the period a. It is not necessary to continue estimation until it disappears, and this estimation can be completed in a short time.

このように１次抵抗と２次抵抗の和が求まり、さらに１次抵抗が求まるので結果的に１次抵抗と２次抵抗両方が求まる。   As described above, the sum of the primary resistance and the secondary resistance is obtained, and further the primary resistance is obtained. As a result, both the primary resistance and the secondary resistance are obtained.

次に、抵抗推定器５０１は、検出した電流から誘導電動機の電気的定数を用いて電圧を演算し励磁電圧推定値ｖｄ＾を求め、その励磁電圧推定値ｖｄ＾と励磁電圧指令ｖｄとの偏差を積分して抵抗を推定しているので、この推定原理を簡単に説明する。   Next, the resistance estimator 501 calculates a voltage from the detected current using the electrical constant of the induction motor to obtain an excitation voltage estimation value vd ＾, and calculates a deviation between the excitation voltage estimation value vd ＾ and the excitation voltage command vd. Is integrated to estimate the resistance, so the principle of this estimation will be briefly described.

実際の誘導電動機１０２の電流と電圧は励磁分電流ｉｄ，励磁電圧指令ｖｄとして判っており、その関係は誘導電動機１０２の電気的定数によって表される。   The actual current and voltage of the induction motor 102 are known as the excitation component current id and the excitation voltage command vd, and the relationship is represented by the electrical constant of the induction motor 102.

ここで、その励磁分電流ｉｄからその時点でわかっている誘導電動機１０２の電気的定数を用いて励磁電圧推定値ｖｄ＾を推定計算すると、判っている電気的定数が誤差を持つぶんだけその励磁電圧推定値ｖｄ＾は励磁電圧指令ｖｄから誤差を持つ。そこで、電気的定数の内１次、２次漏れインダクタンスＬ１、Ｌ２、相互インダクタンスＭは温度上昇にかかわらず一定としているので、励磁電圧推定値ｖｄ＾と励磁電圧指令ｖｄの偏差を取って積分した値により電圧推定器５０２で使用している抵抗値Ｒ１，Ｒ２を操作してやることにより次第に励磁電圧推定値ｖｄ＾と励磁電圧指令ｖｄは一致するようになる。   Here, when the excitation voltage estimation value vd ＾ is estimated and calculated from the excitation component current id using the electric constant of the induction motor 102 known at that time, the excitation voltage is estimated to have an error as far as the known electric constant has an error. Voltage estimation value vd # has an error from excitation voltage command vd. Therefore, among the electrical constants, the primary and secondary leakage inductances L1 and L2 and the mutual inductance M are constant irrespective of the temperature rise. Therefore, the deviation between the excitation voltage estimation value vd ＾ and the excitation voltage command vd is integrated. By manipulating the resistance values R1 and R2 used in the voltage estimator 502 according to the value, the excitation voltage estimated value vd # and the excitation voltage command vd gradually become consistent.

このようにして励磁電圧推定値ｖｄ＾と励磁電圧指令ｖｄが一致した時点では電圧推定器５０２で使用しているフィードバック抵抗値Ｒ１、Ｒ２が真値に一致していることになる。   In this way, when the exciting voltage estimated value vd # matches the exciting voltage command vd, the feedback resistance values R1 and R2 used in the voltage estimator 502 match the true values.

図８では電圧推定器５０２を用いていたが同様なことが電流推定器を用いても構成できる。   Although the voltage estimator 502 is used in FIG. 8, the same can be achieved by using a current estimator.

図１０は電流推定器を用いて構成した誘導電動機制御装置を示すブロック図であり、図中、図８と同一符号は同一又は相当部分を示す。図１０において、５０１ｉは抵抗推定器、５０２ｉは電流推定器である。電流推定器５０２ｉには励磁電圧指令ｖｄが入力されその電圧値より誘導電動機１０２の電気的定数を用い電流推定値Ｉｄ＾を推定計算する。その電流推定値Ｉｄ＾と実電流である励磁分電流ｉｄの偏差を取って積分して抵抗推定値を求めるのは電圧推定器５０２を用いた場合と同様である。   FIG. 10 is a block diagram showing an induction motor control device configured using a current estimator. In the drawing, the same reference numerals as those in FIG. 8 indicate the same or corresponding parts. In FIG. 10, 501i is a resistance estimator, and 502i is a current estimator. The excitation voltage command vd is input to the current estimator 502i, and the current estimation value Id # is estimated and calculated from the voltage value using the electrical constant of the induction motor 102. The difference between the current estimated value Id # and the excitation current id which is the actual current is obtained and integrated to obtain a resistance estimated value in the same manner as when the voltage estimator 502 is used.

上述の構成において実際に抵抗を推定した結果を図１１に示す。図１１は、誘導電動機として、３．７ＫＷのものを用い、期間ａの時間を０．０１ｓｅｃ，期間ｂの時間を０．０４ｓｅｃとし合計０．０５ｓｅｃで推定したものを示し、期間ａで用いる１次抵抗と２次抵抗の和から１次抵抗を求める適当な比ｋ１を０．５としている。したがって１次抵抗と２次抵抗の和から２次抵抗を求める比１−ｋ１も０．５となり期間ａでは１次抵抗＝２次抵抗となっている。実際の１次抵抗の真値は０．５３２Ωで２次抵抗の真値は０．４００Ωであるが、期間ａでは適当な比を入れておけばよい。期間ｂでは１次抵抗を推定し、期間ａで求めた和から推定した１次抵抗を減じて２次抵抗を求めている。期間ａと期間ｂをあわせて０．０５ｓｅｃという短時間でほぼ収束している。図１１では２次抵抗推定値はわずかに大きめに推定されているが起動時のトルク不足を解消するためにはまったく問題のないレベルである。
＜測定時間の短縮及び誤差の補正＞
上述のように起動時の抵抗の推定は短時間に完了できるが、実際にいろいろな条件で実験すると次のような問題があった。
（１） 推定器間ａで用いる適当な比ｋ１が大きく異なっていると推定時間が長くなる。
（２） 起動トルク不足の問題はないものの、前述のようにわずかに誤差を含むため起動後の実運転時に速度やトルクの演算に多少誤差を含む。 FIG. 11 shows the result of actually estimating the resistance in the above configuration. FIG. 11 shows an example in which a 3.7 KW induction motor is used, and the time of the period a is estimated at 0.01 sec and the time of the period b is estimated at 0.04 sec for a total of 0.05 sec. An appropriate ratio k1 for obtaining the primary resistance from the sum of the secondary resistance and the secondary resistance is set to 0.5. Therefore, the ratio 1-k1 for obtaining the secondary resistance from the sum of the primary resistance and the secondary resistance is also 0.5, and the primary resistance = the secondary resistance in the period a. Although the actual true value of the primary resistance is 0.532Ω and the true value of the secondary resistance is 0.400Ω, an appropriate ratio may be set in the period a. In the period b, the primary resistance is estimated, and the secondary resistance is obtained by subtracting the estimated primary resistance from the sum obtained in the period a. The period a and the period b are almost converged in a short time of 0.05 sec. In FIG. 11, the secondary resistance estimated value is estimated to be slightly larger, but at a level that does not cause any problem in order to eliminate the insufficient torque at the time of starting.
<Reduction of measurement time and correction of error>
As described above, the estimation of the resistance at the time of starting can be completed in a short time, but the following problems have been encountered when experiments are actually performed under various conditions.
(1) If the appropriate ratio k1 used between the estimators a is significantly different, the estimation time becomes longer.
(2) Although there is no problem of insufficient starting torque, the calculation of the speed and the torque includes some errors during the actual operation after the start because of a slight error as described above.

次にこの問題をもう少し詳しく説明し、その問題に対する補正について説明する。   Next, this problem will be described in more detail, and correction for the problem will be described.

まず、期間ａでの適当な比ｋ１が大きく異なる場合について説明する。   First, a case where the appropriate ratio k1 in the period a is significantly different will be described.

例えば３．７ＫＷ誘導電動機ではその１次抵抗と２次抵抗の比は各社多少の違いはあるがほぼ１：１になっており、期間ａで使う比ｋ１はｋ１＝１／（１＋１）＝０．５となる。この比は前述のように多少異なっていても期間ｂ終了後には正しい１次抵抗、２次抵抗が推定できる。しかし約５０ｍの接続ケーブルを付けるとその接続ケーブルと誘導電動機１０２内の１次抵抗の合成抵抗は誘導電動機１０２内の１次抵抗の２倍程度になる。その場合ｋ１＝０．５のまま推定を行うと期間ｂで抵抗の推定値収束に必要な時間が長くなるという問題があった。   For example, in a 3.7 KW induction motor, the ratio between the primary resistance and the secondary resistance is approximately 1: 1 although there is some difference between the companies, and the ratio k1 used in the period a is k1 = 1 / (1 + 1) = 0. .5. Even if the ratio is slightly different as described above, a correct primary resistance and secondary resistance can be estimated after the end of the period b. However, when a connection cable of about 50 m is attached, the combined resistance of the connection cable and the primary resistance in the induction motor 102 becomes about twice the primary resistance in the induction motor 102. In this case, if the estimation is performed with k1 = 0.5, there is a problem that the time required for the convergence of the estimated value of the resistance becomes longer in the period b.

そこで、起動前に１回だけ実施の形態１において述べたように、単相電圧などの試験電源を印加して求めたケーブル抵抗を含む１次抵抗と２次抵抗の比を求めて記憶しておき、その比を起動時の期間ａでの比ｋ１として使用することにより、期間ｂでの抵抗の推定値収束に必要な時間が長くなることを防ぎ、いつでも短時間で推定できるようになる。   Therefore, as described in the first embodiment only once before startup, the ratio between the primary resistance and the secondary resistance including the cable resistance obtained by applying a test power supply such as a single-phase voltage is calculated and stored. By using the ratio as the ratio k1 in the period a at the time of activation, it is possible to prevent the time required for the convergence of the estimated value of the resistance in the period b from becoming long, and to perform the estimation at any time in a short time.

次に上述のように抵抗推定値にわずかに誤差を含むため起動後の実運転時に速度やトルクの演算に多少誤差を含むという問題に対する対策について説明する。   Next, a description will be given of a countermeasure against the problem that the estimated value of the resistance includes a slight error as described above, and the calculation of the speed and the torque includes a slight error during the actual operation after the start-up.

図８、図１０に示す誘導電動機制御装置による起動時の抵抗の推定では、２次抵抗の推定結果に誤差を持つ。この原因は表皮効果によるものかと推測されるが、この誤差は誘導電動機１０２の温度が変化し抵抗値が変わっても真値のとの誤差には常に一定の比となることが実験により判った。   The estimation of the resistance at the time of starting by the induction motor control device shown in FIGS. 8 and 10 has an error in the estimation result of the secondary resistance. It is presumed that this cause is due to the skin effect, but it has been found through experiments that this error always has a fixed ratio to the true value error even if the temperature of the induction motor 102 changes and the resistance value changes. .

そこで誘導電動機１０２の運転前に１回だけ実施の形態１に示す単相電圧などの試験電源を印加して１次抵抗Ｒ１０と２次抵抗Ｒ２０を求め、巻線の温度が変化しない程度の時間内に、すなわちほぼ同時にこの実施の形態２による起動時の抵抗の推定も行って１次抵抗Ｒ１１、２次抵抗Ｒ２１を求める。１次抵抗Ｒ１０と１次抵抗Ｒ１１の比Ｃ１と、２次抵抗Ｒ２０と２次抵抗Ｒ２１の比Ｃ２とを記憶しておき、実際の起動時に実施の形態２に示す推定が終了した時点で推定した１次抵抗Ｒ１２にＣ１を、２次抵抗Ｒ２２にＣ２を乗じて補正する。このように補正した抵抗値を求め、実運転時のためにこの補正した抵抗値を使うようにすると、速度やトルクの演算の誤差も極めて小さくできる。
＜繰り返し運転の停止期間中の抵抗測定＞
上記実施の形態２で述べた誘導電動機制御装置は起動時に抵抗値を推定したが、例えば昇降機の運転パターンのように繰り返し運転中の停止期間が２秒程度以上あることが多い。そのような場合は起動時に抵抗を推定する必要はなく、停止期間に直流電圧や単相電圧を印加して抵抗推定を行うようにする。その推定は実施の形態１で述べた。実施の形態１で述べた誘導電動機制御装置では、３つの周波数の指令電圧を印加して行うため推定に３秒程度かかるが、実験によるとばらつきが１％以下と少なく良好な推定結果が得られる。実施の形態２で述べた誘導電動機制御装置では０．０５秒程度の短時間で終了するがばらつきが数％程度有る。この場合、繰り返し運転の停止期間中に何度も繰り返し、平均を取って誤差を減らすといった方法を使用しても良い。こうすることにより起動時だけで行う場合に対してさらに抵抗推定誤差を低減することができる。 Therefore, before the operation of the induction motor 102, the test power source such as the single-phase voltage shown in the first embodiment is applied only once to obtain the primary resistance R10 and the secondary resistance R20, and the time for which the temperature of the winding does not change is obtained. In other words, almost simultaneously, the resistance at the time of startup according to the second embodiment is also estimated to obtain the primary resistance R11 and the secondary resistance R21. The ratio C1 between the primary resistor R10 and the primary resistor R11 and the ratio C2 between the secondary resistor R20 and the secondary resistor R21 are stored, and are estimated when the estimation shown in the second embodiment is completed at the time of actual startup. The corrected primary resistance R12 is multiplied by C1 and the secondary resistance R22 is multiplied by C2. If the corrected resistance value is obtained in this way and the corrected resistance value is used for the actual operation, the error in the calculation of the speed and the torque can be extremely reduced.
<Resistance measurement during stoppage of repeated operation>
In the induction motor control device described in the second embodiment, the resistance value is estimated at the time of start-up. However, for example, a stop period during repetitive operation is often about 2 seconds or more like an operation pattern of an elevator. In such a case, it is not necessary to estimate the resistance at the time of startup, and a DC voltage or a single-phase voltage is applied during the stop period to perform the resistance estimation. The estimation has been described in the first embodiment. In the induction motor control device described in the first embodiment, it takes about 3 seconds to perform estimation by applying command voltages of three frequencies. However, according to an experiment, the variation is less than 1% and a good estimation result can be obtained. . In the induction motor control device described in the second embodiment, the operation is completed in a short time of about 0.05 seconds, but the variation is about several percent. In this case, a method may be used in which the operation is repeated many times during the stop period of the repetitive operation, the average is taken, and the error is reduced. By doing so, it is possible to further reduce the error in estimating the resistance as compared with the case of performing only at the start-up.

また、実施の形態１の誘導電動機制御装置による抵抗推定では約３秒程度時間がかかる。そこで停止期間がそれほど多く取れないような場合は、停止期間中に、ある周波数ｆ1 において２次抵抗・漏れインダクタンス演算測定手段１０６が２次抵抗Ｒ２３を測定し、あらかじめ実施の形態１に記載したように３つの異なる周波数を印加して表皮効果考慮演算手段１０７が求めた有理関数から周波数ｆ1 における２次抵抗Ｒ２１の値を求め、この２次抵抗Ｒ２１と２次抵抗Ｒ２３との比Ｃ３を記憶しておき、実運転時の滑り周波数ｆ2 における２次抵抗Ｒ２４を求め、この２次抵抗Ｒ２４に比Ｃ３を乗じて実運転時の２次抵抗を求めるようにする。このようにすることにより約１秒程度でばらつきが少なく高精度に所要の滑り周波数における２次抵抗を推定することができるようになる。 Also, it takes about three seconds for the resistance estimation by the induction motor control device of the first embodiment. Therefore when the stop period so as not take so much, during the stop period, the secondary resistance, leakage inductance computing measuring means 106 at a certain frequency f 1 measures the secondary resistance R23, described in Embodiment 1 in advance implementation as applied to three different frequencies determined the value of secondary resistance R21 at frequency f 1 from the rational function which skin effect considered computing means 107 is determined, the ratio C3 between this secondary resistance R21 and secondary resistance R23 stored advance, obtains the secondary resistance R24 at slip frequency f 2 during the actual operation, so that by multiplying the ratio C3 to the secondary resistance R24 obtaining secondary resistance in the actual operation. This makes it possible to estimate the secondary resistance at a required slip frequency with little variation in about one second and with high accuracy.

以上のように、この発明にかかる誘導電動機制御装置は、例えば、多相インバータにより試験電源を供給して多相誘導電動機の１次、２次抵抗及び漏れインダクタンスといった電気的定数を測定し、その測定結果を駆動制御のパラメータに用いて多相誘導電動機を速度センサ無しでベクトル制御する誘導電動機制御装置に適している。   As described above, the induction motor control device according to the present invention measures the electrical constants of the polyphase induction motor, such as the primary and secondary resistances and the leakage inductance, by supplying a test power supply by the polyphase inverter, for example. It is suitable for an induction motor control device that performs vector control of a polyphase induction motor without a speed sensor by using the measurement result as a parameter for drive control.

この発明の実施の形態１による誘導電動機制御装置を示すブロック図である。FIG. 1 is a block diagram showing an induction motor control device according to Embodiment 1 of the present invention. ２次導体形状が二重籠となっている誘導電動機の等価回路図である。It is an equivalent circuit diagram of the induction motor whose secondary conductor shape is a double basket. 図３は相互インダクタンスを考慮して２次抵抗・漏れインダクタンスを求めるこの発明の誘導電動機制御装置を示すブロック図である。FIG. 3 is a block diagram showing an induction motor control device according to the present invention for obtaining secondary resistance and leakage inductance in consideration of mutual inductance. この発明の誘導電動機制御装置が制御する誘導電動機の等価回路図であり、図４（ａ）は相互インダクタンスＭを開放と近似したＬ型等価回路図、図４（ｂ）は相互インダクタンスＭを考慮したＴ型等価回路図である。FIG. 4A is an equivalent circuit diagram of an induction motor controlled by the induction motor control device of the present invention. FIG. 4A is an L-type equivalent circuit diagram in which mutual inductance M is approximated to be open, and FIG. FIG. 7 is a T-type equivalent circuit diagram. 図５はこの発明の誘導電動機制御装置における単相電圧印加により２次抵抗・漏れインダクタンスを演算する試験電源指令手段と電力演算手段を示すブロック図である。FIG. 5 is a block diagram showing test power supply command means and power calculation means for calculating secondary resistance and leakage inductance by applying a single-phase voltage in the induction motor control device of the present invention. この発明の誘導電動機制御装置における指定電流で有効電力・無効電力を演算する試験電源指令手段と電力演算手段を示すブロック図である。FIG. 3 is a block diagram showing a test power supply command unit and a power calculation unit for calculating active power and reactive power with a designated current in the induction motor control device of the present invention. この発明の誘導電動機制御装置における試験電源指令手段及び電力演算手段を示すブロック図である。FIG. 3 is a block diagram showing test power supply command means and power calculation means in the induction motor control device of the present invention. この発明の実施の形態２による誘導電動機制御装置を示すブロック図である。FIG. 7 is a block diagram showing an induction motor control device according to Embodiment 2 of the present invention. この発明の実施の形態２による誘導電動機制御装置の動作を説明するための誘導電動機の等価回路図である。FIG. 9 is an equivalent circuit diagram of the induction motor for describing the operation of the induction motor control device according to Embodiment 2 of the present invention. この発明の実施の形態２による他の形態の誘導電動機制御装置を示すブロック図である。FIG. 13 is a block diagram showing another form of induction motor control device according to Embodiment 2 of the present invention. この発明の実施の形態２による誘導電動機制御装置の抵抗値推定時の結果の説明図である。FIG. 14 is an explanatory diagram of a result at the time of estimating a resistance value of the induction motor control device according to the second embodiment of the present invention. 表皮効果を考慮した従来の誘導電動機制御装置を示すブロック図である。It is a block diagram showing a conventional induction motor control device in consideration of the skin effect. 従来の表皮効果の考慮方法を示す説明図である。It is explanatory drawing which shows the conventional method of considering the skin effect. 誘導電動機の等価回路図である。It is an equivalent circuit diagram of an induction motor. 単相による定数測定機能を有する従来の誘導電動機制御装置のブロック図である。FIG. 4 is a block diagram of a conventional induction motor control device having a single-phase constant measurement function. 起動時に抵抗を測定する機能を有する従来の誘導電動機制御装置のブロック図である。FIG. 4 is a block diagram of a conventional induction motor control device having a function of measuring resistance at startup.

符号の説明Explanation of reference numerals

１０５ 有効電力・無効電力演算測定手段（電力演算手段）、１０６ ２次抵抗・漏れインダクタンス演算測定手段、１０７ 表皮効果考慮演算手段、１０８ ２次抵抗・漏れインダクタンス記憶手段。   105 active power / reactive power calculation / measurement means (power calculation means), 106 secondary resistance / leakage inductance calculation / measurement means, 107 skin effect consideration calculation means, 108 secondary resistance / leakage inductance storage means.

Claims (23)

多相インバータにより試験電源を供給して多相誘導電動機の電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置において、
周波数ｆの前記試験電源に対応する有効電力Ｐと無効電力Ｑとを演算する電力演算手段と、
前記周波数ｆに対応する相電圧ピーク値Ｖｐ及び１次角周波数ω、前記有効電力Ｐと前記無効電力Ｑ、予め測定された一次抵抗Ｒ１とからＬ型等価回路より求められる演算式から２次抵抗Ｒ２ｄと２次漏れインダクタンスＬ２ｄを求める第１の演算手段と、
Ｔ型等価回路より求められる２次抵抗Ｒ２と２次漏れインダクタンスＬの演算式のおのおのに、第１の演算手段が求めた前記２次抵抗Ｒ２ｄと前記２次漏れインダクタンスＬ２ｄと予め測定されている２次自己インダクタンスＬＬ２とを代入して２次抵抗Ｒ２と２次漏れインダクタンスＬを求める第２の演算手段と、
を備えたことを特徴とする誘導電動機制御装置。 Induction motor control in which a test power is supplied by a polyphase inverter to measure an electric constant of the polyphase induction motor, and the measurement result is used as a drive control parameter to drive and control the polyphase induction motor by the polyphase inverter. In the device,
Power calculating means for calculating active power P and reactive power Q corresponding to the test power supply of frequency f;
A secondary resistance is calculated from an arithmetic expression obtained from an L-type equivalent circuit based on the phase voltage peak value Vp and the primary angular frequency ω corresponding to the frequency f, the active power P and the reactive power Q, and a previously measured primary resistance R1. First calculating means for calculating R2d and a secondary leakage inductance L2d;
In each of the arithmetic expressions of the secondary resistance R2 and the secondary leakage inductance L obtained from the T-type equivalent circuit, the secondary resistance R2d and the secondary leakage inductance L2d obtained by the first arithmetic means are measured in advance. Second calculating means for substituting the secondary self-inductance LL2 to obtain a secondary resistance R2 and a secondary leakage inductance L;
An induction motor control device comprising: 前記Ｌ型等価回路より求められる演算式は、
Ｌ２ｄ＝Ｑ・Ｖｐ2／（４・ω・（Ｐ2＋Ｑ2））
Ｒ２ｄ＝Ｐ・Ｖｐ2／（２・（Ｐ2＋Ｑ2））−Ｒ１
前記Ｔ型等価回路より求められる演算式は、
Ｌ＝Ｌ２ｄ＋（２・Ｌ２ｄ−ＬＬ２）・Ｒ２ｄ2／（２・ω2・ＬＬ２2）
Ｒ２＝Ｒ２ｄ・ＬＬ２2／（ＬＬ２−Ｌ２ｄ）2
であることを特徴とする請求項１に記載の誘導電動機制御装置。 The arithmetic expression obtained from the L-type equivalent circuit is:
L2d = Q · Vp 2 / (4 · ω · (P 2 + Q 2 ))
R2d = P · Vp 2 / (2 · (P 2 + Q 2 )) − R1
The arithmetic expression obtained from the T-type equivalent circuit is:
L = L2d + (2 · L2d−LL2) · R2d 2 / (2 · ω 2 · LL2 2 )
R2 = R2d · LL2 2 / ( LL2-L2d) 2
The induction motor control device according to claim 1, wherein 前記電力演算手段は、多相インバータが多相誘導電動機に供給する単相交流の印加電圧より瞬時有効電力と瞬時無効電力を求め、前記瞬時有効電力と前記瞬時無効電力から高周波成分をフィルタリングすることを特徴とする請求項１に記載の誘導電動機制御装置。   The power calculation means obtains an instantaneous active power and an instantaneous reactive power from an applied voltage of a single-phase alternating current supplied to a polyphase induction motor by a polyphase inverter, and filters a high-frequency component from the instantaneous active power and the instantaneous reactive power. The induction motor control device according to claim 1, wherein: 前記電力演算手段は、前記瞬時有効電力と前記瞬時無効電力から印加電圧の２倍の周波数成分をノッチフィルタにより、前記２倍の周波数成分を超える高周波成分をローパスフィルタによりフィルタリングすることを特徴とする請求項３に記載の誘導電動機制御装置。   The power calculation means filters a frequency component twice the applied voltage from the instantaneous active power and the instantaneous reactive power by a notch filter, and filters a high-frequency component exceeding the doubled frequency component by a low-pass filter. The induction motor control device according to claim 3. 多相インバータが多相誘導電動機に対し指定した実効値電流を供給するように、前記多相インバータに指令電圧を出力する試験電源指令手段を備えたことを特徴とする請求項１に記載の誘導電動機制御装置。   2. The induction system according to claim 1, further comprising a test power supply command unit that outputs a command voltage to the polyphase inverter so that the polyphase inverter supplies a specified effective value current to the polyphase induction motor. Motor control device. 前記試験電源指令手段は、多相インバータに対する指令電圧を所定の上昇率で上昇し、この指令電圧の上昇に伴って多相インバータが多相誘導電動機へ供給する実効値電流の二乗の時間平均値が指定した実効値電流の１／３と等しいかあるいは大きくなったときに前記指令電圧の上昇を止めることを特徴とする請求項５に記載の誘導電動機制御装置。   The test power supply command means increases a command voltage for the polyphase inverter at a predetermined rate of increase, and with the increase of the command voltage, a time average value of the square of an effective value current supplied to the polyphase induction motor by the polyphase inverter. 6. The induction motor control device according to claim 5, wherein the increase of the command voltage is stopped when the value of the reference voltage is equal to or larger than 1/3 of the designated effective value current. 多相インバータに指令電圧を出力する試験電源指令手段を備え、
前記電力演算手段は、測定及び演算による遅れ時間に対応する補正位相により前記指令電圧の位相を補正して有効電力Ｐと無効電力Ｑとを演算することを特徴とする請求項１に記載の誘導電動機制御装置。 A test power supply command means for outputting a command voltage to the polyphase inverter is provided,
The guidance according to claim 1, wherein the power calculation unit calculates the active power P and the reactive power Q by correcting the phase of the command voltage with a correction phase corresponding to a delay time due to the measurement and the calculation. Motor control device. 多相インバータにより試験電源を供給して多相誘導電動機の電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置において、
前記多相インバータを制御して停止中の前記多相誘導電動機に所定の直流指令電圧を印加して直流電流を供給する電流制御手段と、
前記直流電圧と前記直流電流とを入力して、前記電流制御手段の電流供給直後の第１の期間に１次抵抗と２次抵抗との合成抵抗を演算し、前記第１の期間の次の第２の期間に１次抵抗を演算し、前記第１の期間に演算した前記合成抵抗から前記第２の期間に演算した前記１次抵抗を減じて２次抵抗を推定する抵抗推定手段と、
を備えたことを特徴とする誘導電動機制御装置。 Induction motor control in which a test power is supplied by a polyphase inverter to measure an electric constant of the polyphase induction motor, and the measurement result is used as a drive control parameter to drive and control the polyphase induction motor by the polyphase inverter. In the device,
Current control means for controlling the polyphase inverter and applying a predetermined DC command voltage to the stopped polyphase induction motor to supply a DC current,
The DC voltage and the DC current are input, and a combined resistance of a primary resistance and a secondary resistance is calculated in a first period immediately after the current supply by the current control unit, and the next resistance after the first period is calculated. Resistance estimating means for calculating a primary resistance in a second period and estimating a secondary resistance by subtracting the primary resistance calculated in the second period from the combined resistance calculated in the first period;
An induction motor control device comprising: 前記抵抗推定手段は、
多相誘導電動機に供給した直流電流と前記多相誘導電動機の既知の回路定数と帰還された１次巻線抵抗と２次巻線抵抗とから前記多相誘導電動機に印加した直流電圧値を推定する電圧推定器と、
前記直流電圧値と直流電圧指令値との偏差を積分演算する積分器と、
この積分器の出力を基に第１の期間に１次巻線と２次巻線との合成抵抗を演算し、この合成抵抗を１次巻線抵抗と２次巻線抵抗に分離して前記電圧推定器に帰還し、第２の期間に積分器の出力より１次巻線抵抗を演算し、前記第１の期間に演算した前記合成抵抗から前記第２の期間に測定した前記１次巻線抵抗を減じて２次巻線抵抗を推定し、演算した前記１次巻線抵抗と推定した前記２次巻線抵抗を前記電圧推定器に帰還する抵抗演算手段と、
を備え、
前記偏差が所定値以下となった場合に前記第２の期間に測定及び推定した前記１次巻線抵抗と前記２次巻線抵抗を使用して前記多相誘導電動機を制御することを特徴とする請求項８に記載の誘導電動機制御装置。 The resistance estimating means includes:
Estimating a DC voltage value applied to the polyphase induction motor from the DC current supplied to the polyphase induction motor, the known circuit constants of the polyphase induction motor, and the returned primary winding resistance and secondary winding resistance. Voltage estimator,
An integrator that integrates a deviation between the DC voltage value and the DC voltage command value,
Based on the output of the integrator, a combined resistance of the primary winding and the secondary winding is calculated in a first period, and the combined resistance is divided into a primary winding resistance and a secondary winding resistance. The voltage is fed back to the voltage estimator, the primary winding resistance is calculated from the output of the integrator in the second period, and the primary winding measured in the second period from the combined resistance calculated in the first period. Resistance calculating means for reducing a line resistance to estimate a secondary winding resistance, and feeding back the calculated primary winding resistance and the estimated secondary winding resistance to the voltage estimator;
With
Controlling the polyphase induction motor using the primary winding resistance and the secondary winding resistance measured and estimated during the second period when the deviation is equal to or less than a predetermined value. The induction motor control device according to claim 8, wherein: 前記抵抗推定手段は、
多相誘導電動機に指令した直流指令電圧と前記多相誘導電動機の既知の回路定数と帰還された１次巻線抵抗と２次巻線抵抗とから前記多相誘導電動機に供給した直流電流値を推定する電流推定器と、
前記直流電流値と前記多相誘導電動機に供給した直流電流の検出値との偏差を積分演算する演算器と、
この積分器の出力を基に第１の期間に１次巻線と２次巻線との合成抵抗を演算し、この合成抵抗を１次巻線抵抗と２次巻線抵抗に分離して前記電圧推定器に帰還し、第２の期間に積分器の出力より１次巻線抵抗を演算し、前記第１の期間に演算した前記合成抵抗から前記第２の期間に測定した前記１次巻線抵抗を減じて２次巻線抵抗を推定し、演算した前記１次巻線抵抗と推定した前記２次巻線抵抗を前記電流推定器に帰還する抵抗演算手段と、
を備え、
前記偏差が所定値以下となった場合に前記第２の期間に測定及び推定した前記１次巻線抵抗と前記２次巻線抵抗を使用して前記多相誘導電動機を制御することを特徴とする請求項８に記載の誘導電動機制御装置。 The resistance estimating means includes:
The DC current value supplied to the polyphase induction motor from the DC command voltage commanded to the polyphase induction motor, the known circuit constants of the polyphase induction motor, and the returned primary winding resistance and secondary winding resistance. A current estimator for estimating,
An arithmetic unit that integrates a deviation between the DC current value and a detection value of the DC current supplied to the polyphase induction motor,
Based on the output of the integrator, a combined resistance of the primary winding and the secondary winding is calculated in a first period, and the combined resistance is divided into a primary winding resistance and a secondary winding resistance. The voltage is fed back to the voltage estimator, the primary winding resistance is calculated from the output of the integrator in the second period, and the primary winding resistance measured in the second period is calculated from the combined resistance calculated in the first period. Resistance calculating means for reducing a line resistance to estimate a secondary winding resistance, and feeding back the calculated primary winding resistance and the estimated secondary winding resistance to the current estimator;
With
Controlling the polyphase induction motor using the primary winding resistance and the secondary winding resistance measured and estimated during the second period when the deviation is equal to or less than a predetermined value. The induction motor control device according to claim 8, wherein: 前記抵抗演算手段は、第１の期間に演算した合成抵抗を１次巻線抵抗と２次巻線抵抗とに所定の比で分離することを特徴とする請求項９に記載の誘導電動機制御装置。   10. The induction motor control device according to claim 9, wherein the resistance calculating unit separates the combined resistance calculated during the first period into a primary winding resistance and a secondary winding resistance at a predetermined ratio. . 予め交流電圧を多相誘導電動機に印加して求めた交流印加による１次抵抗Ｒ１及び２次抵抗Ｒ２と、
前記抵抗推定手段によりほぼ同時に推定した直流電圧印加による１次抵抗Ｒ１及び２次抵抗Ｒ２との比を記憶する記憶手段と、
を備え、
前記抵抗推定手段により推定した１次抵抗Ｒ１及び２次抵抗Ｒ２を前記記憶手段に記憶した比により補正することを特徴とする請求項８に記載の誘導電動機制御装置。 A primary resistance R1 and a secondary resistance R2 by applying an AC voltage obtained by applying an AC voltage to the polyphase induction motor in advance;
Storage means for storing a ratio between the primary resistance R1 and the secondary resistance R2 due to the application of the DC voltage estimated by the resistance estimating means almost simultaneously;
With
9. The induction motor control device according to claim 8, wherein the primary resistance R1 and the secondary resistance R2 estimated by the resistance estimating means are corrected based on the ratio stored in the storage means. 運転期間と停止期間を繰り返す多相誘導電動機の停止期間に１次抵抗と２次抵抗を推定して前記多相誘導電動機を制御することを特徴とする請求項８に記載の誘導電動機制御装置。   9. The induction motor control device according to claim 8, wherein the multi-phase induction motor is controlled by estimating a primary resistance and a secondary resistance during a stop period of the polyphase induction motor that repeats an operation period and a stop period. 多相インバータにより試験電源を供給して多相誘導電動機の電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置において、
少なくとも３つの異なる周波数ｆの前記試験電源による試験条件のそれぞれに対応する２次巻線抵抗Ｒ２を演算する第１の演算手段と、
周波数ｆの２次式を含む有理関数を有し、前記試験条件のそれぞれに対応する２次巻線抵抗を前記有理関数に代入してこの有理関数を求め、この求めた有理関数に所要の滑り周波数を代入して２次巻線抵抗Ｒ２を演算する第２の演算手段と、
を備えたことを特徴とする誘導電動機制御装置。 Induction motor control in which a test power is supplied by a polyphase inverter to measure an electric constant of the polyphase induction motor, and the measurement result is used as a drive control parameter to drive and control the polyphase induction motor by the polyphase inverter. In the device,
First calculating means for calculating a secondary winding resistance R2 corresponding to each of the test conditions of the test power supply of at least three different frequencies f;
It has a rational function including a quadratic expression of the frequency f, and substitutes secondary winding resistances corresponding to each of the test conditions into the rational function to obtain the rational function. Second calculating means for calculating the secondary winding resistance R2 by substituting the frequency;
An induction motor control device comprising: 前記第２の演算手段が有する有理関数は、
Ｒ２（ｆ）＝（ａ１＋ａ２・ｆ2）／（１＋ａ３・ｆ2）
であることを特徴とする請求項１４に記載の誘導電動機制御装置。 The rational function of the second calculation means is:
R2 (f) = (a1 + a2 · f 2 ) / (1 + a3 · f 2 )
The induction motor control device according to claim 14, wherein: 多相インバータにより試験電源を供給して多相誘導電動機の電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置において、
少なくとも３つの異なる周波数ｆの前記試験電源による試験条件のそれぞれに対応する漏れインダクタンスＬを演算する第１の演算手段と、
周波数ｆの２次式を含む有理関数を有し、前記試験条件のそれぞれに対応する漏れインダクタンスＬを前記有理関数に代入してこの有理関数を求め、この求めた有理関数に所要の滑り周波数を代入して漏れインダクタンスＬ２を演算する第２の演算手段と、
を備えたことを特徴とする誘導電動機制御装置。 Induction motor control in which a test power is supplied by a polyphase inverter to measure an electric constant of the polyphase induction motor, and the measurement result is used as a drive control parameter to drive and control the polyphase induction motor by the polyphase inverter. In the device,
First calculating means for calculating a leakage inductance L corresponding to each of the test conditions of the test power supply of at least three different frequencies f;
It has a rational function including a quadratic expression of the frequency f, and substitutes the leakage inductance L corresponding to each of the test conditions into the rational function to obtain the rational function. Second calculating means for calculating the leakage inductance L2 by substituting;
An induction motor control device comprising: 前記第２の演算手段が有する有理関数は、
Ｌ （ｆ）＝（ｂ１＋ｂ２・ｆ2）／（１＋ｂ３・ｆ2）
であることを特徴とする請求項１６に記載の誘導電動機制御装置。 The rational function of the second calculation means is:
L (f) = (b1 + b2 · f 2 ) / (1 + b3 · f 2 )
17. The induction motor control device according to claim 16, wherein 多相誘導電動機と、
多相インバータと、
前記多相インバータにより試験電源を供給して前記多相誘導電動機の少なくとも一つの電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置を有する電動機装置において、
前記誘導電動機制御装置が、
少なくとも３つの異なる周波数ｆの前記試験電源による試験条件のそれぞれに対応する２次巻線抵抗Ｒ２を演算する第１の演算手段と、
周波数ｆの２次式を含む有理関数を有し、前記試験条件のそれぞれに対応する２次巻線抵抗を前記有理関数に代入してこの有理関数を求め、この求めた有理関数に所要の滑り周波数を代入して２次巻線抵抗Ｒ２を演算する第２の演算手段と、
を備えたことを特徴とする電動機装置。 A polyphase induction motor,
A polyphase inverter,
The polyphase inverter supplies test power to measure at least one electrical constant of the polyphase induction motor, and drives the polyphase induction motor by the polyphase inverter using the measurement result as a drive control parameter. In an electric motor device having an induction motor control device for controlling,
The induction motor control device,
First calculating means for calculating a secondary winding resistance R2 corresponding to each of the test conditions of the test power supply of at least three different frequencies f;
It has a rational function including a quadratic expression of the frequency f, and substitutes secondary winding resistances corresponding to each of the test conditions into the rational function to obtain the rational function. Second calculating means for calculating the secondary winding resistance R2 by substituting the frequency;
An electric motor device comprising: 前記第２の演算手段が有する有理関数は、
Ｒ２（ｆ）＝（ａ１＋ａ２・ｆ2）／（１＋ａ３・ｆ2）
であることを特徴とする請求項１８に記載の電動機装置。 The rational function of the second calculation means is:
R2 (f) = (a1 + a2 · f 2 ) / (1 + a3 · f 2 )
The motor device according to claim 18, wherein: 多相誘導電動機と、
多相インバータと、
前記多相インバータにより試験電源を供給して前記多相誘導電動機の少なくとも一つの電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置を有する電動機装置において、
前記誘導電動機制御装置が、
少なくとも３つの異なる周波数ｆの前記試験電源による試験条件のそれぞれに対応する漏れインダクタンスＬを演算する第１の演算手段と、
周波数ｆの２次式を含む有理関数を有し、前記試験条件のそれぞれに対応する漏れインダクタンスＬを前記有理関数に代入してこの有理関数を求め、この求めた有理関数に所要の滑り周波数を代入して漏れインダクタンスＬ２を演算する第２の演算手段と、
を備えたことを特徴とする電動機装置。 A polyphase induction motor,
A polyphase inverter,
The polyphase inverter supplies test power to measure at least one electrical constant of the polyphase induction motor, and drives the polyphase induction motor by the polyphase inverter using the measurement result as a parameter for drive control. In an electric motor device having an induction motor control device for controlling,
The induction motor control device,
First calculating means for calculating a leakage inductance L corresponding to each of the test conditions of the test power supply of at least three different frequencies f;
It has a rational function including a quadratic expression of the frequency f, and substitutes the leakage inductance L corresponding to each of the test conditions into the rational function to obtain the rational function. Second calculating means for calculating the leakage inductance L2 by substituting;
An electric motor device comprising: 前記第２の演算手段が有する有理関数は、
Ｌ （ｆ）＝（ｂ１＋ｂ２・ｆ2）／（１＋ｂ３・ｆ2）
であることを特徴とする請求項２０に記載の電動機装置。 The rational function of the second calculation means is:
L (f) = (b1 + b2 · f 2 ) / (1 + b3 · f 2 )
21. The electric motor device according to claim 20, wherein 多相誘導電動機と、
多相インバータと、
前記多相インバータにより試験電源を供給して前記多相誘導電動機の少なくとも一つの電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置を有する電動機装置において、
前記誘導電動機制御装置が、
周波数ｆの前記試験電源に対応する有効電力Ｐと無効電力Ｑとを演算する電力演算手段と、
前記周波数ｆに対応する相電圧ピーク値Ｖｐ及び１次角周波数ω、前記有効電力Ｐと前記無効電力Ｑ、予め測定された一次抵抗Ｒ１とからＬ型等価回路より求められる演算式から２次抵抗Ｒ２ｄと２次漏れインダクタンスＬ２ｄを求める第１の演算手段と、
Ｔ型等価回路より求められる２次抵抗Ｒ２と２次漏れインダクタンスＬの演算式のおのおのに、第１の演算手段が求めた前記２次抵抗Ｒ２ｄと前記２次漏れインダクタンスＬ２ｄと予め測定されている２次自己インダクタンスＬＬ２とを代入して２次抵抗Ｒ２と２次漏れインダクタンスＬを求める第２の演算手段と、
を備えたことを特徴とする電動機装置。 A polyphase induction motor,
A polyphase inverter,
The polyphase inverter supplies test power to measure at least one electrical constant of the polyphase induction motor, and drives the polyphase induction motor by the polyphase inverter using the measurement result as a parameter for drive control. In an electric motor device having an induction motor control device for controlling,
The induction motor control device,
Power calculating means for calculating active power P and reactive power Q corresponding to the test power supply of frequency f;
A secondary resistance is calculated from an arithmetic expression obtained from an L-type equivalent circuit based on the phase voltage peak value Vp and the primary angular frequency ω corresponding to the frequency f, the active power P and the reactive power Q, and a previously measured primary resistance R1. First calculating means for calculating R2d and a secondary leakage inductance L2d;
In each of the arithmetic expressions of the secondary resistance R2 and the secondary leakage inductance L obtained from the T-type equivalent circuit, the secondary resistance R2d and the secondary leakage inductance L2d obtained by the first arithmetic means are measured in advance. Second calculating means for substituting the secondary self-inductance LL2 to obtain a secondary resistance R2 and a secondary leakage inductance L;
An electric motor device comprising: 多相誘導電動機と、
多相インバータと、
前記多相インバータにより試験電源を供給して前記多相誘導電動機の少なくとも一つの電気的定数を測定し、この測定結果を駆動制御のパラメータに用いて前記多相インバータにより前記多相誘導電動機を駆動制御する誘導電動機制御装置を有する電動機装置において、
前記誘導電動機制御装置が、
前記多相インバータを制御して停止中の前記多相誘導電動機に所定の直流指令電圧を印加して直流電流を供給する電流制御手段と、
前記直流電圧と前記直流電流とを入力して、前記電流制御手段の電流供給直後の第１の期間に１次抵抗と２次抵抗との合成抵抗を演算し、前記第１の期間の次の第２の期間に１次抵抗を演算し、前記第１の期間に演算した前記合成抵抗から前記第２の期間に演算した前記１次抵抗を減じて２次抵抗を推定する抵抗推定手段と、
を備えたことを特徴とする電動機装置。 A polyphase induction motor,
A polyphase inverter,
The polyphase inverter supplies test power to measure at least one electrical constant of the polyphase induction motor, and drives the polyphase induction motor by the polyphase inverter using the measurement result as a parameter for drive control. In an electric motor device having an induction motor control device for controlling,
The induction motor control device,
Current control means for controlling the polyphase inverter and applying a predetermined DC command voltage to the stopped polyphase induction motor to supply a DC current,
The DC voltage and the DC current are input, and a combined resistance of a primary resistance and a secondary resistance is calculated in a first period immediately after the current supply by the current control unit, and the next resistance after the first period is calculated. Resistance estimating means for calculating a primary resistance in a second period and estimating a secondary resistance by subtracting the primary resistance calculated in the second period from the combined resistance calculated in the first period;
An electric motor device comprising:
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JP2006101647A (en) * 2004-09-30 2006-04-13 Tokyo Electric Power Co Inc:The Cage-type induction machine simulating apparatus for analog simulator
JP2007159317A (en) * 2005-12-07 2007-06-21 Yaskawa Electric Corp Inverter unit for ac motor
WO2017056649A1 (en) * 2015-09-30 2017-04-06 株式会社日立産機システム Power conversion device and auto-tuning method therefor
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JP5777814B2 (en) 2012-08-09 2015-09-09 三菱電機株式会社 Electric vehicle control device
KR101789130B1 (en) * 2015-09-25 2017-11-16 전남대학교 산학협력단 Instantaneous reactive power and active power measurement techniques using all pass filter
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Publication number Priority date Publication date Assignee Title
JP2006101647A (en) * 2004-09-30 2006-04-13 Tokyo Electric Power Co Inc:The Cage-type induction machine simulating apparatus for analog simulator
JP4497359B2 (en) * 2004-09-30 2010-07-07 東京電力株式会社 Cage induction machine simulator for analog simulator
JP2007159317A (en) * 2005-12-07 2007-06-21 Yaskawa Electric Corp Inverter unit for ac motor
WO2017056649A1 (en) * 2015-09-30 2017-04-06 株式会社日立産機システム Power conversion device and auto-tuning method therefor
JP2017070118A (en) * 2015-09-30 2017-04-06 株式会社日立産機システム Power conversion device and auto-tuning method therefor
WO2023223374A1 (en) * 2022-05-16 2023-11-23 三菱電機株式会社 Power conversion device, refrigeration cycle device using the same, and leakage inductance calculation method

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