JP2003134843A - Control method for pwm power conversion device - Google Patents
Control method for pwm power conversion deviceInfo
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- JP2003134843A JP2003134843A JP2001328643A JP2001328643A JP2003134843A JP 2003134843 A JP2003134843 A JP 2003134843A JP 2001328643 A JP2001328643 A JP 2001328643A JP 2001328643 A JP2001328643 A JP 2001328643A JP 2003134843 A JP2003134843 A JP 2003134843A
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Abstract
Description
【０００１】[0001]
【発明の属する技術分野】本発明は、信号波と搬送波と
の比較によって制御されるＰＷＭ（パルス幅変調）電力
変換装置の制御方法に関する。ここで、ＰＷＭ電力変換
装置とは、信号波と搬送波との比較によって生成される
ＰＷＭパルスにより半導体スイッチング素子をオン・オ
フ制御し、直流−交流変換、交流−直流変換等を行う半
導体電力変換装置をいう。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control method for a PWM (pulse width modulation) power conversion device controlled by comparing a signal wave and a carrier wave. Here, the PWM power conversion device is a semiconductor power conversion device that performs on / off control of a semiconductor switching element by a PWM pulse generated by comparing a signal wave and a carrier wave, and performs DC-AC conversion, AC-DC conversion, and the like. Say.
【０００２】[0002]
【従来の技術】図８は従来のＰＷＭ電力変換装置の制御
ブロック図である。図において、ＰＷＭ電力変換装置３
は、変圧器２を介した電力系統１との間で直流電力Ｐ
ｄｃと交流電力Ｐａｃとを相互に変換するものであり、
例えばＰＷＭインバータ等が該当する。この電力変換装
置３は、図９（ａ）のように、ＧＴＯサイリスタ等の自
己消弧形半導体スイッチング素子（以下、単に半導体素
子という）１１ａ〜１１ｆと、これらに逆並列接続され
たダイオード１２ａ〜１２ｆと、直流入力端子間に接続
されたコンデンサ１３とから構成されている。図９
（ｂ）は電力変換装置３の１相分を取り出したもので、
Ｑ１，Ｑ２はそれぞれ半導体素子であり、図９（ａ）の
１１ａ〜１１ｆに相当する。2. Description of the Related Art FIG. 8 is a control block diagram of a conventional PWM power converter. In the figure, the PWM power converter 3
Is the DC power P between the power system 1 and the transformer 2.
dc and AC power P ac are mutually converted,
For example, a PWM inverter or the like is applicable. As shown in FIG. 9A, the power conversion device 3 includes self-extinguishing semiconductor switching elements (hereinafter, simply referred to as semiconductor elements) 11a to 11f such as GTO thyristors, and diodes 12a to 11f connected in antiparallel. 12 f and a capacitor 13 connected between the DC input terminals. Figure 9
(B) shows one phase of the power conversion device 3 taken out,
Each of Q1 and Q2 is a semiconductor element and corresponds to 11a to 11f in FIG.
【０００３】電力変換装置３が有効電力を出力する場
合、図１０（ａ）のように、前記変圧器２をリアクトル
Ｘとして表すことができる。いま、図１０（ｂ）に示す
ごとく、電力変換装置３の出力電圧ベクトルＶｉｎｖの
位相を系統電圧ベクトルＶｓに対して変えることで、両
者の差電圧ベクトルＶｘがリアクトルＸに印加され、有
効電流ｉが流れることになる。When the power conversion device 3 outputs active power, the transformer 2 can be represented as a reactor X as shown in FIG. Now, as shown in FIG. 10B, by changing the phase of the output voltage vector V inv of the power conversion device 3 with respect to the system voltage vector V s , the difference voltage vector V x between them is applied to the reactor X, The effective current i will flow.
【０００４】電力変換装置３を制御するに当たっては、
図８において出力電圧指令ｉ＊と出力電流ｉＳとの偏差
を加算器６により求めて電流調節器７に入力し、その出
力を電圧検出器５により検出した系統電圧ＶＳに加算器
８にて加算することで、電力変換装置３に対する交流電
圧指令Ｖ＊（信号波）を演算する。そして、図１１に示
すように、交流電圧指令Ｖ＊と搬送波発生器１０により
演算された三角波などの搬送波ＣＡとを比較器９により
比較し、半導体素子Ｑ１，Ｑ２（１１ａ〜１１ｆ）に対
するＰＷＭパルスを生成してスイッチングタイミングを
決定する。このような一連の制御方法が、一般的なＰＷ
Ｍ（Pulse Width Modulation）制御方法である。In controlling the power converter 3,
Input to the current regulator 7 to seek the adder 6 the deviation between the output voltage command i * and the output current i S 8, the adder 8 the output to the system voltage V S detected by the voltage detector 5 The AC voltage command V * (signal wave) for the power converter 3 is calculated by adding and adding. Then, as shown in FIG. 11, the AC voltage command V * and the carrier wave CA such as a triangular wave calculated by the carrier wave generator 10 are compared by the comparator 9, and the PWM pulse for the semiconductor elements Q1 and Q2 (11a to 11f) is compared. Is generated to determine the switching timing. Such a series of control methods is a common PW
This is an M (Pulse Width Modulation) control method.
【０００５】ＰＷＭ制御は、図９（ｂ）の電力変換装置
３の１相分だけを考えると、正弦波変調の場合には、図
１１に示したように正弦波の電圧指令Ｖ＊と搬送波ＣＡ
との比較を行い、Ｖ＊＞ＣＡの場合は上アームの半導体
素子Ｑ１をオンさせ、下アームの半導体素子Ｑ２をオフ
させる。また、Ｖ＊＜ＣＡの場合は半導体素子Ｑ１をオ
フさせ、半導体素子Ｑ２をオンさせる制御である。ここ
で、電力変換装置３の出力電圧Ｖｉｎｖは、コンデンサ
１３の直流電圧をＥｄとすると、数式１のようになる。Considering only one phase of the power converter 3 of FIG. 9 (b), the PWM control, in the case of sine wave modulation, shows a sine wave voltage command V * and a carrier wave as shown in FIG. CA
When V * > CA, the semiconductor element Q1 of the upper arm is turned on and the semiconductor element Q2 of the lower arm is turned off. When V * <CA, the semiconductor element Q1 is turned off and the semiconductor element Q2 is turned on. Here, the output voltage V inv of the power conversion device 3 is represented by Formula 1 when the DC voltage of the capacitor 13 is E d .
【０００６】[0006]
【数１】 Ｖｉｎｖ＝｛（√３／２）×（１／√２）｝Ｅｄ×Ｖ＊ （Ｖｉｎｖ：線間電圧実効値）[Equation 1] Vinv= {(√3 / 2) × (1 / √2)} Ed× V* (Vinv: Effective value of line voltage)
【０００７】数式１から、コンデンサ１３の電圧Ｅｄが
一定の場合、電力変換装置３の出力電圧Ｖｉｎｖを大き
くするためには、電圧指令Ｖ＊を大きくすれば良いこと
が分かる。しかし、電圧指令Ｖ＊は、以下のような理由
によって所定の値よりも大きくすることができない。From Equation 1, it can be seen that when the voltage E d of the capacitor 13 is constant, the output voltage V inv of the power conversion device 3 can be increased by increasing the voltage command V * . However, the voltage command V * cannot be made larger than the predetermined value for the following reasons.
【０００８】すなわち、図１２（ａ）は半導体素子Ｑ
１，Ｑ２を過電圧から保護するためにスナバ回路Ｓを付
加した回路構成図であり、Ｃｓはスナバコンデンサであ
る。例えば下アームの半導体素子Ｑ２に過電圧が印加さ
れて充電されたスナバコンデンサＣＳの電荷は、半導体
素子Ｑ２のオン時に図１２（ａ）の破線矢印の経路で放
電する。図１２（ｂ）に示すように、このスナバコンデ
ンサＣｓの放電時間ｔｃや半導体素子Ｑ１，Ｑ２の短絡
を回避するためのオン遅延時間ｔｄを確保する必要があ
るので、搬送波ＣＡの周波数をｆｓとした場合、電圧指
令Ｖ＊は数式２に示す最大値Ｖ＊ ｍａｘを超えることが
できない。That is, FIG. 12A shows a semiconductor device Q.
1, Q2 and a circuit diagram obtained by adding a snubber circuit S for protection against overvoltages, C s is a snubber capacitor. For example, the electric charge of the snubber capacitor C S charged by applying the overvoltage to the semiconductor element Q2 of the lower arm is discharged through the path indicated by the broken line arrow in FIG. 12A when the semiconductor element Q2 is turned on. As shown in FIG. 12B, since it is necessary to secure the discharge time t c of the snubber capacitor C s and the on-delay time t d for avoiding the short circuit of the semiconductor elements Q1 and Q2, the frequency of the carrier wave CA. If the set to f s, * the voltage command V can not exceed the maximum value V * max shown in formula 2.
【０００９】[0009]
【数２】Ｖ＊ ｍａｘ＝１−２×ｆｓ×（ｔｃ＋ｔｄ）[Number 2] V * max = 1-2 × f s × (t c + t d)
【００１０】例えば、ｔｃ＝１００［μｓ］、ｔｄ＝
［μｓ］、ｆｓ＝４５０［Ｈｚ］とするとすると、数式
２からＶ＊ ｍａｘ＝０．８６５となる。従って、数式１
によれば、電力変換装置３の出力電圧の最大値（√２×
Ｖｉｎｖ）は直流電圧Ｅｄの０．７４５倍（つまり、√
２×Ｖｉｎｖ＝（√３／２）×Ｅｄ×０．８６５＝０．
７４５×Ｅｄ）が限度となるため、直流電圧Ｅｄを十分
に利用していないことにほかならない。このように、直
流電圧利用率が低い場合には電力変換装置３の出力電圧
が低くなり、装置容量が小さくなるという問題を生じ
る。For example, t c = 100 [μs], t d =
Assuming that [μs] and f s = 450 [Hz], V * max = 0.865 is obtained from Equation 2. Therefore, Equation 1
According to the above, the maximum value of the output voltage of the power conversion device 3 (√2 ×
V inv is 0.745 times the DC voltage E d (that is, √)
2 × V inv = (√3 / 2) × E d × 0.865 = 0.
Since the limit is 745 × E d ), it means that the DC voltage E d is not fully utilized. As described above, when the DC voltage utilization rate is low, the output voltage of the power conversion device 3 becomes low, which causes a problem that the device capacity becomes small.
【００１１】[0011]
【発明が解決しようとする課題】従来、直流電圧利用率
を上げる場合は、図１３に示すように、系統電圧Ｖｓを
基準とした３倍調波発生器２３を電圧検出器５の二次側
に接続し、この３倍調波発生器２３により演算した３倍
調波Ｖ３ｆ（系統電圧Ｖｓと同位相）を加算器１４にお
いて電圧指令Ｖ＊に加算することにより、搬送波ＣＡと
比較するべき電圧指令Ｖ＊＊を得ている。なお、図１３
において、図８と同一の構成要素には同一の参照符号を
付してある。Conventionally, in order to increase the DC voltage utilization rate, as shown in FIG. 13, the triple harmonic generator 23 based on the system voltage V s is used as a secondary detector of the voltage detector 5. by connecting to the side, adds in an adder 14 three times harmonic V 3f of calculation (system voltage V s in phase) by the triple harmonic generator 23 to the voltage command V *, compared with the carrier CA The voltage command V ** to be obtained is obtained. Note that FIG.
8, the same components as those in FIG. 8 are designated by the same reference numerals.
【００１２】上記３倍調波Ｖ３ｆの振幅をＶ＊ ｍａｘに
対して約１３％にすると、前出の例では、電圧指令Ｖ＊
と３倍調波Ｖ３ｆとを加算したＶ＊＊（加算器１４の出
力信号）の最大値は約０．７５５となる（０．８６５・
sinθ＋０．８６５×０．１３・sin３θの最大値が約
０．７５５となる）。つまり、Ｖ＊ ｍａｘの値を０．７
５５にしても、（√３／２）×Ｅｄ×０．８６５という
出力電圧Ｖｉｎｖの最大値（√２×Ｖｉｎｖ）が得られ
ることになり、この場合の電圧利用率（直流電圧をどの
くらい交流電圧に変換できるかを示す比率）は（√２×
Ｖｉｎｖ）／Ｅｄ＝（√３／２）×０．８６５となる。
これに対し、３倍調波Ｖ３ｆを電圧指令Ｖ＊に加算しな
い時には、√２×Ｖｉ ｎｖ ＝（√３／２）×Ｅｄ×
０．７５５となり、電圧利用率は（√２×Ｖｉｎ ｖ）／
Ｅｄ＝（√３／２）×０．７５５となる。従って、３倍
調波Ｖ３ｆを電圧指令Ｖ＊に加算すれば、電圧利用率は
約１４．６％（０．８６５／０．７５５≒１．１４６）
上がることになる。このことから、３倍調波Ｖ３ｆを加
算すれば、Ｖ＊ ｍａｘを超えない範囲で直流電圧の利用
率を上げることが可能である。When the amplitude of the third harmonic V 3f is set to about 13% with respect to V * max , the voltage command V * is obtained in the above example .
And the third harmonic V 3f are added together, the maximum value of V ** (output signal of the adder 14) is about 0.755 (0.865 ·
The maximum value of sinθ + 0.865 × 0.13 · sin3θ is about 0.755). That is, the value of V * max is 0.7
Even with 55, the maximum value (√2 × V inv ) of the output voltage V inv of (√3 / 2) × E d × 0.865 is obtained, and the voltage utilization rate (DC voltage is The ratio indicating how much can be converted into AC voltage is (√2 ×
V inv ) / E d = (√3 / 2) × 0.865.
In contrast, when not adding the 3-fold harmonic V 3f voltage command V * is, √2 × V i nv = ( √3 / 2) × E d ×
0.755, and the voltage utilization rate is (√2 × V in v ) /
E d = (√3 / 2) × 0.755. Therefore, if the triple harmonic V 3f is added to the voltage command V * , the voltage utilization rate is about 14.6% (0.865 / 0.755≈1.146).
Will go up. From this, by adding the third harmonic V 3f , it is possible to increase the utilization rate of the DC voltage within a range not exceeding V * max .
【００１３】図１４はもとの電圧指令Ｖ＊と３倍調波Ｖ
３ｆ及びこれらの加算結果である電圧指令Ｖ＊＊の波形
を示しており、もとの電圧指令Ｖ＊がＶ＊ ｍａｘを超え
る場合でも、３倍調波Ｖ３ｆを加算すれば電圧指令Ｖ
＊＊がＶ＊ ｍａｘ以下になることがわかる。FIG. 14 shows the original voltage command V * and the triple harmonic V
3f and the waveform of the voltage command V ** which is the addition result thereof are shown, and even if the original voltage command V * exceeds V * max , if the triple harmonic V 3f is added, the voltage command V *
It can be seen that ** becomes V * max or less.
【００１４】しかし、この方法によると、電流調節器７
の出力に応じて系統電圧ＶＳに対する電圧指令Ｖ＊の位
相が変化し、この位相が大きくずれた時には、図１５に
示すように３倍調波Ｖ３ｆと電圧指令Ｖ＊との位相が大
きくずれてしまい、その結果、電圧指令Ｖ＊＊が電圧指
令Ｖ＊よりも大きくなってＶ＊ ｍａｘを超えることにな
り、直流電圧利用率の向上が達成できない場合があっ
た。そこで本発明は、もとの交流電圧指令に基づいて演
算した同位相の３倍調波をもとの交流電圧指令に加算す
ることにより、交流電圧指令の位相に関わらず、しかも
制限値を超えない状態で直流電圧利用率を高めるように
したＰＷＭ変換装置の制御方法を提供しようとするもの
である。However, according to this method, the current regulator 7
When the phase of the voltage command V * with respect to the system voltage V S changes in accordance with the output of V.sub.S and the phase greatly deviates, the phase between the third harmonic V.sub.3f and the voltage command V * becomes large as shown in FIG. As a result, the voltage command V ** becomes larger than the voltage command V * and exceeds V * max, and in some cases, the improvement of the DC voltage utilization rate cannot be achieved. In view of this, the present invention adds a triple harmonic wave of the same phase calculated based on the original AC voltage command to the original AC voltage command, so that the limit value is exceeded regardless of the phase of the AC voltage command. An object of the present invention is to provide a control method for a PWM conversion device that enhances the DC voltage utilization rate in a non-existent state.
【００１５】[0015]
【課題を解決するための手段】上記課題を解決するた
め、請求項１項記載の発明は、信号波と搬送波とを比較
してＰＷＭパルスを生成し、このＰＷＭパルスを用いて
半導体スイッチング素子をオン・オフ制御することによ
り電力変換を行うＰＷＭ電力変換装置において、前記電
力変換装置に与える３相の交流電圧指令を３相２相変換
して互いに同一周波数の第１、第２の交流波形信号と振
幅信号とに分離演算し、前記交流波形信号の周波数を３
倍にしてもとの交流電圧指令と同位相の３倍調波信号を
生成し、前記振幅信号に所定値を乗算した信号と前記３
倍調波信号とを乗算して得た信号をもとの交流電圧指令
に加算して最終的な交流電圧指令を生成するものであ
る。In order to solve the above problems, the invention according to claim 1 compares a signal wave with a carrier wave to generate a PWM pulse, and uses this PWM pulse to implement a semiconductor switching element. In a PWM power conversion device that performs power conversion by performing on / off control, a three-phase two-phase conversion of a three-phase AC voltage command given to the power conversion device is performed, and first and second AC waveform signals having the same frequency as each other. And the amplitude signal are separately calculated, and the frequency of the AC waveform signal is set to 3
Even if the signal is multiplied, a triple harmonic signal having the same phase as the original AC voltage command is generated, and the amplitude signal is multiplied by a predetermined value, and
A signal obtained by multiplying with the harmonic wave signal is added to the original AC voltage command to generate a final AC voltage command.
【００１６】請求項２記載の発明は、信号波と搬送波と
を比較してＰＷＭパルスを生成し、このＰＷＭパルスを
用いて半導体スイッチング素子をオン・オフ制御するこ
とにより電力変換を行うＰＷＭ電力変換装置において、
前記電力変換装置に与える３相の交流電圧指令を３相２
相変換して互いに同一周波数の第１、第２の交流波形信
号と振幅信号とに分離演算し、前記交流波形信号の周波
数を３倍にしてもとの交流電圧指令と同位相の３倍調波
信号を生成し、この３倍調波信号に所定値を乗算して得
た信号をもとの交流電圧指令に加算して最終的な交流電
圧指令を生成するものである。According to a second aspect of the present invention, a PWM power conversion is performed by comparing a signal wave with a carrier wave to generate a PWM pulse, and using this PWM pulse to perform on / off control of a semiconductor switching element to perform power conversion. In the device,
The three-phase two-phase AC voltage command given to the power converter is used.
Phase conversion is performed to separate and calculate the first and second AC waveform signals and the amplitude signal having the same frequency, and the frequency of the AC waveform signal is tripled. A wave signal is generated and a signal obtained by multiplying the triple harmonic signal by a predetermined value is added to the original AC voltage command to generate a final AC voltage command.
【００１７】請求項３記載の発明は、信号波と搬送波と
を比較してＰＷＭパルスを生成し、このＰＷＭパルスを
用いて半導体スイッチング素子をオン・オフ制御するこ
とにより電力変換するＰＷＭ電力変換装置において、前
記電力変換装置に与える３相の交流電圧指令を３相２相
変換して交流電圧指令の周波数を基準とした第１、第２
の回転座標軸成分と振幅信号とに分離演算し、前記回転
座標軸成分を用いて交流電圧指令の３倍周波数を基準と
して交流電圧指令と同位相の３倍調波信号を生成すると
ともに、前記振幅信号に所定値を乗算した信号と前記３
倍調波信号とを乗算して得た信号をもとの交流電圧指令
に加算して最終的な交流電圧指令を生成するものであ
る。According to a third aspect of the present invention, a PWM power converter for comparing a signal wave and a carrier wave to generate a PWM pulse, and using this PWM pulse to perform on / off control of a semiconductor switching element for power conversion. In the first and second, the three-phase AC voltage command given to the power converter is converted into three-phase to two-phase and the frequency of the AC voltage command is used as a reference.
And the amplitude signal is generated separately, and a triple harmonic signal having the same phase as the AC voltage command is generated using the rotary coordinate axis component as a reference and the triple frequency of the AC voltage command is used. Signal multiplied by a predetermined value and the above 3
A signal obtained by multiplying with the harmonic wave signal is added to the original AC voltage command to generate a final AC voltage command.
【００１８】請求項４記載の発明は、信号波と搬送波と
を比較してＰＷＭパルスを生成し、このＰＷＭパルスを
用いて半導体スイッチング素子をオン・オフ制御するこ
とにより電力変換を行うＰＷＭ電力変換装置において、
前記電力変換装置に与える３相の交流電圧指令を３相２
相変換して交流電圧指令の周波数を基準とした第１、第
２の回転座標軸成分と振幅信号とに分離演算し、前記回
転座標軸成分を用いて交流電圧指令の３倍周波数を基準
として交流電圧指令と同位相の３倍調波信号を生成する
とともに、この３倍調波信号に所定値を乗算して得た信
号をもとの交流電圧指令に加算して最終的な交流電圧指
令を生成するものである。According to a fourth aspect of the present invention, a PWM power conversion is performed by comparing a signal wave with a carrier wave to generate a PWM pulse, and using this PWM pulse to perform on / off control of a semiconductor switching element to perform power conversion. In the device,
The three-phase two-phase AC voltage command given to the power converter is used.
Phase conversion is performed to perform a separate operation on the first and second rotational coordinate axis components based on the frequency of the AC voltage command and the amplitude signal, and the AC voltage is determined using the rotational coordinate axis component and the triple frequency of the AC voltage command as a reference. Generates a triple harmonic signal with the same phase as the command, and adds the signal obtained by multiplying this triple harmonic signal by a predetermined value to the original AC voltage command to generate the final AC voltage command. To do.
【００１９】[0019]
【発明の実施の形態】以下、図に沿って本発明の実施形
態を説明する。図１は第１実施形態を示す制御ブロック
図であり、ＰＷＭ変換装置３に与える３相各相の交流電
圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊は、３相２相変換回路１
５により２相量Ｖα，Ｖβに変換される。ここで、交流
電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊は、図８における加算
器８の出力信号に相当する信号であり、数式３によって
表されるものとする。なお、数式３においてＶｍは振
幅、ωは角周波数、φは位相角である。BEST MODE FOR CARRYING OUT THE INVENTION Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a control block diagram showing the first embodiment. AC voltage commands V a * , V b * , and V c * of each of the three phases given to the PWM conversion device 3 are the three-phase / two-phase conversion circuit 1.
5 is converted into two-phase quantities Vα and Vβ. Here, the AC voltage commands V a * , V b * , and V c * are signals corresponding to the output signals of the adder 8 in FIG. 8, and are represented by Formula 3. In Expression 3, V m is the amplitude, ω is the angular frequency, and φ is the phase angle.
【００２０】[0020]
【数３】 [Equation 3]
【００２１】３相２相変換回路１５では、数式４の演算
により、数式５に示す２相量Ｖα，Ｖβを得る。The three-phase / two-phase conversion circuit 15 obtains the two-phase quantities Vα and Vβ shown in Expression 5 by the calculation of Expression 4.
【００２２】[0022]
【数４】 [Equation 4]
【００２３】[0023]
【数５】 [Equation 5]
【００２４】２相量Ｖα，Ｖβはベクトル変換回路１６
に入力され、数式６の演算により、数式７に示す余弦波
Ｖ１Ｃ及び正弦波Ｖ１Ｓと、電圧指令の振幅Ｖｍを出力
する。The two-phase quantities Vα and Vβ are calculated by the vector conversion circuit 16
And outputs the cosine wave V 1C and the sine wave V 1S shown in Expression 7 and the amplitude V m of the voltage command by the calculation of Expression 6.
【００２５】[0025]
【数６】 [Equation 6]
【００２６】[0026]
【数７】 [Equation 7]
【００２７】ベクトル変換回路１６から出力された余弦
波Ｖ１Ｃは３倍調波発生器１７ａに入力される。この３
倍調波発生器１７ａでは、３倍角の公式に基づく数式８
の演算、具体的には、数式９の演算を行ってＶ１Ｃの周
波数を３倍にした３倍調波Ｖ ３ｆ’を出力する。Cosine output from the vector conversion circuit 16
Wave V1CIs input to the triple harmonic generator 17a. This 3
In the harmonic generator 17a, the formula 8 based on the triple angle formula is used.
Is calculated, specifically, the calculation of Equation 9 is performed to obtain V1CLap
Triple harmonic V with tripled wave number 3f'Is output.
【００２８】[0028]
【数８】Ｖ３ｆ’＝４×Ｖ１Ｃ ３−３×Ｖ１Ｃ ## EQU8 ## V 3f '= 4 × V 1C 3 -3 × V 1C
【００２９】[0029]
【数９】 Ｖ３ｆ’＝４×ｃｏｓ３（ωｔ−φ）−３×ｃｏｓ（ωｔ−φ） ＝ｃｏｓ｛３（ωｔ−φ）｝V 3f '= 4 × cos 3 (ωt−φ) −3 × cos (ωt−φ) = cos {3 (ωt−φ)}
【００３０】一方、振幅Ｖｍに所定値のゲイン２２（例
えば−０．１３）を乗算し、その結果と数式９の３倍調
波Ｖ３ｆ’とを乗算器１８により乗算して、数式１０に
示す３倍調波Ｖ３ｆを得る。この３倍調波Ｖ３ｆは、加
算器１４ａ，１４ｂ，１４ｃにおいてもとの電圧指令Ｖ
ａ ＊，Ｖｂ ＊，Ｖｃ ＊と加算されることにより、最終的
な３相各相の電圧指令Ｖａ ＊＊，Ｖｂ ＊＊，Ｖｃ ＊＊が
出力され、これらの電圧指令が信号波として搬送波と比
較されてＰＷＭパルスが生成されることになる。On the other hand, the amplitude V m is multiplied by a gain 22 of a predetermined value (for example, -0.13), and the result is multiplied by the triple harmonic V 3f ′ of Equation 9 by the multiplier 18, and Equation 10 The third harmonic V 3f shown in is obtained. This triple harmonic V 3f is the original voltage command V in the adders 14a, 14b, 14c.
a *, V b *, by being added to the V c *, the final three phases of voltage command V a **, V b **, V c ** is output, these voltage command The PWM pulse will be generated by being compared with the carrier wave as a signal wave.
【００３１】[0031]
【数１０】 Ｖ３ｆ＝−０．１３・Ｖｍ・ｃｏｓ｛３（ωｔ−φ）｝V 3f = −0.13 · V m · cos {3 (ωt−φ)}
【００３２】図２は、この実施形態における電圧指令Ｖ
ａ ＊，Ｖｂ ＊，Ｖｃ ＊、２相量Ｖα，Ｖβ、余弦波Ｖ
１Ｃ、正弦波Ｖ１Ｓ、振幅Ｖｍ、３倍調波Ｖ３ｆ，Ｖ
３ｆ’、最終的な電圧指令Ｖａ ＊＊，Ｖｂ ＊＊，Ｖｃ
＊＊の波形を示している。この図から明らかなように、
本実施形態によれば、３倍調波Ｖ３ｆはもとの電圧指令
Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊に対して常に同位相となる。す
なわち、３倍調波Ｖ３ｆのゼロクロス点は電圧指令Ｖａ
＊，Ｖｂ ＊，Ｖｃ ＊のゼロクロス点と一致している。こ
のように、交流電圧指令を振幅Ｖｍと交流波形（余弦波
Ｖ１Ｃ及び正弦波Ｖ１Ｓ）とに分離演算するとともに、
振幅Ｖｍに所定値を乗算した値と交流波形（余弦波Ｖ
１Ｃ）の周波数を３倍にした３倍調波Ｖ３ｆ’とを乗算
して交流電圧指令と同位相の３倍調波Ｖ３ｆとを得るこ
とができ、この３倍調波Ｖ３ｆを各相の電圧指令
Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊にそれぞれ加算すれば、もとの
電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊以下であって制限値Ｖ
＊ ｍａｘを超えない電圧指令Ｖ ａ ＊＊，Ｖｂ ＊＊，Ｖｃ
＊＊を得ることができる。このため、電圧指令の位相に
関わらず、制限値Ｖ＊ ｍａｘを超えない範囲で電力変換
装置３の出力電圧を大きくし、直流電圧利用率を上げる
ことができる。FIG. 2 shows the voltage command V in this embodiment.
a *, Vb *, Vc *Two-phase quantities Vα, Vβ, cosine wave V
1C, Sine wave V1S, Amplitude Vm3rd harmonic V3f, V
3f', The final voltage command Va **, Vb **, Vc
**Shows the waveform of. As you can see from this figure,
According to this embodiment, the triple harmonic V3fOriginal voltage command
Va *, Vb *, Vc *It is always in phase with. You
That is, triple harmonic V3fThe zero-cross point of is the voltage command Va
*, Vb *, Vc *It coincides with the zero-cross point of. This
, The AC voltage command is amplitude VmAnd AC waveform (cosine wave
V1CAnd sine wave V1S) And a separate operation,
Amplitude VmValue multiplied by a predetermined value and the AC waveform (cosine wave V
1C) The triple harmonic V which tripled the frequency3fMultiply by
And the third harmonic V of the same phase as the AC voltage command3fAnd get
And this triple harmonic V3fVoltage command for each phase
Va *, Vb *, Vc *If you add each to
Voltage command Va *, Vb *, Vc *Less than or equal to the limit value V
* maxVoltage command V not exceeding a **, Vb **, Vc
**Can be obtained. Therefore, the phase of the voltage command
Regardless of the limit value V* maxPower conversion within the range
Increase the output voltage of device 3 and increase the DC voltage utilization rate
be able to.
【００３３】図３は、この実施形態において、もとの電
圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊の振幅に応じて３倍調波
Ｖ３ｆの振幅が変化する様子を示している。このこと
は、前述の数式１０や、図１において、振幅Ｖｍにゲイ
ン２２を乗じた値を乗算器１８により３倍調波Ｖ３ｆに
乗じていることからも理解される。なお、この図３は、
次の第２実施形態の説明においても比較参照する。FIG. 3 shows how the amplitude of the third harmonic V 3f changes according to the amplitude of the original voltage commands V a * , V b * , V c * in this embodiment. This can be understood from the above-mentioned mathematical expression 10 and in FIG. 1 that the multiplier 18 multiplies the triple harmonic V 3f by a value obtained by multiplying the amplitude V m by the gain 22. In addition, this FIG.
Comparative reference will be made also in the following description of the second embodiment.
【００３４】次に、本発明の第２実施形態を説明する。
図４は本実施形態の制御ブロック図であり、図１と同一
の構成要素には同一の参照符号を付してある。この実施
形態では、３倍調波発生器１７ａから出力される３倍調
波Ｖ３ｆ’に所定値のゲイン１９（例えば−０．１３）
が乗算されて電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊の大きさ
に依存しない振幅一定の３倍調波Ｖ３ｆが出力される。
この３倍調波Ｖ３ｆは前述した数式１０の右辺における
Ｖｍを除去した（１とした）値である。Next, a second embodiment of the present invention will be described.
FIG. 4 is a control block diagram of this embodiment, and the same components as those in FIG. 1 are designated by the same reference numerals. In this embodiment, the third harmonic V 3f ′ output from the third harmonic generator 17 a has a predetermined gain 19 (eg, −0.13).
Is multiplied by to output a third harmonic V 3f having a constant amplitude that does not depend on the magnitude of the voltage commands V a * , V b * , and V c * .
The third harmonic V 3f is a value obtained by removing V m on the right side of Expression 10 described above (assumed to be 1).
【００３５】上記の３倍調波Ｖ３ｆは、図１と同様に加
算器１４ａ，１４ｂ，１４ｃにおいて各相の電圧指令Ｖ
ａ ＊，Ｖｂ ＊，Ｖｃ ＊にそれぞれ加算されることで、も
との電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊以下であって制限
値Ｖ＊ ｍａｘを超えない電圧指令Ｖａ ＊＊，Ｖｂ ＊＊，
Ｖｃ ＊＊が得られる。本実施形態においても、電圧指令
の位相に関わらず制限値Ｖ＊ ｍａｘを超えない範囲で電
力変換装置３の出力電圧を大きくし、直流電圧利用率を
上げることができる。The above triple harmonic V 3f is applied to the voltage command V of each phase in the adders 14a, 14b and 14c as in FIG.
a *, V b *, V c * in by being respectively added, based on the voltage command V a *, V b *, V c * less was it does not exceed the limit value V * max voltage command V a ** , V b ** ,
V c ** is obtained. Also in the present embodiment, it is possible to increase the output voltage of the power conversion device 3 and increase the DC voltage utilization rate within a range that does not exceed the limit value V * max regardless of the phase of the voltage command.
【００３６】第１実施形態では、前述の図３のように３
倍調波Ｖ３ｆの振幅は電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊
の振幅に応じて変化することになるが、この第２実施形
態では、図５に示すように電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖ
ｃ ＊の振幅に関わらず３倍調波Ｖ３ｆの振幅が一定とな
る。この実施形態によれば、第１実施形態に比べて乗算
器１８が不要になり、回路構成が簡略化されるという利
点がある。In the first embodiment, as shown in FIG.
Harmonic V3fIs the voltage command Va *, Vb *, Vc *
It changes according to the amplitude of the
In the state, as shown in FIG.a *, Vb *, V
c *3rd harmonic V regardless of amplitude3fThe amplitude of
It According to this embodiment, multiplication is performed as compared with the first embodiment.
Since the device 18 is not necessary, the circuit configuration can be simplified.
There is a point.
【００３７】次に、本発明の第３実施形態を図６に従っ
て説明する。なお、図１，図４と同一の構成要素には同
一の参照符号を付してある。図６において、各相の電圧
指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊は３相２相変換回路１５に
入力され、２相量Ｖα，Ｖβに変換される。ここで、電
圧指令Ｖａ ＊,Ｖｂ ＊,Ｖｃ ＊は前記数式３によって表さ
れるものとし、３相２相変換回路１５から出力される２
相量Ｖα，Ｖβは数式５のとおりである。Next, a third embodiment of the present invention will be described with reference to FIG.
Explain. The same components as those in FIGS.
One reference numeral is attached. 6, the voltage of each phase
Command Va *, Vb *, Vc *In the three-phase to two-phase conversion circuit 15
It is input and converted into two-phase quantities Vα and Vβ. Where the electric
Pressure command Va *, Vb *, Vc *Is expressed by Equation 3 above.
2 output from the three-phase / two-phase conversion circuit 15
The phase amounts Vα and Vβ are as shown in Equation 5.
【００３８】３０は電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊と
同じ周波数を持つ正弦波ｓｉｎωｔ及び余弦波ｃｏｓω
ｔを出力する正弦波・余弦波発生器であり、前記正弦波
ｓｉｎωｔ及び余弦波ｃｏｓωｔは回転座標軸変換回路
２０に入力される。回転座標軸変換回路２０では、数式
１１により、２相量Ｖα，Ｖβから正弦波ｓｉｎωｔ及
び余弦波ｃｏｓωｔ（電圧指令Ｖａ ＊，Ｖｂ ＊，
Ｖｃ ＊）の周波数を基準とした回転座標軸成分Ｖｄ，Ｖ
ｑを演算する。すなわち、数式１１を演算し、加法定理
を適用して数式１２を得る。こうして求められた回転座
標軸成分Ｖ ｄ，Ｖｑは、フィルタ回路２４ｄ，２４ｑに
よって電圧指令に含まれる高調波成分や逆相成分が除去
される。30 is a voltage command Va *, Vb *, Vc *When
Sine wave sinωt and cosine wave cosω having the same frequency
A sine / cosine wave generator for outputting t,
sinωt and cosine wave cosωt are rotational coordinate axis conversion circuits.
It is input to 20. In the rotary coordinate axis conversion circuit 20,
11, the sine wave sinωt and the two-phase quantities Vα and Vβ
And cosine wave cosωt (voltage command Va *, Vb *，
Vc *) Rotation coordinate axis component V based on the frequencyd, V
qIs calculated. That is, Equation 11 is calculated, and the addition theorem
To obtain Equation 12. Rotating seat thus obtained
Standard axis component V d, VqIn the filter circuits 24d and 24q
Therefore, the harmonic components and negative phase components included in the voltage command are removed.
To be done.
【００３９】[0039]
【数１１】 [Equation 11]
【００４０】[0040]
【数１２】 [Equation 12]
【００４１】回転座標軸成分Ｖｄ，Ｖｑはベクトル変換
回路１６に入力され、数式１３の演算により余弦波Ｖ
１ＣＡ及び正弦波Ｖ１ＳＡが求められる。また、ベクト
ル変換回路１６は電圧指令の振幅Ｖｍを出力する。The rotational coordinate axis components V d and V q are input to the vector conversion circuit 16 and the cosine wave V
1CA and sine wave V 1SA are determined. Further, the vector conversion circuit 16 outputs the amplitude V m of the voltage command.
【００４２】[0042]
【数１３】 [Equation 13]
【００４３】３倍調波発生器１７ｂでは、３倍角の公式
に従って数式１４の演算を行い、Ｖ １ＣＡ，Ｖ１ＳＡの
周波数を３倍にしたＶ２ＣＡ，Ｖ２ＳＡを求めて静止座
標変換回路２１に出力する。In the triple harmonic generator 17b, the triple angle formula is used.
According to the formula 14, 1 CA, V1 SAof
V tripled in frequency2CA, V2SAIn search of stationary
It outputs to the standard conversion circuit 21.
【００４４】[0044]
【数１４】 [Equation 14]
【００４５】３１は電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊の
３倍周波数を持つ正弦波ｓｉｎ３ωｔ及び余弦波ｃｏｓ
３ωｔを出力する正弦波・余弦波発生器であり、これら
の正弦波ｓｉｎ３ωｔ及び余弦波ｃｏｓ３ωｔは前記Ｖ
２ＣＡ，Ｖ２ＳＡとともに静止座標変換回路２１に入力
される。静止座標変換回路２１では、数式１５、詳しく
は数式１６の演算を行い、加法定理を適用して各相電圧
指令の３倍周波数を持つ３倍調波Ｖ３ｆｄ及びＶ３ｆｑ
を出力する。Reference numeral 31 denotes a sine wave sin3ωt and a cosine wave cos having a frequency three times that of the voltage commands V a * , V b * , and V c *.
It is a sine wave / cosine wave generator that outputs 3ωt, and these sine wave sin3ωt and cosine wave cos3ωt are
It is input to the static coordinate conversion circuit 21 together with 2CA and V 2SA . In the static coordinate conversion circuit 21, the calculation of Formula 15, more specifically Formula 16, is performed, and the addition theorem is applied to apply triple harmonics V 3fd and V 3fq having a triple frequency of each phase voltage command.
Is output.
【００４６】[0046]
【数１５】 [Equation 15]
【００４７】[0047]
【数１６】 [Equation 16]
【００４８】一方、振幅Ｖｍに所定値のゲイン２２（例
えば−０．１３）を乗算し、その結果と前記３倍調波Ｖ
３ｆｄとを乗算器１８により乗算して、数式１０と同様
に数式１７に示す３倍調波Ｖ３ｆを得る。この３倍調波
Ｖ３ｆは、加算器１４ａ，１４ｂ，１４ｃにおいてもと
の電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊と加算されることに
より、最終的な３相各相の電圧指令Ｖａ ＊＊，
Ｖｂ ＊＊，Ｖｃ ＊＊となり、搬送波との比較に用いられ
る。On the other hand, the amplitude V m is multiplied by a gain 22 of a predetermined value (for example, -0.13), and the result is multiplied by the third harmonic V.
3fd is multiplied by the multiplier 18 to obtain the third harmonic V 3f shown in Expression 17 as in Expression 10. This triple harmonic V 3f is added to the original voltage commands V a * , V b * , V c * in the adders 14 a, 14 b, 14 c, so that the final voltage command for each of the three phases is obtained. V a ** ,
It becomes V b ** , V c ** and is used for comparison with the carrier wave.
【００４９】[0049]
【数１７】 Ｖ３ｆ＝−０．１３・Ｖｍ・ｃｏｓ｛３（ωｔ−φ）｝V 3f = −0.13 · V m · cos {3 (ωt−φ)}
【００５０】本実施形態によれば、３倍調波Ｖ３ｆはも
との電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊に対して常に同位
相となる。このように、交流の電圧指令を振幅Ｖｍと電
圧指令の周波数を基準とした回転座標軸成分Ｖ１ＣＡ，
Ｖ１ＳＡとに分離演算し、振幅Ｖｍに所定値を乗算した
値と電圧指令の３倍周波数を基準として演算した３倍調
波Ｖ３ｆｄとを乗算して電圧指令と同位相の３倍調波Ｖ
３ｆを得ることができ、この３倍調波Ｖ３ｆを各相の電
圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊にそれぞれ加算すれば、
もとの電圧指令Ｖａ ＊，Ｖｂ ＊，Ｖｃ ＊以下であって制
限値Ｖ＊ ｍａｘを超えない電圧指令Ｖａ ＊＊，
Ｖｂ ＊＊，Ｖｃ ＊＊を得ることができる。従って本実施
形態でも、電圧指令の位相に関わらず、制限値Ｖ＊
ｍａｘを超えない範囲で電力変換装置３の出力電圧を大
きくし、直流電圧利用率を上げることができる。According to this embodiment, the triple harmonic V3fPeach
Voltage command V witha *, Vb *, Vc *Always equal to
Be in phase. In this way, the AC voltage command is applied to the amplitude VmAnd electric
Rotation coordinate axis component V based on the frequency of the pressure command1 CA，
V1 SASeparately calculated into and amplitude VmIs multiplied by a predetermined value
Triple adjustment calculated based on the value and triple the frequency of the voltage command
Wave V3fdMultiply by and the third harmonic V with the same phase as the voltage command
3fThis triple harmonic V3fThe power of each phase
Pressure command Va *, Vb *, Vc *If you add
Original voltage command Va *, Vb *, Vc *Less than
Limit value V* maxVoltage command V not exceedinga **，
Vb **, Vc **Can be obtained. Therefore, this implementation
Also in the form, the limit value V regardless of the phase of the voltage command*
maxThe output voltage of the power conversion device 3 within a range not exceeding
The DC voltage utilization rate can be increased.
【００５１】最後に、本発明の第４実施形態を図７に従
って説明する。この実施形態において、３相２相変換回
路１５から静止座標変換回路２１までの動作は第３実施
形態と同様である。この実施形態では、静止座標変換回
路２１から出力される３倍調波Ｖ３ｆｄに所定値のゲイ
ン１９（例えば−０．１３）が乗算される。これによ
り、第３実施形態と異なって電圧指令Ｖａ ＊，Ｖｂ ＊，
Ｖｃ ＊の大きさに依存しない振幅一定の３倍調波Ｖ３ｆ
が出力される。この３倍調波Ｖ３ｆは数式１７の右辺に
おけるＶｍを除去した（１とした）値である。Finally, a fourth embodiment of the present invention will be described with reference to FIG. In this embodiment, the operation from the three-phase / two-phase conversion circuit 15 to the stationary coordinate conversion circuit 21 is the same as in the third embodiment. In this embodiment, the third harmonic V 3fd output from the static coordinate conversion circuit 21 is multiplied by a gain 19 (for example, −0.13) of a predetermined value. As a result, unlike the third embodiment, the voltage commands V a * , V b * ,
Third harmonic V 3f with constant amplitude that does not depend on the magnitude of V c *
Is output. The third harmonic V 3f is a value obtained by removing V m on the right side of Expression 17 (assumed to be 1).
【００５２】上記３倍調波Ｖ３ｆは、加算器１４ａ，１
４ｂ，１４ｃにおいて各相の電圧指令Ｖａ ＊，Ｖｂ ＊，
Ｖｃ ＊にそれぞれ加算され、もとの電圧指令Ｖａ ＊，Ｖ
ｂ ＊，Ｖｃ ＊以下であって制限値Ｖ＊ ｍａｘを超えない
電圧指令Ｖａ ＊＊，Ｖｂ ＊＊，Ｖｃ ＊＊が得られる。こ
れにより、本実施形態でも電圧指令の位相に関わらず制
限値Ｖ＊ ｍａｘを超えない範囲で電力変換装置３の出力
電圧を大きくし、直流電圧利用率を上げることができ
る。この実施形態によれば、第２実施形態と同様に図６
における乗算器１８が不要になるので、回路構成が簡略
化される。The third harmonic V 3f is added to the adders 14a, 1
4b and 14c, voltage commands V a * , V b * of each phase,
The original voltage commands V a * , V a are added to V c * , respectively.
b *, the voltage command does not exceed the limit value V * max be at V c * below V a **, V b **, V c ** is obtained. As a result, even in this embodiment, the output voltage of the power conversion device 3 can be increased and the DC voltage utilization rate can be increased within a range that does not exceed the limit value V * max regardless of the phase of the voltage command. According to this embodiment, as in the second embodiment, as shown in FIG.
Since the multiplier 18 in is unnecessary, the circuit configuration is simplified.
【００５３】なお、上記各実施形態では各相の電圧指令
の分離演算により得た余弦波を利用して３倍調波を作成
しているが、正弦波を用いて３倍調波を作成することも
可能である。In each of the above embodiments, the triple harmonic is created by using the cosine wave obtained by the separation calculation of the voltage command of each phase, but the triple harmonic is created by using the sine wave. It is also possible.
【００５４】[0054]
【発明の効果】以上のように本発明によれば、もとの電
圧指令に基づいてこれと同位相の３倍調波を生成し、こ
の３倍調波をもとの電圧指令に加算して最終的な電圧指
令を得るようにしたので、系統電圧に対して電圧指令の
位相が変化した場合にも、常に適切な３倍調波を得るこ
とができ、電圧指令の制限値を超えない範囲で電力変換
値の直流電圧利用率を上げることができる。As described above, according to the present invention, a triple harmonic having the same phase as that of the original voltage command is generated, and the triple harmonic is added to the original voltage command. As a result, the final voltage command is obtained, so that even if the phase of the voltage command changes with respect to the system voltage, an appropriate triple harmonic can always be obtained, and the limit value of the voltage command is not exceeded. The DC voltage utilization rate of the power conversion value can be increased in the range.
【図１】本発明の第１実施形態を示す制御ブロック図で
ある。FIG. 1 is a control block diagram showing a first embodiment of the present invention.
【図２】第１実施形態の動作を示す電圧波形図である。FIG. 2 is a voltage waveform diagram showing the operation of the first embodiment.
【図３】第１実施形態において、電圧指令の大きさが異
なる場合の電圧波形図である。FIG. 3 is a voltage waveform diagram when the magnitudes of voltage commands are different in the first embodiment.
【図４】本発明の第２実施形態を示す制御ブロック図で
ある。FIG. 4 is a control block diagram showing a second embodiment of the present invention.
【図５】第２実施形態において、電圧指令の大きさが異
なる場合の電圧波形図である。FIG. 5 is a voltage waveform diagram when the magnitudes of voltage commands are different in the second embodiment.
【図６】本発明の第３実施形態を示す制御ブロック図で
ある。FIG. 6 is a control block diagram showing a third embodiment of the present invention.
【図７】本発明の第４実施形態を示す制御ブロック図で
ある。FIG. 7 is a control block diagram showing a fourth embodiment of the present invention.
【図８】従来技術を示す制御ブロック図である。FIG. 8 is a control block diagram showing a conventional technique.
【図９】電力変換装置の主回路構成図である。FIG. 9 is a main circuit configuration diagram of a power conversion device.
【図１０】電力変換装置の動作説明図である。FIG. 10 is an operation explanatory diagram of the power conversion device.
【図１１】電力変換装置のＰＷＭ制御原理を説明する波
形図である。FIG. 11 is a waveform diagram illustrating the PWM control principle of the power conversion device.
【図１２】電力変換装置におけるスナバ回路と放電時間
及びオン遅延時間の説明図である。FIG. 12 is an explanatory diagram of a snubber circuit and a discharge time and an ON delay time in the power conversion device.
【図１３】直流電圧利用率を上げるための従来技術を示
す制御ブロック図である。FIG. 13 is a control block diagram showing a conventional technique for increasing a DC voltage utilization rate.
【図１４】電圧指令等の波形図である。FIG. 14 is a waveform diagram of a voltage command or the like.
【図１５】発明の解決課題を説明するための電圧指令等
の波形図である。FIG. 15 is a waveform diagram of a voltage command or the like for explaining the problem to be solved by the invention.
１・・・電力系統
２・・・変圧器
３・・・ＰＷＭ電力変換装置
４・・・電流検出器
５・・・電圧検出器
６・・・減算器
７・・・電流調節器
８・・・加算器
９・・・比較器
１０・・・搬送波発生器
１１ａ〜１１ｆ・・・自己消弧形半導体スイッチング素
子
１２ａ〜１２ｆ・・・ダイオード
１３・・・コンデンサ
１４ａ，１４ｂ，１４ｃ・・・加算器
１５・・・３相２相変換回路
１６・・・ベクトル変換回路
１７ａ，１７ｂ，２３・・・３倍調波発生器
１８・・・乗算器
１９，２２・・・ゲイン
２０・・・回転座標変換回路
２１・・・静止座標変換回路
２４ｄ，２４ｑ・・・フィルタ回路
３０，３１・・・正弦波・余弦波発生器1 ... Power system 2 ... Transformer 3 ... PWM power converter 4 ... Current detector 5 ... Voltage detector 6 ... Subtractor 7 ... Current regulator 8 ... Adder 9 ... Comparator 10 ... Carrier wave generators 11a-11f ... Self-extinguishing type semiconductor switching elements 12a-12f ... Diode 13 ... Capacitors 14a, 14b, 14c ... Addition 15-three-phase / two-phase conversion circuit 16-vector conversion circuits 17a, 17b, 23-third harmonic generator 18-multipliers 19, 22-gain 20-rotation Coordinate conversion circuit 21 ... Stationary coordinate conversion circuits 24d, 24q ... Filter circuits 30, 31 ... Sine / cosine wave generator
───────────────────────────────────────────────────── フロントページの続き Ｆターム(参考） 5H006 AA07 CA05 CB01 CB08 CC02 DA02 DC02 DC04 DC05 5H007 AA00 CA05 CB02 CB05 CC03 DA05 DB02 DC02 DC05 EA15 ─────────────────────────────────────────────────── ─── Continued front page F-term (reference) 5H006 AA07 CA05 CB01 CB08 CC02 DA02 DC02 DC04 DC05 5H007 AA00 CA05 CB02 CB05 CC03 DA05 DB02 DC02 DC05 EA15
Claims (4)
スを生成し、このＰＷＭパルスを用いて半導体スイッチ
ング素子をオン・オフ制御することにより電力変換を行
うＰＷＭ電力変換装置において、 前記電力変換装置に与える３相の交流電圧指令を３相２
相変換して互いに同一周波数の第１、第２の交流波形信
号と振幅信号とに分離演算し、前記交流波形信号の周波
数を３倍にしてもとの交流電圧指令と同位相の３倍調波
信号を生成し、前記振幅信号に所定値を乗算した信号と
前記３倍調波信号とを乗算して得た信号をもとの交流電
圧指令に加算して最終的な交流電圧指令を生成すること
を特徴としたＰＷＭ電力変換装置の制御方法。1. A PWM power conversion apparatus for performing power conversion by comparing a signal wave and a carrier wave to generate a PWM pulse, and using the PWM pulse to control ON / OFF of a semiconductor switching element. Three-phase AC voltage command given to the device
Phase conversion is performed to separate and calculate the first and second AC waveform signals and the amplitude signal having the same frequency, and the frequency of the AC waveform signal is tripled. Wave signal is generated, and a signal obtained by multiplying the signal obtained by multiplying the amplitude signal by a predetermined value and the triple harmonic signal is added to the original AC voltage command to generate a final AC voltage command. A method for controlling a PWM power conversion device, comprising:
スを生成し、このＰＷＭパルスを用いて半導体スイッチ
ング素子をオン・オフ制御することにより電力変換を行
うＰＷＭ電力変換装置において、 前記電力変換装置に与える３相の交流電圧指令を３相２
相変換して互いに同一周波数の第１、第２の交流波形信
号と振幅信号とに分離演算し、前記交流波形信号の周波
数を３倍にしてもとの交流電圧指令と同位相の３倍調波
信号を生成し、この３倍調波信号に所定値を乗算して得
た信号をもとの交流電圧指令に加算して最終的な交流電
圧指令を生成することを特徴としたＰＷＭ電力変換装置
の制御方法。2. A PWM power conversion device for performing power conversion by comparing a signal wave and a carrier wave to generate a PWM pulse, and using the PWM pulse to control ON / OFF of a semiconductor switching element. Three-phase AC voltage command given to the device
Phase conversion is performed to separate and calculate the first and second AC waveform signals and the amplitude signal having the same frequency, and the frequency of the AC waveform signal is tripled. PWM power conversion characterized by generating a wave signal and adding a signal obtained by multiplying the triple harmonic signal by a predetermined value to the original AC voltage command to generate a final AC voltage command. Device control method.
スを生成し、このＰＷＭパルスを用いて半導体スイッチ
ング素子をオン・オフ制御することにより電力変換を行
うＰＷＭ電力変換装置において、 前記電力変換装置に与える３相の交流電圧指令を３相２
相変換して交流電圧指令の周波数を基準とした第１、第
２の回転座標軸成分と振幅信号とに分離演算し、前記回
転座標軸成分を用いて交流電圧指令の３倍周波数を基準
として交流電圧指令と同位相の３倍調波信号を生成する
とともに、前記振幅信号に所定値を乗算した信号と前記
３倍調波信号とを乗算して得た信号をもとの交流電圧指
令に加算して最終的な交流電圧指令を生成することを特
徴としたＰＷＭ電力変換装置の制御方法。3. A PWM power conversion device for performing power conversion by comparing a signal wave and a carrier wave to generate a PWM pulse, and using the PWM pulse to control ON / OFF of a semiconductor switching element. Three-phase AC voltage command given to the device
Phase conversion is performed to perform a separate operation on the first and second rotational coordinate axis components based on the frequency of the AC voltage command and the amplitude signal, and the AC voltage is determined using the rotational coordinate axis component and the triple frequency of the AC voltage command as a reference. A triple harmonic signal having the same phase as the command is generated, and a signal obtained by multiplying the amplitude signal by a predetermined value and the triple harmonic signal is added to the original AC voltage command. And a final AC voltage command is generated to control the PWM power converter.
スを生成し、このＰＷＭパルスを用いて半導体スイッチ
ング素子をオン・オフ制御することにより電力変換を行
うＰＷＭ電力変換装置において、 前記電力変換装置に与える３相の交流電圧指令を３相２
相変換して交流電圧指令の周波数を基準とした第１、第
２の回転座標軸成分と振幅信号とに分離演算し、前記回
転座標軸成分を用いて交流電圧指令の３倍周波数を基準
として交流電圧指令と同位相の３倍調波信号を生成する
とともに、この３倍調波信号に所定値を乗算して得た信
号をもとの交流電圧指令に加算して最終的な交流電圧指
令を生成することを特徴としたＰＷＭ電力変換装置の制
御方法。4. A PWM power conversion device for performing power conversion by comparing a signal wave and a carrier wave to generate a PWM pulse, and using the PWM pulse to control ON / OFF of a semiconductor switching element. Three-phase AC voltage command given to the device
Phase conversion is performed to perform a separate operation on the first and second rotational coordinate axis components based on the frequency of the AC voltage command and the amplitude signal, and the AC voltage is determined using the rotational coordinate axis component and the triple frequency of the AC voltage command as a reference. Generates a triple harmonic signal with the same phase as the command, and adds the signal obtained by multiplying this triple harmonic signal by a predetermined value to the original AC voltage command to generate the final AC voltage command. A method for controlling a PWM power conversion device, comprising:
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2001328643A JP3812406B2 (en) | 2001-10-26 | 2001-10-26 | Control method for PWM power converter |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2001328643A JP3812406B2 (en) | 2001-10-26 | 2001-10-26 | Control method for PWM power converter |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JP2003134843A true JP2003134843A (en) | 2003-05-09 |
| JP3812406B2 JP3812406B2 (en) | 2006-08-23 |
Family
ID=19144675
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| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP2001328643A Expired - Lifetime JP3812406B2 (en) | 2001-10-26 | 2001-10-26 | Control method for PWM power converter |
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Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2007244066A (en) * | 2006-03-07 | 2007-09-20 | Ebara Densan Ltd | Voltage-type current control inverter |
| JP2010213512A (en) * | 2009-03-12 | 2010-09-24 | Hitachi Car Eng Co Ltd | Torque controller for permanent-magnet synchronous motor |
| WO2012127910A1 (en) * | 2011-03-23 | 2012-09-27 | 北陸電力株式会社 | Distributed power supply system |
| JP2012231606A (en) * | 2011-04-26 | 2012-11-22 | Fuji Electric Co Ltd | System interconnection power conversion device |
| WO2015036835A3 (en) * | 2013-09-11 | 2015-06-11 | Toyota Jidosha Kabushiki Kaisha | Electric motor control apparatus and electric motor control method |
| CN112385112A (en) * | 2019-05-29 | 2021-02-19 | 东芝三菱电机产业***株式会社 | Power conversion system |
| EP3985851A1 (en) * | 2020-10-16 | 2022-04-20 | General Electric Technology GmbH | Improvements in or relating to voltage source converters |
-
2001
- 2001-10-26 JP JP2001328643A patent/JP3812406B2/en not_active Expired - Lifetime
Cited By (14)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2007244066A (en) * | 2006-03-07 | 2007-09-20 | Ebara Densan Ltd | Voltage-type current control inverter |
| US8305019B2 (en) | 2009-03-12 | 2012-11-06 | Hitachi Car Engineering Co., Ltd. | Torque controller for permanent magnet synchronous motor |
| JP2010213512A (en) * | 2009-03-12 | 2010-09-24 | Hitachi Car Eng Co Ltd | Torque controller for permanent-magnet synchronous motor |
| CN103430441A (en) * | 2011-03-23 | 2013-12-04 | 北陆电力株式会社 | Distributed power supply system |
| JP2012205325A (en) * | 2011-03-23 | 2012-10-22 | Hokuriku Electric Power Co Inc:The | Distributed power supply system |
| WO2012127910A1 (en) * | 2011-03-23 | 2012-09-27 | 北陸電力株式会社 | Distributed power supply system |
| US20130328398A1 (en) * | 2011-03-23 | 2013-12-12 | Fuji Electric Co., Ltd. | Distributed power supply system |
| US9509145B2 (en) | 2011-03-23 | 2016-11-29 | Hokuriku Electric Power Company | Distributed power supply system |
| JP2012231606A (en) * | 2011-04-26 | 2012-11-22 | Fuji Electric Co Ltd | System interconnection power conversion device |
| WO2015036835A3 (en) * | 2013-09-11 | 2015-06-11 | Toyota Jidosha Kabushiki Kaisha | Electric motor control apparatus and electric motor control method |
| CN105556830A (en) * | 2013-09-11 | 2016-05-04 | 丰田自动车株式会社 | Electric motor control apparatus and electric motor control method |
| CN105556830B (en) * | 2013-09-11 | 2018-04-10 | 丰田自动车株式会社 | Electric motor control device and method of motor control |
| CN112385112A (en) * | 2019-05-29 | 2021-02-19 | 东芝三菱电机产业***株式会社 | Power conversion system |
| EP3985851A1 (en) * | 2020-10-16 | 2022-04-20 | General Electric Technology GmbH | Improvements in or relating to voltage source converters |
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| JP3812406B2 (en) | 2006-08-23 |
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