GB2625070A

GB2625070A - A method of controlling a brushless permanent magnet motor

Info

Publication number: GB2625070A
Application number: GB2218141.6A
Authority: GB
Inventors: Junior Ifedi Chukwuma; Pan Zhiyang
Original assignee: Dyson Technology Ltd
Current assignee: Dyson Technology Ltd
Priority date: 2022-12-02
Filing date: 2022-12-02
Publication date: 2024-06-12
Also published as: GB202218141D0; WO2024116064A1

Abstract

A rotor of a brushless permanent magnet motor experiences, during rotation of the rotor, a cogging torque that oscillates between a negative cogging torque that opposes the rotation of the rotor and a positive cogging toque that promotes the rotation of the rotor. A method of controlling the motor includes, determining a first period in which the rotor experiences the negative cogging torque; and controlling, during the first period, current to flow in a phase winding of the motor to drive rotation of the rotor against the negative cogging torque. The method further includes, determining a second period in which the rotor experiences a positive cogging torque; and controlling, during the second period, zero current, or a braking current to oppose the rotation of the rotor, to flow in the phase winding of the motor such that the rotation of the rotor is driven by the positive cogging torque.

Description

A METHOD OF CONTROLLING A BRUSH LESS PERMANENT MAGNET

MOTOR

Field of the Invention

The present invention relates to a method of controlling a brushless permanent magnet motor.

Background of the Invention

There is a general desire to improve electric machines, such as brushless motors, in a number of ways. For example, improvements may be desired in terms of size, weight, power density, manufacturing cost, efficiency, reliability, and noise.

Summary of the Invention

According to a first aspect of the present invention there is provided a method of controlling a brushless permanent magnet motor, a rotor of the motor experiencing, during rotation of the rotor, a cogging torque that oscillates between a negative cogging torque that opposes the rotation of the rotor and a positive cogging toque that promotes the rotation of the rotor; the method comprising: determining a first period in which the rotor experiences the negative cogging torque; controlling, during the first period, current to flow in a phase winding of the motor to drive rotation of the rotor against the negative cogging torque; determining a second period in which the rotor experiences the positive cogging torque; and controlling, during the second period, zero current, or a braking current to oppose the rotation of the rotor, to flow in the phase winding of the motor such that the rotation of the rotor is driven by the positive cogging torque.

Cogging torque is inherent in brushless permanent magnet motors. In certain situations, such as when the motor is operating at low power (e.g. at low steady state rotational speeds), the cogging torque can dominate the output torque of the motor. In such cases, the oscillation of the cogging torque between positive and negative as the rotor rotates can lead to an appreciable corresponding oscillation in the output torque. This can lead to a tone or sound being produced.

For example, the oscillating output torque may cause vibration in the structure of the motor, which may accordingly produce a sound or tone. Such sound may be produced at a frequency that is audible to a user, and hence may be distracting for the user or others in the vicinity. In particular, a fourth harmonic of the output torque oscillation may be particularly prominent to the human ear at the operational speeds of the motor. However, driving rotation of the rotor against the negative cogging torque during the period in which negative cogging torque is experienced, but allowing the positive cogging torque to drive the rotation of the rotor during the period in which positive cogging torque is experienced, changes the frequency of the oscillations in the output torque. Accordingly, a frequency of sound associated with the oscillation can be altered, for example to frequencies that are not audible, or less audible, to the user. In particular, an amplitude of the fourth harmonic of the output torque oscillation, which may be particularly prominent to the human ear at operational rotational speeds, can be reduced. Note that the change in the frequency of the oscillations in the output torque is provided for without having to change the rotational frequency of the rotor itself.

Improved motor operation may therefore be provided.

Alternatively or additionally, driving rotation of the rotor against the negative cogging torque during the period in which negative cogging torque is experienced but allowing the positive cogging torque to drive the rotation of the rotor during the period in which positive cogging torque is experienced allows for efficient operation of the motor. Specifically, since the positive cogging torque drives the rotation of the rotor during the period in which positive cogging torque is experienced by the rotor, energy associated with otherwise driving this rotation can be saved. This may provide for a particularly efficient low power operation of the motor.

Controlling (during the first period) current to flow in the phase winding of the motor to drive rotation of the rotor against the negative cogging torque supresses the negative cogging torque. Controlling (during the second period) zero current to flow in the phase winding may allow for the rotation of the rotor to be driven efficiently by the positive cogging torque. For example, controlling zero current to flow in the phase winding may ensure that a braking current induced into the phase winding by the rotation of the rotor, and which would otherwise produce a torque opposing the rotation of the rotor, does not occur. Accordingly, the positive cogging torque may drive rotation of the rotor without opposing electromagnetic torques. This may allow for efficient motor operation. A braking current may be a current whose flow in the phase winding produces a torque that opposes the given rotation of the rotor. The braking current may brake the rotation of the rotor, driven by the positive cogging torque. Accordingly, controlling (during the second period) a braking current to oppose the rotation of the rotor to flow in the phase winding may allow for a controlled reduction of the output torque of the motor. This may allow for the motor to be controlled to operate at lower rotational speeds. In examples, the braking current that opposes the rotation of the rotor may be that induced into the phase winding by rotation of the rotor, for example, as a result of freewheeling the phase winding. Efficient braking may therefore be provided. Controlling both zero current and braking current opposing rotation of the rotor to flow (e.g. successively) in the phase winding, may provide for the extent to which the rotor braking is applied to be controlled.

The method may comprise, during the first period: for a third period associated with rising negative cogging torque, controlling a rise in current flow in the phase winding; and for a fourth period associated with falling negative cogging torque, controlling a fall in current flow in the phase winding. This allows an electromagnetic torque applied to the rotor (via the current flow in the phase winding) to appropriately oppose the negative cogging torque at different points in the cogging torque cycle. This may allow for effective suppression of the negative cogging torque.

Controlling the current flow may comprise, during the first period: controlling the rise and fall of current flow in the phase winding to approximate a shape of the respective rise and fall of the negative cogging torque. This may allow for the electromagnetic torque applied to the rotor (via the current flow in the phase winding) to closely follow and oppose the negative cogging torque over the negative cogging torque cycle. Yet more effective suppression of the negative cogging torque may therefore be provided.

Controlling current to flow in the phase winding of the motor to drive the rotation of the rotor may comprise, during the first period: applying a voltage to the phase winding and freewheeling the phase winding. This may provide an efficient and effective method by which to control the shape of the current flow as a function of time. Applying a voltage to the phase winding increases the current flowing in the phase winding and hence an electromagnetic torque applied to the rotor. Freewheeling the phase winding allows current flowing in the phase winding to dissipate relatively slowly and/or smoothly. This can assist in shaping the current flow in the phase winding as a function of time.

Controlling the rise in current flow in the phase winding may comprise: applying the voltage to the phase winding to increase the current to a current limit. This may allow for a relatively efficient and/or effective way to control the rise in the current flow, for example to mirror and oppose the corresponding rise in negative cogging torque. For example, applying the voltage may cause the current to rise as a function of time according to a curve that is similar to the curve according to which the negative cogging torque is rising. As above, this may provide for effective negative cogging torque suppression. Moreover, this is provided without necessarily providing complicated current shaping.

Controlling the fall in current flow may comprises freewheeling the phase winding to reduce the current flow in the phase winding. As above, freewheeling allows current flowing in the phase winding to dissipate relatively slowly and/or smoothly, which can assist in controlling a shape of the falling current flow.

Controlling the fall in current flow may comprise: successively freewheeling the phase winding and applying the voltage to the phase winding in order to control an overall fall in the current flow. This may allow for the shape of the falling current (and hence electromagnetic torque applied to the rotor) to be controlled to better mimic the shape of the falling negative cogging torque. For example, the successive freewheeling and voltage application, for example repeated a once or a plurality of times, can cause the current to fall function of time according to a curve that is similar to the curve according to which the negative cogging torque is falling. As above, this may allow for effective suppression of the negative cogging torque.

Controlling the current to flow in the phase winding of the motor to drive rotation of the rotor against the negative cogging torque may comprise controlling the current to flow in advance of the first period. For example, in advance of when the rotor begins to experience the negative cogging torque. There may be a delay between current flowing in the phase winding and the resulting electromagnetic torque being applied to the rotor. Accordingly, controlling the current to flow in advance of the first period, e.g. in advance of when the rotor experiences the negative cogging torque may allow sufficient time for the opposing electromagnetic torque to be provided precisely when the rotor begins to experience the negative cogging torque. A more effective suppression of the negative cogging torque may therefore be provided. It is noted that the advance may be relatively small, such as less than a quarter or less than an eighth of the cogging torque oscillation period. This may help ensure that the impact on the preceding positive cogging torque cycle in which the advance current flow occurs may be kept to a minimum.

The method may comprise for a first part of the second period, controlling a braking current to oppose the rotation of the rotor to flow in the phase winding; and for a second part of the second period, controlling zero current to flow in the phase winding. The braking current to oppose the rotation of the rotor allows for braking to be applied to the rotation of the rotor. This can allow for the motor to operate at reduced rotational speeds and/or with less output torque. Controlling the duration of the first part relative to the second part can allow for the extent of the braking, and hence the rotational speed of the rotor, to be controlled.

Controlling zero current to flow in the phase winding allows for the efficiency with which the rotor is driven by the positive cogging torque to be improved.

Controlling the braking current to oppose the rotation of the rotor to flow in the phase winding may comprise freewheeling the phase winding. As above, this may provide for efficient opposition of the rotation of the rotor, as no additional energy need be provided to the phase winding to achieve this. For example, the motor may be configured such that when the rotor experiences the positive cogging torque, a braking current is induced into the phase winding by the rotor, which induced current opposes the rotation of the rotor. Freewheeling the phase winding may allow this induced current to automatically brake the rotor, that is, without any additional energy needing to be provided to the phase winding.

Controlling zero current to flow in the phase winding may comprise applying zero-current clamping to the phase winding. This may provide simple and effective way to control zero current to flow in the phase winding. In examples, zero-current clamping may be provided by single device freewheeling or alternator switch off.

The method may comprise determining a zero-crossing of back Electromotive Force (back EMF) induced in the phase winding; and determining the first period 30 and/or the second period based on the determined zero-crossing of back EMF. In examples, the oscillation of the cogging torque is approximately in phase with the oscillation of the back EMF induced in the phase winding. In particular, in examples, a zero-crossing of the cogging torque occurs at approximately the same time as the zero-crossing of the back EMF. In practice, the cogging torque may have a small fixed phase shift relative to the back EMF, for example 1 degree. This may allow that, on start-up of the motor, the direction in which the rotor is caused to rotate can be reliably controlled. Nonetheless, in examples, this small fixed phase shift may be small enough that for the purposes of the current control, a zero-crossing of the back EMF may be considered to occur at the same time as a zero crossing of the cogging torque. The first period and/or the second period may then be determined relative to the determined zero-crossing of the back EMF. For example, the first period may be determined as running a quarter back EMF cycle from a zero-crossing of the back [ME, and the second period may be determined as running a quarter back EMF cycle from the end of the first period, or (equivalently) a quarter back EMF cycle in advance of a zero-crossing of the back EMF. In other examples, the small fixed phase shift between the back EMF zero crossing and a cogging torque zero crossing may be taken into account. For example, the small fixed phase shift may be known, and the first period and/or second period may be adjusted based on the small fixed phase shift. The zero-crossing of the back EMF can be determined using sensors (such as Hall effect sensors) or by sensor-less methods. Accordingly, determining the first period and/or the second period based on the zero-crossing of the back EMF may allow for a flexible, cost-effective and/or effective way to determine the first period and/or second period. In examples, the first period may be a half period of the oscillating cogging torque for which the rotor experiences the negative cogging torque; and/or the second period may be a half period of the oscillating cogging torque for which the rotor experiences the positive cogging torque. That is, the determined first period may the entire period for which the cogging torque is negative, and/or the determined second period may be the entire period for which the cogging torque is positive, for a given full cycle or oscillation of the cogging torque. In examples, the second period may occur immediately after the first period (and/or vice versa).

The method may comprise controlling, for the entire first period, current to flow in the phase winding of the motor to drive rotation of the rotor against the negative cogging torque; and/or controlling, for the entire second period, zero current, or the braking current to oppose the rotation of the rotor, to flow in the phase winding of the motor such that the rotation of the rotor is driven by the positive cogging torque. Controlling current flow over the entire first and/or second periods may allow for precise control over the output torque of the motor as a function of time. This may, in turn, allow for precise control over (e.g. reduction in) audible sound produced by thereby, and/or the efficiency of the motor.

The motor may comprise an inverter for controlling the current flow in the phase winding, the inverter may comprise a first pair of switches and a second pair of switches, the first and second pairs of switches movable between a plurality of switch configurations; and the method may comprise placing the inverter in a particular switch configuration to control current flow in the phase winding. This may provide a cost effective way to provide the control of current flow in the phase winding.

In examples, applying the voltage to the phase winding of the motor comprises placing the inverter in a first switch configuration where both of the first and second pairs of switches have one switch open and one switch closed. The phase winding may be commutated by, for each pair, closing the open switch and opening the closed switch.

In examples, freewheeling the phase winding during the first period comprises one or both of placing the inverter in a second switch configuration where one of the first and second pairs of switches is closed and the other of the first and second pairs of switches is open; and placing the inverter in a third switch configuration where one of the first and second pairs of switches is open, and the other of the first and second pairs of switches has one switch open and one switch closed.

Freewheeling using the second switch configuration (also referred to herein as dual device freewheeling) may, for example, be used to control the current to fall during the first period. In examples, freewheeling the phase winding during the second period may comprise placing the inverter in the second switch configuration. Accordingly, in some examples, freewheeling using the second switch configuration may be used to both allow the current flow to fall during the first period, and also allow the current to oppose the rotation of the rotor during the second period (to provide braking). In such situations, using the second switch configuration for both purposes may provide for a reduced number of different switching configurations to be used, and hence provide for efficient control. In examples, this may provide for a first low power mode operation of the motor. The current limit in the first low power mode operation of the motor may be a first value.

Freewheeling using the third switch configuration (also referred to herein as single device freewheeling) may, for example, be used to control the current to fall during the first period. In examples, zero-current clamping the phase winding comprises placing the inverter in the third switch configuration. Accordingly, in some examples, freewheeling using the third switch configuration may be used to both allow the current to fall during the first period, and also allow zero-current clamping during the second period (to allow the positive cogging torque to drive rotation of the rotor efficiently). In such situations, using the third switch configuration for both purposes may provide for a reduced number of different switching configurations to be used, and hence provide for efficient control. In examples, this may provide for a second low power mode operation of the motor. The current limit in the second low power mode operation of the motor may be a second value. The second value may be larger than the first value.

Freewheeling using the second switch configuration may be more efficient than the freewheeling using the third switch configuration. For example, where the switches of the first and second pairs of switches comprise transistors, in an arrangement where freewheeling takes place by utilising a switch configuration where one of a pair of transistors is open and the other of the pair of transistors is closed (as per the third switch configuration), current may flow through a body diode of the open transistor to provide freewheeling. This may, however, result in greater losses than an arrangement in which current flows through two closed transistors to provide freewheeling (as per the second switch configuration), for example due to the voltage drop across the body diode, leading to reduced efficiency.

Zero-current clamping the phase winding may comprise one or both of: placing the inverter in the third switch configuration; and placing the inverter in a fourth switch configuration where the first and second pairs of switches are open.

Placing the inverter in the fourth switch configuration can provide rapid zero-current clamping, as the inverter is effectively turned off. In examples, zero-current clamping in the second part of the second period may be provided by placing the inverter in the fourth switch configuration. This may provide that the current opposing the rotation of the rotor in the first part of the second period is brought quickly to zero. This may provide for precise control over the degree of braking applied to the rotor. Accordingly, in examples, the fourth switch configuration may be used for zero current clamping in the first low power mode operation of the motor. In examples, the third switch configuration may be used for zero-current clamping in the second low power mode operation of the motor.

The first and second pairs of switches may comprise high-and low-side pairs of switches. Freewheeling may comprise freewheeling around either the high-side pair of switches or freewheeling around the low-side pair of switches.

According to a second aspect of the invention, there is provided a brushless permanent magnet motor comprising a phase winding; and a controller configured to perform the method according to any one of the preceding claims.

The motor may comprise an inverter for controlling the current flow in the phase winding, the inverter may comprise a first pair of switches and a second pair of switches, the first and second pairs of switches movable between a plurality of switch configurations.

According to a third aspect of the invention, there is provided a data carrier comprising machine-readable instructions which, when executed by one or more controllers of a brushless permanent magnet motor, cause the one or more controllers to perform the method according to the first aspect.

According to a fourth aspect of the invention, there is provided an appliance comprising the brushless permanent magnet rotor according to the second aspect.

In examples, the appliance is a haircare appliance. For example, the haircare appliance may be a hair dryer. The motor may be configured, for example, to drive an impeller to generate an airflow. It may be desirable that the haircare appliance operate such that the generated airflow has a relatively low flow rate. For example, this may provide particularly gentle hair drying. This may correspond to a situation in which the motor operates at relatively low power, and hence a situation in which the cogging torque may dominate the output torque of the motor. Accordingly, this may correspond to a situation in which aspects of the invention may find particular utility. In other examples, the appliance may be a vacuum cleaner.

Optional features of aspects of the present invention may be equally applied to other aspects of the present invention, where appropriate.

Brief Description of the Drawings

Figure 1 is a first schematic view illustrating a motor drive system according to an

example;

Figure 2 is a second schematic view illustrating the motor drive system of Figure 1; Figure 3 is a table indicating switching states of the motor drive system of Figures 1 and 2; Figure 4 is a flow diagram illustrating a control method according to an example; Figure 5 is a schematic illustration of cogging torque, back EMF, and current flow in the phase winding of the motor as a function of time, according to a first example; Figure 6 is a schematic illustration of cogging torque, current flow in the phase 20 winding, and output torque of the motor as a function of time, according to the first example; Figure 7 is a schematic illustration of back EMF, current flow in the phase winding, and voltage applied to the phase winding of the motor as a function of time, according to a second example; Figure 8 is a schematic illustration of back EMF, current flow in the phase winding, and voltage applied to the phase winding of the motor as a function of time, according to a third example; Figure 9 is a graph illustrating a plot of sound level as a function of revolutions per minute for a benchmark control scheme and an example control scheme; Figure 10 is a schematic illustration of a vacuum cleaner comprising the motor drive system of Figures 1 and 2; and Figure 11 is a schematic illustration of a haircare appliance comprising the motor drive system of Figures 1 and 2.

Detailed Description of the Invention

A motor drive system 10 is shown in Figures 1 and 2. The motor drive system 10 is powered by a DC power supply 12, for example a battery, and comprises a brushless permanent magnet motor 14 and a control circuit 16. It will be recognised by a person skilled in the art that the methods of the present invention may be equally applicable to a motor system powered by an AC power supply, with appropriate modification of the circuitry, for example to include a rectifier to deliver DC power.

The motor 14 comprises a four-pole permanent-magnet rotor 18 that rotates relative to a four-pole stator 20. Although shown here as a four-pole permanent magnet rotor, it will be appreciated that the present invention may be applicable to motors having differing numbers of poles, for example eight poles. Conductive wires wound about the stator 20 are coupled together to form a single-phase winding 22. Whilst described here as a single-phase motor, it will be recognised by a person skilled in the art that the teachings of the present application may also be applicable to multiphase, for example three-phase, motors.

The control circuit 16 comprises a filter 24, an inverter 26, a gate driver module 28, a current sensor 30, a first voltage sensor 32, a second voltage sensor 33, and a controller 34.

The filter 24 comprises a link capacitor Cl that smooths relatively high-frequency ripple that arises from switching of the inverter 26.

The inverter 26 comprises a full bridge of four power switches 01-04 that couple the phase winding 22 to voltage rails V1, V2. Each of the switches 01-04 includes a freewheel diode (not shown). The switches 01 and 03 operate as a first pair of switches (a high side pair of witches), and the switches 02 and 04 operate as a second pair of switches (a low side pair of switches). As illustrated in Figure 2, the switches 01 and 03 comprise high-side switches, and the switches 02 and 04 comprise low-side switches. The first and second pairs of switches are movable between a plurality of switch configurations. Controlling current flow in the phase winding 22 comprises placing the inverter in a particular switch configuration.

The gate driver module 28 drives the opening and closing of the switches 01-04 in response to control signals received from the controller 34.

The current sensor 30 comprises a shunt resistor R1 located between the inverter and the zero-volt rail V2. The voltage across the current sensor 30 provides a measure of the current in the phase winding 22 when connected to the power supply 12. The voltage across the current sensor 30 is output to the controller 34 as a signal, I_SENSE. It will be recognised that in this embodiment it is not possible to measure current in the phase winding 22 during freewheeling, but that alternative embodiments where this is possible, for example via the use of a plurality of shunt resistors, are also envisaged.

The first voltage sensor 32 comprises a voltage divider in the form of resistors R2 and R3 that are located between the DC voltage rail Viand the zero-volt rail V2.

The first voltage sensor 32 outputs a signal, V_DC, to the controller 34, which represents a scaled-down measure of the supply voltage provided by the power supply 12.

The second voltage sensor 33 comprises a pair of voltage dividers constituted by resistors R4, R5, R6, and R7, which are connected to either side of the phase winding 22. The second voltage sensor 33 provides a signal, indicative of back EMF, bEMF, induced in the phase winding 22, to the controller.

The controller 34 comprises a microcontroller having a processor, a memory device, and a plurality of peripherals (e.g. ADC, comparators, timers etc.). In an alternative embodiment, the controller 34 may comprise a state machine. The memory device stores instructions for execution by the processor, as well as control parameters that are employed by the processor during operation. The controller 34 is responsible for controlling the operation of the motor 14 and generates four control signals 61-S4 for controlling each of the four power switches 01-04. The control signals are output to the gate driver module 28, which in response drives the opening and closing of the switches 01-04.

During normal operation, the controller 34 estimates the position of the rotor 18 using a sensorless control scheme, i.e. without the use of a Hall sensor or the like, by using software to estimate a waveform indicative of back EMF induced in the phase winding 22 via the signals V_DC and I_SENSE. In particular, zero-crossings of back [FM induced in the phase winding 22 can be estimated to estimate aligned positions of the rotor 18. The details of such a control scheme will not be described here for the sake of brevity, but can be found, for example, in published GB patent application GB2582612. Another sensorless control scheme that utilises hardware components to estimate back EMF induced in the phase winding 22 is disclosed in published PCT patent application W02013132247A1. With knowledge of the position of the rotor 18 in normal operation, the controller 34 generates the control signals 51-54.

Figure 3 summarises the allowed states of the switches 01-04 in response to the control signals S1-S4 output by the controller 34, and such allowed states may be referred to as switch configurations here. Hereafter, the terms 'set and 'clear' will be used to indicate that a signal has been pulled logically high and low respectively.

In order to apply a voltage to the phase winding 22 to excite the phase winding 22, the controller 34 places the inverter 26 in a first switch configuration where both the first and second pairs of switches have one switch open and one switch closed. For example, as can be seen from Figure 3, the controller 34 sets Si and 54, and clears 52 and 53 in order to excite the phase winding 22 from left to right. Conversely, the controller 34 sets S2 and S3, and clears Si and S4 in order to excite the phase winding 22 from right to left.

In order to freewheel the phase winding 22, the controller 34 may place the inverter 26 in a second switch configuration where one of the first and second pairs of switches is closed and the other of the first and second pairs of switches is open. Such freewheeling may be referred to herein as 'dual device freewheeling' or 'body freewheel'. For example, the controller 34 may clear Si and S3, and set S2 and S4 in order to freewheel the phase winding 22. Such freewheeling enables current in the phase winding 22 to re-circulate around the low-side loop of the inverter 26. In this example, the power switches 01-Q4 are capable of conducting in both directions. Accordingly, the controller 34 closes both low-side switches 02,04 during freewheeling such that current flows through the switches 02,04 rather than through the less efficient diodes (not shown).

In order to freewheel the phase winding 22, the controller 34 may (alternatively) place the inverter 26 in a third switch configuration where one of the first and second pairs of switches is open, and the other of the first and second pairs of switches has one switch open and one switch closed. Such freewheeling may be referred to herein as "single device freewheeling" or "diode free wheel". For example, the controller 34 may clear Si, S2 and S3, and set S4 so as to freewheel the phase winding 22 from left to right. Current in the low-side loop of the inverter 26 is able to flow down through the closed low-side switch 04 and up through the diode of the open low-side switch 02 (clockwise in the sense of Figure 2). However, due to the diode of the open low side switch 02, current is not able to flow in the opposite direction. Accordingly, where current would otherwise flow anti-clockwise in the sense of Figure 2, this switch configuration can be used to zero-current clamp the current in the phase winding 22. As another example, the controller 34 may clear Si, S3 and S4, and set S2 in order to freewheel the phase winding 22 from right to left. Current in the low-side loop of the inverter 26 is able to flow down through the closed low-side switch 02 and up through the diode of the open low-side switch 04 (anticlockwise in the sense of Figure 2). However, due to the diode of the open low side switch 04, current is not able to flow in the opposite direction. Accordingly, where current would otherwise flow clockwise in the sense of Figure 2, this switch configuration can be used to zero-current clamp the current in the phase winding 22.

As a way to prevent current from flowing through the phase winding 22 or to zero-current clamp the current flowing in the phase winding 22, the controller may (alternatively) place the inverter 26 in a fourth switch configuration where the first and second pairs of switches are open. This in effect turns the inverter off.

Appropriate control of the switches 01-04 can be used to drive the rotor 18 at speeds up to or in excess of 100k rpm during normal operation, for example in a steady-state mode. In particular, the phase winding 22 can be excited and freewheeled sequentially, with commutation of the phase winding 22 occurring between successive excitations of the phase winding 22.

The brushless permanent magnet motor 14 experiences the effects of cogging torque. Cogging torque results from the magnetic field produced by the permanent magnets of the rotor 18 linking with the iron cores 21 of the stator 20 about which the phase windings 22 are wound. As the rotor 18 rotates away from a first strong linking position, the rotor 18 experiences a negative cogging torque that opposes the rotation (that is, a torque that urges the rotor 18 to return to the first strong linking position). However, continued rotation of the rotor 18 will at some point bring the rotor 18 closer to the next strong linking position, at which point the rotor experiences a positive cogging torque that promotes the rotation (that is, a torque that urges the rotor 18 to rotate to the next strong linking position). Accordingly, as the rotor 18 rotates in a given direction, the cogging torque experienced by the rotor 18 oscillates between negative cogging torque that opposes the rotation of the rotor and positive cogging torque that promotes the rotation of the rotor 18. In the example of the 4-pole rotor and 4-pole stator motor 14, the cogging torque may undergo 4 full oscillations or cycles of cogging torque for every one full revolution of the rotor 18.

In certain situations, such as when the motor 14 is operating at low power (e.g. at low rotational speeds), the cogging torque can dominate the output torque of the motor 14. In such cases, the oscillation of the cogging torque between positive and negative as the rotor 18 rotates can lead to an appreciable corresponding oscillation in the output torque. This can lead to a tone or sound being produced at a frequency that is audible to a user, and hence may be distracting for the user or others in the vicinity. Alternatively or additionally, not taking the cogging torque into account when operating the motor can lead to inefficiencies in the operation of the motor.

Figure 4 illustrates a method 400 of controlling a brushless permanent magnet motor, such as the motor 14 described above with reference to Figures 1 to 3. As mentioned above, a rotor 18 of the motor 14 experiences, during rotation of the rotor 18 (that is, during rotation of the rotor 18 in a given direction), a cogging torque that oscillates between a negative cogging torque that opposes the rotation of the rotor 18 and a positive cogging toque that promotes the rotation of the rotor 18.

The method 400 comprises, in step 402, determining a first period in which the rotor 18 experiences the negative cogging torque. The method further comprises, in step 404, controlling, during the first period, current to flow in a phase winding 22 of the motor 14 to drive rotation of the rotor 18 against the negative cogging torque. The method further comprises, in step 406, determining a second period in which the rotor 18 experiences the positive cogging torque. The method further comprises, in step 408, controlling, during the second period, zero current, or a braking current to oppose the rotation of the rotor 18, to flow in the phase winding 22 of the motor 14 such that the rotation of the rotor 18 is driven by the positive cogging torque.

Figures 5 to 7 illustrate a first example of the method of controlling the motor 14.

In this first example, the control provides a first low power mode operation of the motor 14. Figure 5 illustrates a plot of back EMF 510, cogging torque 502, and current flow 508 in the phase winding 22 of the motor 14 as a function of time while the rotor 18 rotates in a given direction. Figure 6 illustrates a plot of that same current flow 508 and cogging torque 502, and also output torque (612) of the rotor 18 as a function of time while the rotor rotates. Figure 7 illustrates a plot of that same current flow 508 and back EMF 510, and also voltage 702 across the phase winding 22 as a function of time as the rotor 18 rotates.

Referring to Figures 5 to 7, in broad overview, during a first period (e.g. between times ti and ts) in which the cogging torque 502 is negative 504, voltage 732, 722 is applied to the phase winding to cause the current 508 to flow in the phase winding 22 to drive rotation of the rotor 18 against the negative cogging torque 504. That is, the output torque 612 (resulting from a combination of the cogging torque 502 and the electromagnetic torque resulting from the current flow 508) is positive and hence the rotor 18 rotates in the given direction. During a second period (e.g. between times t3 and t5) in which the cogging torque 502 is positive 596, zero-current clamping 524 is applied and hence zero current flows in the phase winding. Accordingly, the output torque 612 is the positive cogging torque 506, and the rotation of the rotor 18 in the given direction is driven by the positive cogging torque 506. After this, the phase winding is commutated, and the control repeated but with excitation of the phase winding 22 in the opposite sense (e.g. left-to-right excitation commutated to right-to-left excitation), in order to continue to drive rotation of the rotor 18 in the given direction.

According to this scheme, the contribution of the negative cogging torque 504 to the output torque 612 is supressed relative to that of the positive cogging torque 506. Accordingly, the frequency of the oscillation in the output torque 516 due to the cogging torque 502 is altered (for example as compared to the frequency of the cogging torque 502 itself). Accordingly, a frequency of sound associated with the oscillation can be altered (for example to a frequency that is outside the audible range for a human). Further, the rotation of the rotor 18 being driven by the positive cogging torque 506 during the second period may allow for efficient low power motor operation as this saves current that may otherwise have been used to drive rotation of the rotor electromagnetically during this period.

Referring to Figures 5 to 7 now in more detail, as the rotor 18 rotates in the given direction, the back EMF 510 induced into the phase winding 22 by the rotor oscillates. As described above, the back EMF 510 is indicative of the rotational position of the rotor 18 relative to the stator 22. In particular, zero-crossings of the back EMF 510 (e.g. times ti, t5 and t7) correspond to aligned positions of the rotor 18 (that is, where the poles of the rotor 18 are aligned with the poles of the stator 20). The cogging torque 502 oscillates between negative cogging torque 504 and positive cogging torque 506. For example, for a first period (e.g. between times ti and t3), the rotor 18 experiences the negative cogging torque 504, and for a second, immediately subsequent, period (e.g. between times t3 and t5), the rotor 18 experiences the positive cogging torque 506. The back EMF 510 and the cogging torque 502 are aligned such that for every half cycle of the back EMF 510, the cogging torque 502 undergoes a full cycle or oscillation. As mentioned above, at aligned positions of the rotor 18 (that is, at zero-crossings of the back EMF 510) the cogging torque is zero (e.g. times ti, t5 and t7). The cogging torque is also zero at anti-aligned positions of the rotor 18 (that is, half-way between successive zero crossings of the back EMF 510, e.g. times t3 and thy In practice, the cogging torque 502 may have a small fixed phase shift relative to the back EMF 510, for example 1 a maximum of degree. This may allow that, on start-up of the motor, the direction in which the rotor 18 is caused to rotate can be reliably controlled. Nonetheless, in examples, this small fixed phase shift may be small enough that for the purposes of determining the first period and/or the second period (and hence for the purposes of the current control disclosed herein) a zero-crossing of the back EMF 510 may be considered to occur at the same time as a respective zero crossing of the cogging torque 502 (e.g. at times ti, t5 and t7). In other examples, the small fixed phase shift between the back EMF zero crossing and a cogging torque zero crossing may be taken into account. For example, the small fixed phase shift may be known, and the first period and/or second period may be adjusted accordingly based on the known small fixed phase shift.

As mentioned, the method involves determining a first period (e.g. between times ti and t3) in which the rotor experiences the negative cogging torque 504 and determining a second period (e.g. between times t3 and t5) in which the rotor 108 experiences the positive cogging torque 506. This may comprise determining a zero-crossing of back EMF 510, and determining the first period ti-t3 and/or the second period t3-t5 based on the determined zero-crossing of back EMF 510.

Zero-crossings of back EMF may be determined, for example as described above, using sensor-less methods. As one example, a first back EMF zero crossing time ti and a second, subsequent back EMF zero crossing time t5 may be determined (e.g. predicted). A time t3 halfway between these zero crossing times ti and t5 may be determined (e.g. calculated). The first period ti-t3 may then be determined as from the first back EMF zero-crossing time ti to the determined halfway time t3, and the second period ti-ts may be determined as from the determined halfway time t3 to the second back EMF zero crossing time t5. As another example, the first period may be determined as a quarter back-EMF cycle immediately following a given back EMF zero crossing time (e.g. ts) and the second period may be determined as a quarter back EMF cycle immediately preceding the given back EMF zero crossing time (e.g. t5). Other methods may be used. In examples where the small fixed phase shift between the back EMF 510 and the cogging torque 502 is taken into account, the first period and/or the second period may be determined as offset by a time period corresponding to the small fixed phase shift, as compared to the time periods determined above. In the examples described herein, the determined first period is the entire period (e.g. from time ti to time t3) for which the cogging torque 502 is negative, and the determined second period is the entire period (e.g. from time t3 to time t5) for which the cogging torque 502 is positive, for a given full cycle or oscillation of the cogging torque 502. In other examples, the determined first and/or second periods may only be a proportion of these half-cycles.

As mentioned, the method comprises controlling, during the first period ti-t3, current 508 to flow in the phase winding 22 to drive rotation of the rotor 18 against the negative cogging torque 504. As can be seen in Figures 5 to 7, during the first period ti-t3, current 508 is controlled to flow in the phase winding 22. The current 508 flowing in the phase winding 22 during the first period ti 43 produces a magnetic field that interacts with the magnetic field of the rotor 18 to apply a torque to the rotor 18 in the direction of rotation of the rotor 18. This generated electromagnetic torque is larger than the negative cogging torque and hence, as best seen in Figure 6, the total or output torque 612 is positive (that is, in the direction of rotation of the rotor 18). Accordingly, the rotation of the rotor 18 is driven against the cogging torque.

In this example, controlling the current 508 to flow in the phase winding 22 during the first period ti-t3 comprises, for a third period ti-t2 associated with rising negative cogging torque 504 (that is, associated with increasing magnitude of negative cogging torque), controlling a rise in the current flow 508 in the phase winding 22. In this example, controlling the current 508 to flow in the phase winding 22 during the first period ti-t3 comprises, for a fourth period t2-t3 associated with falling negative cogging torque 504 (that is, associated with decreasing magnitude of negative cogging torque), controlling a fall in the current flow in the phase winding 22. Specifically, during the first period ti-t3, the rise and fall of current flow 508 in the phase winding 22 is controlled to approximate the shape of the respective rise and fall of the negative cogging torque 504. This allows the electromagnetic torque applied to the rotor 18 (via the current flow in the phase winding 22) to appropriately oppose the negative cogging torque 504 at different points in the cogging torque cycle. This may allow for effective suppression of the negative cogging torque.

In this example, controlling the current 508 to flow in the phase winding during the first period ti-tscom prises applying a voltage 702 to the phase winding 22 and freewheeling 520 the phase winding 22. Specifically, increases in the current 508 are controlled by applying a voltage to the phase winding 22, and decreases in the current are controlled by freewheeling the phase winding 22. In this example, applying the voltage 702 to the phase winding 22 to increase the current 508 comprises placing the inverter 26 in the first switch configuration. For example, assuming that in the first period ti-t3 the rotation of the rotor 18 is driven by leftto-right excitation of the phase winding 22, applying the voltage may comprise the controller 34 setting Si and 54 and clearing S2 and S3. Freewheeling the phase winding 22 may comprise placing the inverter in the second configuration (body freewheel) or the third configuration (diode freewheel). In this example, freewheeling the phase winding 22 comprises placing the inverter in the third configuration (diode freewheel). For example, assuming that in the first period ti -t3 the rotation of the rotor 18 is driven by left-to-right excitation of the phase winding 22, the freewheeling may be freewheeling the phase winding from leftto-right (that is, clearing Si, S2 and S3, and setting S4). This may be useful as this not only allows the current that is flowing in the phase winding 22 to decrease but also ensures that a current does not flow in a reverse direction (e.g. automatically provides zero-current clamping in the second period to-to, as described in more detail below). In Figures 5 to 7, numeral 524 refers to a 'diode freewheel' period in which the phase winding 22 is diode freewheeled.

Specifically, in this example, between times to and t2, a voltage 732 is applied to the phase winding 22, which causes the current 508 to increase up to a current limit. The current limit in this first example may be a first value, such as 2.2A.

Between times t2 and to, the control comprises successively freewheeling 520 the phase winding (in which case zero voltage 720 is applied) and exciting 522 the phase winding 22 by applying the voltage 722 to the phase winding. In this example, in between times t2 and to, the phase winding is freewheeled four times (the freewheels being separated by respective periods 522 of excitation).

Successive freewheeling and excitation may allow for control over the shape of the fall in current 508 during the first period ti-to, for example so as to mimic the shape of the negative cogging torque, thereby to provide effective negative cogging torque suppression. In Figures 5 to 7, the numeral 532 refers to an 'initial conduction' period during which a voltage is initially applied to the phase winding 22 in order to increase the current 508 to the current limit; and the numeral 526 refers to a 'full conduction' period over which a voltage is (intermittently) applied to the phase winding 22 to cause the current to flow.

As can be seen in Figures 5 to 7, in this example, the voltage 732 is applied across the phase winding 22, and current 508 is thereby caused to flow in the phase winding 22, slightly in advance of the first period (e.g. ti-b) For example, the current 508 may be controlled to start slightly in advance of the zero crossings of cogging torque 502 from positive 506 to negative 504 (e.g. at time to, slightly in advance of ti, or e.g. when the phase winding 22 is commutated, at ta, slightly in advance of to). This advance time (also indicated in Figures 5 to 7 by numeral 530) may allow sufficient time for the current flow 508 to produce the electromagnetic torque for when the rotor 18 begins to experience the negative cogging torque. As also can be seen in the output torque 612 in Figure 6, a more effective suppression of the negative cogging torque may therefore be provided. The advance time may be kept relatively small so as to reduce the impact of the current flow on the preceding period in which the cogging torque is positive (and hence which current flow would oppose the rotation of the rotor 18).

As the first period (e.g. ti-t3) comes to an end (e.g. slightly in advance of t3), the phase winding 22 is freewheeled and the current 508 accordingly reduces, until at time t3 the current 508 is zero. Immediately following the first period (e.g. ti-t3) is the second period (e.g. ts-t5).

As mentioned, during the second period (e.g. between times t3 and t5) in which the cogging torque 502 is positive 596, zero current 508 flows in the phase winding. During the second period, as best seen in Figure 6, the output torque 612 is the positive cogging torque 506, and accordingly, the rotation of the rotor 18 in the given direction is driven by the positive cogging torque 506. More specifically, in this example, between times t3 and ta, zero-current clamping is applied and zero current flows in the phase winding. However, due to the advance in the voltage application for the following negative cogging period, between times tit and t5, a current flows that opposes rotation of the rotor 18. However, as mentioned above, this advance time is small and the impact of this on the rotation of the rotor 18 is relatively small. Indeed, best seen in Figure 6, the impact of the current flow in this advance time on the output torque 612 in the second period is low, and the positive cogging torque 506 still drives the rotation of the rotor 18.

As mentioned, controlling zero current to flow in the phase winding 22 during the second period (e.g. between times ta and ta) may comprise applying zero-current clamping to the phase winding 22. In Figures 5 to 7, the numeral 528 refers to a 'current clamping' period in which zero-current clamping is applied to the phase winding 22. In examples, zero-current clamping may be provided by placing the inverter 26 in the third switch configuration (single device or diode freewheeling) or by placing the inverter 26 in the fourth switch configuration (alternator switch off). In this example, the zero-current clamping is provided by single device freewheeling. Accordingly, as can be seen in Figure 7, between times t3 and ta, the voltage 702 across the phase winding 22 is the back [ME 510, but zero current flows in the phase winding 22 (as this is prevented by the diode). Using the single device freewheeling to provide this zero-current clamping has advantages over using alternator switch off. Firstly, the single device freewheeling will automatically perform zero current clamping when the current falls to zero at t3, whereas alternator switch off may need to be controlled to occur at time ts, which may be more complicated. Secondly, in cases where the freewheeling during the first period is provided by single device freewheeling, use of single device freewheeling also in the second period may reduce the number of different switch configurations that need to be used to provide the control, which may be more efficient. This stretch of single device freewheeling that occurs in both the first period and the second period is indicated in Figure 5 with the reference 524.

Figure 8 illustrates control of the motor 14 according to a second example. In this example, a second low power mode operation of the motor 14 is provided, lower than the first low power mode operation described above (that is, the rotational speeds at which the rotor is driven may be lower in the second low power mode that in the first low power mode operation). In this example, braking current is controlled so as to brake the rotor 18 during the second period, which allows the motor to operate at lower speeds. Figure 8 illustrates a plot of current flow 808 and back EMF 810, and also voltage 802 across the phase winding 22 as a function of time as the rotor 18 rotates in the given direction. Although cogging torque is not shown explicitly in Figure 8, the relationship between the cogging torque and the back EMF 810 in the second example illustrated Figure 8 is the same as for the first example illustrated in Figures 5 to 7.

The control according to the second example as illustrated in Figure 8 is similar to the control according to the first example as illustrated in Figure 7. Similar to the first example, in this second example, there is a first period (e.g. from ti to t3') in which the cogging torque is negative, and a second period (e.g. from t3' to t5') in which the cogging torque is positive. Similarly to the first example, in this second example, during the first period (e.g. from ti to t3') current is controlled to flow in the phase winding 22 to drive rotation of the rotor 18 against the negative cogging torque; and during the second period (e.g. from t3 to t5'), zero current, or a braking current to oppose the rotation of the rotor, is controlled to flow in the phase winding 22 such that the rotation of the rotor 18 is driven by the positive cogging torque.

However, in this second example, rotor braking is applied during the second period (e.g. from t3' to 15') to provide a reduction in the rotational speed of the rotor 18. Specifically, in this second example, for a first part (e.g. between times t3' and t3A) of the second period, current flow 802 in the phase winding 22 is controlled to oppose the rotation of the rotor 18; and for a second part (e.g. between times t3A and te) of the second period, zero current is controlled to flow in the phase winding. In examples, the duration of the first part (e.g. between times t3' and t3A) in which the braking is applied can be controlled relative to the duration of the second part (e.g. between times t3A and t4') to allow for the extent of the braking, and hence the rotational speed of the rotor, to be controlled.

Referring to Figure 8, at a time to' slightly in advance of when the rotor 18 begins to experience the negative cogging torque at ti, an initial voltage 832 is applied to excite the phase winding 22 until time t2', causing the current 808 to rise in the phase winding 22 to a current limit at time t2'. This excitation may be provided by placing the inverter in the first switch configuration (e.g. left-to-right excitation) The time between to' and ti in this example is less than the time between to and t2 in the first example, and accordingly the current limit to which the current rises in this second example (e.g. 1.9A) is lower than that of the first example (e.g. 2.2 A). The produced electromagnetic torque, and hence output torque, in the first period may accordingly be lower in this second example than in the first example. This may contribute to the lower power operation of the motor in this second example.

The initial current rise between times to' and to' is followed by a fall in current between times ti and to'. The fall in current 808 is provided by successive freewheeling 820 and excitation 822 of the phase winding. The freewheeling may be provided by placing the inverter in the second switch configuration (body freewheeling) and the excitation may be provided by placing the inverter in the first switch configuration (left-to-right excitation). The succession ends in body freewheeling the phase winding 22, and the current 808 falls until it reaches zero at time t3'. However, at time to', there is a zero crossing of cogging torque from negative to positive and the back EMF 810 ceases to rise and begins to fall.

Accordingly, at time to', since the phase winding 22 is still being body freewheeled a 'negative' current (that is, a current in an opposite direction to that of the current caused to flow in the first period ti to to') is induced to flow in the phase winding 22. In Figure 8, the numeral 870 refers to the total period over which this body freewheeling is applied. The 'negative' current produces an electromagnetic field that opposes the rotation of the rotor 18, and hence the rotation of the rotor 18 driven by the positive cogging torque is braked. The output torque in the second period may accordingly be lower in this second example than in the first example. This may contribute to the lower power operation of the motor in this second example.

The braking continues until time toA' when zero-current clamping is applied to the phase winding until time t4'. The zero-current clamping causes the 'negative' current to return to zero, and remain at zero until time t4'. Accordingly, for the second part (e.g. between times toA' and t4') of the second period, zero current is controlled to flow in the phase winding 22. In this example, the zero-current clamping between times t3A' and ta' is provided by placing the inverter in the fourth switch configuration (inverter turn off). This period in which the inverter is turned off is also indicated in Figure 8 by numeral 828. Turning the inverter off may provide that the current 809 opposing the rotation of the rotor 18 in the first part (e.g. between times t3' and t3A') of the second period is brought quickly to zero for the second part (e.g. between times t3A' and t4') of the second period. The turning off of the inverter at time t3A' charges the capacitor Cl, which as can be seen in Figure 8, causes the voltage across the phase winding 22 to increase. This can allow for a faster decay of the negative current, for example as compared to using diode freewheeling where the negative current would decay from the resistance in the phase winding 22.

The current is zero-clamped until time ta' where, slightly in advance of the time t5' where the back EMF 810 has a zero crossing from positive to negative, the phase winding 22 is commutated. Accordingly, at t4', the phase winding 22 is excited again (e.g. right-to-left excitation) to continue to drive rotation of the rotor 18 in the next negative cogging period.

In the above examples, described with reference to Figures 4 to 8, the control method alters the frequency of the oscillation in the output torque 516 and accordingly a frequency of sound associated with the oscillation can be altered (for example to a frequency that outside the audible range of a human). Referring now to Figure 9, there is illustrated a plot of noise level (in dB) as a function of rotor rotational speed On RPM) for two motor control schemes. The solid lines represent plots using a control method according to examples described above (referred to in Figure 9 as 'proposal'). The dashed lines represent plots for a comparative control scheme (referred to in Figure 9 as benchmark'). The comparative control scheme, in brief, is a standard 'dual switch' control scheme where one pulse of excitation voltage (to drive rotation of the rotor 18 in a given direction) is applied for every quarter cycle of back EMF. For each of the 'proposal' and 'benchmark', two plots are shown, one being of the amplitude of the fourth harmonic (H4) of the output torque oscillation and the other being of the eighth harmonic (H8) of the output torque oscillation, as shown. The fourth harmonic (H4) is known to be particularly prominent to the human ear at operational rotational speeds of the rotor 18. As can be seen in Figure 9, the method of examples disclosed herein reduces the amplitude of the fourth harmonic as compared to the benchmark across all tested rotor speeds. Across the tested rotational speeds, an average reduction of 6dB of the amplitude of the fourth harmonic is observed as compared to the benchmark. There is also observed a reduction in the amplitude of the eighth harmonic.

Referring to Figures 10 and 11, there are illustrated two examples of an appliance 1000, 1100 comprising the brushless permanent magnetic motor 14. In either case the appliance 1000, 1100 may comprise the motor system 10 described above with reference to Figures 1 to 9. For example, the appliance 1000, 1100 may comprise a brushless permanent magnet motor comprising: a phase winding 22; and a controller configured to perform the method according to any one of the examples described above with reference to Figures 1 to 9. For example, the appliance 1000, 1100 may comprise a controller configured to perform the method of any the examples described above with reference to Figures 1 to 9. 20. For example, the appliance 1000, 11000 may comprise a memory storing instruction which, when executed by one or more controllers of a brushless permanent magnet motor, cause the one or more controllers to perform the method according to any one of the examples described above with reference to Figures 1 to 9. In some examples, the appliance 1000, 1000 may be controllable to operate in one or both of the first low power mode (e.g. described above with reference to Figures 5 t o7) and the second lower power level mode (e.g. described above with reference to Figure 8). For example, the power level mode in which the appliance is to operate may be set by a user using a user interface (not shown).

In Figure 10, the appliance is a vacuum cleaner 1000. In Figure 11, the appliance is a haircare appliance 1100, specifically a hair dryer 1100. For example, the motor 14 may be configured to drive an impeller to generate an airflow. It may be desirable that the haircare appliance 1100 operates such that the generated airflow has a relatively low flow rate. For example, this may provide particularly gentle hair drying. This may correspond to a situation in which the motor operates at relatively low power (low rotational speeds), and hence a situation in which the cogging torque may dominate the output torque of the motor. Accordingly, this may correspond to a situation in which the example methods described above with reference to Figures 1 to 9 may find particular utility. For example, in order to generate the airflow with a relatively low flow rate, the appliance may be configured to operate in the first or second low power modes of operation described above with reference to Figures 1 to 9.

Whilst particular examples have been described, it should be understood that these are illustrative only and that various modifications may be made without departing from the scope of the invention as defined by the claims.

Claims

Claims 1. A method of controlling a brushless permanent magnet motor, a rotor of the motor experiencing, during rotation of the rotor, a cogging torque that oscillates between a negative cogging torque that opposes the rotation of the rotor and a positive cogging toque that promotes the rotation of the rotor; the method comprising: determining a first period in which the rotor experiences the negative cogging torque; controlling, during the first period, current to flow in a phase winding of the motor to drive rotation of the rotor against the negative cogging torque; determining a second period in which the rotor experiences the positive cogging torque; and controlling, during the second period, zero current, or a braking current to oppose the rotation of the rotor, to flow in the phase winding of the motor such that the rotation of the rotor is driven by the positive cogging torque.
2. The method according to claim 1, wherein the method comprises, during the first period: for a third period associated with rising negative cogging torque, controlling a rise in current flow in the phase winding; and for a fourth period associated with falling negative cogging torque, controlling a fall in current flow in the phase winding.
3. The method according to claim 2, wherein controlling the current flow comprises, during the first period: controlling the rise and fall of current flow in the phase winding to approximate a shape of the respective rise and fall of the negative cogging torque.
4. The method according to any one of the preceding claims, wherein controlling current to flow in the phase winding of the motor to drive the rotation of the rotor comprises, during the first period: applying a voltage to the phase winding and freewheeling the phase 5 winding.
5. The method according to claim 4 when dependant on claim 2 or claim 3, wherein controlling the rise in current flow in the phase winding comprises: applying the voltage to the phase winding to increase the current to a current limit.
6. The method according to claim 5 or claim 4, when dependant on claim 2 or claim 3, wherein controlling the fall in current flow comprises: freewheeling the phase winding to reduce the current flow in the phase 15 winding.
7. The method according to claim 6, wherein controlling the fall in current flow comprises: successively freewheeling the phase winding and applying the voltage to the phase winding in order to control an overall fall in the current flow.
8. The method according to any one of the preceding claims, wherein controlling the current to flow in the phase winding of the motor to drive rotation of the rotor against the negative cogging torque comprises controlling the current to flow in advance of the first period.
9. The method according to any one of the preceding claims, wherein the method comprises: for a first part of the second period, controlling a braking current to oppose the rotation of the rotor to flow in the phase winding; and for a second part of the second period, controlling zero current to flow in the phase winding.
10. The method according to any one of the preceding claims, wherein controlling the braking current to oppose the rotation of the rotor to flow in the phase winding comprises freewheeling the phase winding.
11. The method according to any one of the preceding claims, wherein controlling zero current to flow in the phase winding comprises applying zero-current clamping to the phase winding.
12. The method according to any one of the preceding claims, wherein the method comprises: determining a zero-crossing of back EMF induced in the phase winding; and determining the first period and/or the second period based on the determined zero-crossing of back EMF.
13. The method according to any one of the preceding claims, wherein the first period is a half period of the oscillating cogging torque for which the rotor experiences the negative cogging torque; and/or the second period is a half period of the oscillating cogging torque for which the rotor experiences the positive cogging torque.
14. The method according to any one of the preceding claims, wherein the method comprises: controlling, for the entire first period, current to flow in the phase winding of the motor to drive rotation of the rotor against the negative cogging torque; and/or controlling, for the entire second period, zero current, or a braking current to oppose the rotation of the rotor, to flow in the phase winding of the motor such that the rotation of the rotor is driven by the positive cogging torque.
15. The method according to any one of the preceding claims, wherein the motor comprises an inverter for controlling the current flow in the phase winding, the inverter comprising a first pair of switches and a second pair of switches, the first and second pairs of switches movable between a plurality of switch configurations; and wherein the method comprises: placing the inverter in a particular switch configuration to control current flow in the phase winding.
16. The method according to claim 15, when dependant on claim 4, wherein applying the voltage to the phase winding of the motor comprises: placing the inverter in a first switch configuration where both of the first and second pairs of switches have one switch open and one switch closed.
17. The method according to claim 15 or claim 16, when dependant on claim 4, wherein freewheeling the phase winding during the first period comprises one or both of: placing the inverter in a second switch configuration where one of the first and second pairs of switches is closed and the other of the first and second pairs of switches is open; and placing the inverter in a third switch configuration where one of the first and second pairs of switches is open, and the other of the first and second pairs of switches has one switch open and one switch closed.
18. The method according to any one of claim 15 to claim 17, when dependant on claim 10, wherein freewheeling the phase winding during the second period 30 comprises: placing the inverter in a second switch configuration where one of the first and second pairs of switches is closed and the other of the first and second pairs of switches is open.
19. The method according to any one of claim 15 to claim 18, when dependant on claim 11, wherein zero-current clamping the phase winding comprises one or both of: placing the inverter in a third switch configuration where one of the first and second pairs of switches is open, and the other of the first and second pairs 10 of switches has one switch open and one switch closed placing the inverter in a fourth switch configuration where the first and second pairs of switches are open.
20. A brushless permanent magnet motor comprising: a phase winding, and a controller configured to perform the method according to any one of the preceding claims.
21. The brushless motor according to claim 20, wherein the motor comprises an inverter for controlling the current flow in the phase winding, the inverter comprising a first pair of switches and a second pair of switches, the first and second pairs of switches movable between a plurality of switch configurations.
22. A data carrier comprising machine-readable instructions which, when executed by one or more controllers of a brushless permanent magnet motor, cause the one or more controllers to perform the method according to any one of claim 1 to claim 19.
23. An appliance comprising the brushless permanent magnet rotor according to claim 20.
24. The appliance according to claim 23, wherein the appliance is a haircare appliance.