GB2517221A - Tunable antenna systems - Google Patents

Abstract

An antenna system 10 or a matching and tuning system or an antenna system design method comprises: providing an antenna 20 with at least two radiation modes with at least two respective means 30, 40 for supplying or receiving said radiation mode signals via at least two respective matching networks 50 with respective ports 60. At least one of the matching networks 50 is tunable. The antenna 20 may comprise a plurality of antenna sub-elements 100. An impedance element 110 may extend between the antenna sub-elements 100 with a connection 120 provided at the centre of the said element. The antenna 20 may be designed to provide common and differential modes or operation. Each of the matching networks 50 may be optimized to match and/or tune the antenna 20 to its corresponding radiation mode. The centre tapped impedance 110 may be an inductor or a capacitor. The antenna sub-elements 100 may be a dipole, bow-tie, helix or meandering line element in serial connection with a capacitive plate. The matching networks 50 may include MEMS tunable elements where the tunable elements may be digitally adjusted capacitors and/or inductors. Electrical switches may be used to short or shunt parts of the antenna system. The switches may be diodes and/or transistors and/or MEMS switches.

Description

TUNABLE ANTENNA SYSTEMS

Field of the invention

The invention relates to antenna systems, related matching and tuning systems part thereof, design methods for both said systems, method of operating both said systems and use thereof. The invention is for use in a wireless device, e.g. a portable wireless device, such as a laptop computer, a PDA a smartphone, a mobile phone, a tablet, and to a portable wireless device incorporating the antenna system.

General background of the invention

Mobile wireless devices with multiport antennas enable the new generation of mobile communication systems. The design of such antenna systems has proved challenging especially in terms of achieving Hermitian (multiport) match which yields high isolation and efficiency over the desired bandwidth. Many solutions, proposed in recent years, have been trying to balance the physical size constraints of the antenna with its performance. Often, an artificial trade-off between the efficiency and correlation is exploited. Majority, of the proposed in literature designs is for LTE mid-band (1710 MHz -2170 MHz) and higher bands. However, addressing LTE low-band frequency range (704 MHz -950 MHz) has proved to be particularly demanding.

Multiport antennas on small mobile platforms exhibit significant difference in radiation properties of radiation modes. Modal radiation quality factor increases (modal bandwidth decreases) with modal index. The frequency range over which all radiation modes can be matched simultaneously to the desired level of TMRL [1] is dominated by the smallest modal bandwidth. In small mobile wireless devices, such as Universal Serial Bus (USB)-dongles and various handset form factors this bandwidth may be insufficient to cover both the Uplink (UL) and the DL portion of several given band classes (BC) simultaneously. Instead of compromising on TMRL the available bandwidth of each of e.g. two radiation modes can then be exploited by matching the radiation mode with lower radiation quality factor over the UL and DL portion of the band simultaneously but the other only over the DL portion.

Assuming that the true multiport match (Hermitian match) in terms of total multiport reflectance or minimization of a subset of modal reflectances over different frequency bands by a Mode Matching Network (MMN) is achieved the maximum attainable bandwidth of a multiport antenna is limited by the weakest radiation mode (radiation mode with the highest radiation quality factor). Any antenna solution that exploits a superposition of radiation modes is therefore constrained in terms of that bandwidth.

Specific Background of the invention

Until now only one solution that has a potential of realizing tunable DL-MIMO antenna system for mobile platforms has been proposed. It exploits separate monopole and dipole (Fig. 6a) [1], where each radiation mode is mapped directly to particular antenna port. The solution is space consuming as it requires antennas to be placed on the opposite short edges of a mobile device's chassis [2], [3].

Such a placement is clearly not possible for e.g. Universal Serial Bus (USB)-dongles and may be prohibitive for other mobile platforms. However, the advantage of this approach is direct separation and independent low-loss tuning of both radiation modes. Tuning range of the solution shown in Fig. 6b is limited by the fact that the common mode equivalent circuit and the differential mode equivalent circuit of the MMN share the same elements [4]. Therefore, both networks are mutually dependent which limits the tuning capability of the MMN.

[1] W. L. Schroeder and A. Krewski, "Total multiport return loss as a figure of merit for MIMO antenna systems," in European Microwave Conference (EuMC), Paris, Sep. 2010, pp. 1742-1745.

[2] S. Yoon and N. G. Alexopoulos, "Multiple antenna high isolation apparatus and application thereof," U.S. Patent 2010/022 022A1, Sep. 2, 2010.

[3] J. llvonen, 0. Kivekaes, A. Azremi, R. Valkonen, J. Holopainen, and P. Vainikainen, "Isolation improvement method for mobile terminal antennas at lower UHF band," in Proceeding of the 5th European Conference on Antennas and Propagation (EuCAP), Apr. 2011, pp. 1307-1311.

[4] 7. H. Hu, P. 5. Hall, P. Gardner, and V. Nechayev, "Wide tunable balanced antenna for mobile terminals and its potential for MIMO applications," in Loughborough Antennas and Propagation conference (LAPc), Nov. 2011.

[5] A. Krewski, W. L. Schroeder, and K. Solbach, "Matching network synthesis for mobile MIMO antennas based on minimization of the total multi-port reflectance," in Loughborough Antennas and Propagation Conference (LAPC), Nov. 2011.

[6] A. Krewski and W. L. Schroeder, "N-port DL-MIMO antenna system realization using systematically designed mode matching and mode decomposition network," in European Microwave Conference (EuMC), Amsterdam, Nov. 2012.

[7] J. Volakis, C-C. Chen, and K. Fujimoto, Small Antennas: Miniaturization Techniques and Applications. Mcgraw-Hill Professional.

[8] SEMCAD X ver. 14.6.1 Aletsch, Schmid & Partner Engineering AG, Zurich, Switzerland, 2004-2011.

[9] C. Volmer and i. Weber and R. Stephan and K. Blau and M.A. Hem. "An Eigen-Analysis of Compact Antenna Arrays and Its Application to Port Decoupling" IEEE Transactions on Antennas and Propagation, 2008 volume 56 pages 360--370 number 2 February [10] Shameem Kabir Chaudhury and Werner L. Schroeder and Heinz J. Chaloupka, "Multiple Antenna Concept Based on Characteristic Modes of Mobile Phone Chassis", The Second European Conference on Antennas and Propagation 2007, Edinburgh, November [11]

Summary of the invention

The invention provides for an antenna system, suitable for use in DL-MIMO, comprising: (1) an antenna element, capable of handling (generating, receiving) at least N>1 radiation modes and provided with at least M>1 means for outputting or inputting, preferably each one and only one of, said radiation modes; (2) at least 7>1 matching networks, each one of said matching networks being connected to one and only one of said means for outputting or inputting radiation modes; and at least K>1 ports, each being connect to one and only one of said matching networks, wherein at least one of said matching networks being tunable.

The invention is especially suited for electrically small antenna's (given the differential mode features radiation properties thereof make it is difficult to match it over the desired bandwidth).

In a preferred embodiment the antenna system has one or more antenna sub elements comprises switches, which for the same reason as above enable to cover wider bandwidth since it is necessary to employ switching in the antenna physical structure so as to minimize losses and be able to cover sufficient range of frequencies.

In a more preferred embodiment where switching alone may be insufficient to cover the desired frequency bands due to e.g. manufacturing inaccuracies of the antenna and other reasons, further the provided tuning (in the matching networks) can be used to rectify this problem and fine tune the antenna to the desired frequency bands.

In an even more preferred embodiment both operations i.e. mode decomposition and tuning are performed within a Mode Decomposition Network (MDN). The fact that MDN allows for low-loss tuning of both radiation modes independently is one of the aspects of the present solution which make it viable for use in small mobile wireless devices.

From the above it is clear that a key aspect to be addressed in the design of a DL-or UL-MIMO multiport-antenna system is decomposition of radiation modes which results in mapping of the modes to separate external ports, especially in combination electrical tuning of said MIMO multiport-antennas. Further in order to circumvent the problem of insufficient bandwidth as described above an antenna system that makes optimum use of radiation modes can be used. Obtaining radiation modes from their superpositions is the task of a Mode Decomposition Network (MDN) as discussed in detail in the detailed description of the invention.

The different aspects of the invention are described by the claims. The dependent claims 2-17 can obviously be combined with independent claim 28.

The invention further provides for run-time methods for performing the action of switching and/or tuning, the methods comprising the steps of receiving input of the transmit/receive environment and the particular demand of the application running on the device using the wireless transmission, supported by the antenna system in accordance with the invention, and steps of determining the preferred setting of said switches and/or tuneable elements; and further generating the necessary signals to actually set said switches and/or tuneable elements in this preferred setting. The invention further provides a computer program product, operable on a processing engine, for executing any of those methods. The invention further provides non-transitory machine readable storage medium storing the computer program products just described.

The invention further provides for design time methods for designing antenna systems in accordance with the invention, more in particular for determining the position and/or characteristics of said switches and/or range of value for and/or a discrete set of value to be provided by said tuneable elements, the methods comprising the steps of receiving input of the transmit/receive environments in which the antenna system is supposed to operated and the particular demand of the applications (such as standards to be supported) running on the device using the wireless transmission, supported by the antenna system in accordance with the invention and steps of determining positions, characteristics; range and/or values based thereon. The invention further provides a computer program product, operable on a processing engine, for executing any of those methods. The invention further provides non-transitory machine readable storage medium storing the computer program products just described. Part of the design time methods or approximations thereof can be re-used in said run-time methods.

It is especially notable that the selection of the antenna system architecture enables a systematic approach for said design-or run-time methods, in particular the possibility to decouple the various aspects of such design, and providing the possibility to handle the design in a particular order and hence handling smaller problems (which in itself allows re-use as in run-time mode) while still ensuring that overall design objectives can be reached. In the description the order to perform the design and various equations to be used in the methods are given.

Brief description of the Annexes:

Annex 1 (pages 30-33) Krewski A. et al. "2-port DL-MIMO Antenna Design for LTE-enabled USB-dongles" Annex 2 (page 34-39) Krewski A. and Schroeder W. "Tunable Mode Decomposition Network for 2-port MIMO Antennas"

Brief description of the drawings

Figure 1 gives a schematic of the invented antenna system as illustrated for a 2 port case. The dashed line in each of the drawings shows the boundaries of the matching and tuning system.

Figure 2 gives a re-organized schematic of the invented antenna system as illustrated for a 2 port case. Figure 3 gives a re-organized schematic of the invented antenna system as illustrated for a 3 port case.

Figure 4 gives another embodiment for the 2-port case with common mode suppression element.

Figure 5 gives another embodiment for the 3-port case with common mode suppression element.

Figure 6 provides a schematic representation of the DL-MIMO antenna system realizations (a) described in literature [2], [3] and patented [1] solution that features physically separate monopole and dipole antennas and separate individual MNs for each antenna (b) with MMN for simultaneous match of both radiation modes and subsequent MDN (hybrid) to map these modes to external ports [5].

Figure 7 demonstrates center-tapped impedance 242. A and denote the differential mode and the common mode ports, respectively. Two elements of the physical structure of the antenna are connected to square pins (note that this is just simplified graphical representation).

Figure 8 illustrates center-tapped capacitance C1J2. A and denote the differential mode and the common mode ports, respectively. Two elements of the physical structure of the antenna are connected to square pins (note that this is just simplified graphical representation).

Figure 9 demonstrates center-tapped inductance 2L12. A and denote the differential mode and the common mode ports, respectively. Two elements of the physical structure of the antenna are connected to square pins (note that this is just simplified graphical representation).

Figure 10 provides a schematic representation of the proposed 2-port DL-MIMO antenna for USB-dongles with center-tapped inductance 2L12 and a balun transformer. A and denote the differential mode and the common mode ports, respectively. Antenna physical structure is a helix with capacitive plates at its ends.

Figure 11 provides a model of the 2-port BC 17 DL-MIMO antenna for USB-dongles [7].

Figure 12 shows simulated per port return loss and isolation of the antenna system after Figure 11.

Figure 13 a and b provide schematic representations of embodiments of the switchable 2-port DL-MIMO antenna. Four coarse switching states are possible.

Figure 14 shows the differential mode return loss (RLA) of the switchable antenna system after Fig. 13 without Matching Module. Legend indicates states of PIN diodes D2 and D2 (0 denotes open state, 1 denotes conducting state).

Figure iSa Tunable matching module for a geometrically symmetric 2-port DL-MIMO antenna comprising separate LMN and AMN. The tunable MDN comprises center-tapped inductance L12 which for ESA is a part of an antenna physical structure. The value of L12 depends on the antenna design. For a geometrically symmetric N-port antenna the tunable MN has the same fl-type topology.

Figure lSb For the common mode path the LMN tunable matching network has the H-type topology.

For the differential path the AMN tunable matching network has either the L-type or T-type topology.

Figure 16 shows the range of the EMN impedance matching capability. Blue dots and red squares and indicate values of the common mode impedance of the antenna that can be matched by LMN to the load impedance with higher than 10 dB RL at 704 MHz and 960 MHz, respectively. Solid color lines denote common mode impedance (4) of the antenna modified by ground capacitance of the AMN over the whole LTE low-band frequency range for different switching states of the differential mode shown in Fig. 14.

Figure 17 shows the range of the AMN impedance matching capability. Blue dots and red squares indicate values of the differential mode impedance of the antenna that can be matched by AMN to the load impedance with higher than 10 dB RL at 741 MHz and 925 MHz, respectively. Solid color lines denote matched impedance of the differential mode (Zd) of the antenna in range ±20 MHz around the resonance frequency after Fig. 14.

Figure 18 illustrates the differential mode return loss (RL) of the switchable antenna system after Fig. 13 with Matching Module comprising independently tuned SMN and MN. In comparison to Fig. 14 the RLA response is tuned to the center of DL portion of particular BC in LIE low-band.

Figure 19 provides a model of the 2-port switchable antenna for BC 17 DL-MIMO on USB-dongles [7].

Figure 20 shows a meandering antenna embodiment in accordance with the invention.

Figure 21 provides a schematic representation of the optimum sequence of matching and mode decomposition for an N-port MIMO antenna.

Figure 22 provides a schematic representation of MMN and MDN for a circular symmetric 4-port antenna.

Figure 23 provides a schematic representation of MDN for a symmetric 3-port antenna. Upper hybrid ("H") has unequal power division a.

Figure 24 provides a schematic representation of tunable MDN for a geometrically symmetric 4-port MIMO antenna, briefly denoted tunable MDN4p.

Figure 25 provides a schematic representation of tunable MON for a geometrically symmetric 3-port MIMO antenna, briefly denoted tunable MDN3p.

Figure 26 provides a chassis structure with 4 capacitive couplers at 4 corners.

Figure 27 provides an illustration of the common mode (red curve) and the differential mode (blue curve) impedances of a geometrically symmetric 2-port DL-MIMO antenna for USB-dongles.

Figures 28 to Figure 33 shows the Smith Charts realized with the invention. Figures represent range of impedances which can be matched to load impedance (i.e. 50 Ohms) with higher than 50 % matching efficiency at a given frequency. FREQUENCY is given in MHz. Figures 28 to 31 shows the common mode respectively for 704, 960, 1710 and 2200 for the common mode while Figure 32 and Figure 33 are for 741 and 945 MHz respectively for the differential mode.

Detailed description of the invention

Introduction to Mode Decomposition Network (MDN)

Separation in function and the optimum sequence of MMN and MDN allows for low-loss realization of either a DL-MIMO or UL-MIMO in general multiport-antenna system. Many of the techniques of multiport match and mode decomposition reported so far do not provide much flexibility for antenna design and some of them are case-specific, such as the use of connecting lines. A connecting line provides an additional coupling path between two closely spaced antennas which counteracts the intrinsic coupling due to the proximity of the antennas. The MDN in this case consists of a single hybrid. Others techniques are more general. "Decoupling networks' or "matching and decoupling networks" are used by many authors. Essentially, such network consists of hybrids and additional separate 2-port matching networks which individually match each mode and are placed at the outputs of the hybrids. This technique is a special case of a general approach and is easily applicable to N-port antennas. However, since in the majority of cases the antenna's radiation modes are not yet matched) connecting hybrids to the antenna system and matching modes individually later results in high losses. This is due to the extended resonant current path over a bank of hybrids.

The invention provides an approach based on minimization of a single figure of merit defined for MIMO antenna matching in or alternatively minimization of a subset of modal reflectances in different frequency regions. Accordingly, the matching network is named Mode Matching Network (MMN). The use of an additional MDN is optional but of high practical interest as it allows exploiting the maximum attainable bandwidth of all N radiation modes. It gives means to provide multiband MIMO operation or a MIMO operation restricted e.g. to the uplink portion of a band (UL-MIMO) or the downlink portion of a band (DL-MIMO) by assigning different modes to different parts of a band.

See Fig. 21.

The TMRL, which is descriptive quantity for matching of multi-port antennas, can likewise calculated as the Frobenius norm of the N x N scattering matrix of the antenna with respect to its N physicalfeed ports or as the Frobenius norm of the diagonal modal reflectance matrix-i.e. using modal refiectances of N radiation modes.

In the transmit mode of operation of the multi-port antennas a relation between a vector of incoming wove quantities and radiated electric field exists. The relation is described by the N x 2 complex matrix function called compound pattern. Calculation of the total radiated power by a multi-port antenna leads to definition of an N x N 1-lermitian matrix called radiation matrix which represents normalized integral of the product of compound pattern over the all the whole sphere. Since radiation matrix is Hermitian there is a unique unitary matrix V which diagonalizes this matrix. The eigen values are called modal efficiencies and the diagonal matrix is called modal efftciency matrix C. The columns of V, that is, the eigenvectors are referred to as radiation modes. The associated patterns are mutually orthogonal. The columns of V, that is right singular vectors V (n=1...N) of the scattering matrix are referred to as network modes. Furthermore, V in its columns contains unique eigenvectors V,1 (n=1...N) for radiation matrix decomposition, which are at the same time the right singular vectors of S. The same set of eigenvectors (columns of V) diagonalizes both the matrices R" scattering matrix SA, which together fully characterize a lossless N-port MIMO antenna system.

Therefore radiation modes correspond directly to network modes.

In the idealization of a lossless N-port antenna, the radiation modes coincide with the network modes, i.e. with the eigenvectors of the product of the Hermitian transpose of the scattering matrix with the scattering matrix itself. The latter statement typically holds as a good approximation for real N-port antennas. Generalization to lossy antennas is straightforward by adding a loss matrix term to the power balance relation. For simplicity we consider only the lossless case here. The eigenvectors of RA (radiation modes) then correspond directly to the right singular vectors of SA (network modes).

The above reveals that it is possible to separate multiport antenna matching from mode decomposition.

In an aspect of the present invention the multiport antenna matching is separated from mode decomposition with MMN as a first stage and MDN as a second stage network which is essential for low-loss design. An MMN is placed directly behind the antenna while an MDN should only be applied to a matched antenna, with modal impedances adjusted to the reference impedance, for a low-loss design. The task of matching network is to match all N modal reflectances of the antenna without changing radiation modes, hence this network is called Mode Matching Network (MMN).

In order to satisfy this condition the scattering matrix of the matched antenna S (See Fig. 22) has to be diagonalizable by the same unitary transform V as SA. The decomposition then leads to the diagonal modal reflectance matrix [P = VTSPV. The MMN design procedure by minimizing s since V is unitary, will result in an Hermitian match between the antenna and the matching network. The norm of the modal reflectance matrix is a metric for mismatch with respect to Hermitian match and as shown a suitable MIMO antenna figure of merit is the total multiport reflectance given below.

l:ot IITII yields Multiport match is achieved for 1ot 0. The properties of radiation modes do not change practically over frequency which is not only convenient from a design point of view but also ensures that basic properties of an antenna such as bandwidth and efficiency do not change significantly over frequency and are only constrained by the applied volume for an antenna.

The design of MMN is essential for proper function of a multiport antenna as it allows matching all N network modes of the antenna. However, it has to be understood that at each output of the MMN a particular linearly independent superposition of all N radiation modes is apparent. The modes themselves are not yet directly accessible at the MMN outputs. But the MMN design procedure by minimizing [j over the requested bandwidth and over all radiation modes causes the overall bandwidth being limited by the radiation modes with high radiation quality factor (or inversely proportional limited bandwidth). Therefore in order to exploit the maximum attainable bandwidth of each of N radiation modes a further Mode Decomposition Network (MDN) can be used. As the name indicates it maps the N radiation modes of the multiport antenna to N output ports.

MDIV construction In general, the scattering matrix of an MDN is given by JO 0 1 1 SM_1° V'i(O @ 1 -1 1 vT o)vri 1 0 0 \i -1 0 0 The matrix V can be approximately factored in terms of complex Givens rotations in the form VVlV2...VL where L «= N(N-1)/2. Each V, describes 2N-port network with N-2 through connections (See e.g. Fig. 22). The complete MDN is systematically constructed, when assuming the matrix of eigenvectors V real and almost frequency independent, by cascading L 4-port networks each representing hybrid with a particular power division and phase shift. The mode decomposition is provided at the reference impedance level. (Mostly 500). See also Fig. 23.

In an embodiment of the present invention one uses the matrix V obtained from (1) for the MDN synthesis which reduces the overall network complexity and losses since this network serves only the purpose of mode decomposition at the reference impedance level. In many cases the frequency dependence of the radiation modes, as described by the matrix of eigenvectors V is small over the frequency range of interest. For antenna structures with certain symmetry planes, where the radiation modes are governed by the symmetries of the structure, the eigenvectors are often frequency independent and real. In these practically significant cases the MMN feature the same symmetry planes as the antenna structure and the MDN can be constructed from 180 deg hybrids (with equal or unequal power division). The MDN design of Fig. 22 is obtained from factorization of the matrix V (2). It consists of four 180 deg hybrids with equal power division labeled by "H". Column vectors on the right and left side indicate excitations and their responses.

/___*_ oo\/100 0\/i \/100 0 (vT (oio o\(v° p 0\ft L 1 1 n i n in C C ---0 0 1 1 U 1 U U VV1V2V3V4 /2 /2 0 0 - 2 V2 --- 0 io)I, 1)o \o o o iJ \0 0 \o 0 0 1/ \ (2) Introduction to the concept DL-or UL-MIMO (only MIMO operation in a number of subbands) As we can map now the N radiation modes of the multiport antenna to N output ports, we can optimize each and every radiation mode over pre-defined portions of spectrum. In other words it exploits the maximum attainable bandwidth of each radiation mode, even radiation modes with higher radiation quality factor (or inversely proportional to limited bandwidth). It is the Mode Matching Network (MMN) that will select and group radiation modes and will maximize performance in term of efficiency each selection and grouping over chosen portions of spectrum. The MMN is placed between the internal ports of the antenna and the MDN. The next table can help to design the MMN circuitry. Suppose an antenna system supporting N orthogonal radiation modes and comprising N feed ports for connecting to transmit and/or receive circuitry of the terminal (where N»=2). A first subset of the feed ports (1 & 3) is assigned to a DL-1 and a second subset of the feed ports (1 &4) is assigned to an UL-1. MDN Circuitry is arranged to map the N radiation modes to the N feed ports. See

table below.

aci.t',, ci j N I J fif S tag3 the ntenra Lystem s S D..-V3MD t -1 i tarne rernrk for UI-portIon 4,..:.,/.H:" 1 .., /f K. t TDL eat1I.n

I

t. / I I,C)fUI. 33< . j J. ;i, U' \ . .. l. . t: 4.. . . .. ., . ...

4' ,r,c c *; 4' Table 1. Overview N radiation modes supporting M subbands. Radiation modes aren't ranked according their modal index.

Description of some aspects of the invention

FINE TUNING

The invention fits in the spirit of a DL-or UL-MIMO multiport antenna system design based on decomposition of radiation modes which results in mapping of the modes to separate external ports combined with obtaining radiation modes from their superpositions by use of a Mode Decomposition Network (MDN) approach as discussed above.

A very basic conversion from 180 deg hybrid to two in-phase lines and a balun is known from the prior art. It realizes an MDN as 180 deg hybrid with S parameters given by (1) and maps radiation modes to external ports over a wide fractional bandwidth. However, it does not allow for impedance tuning as mode decomposition is performed all at the reference impedance level.

However the invention shown here relies on a variable impedance level over the whole circuit and thus tunes over a broad tuning range of interest.

The invention replaces each single hybrid in the factored and cascaded MDN as defined previously with essentially a tunable 180 deg hybrid ("H") and thus it enables the tuning of radiation modes. The MMN with the scattering matrix SM as depictured in Fig. 22 is omitted.

The tuning is achieved by attaching to each Givens rotation (180 deg hybrid) two separate and independent mode Matching Networks (MN), forming jointly a tunable "H". All the "H" form collectively the tunable MDN. This allows realizing the tuning of all N radiation modes independently providing much wider tuning range then other solutions in the field. As a way of example for the N=3 and N=4 radiation modes case the hybrids in (Fig. 22 and Fig. 23) are exchanged with essentially tunable 180 deg hybrids and thus enable tuning of radiation modes. The results are shown in Fig. 24 and Fig. 25 respectively. By induction the N=5 and more radiation modes cases of tunable MDN can be designed accordingly. Color/Letter coded column vectors on the right hand side of the schematic indicate excitations and on the left hand side responses (radiation modes) to those excitations. "H" denotes hybrid. By which MN (3 in total) and LMN (also 3 in total) networks in particular the 4 radiation modes are tuned can be found in the next enumeration (See Fig. 24).

1. RED column vector: 1000 which corresponds to radiation mode i-+++. All three MN are to be used to tune this mode.

2. BLUE column vector 0100 corresponds to radiation mode ++--. In order to tune it two MN networks in the first state of the circuit (on the left) have to be used together with one MN from the second state of the circuit.

3. Trivially the other radiation modes (GREEN and BLACK) will be tuned by specific EMN and AMN networks.

In the tunable MDN4p case one "H" has not been replaced because the outputs from the previous two hybrids in the first stage of the network (on the left) are already matched to the reference impedance level. Therefore there is no reason to replace the subsequent hybrid with a tunable hybrid. The solution in this invention is) in principle, antenna structure independent) low-loss and can be scaled to the desired frequency bands. The antenna independency entails that the fine tuning can be performed within an MDN outside an antenna on the remote printed circuit board (PCB). This allows flexibility in the design of an antenna. This approach is also considered attractive, because it allows separating the design in optimization of the antenna structure itself and of the tunable MON.

So each tunable hybrid "H" features a 180 deg hybrid materialized by a center-tapped inductance. It maps all radiation modes to external ports 1,2,3,4....N via separate and independent matching networks MN and LMN. These matching networks are called differential and common as each follows its own differential, resp. common branch. The branches of AMN and EMN are not mutually constrained as in the case of a multiport matching network MMN. This yields significantly improved tuning range over the case of tunable multiport matching network. Tuning range in the present solution is governed mostly by the radiation properties of the radiation modes. Balun transformer or a common mode choke can be used to suppress the common mode signal in the differential mode branch.

In the case of an antenna like the monocoque laptop (PCT/EP2012/070976) the electronically tunable MDN could be placed on a separate PCB located inside the laptop. (See Fig. isa if for the 2-port antenna) Design details: matching network in a N-radiation mode antenna system tunable LMN All the tunable XMN have a fl-type topology which consists of two shunts and a single series LC tank (See Fig. iSb, 24 and 25). It is well known that this basic network topology brings about the maximum possible tuning range which covers nearly the entire Smith chart. Three inductors LI, L and LI have fixed values while capacitors Cf, Cand G are tunable. The matching network is also used for transmit mode of operation so that only highly linear tunable Radio Frequency -Micro Electro-Mechanical System (RF-MEMS) capacitors or Complementary Metal-Oxide-Semiconductor (CMOS) Digitally Tunable Capacitors (DTC5) can be considered in a design. This tunable MN is characterized

and denoted in this description by its 3 DTC's.

tunable iMN All the tunable AMN have a balanced T-type topology because all radiation modes need the largest tuning range possible, as it is unknown in the slightest what the range must be. Corresponding series tanks in the T-type matching network have to feature the same values for geometrically symmetric antennas. Tunable Capacitors (DTC5) can be considered in a design. This tunable MN is characterized and denoted in this description by its 5 DTC's, but out of which 3 are chosen independently.

In addition, in order to suppress the common mode in the differential mode branch of each tunable hybrid "H" a common mode choke has to be used.

The fact that both tunable matching networks AMN and EMN have each their identical topology in an N-radiation mode case means that the layout of the circuitry on the PCB's which fit the matching networks can be standardized and simplified.

Fine Tuning of the 2-port antenna Additional adaptive fine tuning can be performed within an MDN placed outside an antenna on the printed circuit board (PCB). This approach is also considered mandatory from a design point of view, because it allows separating the design and optimization of the antenna structure itself and of the MDN. The idea leads to the definition of a soft specification of the interface between antenna and MDN. The antenna will be required only to be switched to an extent which assures that modal impedances stay within a specified region of the Smith chart. The MDN with additional (adaptive) tuning capability will be required only to map this region of the Smith chart to the desired feed-port impedance.

It is a fact that any well matched electrically small dipole (helix, meander line or any other physical realization) features impedance change over frequency (common and differential mode) similar to that given in Fig. 27. For the common mode the H-type topology is chosen as in the N-port antenna, the EMN matching network. (The impedance change of a matched 2-port antenna on the Smith chart is given in more detail in Annex).

For the differential mode the L-type matching network topology is most ideal for tuning, the AMN matching network. Tuning range of the AMN network is limited in comparison to EMN due to assumed [-type topology. However, it is not a constraint as this network should only allow matching the differential mode within no more than ±3 % of its resonance frequency for a low-loss design. T-type is also appropriate not in the least if the differential mode features similar radiation properties as the common mode. However, since the impedance of a matched electrically small dipole change rapidly over the desired portion of spectrum as given in Fig. 27, it is logical to choose L-type network instead of T-type.

The goal is always to minimize the number of components used in order to minimize losses and the cost of a solution. Contrary to the fl-type only 2 independent DTC's need to be tuned by the L-type topology. Contrary to the N-port antenna, the AMN matching network has only to fine tune one mode, the differential mode. The L-type topology is shown in Fig. 15b.

There are two sides defined at an interface: The antenna at one side and the Smith region where the differential mode impedance of the antenna lands at the other side. The EXACT map of where it should land, so that the tunable LMN and AMN can take it over from there, is shown next. Figures (see for instance the charts in 28 to 33) represent on the Smith chart the range of impedances which can be matched to load impedance (i.e. 50 Ohms) with higher than 50 % matching efficiency at a given frequency. (FREQUENCY is given in MHz). The common mode for 4 frequencies: 704, 960, 1710, 2200 MHz and for differential mode for 2 frequencies: 741, 945 MHz. The algorithm and criterion to find the fixed values of inductances of the tunable MDN are given next: 1. Simulation setup as given in Fig. 3 of the annex LAPC-2013.

2. 370 dense points are chosen on the Smith chart, evenly distributed.

3. For a particular value of two fixed inductances an optimization criterion is used which allows to obtain the optimum settings for the DTCs to match a particular modal impedance with the highest possible matching efficiency at a single frequency.

4. Use of such algorithm over all 370 points on the Smith chart yields figures which show the value of matching efficiency with which particular modal impedance can be matched to 50 Ohms at single frequency.

5. At a single frequency one the result from step 4) is available one can calculate the area on the Smith chart which can be matched with matching efficiency higher than 50 % (see figures 28 to 33 defining Smith chart allowed regions).

6. Steps 4) and 5) are repeated for all combinations of discrete values from 1.8 nH to 18 nH in the 0603 range. (There are two independent values of inductances, see subscripts and superscripts in the annex in Fig. 15a) at the lowest and the highest frequency of interest. That is 704 MHz and 2200 MHz for the common mode matching network and from 741 MHz to 945 MHz for the differential mode matching network.

7. The values of inductances which correspond to the largest "higher than 50% area on the Smith chart" (as described in step 5) common to both the lowest and the highest frequencies are taken as the optimum.

i).CTRL1 F U U I Fl P (IRI U j Table with switching states for the structures in Fig. 13a or b. Fig. 27 is derived for the structures in Fig. 13a orb for the switch positions 00 of the RF PIN diodes. It is directly deductible from the Smith chart (in Fig. 27) that the common mode can be fairly well-matched in the LTE low band. Tuning range of the differential mode is limited to immediate vicinity of resonance frequency in order to maximize efficiency of this radiation mode. The common mode and the differential mode impedances are displayed from 500 MHz to 1000 MHz and from DC to 500 MHz, respectively.

ESA

ESA: The Antenna system is Electrically Small if its maximum length is less than X/2rt. This is the (Harold) Wheeler definition of a small antenna, which was derived mathematically in 1947. Multiport antennas on small mobile platforms, like USB dongles exhibit significant difference in radiation properties of radiation modes. Modal radiation quality factor increases (modal bandwidth decreases) with modal index. Practically, the highest and some higher order index radiation modes are considered as an ESA. They feature impedance and (poor) radiation quality factor similar to that of an ESA. A mode exhibiting a quickly-changing impedance over the frequency range of importance (See blue curve in Fig. 27) is an ESA. Vice versa a particular ESA in this invention is the only, or one of these radiation modes. An ESA is defined within a given volume and at a given portion of spectrum.

An ESA can be reasonably well-matched (TMRL criterion of 7dB) within the desired band of operation after the physical introduction of the MDN in the antenna structure. It does not matter if tunable but of course tunable version of an MDN is preferred. The incorporated various (fixed or electronically tunable) &MN and EMN circuits will have to further fine tune the antenna system to the reference impedance level on all N ports.

In the case of ESA placing the (tunable) MDN remotely could result in high losses and potentially high non-linearity in the transmit mode of operation due to strong resonant current flowing all along the path to the MDN. For making it a viable solution when applying coax cables it should be taken into account1 the influence of loss of the coax cable which connects the (tunable) MDN with exciters.

It is a fact that any well matched ESA (helix, meander line, bow-tie or any other physical realization) features rapidly impedance change over frequency (one of or the only ESA radiation mode). The tuning range of the ESA is limited to immediate vicinity ± 3 % of the concerned radiation mode resonance frequency in order to maximize efficiency of this radiation mode. For higher frequency bands it may be more than ± 3%. Eventually it depends on the radiation quality factor of the ESA.

A practical embodiment for a 4-port MIMO antenna application such as outlined by article [10] can comprise one or more ESA's. The difficulty in incorporating a tunable MDN into the antenna physical structure is the physical distance between exciters. This might be an issue if the location of exciters is predefined or preemptively chosen. A carefully considered solution is to have either switching in each coupler or individual external matching networks for each coupler plus a tunable MDN as given in Fig. 24 and Fig. 25 located in the center of the PCB. The losses introduced by coax cables connecting exciters with tunable MDN should be taken into account.

The ESA 2-port antenna Here the differential mode is equivalent to an ESA for small devices as an USB dongle. USB dongles are commercially easily available today. The antenna is produced as a symmetrically geometrical antenna structure.

If the antenna is an ESA the center tapped impedance [executing the hybrid function of the tunable MDN, has to be part of the antenna physical structure so that the RF current in the ESA can flow without major interruption. In other words it means that the MDN has to be placed as close as possible to the antenna physical structure in order to minimize losses.

For an ESA 2-port antenna the MDN is achieved by a center tapped inductance. There is the minimum value of inductance L which can be realized using particular size of an SMD component.

That is e.g. 1.8 nH for 0603 size. The maximum value of inductance is constrained by its losses. The higher nominal value of inductance the higher the losses. The right value should be chosen in order to maximize total efficiency, in particular, of the ESA.

A measure to get a better ESA radiation quality factor might be by increasing the inductance of a center-tapped inductor L within the antenna structure itself in order to ease requirements for modeling of the antenna physical structure. However, it is important to remember that the series loss resistance of L has to be much smaller than the radiation resistance of unmatched ESA, within a given volume and at a given center frequency.

COARSE SWITCHING FOR AN ESA

If the ESA at hand needs to serve more than one band, coarse switching should be applied in the antenna physical structure. The hardware switch can be constructed by RF PIN diodes. Two diodes makes up 4 switch positions. Each switch position defines a subsequent and equivalent bandwidth. At each switch position the ESA should be approximately matched in the particular portion of the band.

The fine tuning is again accomplished by the tunable MDN with the various tunable AMN and EMN.

Note that all the tunable AMN have a balanced 1-type topology and all tunable LMN have a H-type topology because all radiation modes need the largest tuning range possible, as it is unknown in the slightest what the range might be. (See Fig. lSb for a fl-type topology). Both H-type and 1-type topologies serve basic matching and tuning.

The ESA 2-port antenna (Coarse switching capabilities) The frequency range over which all radiation modes can be matched simultaneously to the desired level of TMRL [1] is dominated by the smallest modal bandwidth. Useful level of TMRL is 7dB. This said bandwidth governed by differential mode (with the highest modal index) can be extended by coarse switching.

The antenna structure must be competent to switch to an extent which assures that modal impedances stay within a specified region of the Smith chart. The MDN with additional (adaptive) tuning capability must be competent on its turn to map this region of the Smith chart to the desired feed-port impedance.

It is essential to use electronic or mechanical switches for coarse switching within the antenna structure itself, at locations where the current is not at maximum to extend or reduce, respectively, the electrical length of the structure. Additional adaptive fine tuning can then be performed within an MDN placed outside an antenna on the printed circuit board (PCB). The result after fine tuning of 4 switching states by 2 RF PIN diodes (See Fig. 13a or 13b) is represented in Fig. 14.

The solid color lines in Fig. 17 show the matched impedance of the differential mode Z in the 4 different switch positions. The range of each switch position is 20 MHz around the resonance frequency after Fig. 14 and Fig. 13a or 13b.

Examples of coarse switching in different antenna structures are in Fig. 20 for a meander type and in Fig. 19, Fig. 13a or Fig. 13b for a helix type, both with 2 RF PIN diode switches for 4 switch positions governing 4 different subsequent and equivalent bandwidths.

ELECTRONIC FINE TUNING (For a 2-port Antenna) Adaptive tuning of the proposed antenna system can be employed to compensate for parameter variations in manufacturing (which are critical in view of the small bandwidth of the differential mode) and, to some extent, to compensate for detuning due to variations in the near field environment. Adaptive tuning requires a closed loop control mechanism based on suitable measurements which are processed e.g. by the DSP of the RFIC. A seemingly straightforward approach to perform measurements which allow estimating the mismatch of an antenna makes use of directional couplers to in a transmission line.

This approach, however, has fundamental limitations: For FDD systems which employ electrically small antennas, i.e. antennas with small bandwidth and therefore strong variation of the antenna input impedance with frequency, measurements at TX frequency are not in general a reliable indicator of the mismatch at RX frequency. A model based prediction of mismatch at RX frequency based on measurements at TX frequency would have to be applied. The above statement particularly applies if the DL-MIMO approach is followed, because no TX signal is available at the DL-only antenna port.

Antenna fine tuning at TX frequency based on reflected DC power measurements For the presently proposed structure it is possible to use the directional coupler I quadrature down converter approach for measurements of the common mode antenna reflectance at TX frequency. At RX frequency and specifically for the differential mode a different approach has to be selected.

We have three independent variables C, = The common mode reflection coefficient seen toward the antenna is given by F'. C, C and Ccan be found by maximizing the power transfer from the transmitter to the antenna. It is accomplished by minimization of the output reflectance Trx.

c:t =arufl(r(cc)2) Antenna fine tuning at fiX frequency based on RSSI measurements Evaluation of antenna mismatch at RX frequency is based on scalar (RSSI) measurements, assumed to be available in the digital baseband. It is required that RSSI information is made available in the UE per receive path, i.e. per antenna of the UE in order to allow for any conclusions relevant for antenna adaption. Estimation of mismatch at the common and differential mode ports of the antenna system (including MN and LMN) is based on measuring only one scalar quantity, viz. RSSl per receive path.

The receive mode has to be supported in both common mode and differential mode branches for the 2-port DL-MIMO antenna system Differential mode branch By measuring delivered power Pf for three different settings of 2 DTC's: Cf and C the differential mode impedance of antenna can be found.

Once the said impedance is known Cf and Ccan be found so as to maximize the power transfer from the antenna to the load. It is accomplished by minimization of the input reflectance by f.

CaUPt = argminOr,Rv(cd)2) Ca) The shown approach is clearly mathematically correct and its advantage is that the complex impedance of the differential mode is obtained using three measurements of the power delivered to the load. The phase measurements are not needed.

Common mode branch Analogous derivations can be written for the common mode branch. Note that in this case however we have three independent variables Cc = (Cf, C, Cf). Mutatis mutandis Cf, Cand C2can be found by maximizing the power transfer from the antenna to the load. It is accomplished by minimization of the input reflectance by = arg Weigh ting In a 2 port antenna case, as fl. small USA dongles, the common mode serves the UL-and DL-band whilst the differential mode (highest modal index) supports only the DL in a DL-MIMO scenario. The common mode gets tuned with the tunable LMN matching network. In a dynamic environment three independent DTC's need to be defined at regular time intervals. These 3 DTC's define at best effort the UL path or DL RF-path. There is the given option whether the antenna can be matched better in the DL portion or UL portion of the desired band depending on the requirements at a given time.

The weights correspond to what takes certain level of priority in the optimization algorithm which is either maximization of the transmitted power or maximization of the RSSI.

It depends on the goal which is to be achieved in this dynamic matching scheme. The weighting plays important role as it can be changed over time in order to give more priority to either DL or UL portion of a band, i.e. to match the antenna better in DL or UL portion of a band at a given time. This may be of particular importance for E-UTRA band classes where the distance in frequency between UL and DL portions of a band is high. As an example consider BC 4.

Different operating conditions: Line of Sight, shadowing effects from buildings, Power Received (RSSl) level, interference from basestations or other UE (User Equipment), etc..

4.3 Optimization critetia for adaptive matching for different operating scenarios The 2-port MiMC.: antenna system, where one of the radiation modes provides insufficient band- wiSh to cover auth L arie L. sriultaneousy can be Ut lzad n hree dilteiant operating sce-nanos: Single antenna (common mode) used for both TX and RX 2:riP - = wit I I L w r (( ) 6) where c*T and are reflectances seen from ThIN toward the antenna at center fre quency of UL and DL portion of a band, respectively w and w are weighting coefficients.

Subscript RX indicates that the tuning algorithm uses ASSI (tuning in the receive mode) while suhschpi TX denotes that the reflected power measurement is performed (tuning in the transmit mode).

Common mode used for TX only and the differential mode used for FkX oniy In this scenario both networks are tuned independently r4Opt -nP. r-cop-L rP /n.

-..r& = ii

GCT

2 port DL-MIMC) (common mode used br both TX and RX and the differential mode used only for r:1x) Ct = + (8) {C} where w and W2 are weighting coefficients, as in single antenna mode, and (9) {c} Description of exemplary embodiments of the invention The invention relates to N port (for instance 2 port) antenna systems, which are electrically switchable and/or tunable, suitable for instance as downlink MIMO antenna, but not limited thereto.

In an embodiment of the invention realization of a switchable and tunable geometrically symmetric 2-port DL-MIMD antenna system for small wireless devices is presented. In a first aspect thereof a fixed frequency geo-rnetrically symmetric 2-port DL-MIMO antenna for LTE low-band BC-17 is reported.

Switching between different frequency bands of LIE low-band is introduced. Furthermore in a second aspect matching module or system for electrical tuning of the antenna is described. Ihe module is of particular practical interest for tuning of matched antenna solutions that realize DL-MIMO concept for a small mobile wire-less devices such as USB-dongles. However possible range of applications where the present solution can be employed is much broader. For instance the matching module can be used for a tuning of a 2-port MIMO antenna for laptops.

The present invention has certain advantages over previously described solutions. The physical structure of the antenna is a single element which reduces space necessary for its placement is a mobile wireless device. Moreover, due to the fact that both radiation modes are mapped to external ports directly at the antenna structure (comprising center-tapped impedance (Fig. 7), center-tapped capacitance Fig. 8 or center-tapped inductance Fig. 9 depending on the differential mode impedance of the antenna), each radiation mode can be tuned independently by a separate mode-specific matching network. Note that both matching networks are an integral part of an MDN. Therefore, unlike in a classic solution of mode decomposition our tunable MDN allows not only to map radiation modes to external port but also tune radiation modes independently with low losses. This allows to realize tuning of both radiation modes independently providing much wider tuning range than the prior art solution (see Fig. Sb). If the differential mode of a geometrically symmetric 2-port MIMO antenna features impedance similar to that of an Electrically Small Antenna (ESA) then operation of the antenna has to be restricted to DL-Multiple Input -Multiple Output (MIMO). The physical structure of the antenna has to be modeled so that the differential mode is reasonably well-matched within the desired band of operation. Tuning range of this radiation mode is limited to immediate vicinity of the differential mode resonance frequency in order to maximize efficiency of this radiation mode. Physical structure of the antenna (radiator) was selected in order to achieve maximum attain-able bandwidth and efficiency of the differential mode within a given volume. According to [6] the optimum physical structure for the ESA comprises helix with capacitive plates at its ends. The choice of the optimum radiator enables the realization of switchable antenna by electrical shorting of the windings of the helix. Several realizations of the switching concept are feasible. Switching can be divided into two different classes i.e. mechanical switching and electrical switching. With electrical switching several different components can be used e.g. PIN diode, FET transistor and MEMS SPST.

Theoretically the best performance in terms of achieving high efficiency and RF-Direct Current (DC) isolation can be accomplished with MEMS SPST switches. However, this technology is still expensive and not sufficiently mature (the components can't be used off-the-shelf as they require specific handling) to be used in commercial low-cost wireless communication applications. The use of the RF PIN diodes is the most feasible with respect to cost, design and handling. The problem in this case is biasing. Achieving sufficient RF-DC isolation is challenging. Nevertheless, this approach is considered to be the most practical solution.

Generally speaking the following combinable features of the invention can be noted. (1) The use of a center-tapped inductance as it allows mapping both radiation modes of a geometrically symmetric antenna to external ports. (2) Use of a single antenna structure for both transmit and receive MO.

with two ports. (3) Selection of an optimal physical structure of the antenna (helix loaded with capacitive plates) (for given size constraints relative the wavelength) with respect to maximum attainable efficiency and bandwidth. (4) Capability to be used for DL-MIMO system when one of the modes features behavior similar to that of an ESA. (5) Separation of switching and tuning (in order to maximize efficiency). Switching is preferably used in the physical structure of the antenna while tuning network is placed on the printed circuit board (PCB). (6) Realization of the switching between different frequency bands by electrical or mechanical shorting of the windings of a helix. (7) Tuning of both radiation modes is realized separately and independently for each of them.

The following three sections are dedicated to fixed frequency geometrically symmetric antenna solution for BC 17 DL-MIMO, Extension of this antenna toward switching between BC 17, BC 20, BC S and BC 8 and finally a description of the matching module for electrical tuning of the antenna.

Fixed Frequency Antenna Schematic drawing of the fixed frequency antenna solution with optimum radiator (i.e. optimum for the differential mode performance is its impedance is similar to that of an electrically small antenna) is given in Fig. 10. It comprises center-tapped inductance (see Fig. 9), separate matching networks for each radiation mode and balun transformer to suppress the common mode in the differential mode branch. Physical structure of the antenna is a helix with two capacitive plates at its ends. Fig. 11 shows physical structure of the 2-port DL-MIMO antenna for BC 17 after schematic representation given in Fig. 10. It comprises a 0.8 mm diameter silver wire helix. At both ends of each helix a capacitive plate is attached. As stated before this design constitutes optimum radiator given the space available for the antenna placement (note that the space for the antenna was increased to 60 mm which is 0.15A at 741 MHz). MN comprises two shunt capacitances C12 = 1 pF and a series inductance 18 nH. MN features two series inductances L13 = 3.9 nH. The center-tapped inductance 2L12 = 7.8 nH. All elements are to be realized using 0603 Surface Mount Device (EMD) components from Jonansontechnology. The common mode choke used in the simulations and optimization process is TCN1-10 from Minicircuits. Simulated performance of the antenna which includes all above elements is given in Fig. 12. The return loss is higher than 10 dB for both UL and DL portions of BC 17 while the isolation is higher than 17 dB. The present solution is applicable to other frequency bands provided that the physical structure of the antenna is chosen accordingly. For instance in the LIE mid-band frequency range and BC 7 it would be more appropriate to use bow-tie antenna elements or planar meander line structure instead of a helix.

Switchable Antenna As indicated earlier the 2-port DL MIMO antenna shown in Fig. 10 can be extended to switching and further to tuning. Fig. 13 shows schematic of the present solution for switching of the differential mode of the antenna between different DL portions of the LTE low-band band classs (BCs). The solution features minimum number of the SMD components and control lines. Two PIN diode (PIN) diodes (BAR 90 from Infineon) per each arm of a helix antenna allow to realize four switching states (see IABLE I). Ihe diode was modeled (using information from the datasheet) as parallel RC tank with R = 5 kO, C = 0.2 pF and as series RL (R = 1.3 0, L = 0.4 nH) in the open state and conducting state, respectively.

D1, CTRL 1 D2JCTRL2 1 DLofBC 117 0 Table 1: Truth table for differential mode switching of 2-port DL-MIMO antenna after Fig. 13.0 and 1 denote reverse bias voltage applied to diodes and forward bias current, respectively.

The RF power handling of the PIN diode is constrained by either the diode's breakdown voltage or its power dissipation capability. For example, for the series connection and a diode with a low and high resistance state of 1 0 and 10 kG, respectively, approximately 2% (CW Power Multiplier = 50) of the input power dissipates in the diode in either the "ON" or the "OFF' state. This means that the diode with drss = 250 mW can handle 12.5 W of input power. As a rule of thumb PIN diodes should be operated at frequencies considerably higher than the reciprocal of T. This avoids non-linear intermodulation and harmonics effects by preventing modulation of the charge carrier concentration.

ForT around 500 ns to 1000 ns the minimum operating frequency is between 20 MHz to 10 MHz and of no concern for our intended application.

Ceramic chip 5MB inductors 0603HP (1608) inductors from CoilCraft are used as RF chokes L. These choke inductors have to be distributed over the length of the biasing lines. The main rule during the optimization process is to place few chokes (e.g. two or three) near the diode and single choke near the RF ground. The DC block capacitors Care 2 x 82 pF. Simulation results for all four states of the PIN diodes are shown in Fig. 14. Small deviation from the desired DL portion of particular BC can be resolved by final optimization of the design. In practical implementation the differential mode response can be tuned within about ±20 MHz from its center frequency using tunable MN which is the subject of the next section.

Tuning Module The differential mode of a 2-port MIMO ESA features narrow bandwidth (by design it has to be sufficient for a DL-MIMO antenna system realization) and low efficiency. In order to minimize losses switching of the differential mode between different DLs of BCs of LIE low-band has to be introduced in the antenna physical structure. Common mode is related to coupling between the antenna and the chassis of a mobile device. The attainable bandwidth of this mode is usually sufficient to cover both UL and DL portions of a particular band. Fig. 15a shows the optimum solution for matching and tuning of a geometrically symmetric 2-port MIMO antenna. Ihe center-tapped inductance is used to obtain radiation modes directly in the physical structure of the antenna by exploiting geometrical symmetry..

This means that the topologies of the MN and AMN are not mutually constrained. luning range is significantly improved over the case of the MMN (Fig. 6b). In reality due to ground capacitance of the MN the common mode impedance of the antenna is modified, however it is easy to account for this effect. The solution can be applied to a geometrically symmetric 2-port MIMO antennas but is of significant importance when the differential mode impedances is similar to that of an ESA. This scenario is typical for many mobile platforms, especially USB-dongles.

Common mode matching network (MN) Tunable MN is a it-type matching network that consists of two shunt and single series LC components. Assuming inductance values and typical range of the tunable capacitances it is possible to obtain the range of the common mode impedances that can be matched with this MN (Fig. 16).

Common mode impedance of the antenna is within 1MN matching capability in the whole LTE low-band.

d d c c element i L1 2 3 C value 3.9 3.9 6.8 5.8 6.8 From 0.7 to 4.6 Table 2: Element values (given in pF and nH) of the tuning module after Fig. 15a. Value of a center-tapped inductance L depends on the antenna design.

Differential mode matching network (AMN) Tunable AMN is a L-type matching network that consists of series LC and shunt LC components. In addition in order to suppress common mode in the differential mode branch a balun transformer is used. Isolation between external ports S and A depends on the Common Mode Rejection Ratio (CMRR) of a balun transformer. Tunability of this network is limited in comparison to EMN (Fig. 17).

It is within ±20 MHz around center frequency for a particular switching state of the antenna. The differential mode return loss for four different switching states of the PIN diodes incorporated into the antenna structure and tuned to the center frequency of DL portion of each BC in the LTE low-band is shown in Fig. 18.

Meandering antenna embodiment Recall that due to bandwidth limitations of the differential mode a switchable and tunable antenna system is required in order to realize (at least) a DL-MIMO antenna system. An approach which is based on rnodularization is provided in the invention. Fig. 20 provides a schematic representation of the proposed antenna structure for LIE low-band using 4 PIN diodes. To keep losses low coarse switching between different LIE bands is done in the antenna structure itself while fine tuning (to compensate a small frequency deviation from assigned BC, e.g. 10 MHz -20 MHz is accomplished by means of an external AMN. The idea is to separate antenna and front-end tuning module using a common soft interface specification. Single tuning module can be used for many different antenna designs provided that the antenna impedance levels comply with the tuning network requirements.

Abbreviations BC band class CMRR Common Mode Rejection Ratio DC Direct Current DL Downlink ESA Electrically Small Antenna

FET Field-effect transistor

LTE Long Term Evolution MDN Mode Decomposition Network MEMS Micro Electro Mechanical System MIMO Multiple Input -Multiple Output MMN Mode Matching Network MN Matching Network PCB printed circuit board RF radio frequency RI Return Loss SMD Surface Mount Device UL Uplink USB Universal Serial Bus PIN PIN diode

Claims (33)

1. Claims 1. An antenna system (10) (wherein each radiation mode is directly mapped to a port), comprising: (1) an antenna element (20), capable of handling (generating, receiving) at least N>1 radiation modes and provided with at least M>1 means (30) (40) for outputting or inputting each one and only one of said radiation modes; (2) at least Z>1 matching networks (5), each one of said matching networks being connected to one and only one of said means for outputting or inputting radiation modes; and (3) at least K>1 ports (60) , each being connect to one and only one of said matching networks, wherein at least one of said matching networks being tunable.
2. 2. The antenna system of claim 1, wherein said antenna element (20), comprises a plurality of antenna sub elements (100) and one or more impedances (110), each of said impedances being provided with a connection (120) in its center, part of said impedances being provided between two of said sub elements, whereby one or more of said means for outputting or inputting (40) being provided by said connection (120) (to thereby represent a first signal referenced to ground), while one or more of the other of said means for outputting or inputting (30) being provided by two connection points (130) with said two antenna sub elements (to thereby represent a second signal therein between).
3. 3. The antenna system of claim 1 or 2, wherein said antenna element, comprises a plurality of N antenna sub elements and a plurality of N-i impedances, each providing a connection in their center, at least one of said impedance being provided between said sub elements, whereby at least one of said means for outputting or inputting being provided by said connection.
4. 4. The antenna system of claim 1, 2 or 3 wherein said antenna system further comprises N-i common mode suppressing elements (200) (such as a balun transformer or common mode choke), of which at least one (optionally up to N-i) is being connected in between one of said ports and one of said matching networks.
5. 5. The antenna system of any of the previous claims, wherein N and/or M and/or Z and/or K equal 2, preferably N, M, Z and K are 2
6. 6. The antenna system of claim 5, wherein said antenna element being designed in order to provide that said modes are respectively the common mode and differential mode.
7. 7. The antenna system of claim 5 or 6, wherein each of said matching networks being optimized to match and/or to tune for its corresponding radiation mode.
8. 8. The antenna system of any of the previous claims 5 to 7, wherein said antenna element, comprises two antenna sub elements and an impedance, providing with a connection in its center, said impedance being provided between said sub elements, whereby one of said means for outputting or inputting being provided by said connection (to thereby represent a first signal referenced to ground), while the other of said means for outputting or inputting being provided by two connection points with said two sub elements (to thereby represent a second signal therein between).
9. 9. The antenna system of claim 8, wherein said connection points being located at the interface of said two sub elements with said impedance.
10. 10. The antenna system of any of the previous claims, wherein said impedance is one of a center tapped inductance or a center tapped capacitance.
11. 11. The antenna system of any of the previous claims 2, 3, 8 or 9 wherein said antenna sub elements being a dipole, a bow-tie element or a helix or a meandering line in serial connection with a capacitive plate.
12. 12. The antenna system of any of the previous claims wherein said antenna element being geometrically symmetric.
13. 13. The antenna system of any of the previous claims, wherein one or more of said matching networks comprise of tunable elements, preferably MEMS tunable elements.
14. 14. The antenna system of claim 13, wherein said tunable elements being digitally tunable capacitors and/or inductors.
15. 15. The antenna system of any of the claims 2, 3, 8 or 9, wherein one or more of said antenna sub elements comprises (electrical and/or mechanical) switches.
16. 16. The antenna system of claim 15 in combination with claim 11, wherein said switches being capable for electrical shorting one or more of the windings of said helix or for shunting one or more of said meander lines.
17. 17. The antenna system of any of the claims 15 to 16 wherein said switches comprise PIN diodes and/or FET transistors and/or MEMS SPST switches.
18. 18. A matching and tuning system, comprising: (1) P connections (130) to connect antenna sub elements thereto; (2) at least M>i means (30) (40) at which one and only one radiation modes can be defined; (3) an impedance (110), providing with a connection (120) in its center; (4) at least Z>i matching networks (SO), each one of said matching networks being connected to one and only one of said at which one and only one radiation modes can be defined; and (4) at least K>i ports (60), each being connect to one and only one of said matching networks, whereby one of said M>1 means for outputting or inputting said radiation modes being said center connection of said impedance, wherein at least one of said matching networks being tunable.
19. 19. The matching and tuning system of claim 18, further comprising one or more impedances, each of said impedances being provided with a connection in its center, part of said impedances being provided between two of said connection points, whereby one or more of said means at which one and only one radiation modes can be defined being provided by said connection (to thereby represent a first signal referenced to ground), while one or more of the other of said means at which one and only one radiation modes can be defined being provided by two of said connection points (to thereby represent a second signal therein between).
20. 20. The matching and tuning system of claim 18 or 19, comprising a plurality of N-i impedances.
21. 21. The matching and tuning system of claim 18, 19 or 20, further comprises N-i common mode suppressing elements (200) (such as a balun transformer or common mode choke), of which at least one is being connected in between one of said ports and one of said matching networks.
22. 22. The matching and tuning system of any of the previous claims, wherein N and/or M and/or Z and/or K equal 2, preferably N, M, Z and K are 2.
23. 23. The matching and tuning system of claim 22, wherein each of said matching networks being optimized to match and/or to tune for its corresponding radiation mode.
24. 24. The matching and tuning system of any of the previous claims, wherein said impedance is one of a center tapped inductance or a center tapped capacitance.
25. 25. The matching and tuning system of any of the previous claims, wherein one or more of said matching networks comprise of tunable elements, preferably MEMS tunable elements.
26. 26. The matching and tuning system of claim 25, wherein said tunable elements being digitally tunable capacitors and/or inductors.
27. 27. The matching and tuning system of any of the previous claims, wherein the topology of two or more of said matching networks being equal, preferably said matching networks have a EMN portion have a li-type topology and a AMN portion have a balanced T-type topology.
28. 28. An antenna system, comprising: (1) an antenna element, capable of handling (generating, receiving) at least N>1 radiation modes, said antenna element, comprises two antenna sub elements, one or more of said antenna sub elements being switchable, and provided with at least M>1 means for outputting or inputting at least one of said radiation modes; (2) at least Z>1 matching networks, each one of said matching networks being connected to one and only one of said means for outputting or inputting radiation modes; and (3) at least K>1 ports, each being connect to one and only one of said matching networks, wherein one or more of said matching networks being tunable.
29. 29. The antenna system of claim 28, wherein one or more of said matching networks comprise of tunable elements.
30. 30. The antenna system of claim 28 or 29 wherein one or more of said antenna sub elements comprises (electrical and/or mechanical) switches
31. 31. The antenna system of claim 29 and 30, wherein said switches are arranged for coarse switching and said tunable elements being arranged for fine tuning.
32. 32. The antenna system of any of the claims 28-31, being arranged in that irrespectively of any one of said tunings and/or switching the mapping from radiation modes to a port remain unaffected.
33. 33. A method of designing an antenna system (10) (wherein each radiation mode is directly mapped to a port), comprising: (1) an antenna element (20) (optionally an Electrical Small Antenna), capable of handling (generating, receiving) at least N>1 radiation modes and provided with at least M>1 means (30) (40) for outputting or inputting said radiation modes, preferably each one and only one of said radiation modes; (2) at least Z>1 matching networks (5), each one of said matching networks being connected to one and only one of said means for outputting or inputting radiation modes; and (3) at least K>1 ports (60) each being connect to one and only one of said matching networks) such that said antenna system is useable and hence matched over a desired (wide) portion of the band, the method comprising of (a) providing that at least one, preferably all, of said matching networks being tunable for its corresponding radiation mode; and (b) providing that one or more) preferably all of said antenna sub elements being switchable, further characterized in that said switches are arranged for coarse switching and said tunable elements being arranged for fine tuning.