GB2448741A

GB2448741A - Current sensing and overload protection of a switch mode power converter

Info

Publication number: GB2448741A
Application number: GB0708112A
Authority: GB
Inventors: Paul Thomas Ryan
Original assignee: Cambridge Semiconductor Ltd
Current assignee: Cambridge Semiconductor Ltd
Priority date: 2007-04-26
Filing date: 2007-04-26
Publication date: 2008-10-29
Also published as: GB0708112D0; WO2008132501A2; WO2008132501A3

Abstract

A bi-level current sense circuit comprises two voltage comparators 808a,808b that have different respective thresholds TH1,TH2, and share a single current sense input 806 connected to a current generator 804 and via a resistor Rp to a current sense resistor Rs. The two bi-level current level thresholds can be set independently by the current sense Resistor Rs and the series resistor Rp, without changing thresholds TH1,TH2. The current level thresholds can be set by selecting the values of resistors Rp and Rs. The current sense circuit can be used to detect the current trough a switch 204 and a magnetic storage device 202 of a resonant discontinuous forward power converter (RDFC) where the comparators are connected to a switch controller 810 that can select a high or a low power mode in response. Over current protection can be provided by sensing a combination of current flowing in a winding of the magnetic storage device and a rate of change of that current. A series capacitor (fig 9, Cff) and a potential divider (fig 9, R2,R3) across the current sense resistor provide a signal combining current with its rate of change. When the combined sense signal exceeds a threshold the on time of the switch can be reduced to reduce an average power input to the magnetic device.

Description

Switching Power Converters

FIELD OF TI-IE INVENTION

This invention relates to switching power converters, in particular to current sense and over-current control techniques for such converters.

BACKGROUND TO TFIE INVENTION

In a switching power converter broadly speaking a magnetic energy storage device, in many cases a transformer, is used to cyclically transfer energy from an input to an output of the converter by switching power through the device. The converter may be a forward converter, when power is transferred to the output when the switch is on, or a flyback converter when power is transferred when the switch is off. A range of power switching devices may be employed including, but not limited to, a bipolar junction resistor (BJT), an insulated gate bipolar transistor (IGBT) and a MOSFET.

Forward power converters have a number of advantages including relatively small size and low cost; they also have potentially good efficiency because they may be operated in resonant mode. However conventionally they have been difficult to regulate and the components, particularly the switch, have been prone to failure under some load conditions and at start-UI).

in power converters in general and in switching power converters, especially forward power converters, in particular, some form of current limiting is desirable. For example, with an integrated circuit used as a controller in a switching power converter a low-value resistor in series with the primary power circuit can be used to monitor current. The resistor may be inserted in the low voltage connection of the primary switch and the controller device may then sense the voltage across it. Internally, the controller may use the sensed signal for a range of power management functions, but particularly for triggering over-current protection. The information that is desired is whether the primary current is above or below a preset threshold. This can be accomplished using a voltage comparator to produce a digital signal which triggers if the signal voltage from the sense resistor exceeds a reference voltage.

There may also be a requirement to compare the sense resistor signal to another threshold, for example to detect low power conditions so that the controller can change to a low duty cycle mode. A second comparator may be included in the controller device to perform this comparison to a different threshold voltage. However, it is desirable to be able to set different thresholds for, for example, low power and overload protection thresholds in different applications. Making the thresholds in the controller device programinable could achieve this, but this is complex and expensive to implement and control.

Furthermore, it is desirable for power converters to use a controller in the form of an integrated circuit. In such a case, it is desirable for the integrated circuit to have as few connection pins as possible, as this reduces the area and cost of such a controller.

SUMMARY OF THE INVENTION

Bi-level current sensing The present invention therefore provides a bi-level current sense circuit for a power converter, to enable a single current sense input to provide two different current sense levels, said power converter comprising a magnetic energy storage device coupled between an input and an output of said power converter, a switch to switch power to said magnetic energy storage device, and a controller to control switching of said switch; said current sense circuit comprising: a single current sense input for connection to a current sense resistor via a second resistor, said current sense resistor for coupling in series with said switch to generate a current sense signal representing a current flowing through said switch; first and second comparators sharing a connection to said CulTent sense input and having respective first and second outputs coupled to said controller, said first and second comparators being configured to compare a voltage on said shared input connection against different respective first and second threshold values; and a current generator coupled to said shared input connection; and wherein said single current sense input enables third and fourth bi-level current sense threshold values to be set independently by said current sense and second resistors without changing said first and second threshold values.

In embodiments the levels of thresholds are effectively changed from first and second threshold values to third and fourth threshold values by the use of two resistors, a current sense resistor and a second resistor connected in series with the current sense resistor. The change in level of threshold levels may therefore be performed without the need to adjust either or both internal threshold levels. Preferably, selection of values of said second resistor and said current sense resistor sets the values of said third and fourth threshold values. Preferably, said controller controls said switch in response to signals on said first and second outputs of said respective first and second comparators.

The comparators may, but need not necessarily, connect directly to the shared input connection. The current generator may comprise a current source or sink, and in a simple embodiment may comprise a resistor connected to a higher voltage than the shared input connection.

The present invention also provides a controller for a power converter, said power converter comprising: a magnetic energy storage device coupled between an input and an output of said power converter; a switch to switch power to said storage device; and a controller to control switching of said switch, said controller comprising a bi-level current circuit according to any one of the above statements.

The present invention further provides a power converter comprising: a magnetic energy storage device coupled between an input and an output of said power converter; a switch to switch power to said storage device; a controller to control switching of said switch; a single current sense input connected to a current sense resistor via a second resistor, said current sense resistor connected in series with said switch to generate a current sense signal representing a current flowing through said switch; a first and second comparator sharing a connection to said current sense input and having respective first and second outputs coupled to said controller, said first and second comparators being configured to compare a voltage on said shared input connection against different respective first and second threshold values; and a current source coupled to said shared input connection; and wherein said single current sense input enables third and fourth bi-level current sense threshold values to be set independently by said current sense and second resistors without changing said first and second threshold values.

The present invention also provides a power converter controller IC including a bi-level current sense device for a power converter, said power converter comprising a magnetic energy storage device coupled between an input and an output of said power converter and a switch to switch power to said magnetic energy storage device; wherein said IC has a single external current sense connection, said single external connection being coupled to a pair of internal comparators with different thresholds, and such that said current sense device is configured to enable selection of two different independently externally adjustable current sense thresholds simultaneously by sensing a single current sense voltage level.

The present invention also provides a method of providing two different levels of sensed current threshold using a single current sense input, a current sense resistor and a second resistor, said current senses resistor and said second resistor being connected in series, the method comprising: passing a current for sensing through said current sensing resistor; passing a second current via said current sense input through a said second resistor connected in series with said current sense resistor to generate a sensed voltage at said current sense input; and comparing said sensed voltage at said single current sense input against two different comparator threshold values.

Advantageously, the levels of thresholds can be changed using only two resistors, a current sense resistor and a second resistor connected in series with the current sense resistor. The change in level of threshold levels may therefore be performed without the need to adjust internal threshold levels.

Thus the invention also provides a method of sefting the two different sensed current threshold levels, the method comprising selecting values of the current sense resistor and the second resistor without altering the comparator threshold values.

Rate adaptive current limiting According to another aspect of the invention there is provided a forward switch mode power converter, said power converter having an input to receive an input power supply, an output to provide an output power supply, a magnetic device coupled between said input and said output, said magnetic device having at least one winding, and a switching device coupled to switch power from said input on and off to said winding of said magnetic device to transfer power from said input to said output, said power converter further comprising: a sensing circuit to sense a combination of a current flowing in said winding when said switching device is on and a rate of change of said current; and a control system coupled to said sensing circuit to control said switching of said switching device responsive to said sensed combination to control an output current from said converter to provide over-current protection.

We have described, above, bi-level current sensing techniques which may be employed, for example, to select either a low or a high power mode for a switch mode power supply. In, say, a high power mode the switching device may be turned off to avoid dissipating too much power if the load is heavy. More specifically over-current protection of this sort may be provided on a cycle-by-cycle basis by switching the switching device off immediately an overload condition is detected, thus shortening the on period (without necessarily significantly changing the off-period, for example because the converter may still be resonant). If a heavy output load is applied then the input side current builds according to the leakage inductance between the input and output (in a circuit model of the power converter), that is if the inductance is large the current builds relatively more slowly. If the current crosses a high current threshold then the switching device may be turned off early, thus reducing the average power transferred. If the output voltage is clamped to a relatively low level, say by a high load or close to short circuit, there is a larger input-output mismatch and thus the input side current in the on period rises faster.

In embodiments the power converter may be a resonant power converter. In a converter where the off time is resonant, a range of behaviour may be observed in a forward switch mode power converter, depending on the power delivered (the off time affects the power delivered because this is broadly proportional to the ratio of on to off time of the switch). Some examples of the observed behaviour are described later with reference to Figure 3f but, broadly speaking, it can be possible for a relatively high output current to exist in combination with a relatively low output voltage. Similarly, high currents may be observed in non-resonant forward converters under overload conditions. Embodiments of the above-described forward switch mode power converter address this problem by adding to a signal compared with a threshold value a component proportional to a rate of change of sensed input side or primary current rather than simply comparing the sensed current with a threshold. Thus, for example, if the output voltage is clamped at a low level the current rises quickly and this contributes more strongly to the combination of sensed current and rate of change of current to thus trip the over- current protection, with the effect of shortening the on-period further (earlier) than would otherwise be the case. This particularly affects what would otherwise be the low output voltage-high output current portion of the output characteristic of the power converter. Depending upon the proportion of rate of change of current included more or less weight may be given to this type of high current (low voltage) output condition in operation of the over-current protection.

In some preferred embodiments a current sense signal is generated using a current sense resistance series coupled with the switching device and the at least one winding of the magnetic energy storage device. Conveniently a signal dependent on the rate of change of this sensed current may be generated by a capacitor and a second resistance coupled in series and receiving a voltage generated by this current sense resistance, the changing voltage (sensed current) generating a current through the capacitor which is converted to a voltage by the second resistance. Particularly conveniently a signal comprising a first part dependent on the sensed current and a second part dependent on a rate of change of the sensed current may be generated using a potential divider by coupling a third resistance in parallel with the aforementioned capacitor such that the second and third resistance form a potential divider. For example, a potential divider may be connected across the current sensing resistor and a capacitor connected in parallel with one resistor of the potential divider. The combined signal can be compared against the threshold and used to control the switching device, for example to curtail the on period of the switching device in response to the signal exceeding the threshold in order to reduce the average power transferred to the output. It will, however, be appreciated that other control lec]miques mayalso be employed including, but not limited to, one or more of pulse amplitude, pulse width, pulse frequency, and variable slope modulation.

In a related aspect the invention provides a forward switch mode power converter including a controller to control said power converter, said controller having a current sense input, and wherein said power converter further comprises a current sense circuit coupled to said current sense input, said current sense circuit comprising a current sense resistance, a series coupled capacitor and second resistance (R3) together coupled in parallel with said current sense resistance, and a third resistance (R2) coupled in parallel with said capacitor, and wherein a signal at said current sense input comprises a signal dependent upon a combination of a current sensed by said current sense resistance and a rate of change of said current.

In a further related aspect the invention provides a method of over-current protection in a forward switch mode power converter, said power converter having an input to receive an input power supply, an output to provide an output power supply, a magnetic device coupled between said input and said output, said magnetic device having at least one winding, and a switching device coupled to switch power from said input on and off to said winding of said magnetic device to transfer power from said input to said output, the method comprising: sensing a combination of a current flowing in said winding when said switching device is on and a rate of change of said current; and controlling switching of said switching device responsive to said sensed combination to control an output culTent from said converter to provide said over-current protection.

In some embodiments of the method the equivalent inductance between the input and the output of the power converter may be used to substantially remove a region of negative output resistance of the power converter when the over-current protection is active.

Features of the above-described aspects and embodiments of the invention may be combined in any permutation. In particular, the second resistor used in combination with the current sense resistor in the bi-level current sense circuit may be the same resistor as used to provide the third resistance in the sensing circuit responsive to both sensed current and rate of change of the sensed current. In this way a combination of bi-level current sensing and rate-based or rate adaptive current sensing (i.e. dependent to a degree on the rate of change of current) may efficiently he implemented.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other aspects of the invention will now be further described, by way of example only, with reference to the accompanying figures in which: Figures la and lb show, respectively, an embodiment of a discontinuous resonant forward converter (RDFC) in the context of which the tecirniques we describe may be employed, and an example timing and control arrangement for the converter of Figure 1 a; Figure 2 shows a further circuit diagram of an RDFC; Figure 3a, 3b, 3c, 3d and 3e show example waveforms of converter operation; Figure 3f shows the effects of protection threshold and leakage on the output voltage/current characteristic; Figure 4a and 4b show simplified equivalent circuits of an RFDC; Figure 5a shows a range of current waveforms corresponding to different Vin -Vout' differences; Figure 5b shows the effect of regulating power delivery in response to time-sampled primary current; Figure 6 shows an increasing resonance time with a reduced on-type of the switch of the RDFC; Figure 7 shows an example RDFC; Figure 8 shows an example of a bi-level current limit circuit according to an embodiment of an aspect of the invention; Figure 9 shows an example of a rate-adaptive current limit circuit according to an embodiment of an aspect of the invention; Figure 10 shows the effect of the circuit of Figure 9 on the output characteristic of an RDFC with low leakage inductance; Figure 11 shows the effect on the output characteristic of an RDFC with increased leakage inductance; and Figure 12 illustrates a potential effect of the circuit of Figure 9 on the output characteristic of an RDFC with a very high leakage inductance.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Resonant discontinuous forward converters Referring to Figure Ia, (which is taken from our earlier application US 11/639,827), this shows an example of resonant discontinuous forward converter 100; Figure lb shows an example timing and control system 210 for the converter. A dc input 102 coupled to the primary winding 104 of a transformer 106, connected in series with a power switch 112.

A resonant capacitor 114 is connected across the primary winding of the transformer and the dc input 102 is provided with a smoothing capacitor 116. On the output side of the forward converter a secondary winding 108 of the transformer provides power to a pair of dc output terminals 118 via a rectifier 120. A smoothing capacitor 122 is connected across the dc output terminals 118 and an output node at the junction of rectifier 120, smoothing capacitor 122 and a connection to one of the dc output teni'iinals 218 is denoted "X". The current into node X, which flows to either or both of the smoothing capacitor 122 and output 118, is discontinuous.

The switch 112 may comprise a bipolar or MOS transistor for example a MOSFET, IGBT, or BJT.. The rectifier 120 may be implemented as a diode or by means of a MOS transistor. The resonant capacitor 114 may either comprise a discrete component, or may be entirely provided by parasitic capacitance, or may be a combination of the two.

The switch 112 is controlled by a controller 110 comprising a timing control module 11 Oa and a switch control module 11 Ob, the timing control module providing switch on and switch off signals 11 Oc to the switch control module 11 Ob. The timing control module may have one or more sense inputs, such as a voltage sense input and a current sense input as illustrated, or such sensing may be omitted and the timing control module 11 Oa may operate substantially independently of any sensed condition of the forward converter circuit. Using a controller 110 to control the timing of the switch 212 on and off allows a variety of advantageous control techniques to be employed including, for example, current limiting, start-up control and regulation. The controller may, for example implement one or more of pulse amplitude, pulse width, pulse frequency, and variable slope modulation to control the switching device.

Where voltage sensing is employed the voltage on the primary winding of the transformer may be sensed, either directly or indirectly. For example the voltage may be sensed as shown by means of a connection to a j unction between the primary winding and switch; alternatively, for example, a sensing voltage may be derived from an auxiliary winding of the transformer (not shown in Figure Ia). Where current sensing is employed this may be conveniently implemented by sensing the voltage across a current sense resistor.

In operation the circuit of Figure 1 a converts the input dc voltage, typically relatively high, to an output dc voltage, typically in a range suitable for consumer electronic devices, for example between around 5V and 20V. Preferably the dc output is isolated from the dc input, as shown in Figure la; in other implementations secondary side feedback may be employed, in which case an opto-isolator may be included to provide isolation between the primary and secondary sides of the forward converter.

Figure lb illustrates an example implementation of the controller 110 of Figure Ia. A comparator 150 compares a sensed voltage with a reference voltage, for example zero volts, to provide a control signal 152 to a switch control unit 156 to control switch 112 on. The output of comparator 150 is also provided to a timer 158 which begins timing an on pulse width. When the timer times out a signal is provided on a second control line 154 to switch control unit 156 to control switch 112 off. Switch control unit 156 may comprise, for example, a set-reset latch together with interface circuitry for driving the base of a bipolar transistor andlor the gate of an MOS transistor. The controller may include an OR gate 160 with an input 162 from an over current protection line. This input may be generated by a comparing a current sense input with a reference level defining a threshold for current limiting. Thus in this example when the over current protection input 162 becomes active the switch control unit 156 is immediately controlled to switch 112 off, thus implementing cycle-by-cycle current limit control.

Power management techniQues We have previously described a method of controlling a resonant discontinuous forward converter (RDFC), said converter including a transformer with primary and secondary matched polarity windings and a switch to, in operation, cyclically switch DC power to said primary winding of said transformer, said converter further having a DC output coupled to said secondary winding of said converter, said method comprising: sensing a primary winding signal during an on period of said switch, said primary winding signal representing a current in said primary winding; comparing said sensed primary winding signal with a threshold value; and controlling one or both of an on and off duration of said switch in response to said comparison. * 12

The above method advantageously enables an RDFC to be controlled using signals sensed on the primary side of the transformer, which obviates the need for the use of isolated components to convey feedback signals from the secondary side of the transformer to a controller. This may reduce the cost of such a device.

Preferably, said threshold value is an overload threshold value; and said method further comprises: generating an overload signal for triggering an overload condition in said RDFC, said overload signal representing a period during which said sensed primary winding signal is greater than said overload threshold value. Generating an overload signal may prevent damage occurring to the transformer, other components in the converter or in any connected load.

Preferably, the method further comprises: sensing a first event associated with switch on of said switch; sensing a second event associated with said overload signal; determining a duration between said first and second events; comparing said duration with an overload period threshold value; and generating an early overload detection signal for detecting an early overload condition of said RDFC, said early overload detection signal being generated in response to said sensed duration being below said overload period threshold value.

Preferably, said duration is sampled over two or more on and off cycles of said switch and wherein said duration is compared with said overload period threshold value.

Preferably, said on duration of said switch is reduced in response to said overload condition. Preferably, one or both of said on and off durations are controlled in response to said early overload detection signal. Preferably, said switch is switched off in response to said overload condition, or said overload detection signal.

Preferably, the method comprises entering a latched mode following detection of said overload or early overload condition, said latched mode defining a period during which durations of one or both of said on and off durations of said switch are varied in response to said overload or early overload conditions or conduction of the switch is disabled.

Preferably, the method comprises a starting method, said starting method comprising: increasing said on duration or reducing said off duration of said switch for a burst period.

Preferably, said burst period is increased if said sensed primary signal falls below said overload threshold value during said burst period. Alternatively, said burst period is increased if said duration is above said overload period threshold value during said burst period. Preferably, said RDFC is prevented from entering a latched mode during said starting method, said latched mode defining a period during which durations of one or both of said on and off durations of said switch are varied in response to said overload or early overload conditions or conduction of the switch is disabled.

In an alternative, said threshold is a regulation threshold value; and wherein said on duration of said switch is reduced or said off duration of said switch is increased in response to said comparison.

Preferably, an output voltage of said RDFC is regulated.

We have also previously described a controller for controlling a resonant discontinuous forward converter (RDFC), said converter including a transfonner with primary and secondary matched polarity windings and a switch to, in operation, cyclically switch DC power to said primary winding of said transformer, said converter further having a DC output coupled to said secondary winding of said converter, the controller comprising: an input to sense a primary winding signal during an on period of said switch, said primary winding signal representing an operational current in said primary winding; a comparison means to conipare said sensed primary winding signal with a threshold value; and an output to control one or both of an on and off duration of said switch in response to said comparison.

We have also previously described a method of detecting an overload condition of a resonant discontinuous forward converter (RDFC), said converter including a transformer with primary and secondary matched polarity windings and a switch to, in operation, cyclically switch DC power to said primary winding of said transformer, said converter further having a DC output coupled to said secondary winding of said converter, said niethod comprising: sensing a primary winding signal during an on period of said switch, said primary winding signal representing an operational current in said primary winding; comparing said sensed primary winding signal with an overload threshold; and generating an overloadsignal for triggering an overload condition in said RDFC, said overload signal representing a period during which said sensed primary winding signal is greater than said overload threshold.

Preferably, the above method comprises: sensing a first event associated with switch on of said switch; sensing a second event associated with said overload signal; determining a duration between said first and second events; comparing said duration with an overload period threshold value; generating an early overload detection signal for detecting an early overload condition of said RDFC, said early overload detection signal being generated in response to said sensed duration being below said overload period threshold value.

We have also previously described a controller configured to detect an overload condition of a resonant discontinuous forward converter (RDFC), said converter including a transformer with primary and secondary matched polarity windings and a switch to, in operation, cyclically switch DC power to said primary winding of said transformer, said converter further having a DC output coupled to said secondary winding of said converter, the controller comprising: an input to sense a primary winding signal duriiig an on period of said switch, said primary winding signal representing an operational current in said primary winding; comparison means to compare said sensed primary winding signal with an overload threshold value; and a system to generate an overload signal for triggering an overload condition in said RDFC, said overload signal representing a period during which said sensed primary winding signal is greater than said overload threshold value.

We have also previously described a method for regulating an output voltage of a resonant discontinuous forward converter (RDFC), said converter including a transformer with primary and secondary matched polarity windings and a switch to, in operation, cyclically switch DC power to said primary winding of said transformer, said converter further having a DC output coupled to said secondary winding of said converter, said method comprising: sensing a primary winding signal during an on period of said switch, said primary winding signal representing an operational current in said primary winding; comparing said sensed primary winding signal with a regulation threshold value; and controlling one or both of an on and off duration of said switch in response to said comparison.

Preferably, said sensing of said primary winding signal comprises: detecting turn on of said switch; waiting for a delay period; sampling said primary signal at an end of said delay period.

Preferably, said detecting turn on of said switch comprises: detecting an increase in said primary winding signal following turn on of said switch that is greater than a first threshold. Alternatively, said detecting turn on of said switch comprises detecting a drive signal of said switch. Alternatively, said detecting turn on of said switch comprises: detecting a voltage across said switch.

We have also previously described a controller for regulating an output voltage of a resonant discontinuous forward converter (RDFC), said converter including a transformer with primary and secondary matched polarity windings and a switch to, in operation, cyclically switch DC power to said primary winding of said transformer, said converter further having a DC output coupled to said secondary winding of said converter, the controller comprising: means for sensing a primary winding signal during an on period of said switch, said primary winding signal representing an operational culTent in said primary winding; means for comparing said sensed primary winding signal with a regulation threshold value; and means for controlling one or both of an on and off duration of said switch.

Preferred embodiments of the techniques we describe in this specification are concerned with controlling resonant discontinuous forward converters (RDFCs), although they are not limited to RDFCs. In an exemplary RDFC power to a primary or input winding of a transformer is switched and a secondary or output winding of the transfoimer, with a polarity matched to that of the primary winding, is coupled to a rectifier which provides dc power to a smoothing capacitor, dc power being supplied by the RDFC to its output from this connection (node X in Figure 1.). A voltage waveform on the secondary winding of the transformer has a first portion during which the switch is on and current flows into the output connection iiode X, aiid second substantially resonant portion during which both the switch and the rectifier are off. Substantially no current flows into the connection node (other than from the smoothing capacitor) during the second portion of the voltage waveform.

In some example designs we describe a connection between the rectifier and the connection node may include a small inductor (for example less than 5% of the primary side magnetising inductance) but substantially no current flows in this inductance during the second, resonant portion of the waveform and there is no need for a large choke of the type used in a continuous forward converter. There is no need connect a capacitor across the rectifier to achieve resonance; other connection positions are possible, for example, across a primary, secondary or auxiliary winding of the transformer. More particularly in embodiments we use the magnetising inductance of the transformer with an added capacitor on the primary side to achieve resonance in the off cycle.

In some preferred RDFC implementations the RDFC is configured for AC-DC power conversion and thus includes an AC-DC converter such as a bridge rectifier on the primary side. In some particularly preferred implementations the RDFC is mains-powered and the primary side is powered by a high dc voltage (for example greater than 7OVdc, 1 OOVdc, I 5OVdc or 200 Vdc) whilst the secondary side dc voltage is low (for example, less than 2OVdc or I OVdc). In embodiments we employ minimum voltage switching on the primary side (i.e. a primary side switch is turned on at a time when a voltage across the switch is close to zero volts or at a minimum).

We have also previously described techniques for implementing a resonant discontinuous forward converter (RDFC) which employ a control system to turn a power switch of the RDFC on and off in a controlled manner. As previously described, the control system may operate in an uncontrolled, fixed frequency mode or the control system may sense from one or more inputs and decide when to turn the power switch on and off responsive to this sensing, for example to implement pulse width and/or frequency modulation. This facilitates regulation of the RDFC which, in detail, may be performed using a range of algorithms. One technique uses the control system to operate the RDFC to compensate for circuit variables and to operate in a minimum voltage switching mode. The converter may also control the switching frequency during start-up and/or current limit in order to protect the power switch and increase the energy transferred to the load. The control system is preferably implemented using a control IC (integrated circuit).

As mentioned above, the RDFC operates without a freewheeling or flyback diode, and with or without an output inductor. However, if present the output inductor is sufficiently small to ensure that the forward converter operates in a discontinuous mode and substantially resonantly that is at or close to resonance.

Within an RDFC, the timing of the primary power switch is preferably in accordance with the resonance of the isolating transformer and other components forming part of the resonating circuit; managed to control the power transfer to the output; and appropriate to the other components (particularly the transformer) so they operate without excessive stress and within an efficient range. In preferred embodiments, timing is set by a controller, which determines the time to turn the primary switch on and how long to leave it on.

Fig. 2 shows a basic arrangement of an RDFC 200, comprising an isolating transformer 202 having primary 202p and secondary 202s windings. A switch 204, when closed, applies a DC voltage Vht across the primary winding of the transformer. Power is delivered to the load 212 via a rectifying diode 208 and smoothing capacitor 210. The polarity of the transformer is such that power is supplied to the output circuit when the primary switch is closed. Inductance Licak may be a discrete component or may be a characteristic of the transformer itself, in which case it can be modelled as an inductor in either the primary circuit or the secondary circuit, or both. The magnitude of Leak affects the behaviour and performance of the converter. For applications requiring more stable voltage regulation Licak is normally lower in value; for applications requiring a higher converter output impedance (such as battery chargers), its value is set higher.

Example waveforms of converter operation are shown in Fig. 3a. When switch 204 is in the On state, current flows through the switch and transformer primary. The current rises in accordance with load conditions and the effect of Lteak. At switch-off, the voltage rises rapidly (reflected load current in Lleak) to point "A". Once energy has been transferred out of the leakage inductance, the circuit resonates as the combination of Cres 206 and the magnetising inductance of the transformer Lrnag, and other stray reactances.

During resonance, the switch voltage reaches a peak then reduces. Depending on circuit values and conditions, it may reach OV ("B"). Depending on the characteristics of circuit components, the switch voltage may be prevented from resonating below OV either by diodes associated with the switch, or by the effect of the output diode and voltage on the output capacitor.

Fig 3a shows the switch being turned on at point "B". Fig 3b shows example waveforms of the switch being turned on before the resonant waveform reaches point "B", and Fig 3c (on a longer time axis) shows the switch being turned on later than point "B". In Fig. 3c, the switch waveform can be seen to oscillate as a damped sinewave around V1.

Optimum power transfer is achieved by turning the switch on again at point "B".

It is preferable to adjust one or both of an on-time of the switch and an off-time of the switch to control the power transfer. 1-lowever, the on-time is preferably chosen to correspond to the resonance of the switch voltage.

Preferably, the switch is turned on when the switch voltage is close to OV ("B" in Fig. 3a). For situations where there is an extended off-time (as shown, for example in figure 3c), it is preferable to turn the switch back on when the resonant voltage is at an instant of minimum voltage (e.g. "C" in Fig. 3c); this reduces EM! and turn-on loses in the switch 204.

Apart from component values and parameters, the resonant waveforms observed in an RDFC vary according to load and switching conditions. Figure 3d is a more detailed view of the waveform shown in figure 3a and shows the main parts of the observed resonance wave. The polarity represents the voltage across the power switch assuming a positive supply to the transformer primary and the power switch is connected in the other connection of the primary to the negative supply.

The three main parts arc: On-period: when the power switch is turned on, connecting the transformer primary to the power source. Voltage across the switch is low. During this period current flow is principally reflected secondary (load-related) current and magnetising current rising according to the primary inductance and supply voltage etc. Leakage resonance: In this period, current continues to flow in the secondary. The primary voltage is a (fast) resonance of the leakage inductance with the resonant capacitance. The magnitude of this resonant voltage depends on the current at turn-off and the leakage inductance. The former is typically mostly the reflected secondary current, but there may be a small contribution from magnetising current, depending on circuit parameters. Typically, this resonance is fast and the turn-off time of the switch may be significant so the exact waveform is complex.

Magnetising resonance: Once the secondary current has substantially decayed, the wave results from resonance of the magnetising (primary) inductance and the resonant capacitance. The amplitude depends on the magnetising current at the end of the on-period which depends on the magnetising current at the beginning of the on-period, the supply voltage and the duration of the on-period etc. As load and switching conditions vary, the principal effects are: * Amplitude and duration of the leakage resonance increases with the sum of reflected load current and magnetising current. These scale according also to the magnitude of the leakage inductance.

Amplitude of the magnetising resonance increases with on-period and supply voltage. Duration of the magnetising resonance depends on the magnetising current remaining at the end of the leakage resonance and the voltage at that time.

Typical waveforms are illustrated in figure 3e; all correspond to the condition where the on-period commences immediately the switch voltage reaches a minimum. Though the on-time and reflected secondary current are not necessarily dependent, in typical applications the on-period is shortened when the secondary current is high in overload.

In applications where the power transfer is varied by changing the on-time and off-time of the primary switch, at least three operational modes to manage power are identifiable: * Maximum power (Maximum on-time and minimum off-time) * Medium power (Reduced on-time and minimum off-time) * Low power (Minimum on-time and extended off-time) A prefelTed power management technique has the following operating modes: * "Standby B" (Low power): minimum on-time and extended off-time * "Standby A" (Low/medium power): variable on-time and resonant off-time * "Normal" (Medium/high power): maximum on-time and resonant off-time * "Overload" (High power): on-time shortened according to trigger of over-current protection by the instantaneous primary switch current, resonant (or minimum) off-time * "Foldback" (Limited power): on-time shortened as for Overload, off-time extended to reduce power and current * "Burst" (short-term maximum power): As overload, but applied for a fixed duration (or number of cycles) to provide robust start-up Preferably, the on-time is reduced before increasing off-time. Reducing the on-time minimises transformer losses, EMI and audible noise, as the core flux is reduced (consequence of reducing on-time). Minimising the off-time allows the on-time to be maxirnised at medium and high powers, whilst staying within allowable core flux, as this maximises the reverse magnetising current from the previous cycle.

Preferably, change of mode is performed on the basis of the measured primary switch current and its rate of rise, but a further choice is the rate at which the controlled parameters (on-time and off-time) are changed in response to the measured current. If the controlled parameters change slowly then the system will also respond slowly to changing conditions, possibly leading to undesirable voltage excursions.

A common requirement for power supplies and power converters is to include protection against adverse effects of overloads on either the power supply itself or on any connected load. Preferably, overload protection limits the output current; the limit may be dependent on the output voltage itself. For example, "foldback", where the limited current is lower when the output voltage is lower, may be used to limit power dissipation in fault conditions.

However, it is desired that sufficient power is delivered in transient and transient non-fault conditions, such as during start-up when large capacitors internal to the load must be charged quickly to the required operating voltage.

In addition to "overload" characteristics, there are some loads that require particular voltage/current characteristics, for example rechargeable batteries. Power must be delivered to these over a range of voltages, and with both current and voltage limiting to avoid damage or deterioration.

The forward converter does not have inherently good power/current limiting so it is desirable to manage the on-time and off-time to achieve a good characteristic. It is known to use circuits on the load (secondary) side of the transformer to monitor load current and delivered voltage. Signal(s) are passed back to the controller on the primary side to set the power delivery to achieve the required response. However this normally requires galvanic and safety isolation of the signals, which are expensive.

In embodiments of the present invention, this large cost is overcome by sensing only on the primary side. If an adequate power management characteristic can be achieved by sensing signals only from the primary side of the transformer then a lower cost is possible.

A useful parameter is the current through the primary switch which is approximately proportional to input power. If this current is compared against a threshold then a condition of overload can be recognised and appropriate action taken e.g. Turn off and disable turn-on. This has an advantage in that a quick response to overload and protection of the RI)FC is achieved. However, this method is sensitive to onloff ratio.

Alternatively, the peak and/or average value of current can be tested, which gives a true measure of power input, but has to be taken over a sufficient period to even out variation due to e.g. ripple at supply line frequency.

A combination of the above methods is possible. However, for simplicity and robustness, peak-current limiting is preferred.

In such a method, the instantaneous current though the primary switch is compared against a threshold and the switch is turned off if it is exceeded. Preferably, the controller will turn on again (after resonance) to attempt to maintain power output.

However, the controller may alternatively disable the switch such that the RDFC goes into a fault state to prevent any further power delivery.

When the on-time is cut short by excessive primary switch current, typically the off-time does not change proportionately as much, since it follows the resonance of the circuit. Consequently the onloff ratio reduces, further reducing power delivered to the load. As the load to the converter is increased, this may manifest as negative effective output resistance.

At low output voltages, the primary switch current rises rapidly during the on period.

Any delay in recognising this condition and turn-off of the switch can lead to excessive currents in either the transformer primary or secondary circuits. Further, at low output voltages, reset of the secondary winding inductance voltage may cause secondary current to flow for much longer than the on-period of the switch. This delivers higher culTents to the load which may cause adverse effects. It is therefore preferable to include further protection in these circumstances.

To detect this condition it is possible to sense when the on-time reduces below a predetermined threshold, either as a fixed value or a proportion of a measured resonance time, or some combination of the two. As the output voltage is reduced, the on-time reduces because the rate of rise of primary switch current rises, so crossing the over- current threshold sooner. Further protection can be triggered when the on-time reduces below the threshold. Triggering can be on the basis of short on-time in any individual cycle or if this occurs for a predetermined number of cycles. A predetermined number of cycles is preferable because it avoids false triggering caused by system noise or short-term events.

Though it is preferable to turn off the switch when the current passes a threshold, and to use this time to change to a protection mode, it is not essential. Alternatives include: * Separate thresholds to force switch turn-off and to trigger a further protection mode. The latter threshold would be set at a lower current compared to the former.

* Switch turn-off occurs after a delay following the time when current exceeds the threshold whereas the duration that triggers further protection is substantially the delay until current reaches the threshold These may be preferable where it is desired to obtain an overload characteristic that: * delivers output power more robustly before further protection triggers or * has a characteristic that can be modified automatically or * where further levels of protection are triggered by low output voltage, not only by peak primary current For example, when the transformer has a relatively low leakage inductance or high equivalent series resistance, the rise time of the current is a small portion of the on-period. It is this rise time that indicates the input-output voltage mismatch but it is the current delivered in the remainder of the onperiod and the duration of the on-period that determines the power delivery. So, further protection can be triggered by low output voltage, but high power can be delivered before protection operates.

As load current increases, the on-period primary current also increases. The overload current threshold is used to shorten the onperiod to protect the system and any coimected load. The effect is to shorten the on-period when high currents are taken, causing the output voltage to fall (or remain low). At the same time the increasing current at turn-off will cause a larger rise of switch voltage due to leakage inductance.

The combination causes a reduction in resonance time, so the operating frequency typically increases. Depending on the value of the leakage inductance in relation to resonance, the system may exhibit a range of behaviours with increasing load.

Leakage Behaviour Low Primary current very sensitive to Vin-Vout', short time constant of Lleak/Rtot.On-period reduces rapidly with reducing output voltage.

Resonance period shortens less quickly than on-period. Output current falls.

Negative output impedance High Primary current less sensitive to Vin-Vout', long time constant of LleaklRtot. On-period reduces slowly with reducing output voltage.

Resonance period shortens more quickly, due to effect of leakage energy.

Output current maintained. Positive output impedance The effects of protection threshold and leakage on the output voltage/current characteristic are illustrated in figure 3f.

The voltage/current characteristic of a converter may be altered to suit the application by choosing or adjusting the following factors: * Leakage inductance as a proportion of the magnetising inductance * On-time, optionally by changing the resonance period via the niagnetising inductance and resonating capacitance * Overload current threshold * Response of the controller to the overload signal (e.g. reduction of on-time) Once triggered into an overload condition, there is a range of possible protection strategies, such as complete cessation of operation or run in a low-duty ("foldback") mode. Once in the overload condition, the length of time before exiting the overload condition may include: a) Until user intervention e.g. by reset control or by removal and re-application of input voltage b) Until on-time increases above the threshold, or above some other lime threshold c) For a predetermined time then attempt to restart according to a robust scheme and return to foldback mode if on-time is still short or return to normal operation if the on-time has increased sufficiently.

Option a) can have a high degree of safety but may be inconvenient. Option b) requires continued converter operation, which has to be at a low duty. At low duty, the output voltage (hence the on-time) is unlikely to recover with typical loads, so it would be desirable for user intervention to reduce or remove the load before restart is possible.

Option c) is preferred as it avoids the need for user intervention but can still provide good safety because power and current, averaged over a foldbacklrestart cycle, can be limited to acceptable values.

An example of a robust restart scheme is to operate the converter in "normal" mode for a short predetermined period ("burst"), limiting the oil-time by over-current protection as described above, but not responding to short on-time. The length of the burst must be sufficient to restart all normal loads but not so long as to pose a hazard due to high currents and associated heating during tile burst.

When the output voltage is low the on-time is typically short (limited by overload protection) but on-state current may be high. Tf the off-period is set to the resonance period a consequence may be that tile on-period is comparable to the off-period, rather than substantially shorter. With high current delivered to the output during the on-period, the average output current may also be high. In some circumstances this may be desirable, for example to start-up quickly with highly capacitive loads, but may be excessive in others, causing damage or undesired behaviour. To avoid this, a convenient modification to the method is to limit the minimum off-period to a preset time, or a time measured from resonance under other load conditions. The effect is to reduce the on/off ratio in conditions of high overload. Alternatively or additionally, the on-period may be shortened further (less than that determined by the overload culTent threshold), in response to short resonance period.

Additionally, the change of on-time during the burst may be monitored. If the on-time increases (i.e. the time taken to tlip the over-current protection on each converter cycle) then this normally indicates that the output voltage is increasing; in this case the burst can be continued since it appears that the system is recovering. If it does not increase, or if the rate of increase is below a predetermined rate, then it is determined that the system is not recovering and the burst may be terminated sooner. This assures protection in overload conditions but improves the ability to restart with heavy and/or highly capacitive loads.

To minimise size and cost of power converters, it is preferable to use the maximum available range of flux in the transformer core, limited by the characteristics of the core material. In resonant operation, there may be a "negative" flux in the core at the beginning of the on-period of the power switch. During the on-period, the core flux increases (tends to more positive) and reaches a maximum approximately at the end of the on-period.

On-period is a primary means of control of power delivery, enabling reduction of power in response to light load or overload. In a particular design of power converter it is necessary to manage the on-period to ensure that a large, but not excessive, flux range is used. In resonant converters, the negative flux at the beginning of the on-period may be significant in reducing the peak positive flux at the end of the period. This can cause difficulties when the on-period or off period are changed, because both affect the negative flux at the beginning of the on-period.

If the on-period is increased in one converter cycle compared to the previous, the negative flux at the beginning of the on-period may be insufficient to limit the flux at the end to an acceptable level. To avoid this problem, it is preferable to limit the rate of increase of on-period between successive cycles. The increase between cycles may be simply at a fixed predetermined rate or, preferably, at a rate determined from the maximum on-period (itself may be determined from the estimated resonance time), such as a fraction of the maximum on-period.

Changes to the off-period affect the negative core flux at the beginning of the next cycle in complex ways. Firstly, the flux oscillatesdue to the resonant behaviour and secondly, it tends to decay as a result of circuit losses. As described elsewhere, to manage power delivery it is preferable to reduce the on-period before increasing the off-period. In these circumstances, there is no risk of excessive core flux provided the on-period is reduced to a relatively low value before the off-period is increased.

Electronic systems commonly need supply voltage to be controlled within a narrow range. Converters and power supplies often have voltage stabilising systems to reduce the variation of output voltage when input voltage, output voltage or other conditions (e.g. temperature) change. The present invention also attempts to reduce the effect of load current on output voltage.

Fig.4a shows a simplified equivalent circuit of and RFDC during the on-time of the primary switch. The circuit comprises an input supply 402, a switch 404, a primary-referred circuit total resistance (including contributions from the switch, transformer, output diode, output capacitor etc.) 406, a leakage inductance of the transformer (referred to primary) 408, a transformer 410 and an output voltage (including any fixed voltage drop e.g. in output diode) 412.

Fig. 4b shows a further simplified circuit of the circuit shown in figure 4a. In fig 4b, the secondary side parameters are referred to the primary side. When the switch is closed, the current builds through the loop according to the voltage difference between the supply (Vm) and the output voltage referred to the primary side (V0'). Since Liag is typically very large in comparison to Licak, its effect can be neglected when the on-time is short and there is significant difference between V and V'.

Fig 5a shows a range of current waveforms corresponding to different -V01' differences.

In RDFC operation it is possible to reduce the on-time while maintaining resonant conditions during off-time. Typically, the off-time increases as the on-time reduces, due to the reduced amplitude of oscillation during resonance (shown in Fig. 6). Whether the off-time increases or not, the converter duty can be reduced simply by reducing the on-time. When the duty is reduced, the average current delivered decreases for a given input-output voltage difference; or alternatively for constant load, the output voltage will fall if the duty is decreased.

To control the duty in relation to the delivered power requirement it is desirable for the controller to have access representing the latter. This can be done using an electronic circuit to measure delivered current and/or voltage, comparing it against a reference then using the resulting error signal to adjust the converter duty. However in the case where the output has to be galvanically isolated from the input (e.g. mains-powered off-line converters), signals should be passed across an isolation barrier, typically by optically-coupled isolators. The cost of these are significant in low-cost applications. In these cases it is preferred to adjust the converter duty in response to parameters available on the primary side of the transformer, primary switch current for example.

Control of the output voltage can be achieved by adjusting the converter duty (by changing the on-time) in response to current measured during the on-time.

Referring to Fig. 5a, the current is measured at a fixed delay after switch-on (Tlowsaniple). If the current at this time is above a predetermined threshold then the on-time (hence duty) is increased, if it is below the threshold then the duty is decreased.

For example, suppose the output voltage of the converter is low, this gives a fast rate of rise of current during the on-time so the sampling will record a current above the threshold. This causes an increase in duty, with consequent increase in output current.

The effect is to tend to stabilise the control in a condition where the Vin-Vout' difference is substantially constant.

For proper control by this method it is desirable that the time Tlowsample is consistent, measured from the time when the switch current starts. This is not the same as the time when the turn-on signal is applied to the switch (typically implemented as a transistor); there may be a delay between this signal and the time when the switch voltage has fallen to a low value and current starts to build. If there is error in timing, this manifests as a variation of Vin-Vout over the working range of loads. The error can be ininimised by: * Ensuring fast turn-on of the switch e.g. by applying a high turn-on current/voltage * Measure the switch voltage and time the Tlowsarnple from when the voltage reaches a predetermined low value * Measure the switch current and time Tlowsarnple from when it increases above a predetermined low value.

For simplicity, the first of these is preferred.

The typical effect of this technique is illustrated in figure 5b. The threshold value may be chosen to alter the slope of the V-I characteristic in the control region, a higher threshold reducing the slope. If excessive, the slope may reverse which manifests as negative output resistance which may be undesirable as it typically causes instability of output voltage.

A further useful characteristic of this technique is that the converter duty cycle automatically reduces (reduced on-time and, possibly, extended off-time) as the load reduces. This reduction in duty cycle also reduces the power dissipated in the converter (from resistive, core and switching losses), which is very desirable in order to maximise conversion efficiency and to reduce wasted power in low-or no-load conditions.

Very low duty cycles may be achieved by increasing the off-time (oiice the on-time has already been reduced to a minimum), with consequent reduced power waste and consistent regulation. However other factors may limit the minimum duty cycle, the need to maintain power to the controller via an auxiliary winding is an example. As the load is reduced, but the duty cycle reaches a minimum, the output voltage will then rise, as shown in figure 5b.

Such control is effective only over the available range of duty control. If the duty reaches the maximum or minimum limits imposed for other reasons then voltage control is lost and the output voltage will increase or decrease accordingly. Maximum duty is set by the maximum on to off ratio, typically limited by the need to reset flux in the magnetic component. For reducing load, the duty can be reduced indefinitely, preferably by reducing the on-time first then subsequently increasing the off-time, also as discussed above. However, if off-time is increased to a long time, the response to load increase may be unacceptably slow since no indication is available until the next converter cycle. Another limitation on allowable off-time may be a need to maintain power to the converter via the RFDC itself (see below).

A further benefit of varying the duty cycle with load is the possibility of reducing power waste at low (or zero) loads. Though the RDFC topology generally gives good efficiency, energy is wasted in several ways including, for example: Drive power to ensure the switch turns on; hysteresis and eddy current loss in the transformer core; loss associated with turn-off of the switch and output diode; energy stored in the resonating capacitor if the switch is turned on with non-zero voltage e.g. when off-time is extended beyond resonance.

There are other loss mechanisms, but the above cause loss on a per-cycle basis. So, to minimise the power loss from these it is preferable to reduce the frequency of cycles.

Reducing the on-time, without increasing the off-time, also offers some energy saving mainly by reducing the hysteresis loss of the core. However this saving may be outweighed by the increased loss due to adverse duty cycle causing higher on-state currents. Overall, it is particularly beneficial at low loads to reduce the on-time and increase the off-time. This is achieved by the voltage control scheme described above.

However it could also be achieved without the voltage control by measuring primary switch current in some other way and adjusting the duty accordingly. A preferable method is to measure the average of the current or integral over one, a few or many converter cycles. For AC/DC converters it is preferred to measure over a half or full cycle of incoming line waveform as this avoids noise and errors due to line-frequency variations.

We now describe low power operation without voltage regulation.

In applications where improved voltage regulation is not needed, this is an alternative method for managing converter duty in relation to applied load. One scheme is to compare the on-state switch current, sampled at the end of the on-period, to a fixed threshold. If it is above the threshold then the duty is increased (reduce any extended off-time then increase the on-time), it is below then the duty is decreased. However, at low loads the effect of magnetising current can be significant and may be mistaken for apparent load causing incorrect low-load behaviour. One method to avoid this is to integrate the supply current over the converter cycle, the integral then being insensitive to magnetising current. However this can be a difficult or expensive process to embody.

An alternative, using the example of reducing load, is: * Monitor primary switch current and reduce on-time (but off-time set for resonance) as described above.

* Once a preset minimum on-time has been reached, force an extended off-time followed by a minimum on-time. The primary switch current at the end of this on-time is measured and held as a subsequent reference.

* Continue with cycles of extended off-time and minimum on-time, each time comparing the primary switch current to the held reference.

* If the switch current, less the reference, is above a preset threshold then return to converter cycles using resonant off-time, rather than extended off-time. Otherwise continue with extended off-time The control techniques described here may be embodied in system at low cost with few components. Preferably, an integrated controller device works with a low-cost power bipolar transistor to make an off-line power converter with several commercial advantages, including: high power capability despite small and low-cost components; the use of bipolar power switch transistor rather than the higher-cost alternative of power mosfet or IGBT; compact size; low power loss in low/no-load conditions; and high conversion efficiency.

Figure 7 shows an example RDFC 700 with such a controller. The RDFC comprises an isolating transformer 702 having primary 7O2p and secondary 702s windings. A switch 704, when closed, applies a DC voltage V across the primary winding of the transformer. Power is delivered to the load 712 via a rectifying diode 708 and smoothing capacitor 710. The polarity of the transformer is such that power is supplied to the output circuit when the primary switch is closed.

The RDFC further comprises an RDFC controller 740. The controller preferably includes both analog and digital circuits to implement the above-mentioned control functions in accordance with the present invention. Preferably, the controller is fabricated in a low-cost conventional CMOS process.

The controller 740 is AC coupled to the primary winding of the transformer to sense a primary signal via a resonance capacitor 706, which also acts as the resonance capacitor for the RDFC. It is particularly advantageous for the capacitor to serve both functions; high voltages typically occur during converter operation and capacitors capable of withstanding them are expensive. Alternatively, two or more capacitors could be employed, one to couple signal to the controller and the other to supplement the resonance. This may be necessary in high power converters where the resonant current is high and unsuitable for applying in its entirety to the controller input. Optionally, the resonance may be coupled to the controller from a winding of the transformer other than the primary winding, the auxiliary winding for example. Furthermore, the primary switch current is sensed by a single resistor 720 The controller is configured to provide a drive signal to the primary switch in response to the above-sensed signals.

Auxiliary power for the controller device and for base current to the primary switch is derived from an additional winding on the converter transformer, shown in this instance operating in a forward mode. This mode is preferred as it minimises the range of auxiliary voltage Vaux under conditions of varying duty cycle and load.

Where the controller is powered from an auxiliary supply derived from the converter itself, there is an opportunity to use the rectified auxiliary voltage to manage the off-time in low power or foldback. In these modes, minimum converter duty (on-time and off-time) may depend on both maintaining sufficient output power delivery and maintaining sufficient power to operate the controller. The latter can be sensed via the auxiliary supply voltage or current and, if it falls, the duty increased to maintain adequate power to ensure proper operation High loads (reduced or zero output voltage) tend to reduce Vaux via the transformer action. Therefore, it is preferred to include a shunt-mode voltage regulator in the controller device and supply via a resistor (Raux) from Vaux to avoid malfunction of the controller. In this way, an accurate controller supply voltage can be provided despite changes in input voltage and load conditions.

However, a wide range of Vatix can still lead to high power dissipation, which is undesired, especially in no/low-load conditions. Preferably, the effect of load on Vaux is minimised by the construction of the transformer. For example, the winding sequence can be constructed (from core outwards): Auxiliary, Primary and then Secondary.

Furthermore, the secondary winding leakage inductance can be increased by winding over a reduced length of core, in comparison to the width of the primary and auxiliary windings.

To enable the controller to start up correctly from an unpowered state, power is preferably taken from the rectified high voltage supply via high value resistor Rht.

Programmable bi-level current limit As mentioned above, in a power converter controller it is often desired to be able to set different thresholds for, for example, low power and overload protection thresholds.

Moreover in a controller IC it is desirable to keep the pin-count low. Thus an example circuit 800 to provide a bi-level current limit using a single external connection 806 is shown in Figure 8.

Referring to Figure 8, a preferred method is to provide in the controller 802 a source of current 804 that flows through the connection pin 806 to the sense resistor Rs. External to the controller is a series resistor Rp so an offset voltage is developed across it as a result of the current through the pin, internally, two comparators 808a, b (using different thresholds TI-I 1, 2) provide signals to a power control module 810. With this arrangement, it is possible to set independent thresholds of primary switch current corresponding to the thresholds of the comparators.

The primary switch i, current corresponding to an internal threshold voltage of THy is: = Ib.Rp -THy Rs This assumes that Ib>>lp.

If Ib, TH 1 and TH2 are preset then by choosing values for Rs and Rp, it is possible to implement two ip threshold values independently, without recourse to altering Tb, THI or TI-12.

Rate-adaptive current limitin2 As previously mentioned the behaviour of forward converters such as RDFCs in overload conditions is governed by over-current protection (OCPH) action. During the.

on-period, primary current is monitored via the CS (current sense) resistor in series with the input voltage supply (HT-). If the current exceeds a preset OCPH threshold (the OCPH CS voltage threshold divided by the CS resistor value), the transistor is preferably switched off immediately (optionally subject to a blanking period timed from transistor turn-on). The effect on output current and voltage is, however, complex because of the effects on resonant behaviour of the converter.

For RDFCs, in general the output characteristic (excluding effect of foldback) is of the fonn, shown in Figure 3F. As can be seen in that figure the OCPH action moves the output voltage-current characteristic round a knee but after this the output current increases once more, moving down the curve. Thus undesirable high output currents can occur when the output voltage is clamped low. Note, however, that the technique may also be applied to non-resonant forward converters, with similar protection benefits in case of heavy overload.

Leakage inductance affects the profile because it determines the reduction of on-time and contributes to collector peak voltage. The latter is important because the resonant off-time depends on magnetising energy (which reduces with on-time) and leakage energy.

As mentioned previously, a second form of protection which may be implemented is "foldback", where high levels of overload trigger a protection mode where the output power is reduced by increasing the off-period far beyond resonance. In implementations foldback mode may be triggered when OCPH triggers within a preset time from transistor turn-on, that is when the output voltage is low, current increases rapidly through the leakage inductance and OCPH triggers within the threshold time.

Hence, the point on the V-I curve where foldback occurs depends on the leakage inductance, OCPH, resonant behaviour (average vs peak current) and threshold time.

Shorter threshold time means the foldback limit voltage reduces and current increases.

If the leakage inductance and OCPI-I are sufficiently high, the foldback threshold will be less than OV, in which case the controller will not fold back.

This is not necessarily a problem and may even be desirable in some applications.

Flowever the issues to consider are the stresses on the components in the application (particularly the primary switch, transformer, output diode and output capacitor).

Operation at high levels of overload increases the stresses; the skilled person will appreciate that what matters is the absolute stress in comparison to component capabilities (the main issues are thermal). Once foldback has occulTed the output voltage collapses (or even may be short-circuited), giving the highest instantaneous voltage and current stress -the components should preferably be able to survive this no matter what is the foldback limit. Before fo!dback occurs, however, switching cycles are continuous with high average values causing thermal stress.

The worst thermal conditions typically occur just before foldback protection triggers. if the foldback threshold time is short (e.g. a high clock frequency in the controller), the stress levels are higher. If they are too high, some possible solutions are to increase the leakage inductance or reduce the OCPH threshold. 1-lowever, this may not be possible while achieving the desired nominal load output characteristic. Thus we here describe a technique to adjust the OCPH behaviour at high overload which in embodiments helps to reduce maximum stresses and improve the V-I characteristic in this region.

Referring to Figure 9, this shows a part of an embodiment of an RDFC 900 configured to implement such a teclrnique. The inset circuit shows the potential divider of R2 and R3 re-drawn for clarity.

Two components, Cff and R3, are added to the CS network in the bi-level current limit circuit described above. Rcs is the current sensing resistor and R2 previously programmed OCPL threshold. In the configuration of Figure 9, R3 in parallel with R2 now program the (OCPL) threshold. The current through CFF depends on the rate of change of voltage across Cff (which voltage depends on the sensed current), and this current is converted to a voltage by R3, this component contributing to the output voltage from the R2, R3 potential divider dividing the voltage across Rcs.

First, consider the behaviour of current in the converter in relation to leakage inductance and output voltage as primarily illustrated in Figure 4a. The circuit of Figure 4a represents the effective electrical system during the on-period. If the parameters are referred to the input of the transformer, the circuit can be simplified to that shown in Figure 4b.

in Figure 4b Lmag is the inductance of the primary winding alone, and is normally much larger than Lleak -in which case it can be neglected in the analysis of current protection and the like. The series resistance is made up mainly of contributions from the transformer, output diode and output capacitor. Other components such as the HT capacitor and switching transistor contribute less. Vout' is the converter output voltage, plus the fixed component of output diode drop, referred to the primary.

During the on-period the current builds according to Lleak, Rtot and the difference between Vin and Vout' as: I (Vin-Vout')(l -eth'T)) Where t = Lleak/Rtot This is illustrated in Figure 5a when the Vin -Vout' mismatch is low, current ramps rclativcly slowly in the primary circuit. As Vout falls, the mismatch increases and so does the ramp rate of current in the primary.

The effect of Cff in the new network is to modify the OCPH triggering time when the ramp rate is high, effectively making the controller more sensitive if the ramp rate is high. In terms of protection behaviour, this means OCPH triggers sooner at a given voltage mismatch and leakage inductance. The modified characteristic depends on the R2/R3 ratio and on the Cff and R2//R3 time constant. A higher R2/R3 ratio makes the protection stronger. The time constant is preferably set to come into action around the desired on-time.

This technique modifies the output voltage/current characteristic of RDFC converters; typical behaviours are illustrated in Figure 10, which shows the effect of the circuit of Figure 9 on an RDFC with low series of (leakage) inductance.

For some applications (e.g. battery charging), a highly compliant output characteristic is desirable so that a well-controlled current can be delivered over a wide range of output voltages. A preferred method to achieve this is to increase the leakage inductance. The effect of this on the output characteristic of Figure 10 is shown in Figure 11: High leakage inductance increases the slope resistance and removes the region of negative output resistance when OCPH is active.

In the extreme, a "triangular" characteristic can he obtained, with foldback occurring at a low voltage. However achieving this level of leakage inductance can be difficult, and in any case this may not be desirable. It may also cause high EM! (electromagnetic interference) due to the fast/large rise of collector voltage at switch-off. In addition, with a triangular characteristic, currents are large at low output voltages, and this could be a problem for component stress and EMI. Using the modified OCPFI behaviour as described above, current can be limited at low output voltage before foldback occurs, as shown in Figure 12.

The skilled person will understand that, in particular, the bi-level current programming techniques we describe are not limited in their application to RDFCs or to switching power converters. No doubt many other effective alternatives and applications will occur to the skilled person.

It will be understood that the invention is not limited to the described embodiments and encompasses modifications apparent to those skilled in the art lying within the spirit and scope of the claims appended hereto.

Claims

CLAIMS: 1. A bi-level current sense circuit for a power converter, to

enable a single current sense input to provide two different current sense levels, said power converter comprising a magnetic device coupled between an input and an output of said power converter, a switch to switch power to said magnetic device, and a controller to control switching of said switch; said current sense circuit comprising: a single current sense input for connection to a current sense resistor via a second resistor, said culTent sense resistor for coupling in series with said switch to generate a current sense signal representing a current flowing through said switch; first and second comparators sharing a connection to said current sense input and having respective first and second outputs coupled to said controller, said first and second comparators being configured to compare a voltage on said shared input connection against different respective first and second threshold values; and a current generator coupled to said shared input connection; and wherein said single current sense input enables third and fourth bi-level current sense threshold values to be set independently by said current sense and second resistors without changing said first and second threshold values.
2. A device according to claim 1 wherein said first and second threshold values are fixed and wherein selection of values of said second resistor and said current sense resistor sets the values of said third and fourth threshold values.
3. A controller for a power converter, said controller comprising a bi-level current sense circuit according to claim 1 or 2 and wherein said controller is configured to control said switch in response to signals on said first and second outputs of said respective first and second comparators.
4. A controller as claimed in claim 3 having a high power mode and a low power mode and wherein said controller is configured to automatically select a said mode in response to said sensed current.
5. A power converter comprising a bi-level current sense circuit according to claim I or 2 or a controller according to claim 3 or 4.
6. A power converter controller IC including a bi-level current sense device for a power converter, said power converter comprising a magnetic device coupled between an input and an output of said power converter and a switch to switch power to said magnetic device; wherein said IC has a single external current sense connection, said single external connection being coupled to a pair of internal comparators with different thresholds, and such that said culTent sense device is configured to enable selection of two different independently externally adjustable current sense thresholds simultaneously by sensing a single current sense voltage level.
7. A method of providing two different levels of sensed current threshold using a single current sense input, a current sense resistor and a second resistor, said current senses resistor and said second resistor being connected in series, the method comprising: passing a current for sensing through said current sensing resistor; passing a second current via said current sense input through a said second resistor connected in series with said current sense resistor to generate a sensed voltage at said culTent sense input; and comparing said sensed voltage at said single current sense input against two different threshold values.
8. A method of setting the two different levels of sensed current threshold of claim 7, the method comprising selecting values of said current sense resistor and said second resistor without altering said comparator threshold values.
9. A forward switch mode power converter, said power converter having an input to receive an input power supply, an output to provide an output power supply, a magnetic device coupled between said input and said output, said magnetic device having at least one winding, and a switching device coupled to switch power from said input on and off to said winding of said magnetic device to transfer power from said input to said output, said power converter further comprising: a sensing circuit to sense a combination of a current flowing in said winding when said switching device is on and a rate of change of said current; and a control system coupled to said sensing circuit to control said switching of said switching device responsive to said sensed combination to control an output current from said converter to provide over-current protection.
10. A forward switch mode power converter as claimed in claim 9 wherein said control system includes a comparator to compare said sensed combination with a threshold level, and a system to reduce an average power input to said magnetic device when said sensed combination exceeds said threshold.
11. A forward switch mode power converter as claimed in claim 10 wherein said control system is configured to reduce an on-time of said switching device in response to detection of said sensed combination exceeding said threshold.
12. A forward switch mode power converter as claimed in claim 9, 10 or 11 wherein said sensing circuit is configured to generate a first signal dependent on said current and a second signal dependent on said rate of change of said current and to form a combination of said first and second signals.
13. A forward switch mode power converter as claimed in any one of claims 9 to 12 wherein said sensing circuit comprises a current sense resistance to generate a voltage dependent on said current through said winding, and a series coupled capacitor and second resistance together coupled in parallel with said current sense resistance to generate a voltage dependent on said rate of change of said current.
14. A forward switch mode power converter as claimed in claim 14 further comprising a third resistance coupled in parallel with said capacitor such that said second and third resistance form a potential divider in parallel with said current sense resistance.
15. A forward switch mode power converter in any one of claims 9 to 14 wherein said forward switch mode power converter is a resonant discontinuous power converter.
16. A forward switch mode power converter including a controller to control said power converter, said controller having a current sense input, and wherein said power converter further comprises a current sense circuit coupled to said current sense input, said current sense circuit comprising a current sense resistance, a series coupled capacitor and second resistance together coupled in parallel with said current sense resistance, and a third resistance coupled in parallel with said capacitor, and wherein a signal at said current sense input comprises a signal dependent upon a combination of a current sensed by said current sense resistance and a rate of change of said current.
17. A method of over-current protection in a forward switch mode power converter, said power converter having an input to receive an input power supply, an output to provide an output power supply, a magnetic device coupled between said input and said output, said magnetic device having at least one winding, and a switching device coupled to switch power from said input on and off to said winding of said magnetic device to transfer power from said input to said output, the method comprising: sensing a combination of a culTent flowing in said winding when said switching device is on and a rate of change of said current; and controlling switching of said switching device responsive to said sensed combination to control an output culTent from said converter to provide said over-current protection.
18. A method as claimed in claim 17 wherein said controlling comprises comparing said sensed combination with a threshold level and controlling said switching to reduce an average power input to said magnetic device when said sensed combination exceeds said threshold.
19. A method as claimed in claim 18 wherein said controlling comprises curtailing an on-time of said switching device.
20. A method as claimed in claim 17, 18 or 19 wherein said sensing of said combination comprises sensing said current using a current sense resistance to generate a voltage dependent on said current and generating a signal component representing said rate of change of said current for said combination from said voltage using a series coupled capacitor and second resistance.
21. A method as claimed in any one of claims 17 to 20 further comprising using an inductance between said input and said output to substantially remove a region of negative output resistance of said power converter when said over-current protection is active.
22. A method as claimed in any one of claims 17 to 21 wherein said forward switch mode power converter comprises a resonant forward switch mode power converter, the method further comprising implementing a foldback protection mode in which output power of said forward switch mode power converter is reduced in response to detection of an overload condition by increasing an off-period of said switching device by a time interval greater than one or more resonant periods of said resonant forward switch mode power converter.